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CMOS AMPLIFIERS. Simple Inverting Amplifier Differential Amplifiers Cascode Amplifier Output Amplifiers Summary. Simple Inverting Amplifiers. Small Signal Characteristics. Inverter with diode connection load. How do you get better matching?. High gain inverters. - PowerPoint PPT Presentation
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CMOS AMPLIFIERS
• Simple Inverting Amplifier
• Differential Amplifiers
• Cascode Amplifier
• Output Amplifiers
• Summary
Simple Inverting Amplifiers
Small Signal Characteristics
How do you get better matching?
Inverter with diode connection load
2
1
221
1
m
m
mdsds
m
in
outv g
g
ggg
g
v
vA
12
21
12
21'
'
LW
LW
LW
LW
K
K
v
vA
P
N
P
N
in
outv
2221
11
mmdsdsout gggg
r
High gain inverters
Current source load or push-pull
• Refer to book for large signal analysis
• Must match quiescent currents in PMOS and NMOS transistors
• Wider output swing, especially push-pull
• Much high gain (at DC), but much lower -3dB frequency (vs diode load)
• About the same GB
• Very power dependent
Small signal
High gain! Especially at low power.
21
21
dsds
mm
in
outv gg
gg
v
vA
21
1
dsds
m
in
outv gg
g
v
vA
Dm Ig 21 Dds Ig 1
D
vI
KA
21
1
dsdsout gg
r
))(1(|)|(2
)1(|)|(2
))(1()(2
)1()(2
22
2
2
2
2
22
22
1
1
2
1
11
oV
PTPinGGPDDoxP
DSPTPSGoxP
D
oV
NTinGGoxN
DSNTGSoxN
D
vVvVVL
WC
VVVL
WCI
vVvVL
WC
VVVL
WCI
DD
DD
Key to analysis by hand:• Use level 1 or 3 model equations• Use KCL/KVL
PTCDDoxP
ds
mTCDDoxP
m
NTCDDoxN
ds
TCDDoxN
m
DDoCDDGGPCDDGG
C
NPTTPoxPoxN
VEVL
WCg
gVEVL
WCg
VEVL
WCg
VEVL
WCg
VVEVVEVV
E
VVL
WC
L
WC
2
2
22
12
22
2
1
11
1
11
2
2
1
1
)2/)((2
)2/)((
)2/)((2
)2/)((
2/,2/)(,2/)(
:ispoint operating then used, is offset battery a and
,,|| suppose,22
Let
Dependence of Gain upon Bias Current
Transfer function of a system
Systeminput u output y
)()1(
)1()()()(
11
1
11
10 susasasa
sbsbsbAsusHsy
nn
nn
mm
mm
)()1()1)(1(
)1()1(
21
10
su
ps
ps
ps
zs
zs
A
n
m
0)()1( 11
1 sysasasa n
nn
n When u(s) = 0, y(s) satisfies:
These dynamics are the characteristic dynamics of the system. The roots of the coefficient polynomial are the poles of the system.
When y(s) = 0, u(s) satisfies:
0)()1( 11
1 susbsbsb m
mm
m These dynamics are the zero dynamics of the system. The roots of the coefficient polynomial are the zeros of the system.
Frequency Response of CMOS Inverters
Poles of CMOS Inverters
Let vin = 0, x = 0, VDD = 0, VSS = 0.
yCGS1, CGS2, CBS1, CBS2 are all short
CGD1, CGD2, CBD1, CBD2, CL in parallel
C’L = Ctotal = CGD1+ CGD2+ CBD1+ CBD2+ CL
Total conductance from y to ground:
go = gds1 + gds2
KCL at node y:
0)()1(
0)()(
0)()(
0
'
'
'
sys
sygsC
tygdt
tdyC
i
L
o
Cg
oL
oL
Therefore system pole is: '1L
o
C
gp
Zeros of CMOS Inverters
Let vin = x = u, VDD = 0, VSS = 0.
gds1, gds2 also short
CGD1, CGD2, are in parallel, CBD1, CBD2, CL are all short
No current in them
KCL:
0)()]()[(
)()()()(
2121
2121
suggsCC
gtugtudt
tduC
dt
tduC
mmGDGD
mmGDGD
Zero is:21
211
GDGD
mm
CC
ggz
Zeros of CMOS Inverters
Let vin = u, X=0, VDD = 0, VSS = 0.
gds1, gds2 also short
CGS2, CGD2, CBD1, CBD2, CL are all short
No current in them
KCL:
Zero is:
0)(][)()(
1111 sugsCgtudt
tduC mGDmGD
1
11
GD
m
C
gz
Input output transfer function
)1(
)1(
)1(
)1(
)(
'21
21
2121
21
1
10
L
dsds
GDGD
mmdsds
mm
C
ggs
CC
ggs
gg
gg
pszs
A
sA
When s=j0, A(0) 21
210
dsds
mm
gg
ggA
When w∞, A(s) 1'
21
L
GDGD
C
CC
|p1|=g0/CL’
|z1|=gm/Cgd
=GB*CL’/Cgd
|A0 | =gm/go
0 dB
Unity gain
frequency
=|A0p1|
=GB
=gm/CL’
Acl=1/
-3dB frequency of closed loop=*GB
Unity gain feedback
A(s)
10
1
0
0
10
0
10
1
0
0
110
10
1
1
1
))1()1(
1(1
1
1
)1(1
)1(
)(1
)(
pAs
zs
A
A
zAA
pAs
zs
A
A
pszsA
zsA
sA
sAAc
If a step input is given, the output response is
100
0
10
1
0
0 1)(
1
1
1
1
1
1 pAssA
A
spAs
zs
A
A
In the time domain:
0
0
0
0
1)(
110
A
Ae
A
A tpA
Final settling determined by A0
need high gainSettling speed determined by A0p1=GB=UGF,
need high gain bandwidth product
Gain bandwidth product
'21
'21
21
2110
L
mm
L
dsds
dsds
mm
C
gg
C
gg
gg
ggpAGB
C’L = Ctotal = CGD1+ CGD2+ CBD1+ CBD2+ CL
LDjDjoxoxL CLWCLWCLWCLWCC 221121'
22
21
1
121 EB
oxPEB
oxNmm V
L
WCV
L
WCgg
When CL ≈ C’L, W↑GB↑, but it saturates, when
LDjDjoxox CLWCLWCLWCLWC 221121
22
2
221
1
1
22 EBoxP
EBoxN
D VL
WCV
L
WCI
Note:
If VEB1 and VEB2 are fixed, W1/L1 and W2/L2
must be adjusted proportionally, and they are
proportional to DC power.
22
2
221
1
1
22 EBDDoxP
EBDDoxN
DDD VVL
WCVV
L
WCIVP
11
2
2
11
1
1 2,
2
EBDDEB
oxP
EBDDEB
oxN
VV
PV
L
WC
VV
PV
L
WC
)11
(222
211121
EBEBDDEBDDEBDDmm VVV
P
VV
P
VV
Pgg
)11
()),((
2
)11
(2
2121
21'
EBEBDDL
EBEBDDL
VVVWWCC
P
VVVC
PGB
Therefore:
P is proportional to W1, W2CL constant, but C(W1,W2) proportional to W1, W2When C(W1, W2) << CL, GB proportional to PWhen C(W1,W2)CL or >CL, GB saturates
P
GB
Linear increase region
),(
2
),(
2
21
1
21'1
WWCC
W
L
IC
WWCC
I
C
gGB
L
DoxN
L
D
L
m
For given current or power (current source load)
Initially, as W1 increased, GB increases
But GB will reach a max, and then drop as W1 increases
NOISE IN MOS INVERTERS
22
2
2
121
0
2n
m
mn
Vout e
g
gee
in
22
2
1
2212
22
nm
mn
v
outeq e
g
ge
A
ee
21
22
2
1
221
2 1n
n
m
mneq
e
e
g
gee
NN
PPneq BK
BK
L
Lee
'
'2
2
121
2 1
constant; :noise 1/fFor 2 BfWL
Ben
Dm IL
WKg
'2
N
P
DN
DPneq B
LfW
LfW
B
LIWK
LIWKee 11
2211'
22'
21
2
/2
/21
To minimize:
1) L2 >>L12) En1 small 1
2
2'
1'
L
L
WK
WKgain
P
N
For thermal noise
Noise in Push-Pull current source load Inverter
Differential Input, single-ended output single stage Amplifier
N-Channel
vin+vin-
P-channel
Large Signal Eq. in a N-channel Differential pair
iD1=0, when iD2=ISS and VGS2=VT+(2ISS/)0.5
=0.51(VGS1-VT)2
=(2ID1/1)0.5
21
2
21212
42
222
22
DDSSID
DDDDID
iiIv
iiiiv
21
22
21212
42
4)2
(2
4424
DDID
SSSSSS
DDDDSSSSID
iiv
III
iiiiIIv
22121
242
2 )(44 DDDDSS
IDIDSS iiiiI
vvI
4
422
21ID
IDSSDD
vvIii
Solving for iD1 and iD2
VON1=VON2=(ISS/)0.5iD1=iD2=ISS/2
2
422
21 422,
SS
ID
SS
IDSSSSDD I
v
I
vIIii
N-Channel Input Pair Differential Amplifier
C.M. Load
C.M. Bias
Simplecurrentreference
Voltage transfer curve
P-Channel Input Pair Differential Amplifier
Voltage transfer curve
INPUT COMMON MODE RANGE
VG1=VG2=ViCM
VSDSAT1=VSDSAT2
=VON
VD1=VD3=VSS+VT3+VON
VG1min=VD1-|VT1|
VG1max=VDD-VSD5SAT-|VT1|-VON
Output Range
Vomin=Vss+Von4
Vomax=Vicm –|VT2|
So what’s the vo range
What’s for the N-chcircuit.
SMALL SIGNAL ANALYSIS
AV
Common Mode Equivalent Circuit, with perfect match
iC1
iC1=VIC/(1/gm1
+2rds5)
ro1≈1/gm3
ACM≈1/ 2rds5gm3
CMRR=Av/ACM=2gm1gm3/(gds4+gds2)/gds5
If not perfectly matched
iC1
io=iIC
is a fraction
go1≈ gds2 + gds4
ACM≈gds5 / 2(gds2 + gds4)
CMRR=Av/ACM=2gm1/gds5
equations. all on to
respect withderivative partial take small, const Let
d
dic
sTGGDD
oDDTpGDD
soTsd
icD
GDDTpGDD
sGTsd
icD
v
vV
VVVIII
VVVVV
VVVVv
VI
VVVVV
VVVVv
VI
,)1
12
12
122
12
122
525
521
42
34
2
22
2
332
33
31
21
1
Formal detailed analysis
d
sTGG
d
D
d
D
d
GoDDTpGDD
d
oTpGDD
d
s
d
oTs
dic
d
ssoTs
dic
d
D
d
GTpGDD
d
GGDDTpGDD
d
s
d
GTs
dic
d
ssGTs
dic
d
D
v
VVV
v
I
v
I
v
VVVVVV
v
VVVV
v
V
v
VVV
vV
v
VVVVV
vV
v
I
v
VVVV
v
VVVVVV
v
V
v
VVV
vV
v
VVVVV
vV
v
I
52521
3434
42
34
2
22
222
33
23
3
33333
31
21
3111
2
1
2
22
2
11
2
2
1
22
2
11
2
ic
Gds
ic
Gm
ic
GTpGDD
ic
GGDDTpGDD
ic
s
ic
Gds
ic
sm
ic
s
ic
GTsic
ic
ssGTsic
ic
D
ic
icd
v
Vg
v
Vg
v
VVVV
v
VVVVVV
v
V
v
Vg
v
Vg
v
V
v
VVVv
v
VVVVVv
v
I
v
vv
33
33
33
23
3
33333
311
31
21
3111
2
1
1
2
11
,0)2
equations. all on to respect withderivative partial take
variable, tindependen as Let
ic
sds
ic
sTGG
ic
D
ic
D
ic
ods
ic
Gm
ic
oTpGDD
ic
GoDDTpGDD
ic
s
ic
ods
ic
sm
ic
s
icTsic
ic
ssoTsic
ic
D
v
Vg
v
VVV
v
I
v
I
v
Vg
v
Vg
v
VVVV
v
VVVVVV
v
V
v
Vg
v
Vg
v
V
v
VoVVv
v
VVVVVv
v
I
552521
43
4
42
34
3434
22
222
222
2
2
1
1
2
11
SLEW RATE: the limit of the rate of change of the output voltage
Max |CLdvo/dt|=ISS
C’Ldvo/dt=i4-i2
Slew Rate = ISS/C’L
ISS
ISSISS
0Output swing: Vosw
GB frequency: fGB
vo(t)=Voswsin(2fGBt)
Max dvo/dt =Vosw2fGB
To avoid slewing: ISS > C’L Vosw2fGB
Parasitic Capacitances
CT: common mode only
CM: mirror cap = Cdg1 + Cdb1 + Cgs3 + Cgs4 + Cdb3
COUT = output cap= Cbd4 + Cbd2 + Cgd2 + CL
• Impedances– rout = rsd2 || rds4 = 1 / (gds2 + gds4)
– rM = 1/gm3 || rds3 || rds1 ≈ 1/ gm3
– Hence the output node is the high impedance node
• When vi=0, slowest discharging node is output node with dominant pole
p1 = -1/(C’outrout), where C’out = Cout+ Cgd4
• Approximate transfer function
AV(s) = AV/(s/p1─1)
When vG1=vG2=0(AC)
KCL at D1:
omdsdsgdM
gdD
DmdsdsoD
gdD
M
vgggsCC
sCv
vgggdt
vvdC
dt
dvC
3314
41
13311
41
)(
0)()(
KCL at D2:
144424
14421
4
)())((
0)()(
Dmgdodsdsgdout
DmodsdsDo
gdo
out
vgsCvggsCC
vgvggdt
vvdC
dt
dvC
omgdgd
omdsdsgdMdsdsgdout
vgsCsC
vgggsCCggsCC
)(
)))(()((
444
3314424
Gain bandwidth product
• Gain AV(0) = gm1 / (gds2 + gds4)
• Bandwidth ≈ |p1| ≈ (gds2 + gds4) / C’out
• GBW ≈ gm1 / C’out
• gm1 = {2*ID1CoxW1/L1}½
– increase gm1 increase GBW
– increase W1 increase GBW• But C’out has Cdb2 and Cgd2 W1
– Once Cdb2 and Cgd2 become comparable to CL, increasing W1 reduces GBW
Other poles and zeros
M3
M2
M5
M1
M4
Vb2
VDD
VOUT
CL
Vi+ Vi-
Cgd4
(1+AV4)Cgd4
Cgd4
Second pole at D1
r = 1/gm3
C = CM +
(1+AV4)Cgd4
p2 = CM + (1+AV4)Cgd4
─ gm3
AV4 = gm4/gds4
M3
M2
M5
M1
M4
Vb2
VDD
VOUT
CL
Vi+= - Vi-
Vi-
Unstable zero at Cgd2
Enforce vo=0, float vin.
ids2, ids4 = 0
Cgd2 dvi-/dt= gm2 vi-
z1 = gm2/Cgd2
M3
M2
M5
M1
M4
Vb2
VDD
VOUT
CL
Vi+ Vi-
For zero at D1:
For diff, Vi+ = - Vi-
which is set by Cgd2
Both Cgd4 and Cgd1 to gnd
Ctot = CM + Cgd4
z2 = CM + Cgd4
─gm3
• A better approximation of TF:AV(s)=AV(s/z1-1)(s/z2-1)/(s/p1-1)(s/p2-1)
• If p1 is dominant, |p1|<<|p2|,|z1|,|z2|; AV(s)≈AV/(s/p1-1)
• If p1 is non-dominant, at low frequency, AV(s)≈AV /(s/p1+s/p2-s/z1-s/z2 -1)
• 1/peq≈ 1/p1+1/p2-1/z1-1/z2 ≈ 1/p1+1/p2-1/z2 ,
since |z1| >> |z2|, |p1|, |p2|; ≈ 1/p1, if AV4 is not
very large
• In either case, BW ≈ p1
frequency response
AV
-90
-180PM
p1 p2 z2z1
UGF
All in abs val
Observations
• PM ≈ 90 – tan-1(UGF/z1) GBW should be at least 2~3 times lower than z1 to
ensure good phase margin at UGF There is conflict between AV and PM
• If z2 not = p2, UGF < AV*p1
• Design approaches• make z1 high higher than UGF
• make Cgd2 small, gm1 large
• make z2 close to p2 better 1st order approx. • make AV4 small
• make p1 low large AV
• make gds2 and gds4 small
Design Steps
• Select Iss based on– GB & V_osw, SR, or P_max
• Select W1/L1 based on– GB = gm/CL’, Assuming CL’ = (1.1~1.5)CL– Maximize z1 (minimize Cgd2)
• Select W4/L4 based on– ICMR, – Small Av4
• Select W5/L5 based on– ICMR
NOISE Model
Input equivalent noise source
• Total output noise current is found as,
• Let
• Then
How does this affect Av4 and go?
Cascoding
• Objectives– Increase ro
– Increase AV
– Remove feed forward from vin to vo
– Remove unstable zero
• Methods– Direct cascoding– Folded cascoding
CMOS CASCODE AMPLIFIERS
VDD
Vbb
Vin
CL
Rb
Vout-min increase by VON2
Vout-max decreased if aCascoded source used
Output swing is a big Problem in low voltageApplications
VDD
Vin
CL
Rb
Vbb
Vin
CL
Vyy
Vxx
VDDQ: How should you set the bias?Q: what is Vout-max?
ro =
AV =
ro at D1?
vD1
vin
=
Cascoded current source load
Vbb
Vin
CL
Vyy
Vxx
VDDHigh frequency model
AV(s) =AV0(s/z1 -1)…
(s/p1-1)(s/p2-1)…
For poles, short input, and compute the time constants at each node. For zeros, float input but require vo = 0. (don’t short vo!)
Consider only the effect of the lower half circuit.
Vbb
Vin
CL
Vyy
Vxx
VDDShort vin, float vo:
At the high impedance node
r =rds1(gm2+gmb2)rds2
C =CL+Cdb2+Cgd2
p1 = -1/RC
At the low impedance node
r =1/(gm2+gmb2+gds1+gds2)
C =Cgd1+Cdb1+Cgs2+Csb2
p2 =
Vbb
Vin
CL
Vyy
Vxx
VDDEnforce vo=0, float vin.
At the G1-D node
iCo=0, no current cross line, and iCgd2=0id2, id3 = 0, gm2vgs2=0
Was the unstable zero removed?
vs2=0
igds1=0
sCgd1vin=gm1vin
Gain bandwidth product
• If |p1| << |p2|, |p3|,…, |p1| << |z1|, |z2|,…
– BW ≈ |p1|
– GBW ≈ gm1/Co
• Otherwise– AV(s) ≈ AV/(s/p1+s/p2…-s/z1-s/z2… - 1)
– 1/BW ≈ 1/p1+2/p2…-1/z1-2/z2… = RC1 + RC2 + …
VDD
Vbb
Vin
CL
Rb
VDD
Any enhancement?
Note: rds2, Rb 1/ID2
gm2 √ID2
Effects on:
ro, AV
Co, GBW
Slew rate
VDD
Vbb
Vin
CL
Rb
Another possible modification
Effects on:
ro, AV?
Co, GBW?
Slew rate?
poles?
zeros?
VDD
Vin CL
Folded cascoding
Which I source should be cascoded?
ro, AV?
Co, GBW?
Slew rate?
poles?
zeros?
Vbb
OUTPUT AMPLIFIERS
• Requirements– Provide sufficient output power in the form of
voltage or current.– Avoid signal distortion for large signal swings.– Be power efficient.– Provide protection from abnormal conditions.
• Types of Output Stages– Class A amplifier.– Source follower.– Push-Pull amplifier ( inverting and follower).– Negative feedback (OP amp and resistive).
Power efficiency
• It is most power efficient at maximum signal level
• Let VSS= ─VDD, Vin is sinusoidal such that Vout reaches Voutmax
• PRL = ½ (Voutmax)2/RL
• Psupply=average((VDD or VSS)*IRL) =VDD*average(Voutmax/RL *sin()) =2*VDD*Voutmax/RL/
• Power efficicy = PRL/Psupply</4 (78%)
CLASS A AMPLIFIER
ro, AV, z, p as before
Power effic = PRL
Psupply
= 0.5voutmaxIQ
IQ(VDD-VSS) < 25%
VSS=-VDD, Voutmax=VDD-Vdssat
SOURCE FOLLOWER
or VSS+VT
Push-pull
Push-pull inverting amp
Implementation
PUSH-PULL SOURCE FOLLOWER
Negative Feedback To Reduce Rout
Ro=?
Super source follower
VDD
Vin
Vo
Vo => I1 =(gm1+gmb1)Vo
VGS2= ro1(gm1+gmb1)Vo
I2 = gm2ro1(gm1+gmb1)VoI1
I2
go=gm2ro1(gm1+gmb1) +(gm1+gmb1)+go2
≈gm2ro1(gm1+gmb1)Gm ≈gm1+gm1ro1gm2
AV=Gm/go≈gm1
gm1 + gmb1
Ex: rework these when I1 and I2 have finite ros.
VDD
Vin
Vo
I1
I2
If we re-arrange witha flipped version, we get this push-pullsuper source follower
Ex: provide a transistorlevel implementation.Comment on powerefficiency.
