Development of Dielectric Resonator Antenna

7/28/2019 Development of Dielectric Resonator Antenna

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Development of Dielectric

Resonator Antenna (DRA)K. W. Leung

State Key Laboratory of Millimeter Waves &

Department of Electronic Engineering,City University of Hong Kong

Presented to the IEEE LI Section Antenna and

Propagation Society on Monday October 8th 2012


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I. Introduction

II. Circularly Polarized DRA Using a Parasitic Strip

III. Frequency Tuning Technique

IV. Omnidirectional Circularly Polarized DRAs

V. Dualband & Wideband DRAs

VI. Dualfunction DRAs

Outline


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3

The DRA is an antenna that makes use of a radiating modeof a dielectric resonator (DR).

It is a 3-dimensional device of any shape,

e.g., hemispherical, cylindrical, rectangular,

triangular, etc.

Resonance frequency determined by the its dimensions and

dielectric constant r.

What is Dielectric Resonator Antenna (DRA) ?


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4

Some DRAs :


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5

Advantages of the DRA

Low cost

Low loss (no conductor loss) Small size and light weight

Reasonable bandwidth (~10% for r ~10)

Easy of excitation High radiation efficiency ( generally > 95%)


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6

Excitation schemes

(i) Microstrip line feed


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7

Ground plane

Microstrip

feed line

Feed

Substrat

DRA

Slot

(ii) Aperture-couple feed

Excitation schemes


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8

(iii) Coaxial feed

Excitation schemes


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9

Bottom viewTop view

Coaxial feed


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10

Bottom view Top view

Aperture-coupled feed


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11

Slot-fed DRA array using corporate

microstrip feed network

Corporate feedline for DRA array


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12

Conformal-Strip Method


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Rectangular DielectricResonator Antennas


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14

Proposed Antenna Geometry

W1

l1

a

d

b

x

y

zGround

Plane

Rectangular

DRA

Coaxial

Aperture

Conducting

Strip

r

a

(mm)

b

(mm)

d

(mm)

l1(mm)

W1(mm)

r

14.3 25.4 26.1 10 1 9.8


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Resonant frequency of TEmnl(y) mode

Analytical Solution

Dielectric Waveguide Model (DWM)

222

0

2zyx

r

kkkcf

d

l

kb

n

ka

m

k zyx 2,,

2

0

222 kkkk rzyx


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16

Numerical Solution

Advantages

- Very simple- High modeling capability for general EM structures

- No spurious modes nor large matrix manipulation

- Provide a very wideband frequency response

Finite-Difference Time-Domain (FDTD) method

Disadvantages

- Time consuming, powerful computer required


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17

Baseband Gaussian pulse

])3(exp[22 TTntEz T : pulse width

0

011

)(

)(

ZZ

ZZS

tIFFT

tVFFTZ

in

in

in

Conformal Strip

Ground Plane

Ez (V)Hy Hy

HxHx

(I)

Source occupies only one grid

Source model and extraction of S parameters


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18

Uniform Cartesian grids

T = 0.083ns, t0

= 3T

10-cell-thick PML with polynomial spatial scaling

(m = 4 and max = 1)

total grid size : 80x 110y 112z

total time steps : 10000

x= 0.715 mm, y= 0.508 mm, z= 0.5 mm

Parameters


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19

3 3.5 4 4.5 5 5.5 6

-100

-50

0

50

100

150

200

Frequency (GHz)

2 2.5 3 3.5 4 4.5 5 5.5 6-30

-25

-20

-15

-10

-5

0

ExperimentTheory

Frequency (GHz)

|S11|(dB)

InputImpedance(ohm)

ExperimentTheory

Resistance

Reactance

Input Impedance/S11

Reasonable agreement.

Wide Bandwidth of ~ 43%.

Dual resonantTE111y and TE113y modes are excited.


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20

y

111TE

y

113TE

ResonantModes

Measuredresonant

frequencies

Calculated resonantfrequencies (FDTD)

Predicted resonantfrequencies (DWM)

fmea

(GHz)fFDTD

(GHz)

error

(%)

fDWM

(GHz)

error

(%)

3.81 3.90 2.3 3.95 3.6

N/A N/A N/A 4.26 N/A

4.57 4.60 0.7 4.7 1.7

Comparison between Theory and Measurement

y

112TE



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21

Field Distribution --- TE111y

Electric field

Magnetic field

z

2dx

b

x-z

bx

y

a

x-y

Imaged DRA (gound plane removed) With gound plane


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22


x-z

bx

y

a

x-yElectric field

Magnetic field

z

2d x

b

Imaged DRA (gound plane removed)


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23


x-zElectric field

Magnetic field

bx

y

a

x-y

z

2dx

b

With gound planeImaged DRA (gound plane removed)


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24

E (xz) - plane H (yz) - plane(+x)(-x) (-y) (+y)-40

0o

-30 -20-10 0

30o

30o

60o

60o

90o

90o

-40 -30 -20-10 0

0o

30o

30o

60o

60o

90o

90o

E (xz) - plane H (yz) - plane(+x) (-y)(-x) (+y)

-40 -30-20-10 0

0o

30o

30o

60o

60o

90o 90o

-40 -30-20-10 0

0o

30o

30o

60o

60o

90o 90o

f= 3.5 GHz

f= 4.3 GHz

Radiation Patterns

Broadside radiation patterns are observed.

Measured E-plane crosspolarized fields mainly caused by finite

ground plane diffraction.


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III. Circularly Polarized Design

using a Parasitic Strip


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26

Proposed Antenna Geometry

a

(mm)

b

(mm)

d

(mm)

l1(mm)

W1(mm)

l2(mm)

W2(mm)

0(degree)

r

24 23.5 12.34 10 1 12 1 225.6 9.5

W1

l1

ad

b

x

y

z

Ground

Plane

Rectangular

DRA

Coaxial

ApertureConducting

Strip

r

C

W2l2 Parasitic

Patch

(Center of

Parasitic Patch)

0

Top

ViewParasitic

Patch

Conducting

Strip

x

y

r


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27

Input Impedance/S11


Bandwidth ~ 14%.

Two nearly-degenerate TE111(y) modes are excited.

CP operation

2.5 3 3.5 4 4.5-25

-20

-15

-10

-5

0

|S11|(

dB)

Frequency (GHz)


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28

Axial Ratio in the boresight direction

3-dB AR bandwidth is ~ 2.7%, which is a typical value for

a singly-fed CP DRA.

3 3.1 3.2 3.3 3.4 3.5 3.6 3.7 3.8

0

5

10

15

20

AxialRat

io(dB)

Frequency (GHz)

ExperimentTheory


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The H field of the DRA without and with parasitic

strip (Top view)

Without parasitic strip - LP field With parasitic strip - CP field

3.4 GHz 3.4 GHz

Feeding strip Feeding strip

Parasitic

strip

29


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30

Radiation Patterns (f= 3.4GHz, )

LHCP

RHCP

xz plane yz plane(+x)(-x) (-y) (+y)

-40 -30 -20-10 0

0o

30o

30o

60o

60o

90o

90o

-40 -30 -20-10 0

0o

30o

30o

60o

60o

90o

90o

A broadside radiation mode is observed.

For each radiation plane, the LHCP field is more than 20dB

stronger than the RHCP field.

The maximum gain is 5.7 dBic (not shown here).


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Effects of feeding strip length l1

Input impedance changes substantially with l1.

AR is almost unchanged for different l1.

l1 can be adjusted to match the impedance without changing AR.

2.4 2.6 2.8 3 3.2 3.4 3.6 3.8 4 4.2-20

-15

-10

-5

0

|S11|

(dB)

Frequency (GHz)

l1 = 8 mm

l1 = 10 mm

l1 = 12 mm

3.1 3.2 3.3 3.4 3.5 3.6 3.70

5

10

15

20

AxialRa

tio(dB)

Frequency (GHz)

l1 = 8 mm

l1 = 10 mm

l1 = 12 mm


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II. Frequency Tuning Technique


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The DRA for a paticular frequency may not be

available from the comericial market.

Fabrication tolerances cause errors betweenmeasured and calculated resonant frequencies.

Frequency tuning methods:(i) loading-disk; and

(ii) parasitic slot.

Backgruond


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Frequency Tuning Technique- using a loading disk


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Side view Top view

The slot-coupled DRA with a conducting loading

cap

d

z

Slot

Hemispherical DRA

Dielectric

substrate rs

microstrip

line

Lt

Ground

planeara

xConducting

Loading Cap

Lt

x

y

Ls

Ws

Hemispherical

DRA

microstrip

line

Slot

Wf

Conducting

Loading Cap

Hemispherical DRA: radius a= 12.5 mm, dielectric constant r

= 9.5.

Coupling slot : length L s , width WsOpen-circuit stub: length L

t

Grounded dielectric slab: rs

= 2.33, height d= 1.57 mm

Microstrip feedline: width Wf= 4.7 mm


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Calculated and measured return losses

(L s = 12 mm and Ws = 1 mm)

3 3.2 3.4 3.6 3.8-70

-60

-50

-40

-30

-20

-10

0

Frequency (GHz)

|S11|(dB) = 26.38

Lt = 4.42 mm

o

= 52.8Lt= 13.6 mm

o

Theory

Experiments

Without cap (= 0 )Lt= 10.63mm

o

Resonance frequency:

3.52 GHz without any conducting cap ( = 00), withLt = 4.42 mm

3.25 GHz ( = 26.38o andLt = 4.42 mm)

3.68 GHz ( = 52.8o andLt = 13.6 mm)


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Calculated and measured radiation patterns

-40 -30 -20 -10 0

0

30

60

0

30

60

90o o

o o

o o

o

(a) (b)

-40 -30 -20 -10 0

0

30

60

90

30

60

90o o

o o

o o

oCo-pol

X-pol

-40 -30 -20 -10 0

0

30

60

0

30

60

90o o

o o

o o

o

(c)

-40 -30 -20 -10 0

0

30

60

90

30

60

90o o

o o

o o

o

(d)

Co-pol

X-pol

3.58 GHz ( = 52.8o

andLt = 13.6 mm)

Reasonable agreement

between theory andexperiment.

The effect of loading

cap on field pattern is not

significant.

3.25 GHz ( = 26.38o andLt = 4.42 mm)


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Calculated and L t for having a good return loss

(minimum |S11| < -20dB)

3.2 3.3 3.4 3.5 3.6 3.70

10

20

30

40

50

60

2

4

6

8

10

12

14

16

Tuning frequencyfr(GHz)

Angle(d

egree)

StubLengthLt(mm)

The resonant frequency can be tuned by varying and L t decreases from 26.38o to 0o(3.25


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Impedance bandwidth

The bandwidth decreases after a loading cap is added.

3.2 3.3 3.4 3.5 3.6 3.7

2

4

6

8

10

Frequency (GHz)

ImpedanceBandw

idth(%)


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Frequency Tuning Technique

- using a parasitic slot


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(a) Side view

(b) Top view

The annular-slot-excited cavity-backed DRA

z

x

Hemispherical DRA

Coaxial cable Parasitic slot

a

r

metallic cavity b

Ground plane

WB WA

Feeding slotrB

rA

Coaxial cable

Hemispherical DRA Feeding slot

a

r

metallic cavity

b

Parasitic slot

A

rA rBWB

WA

y

x


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IV. Omnidirectional CircularlyPolarized DRA


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43

CP DRAs concentrated on broadside-mode designs only.

Provide larger coverage.

Advantages of omnidirectional CP antenna


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44

Slotted omnidirectional CP DRA

Design I:


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45

Antenna configurations

h

wd

a

b

Dielectric

block

x

h

g

z

Slot

SMA connector

w

Probel

12r

Perspective view Front view

Dielectric cube with oblique slots (polarizer) fabricated onits four sidewalls.

Centrally fed by a coaxial probe extended from a SMAconnector, whose flange used as the small ground plane.


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Wave polarizer

h

wd

a

b

Dielectric

block

LP omnidirectional DRA

Dielectric block

Pol

ariz

er

Dielectric block with the wave

polarizer

Proposed compact omnidirectional CP DRA

x

y

z

Dielectricslabs

D

E

+

Antenna principle


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47

Prototype for 2.4 GHz WLAN design

Top face and sidewalls Bottom face

Design parameters

r= 15, a = b = 39.4 mm, h = 33.4 mm, w = 9.4 mm,

d =14.4 mm, r1= 0.63 mm, l= 12.4 mm,g= 12.7 mm

Photographs of the prototype


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Simulated and measured results

Reflection coefficient

Impedance bandwidth: AR bandwidth:Simulated: 20.3% (2.34-2.87 GHz) Simulated: 8.2% (2.34-2.54 GHz)Measured: 24.4% (2.30-2.94 GHz) Measured: 7.3% (2.39-2.57 GHz)

Axial ratio

2 2.2 2.4 2.6 2.8 3

-30

-20

-10

0

Frequency (GHz)

|S11| (dB)

HFSS Simulation

Experiment

Axial Ratio (dB)

HFSS Simulation

Experiment

2 2.2 2.4 2.6 2.8 30

3

6

9

Frequency (GHz)


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49

Very good omnidirectional characteristic

In the horizontal plane,LHCP fields > RHCP fields by~20 dB .

Simulated and measured radiation patterns

LHCP

-40 -30 -20 -10 0

30

150

60

120

9090

120

60

150

30

180

o

o o

o

o o

o o

o o

o

dB

xz-plane

0o

RHCP

(+x)(-x)

-40 -30 -20 -10 0

30

210

60

240

90270

120

300

150

330

180

o

o o

o

o o

o o

o o

o

dB

xy-plane

0o

RHCP


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50

Simulated and measured antenna gain

2 2.2 2.4 2.6 2.8 3-3

-2

-1

0

1

2

3

Frequency (GHz)

Gain (dBic)

HFSS Simulation

Experiment


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51

Wideband omnidirectional CP antennawith parasitic metallic strips

Design II:


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52

Perspective view Front view

x

h

g

z

ls

ws

Slot

Strip

SMA connector

w

Probe

Hollow circular

cylinder

l

12r

Four parasitic metallic strips are embedded in thelateral slots to enhance the AR bandwidth.

The hollow circular cylinder is introduced to enhancethe impedance bandwidth.

Antenna configurations


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53

Photographs of the prototype

Top face and sidewalls Bottom face

Design parametersr= 15, a= b= 30 mm, h= 25 mm, r= 3 mm, w =7 mm, d =10.5 mm

ls

= 30.5 mm, ws

= 1 mm, x0 = 6.4 mm, r1= 0.63 mm, l= 19 mm.

Prototype for 3.4 GHz WiMAX design


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Overlapping bandwidth: 22.0%; bandwidth widened by ~3 times.

Simulated and measured reflection

coefficient and axial ratio

Axial Ratio (dB)

2.8 3.2 3.6 4-30

-20

-10

0

Frequency (GHz)

|S11| (dB)

HFSS Simulation

Experiment

2.8 3.2 3.6 40

6

12

18

Frequency (GHz)

Impedance bandwidth:Simulated: 22.3% (3.11-3.89 GHz)Measured: 24.5% (3.08-3.94 GHz)

AR bandwidth:Simulated: 24.8% (3.11-3.99 GHz)Measured: 25.4% (3.16-4.08 GHz)


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Measured gain: wider bandwidth.

Measured antenna efficiency: 84-98% (3.1-3.9 GHz).

Antenna gain

Simulated and measured results

Radiation efficiency

Gain (dBic)

HFSS Simulation

Experiment

2.8 3.2 3.6 4-4

-3

-2

-1

0

1

2

3

Frequency (GHz)2.8 3.2 3.6 40

0.2

0.4

0.6

0.8

1

Frequency (GHz)

Efficiency


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56

3.4 GHz 3.8GHz

LHCP fields > RHCP fields by more than 15 dB in horizontal plane. Stableradiation patterns across the entire passband (3.13.9 GHz).

LHCP

-40 - 30 - 20 -10 0

30

210

60

240

90270

120

300

150

330

180

o

o o

o

o o

o o

o o

o

dB

xy-plane

0o

RHCP

-40 - 30 - 20 -10 0

30

150

60

120

9090

120

60

150

30

180

o o

o o

o o

o o

o o

o

dB

xz-plane

0o

RHCP

(+x)(-x)

-40 -30 -20 -10 0

30

150

60

120

9090

120

60

150

30

180

o

o o

o

o o

o o

o o

o

dB

xz-plane

0oLHCP

RHCP

(+x)(-x)

-40 -30 -20 -10 0

30

210

60

240

90270

120

300

150

330

180

o

o o

o

o o

o o

o o

o

dB

xy-plane

0o

RHCP

Simulated and measured radiation patterns


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V. Dualband & Wideband DRAs


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(i) Rectangular DRA


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59

Dualband and wideband antennas are extensively used

(e.g., WLAN)

Multi-element DRA [1]

- requiring more DR elements and space

Hybrid slot-DRA [2]

- coupling slot used as the feed and antenna

- inflexible in matching the impedance

[1] Petosa, N. Simons, R. Siushansian, A. Ittipiboon and C. Michel, Design and analysis of

multisegmentdielectric resonator antennas, IEEE Trans. AP, vol.48, pp.738-742, 2000.

[2] Buerkle, K. Sarabandi, and H. Mosallaei, Compact slot and dielectric resonator antenna with

dual-resonance, broadband characteristics,IEEE Trans. AP, vol. 53, pp.1020-1027, 1983.

Background


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Wideband DRA [1]

Dualband DRA [2]

Trial-and-error approach is normally used

Systematic design approach is desirable[1] B. Li and K. W. Leung, Strip-fed rectangular dielectric resonator antennas with/without a

parasitic patch,IEEE Trans. Antennas Propagat., vol.53, pp.2200-2207, Jul.2005.

[2] T. H. Chang and J. F. Kiang, Dual-band split dielectric resonator antenna, IEEE Trans.

Antennas Propagat., vol.55, no.11, pp.3155-3162, Nov.2007.

Use of higher-order DRA

mode


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Design Formulas for Dual-Mode rectangular DRA

yTE111yTE112

yTE113

The E-field should vanish on the PEC and the TE112 mode

cannot be excited properly.

The TE111 mode and TE113 mode are used in the dual-

mode design.


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62

x

y

z

b

a

d

Ground plane

Rectangular DRAr

Formula Derivation

The wavenumbers kx1,x2 and

kz1,z2 can be written as follows:

akk xx

21

dkz

21

dkz

2

32


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From the DWM model, the frequenciesf1,f2 are given by:

2

2,1

2

2,1

2

2,12,1 2 zzyyxxr

kkkc

f

2

2,1

2

2,1

2

2,12,1 zzxxyy kkkk

in which are wavenumbers in the

dielectric, with c being the speed of light in vacuum.

where

cfk r /2 2,12,1

(*)


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64

dkkd 2122

2

)96.3(

22

21

1

2

32.10

9

32.10 ff

kka

21 35.065.0 bbb

Engineering Formulas for the DRA dimensions

2

1

23

1

24

1

2 4422.113209.21393.0f

f

f

f

f

fd

3

1

2 104437.184984.23

f

f(m)

11

1tan2

2

2,1

2,11

2,1

2,1

yyryy k

k

kb

where


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65

)96.3(

22

21

1

2

32.10

9

32.10 ff

kka

Limit of frequency ratio f2/f1

From

3k1 > k2 or 3f1 >f2We have

giving

f2/f1 < 3

which is the theoretical limit that is not known before.

09 2221 dkk


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66

1 1.2 1.4 1.6 1.8 2 2.2 2.4 2.6 2.80

1

2

3

4

5r

10r

30r

70r

Error of (%)1f

f f2 1/

Compared with DWM results, errors off1, f2 are both

less than 2.5% for 1


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67

A. Example for Dual-band Rectangular DRA Design

a = 20.8 mm, b = 10.5 mm, and d= 18.5 mm.

Given:f1 = 3.47 GHz (WiMAX)

f2 = 5.2 GHz (WLAN), r=10

Using dual-mode

formulas


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68

Configuration of the dualband DRA

b

ad

Ground plane

Microstripline

Coupling

slot

LSW

L

Wf

h

Substraters

Rectangular

DRAr

z

y

x

W= 2.6 mm,L =10.6 mm,Ls=7.2 mm, Wf=1.94 mm,

h=0.762mm, rs= 2.93


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69

Measured and simulated reflection coefficients

Measured bandwidths:

Lower band: 15% (3.25-3.78 GHz) covering WiMAX (3.4-3.7 GHz).

Upper band: 8.3% (5.03-5.47 GHz) covering WLAN (5.15-5.35 GHz).

3.2 3.6 4 4.4 4.8 5.2 5.6-40

-30

-20

-10

0

HFSS Simulation

Measurement

Frequency (GHz)

Reflection coefficient |S | (dB)11


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70

COMPARISON OF DESIGN, SIMULATED, AND MEASURED

RESONANCE FREQUENCIES OF TE111y AND TE113

y MODES

Resonant

Mode

Measured

frequency

(GHz)

Design

frequency

Simulated HFSS

frequency

f1,2(GHz)

Error(%)

fHFSS(GHz)

Error(%)

TE111y 3.40 3.47 2.05 3.47 2.05

TE113y

5.18 5.30 2.32 5.24 1.15


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71

TE111y mode: measured (3.40 GHz), simulated (3.47 GHz).

Broadside radiation patterns are observed for both planes.

Co-polarized fields > cross-polarized fields by more than 20 dB in

the boresight direction.

Measured and simulated radiation patterns

-40 -30 -20 -10 0

30

150

60

120

9090

120

60

150

30

180

0oo

o o

o

o o

o o

o o

o

dB

9090

150

180

o

o

o

o

o

dB

-40 -30 -20 -10 0

30

150

60

90

120

60

150

30

180

0 oo o

o o

o o

o o

o o

o

dB

(a)

Simulation

H-plane (y-z plane)

Measurement

E-plane (x-z plane)

-x

120

Co-pol

Cross-pol

Co-pol

Cross-pol

+x -y +y


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72


TE113y mode: measured (5.18 GHz), simulated (5.24 GHz).

Broadside radiation patterns are observed for both planes.

Co-polarized fields > cross-polarized fields by more than 20 dB in

the boresight direction.

-40 -30 -20 -10 0

30

150

60

120

9090

120

60

150

30

180

0oo

o o

o

o o

o o

o o

o

dB

9090

150

180

o

o

o

o

o

dB

-40 -30 -20 -10 0

30

150

60

90

120

60

150

300 oo o

o o

o o

o o

o o

o

dB

(b)

120

180o180o

Simulation

H-plane (y-z plane)

Measurement

E-plane (x-z plane)

-x +x -y +y

Co-polCo-polCross-pol Cross-pol


73/122

73

TE111y mode: Maximum gain of 4.02 dBi at 3.48 GHz.

TE113y mode: Maximum gain of 7.52 dBi at 5.13 GHz.

Electrically larger antenna has a higher antenna gain.

Measured antenna gain

3 3.5 4 4.5 5 5.5 6-5

0

5

10Gain (dBi)

Frequency (GHz)


74/122

74

B. Example for Wideband DRA Design

a = 30.7 mm, b = 24.7 mm, and d= 47.7 mm.

Given:f1 = 1.98 GHz (PCS)

f2 = 2.48 GHz (WLAN), r=10

Using formulas fordual-mode

rectangular DRA


75/122

75

Configuration of the wideband DRA

b

Conducting

feeding strip

Coaxial

aperture

a

dW

l

Ground plane

Rectangular

DRArx

y

z

l= 17 mm, W= 1 mm


76/122

76


Measured bandwidths : 30.9% (1.83-2.50 GHz)

PCS (1.85-1.99 GHz), UMTS (1.99-2.20 GHz)

& WLAN (2.4-2.48 GHz)

1.5 2 2.5 3-40

-30

-20

-10

0

HFSS Simulation

Measurement

Reflection coefficient |S | (dB)11

Frequency (GHz)



77/122

77


Measured (2.16 GHz), simulated (2.11 GHz).


Co-polarized fields > cross-polarized fields

by more than 20 dB in the boresight direction.

-40 -30 -20 -10 0

30

150

60

120

9090

120

60

150

30

180

0o

o

o o

o

o o

o o

o o

o

dB

9090

150

180

o

o

o

o

o

dB

-40 -30 -20 -10 0

30

150

60

90

120

60

150

30

180

0o

o o

o o

o o

o o

o o

o

dB

120

Simulation

H-plane (y-z plane)

Measurement

E-plane (x-z plane)

-x +x -y +y

Co-pol Co-pol

Cross-pol Cross-pol

(a)


78/122

78


Measured (2.41 GHz), simulated (2.46 GHz).


Co-polarized fields > cross-polarized fields by more

than 20 dB in the boresight direction.

-40 -30 -20 -10 0

30

-150

60

-120

90-90

120

-60

150

-30

180

0o

o

o o

o

o o

o o

o o

o

dB

90-90

150

180

o

o

o

o

o

dB

-40 -30 -20 -10 0

30

-150

60

90

120

-60

150

-30

180

0o

o o

o o

o o

o o

o o

o

dB

H-plane (y-z plane)

Measurement

E-plane (x-z plane)

Simulation

Co-pol Co-polCross-pol Cross-pol

(b)

-x +x -y +y

M d t i


79/122

79

Measured antenna gain

The maximum gain of 6.98 dBi at 2.47GHz.

TE113y -mode gain > TE111

y -mode gain.

1.6 1.8 2 2.2 2.4 2.6 2.80

2

4

6

8

Frequency (GHz)

Gain (dBi)


80/122

(ii) Cylindrical DRA

R f f th HEM d f th li d i l DRA


81/122

81

2

0

22

irzii kkk Ground plane

z

a

Cylindrical

DRA

h

r

ki & kzi :dielectric wavenumbers along the &zdirections

k0i = 2fi/c : wavenumber in air

(1)

Resonance frequency of the HEMmnrmode of the cylindrical DRA

i = 1, 2 forf1,f2

f1 : HEM111 mode frequencyf2 : HEM113 mode frequency

R f f h HEM d f h li d i l DRA


82/122

82

z

x

y

r

a

Infinitedielectric rod

Resonance frequency of the HEMmnrmode of the cylindrical DRA

Fork:

220)1(' iiri kkk

where

is the radial wavenumber outside the

dielectric rod

Jm(x) : Bessel function of the first kind

Km(x): modified Bessel function of the second kind.

(2)

(3)

D. Kajfez and P. Guillon, Dielectric resonators, Norwood, MA, Artech House, Inc., 1986.

24

22222

)'(

)')('(

'

''

'

1

)(

'

'

''

'

1

)(

'1

akk

kkkkm

akK

akK

kakJ

akJ

kakK

akK

kakJ

akJ

k

ii

iriii

im

im

iim

im

i

r

im

im

iim

im

i


83/122

83

h

z

r

Infinite dielectric

slab

zi

ziirr

i

zi

k

kk

p

hk22

01 )1(tan

Forkz: approximated by the

TM01-mode wavenumber

Resonance frequency of cylindrical DRA

(i = 1, 2 forf1,f2)

wherep1 = 1 andp2 = 3

correspond to the HEM111 andHEM113 modes, respectively.

(4)

R. K. Mongia and P. Bhartia, Dielectric resonator antennas- a review and general design relations for

resonant frequency bandwidth,International Journal of Microwave and Millimeter-Wave Computer-

Aided Engineering, vol. 4, no. 3, pp 230-247, 1994.

Design formula of ratio h/a for given f f and


84/122

84

4

1

44

1

2

1

i

i

i

f

fBi

i

rr

S D

Ce

AE

a

hi

0

0

0

4444

3333

2222

1111

DCBA

DCBA

DCBA

EDCBA s

=

0996.1982.3162.123.19

02.160713.4511.115.36

07.3682402.42.6253.680

11650034800937.0234.07.489

(1)

f1 : HEM111 mode frequency (lower band)

f2 : HEM113 mode frequency (upper band)Ground plane

z

a

Cylindrical

DRA

h

r

Design formula of ratio h/afor given f1, f2, and r

Using the covariance matrix adaptation

evolutionary strategy again,


85/122

85

i

i

a

hBi

ii

rr

S D

Ce

AE

fa

i

4

1

44

1r

1

2

c

0

0

0

4444

3333

2222

1111

DCBA

DCBA

DCBA

EDCBA s

0057.0114.6659.5429.4

0814.179764.00368.0152.0

01.3049973.0005.00571.0

109328.310700152.0751.1109.1

=

(2)

Design formula of radius a

Radius a can be found by inserting h/a into (2) below:

Aftera isfound, h can be determined from h/a.

Maximum error ofa: 2.1% for 1 h/a 3.5, 9 r 27

Maximum error ofh: 3.0% for 1.28 h/a 1.85, 9 r 27

A f i i A i


86/122

86

A. Example for dualband cylindrical DRA design

a = 17.9 mm & h = 42.5 mm

Given:f1 = 1.71 GHz (DCS:1.71- 1.88 GHz)

f2 = 2.4 GHz (WLAN:2.4 - 2.48 GHz),

r=9.4

Using formulas

(1) & (2)

C fi i f h d lb d LP DRA


87/122

87

Matching slotExcitation

stripVia

x

y

Cylindrical

DRAra

Aperture

Feedline

Ground plane

Wf

WsDsLs

a

h

x

z Cylindrical

DRAr

Matching slotVia

Feedline

d

Ground planeAperture

for via

Excitation strip

w

l

Configuration of the dualband LP DRA

Top view Side view

a = 18.7 mm, h = 42.5 mm, r= 9.4, l= 12.5 mm, w = 1 mm,

Ls = 20 mm, Ws = 1.5 mm, andDs = 12.75 mm.

Radius a has been slightly increased to reduce the merging effect

M d d Si l t d R fl ti ffi i t


88/122

88

1.6 1.8 2 2.2 2.4 2.6-30

-20

-10

0

Frequency (GHz)

HFSS Simulation

Measurement

Reflection Coefficient |S11| (dB)

Measured and Simulated Reflection coefficients


Lower band impedance bandwidth: 15.5% (1.70-2.00 GHz)

Upper band impedance bandwidth: 3.7% (2.39-2.48 GHz)

M d d i l t d di ti tt


89/122

89

oo

-40 -30 -20 -10 0

30

150

60

120

9090

120

60

150

30

180

o

o o

o

o o

o o

o o

o

dB

9090

150

180

o

o

o

o

dB

-40 -30 -20 -10 0

30

150

60

90

120

60

150

30

180

o o

o o

o

o o

o o

o

dB

120

Col-pol

Cross-polCross-pol

Simulation

H-plane (y-z plane)

Measurement

E-plane (x-z plane)

-x +x -y +y

0o 0o

-40 -30 -20 -10 0

30

150

60

120

9090

120

60

150

30

180

o

o o

o

o o

o o

o o

o

dB

9090

150

180

o

o

o

o

o

dB

-40 -30 -20 -10 0

30

150

60

90

120

60

150

30

180

o o

o o

o o

o o

o o

o

dB

120

Col-pol

Cross-pol

Cross-pol

Simulation

H-plane (y-z plane)

Measurement

E-plane (x-z plane)

-x +x -y +y

0o 0o


HEM111 mode: measured (1.8 GHz), simulated (1.8 GHz)


(a) (b)


Co-polarized fields > cross-polarized fields by more than 20 dB in the

boresight direction.

M d d i l t d i


90/122

90

1.6 1.65 1.7 1.75 1.8 1.85 1.90

2

4

6

Frequency (GHz)

Lower band gain (dBi)

HFSS Simulation

Measurement

Frequency (GHz)

Upper band gain (dBi)

2.3 2.35 2.4 2.45 2.5 2.550

2

4

6

8

1010

8

6

42

02.3 2.35 2.52.4 2.45 2.55

Measured and simulated gain

HEM111 mode: Maximum measured gain of ~6 dBi (1.75 GHz)

HEM113 mode: Maximum measured gain of ~ 8 dBi (2.43 GHz)

D lb d CP DRA


91/122

91

L2

L3

L1W2

W0

WS

W3 W1

LS

DS

Input port

Via

Cylindrical

DRAr

x

ya

quadrature

coupler

Dualband

grounding

via

To

planeGround

portIsolation

slotMatching

stripExcitation

viaTo grounding

Dualband CP DRA

a = 18.7 mm, h = 42.5 mm, r= 9.4, l= 12.5 mm, w = 1 mm,Ls = 21 mm, Ws = 1.5 mm,Ds =

12.75 mm, L1 = 26.9 mm,L2 =26.5 mm,L3 = 56.65 mm, W0 = 4.66 mm, W1 = 7.3 mm, W2 =

4.44 mm, and W3 = 0.46 mm.

a

h

x

z Cylindrical

DRAr

Matching slotVia

Feedline

d

Ground planeAperturefor via

Excitation strip

w

l

Top view Side view

M d d i l t d fl ti ffi i t


92/122

1.6 1.8 2 2.2 2.4 2.6-40

-30

-20

-10

0


Frequency (GHz)

HFSS Simulation

Measurement


Lower band bandwidth:18.9% (1.58-1.91 GHz).

Upper band bandwidth:7.8% (2.33-2.52 GHz).


M d d i l t d i l ti (AR )


93/122

93

1.6 1.65 1.7 1.75 1.8 1.85 1.90

2

4

6

8

10 Lower band AR (dB)

Frequency (GHz)

HFSS Simulation

Measurement

Upper band AR (dB)

Frequency (GHz)

2.3 2.4 2.50

2

4

6

88

6

4

2

02.3 2.4 2.5

Measured and simulated axial ratios (ARs)


Lower band AR bandwidth: 12.4% (1.67-1.89 GHz)

Upper band AR bandwidth: 7.4% (2.34-2.52GHz)

M d d i l t d di ti tt


94/122

94

-40 -30 -20 -10 0

30

150

60

120

9090

120

60

150

30

180

o

o o

o

o o

o

o

o o

o

dB

9090

150

180

o

o

o

o

o

dB

-40 -30 -20 -10 0

30

150

60

90

120

60

150

30

180

o o

o o

o o

o o

o o

o

dB

120

LHCP

RHCP RHCP

Simulation

H-plane (y-z plane)

Measurement

E-plane (x-z plane)

-x +x -y +y

0o 0o

-40 -30 -20 -10 0

30

150

60

120

9090

120

60

150

30

180

o

o o

o

o o

o

o

o o

o

dB

9090

150

180

o

o

o

o

o

dB

-40 -30 -20 -10 0

30

150

60

90

120

60

150

30

180

o o

o o

o o

oo

o o

o

dB

120

RHCP

LHCP

RHCP

Simulation

H-plane (y-z plane)

Measurement

E-plane (x-z plane)

-x +x -y +y

0o 0o


(a) (b)




LHCP fields > RHCP fields by ~20 dB in the boresight direction.

B Example for wideband cylidnrical DRA design


95/122

95

B. Example for wideband cylidnrical DRA design

a = 10.3 mm & h = 34.3 mm

Given:f1 = 2.90 GHz,f2 = 3.72 GHz, r= 9.4

Using formula(5) & (6)

Wideband LP cylindrical DRA


96/122

96

CylindricalDRAr

Conducting

feeding stripCoaxial

aperture

l

w h

a

Ground plane

z

x

y

2.8 3 3.2 3.4 3.6 3.8 4 4.2-40

-30

-20

-10

0Reflection Coefficient |S11| (dB)

Frequency (GHz)

HFSS Simulation

Measurement

Configuration Reflection coefficient

a = 10.3 mm, h = 34.3 mm, r= 9.4,

l= 12 mm, and w = 1 mm.

Good agreement

Measured impedance bandwidth:

23.5% (3-3.8 GHz)

Wideband LP cylindrical DRA



97/122

97

3 3.2 3.4 3.6 3.8 40

2

4

6

8

10

12

Frequency (GHz)

Antenna Gain (dBi)

HFSS Simulation

Measurement




Wideband CP cylindrical DRA


98/122

98

L1

L1W0

W0

W1

Input port

Excitation

strip

Isolation

port

Ground

plane

via

Excitation strip

Cylindrical

DRAr

Widebandquadraturecoupler

via

x

ya

a

h

x

z Cylindrical

DRAr

Via

d

Ground planeAperturefor via

Excitation strip

w

l

Wideband quadrature coupler

a = 10.3 mm, h = 34.3 mm, r= 9.4, l= 11.5 mm, w = 1 mm,

L1 = 14.67 mm, W0 = 1.94 mm, and W1 = 3.21 mm.

Wideband CP cylindrical DRA

Top view Side view

Wideband CP DRA


99/122

99

3 3.2 3.4 3.6 3.8 4

-30

-20

-10

0

Frequency (GHz)


HFSS Simulation

Measurement

3 3.2 3.4 3.6 3.8 40

2

4

6

8

Fre uenc (GHz

Axial ratio (dB)

HFSS Simulation

Measurement

Measured 3-dB AR bandwidth :

24.7% (3.05-3.91 GHz).

Measured impedance bandwidth:

25.5% (3.04-3.93 GHz).

Wideband CP DRA

Reflection coefficient Axial ratio


100/122

VI. Dualfunction DRAs


101/122

101

AdvantageSystem size and cost can be reduced by

using dualfunction DRAs.

Additional functions

- Packaging cover- Oscillator


102/122

102

Packaging Cover


103/122

103

Conventional

Aperture Metallic supports

for grounding

Dielectric resonatorantenna/oscillatorload and packagingcover

Metal ground

Power supply

z

y

Front view

hH

Microstrip

feedline

d

Transistor (other

RF components

not shown)

Proposal

Antenna Configuration


104/122

104

Antenna Configuration

Aperture

Feedline and the

RF/MIC circuits

Metal ground

Coaxial line

Dielectric resonator antenna

and packaging coverMetallic supportsfor grounding

H

hy

z

Side view

Coaxial

x

yMicrostrip

feedline

Aperture

L

W

a

b

Top view

Resonant frequencyf0 = 2.4GHz

Parameters:

Hollow DRA:

L=30mm, W=29mm,

H=15mm, & r=12

Metallic Cavity:

a = 15mm, b = 21.6mm, h = 5mmTop face : Duroid r=2.94

thickness 0.762mm

Aperture: 0.2063 e

Design Procedure (Simulation):


105/122

105

Design Procedure (Simulation):

Step 2Remove the lower center

portion concentrically to

form a notched DRA.

As a result, the resonant

frequency >2.4GHz

Step 3Cover the two sides with thesame material. Move the

frequency back to 2.4GHz

by increasing the thickness.

(thickness f0 )

Step 1Use the DWM to design

a solid rectangular DRA

at 2.4-GHz fundamental

TE111 Mode.

x

z

Experimental Verification:


106/122

106

Experimental Verification:

- Hard-clad foam (r1) is used to form the

container.

- ECCOSTOCK HiK Powder ofr=12 is used as

the dielectric material.

Return Loss and Input Impedance


107/122

107

2 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3

Frequency (GHz)

-60

-50

-40

-30

-20

-10

0

Return

Loss(dB)

Theory

Experiment

2 2.1 2.2 2 .3 2.4 2 .5 2.6 2 .7 2.8 2 .9 3

Frequency (GHz)

-60

-40

-20

0

20

40

60

80

InputImpedance(ohm)

(Passive hollow RDRA with a metallic cavity)

Good agreement.

Bandwidth ~ 5.6%.

Measured resonance frequency: 2.42GHz (error < 0.83%)

Radiation Patterns

(P i h ll DRA ith t lli it )


108/122

108

-40 -30 -20 -10 0

30

-150

60

-120

90-90

120

-60

150

-30

180

0

co-pol

cross-pol

-40 -30 -20 -10 0

30

-150

60

-120

90-90

120

-60

150

-30

180

0

cross-pol

co-pol

Theory

Experiment

(a) (b)

x-z plane y-z plane

(Passive hollow DRA with a metallic cavity)

Broadside TE111y

mode is observed.

Co-polarized fields generally stronger than the cross-

polarized fields by 20dBin the boresight direction.

Return Loss of the Active Integrated Antenna


109/122

109

2 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3

Frequency (GHz)

-40

-30

-20

-10

0

ReturnLoss(dB)

A A

DRA DRA

ReturnLoss

ReturnLoss

Receiver Transmitter

DRA (passive)

DRA (active, receiver)

DRA (active, transmitter)

Return Loss of the Active Integrated Antenna

Integrated with Agilent AG302-86 low noise amplifier (LNA)(gain of 13.6dB at 2.4GHz)

LNA prematched to 50 at the input.

A small hole is drilled on the ground plane to supply the DC bias to the LNA.

Amplified Radiation Pattern


110/122

110

-180 -120 -60 0 60 120 180

Observation angle (degree)

-80

-70

-60

-50

-40

-30

-20

-10

0

Amp

litude(dB)

co-pol (passive)

co-pol (active)

cross-pol (passive)

cross-pol (active)

Amplified Radiation Pattern

Compared to the passive DRA,the active DRA has a gain of

7 - 12dB across the observation angle from -90o to 90o.

The gain is less than the specification due to unavoidable

impedance variations and imperfections in the measurement.


111/122

Dielectric Resonator Antenna

Oscillator (DRAO)

Methodology


112/122

112

The DRA is used as the oscillator load,

named as DRAO.

Methodology

The reflection amplifier method is used to

design the antenna oscillator.

DRAO Schematic Diagram


113/122

113

DRAjXZin=Rin+X in

Lm

Transistor

C

ZL=RL+XL

DRAO Schematic Diagram

- Oscillate condition:XL+Xin=0 &RL


114/122

114

L

w

Top view

Microstrip

feedlineAperture

y

x

Lm Ls

La

Wa

Transistor (other

RF components

not shown)

fo = 1.85GHz at TE111y

Parameters:DRA

L=52.2mm,

W=42.4mm,

H=26.1mm,

r= 6.

Aperture

La = 0.3561e, Wa = 2mm

Ls = 9.5 mm, Lm = 40 mm.

Duroid substrate

rs=2.94, d=0.762mm

antenna and

oscillating load

Microstrip

feedline Aperture

Ground

L

H

z

x

Substrate

Side view

d

Transistor (other

RF components

not shown)

r



115/122

115


Good agreement.

Bandwidth ~ 22.14%.

Resonance frequency: Measured 1.86GHz

Simulated 1.83GHz (1.5% error).

1.6 1.7 1.8 1.9 2 2.1 2.2-60

-50

-40

-30

-20

-10

0

Frequency (GHz)

Return Loss (dB)

HFSS Simulation

Experiment1.6 1.7 1.8 1.9 2 2.1

-80

-60

-40

-20

0

20

40

60

80

Frequency (GHz)

Input Impedance (ohms)

Spectrum of the Free-running DRAO


116/122

116

p g

Transmitting powerPt= 16.4dBm

DC-RF efficiency: ~ 13% (2-25% in the literature).

Phase noise: 103dBc/Hz at 5MHz offset

Second harmonic < fundamental by 22dB

Radiation Pattern


117/122

117

Broadside TE111y

is observed.

Co-polarized fields are generally 20dB stronger

than the cross-polarized fields in the boresight direction.

Solid DRAO (measur.)

Passive Solid DRA (measur.)

HFSS Simulation

-40 -30 -20 -10 0

30

60

9090

120

150

-30

180

0

-60

-120

-150

co-pol

cross-pol

x-z plane

(a)

-40 -30 - 20 -10 0

30

60

90-90

120

150

-30

180

0

-60

-120

-150

y-z plane

(b)

co-pol

cross-pol


118/122

118

DRA can be of any shape. Can it be made like a swan?

Yes!

DRA is simple made of dielectric. Can glass be used forthe dielectric?

It leads to probably the most beautiful antenna in

the world .

Yes!

Glass-Swan DRA


119/122

Glass Swan DRA

Distinguished Lecture

Transparent antennas: From 2D to 3D119

Conclusion


120/122

120

The DRA can be easily excited with various excitation schemes.

Frequency tuning of the DRA can be achieved by using

a loading-disk or parasitic slot.

The dualband and wideband DRAs can be easily designed using

higher-order modes.

Compact omnidirectional CP DRAs have been presented

Dualfuncton DRAs for packaging and oscillator designs have

been demonstrated.


121/122

121


122/122

Q & A

Documents

Development of Dielectric Resonator Antenna