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Development of Dielectric Resonator Antenna (DRA)Development of Dielectric Resonator Antenna (DRA)
K. W. LeungState Key Laboratory of Millimeter Waves &
Department of Electronic Engineering,City University of Hong Kong
I. Introduction
II. Circularly Polarized DRA Using a Parasitic Strip
III. Frequency Tuning Technique
IV. Omnidirectional Circularly Polarized DRAs
V. Dualband & Wideband DRAs
VI. Dualfunction DRAs
Outline
3
• The DRA is an antenna that makes use of a radiating mode of a dielectric resonator (DR).
• It is a 3-dimensional device of any shape,e.g., hemispherical, cylindrical, rectangular,triangular, etc.
• Resonance frequency determined by the its dimensions and dielectric constant r.
What is Dielectric Resonator Antenna (DRA) ?
4
Some DRs :
5
Advantages of the DRA
• Low cost• Low loss (no conductor loss)• Small size and light weight• Reasonable bandwidth (~10% for r ~10)• Easy of excitation• High radiation efficiency ( generally > 95%)
6
Excitation schemes
Ground plane
Dielectric substrate
DRAMicrostripfeed line
(i) Microstrip line feed
7
Ground plane
Microstripfeed line
FeedSubstrate
DRA
Slot
(ii) Aperture-couple feed
Excitation schemes
8
Ground plane
Coaxial probe
DRA
(iii) Coaxial feed
Excitation schemes
9
Bottom viewTop view
Coaxial feed
10
Bottom view Top view
Aperture-coupled feed
11
Slot-fed DRA array using corporate microstrip feed network
Corporate feedline for DRA array
12
Conformal-Strip Method
Ground plane
HemisphericalDRA
Conducting conformal strip
la
W
Rectangular Dielectric Resonator Antennas
Rectangular Dielectric Resonator Antennas
14
Proposed Antenna Geometry
W1
l1
a
d
b
x
y
zGroundPlane
RectangularDRA
Coaxial Aperture
Conducting Strip
r
a(mm)
b(mm)
d(mm)
l1(mm)
W1(mm) r
14.3 25.4 26.1 10 1 9.8
Resonant frequency of TEmnl(y) mode
Analytical Solution
• Dielectric Waveguide Model (DWM)
2220 2 zyx
r
kkkcf
dlk
bnk
amk zyx 2
,,
20
222 kkkk rzyx
16
Numerical Solution
Advantages- Very simple- High modeling capability for general EM structures- No spurious modes nor large matrix manipulation- Provide a very wideband frequency response
• Finite-Difference Time-Domain (FDTD) method
Disadvantages- Time consuming, powerful computer required
17
Baseband Gaussian pulse
])3(exp[ 22 TTntEz T : pulse width
0
011
)()(
ZZZZS
tIFFTtVFFTZ
in
in
in
Conformal Strip
Ground Plane
Ez (V)Hy Hy
HxHx
(I)
Source occupies only one grid
Source model and extraction of S parameters
18
Uniform Cartesian grids
T = 0.083ns, t0 = 3T
10-cell-thick PML with polynomial spatial scaling (m = 4 and κmax = 1)
total grid size : 80∆x × 110∆y × 112∆z
total time steps : 10000
∆x = 0.715 mm, ∆y = 0.508 mm, ∆z = 0.5 mm
Parameters
19
3 3.5 4 4.5 5 5.5 6-100
-50
0
50
100
150
200
Frequency (GHz)
2 2.5 3 3.5 4 4.5 5 5.5 6-30
-25
-20
-15
-10
-5
0
ExperimentTheory
Frequency (GHz)|S
11|(d
B)
Inpu
t Im
peda
nce
(ohm
)
ExperimentTheory
Resistance
Reactance
Input Impedance/S11
• Reasonable agreement.• Wide Bandwidth of ~ 43%.• Dual resonant TE111
y and TE113y modes are excited.
20
y111TE
y113TE
Resonant Modes
Measured resonant
frequencies
Calculated resonant frequencies (FDTD)
Predicted resonant frequencies (DWM)
fmea (GHz) fFDTD(GHz)
error (%)
fDWM(GHz)
error (%)
3.81 3.90 2.3 3.95 3.6
N/A N/A N/A 4.26 N/A
4.57 4.60 0.7 4.7 1.7
Comparison between Theory and Measurement
y112TE
• Reasonable agreement.
21
Field Distribution --- TE111y
Electric fieldMagnetic field
z
2d x
bx-z
b x
y
a
x-y
Imaged DRA (gound plane removed) With gound plane
22
Field Distribution --- TE112y
x-z
b x
y
a
x-yElectric fieldMagnetic field
z
2d x
b
Imaged DRA (gound plane removed)
23
Field Distribution --- TE113y
x-zElectric fieldM agnetic field
b x
y
a
x-y
z
2d x
b
With gound planeImaged DRA (gound plane removed)
24
E (xz) - plane H (yz) - plane(+x)(-x) (-y) (+y)-40
0o
-30-20-10 0
30o30o
60o60o
90o 90o
-40 -30-20-10 0
0o
30o30o
60o60o
90o 90o
E (xz) - plane H (yz) - plane(+x) (-y)(-x) (+y)
-40 -30-20-10 0
0o
30o30o
60o60o
90o 90o
-40 -30-20-10 0
0o
30o30o
60o60o
90o 90o
f = 3.5 GHz
f = 4.3 GHz
Radiation Patterns
• Broadside radiation patterns are observed.• Measured E-plane crosspolarized fields mainly caused by finite ground plane diffraction.
III. Circularly Polarized Design using a Parasitic Strip
III. Circularly Polarized Design using a Parasitic Strip
26
Proposed Antenna Geometry
a(mm)
b(mm)
d(mm)
l1(mm)
W1(mm)
l2(mm)
W2(mm)
0(degree) r
24 23.5 12.34 10 1 12 1 225.6 9.5
W1
l1
a d
bx
y
z
GroundPlane
RectangularDRA
Coaxial Aperture Conducting
Strip
r
C
W2
l2 ParasiticPatch
(Center of Parasitic Patch)
0
Top ViewParasitic
Patch
ConductingStrip
x
y
r
27
Input Impedance/S11
• Reasonable agreement.• Bandwidth ~ 14%.• Two nearly-degenerate TE111(y) modes are excited. CP operation
2.5 3 3.5 4 4.5-25
-20
-15
-10
-5
0
|S11
| (dB
)
Frequency (GHz)
28
Axial Ratio in the boresight direction
3-dB AR bandwidth is ~ 2.7%, which is a typical value for a singly-fed CP DRA.
3 3.1 3.2 3.3 3.4 3.5 3.6 3.7 3.80
5
10
15
20A
xial
Rat
io (d
B)
Frequency (GHz)
ExperimentTheory
The H field of the DRA without and with parasitic strip (Top view)
Without parasitic strip - LP field With parasitic strip - CP field
3.4 GHz 3.4 GHz
Feeding strip Feeding strip
Parasitic strip
29
30
Radiation Patterns (f = 3.4GHz, )
LHCP
RHCPxz plane yz plane
(+x)(-x) (-y) (+y)-40 -30 -20-10 0
0o
30o30o
60o60o
90o 90o
-40 -30 -20-10 0
0o
30o30o
60o60o
90o 90o
• A broadside radiation mode is observed.• For each radiation plane, the LHCP field is more than 20dB
stronger than the RHCP field.• The maximum gain is 5.7 dBic (not shown here).
31
Effects of feeding strip length l1
• Input impedance changes substantially with l1. • AR is almost unchanged for different l1. • l1 can be adjusted to match the impedance without changing AR.
2.4 2.6 2.8 3 3.2 3.4 3.6 3.8 4 4.2-20
-15
-10
-5
0
|S11
| (dB
)
Frequency (GHz)
l1= 8 mml1= 10 mml1= 12 mm
3.1 3.2 3.3 3.4 3.5 3.6 3.70
5
10
15
20
Axi
al R
atio
(dB
)Frequency (GHz)
l1= 8 mml1= 10 mml1= 12 mm
II. Frequency Tuning TechniqueII. Frequency Tuning Technique
The DRA for a paticular frequency may not be available from the comericial market.
Fabrication tolerances cause errors between measured and calculated resonant frequencies.
Frequency tuning methods: (i) loading-disk; and(ii) parasitic slot.
Backgruond
Frequency Tuning Technique- using a loading diskFrequency Tuning Technique- using a loading disk
Side view Top view
The slot-coupled DRA with a conducting loading cap
d
z
Slot
Hemispherical DRA
Dielectric substrate rs
microstrip line
Lt
Ground planea
ra
xConducting Loading Cap
Lt
x
y
Ls
Ws
Hemispherical DRA
microstrip line
Slot
Wf
Conducting Loading Cap
Hemispherical DRA: radius a = 12.5 mm, dielectric constant εr = 9.5.Coupling slot : length Ls , width WsOpen-circuit stub: length LtGrounded dielectric slab: εrs = 2.33, height d = 1.57 mmMicrostrip feedline: width Wf = 4.7 mm
Calculated and measured return losses(Ls = 12 mm and Ws = 1 mm)
3 3.2 3.4 3.6 3.8-70
-60
-50
-40
-30
-20
-10
0
Frequency (GHz)
|S11
| (dB
) = 26.38Lt = 4.42 mm
o = 52.8Lt = 13.6 mm
o
Theory
Experiments
Without cap ( = 0 )Lt = 10.63mm
o
Resonance frequency: 3.52 GHz without any conducting cap (α = 00), with Lt = 4.42 mm 3.25 GHz (α = 26.38o and Lt = 4.42 mm) 3.68 GHz (α = 52.8o and Lt = 13.6 mm)
Calculated and measured radiation patterns
-40 -30 -20 -10 0
030
60
90
30
60
90o o
o o
o o
o
(a) (b)-40 -30 -20 -10 0
030
60
90
30
60
90o o
o o
o o
oCo-pol
X-pol
-40 -30 -20 -10 0
030
60
90
30
60
90o o
o o
o o
o
(c)-40 -30 -20 -10 0
030
60
90
30
60
90o o
o o
o o
o
(d)
Co-pol
X-pol
3.58 GHz (α = 52.8o and Lt = 13.6 mm)
Reasonable agreementbetween theory andexperiment.
The effect of loadingcap on field pattern is notsignificant.
3.25 GHz (α = 26.38o and Lt = 4.42 mm)
Calculated α and Lt for having a good return loss (minimum |S11| < -20dB)
3.2 3.3 3.4 3.5 3.6 3.70
10
20
30
40
50
60
2
4
6
8
10
12
14
16
Tuning frequency fr (GHz)
Ang
le (d
egre
e)
Stub
Len
gth
L t(m
m)
The resonant frequency can be tuned by varying α and Lt α decreases from 26.38o to 0o (3.25 < fr < 3.5 GHz ) α increases from 0o to 52.8o (3.5 < fr < 3.78 GHz)
Impedance bandwidth
The bandwidth decreases after a loading cap is added.
3.2 3.3 3.4 3.5 3.6 3.72
4
6
8
10
Frequency (GHz)
Impe
danc
e Ba
ndw
idth
(%)
Frequency Tuning Technique- using a parasitic slot
Frequency Tuning Technique- using a parasitic slot
(a) Side view
(b) Top view
The annular-slot-excited cavity-backed DRA
z
x
Hemispherical DRA
Coaxial cable Parasitic slot
ar
metallic cavity b
Ground plane
WB WA
Feeding slotrB
rA
Coaxial cable
Hemispherical DRA Feeding slot
a
r
metallic cavity
b
Parasitic slot
A
rA rB
WB
WA
y
x
IV. Omnidirectional CircularlyPolarized DRA
IV. Omnidirectional CircularlyPolarized DRA
43
CP DRAs concentrated on broadside-mode designs only.
Provide larger coverage.
Advantages of omnidirectional CP antenna
44
Slotted omnidirectional CP DRA
Design I:
45
Antenna configurations
x
h
g
z
w
l
12r
Perspective view Front view
Dielectric cube with oblique slots (polarizer) fabricated onits four sidewalls.
Centrally fed by a coaxial probe extended from a SMAconnector, whose flange used as the small ground plane.
Wave polarizer
LP omnidirectional DRA Dielectric block with the wave polarizer
Proposed compact omnidirectional CP DRA
x
y
z
Dielectric slabs
D
E
+
Antenna principle
47
Prototype for 2.4 GHz WLAN design
Top face and sidewalls Bottom face
Design parametersr = 15, a = b = 39.4 mm, h = 33.4 mm, w = 9.4 mm,d =14.4 mm, r1 = 0.63 mm, l = 12.4 mm, g = 12.7 mm
Photographs of the prototype
48
Simulated and measured results
Reflection coefficient
Impedance bandwidth: AR bandwidth:Simulated: 20.3% (2.34-2.87 GHz) Simulated: 8.2% (2.34-2.54 GHz)Measured: 24.4% (2.30-2.94 GHz) Measured: 7.3% (2.39-2.57 GHz)
Axial ratio
2 2.2 2.4 2.6 2.8 3
-30
-20
-10
0
Frequency (GHz)
|S11| (dB)
HFSS SimulationExperiment
Axial Ratio (dB)
HFSS SimulationExperiment
2 2.2 2.4 2.6 2.8 30
3
6
9
Frequency (GHz)
49
Very good omnidirectional characteristic In the horizontal plane, LHCP fields > RHCP fields by
~20 dB .
Simulated and measured radiation patterns
LHCP
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
o
o o
o
o o
o o
o oo
dB
xz-plane
0o
RHCP
(+x)(-x)
-40 -30 -20 -10 0
30
210
60
240
90270
120
300
150
330
180
o
o o
o
o o
o o
o oo
dB
xy-plane
0o
RHCP
50
Simulated and measured antenna gain
2 2.2 2.4 2.6 2.8 3-3
-2
-1
0
1
2
3
Frequency (GHz)
Gain (dBic)
HFSS SimulationExperiment
51
Wideband omnidirectional CP antenna with parasitic metallic strips
Design II:
52
Perspective view Front view
x
h
g
z
ls
ws
wl
12r
Four parasitic metallic strips are embedded in thelateral slots to enhance the AR bandwidth.
The hollow circular cylinder is introduced to enhancethe impedance bandwidth.
Antenna configurations
53
Photographs of the prototype
Top face and sidewalls Bottom face
Design parametersr = 15, a = b = 30 mm, h = 25 mm, r = 3 mm, w = 7 mm, d =10.5 mmls = 30.5 mm, ws = 1 mm, x0 = 6.4 mm, r1 = 0.63 mm, l = 19 mm.
Prototype for 3.4 GHz WiMAX design
54
Overlapping bandwidth: 22.0%; bandwidth widened by ~3 times.
Simulated and measured reflection coefficient and axial ratio
Axial Ratio (dB)
2.8 3.2 3.6 4-30
-20
-10
0
Frequency (GHz)
|S11| (dB)
HFSS SimulationExperiment
2.8 3.2 3.6 40
6
12
18
Frequency (GHz)
Impedance bandwidth: Simulated: 22.3% (3.11-3.89 GHz) Measured: 24.5% (3.08-3.94 GHz)
AR bandwidth:Simulated: 24.8% (3.11-3.99 GHz) Measured: 25.4% (3.16-4.08 GHz)
55
Measured gain: wider bandwidth. Measured antenna efficiency: 84-98% (3.1-3.9 GHz).
Antenna gain
Simulated and measured results
Radiation efficiencyGain (dBic)
HFSS SimulationExperiment
2.8 3.2 3.6 4-4
-3
-2
-1
0
1
2
3
Frequency (GHz)2.8 3.2 3.6 40
0.2
0.4
0.6
0.8
1
Frequency (GHz)
Efficiency
56
3.4 GHz 3.8GHz
LHCP fields > RHCP fields by more than 15 dB in horizontal plane. Stable radiation patterns across the entire passband (3.1 – 3.9 GHz).
LHCP
-40 -30 -20 -10 0
30
210
60
240
90270
120
300
150
330
180
o
o o
o
o o
o o
o oo
dB
xy-plane
0o
RHCP
-40 -30 -20 -10 0
30
150
60
120
90 90
120
60
150
30
180
o o
o o
o o
o o
o oo
dB
xz-plane
0o
RHCP
(+x)(-x)
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
o
o o
o
o o
o o
o oo
dB
xz-plane
0oLHCP
RHCP
(+x)(-x)
-40 -30 -20 -10 0
30
210
60
240
90270
120
300
150
330
180
o
o o
o
o o
o o
o oo
dB
xy-plane
0o
RHCP
Simulated and measured radiation patterns
V. Dualband & Wideband DRAsV. Dualband & Wideband DRAs
(i) Rectangular DRA(i) Rectangular DRA
59
Dualband and wideband antennas are extensively used(e.g., WLAN)
Multi-element DRA [1]
- requiring more DR elements and space
Hybrid slot-DRA [2]
- coupling slot used as the feed and antenna
- inflexible in matching the impedance [1] Petosa, N. Simons, R. Siushansian, A. Ittipiboon and C. Michel, “Design and analysis of
multisegmentdielectric resonator antennas,” IEEE Trans. AP, vol.48, pp.738-742, 2000. [2] Buerkle, K. Sarabandi, and H. Mosallaei, “Compact slot and dielectric resonator antenna with
dual-resonance, broadband characteristics,” IEEE Trans. AP , vol. 53, pp.1020-1027, 1983.
Background
60
• Wideband DRA [1]
• Dualband DRA [2]
• Trial-and-error approach is normally used
• Systematic design approach is desirable[1] B. Li and K. W. Leung, “Strip-fed rectangular dielectric resonator antennas with/without a
parasitic patch,” IEEE Trans. Antennas Propagat., vol.53, pp.2200-2207, Jul.2005. [2] T. H. Chang and J. F. Kiang, “Dual-band split dielectric resonator antenna,” IEEE Trans.
Antennas Propagat., vol.55, no.11, pp.3155-3162, Nov.2007.
Use of higher-order DRA mode
61
Design Formulas for Dual-Mode rectangular DRA
yTE111yTE112
yTE113
The E-field should vanish on the PEC and the TE112 mode cannot be excited properly.
The TE111 mode and TE113 mode are used in the dual-mode design.
62
xy
z
ba
d
Ground plane
Rectangular DRA r
Formula Derivation
The wavenumbers kx1, x2 andkz1, z2 can be written as follows:
akk xx
21
dkz 21
dkz 2
32
From the DWM model, the frequencies f1, f2 are given by:
22,1
22,1
22,12,1 2 zzyyxx
r
kkkcf
22,1
22,1
22,12,1 zzxxyy kkkk
in which are wavenmubers in the dielectric, with c being the speed of light in vacuum.
where
cfk r /2 2,12,1
(*)
64
dkk
d
21
22
2
)96.3(
22
21
1
2
32.109
32.10 ff
kka
21 35.065.0 bbb
Engineering Formulas for the DRA dimensions
2
1
23
1
24
1
2 4422.113209.21393.0ff
ff
ffd 3
1
2 104437.184984.23
ff
(m)
111tan22
2,1
2,11
2,12,1
yyryy kk
kb
where
65
)96.3(
22
21
1
2
32.109
32.10 ff
kka
Limit of frequency ratio f2/f1
From
3k1 > k2 or 3f1 > f2
We have
givingf2/f1 < 3
which is the theoretical limit that is not known before.
09 22
21 dkk
66
1 1 .2 1 .4 1 .6 1 .8 2 2 .2 2 .4 2 .6 2 .80
1
2
3
4
5r 10r 3 0r 7 0r
E r ro r o f (% ) 1f
f f2 1 /
Compared with DWM results, errors of f1, f2 are both less than 2.5% for 1 < f2/f1≤2.8 , 5 ≤εr ≤70.
f1 kept constant at 2.4 GHz.
Error analysis
67
A. Example for Dual-band Rectangular DRA Design
a = 20.8 mm, b = 10.5 mm, and d = 18.5 mm.
Given: f1 = 3.47 GHz (WiMax)f2 = 5.2 GHz (WLAN), εr=10
Using dual-modeformulas
68
Configuration of the dualband DRA
b
ad
Ground plane
Microstripline
Couplingslot
LS W
L
Wf
h
Substrate rs
Rectangular DRA r
z
y
x
W = 2.6 mm, L =10.6 mm, Ls=7.2 mm, Wf=1.94 mm, h=0.762mm, εrs= 2.93
69
Measured and simulated reflection coefficients
Measured bandwidths: Lower band: 15% (3.25-3.78 GHz) covering WiMAX (3.4-3.7 GHz).Upper band: 8.3% (5.03-5.47 GHz) covering WLAN (5.15-5.35 GHZ).
3.2 3.6 4 4.4 4.8 5.2 5.6-40
-30
-20
-10
0
HFSS Simulation Measurement
Frequency (GHz)
Reflection coefficient |S | (dB)11
70
COMPARISON OF DESIGN, SIMULATED, AND MEASUREDRESONANCE FREQUENCIES OF TE111
y AND TE113y MODES
ResonantMode
Measured frequency
(GHz)
Design frequency
Simulated HFSS frequency
f1,2(GHz)
Error (%)
fHFSS(GHz)
Error(%)
TE111y 3.40 3.47 2.05 3.47 2.05
TE113y 5.18 5.30 2.32 5.24 1.15
71
TE111y mode: measured (3.40 GHz), simulated (3.47 GHz).
Broadside radiation patterns are observed for both planes. Co-polarized fields > cross-polarized fields by more than 20 dB in
the boresight direction.
Measured and simulated radiation patterns
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
0oo
o o
o
o o
o o
o oo
dB
90 90
150180
o
o
o
oo
dB-40 -30 -20 -10 0
30
150
60
90
120
60
150
30
180
0 oo o
o o
o o
o o
o oo
dB
(a)Simulation
H-plane (y-z plane)Measurement
E-plane (x-z plane)-x
120
Co-pol
Cross-pol
Co-pol
Cross-pol
+x -y +y
72
Measured and simulated radiation patterns
TE113y mode: measured (5.18 GHz), simulated (5.24 GHz).
Broadside radiation patterns are observed for both planes. Co-polarized fields > cross-polarized fields by more than 20 dB in
the boresight direction.
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
0oo
o o
o
o o
o o
o oo
dB
90 90
150180
o
o
o
o
o
dB-40 -30 -20 -10 0
30
150
60
90
120
60
150
300 o
o o
o o
o o
o o
o o
o
dB
(b)
120
180 o180 o
Simulation H-plane (y-z plane)
Measurement E-plane (x-z plane)
-x +x -y +y
Co-polCo-polCross-pol Cross-pol
73
TE111y mode: Maximum gain of 4.02 dBi at 3.48 GHz.
TE113y mode: Maximum gain of 7.52 dBi at 5.13 GHz.
Electrically larger antenna has a higher antenna gain.
Measured antenna gain
3 3 .5 4 4 .5 5 5 .5 6-5
0
5
10G ain (dBi)
Frequen cy (G Hz)
74
B. Example for Wideband DRA Design
a = 30.7 mm, b = 24.7 mm, and d = 47.7 mm.
Given: f1 = 1.98 GHz (PCS)f2 = 2.48 GHz (WLAN), εr=10
Using formulas for dual-moderectangular DRA
75
Configuration of the wideband DRA
b
Conducting feeding strip
Coaxial aperture
a
dW
l
Ground plane
Rectangular DRA r
x
y
z
l = 17 mm, W = 1 mm
76
Measured and simulated reflection coefficients
Measured bandwidths : 30.9% (1.83-2.50 GHz)PCS (1.85-1.99 GHz), UMTS (1.99-2.20 GHz) & WLAN (2.4-2.48 GHz)
1.5 2 2.5 3-40
-30
-20
-10
0
HFSS Simulation Measurement
Reflection coefficient |S | (dB)11
Frequency (GHz)
77
Measured and simulated radiation patterns
Measured (2.16 GHz), simulated (2.11 GHz). Broadside radiation patterns are observed. Co-polarized fields > cross-polarized fields by more than 20 dB in the boresight direction.
-40 -30 -20 -10 0
30
150
60
120
90 90
120
60
150
30
180
0oo
o o
o
o o
o o
o oo
dB
90 90
150180
o
o
o
oo
dB-40 -30 -20 -10 0
30
150
60
90
120
60
150
30
180
0 oo o
o o
o o
o o
o oo
dB
120
Simulation H-plane (y-z plane)
Measurement E-plane (x-z plane)
-x +x -y +y
Co-pol Co-polCross-pol Cross-pol
(a)
78
Measured and simulated radiation patterns
Measured (2.41 GHz), simulated (2.46 GHz). Broadside radiation patterns are observed. Co-polarized fields > cross-polarized fields by more than 20 dB in the boresight direction.
-40 -30 -20 -10 0
30
-150
60
-120
90-90
120
-60
150
-30
180
0oo
o o
o
o o
o o
o oo
dB
90-90
150180
o
o
o
oo
dB-40 -30 -20 -10 0
30
-150
60
90
120
-60
150
-30
180
0 oo o
o o
o o
o o
o oo
dB
H-plane (y-z plane)Measurement
E-plane (x-z plane)Simulation
Co-pol Co-polCross-pol Cross-pol
(b)
-x +x -y +y
79
Measured antenna gain
The maximum gain of 6.98 dBi at 2.47GHz. TE113
y -mode gain > TE111y -mode gain.
1.6 1.8 2 2.2 2.4 2.6 2.80
2
4
6
8
Frequency (GHz)
Gain (dBi)
(ii) Cylindrical DRA(ii) Cylindrical DRA
81
20
22irzii kkk
Ground planez
aCylindrical DRA
h
r
kρi & kzi :dielectric wavenumbers along the & z directions
k0i = 2fi/c : wavenumber in air
(1)
Resonance frequency of the HEMmnr mode of the cylindrical DRA
i = 1, 2 for f1, f2
f1 : HEM111 mode frequencyf2 : HEM113 mode frequency
82
z
x
y
r
a
Infinite dielectric rod
Resonance frequency of the HEMmnr mode of the cylindrical DRA
For k:
220)1(' iiri kkk
where
is the radial wavenumber outside the dielectric rod
Jm(x) : Bessel function of the first kind Km(x): modified Bessel function of the second kind.
(2)
(3)
D. Kajfez and P. Guillon, “Dielectric resonators,” Norwood, MA, Artech House, Inc., 1986.
24
22222
)'()')('(
'''
'1
)('
'''
'1
)('1
akkkkkkm
akKakK
kakJakJ
kakKakK
kakJakJ
k
ii
iriii
im
im
iim
im
i
r
im
im
iim
im
i
83
h
z
r
Infinite dielectric slab
zi
ziirr
i
zi
kkk
phk 22
01 )1(tan
For kz: approximated by the TM01-mode wavenumber
Resonance frequency of cylindrical DRA
(i = 1, 2 for f1, f2)
where p1 = 1 and p2 = 3 correspond to the HEM111 and HEM113 modes, respectively.
(4)
R. K. Mongia and P. Bhartia, “Dielectric resonator antennas- a review and general design relations for resonant frequency bandwidth,” International Journal of Microwave and Millimeter-Wave Computer-Aided Engineering, vol. 4, no. 3, pp 230-247, 1994.
84
4
144
1
2
1i
i
iffB
ii
rr
S DCe
AEah
i
000
4444
3333
2222
1111
DCBADCBADCBA
EDCBA s
=
0996.1982.3162.123.1902.160713.4511.115.3607.3682402.42.6253.680
11650034800937.0234.07.489
(1)
f1 : HEM111 mode frequency (lower band)f2 : HEM113 mode frequency (upper band)
Ground planez
aCylindrical DRA
h
r
Design formula of ratio h/a for given f1, f2, and r
Using the covariance matrix adaptationevolutionary strategy again,
85
i
iahB
i
ii
rr
S DCe
AEf
ai
4
144
1r
12
c
000
4444
3333
2222
1111
DCBADCBADCBA
EDCBA s
0057.0114.6659.5429.40814.179764.00368.0152.001.3049973.0005.00571.0
109328.310700152.0751.1109.1
=
(2)
Design formula of radius a
Radius a can be found by inserting h/a into (2) below:
After a is found, h can be determined from h/a.
Maximum error of a: 2.1% for 1 h/a 3.5, 9 r 27Maximum error of h: 3.0% for 1.28 h/a 1.85, 9 r 27
86
A. Example for dualband cylindrical DRA design
a = 17.9 mm & h = 42.5 mm
Given: f1 = 1.71 GHz (DCS:1.71- 1.88 GHz )f2 = 2.4 GHz (WLAN:2.4 - 2.48 GHz ), εr=9.4
Using formulas (1) & (2)
87
Matching slotExcitation strip Via
x
y
Cylindrical DRA r
a
ApertureFeedline
Ground plane
Wf
WsDsLs
a
h
x
z Cylindrical DRA r
Matching slotVia
Feedline
d
Ground planeAperture for via
Excitation stripw
l
Configuration of the dualband LP DRA
Top view Side view
a = 18.7 mm, h = 42.5 mm, r = 9.4, l = 12.5 mm, w = 1 mm, Ls = 20 mm, Ws = 1.5 mm, and Ds = 12.75 mm.
Radius a has been slightly increased to reduce the merging effect
88
1.6 1.8 2 2.2 2.4 2.6-30
-20
-10
0
Frequency (GHz)
HFSS Simulation Measurement
Reflection Coefficient |S11| (dB)
Measured and Simulated Reflection coefficients
Reasonable agreement Lower band impedance bandwidth: 15.5% (1.70-2.00 GHz)Upper band impedance bandwidth: 3.7% (2.39-2.48 GHz)
89
oo-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
o
o o
o
o o
o o
o oo
dB
90 90
150180
o
o
oo
dB-40 -30 -20 -10 0
30
150
60
90
120
60
150
30
180
o o
o o
o
o o
o oo
dB
120
Col-pol
Cross-pol Cross-pol
Simulation H-plane (y-z plane)
Measurement E-plane (x-z plane)
-x +x -y +y
0o 0o
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
o
o o
o
o o
o o
o oo
dB
90 90
150180
o
o
o
oo
dB-40 -30 -20 -10 0
30
150
60
90
120
60
150
30
180
o o
o o
o o
o o
o oo
dB
120
Col-pol
Cross-polCross-pol
Simulation H-plane (y-z plane)
Measurement E-plane (x-z plane)
-x +x -y +y
0o 0o
Measured and simulated radiation patterns
HEM111 mode: measured (1.8 GHz), simulated (1.8 GHz)HEM113 mode: measured (2.42 GHz), simulated (2.45 GHz)
(a) (b)
Broadside radiation patterns are observed.Co-polarized fields > cross-polarized fields by more than 20 dB in the boresight direction.
90
1.6 1.65 1.7 1.75 1.8 1.85 1.90
2
4
6
Frequency (GHz)
Lower band gain (dBi)
HFSS Simulation Measurement
Frequency (GHz)
Upper band gain (dBi)
2.3 2.35 2.4 2.45 2.5 2.550
2
4
6
8
10108
642
02.3 2.35 2.52.4 2.45 2.55
Measured and simulated gain
HEM111 mode: Maximum measured gain of ~6 dBi (1.75 GHz) HEM113 mode: Maximum measured gain of ~ 8 dBi (2.43 GHz)
91
L2
L3
L1W2
W0
WS
W3 W1
LS
DS
Input port
Via
Cylindrical DRA r
x
ya
quadraturecoupler
Dualband
groundingvia
To
planeGround
portIsolation
slotMatching
stripExcitation
viaTo grounding
Dualband CP DRA
a = 18.7 mm, h = 42.5 mm, r = 9.4, l = 12.5 mm, w = 1 mm, Ls = 21 mm, Ws = 1.5 mm, Ds = 12.75 mm, L1 = 26.9 mm, L2 =26.5 mm, L3 = 56.65 mm, W0 = 4.66 mm, W1 = 7.3 mm, W2 = 4.44 mm, and W3 = 0.46 mm.
a
h
x
z Cylindrical DRA r
Matching slotVia
Feedline
d
Ground planeAperture for via
Excitation stripw
l
Top view Side view
1.6 1.8 2 2.2 2.4 2.6-40
-30
-20
-10
0Reflection Coefficient |S11| (dB)
Frequency (GHz)
HFSS Simulation Measurement
Reasonable agreementLower band bandwidth:18.9% (1.58-1.91 GHz).Upper band bandwidth:7.8% (2.33-2.52 GHz).
Measured and simulated reflection coefficients
93
1.6 1.65 1.7 1.75 1.8 1.85 1.90
2
4
6
8
10Lower band AR (dB)
Frequency (GHz)
HFSS Simulation Measurement
Upper band AR (dB)
Frequency (GHz)2.3 2.4 2.5
0
2
4
6
88
6
4
2
02.3 2.4 2.5
Measured and simulated axial ratios (ARs)
Reasonable agreement Lower band AR bandwidth: 12.4% (1.67-1.89 GHz) Upper band AR bandwidth: 7.4% (2.34-2.52GHz)
94
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
o
o o
o
o o
o o
o oo
dB
90 90
150180
o
o
o
oo
dB-40 -30 -20 -10 0
30
150
60
90
120
60
150
30
180
o o
o o
o o
o o
o oo
dB
120
LHCP
RHCP RHCP
Simulation H-plane (y-z plane)
Measurement E-plane (x-z plane)
-x +x -y +y
0o 0o
-40 -30 -20 -10 0
30
150
60
120
9090
120
60
150
30
180
o
o o
o
o o
o o
o oo
dB
90 90
150180
o
o
o
oo
dB-40 -30 -20 -10 0
30
150
60
90
120
60
150
30
180
o o
o o
o o
o o
o oo
dB
120
RHCP
LHCP
RHCP
Simulation H-plane (y-z plane)
Measurement E-plane (x-z plane)
-x +x -y +y
0o 0o
Measured and simulated radiation patterns
(a) (b)
HEM111 mode: measured (1.8 GHz), simulated (1.8 GHz)HEM113 mode: measured (2.42 GHz), simulated (2.45 GHz)
Broadside radiation patterns are observed.LHCP fields > RHCP fields by ~20 dB in the boresight direction.
95
B. Example for wideband cylidnrical DRA design
a = 10.3 mm & h = 34.3 mm
Given: f1 = 2.90 GHz, f2 = 3.72 GHz, εr= 9.4
Using formula (5) & (6)
96
Cylindrical DRA r
Conducting feeding strip
Coaxial aperture
l
w h
a
Ground plane
z
xy
2.8 3 3.2 3 .4 3.6 3.8 4 4.2-40
-30
-20
-10
0Reflection Coefficient |S11| (dB)
Frequency (GHz)
HFSS Simulation Measurement
Configuration Reflection coefficient
a = 10.3 mm, h = 34.3 mm, r = 9.4, l = 12 mm, and w = 1 mm.
Good agreementMeasured impedance bandwidth:23.5% (3-3.8 GHz)
Wideband LP cylindrical DRA
97
3 3.2 3.4 3.6 3.8 40
2
4
6
8
10
12
Frequency (GHz)
Antenna Gain (dBi)
HFSS Simulation Measurement
Measured and simulated gain
HEM111 mode: Maximum measured gain of ~7 dBi (3.29 GHz) HEM113 mode: Maximum measured gain of ~10 dBi (3.83 GHz)
98
L1
L1W0
W0
W1
Input port
Excitationstrip
Isolationport
Groundplane
via
Excitation stripCylindrical DRA r
Wideband quadrature coupler
via
x
ya
a
h
x
z Cylindrical DRA r
Viad
Ground planeAperture for via
Excitation stripw
l
Wideband quadrature coupler
a = 10.3 mm, h = 34.3 mm, r = 9.4, l = 11.5 mm, w = 1 mm, L1 = 14.67 mm, W0 = 1.94 mm, and W1 = 3.21 mm.
Wideband CP cylindrical DRA
Top view Side view
99
3 3.2 3.4 3.6 3.8 4
-30
-20
-10
0
Frequency (GHz)
Reflection Coefficient |S11| (dB)
HFSS Simulation Measurement
3 3.2 3.4 3.6 3.8 40
2
4
6
8
Frequency (GHz)
Axial ratio (dB)
HFSS Simulation Measurement
Measured 3-dB AR bandwidth :24.7% (3.05-3.91 GHz).
Measured impedance bandwidth:25.5% (3.04-3.93 GHz).
Wideband CP DRA
Reflection coefficient Axial ratio
VI. Dualfunction DRAsVI. Dualfunction DRAs
101
AdvantageSystem size and cost can be reduced byusing dualfunction DRAs.
Additional functions- Packaging cover- Oscillator
102
Packaging Cover
103
Conventional
Aperture Metallic supportsfor grounding
Dielectric resonator antenna/oscillator load and packaging cover
Metal ground
Power supply
z
y
Front view
hH
Microstripfeedline
d
Transistor (other RF components not shown)
Proposal
104
Antenna Configuration
Aperture
Feedline and theRF/MIC circuits
Metal ground
Coaxial line
Dielectric resonator antenna and packaging cover Metallic supports
for grounding
H
hy
z
Side view
Coaxial
x
yMicrostrip feedline
Aperture
L
W
a
b
Top view
Resonant frequencyf0 = 2.4GHz
Parameters:• Hollow DRA:
L=30mm, W=29mm, H=15mm, & r =12
• Metallic Cavity:a = 15mm, b = 21.6mm, h = 5mmTop face : Duroid r =2.94
thickness 0.762mmAperture: 0.2063 e
105
Design Procedure (Simulation):
Step 2Remove the lower center portion concentrically to form a notched DRA. As a result, the resonant frequency >2.4GHz
Step 3Cover the two sides with thesame material. Move the frequency back to 2.4GHz by increasing the thickness.(thickness ↑ f0 ↓ )
Step 1Use the DWM to design a solid rectangular DRA at 2.4-GHz fundamental TE111 Mode.
x
z
106
Experimental Verification:
- Hard-clad foam (r ≈1) is used to form the container.
- ECCOSTOCK HiK Powder of r =12 is used as the dielectric material.
107
2 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3Frequency (GHz)
-60
-50
-40
-30
-20
-10
0R
etur
n Lo
ss (d
B)
Theory
Experiment
2 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3Frequency (GHz)
-60
-40
-20
0
20
40
60
80
Inpu
t Im
peda
nce
(ohm
)
Return Loss and Input Impedance(Passive hollow RDRA with a metallic cavity)
•Good agreement.•Bandwidth ~ 5.6%.• Measured resonance frequency: 2.42GHz (error < 0.83%)
108
-40 -30 -20 -10 0
30
-150
60
-120
90-90
120
-60
150
-30
180
0co-pol
cross-pol
-40 -30 -20 -10 0
30
-150
60
-120
90-90
120
-60
150
-30
180
0
cross-pol
co-pol
TheoryExperiment
(a) (b)
x-z plane y-z plane
Radiation Patterns(Passive hollow DRA with a metallic cavity)
• Broadside TE111y mode is observed.
• Co-polarized fields generally stronger than the cross-polarized fields by 20dB in the boresight direction.
109
2 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3Frequency (GHz)
-40
-30
-20
-10
0
Ret
urn
Loss
(dB
)
A A
DRA DRA
Return Loss
Return Loss
Receiver Transmitter
DRA (passive)
DRA (active, receiver)
DRA (active, transmitter)
Return Loss of the Active Integrated Antenna
• Integrated with Agilent AG302-86 low noise amplifier (LNA)(gain of 13.6dB at 2.4GHz)
• LNA prematched to 50 at the input.• A small hole is drilled on the ground plane to supply the DC bias to the LNA.
110
-180 -120 -60 0 60 120 180Observation angle (degree)
-80
-70
-60
-50
-40
-30
-20
-10
0A
mpl
itude
(dB
)
co-pol (passive)
co-pol (active)
cross-pol (passive)
cross-pol (active)
Amplified Radiation Pattern
• Compared to the passive DRA, the active DRA has a gain of7 - 12dB across the observation angle from -90o to 90o.
• The gain is less than the specification due to unavoidable impedance variations and imperfections in the measurement.
Dielectric Resonator Antenna Oscillator (DRAO)
Dielectric Resonator Antenna Oscillator (DRAO)
112
• The DRA is used as the oscillator load, named as DRAO.
Methodology
• The reflection amplifier method is used to design the antenna oscillator.
113
DRAjX Zin=Rin+Xin
Lm
Transistor
C
ZL=RL+XL
DRAO Schematic Diagram
- Oscillate condition: XL+Xin=0 & RL<|Rin|- DRA first replaced by a 50 load at 1.85GHz.
114
L
w
Top view
MicrostripfeedlineAperture
y
x
Lm Ls
La
Wa
Transistor (other RF components not shown)
Resonance frequencyfo = 1.85GHz at TE111
y
Parameters:DRAL=52.2mm, W=42.4mm, H=26.1mm, r = 6.
ApertureLa = 0.3561e, Wa = 2mmLs = 9.5 mm, Lm = 40 mm.
Duroid substraters=2.94, d=0.762mm
Antenna Configuration:Dielectric resonator antenna and oscillating load
Microstripfeedline Aperture
Ground
L
H
z
x
Substrate
Side view
d
Transistor (other RF components not shown)
r
115
Return Loss and Input Impedance
• Good agreement.• Bandwidth ~ 22.14%.• Resonance frequency: Measured 1.86GHz
Simulated 1.83GHz (1.5% error).
1.6 1.7 1.8 1.9 2 2.1 2.2-60
-50
-40
-30
-20
-10
0
Frequency (GHz)
Return Loss (dB)
HFSS SimulationExperiment 1.6 1.7 1.8 1.9 2 2.1
-80
-60
-40
-20
0
20
40
60
80
Frequency (GHz)
Input Impedance (ohms)
116
Spectrum of the Free-running DRAO
• Transmitting power Pt = 16.4dBm• DC-RF efficiency: ~ 13% (2-25% in the literature).• Phase noise: 103dBc/Hz at 5MHz offset• Second harmonic < fundamental by 22dB
117
Radiation Pattern
• Broadside TE111y is observed.
• Co-polarized fields are generally 20dB stronger than the cross-polarized fields in the boresight direction.
Solid DRAO (measur.)Passive Solid DRA (measur.)
HFSS Simulation
-40 -30 -20 -10 0
30
60
90-90
120
150
-30
180
0
-60
-120
-150
co-pol
cross-pol
x-z plane
(a)
-40 -30 -20 -10 0
30
60
90-90
120
150
-30
180
0
-60
-120
-150
y-z plane
(b)
co-pol
cross-pol
118
DRA can be of any shape. Can it be made like a swan?
Yes!
DRA is simple made of dielectric. Can glass be used for the dielectric?
It leads to probably the most beautiful antenna in the world …….
Yes!
Glass-Swan DRA
Distinguished LectureTransparent antennas: From 2D to 3D
119
120
• The DRA can be easily excited with various excitation schemes.
• Frequency tuning of the DRA can be achieved by usinga loading-disk or parasitic slot.
• The dualband and wideband DRAs can be easily designed usinghigher-order modes.
• Compact omnidirectional CP DRAs have been presented
• Dualfuncton DRAs for packaging and oscillator designs havebeen demonstrated.
Conclusion
121
122
Q & A
