Novel isolated cascaded half-bridge converter for battery energy storage systems

Felix Jauch and Jürgen Biela, Member, IEEE,

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Novel Isolated Cascaded Half-Bridge Converterfor Battery Energy Storage Systems

Felix Jauch, Jurgen BielaLaboratory for High Power Electronic Systems, ETH Zurich

Email: jauchf@ethz.chURL: http://www.hpe.ee.ethz.ch

AcknowledgmentsThe authors would like to thank Swisselectric Research and the Competence Center Energy and Mobility(UFCEV project) very much for their strong financial support of the research work.

Keywords�Converter circuit�, �Energy storage�, �Modulation strategy�, �Soft switching�, �Three-phase sys-tem�, �ZCZVS converters�

AbstractThis paper presents a novel isolated cascaded half-bridge converter for medium voltage battery energystorage systems with three phase legs of series connected modules in star connection. Each moduleconsists of a grid-side half-bridge, a battery-side full-bridge and a medium-frequency transformer pro-viding isolation. The presented modulation ensures grid-side zero-current-switching (ZCS) and battery-side zero-voltage-switching (ZVS). The control system consists of a grid current control as well asa state-of-charge (SOC) balancing control for the batteries. Evaluation of the proposed module of a6.6 kV, 150 kW, 1 MWh design example shows an average module efficiency of 97.4 % including semi-conductor and transformer losses. Simulation results verify the converter concept including the appliedcontrol structures.

IntroductionWith increasing use of renewable energy sources like wind power and solar energy, energy storage sys-tems become an important part of future energy distribution grids due to the inherently fluctuating andstochastic nature of renewable energy sources. In order to store considerable amounts of energy and

Module

Module

MV AC Grid

+-

StorageBattery

Module

Module

Module

+-

StorageBattery

+-

StorageBattery

+-

StorageBattery

Module

Module

Module

+-

StorageBattery

+-

StorageBattery

+-

Module +-

StorageBattery

+-

StorageBattery

StorageBattery

av bv cv

Lσ,cii

p,civ

3S 5S

4S

: 1n

+

a1S

b1S

a2S

b2S

σL

mL

6S

acC

acC

dcC

dcL

civBciv

ci

Bcii

s,civ

ai bi ci

acL acL acL

Figure 1: Novel isolated cascaded half-bridge converter for medium voltage battery energy storage systems with series connected modulesapplying grid-side half-bridges with bidirectional switches, battery-side full-bridges with unidirectional switches and medium-frequency trans-formers providing galvanic isolation. Either star or delta connection of module branches is possible. The module input voltage vci correspondsto a sinusoidal voltage waveform with the grid frequency.

ensure balancing supply and demand, suitable high power electronic equipment interfacing the storagemedia play an important role. Besides the well known energy storage systems using the potential energyof water, in recent years additional concepts based on compressed air, flywheels, thermal energy storagein molten salt as well as electrochemical storage in batteries have been investigated. The battery basedsystems offer the advantages of high energy and power density, high cycle efficiency as well as beinglocation-independent and easily scalable. Such systems also become more important in grid applicationsfor load leveling providing fast frequency regulation [1].

Battery energy storage systems at high power levels are usually connected to the medium voltage dis-tribution grid. Because of the relatively high voltages, multilevel converter systems are advantageous touse due to lower harmonics, robust operation and reduced switching losses. Furthermore, no bulky trans-former providing voltage adaptation is necessary. Especially the modular multilevel converter (MMC)[2] as well as the cascaded H-bridge converter (CHB) [3] offer the benefit to easily split and distributethe energy storage among several modules [4–6]. The storage battery is either attached directly to themodule DC-link [6] or connected via a subsequent (isolated) DC-DC converter [4]. In the case of usingisolated DC-DC converters, the batteries can be connected to earth. Moreover, their energy can be usedfor ultra-fast charging of electric vehicles as examined in [7–9].

The typical solution to integrate isolation into a module structure of a MMC or CHB is to connect a sub-sequent isolated DC-DC converter to the module DC-link by using a two-stage approach. In this paper,the H-bridge module of a CHB and the subsequent isolated DC-DC converter are integrated into a singleisolated module. For the proposed module topology, an optimal modulation scheme for a combinationof IGBTs/MOSFETs featuring grid-side zero-current-switching (ZCS) and battery-side zero-voltage-switching (ZVS) is introduced. Furthermore, an advanced module-level controller is presented.

First, the topology and the operating principle both on system and module level is described. After-wards, the control of the battery energy storage system including grid current and state-of-charge (SOC)balancing control for the batteries is presented. Components of a single- and a two-stage module for a6.6 kV, 150 kW, 1 MWh design example including their loss models are given and efficiencies of bothsolutions are calculated. Finally, simulation results that verify the converter concept including the appliedcontrol structures are presented.

Topology and Operating PrincipleThe proposed isolated cascaded half-bridge converter consists of N series connected modules per phaseleg with a grid-side half-bridge, a battery-side full-bridge and a medium-frequency transformer provid-ing galvanic isolation as shown in Fig. 1. On the grid side, because of the star connection of the modules,bidirectional switching devices are applied whereas on the battery side unidirectional switching devicesare used. The storage batteries are distributed among the modules and attached to their DC-links to easilyallow balancing. Due to the galvanic isolation, the batteries can be connected to earth for safety reasons.

Each of the modules is operated by phase-shift control like conventional dual active bridge (DAB) con-verters [7, 10, 11]. The input voltages of the modules exhibit a sinusoidal voltage waveform with the ACgrid frequency and with amplitudes according to the power distribution among the modules. The appliedmodulation scheme guarantees grid-side ZCS and battery-side ZVS over the whole AC line period simi-lar to the ZVS modulation presented in [11]. The modules in the same phase are interleaved on the gridside by shifting their phase-shift references by multiples of 2π/N in order to reduce the ripple of the gridcurrent and to keep the grid filter inductance values Lac low. The modulation details of a module and thepower distribution among the modules are presented in the next sections.

ZCS/ZVS Modulation of a ModuleOn the module level, the grid-side half-bridge and the battery-side full-bridge apply medium-frequencysquare-wave voltages to the transformer windings to shape the leakage inductance current in order tocontrol the power transfer. The switching frequency is chosen to be well above the frequency of themedium voltage grid and the capacitors assumed to be large enough, so that the amplitudes of the gen-erated square-wave voltages can be considered as constant during one switching cycle Ts. During onehalf-cycle of the grid-side AC voltage, two of the grid-side IGBTs in the bidirectional switches are con-stantly on. These are S1b and S2b for the positive and S1a and S2a for the negative half-wave (see Fig. 1).In the following, the modulation in grid-to-battery operation is presented. The derivation of the modula-tion in battery-to-grid operation can be done analogously.

For low power transfer over a switching cycle, modulation mode 1 as shown in Fig. 2a is applied whereasfor high power transfer over a switching cycle modulation mode 2 as shown in Fig. 2b is used with aseamless transition in between. The transformer turns ratio guarantees that nvs,ci > vp,ci in every pointof the AC input voltage vci. The control variables for the power flow are the battery-side pulse-widthτci and the phase-shift φci in relation to the grid-side applied voltage vp,ci. For low power transfer φci isnegative and for high power transfer positive. The mode change occurs at φci = 0. ZCS on the grid-side

sT sT

1τ2τ 3τ 4τ

5τ 6τ0τ

p,civ

Lσ,cii

s,cinv

ciτ

ciφ

(a)

sT sT

1τ

2τ 3τ 4τ5τ 6τ

0τ

p,civ

Lσ,cii

s,cinv

ciτ

ciφ

(b)

Figure 2: Medium-frequency voltages vp,ci,nvs,ci applied to the transformer windings of a module and resulting leakage inductance currentiLσ,ci in modulation mode 1 (a) for low power transfer over a switching cycle Ts and modulation mode 2 (b) for high power transfer over aswitching cycle Ts. The modulation scheme guarantees grid-side ZCS and battery-side ZVS, so that IGBTs can be used on the grid-side andMOSFETs on the battery-side.

is always maintained during the whole AC line period, whereas ZVS for a finite commutation current islost around the zero-crossing of φci as well as around the zero-crossing of the grid-side voltage. In theseregions, the commutation current is not sufficient to charge/discharge the output capacitances of theswitches within the interlocking interval. Nevertheless, considering also the magnetizing current of thetransformer, which has a positive contribution to the commutation current for the battery-side switches,ZVS can be achieved also around φci = 0 [11].

By using piecewise linear equations to describe the leakage inductance current iLσ,ci over a switchingcycle Ts and setting iLσ,ci(τ0) = iLσ,ci(τ3) = 0 (see Fig. 2) to achieve grid-side ZCS, the control variablesfor modulation mode 1 are determined as

τci =π |vci|2nvBci

, (1)

φci =16ωsLσnvBci p∗ci−2πnvBci |vci|2 +π |vci|3

4nvBci |vci|2, (2)

whereas for modulation mode 2 as

τci =4nvBciφci +π |vci|

2nvBci, (3)

φci =2πnvBci |vci|−π |vci|2−

√−32πωsLσnvBci |vci| p∗ci +4π2n2v2

Bci |vci|2−π2 |vci|4

4nvBci |vci|. (4)

There, p∗ci is the reference power transferred over a switching cycle, vci the module input voltage, vBci thebattery voltage and ωs = 2π fs represents the angular switching frequency.

Power Distribution among the ModulesDue to the series connection of the modules in a phase leg, the power distribution among the modulescan only be set by controlling the grid-side voltages of the modules as the input currents are equal. By in-creasing the voltage of a module by ∆vM, the voltage of another module or the voltages of other modulesin the same phase leg have to be reduced, so that the voltage sum of all modules equal the grid voltage.In this way, only the power distribution among the modules is affected, but not the overall power flowbetween the grid and the batteries. The upper limit of the module power is basically given by the block-ing voltage of the applied semiconductor devices.

Each module can be described as a voltage source on the grid side, where the voltage is controlled by thepower transfer between grid-side capacitors Cac and the battery. To increase the module power, first themodule input voltage vci has to be increased which is done by reducing the power transfer to the battery.As soon as the input voltage rises, the power transfer can be increased again to reach its reference value.This is done by the module voltage controller as described below.

Converter ControlThe control of the proposed isolated cascaded half-bridge converter with integrated battery energy stor-age is divided into a top-level and a module-level control as shown in Fig. 3. The objectives of thetop-level control are obtaining the grid and phase angles by means of phase-locked loops (PLLs), con-trolling the grid current in the dq-frame and balancing the average SOC values over the three phases. Onthe module level, the SOCs of the batteries in one phase are balanced and the grid-side module voltagesare controlled.

Grid current controlRef. [14]

Phase SOCbalancing control

Eq. (8)

Battery SOCbalancing control

Eqs. (10)-(12)

Modulevoltage control

Fig. 4

Grid PLLRef. [14]

ai

bi

ci

abcε

Ma∗v

Mb∗v

Mc∗v

av

bv

cv

ai∗v∆ bi

∗v∆ ci∗v∆

ai∗v

bi∗v

ci∗v

Phase PLLsRef. [13]

aε

bε

cε

d∗i q

∗i

abcSOC aSOC

bSOC

cSOC

aiSOC

biSOC

ciSOC

N1

ai, τaiφ

bi, τbiφ

ci, τciφ

Top-level control Module-level control

aiv biv civ

a∗i b

∗i c∗i

0∗v 0

∗v 0∗v

Limiter

dv qv

iδ

Figure 3: Overview of the control system for the battery energy storage system based on the proposed isolated cascaded half-bridge converter.The control consists of a top-level control for grid current and phase SOC balancing control and a module-level control for battery SOCbalancing and module voltage control.

Grid and Phase PLLsFor the grid current control in the dq-frame, the grid PLL in Fig. 3 obtains the grid angle εabc of thethree-phase grid. Moreover, for the module-level voltage control in dq-coordinates, the angles εa,εb,εcof the phase voltages are required which are obtained by the phase PLLs (see Fig. 3). A separate gridPLL is implemented because during unbalanced grid voltage conditions the angle εabc differs from εa.Furthermore, deriving εabc from the phase angles εa,εb,εc would require the measurement of the peakphase voltages at a reasonable sampling rate.For a single phase PLL, only the α-component is available, the β-component has to be constructed. Thisis done by using an orthogonal system generator (OSG) implemented with a second order generalizedintegrator (SOGI) structure as proposed in [12, 13]. The OSG-SOGI structure filters the measured inputvoltage and delivers two clean orthogonal voltage waveforms [13].

Grid Current ControlThe grid current control in dq-coordinates depicted in Fig. 3 takes the grid currents i∗d , i

∗q as reference

values and outputs the reference voltages in time domain v∗Ma,v∗Mb,v

∗Mc of all series connected modules

in a phase leg [14]. These reference values are adjusted by the phase SOC balancing control as describedbelow, then divided by the number of modules N in a phase and passed to the module-level control ofeach module. The current controllers for d- and q-axis have both a proportional and integral part.

Phase SOC Balancing ControlThe objective of the phase SOC balancing control as part of the top-level control is to equalize the aver-age SOC values among the phase legs by means of zero-sequence voltage injection as described in [6].By adding the zero-sequence voltage v∗0 to the phase reference voltages v∗Ma,v

∗Mb,v

∗Mc as shown in Fig. 3,

an unequal active power can be drawn in each phase without drawing negative-sequence current on thegrid side.

The average SOC values in the three phases are given by

SOCa =1N

N

∑i=1

SOCai, SOCb =1N

N

∑i=1

SOCbi, SOCc =1N

N

∑i=1

SOCci, (5)

whereas the reference SOC in a phase leg is the average value of the SOCs of all batteries in the systemcalculated as

SOCabc =13(SOCa +SOCb +SOCc) . (6)

The deviations from the reference values in each phase can then be written as

∆SOCa = SOCabc−SOCa, ∆SOCb = SOCabc−SOCb, ∆SOCc = SOCabc−SOCc, (7)

which are transformed to αβ-coordinates to form a deviation vector ∆SOCv0 · e jδv0 . For compensatingthe phase SOC deviations, the zero-sequence voltage according to

v∗0 = K0 ·∆SOCv0 · cos(εabc +δi−δv0) with δi = arctan(

iqid

)(8)

is added to the phase reference voltages as depicted in Fig. 3, where K0 defines a proportional gain, εabccorresponds to the grid voltage angle obtained by the grid PLL and δi represents the angle of the gridcurrent space vector. The overall three-phase power transfer is not affected by the zero-sequence voltageinjection since the sum of the three phase currents ia, ib, ic is zero due to the open star point.

Battery SOC Balancing ControlThe battery SOC balancing control acts as part of the module-level control to ensure balancing the SOCsamong the batteries in the same phase leg. For the proposed isolated cascaded half-bridge converter, thisis done by adjusting the module voltage references (see Fig. 3) to draw the specific amount of activepower in each module needed to balance the SOCs.

With the measured battery SOCai,SOCbi,SOCci, the deviation of the SOC of a battery i from the averagephase SOC value is calculated. The deviations for phases a,b,c are given by

∆SOCai = SOCa−SOCai, ∆SOCbi = SOCb−SOCbi, ∆SOCci = SOCc−SOCci. (9)

To equalize the SOCs among the batteries in one phase, the control adds the voltages ∆v∗ai,∆v∗bi,∆v∗ciaccording to the control laws

∆v∗ai = K1 ·∆SOCai · cos(εabc +δi) , (10)

∆v∗bi = K1 ·∆SOCbi · cos(

εabc−2π

3+δi

), (11)

∆v∗ci = K1 ·∆SOCci · cos(

εabc−4π

3+δi

), (12)

as depicted in Fig. 3, where δi corresponds to the angle of the grid current space vector. For positive∆SOCai,∆SOCbi,∆SOCci, the power flowing into the batteries has to be increased, which is done byincreasing the grid-side module voltages. Accordingly, negative SOC deviations lead to a decrease ofthe module voltages. The upper limit of the grid-side module voltage is defined by the blocking voltagerating of the applied switching devices considering common safety margins. The limiters are shown inFig. 3 which output the reference voltages v∗ai,v

∗bi,v∗ci for the module-level voltage controllers.

Module Voltage ControlIn order to enable balancing the SOCs of the batteries, the power distribution among the modules has tobe varied. This can be done only by controlling the grid-side voltages of the series connected modulesin a phase leg while maintaining the sum of all module voltages equal to the grid voltage. This approachis similar to the conventional CHB converter where the balancing control of the DC-side capacitors orbatteries is based on varying the amplitudes of the PWM generated sinusoidal voltages of a H-bridgemodule [6,14–16]. Nevertheless, in the proposed isolated half-bridge converter, the module voltage can-not directly be controlled, rather it is controlled indirectly via the power flow over the transformer.

αβ

dq

Orthogonalsystem

generator

αβ

dq

Orthogonalsystem

generator

P

P αβ

dq Module controlvariables calculation

Eqs. (1)-(4)

ci∗v

civ

ci,α∗v

ci,β∗v

ci,βv

ci,αv

cε

cε

ci,d∗v

ci,q∗v

ci,dv

ci,qv

cε

ci,p∗v

c∗i

ci∗p

civ Bciv

ciφ

ciτ

Figure 4: Overview of the module voltage control on the grid side in the dq-frame, exemplarily shown for a module i in phase c.

The voltage control is performed in the dq-frame of the respective phase as exemplarily shown in Fig. 4for phase c. The transformation angles εa,εb,εc are obtained from the corresponding phase PLLs. Thecalculation of the αβ-components of the measured module voltage vci and the reference v∗ci is done byusing the same OSG-SOGI structure [13] as for the single phase PLLs described above.Two proportional controllers with feed-forward of the voltage reference values v∗ci,d ,v

∗ci,q are used to

generate a voltage reference signal v∗ci,p in the time domain which is multiplied by the phase currentreference i∗c to get the desired module transfer power p∗ci. The control variables φci,τci can be calculatedusing (1)-(4) taking also the measured battery voltage vBci into account.

6.6 kV, 150 kW, 1 MWh Battery Energy Storage SystemAs a design example, a 150 kW, 1 MWh battery energy storage system connected to the three-phase6.6 kV, 50 Hz medium voltage grid is considered. Each phase leg consists of 9+1 identical modulesconnected in series with a nominal output power of 5 kW and a power control reserve of ±30 % forSOC balancing of the batteries. All of the modules are connected to separate Lithium-ion storage batterypacks with a voltage range of 280 V to 420 V. Details of the system are listed in Table I. For simulationpurposes, the storage system is scaled down to obtain the simulation results in a reasonable amount oftime.

Table I: Parameters of the battery energy storage system. For simulation purposes, the system is scaled down.

Parameter Design example Simulation

Cascaded half-bridge converter AC grid voltage (RMS line-to-line) Vabc 6.6 kV 2.4 kVAC grid frequency fabc 50 Hz 50 HzNominal system power Pabc 150 kW 45 kWTotal energy storage capacity Eabc 1 MWh 30 WhGrid-side filter inductors Lac 100 µH 100 µHNumber of modules per phase leg N 9+1 3

Half-bridge converter module Input voltage (RMS) Vci 460 V ± 30 % 460 V ± 30 %Nominal output power Pci 5 kW ± 30 % 5 kW ± 30 %Switching frequency fs 15 kHz. . . 55 kHz 20 kHzTransformer turns ratio n 14/9 14/9Transformer leakage inductance Lσ 47 µH 47 µHTransformer magnetizing inductance - neglected neglectedGrid-side capacitors Cac 20 µF 20 µFBattery-side inductors Ldc 100 µH 100 µHBattery-side capacitors Cdc 60 µF 60 µF

Storage battery Battery voltage VBci 280 V. . . 420 V 350 VEnergy storage capacity EBci 33.3 kWh 3.33 Wh

Module Components and Loss ModelIn the following, the components for the single-stage module shown in Fig. 1 are described. To bench-mark the proposed isolated module, it is compared with a two-stage solution applying an H-bridge inputstage and a subsequent DAB converter [4] depicted in Fig. 5. Table II and Table III summarize the com-ponents of the single- and two-stage module designs.

The loss model to calculate the module efficiency includes the main loss shares as conduction and switch-ing losses of the semiconductor devices, skin and proximity effect losses of the transformer windings aswell as the core losses of the transformer. The comparison of the single- and the two-stage solution isdone for a fixed chip area applying the same semiconductor technology and for a fixed volume which isbasically given by cooling system, transformer and filter components.

SemiconductorsOn the grid side where higher blocking voltages are required, 1200 V IGBTs are used. On the batteryside, the applied modulation method guarantees ZVS, so that 650 V MOSFETs are considered. Forthe single-stage solution, paralleled IGBTs of type IHW40N120R3 are utilized on the grid side andparalleled MOSFETs of type IPW65R019C7 on the battery side, both from Infineon [17]. The IGBTIHW40N120R3 offers a relatively low collector-emitter saturation voltage and is therefore ideally suitedin combination with ZCS.

dcC

dcL

civ Bciv

Bcii

σL

mL s,civ

Lσ,cii

: 1n

p,civMciV

1S

2S

3S

4S

5S

6S

7S

8S

9S

10S

1MC

2MC

Figure 5: Two-stage module in the cascaded converter shown in Fig. 1 applying a grid-side H-bridge input stage connected to module capacitors CM1,CM2 and a subsequent bidirectional dualactive bridge DC-DC converter providing galvanic isolation and voltage adaptation to the batteryside. The DC-link voltage VMci is controlled to a fixed voltage of 800 V. The DAB converter isoperated at constant power for a given operating point.

pWWinding sWWinding

pΦ

sΦ

Figure 6: Transformer including reluc-tance model consisting of four U-coreswith windings Wp,Ws wound around theinner leg. The distance between thewindings defines the size of the leakageinductance and provides the requiredisolation.

For the two-stage solution, hard-switching occurs in the H-bridge input stage as well as partially in thesubsequent DAB converter, so that IGBTs of type IKW40N120H3 [17] are used. In the half-bridge ofthe DAB converter the switches are paralleled, resulting in 8 IGBTs in total which corresponds to thesame amount of IGBT devices as in the single-stage solution. On the battery side, the same configurationof semiconductors as in the single-stage solution is applied.

The models for switching and conduction losses are based on datasheet parameters which have been pre-sented in [10]. For ZCS conditions for the IGBT as well as ZVS conditions for the MOSFET negligiblesmall losses are assumed.

TransformerBoth for the single- and the two-stage solution, the transformer is built with four U-cores AMCC-50 [18]with material VITROPERM 500F [19], which exhibits a relatively high saturation flux density of 1.2 Tand is therefore ideally suited for switching frequencies in the range of a few 10 kHz. The grid-side andthe battery-side windings Wp,Ws are wound around the inner leg of the transformer core as shown inFig. 6. Due to the high isolation requirements between grid and battery side of around 10 kVpeak andthe integration of a series inductance, the two windings are accordingly separated with a leakage gap inbetween. Furthermore, a proper isolation distance between windings and core is maintained by the coilformer. The peak electric field is obtained from a FEM simulation as 7.5 kV mm−1. The transformersetup is sealed in epoxy to guarantee the isolation level.

For the single-stage solution, the turns ratio of the transformer n is chosen such that v′Bci > |vci|/2 (withv′Bci referred to the grid side of the transformer) is always satisfied, also at the lowest battery voltageof 280 V. The maximum transformer leakage inductance Lσ is defined by the peak of the output powerPci including the module power control reserve of 30% that has to be available at any given input/outputvoltage combination. After optimizing number of turns, turns ratio and litz wires, the results summarizedin Table II are obtained.

For the DAB converter, the turns ratio is given by the fixed DC-link voltage VMci of 800 V and the nominalbattery pack voltage of 350 V. In this way, the soft-switching range for phase-shift modulation extendsover a wide operating range in terms of output voltage and power [10]. The maximum transformer leak-age inductance Lσ is defined by the nominal module power including the control reserve of 30 %. The

Table II: Components of the single-stage module.

IGBTs S1a,S1b,S2a,S2b 2x IHW40N120R3, 1200 VMOSFETs S3,S4,S5,S6 2x IPW65R019C7, 650 VTransformer core 2x AMCC-50 VITROPERM 500FTransformer winding Wp 14 turns, Litz wire

2883 strands, 0.08 mmTransformer winding Ws 9 turns, Litz wire

1722 strands, 0.12 mmTransformer leakage Lσ 47 µH

Table III: Components of the two-stage module.

IGBTs S1,S2,S3,S4 1x IKW40N120H3, 1200 VIGBTs S5,S6 2x IKW40N120H3, 1200 VMOSFETs S7,S8,S9,S10 2x IPW65R019C7, 650 VTransformer core 2x AMCC-50 VITROPERM 500FTransformer winding Wp 24 turns, Litz wire

231 strands, 0.1 mmTransformer winding Ws 21 turns, Litz wire

1558 strands, 0.12 mmTransformer leakage Lσ 82 µH

3.5 4 4.5 5 5.5 6 6.5280

300

320

340

360

380

400

420

0.970.971

0.972

0.972

0.973

0.9730.974

0.974

0.97

5

0.975

0.976

0.976

0.977

Input Power (kW)

Out

put V

olta

ge (V

)

Figure 7: Relative converter efficiencies for the single-stage moduleshown in Fig. 1 for an input power range of 5 kW ± 30 % and abattery output voltage range of 280 V to 420 V applying IGBTs onthe grid side and MOSFETs on the battery side. A peak efficiency of97.8 % is reached, the average efficiency is 97.4 %.

3.5 4 4.5 5 5.5 6 6.5280

300

320

340

360

380

400

420

0.960.9620.964

0.9660.968

0.9680.970.97

0.9720.972

0.9740.974

0.9760.976

0.978

0.978

0.978

0.98

0.98

0.98

0.982

0.9820.982

Input Power (kW)

Out

put V

olta

ge (V

)

Figure 8: Relative converter efficiencies for the two-stage moduleshown in Fig. 5 for an input power range of 5 kW ± 30 % and abattery output voltage range of 280 V to 420 V applying IGBTs onthe grid side and MOSFETs on the battery side. A peak efficiency of98.3 % is reached, the average efficiency is 97.4 %.

optimization of number of turns and litz wires lead to the results listed in Table III.

In the loss model, the core losses per volume are calculated by applying the improved GeneralizedSteinmetz Equation (iGSE) [20]. The skin and proximity effect losses per unit length in litz wires for eachcurrent harmonic are determined according to [21]. The external magnetic field strength for evaluatingproximity effect losses is derived by a 1D approximation based on the Dowell method [22].

Efficiency CalculationThe module efficiencies of the single-stage solution shown in Fig. 1 and the two-stage solution depictedin Fig. 5 are calculated for an output power range of 5 kW ± 30 % and an output voltage of 280 V to420 V. Both efficiency calculations are based on a fixed chip area applying the same semiconductor tech-nology (same amount of IGBT and MOSFET devices) and for a fixed volume which is basically givenby cooling system, transformer and filter components.

For the single-stage solution, the switching frequency of a module is varied depending on its operatingpoint in terms of output voltage and power. In this way, the leakage impedance Zσ = jωsLσ is modu-lated to adapt the transformer design to the various AC operating points to achieve higher efficiencies.In contrast to this, the two-stage solution applies a DAB converter with fixed switching frequency andconstant power transfer where the transformer can be designed and optimized for given operating points.Furthermore, the two-stage solution features better utilization of the transformer core.

The H-bridge input stage of the two-stage module is operated at 1.5 kHz to get an equivalent switchingfrequency of 30 kHz by applying the phase-shifted unipolar sinusoidal PWM [23], so that the filter effortfor both module solutions is similar. For the DAB converter, phase-shift modulation is applied whichleads to high efficiency for high power transfer [10].

The relative module efficiencies are shown in Fig. 7 and Fig. 8. Considering the single-stage module, itcan be seen, that for low output voltages the efficiency is increased whereas for high output voltages theefficiency is decreased compared to the two-stage module. Nevertheless, the average efficiency over thewhole operating range for both solutions is 97.4 %.

Simulation ResultsThe proposed converter is simulated in GeckoCIRCUITS [24] with a simulation model according toFig. 1 of a downscaled battery energy storage system with parameters given in Table I. The simulationmodel applies 3 modules per phase leg. The input current reference is set to I∗abc = 10Arms for chargingoperation. Initially, the SOCs of the batteries are in an unbalanced state.

Fig. 9 shows the simulated grid currents ia, ib, ic together with the grid voltages va,vb,vc. The currentsare controlled to the reference amplitudes and in phase to the grid voltages (PFC operation). In Fig. 10,the module input voltages vai,vbi,vci and the battery currents iBai, iBbi, iBci are depicted. It can be seen,that the module voltage controller adjusts the module input voltages according to the reference given bythe top-level control and the battery SOC balancing control. The overall power flow into the batteries isaccordingly divided as can be seen from the battery currents.

−2000

−1000

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1000

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rid V

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ges (

V)

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5

10

15

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rent

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)

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av bv cv

ai bi ci

Figure 9: Simulated grid currents of the downscaled cascaded half-bridge converter with parameters given in Table I for a referencevalue of I∗abc = 10Arms in charging operation.

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tery

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3av2av

1av

3bv2bv

1bv

3cv2cv

1cv

3Bai2Bai

1Bai3Bbi

2Bbi1Bbi

3Bci2Bci

1Bci

Figure 10: Simulated module voltages and battery currents of thedownscaled cascaded half-bridge converter with parameters given inTable I for a reference value of I∗abc = 10Arms in charging operation.

0.2

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1aSOC

3bSOC2bSOC

1bSOC

3cSOC2cSOC

1cSOC

3Bai2Bai1Bai

3Bbi2Bbi1Bbi3Bci2Bci1Bci

Figure 11: Simulated battery SOCs and battery currents of the down-scaled cascaded half-bridge converter with parameters given in Ta-ble I for a reference value of I∗abc = 10Arms in charging operation.

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3a∆SOC

2a∆SOC

1a∆SOC

3b∆SOC

2b∆SOC

1b∆SOC

3c∆SOC

2c∆SOC1c∆SOC

Figure 12: Simulated phase SOC deviations and battery SOC de-viations of the downscaled cascaded half-bridge converter with pa-rameters given in Table I for a reference value of I∗abc = 10Arms incharging operation.

From Fig. 11 the balancing of the battery SOCs by the phase SOC and battery SOC balancing controlcan be seen. The initial SOC imbalance gets smaller during the charging operation of the batteries. Thechanges of the battery currents over time show the different power requirements of the batteries to finallyend up in a balanced state where the battery currents converge to the same values. In Fig. 12, the phaseSOC deviations (7) as well as the battery SOC deviations (9) are depicted. Due to the charging withpulsating currents, the SOCs exhibit small ripples with double the grid frequency.

ConclusionA novel isolated cascaded half-bridge converter for battery energy storage systems with three phase legsof series connected modules in star connection is presented. The modules consist of grid-side half-bridges, battery-side full-bridges and medium-frequency transformers providing isolation. By varyingthe number of modules in a phase leg, the converter system can be easily scaled according to voltage,power and energy storage requirements. The new modulation scheme applied on a single module ensuresgrid-side ZCS and battery-side ZVS. The modules in the same phase leg are interleaved on the grid sidein order to reduce the grid current ripple and to keep the grid filter effort low. The control system consistsof a grid current control as well as a SOC balancing control for the batteries including a new module-level voltage controller and is implemented on a design example system in simulation. Evaluation of theproposed module shows an average efficiency of 97.4 %.

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