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DCIS 2004
Extended Abstracts of the
XIX Conference on Design of Circuits and Integrated Systems
Edited by
Pascal Fouillat María Luisa López Vallejo
Jean Tomas
Bordeaux, France November 24-26, 2004
DCIS 2004 XIX Conference on
Design of Circuits and Integrated Systems
Bordeaux, France November 24-26, 2004
Organized by
Laboratoire IXL - ENSEIRB - CNRS UMR5818 UNIVERSITE BORDEAUX 1
Sponsored by
La Région Aquitaine La Communauté Urbaine de bordeaux
La Mairie de Bordeaux La Maison du Tourisme de la Gironde
And
IEEE CAS Le Club EEA
ANSOFT CIS Mentor Graphics Coherent
ST Nasa
Foreword
On behalf of the DCIS Organizing and Program Committees, we would like to welcome you to the 19th Conference on Design of Circuits and Integrated Systems (DCIS), held in Bordeaux, France, November 24th-26th.
The Conference aims at gathering the experts in the field ofMicroelectronics, and providing a forum to exchange ideas and information on industrial and research results. The 2004 edition of DCIS confirms its international impact with the contribution of 16 countries.Experts in a wide range of areas have decided to participate, leading to a very high number of submitted papers and a reviewing effort shared upon 178 reviewers. The quality of the submissions made the selection very difficult. Following the reviewers recommendations, the program committee accepted 182 papers for oral presentation at the Conference.
This year, the three-day technical program is organized in four parallel tracks and 36 sessions. Two plenary sessions take place with distinguished speakers from STMicroelectronics and the NASA to presentthe new challenges offered to our community by Software Designed Radioand Space Electronics. A panel session is also scheduled in order to evaluate and discuss altogether the return on experiments from European countries with our new higher educational system according to the Bologna process. And for the first time this year, a special award will be conferred to honour the best paper from the previous edition of DCIS, in 2003.
DCIS 2004 results from the work of many dedicated volunteers: the authors of the papers, the reviewers, the session organisers, the moderators, the invited speakers, and the sponsors. We would like to thank the ENSEIRB Graduate Engineering School for providing the modern environment abreast of the scientific level of this event. We alsowould like to use this opportunity to express our gratitude to the members of the IXL Organizing Committee, for all the time and efforts they have offered freely for the great pleasure of our scientific community. We wishyou a productive and enjoyable stay in the sweet area of Bordeaux.
Pascal FouillatGeneral Chairman
Maria Luísa López Vallejo & Jean Tomas Program Co-Chairs
III
Table of Contents1 Plenary Sessions
2 Software Defined Radio : Theory and Applications, Ernesto Perea (STMicroelectronics)
3 Space Electronics : a Challenging World for Designers, Christian Poivey & Kenneth LaBel (Goddard Space Flight Center, NASA)
4 Panel Discussion
5 The Bologna Process : Return on Experiment, Moderator: Prof. Yves Danto (U Bordeaux 1)
6 Exhibits
7 Microwind : An introduction to nano-scale CMOS cell design. Prof. Etienne SICARD, Sonia BENDHIA (INSA Toulouse)
8 IC-Emit: Comparing simulated/measured Parasitic Emission of Integrated Circuits. Prof. Etienne SICARD (INSA Toulouse), Amaury SOUBEYRAN (Eads-CCR)
9 Session 1a : CAD Tools and Optimisation Algorithms
10 "BNSAT: Representing Boolean Functions in a Non-Canonical Form", Joaquín Saiz(Universidad Autónoma de Barcelona), Jordi Cortadella (Universidad Politécnica de Catalunya), Lluís Ribas, Jordi Carrabina (Universidad Autónoma de Barcelona)
11 "SUSANA: a MOS-Mixed-Circuit Simulator Using Logic/ELogic Algorithms Implemented in Python", Tiago Carrisosa , Tiago Félix , Miguel Jerónimo (INESC-ID/IST), José SoaresAugusto (INESC-ID/FCUL-Dep. Física)
12 "A Distributed Enhanced Genetic Algorithm Kernel Applied to a Circuit/Level Optimization E-Design Environment", Manuel Barros (Instituto Politecnico de Tomar), Goncalo Neves(Instituto Superior Tecnico - IST/IT), Jorge Guilherme (Instituto Politecnico de Tomar), NunoHorta (Instituto Superior Tecnico-IST/IT)
13 "A CAD Tool for the Design of RTD Programmable Gates based on MOBILE ", HectorPettenghi , Maria Jose Avedillo , Jose Maria Quintana (Instituto de Microelectrónica de Sevilla, Centro Nacional de Microelectrónica)
14 Session 1b : Data Converter Design
15 "A New Capacitor-Ratio and Offset Independent Amplifier for Pipelined A/D Converters",Fernando Muñoz Chavero , Antonio Torralba Silgado , Ramón González Carvajal , Bernardo Palomo Vázquez (Departamento de Ingeniería Electrónica, Escuela Superior de Ingenieros, Universidad de Sevilla, Spain)
IV
16 "Simulation-based High-level Synthesis of Pipeline Analog-to-Digital Converters", JesúsRuiz-Amaya , José M. de la Rosa, Manuel Delgado-Restituto (Instituto de Microelectrónicade Sevilla, IMSE-CNM (CSIC))
17 "Digital Background Technique for Gain Error Correction in Pipeline ADCs", Antonio JoséGinés Arteaga , Eduardo José Peralías Macías , Adoración Rueda Rueda (Instituto de Microelectónica de Sevilla, Centro Nacional de Microelectrónica)
18 "Mismatch Properties of MOS and Resistors Calibrated Ladder Structures", Rafael Serrano-Gotarredona , Teresa Serrano-Gotarredona, Bernabé Linares-Barranco (IMSE-CNM-CSIC)
19 Session 1c : SiGe Designs
20 "A multi-standard SiGe Power Amplifier for GSM900/DCS/PCS/WCDMA applications.",Laurent Leyssenne , Jean-Marie Pham , Pierre Jarry , Eric Kerherve (IXL Microelectronicslaboratory ), Daniel Saias (ST Microelectronics - Crolles)
21 "A SiGe BiCMOS, Low Noise and Wide Band Amplifier Working at 77 K", Damien Prele , Geoffroy Klisnick , Gérard Sou , Michel Redon (UPMC-LISIF), Alain Kreisler (SUPELEC-LGEP), Cyrille Boulanger (CNES)
22 "A SiGe Power Amplifier with Dynamic Bias for Efficient Power Control in UMTS/W-CDMAApplications", Nathalie Deltimple, Eric Kerherve (IXL Laboratory), Didier Belot (ST Microelectronics), Yann Deval, Pierre Jarry (IXL Laboratory)
23 "A 5GHz SiGe VCO for WLAN using Optimized Spiral Inductors", Amaya Goñi (Applied Microelectronics Research Institute, University of Las Palmas de Gran Canaria, Las Palmas,Spain), Sunil Lalchand Khemchandani , Javier Del Pino, Antonio Hernández (AppliedMicroelectronics Research Institute and Departamento de Ingeniería Electrónica y Automática, University of Las Palmas de Gran Canaria, Las Palmas, Spain)
24 Session 1d : Built In Self Test
25 "Robustness Improvement of a Ratiometric Built-In Current Sensor", Mikaël Cimino , Magali De Matos , Hervé Lapuyade, Jean-Baptiste Bégueret, Yann Deval (IXL lab)
26 "A Non-Intrusive Built-In Sensor for Transient Current Testing of Digital VLSI Circuits ", Bartomeu Alorda , Vicenç Canals , Jaume Segura (Universitat Illes Balears)
27 "Built-In Current Sensor using Floating-Gate MOS Transistors for Low-Voltage Applications", Alkiviades A. Hatzopoulos , Stilianos Siskos (Aristotle Univ. of Thessaloniki)
28 "Experimental Evaluation of a Built-in Current Sensor for Analog Circuits", Roman Mozuelos, Yolanda Lechuga , Miguel Angel Allende, Mar Martinez , Salvador Bracho (University of Cantabria)
V
29 Session 2a : High Level Modeling
30 "Compact Modeling of a Magnetic Tunnel Junction using VHDL-AMS" , Jean-BaptisteKammerer, Luc Hébrard (Laboratoire d'Electronique et de Physique des SystèmesInstrumentaux (LEPSI)), Michel Hehn (Laboratoire de Physique des Matériaux (LPM)), Francis Braun (Laboratoire d'Electronique et de Physique des Systèmes Instrumentaux(LEPSI)), Patrick Alnot (Laboratoire de Physique des Milieux Ionisés et application (LPMI)), Alain Schuhl (Laboratoire de Physique des Matériaux (LPM))
31 "Analogue-Synthesis Tool Development for Switched-Current Systems using VHDL-AMS", Nesrine Ksentini, Ahmed Fakhfakh, Mourad Loulou, Nouri Masmoudi (LETI Laboratory,ENIS, SFAX, TUNISIA), Yannick Hervé (CNRS-PHASE, Strasbourg, France), Jean-JacquesCharlot (ENST)
32 "Modeling and simulation of phototransistors using VHDL-AMS", Annick Alexandre, Andrea Pinna, Bertrand Granado, Patrick Garda (LISIF - UPMC - PARIS)
33 "Final User Oriented SOC Modeling", Sébastien Snaidero, Yannick Hervé (CNRS/PHASE)
34 "Design and Simulation of Mixed-Mode Optical Systems for PSD Applications", Ricardo Doldán , Eduardo Peralías, Alberto Yúfera, Adoración Rueda (Instituto de Microelectrónicade Sevilla (IMSE) - Centro Nacional de Microelectrónica (CNM))
35 Session 2b : Biometric and Robotic Applications
36 "DSP-based Fuzzy Controllers: Application to Parking an Autonomous Robot", IluminadaBaturone, Francisco J. Moreno-Velo, Santiago Sánchez-Solano (Instituto de Microelectrónica de Sevilla (IMSE-CNM)), Víctor Blanco , Joaquín Ferruz (Dept. Ingeniería de Sistemas y Automática. Univ. de Sevilla)
37 "Coprocessor of the Ridge Line Following Fingerprint Algorithm", E Canto, N Canyellas(URV), M Lopez (UPC), M Fons, F Fons (URV)
38 "Iris Biometrics Verifiers for Low Cost Identification Tokens", Judith Liu-Jimenez, Raul Sanchez-Reillo (Universidad Carlos III de Madrid), Carmen Sanchez-Avila (E.T.S.I. Telecomunicacion (U. P. M.)), Luis Entrena (Universidad Carlos III de Madrid)
39 "Fingerprint Matching Acceleration in Smart Cards", Luis Entrena, Raúl Sánchez-Reillo,Almudena Lindoso, Judith Liu (Universidad Carlos III de Madrid)
40 "Hardware Implementation of the Bresenham Line Generation Algorithm applied to µ-robotMovement", Raimon Casanova, Angel Dieguez, Juanjo Lacort, Josep Samitier (Departamentd’Electrònica, SIC, Universitat de Barcelona)
41 Session 2c : Industrial Applications
42 "Dual-Port Serial Arbiter with GSM Modules for Simultaneous Local/Remote Control of RS232-based Devices", Eloi Ramon, Lluís Ribas (Universitat Autònoma de Barcelona)
VI
43 "Signal Processing Unit for River Tugboat Telemetry System", Humberto Campanella(Instituto de Microelectrónica de Barcelona IMB-CNM / U. del Norte), Mauricio Pardo, Víctor Manotas, Javier Páez (Universidad del Norte (Barranquilla-Colombia)), Juan Carlos Niebles,David Angulo (Flota Fluvial Carbonera Ltda)
44 "A Sensorless Electronically Controlled Horn for Automobiles", M. Cesar Rodriguez, CesarSanz (Universidad Politecnica de Madrid), Jacinto M. Acero, Fernando Nozal (Robert BoschEspaña S.A.)
45 "Design of Low-Power CMOS Read-Out ICs for Large Arrays Cryogenic Infra-Red Sensors",Bertrand Misischi, Francisco Serra-Graells (Centro Nacional de Microelectrónica - CSIC), Eduardo Casanueva, César Méndez (Indra Sistemas S.A.), Lluís Terés (Centro Nacional deMicroelectrónica - CSIC)
46 "A Dynamic Current Mode Logic to Counteract Power Analysis Attacks", François Macé, François-Xavier Standaert, Illham Hassoune, Jean-Didier Legat, Jean-Jacques Quisquater(Laboratoire de Microélectronique, UCL, Belgium.)
47 Session 2d : Data Converter Test
48 "Digital Diagnosis of Settling Error in Sigma-Delta Modulators ", Gildas Leger (Instituto de Microelectronica de Sevilla (IMSE-CNM)), Adoración Rueda (Universidad de Sevilla, Instituto de Microelectronica de Sevilla (IMSE-CNM))
49 "Digital Sigma Delta Oscillator : Design Consideration ", Maher Jridi (Laboratoire IXL), Dominique Dallet (Laboratoire IXL - ENSEIRB), Chiheb Rebai (Institut Supérieur des arts dumultimédia de Manouba), Philippe Marchegay (Laboratoire IXL - ENSEIRB)
50 "Optimal Implementation of Linear and Adaptive Filter Bank for ADC Characterization",Fahmi Missaoui (MEDIATRON Laboratory - High School of Communications TUNIS SUP'COM - Tunisie), Dominique Dallet (IXL laboratory - ENSEIRB- Bordeaux 1 University), Chiheb Rebai, Adel Ghazel (MEDIATRON Laboratory - High School of CommunicationsTUNIS SUP'COM - Tunisie)
51 "Selection of Test Techniques for High-Resolution Sigma-Delta Modulators", Oscar Guerra , Sara Escalera, Jose Manuel de la Rosa (IMSE-CNM), Eric Compaigne, Christophe Galliard(Dolphin Integration), Angel Rodríguez-Vázquez (IMSE-CNM)
52 "Guidelines for the Design of a Sine-Wave Analyzer for BIST Applications", Manuel J. Barragan, Diego Vazquez, Adoracion Rueda (Instituto de Microelectronica de Sevilla- CentroNacional de Microelectronica (IMSE-CNM))
53 Session 3a : Digital Signal Processing in FPGA Platforms
54 "Implementing the FFT Algorithm in FPGA Plaforms: a Comparative Study of Parallel Architectures", Miguel Angel Sanchez Marcos, Mario Garrido Galvez, Marisa López-Vallejo,Jesus Grajal de la Fuente (Univ. Politecnica Madrid)
VII
55 "An FPGA Landmine Detection System based on Infrared Images", Fernado Pardo (Universidad de Santiago de Compostela. Santiago. Spain), Marco Balsi (Università LaSapienza. Roma. Italy), Paula López (Fraunhofer Institut für Integrierte Schaltungen.Erlangen. Germany), Diego Cabello (Universidad de Santiago de Compostela. Santiago. Spain)
56 "Implementation of Optimized FFT on Stratix DSP Development Board ", Nouvel Fabienne(IETR)
57 "Comparison of Two Implementations of Scalable Montgomery Coprocessor Embedded in Reconfigurable Hardware", Milos Drutarovsky (Technical Univeristy of Kosice, Slovak Republic), Viktor Fischer (Universite Jean Monnet, Saint-Etienne, France), Martin Simka (Technical Univeristy of Kosice, Slovak Republic)
58 "An implementation of a Parallel Architecture for the Self-Sorting FFT Algorithm applied to IEEE 802.11a", Ainhoa Cortes, Igone Velez, Juan Francisco Sevillano (CEIT), Andoni Irizar(Universidad de Navarra), Pilar Calvo (CEIT)
59 "Optimized FPGA implementation of Trigonometric Functions with Large Input Argument",Javier Hormigo, Manuel Sanchez, Mario A. Gonzalez, Gerardo Bandera, Julio Villalba (Dept.Computer Architecture. University of Malaga)
60 Session 3b : Sensors and Smart Objects
61 "CMOS Buried Double Junction Active Pixel Sensor For High-Sensitivity Low-ResolutionLinear Arrays ", Patrick Pittet, Genaro Carillo, Guo-Neng Lu, Loubna Hannati (LENACUniversité Claude Bernard)
62 "An Experience on Wireless Networks for Industrial Applications", Emili Lupon, GabrielTorrens (Departament d'Enginyeria Electrònica, Universitat Politècnica de Catalunya)
63 "Sigma Delta Based Parametrable Sensor Interface, a Design Methodology", Fellah Yasmina, Tixier Thierry, Aubert Alain, Abouchi Nacer (CPE LYON LPM)
64 "An Improved Love Wave Oscillator for Low Concentration Chemical Sensing Application",Nicolas Moll, Corinne Déjous, Dominique Rebière, Jacques Pistré (Laboratoire IXL), RogerPlanade (Centre d’Etudes du Bouchet)
65 "Parasitic Hot Electron Effects in Active Pixel Sensors", Stephan Maëstre, Pierre Magnan (Supaero)
66 "Ring-Oscillator Based Temperature Sensor for Deep Submicron CMOS Technologies",Sebastia Bota, Vicens Canals, Josep lluis Rosselló, Jaume Segura (Universitat de les Illes Balears)
67 Session 3c : Bio-inspired Circuits
68 "Experiments on Electrical MFHN Neurons", Stéphane Binczak, Sabir Jacquir, Olivier Tarlet, Jean-Marie Bilbault (LE2I, CNRS UMR 5158, Université de Bourgogne)
VIII
69 "A Mixed Neuromorphic ASIC for Computational Neurosciences", Sylvain Saighi, Jean Tomas, Yannick Bornat, Sylvie Renaud (Laboratoire IXL)
70 "Mixed-Mode Class AB Neuron Building Blocks: Analysis and Real Application", Guillermo Zatorre-Navarro, Nicolas Medrano-Marques, Santiago Celma-Pueyo (Universidad de Zaragoza)
71 "A Discrete-Time Cellular Neural Network Architecture for a Pixel-Level Snake On-ChipImplementation", V.M. Brea, D.L. Vilarino, D. Cabello (Dept. of Electronics and ComputerScience, University of Santiago de Compostela)
72 "Charge-Packet Driven Mismatch-Calibrated Integrate-and-Fire Neuron for Address-Event-Representation", Rafael Serrano Gotarredona, Bernabe Linares-Barranco, Teresa SerranoGotarredona (Instituto de Microelectrónica de Sevilla)
73 "Digital Implementation of a Simplicial Cellular Neural Network", Pablo Echevarria, Victoria Martinez, Jose M. Tarela, Ines del Campo (Universidad del Pais Vasco)
74 Session 3d : Power Electronics : Devices and Systems
75 "Optimization of a High Voltage p-Channel Transistor fabricated using a Standard CMOS Process", Amador Pérez-Tomás, Xavier Jordà, Philippe Godignon , Miquel Vellvehí, José Millán (Centre Nacional de Microelectrònica (IMB-CNM-CSIC))
76 "Digital Phase-Shifting for Multiphase Converters", Angel de Castro (Universidad Politecnicade Madrid), Pablo Zumel (Universidad Carlos III de Madrid), Oscar Garcia, Teresa Riesgo(Universidad Politecnica de Madrid)
77 "Electro-thermal Characterization of Ultracapacitors used as Power Source in Hybrid ElectricVehicles", Walid Lajnef, Jean-Michel Vinassa, Stéphane Azzopardi, Olivier Briat, Eric Woirgard, Christian Zardini (IXL)
78 "Linear Regulators for Lithium Batteries", Peter Spies, Günter Rohmer (Fraunhofer Institute IIS)
79 "Specific Drivers and Integrated 20V Regulated Charge-Pump for an AutonomousMicroRobot: MiCRoN", Albert Saiz, Pere Miribel-Catala, Jordi Brufau, Raimon Casanova,Manel Puig-Vidal, Josep Samitier (Sic Lab. Electronics Department. Universitat deBarcelona)
80 "AC Current and DC Voltage Sensorless Control of Bidirectional Boost-Buck Converter",Touzani Youssef, Toumalet Jean Pierre, Le Bars Pierre, Laurent Alain, Gary Francis (IUT Montluçon)
81 Session 4a : Image Processing
82 "A Study of Trade offs in Inter-frame Compression MPEG for a Multiprocessor Platform", Antoni Portero (Dept. Informatica Universitat Autonoma de Barcelona), Pol Marchal (IMEC ), Jose Ignacio Gomez, Luis Piñuel (DACYA U.C.M,), Francky Catthoor (IMEC), JordiCarrabina (Dept. Informatica Universitat Autonoma de Barcelona)
IX
83 "Adviser Coprocessor for Image Compression on FPGA", Antonio Guzman, Marta Beltran (Rey Juan Carlos University)
84 "Power-Aware Tuning of Dynamic Memory Management for Embedded Real-TimeMultimedia Applications", David Atienza (DACYA/Complutense University of Madrid & IMEC vzw), Stylianos Mamagkakis (VLSI Center-Demokritus University), Miguel Peon, JoseManuel Mendias (DACYA/Complutense University of Madrid), Francky Catthoor (IMEC vzw), Dimitrios Soudris (VLSI Center-Demokritus University)
85 "An IIR Based 2D Adaptive and Predictive Cache for Image Processing", Stéphane Mancini ,Nicolas Eveno(LIS - Laboratoire des Images et des Signaux)
86 "Real Time Smart Pixels Processing Array for Mobile Multimedia Applications", SebastianLópez, Rafael Calzada, Ayoze Tejera, Jose Fco. López, Roberto Sarmiento (ResearchInstitute for Applied Microelectronics (IUMA))
87 "Adaptation of Altera Stratix DSP Board for Real-time Stereoscopic Image Processing",Pavol Pavelka, Vincent Betheas, Viktor Fischer, Virginie Fresse (Laboratoire Traitement du signal et Instrumentation-Universite Jean Monnet)
88 Session 4b : Embedded Design & System On Chip
89 "On the Performance of Three-State and Multiplexor Logic Interconnection for Shared Bus SoC Design", Unai Bidarte, Armando Astarloa, José Luis Martín, Jaime Jiménez, CarlosCuadrado (Universidad del Pais Vasco UPV - Euskal Herriko Unibertsitatea EHU)
90 "Reconfiguration Control for Dynamically Reconfigurable Systems", Ewerson Carvalho, Ney Calazans, Fernando Moraes (Pontifícia Universidade Católica do Rio Grande do Sul - PUCRS), Daniel Mesquita (Universite de Montpellier II (LIRMM))
91 "A SoC-based Architecture coupled with a CMOS Image Sensor for measurements by Image Processing", Lelong Lionel, Motyl Guy, Jacquet Gérard, Bochard Nathalie(Laboratoire Traitement du Signal et Instrumentation )
92 "Simulation of a NoC-based Heterogeneous System Using Ns", Maria del Milagro BoladoTirado, Pablo Sanchez Espeso (Microelectronics Engineering Group - University of Cantabria)
93 "A Context-Switch Based Checkpoint and Rollback Scheme", Michele Portolan, Régis Leveugle (Tima-CMP Laboratory)
94 "Cyclope: An Integrated Real-time 3D Image Sensor", Tarik graba, Bertrand Granado, Olivier Romain (LISIF - UPMC), Thomas Ea (ISEP), Andrea Pinna, Patrick Garda (LISIF - UPMC)
95 Session 4c : Analog CMOS Design
96 "High Sensitivity and Wide Bandwidth CMOS Transimpedance Amplifier for Optical Receiver Circuit", Mohamed Boutaleb Guermaz, Lyes Bouzerara (Centre de Développement des Technologies Avancées), Hamoudi Escid (Université des Science et de Technologies HouariBoumediene), Mohand Tahar Belaroussi (Centre de Développement des TechnologiesAvancées)
X
97 "1.5V Square-Root Domain Magnitude Locked Loop", Carlos A. De La Cruz-Blas, Antonio Lopez-Martin, Alfonso Carlosena (Public University of Navarra)
98 "High-Speed High-Precision Analog Rank Order Filter with O(n) complexity in CMOS Technology", Ramon Carvajal (Dpto. de Ingenieria Electronica, Escuela Superior de Ingenieros, Universidad de Sevilla (Spain)), Jaime Ramirez-Angulo, Gladys OmayraDucoudray (Klipsch School of Electrical and Computer Engineering, New Mexico State University), Antonio Lopez-Martin (Dept. of Electrical and Electronic Engineering, PublicUniversity of Navarra, Pamplona (Spain))
99 "A Seventh Order Elliptic CMOS Continuous Time Gm-C Filter for PLC applications", Juan Francisco Fernández-Bootello, Manuel Delgado-Restituto, Angel Rodríguez-Vázquez(Instituto de Microelectrónica de Sevilla, Centro Nacional de Microelectrónica)
100 "Tunable Gm-C Biquadratic Filter Operating in Moderate Inversion", Jaime Ramirez-Angulo(New Mexico State University), Chandrika Durbha (Biomorphic VLSI, Inc.), Antonio J. Lopez-Martin (Public University of Navarra), Ramon G. Carvajal (University of Sevilla)
101 "Fully-Differential CMOS Current Conveyor Operating in Moderate Inversion", Antonio J. Lopez-Martin (Public University of Navarra), Jaime Ramirez-Angulo, Chandrika Durbha (NewMexico State University), Ramon G. Carvajal (University of Sevilla)
102 Session 4d : Radiation Effects and EMC
103 "Analysis of Transient Fault Emulation Techniques in Platform FPGAs", Marta PortelaGarcia, Celia Lopez-Ongil, Mario Garcia-Valderas, Luis Entrena (Universidad Carlos III of Madrid)
104 "Analysis of input and feedback capacitances effect on low noise preamplifier performancefor X-rays silicon strip detectors", Thomas Noulis, Stilianos Siskos (Aristotle University ofThessaloniki (AUTH)), Gerard Sarrabayrouse (Laboratoire d'Analyse et d'Architecture desSystemes (LAAS) - National Center for Scientific Research (CNRS))
105 "A Hardware Approach for SEU Immunity Verification using Xilinx FPGA's", Miguel A. Aguirre, Jonathan N. Tombs, Fernando Muñoz, Vicente R. Baena, Antonio J. Torralba,Leopoldo G. Franquelo (Departamento de Ingeniería Electrónica. Universidad de Sevilla), Agustin Fernàndez-Léon, Francisco Tortosa-Lopez, Daniel Gutiérrez-Gonzàlez (Data Systems Division, ESTEC/TOS-ED European Space Agency)
106 "Radiation Hardness Assessment of an ADC for Space Application using a Laser Test Equipment", Vincent Pouget, Pascal Fouillat, Dean Lewis, Frédéric Darracq (IXL - CNRS UMR 5818)
107 "Exploitation of the ICEM Model for Jitter Analysis in an Integrated PLL", Jean-Luc Levant(ATMEL Nantes), Mohamed Ramdani, Richard Perdriau (ESEO), M'hamed Drissi (INSA Rennes)
108 "An IP-Based Chip-Level EMC Modeling and Prediction Methodology", Richard Perdriau,Mohamed Ramdani (ESEO), Jean-Luc Levant (ATMEL Nantes)
XI
109 Session 5a : Devices for High Frequency Circuits
110 "Integrated MOS Varactors in Accumulation Mode for RF Applications", Benito Gonzalez,Javier Garcia (IUMA. Universidad de Las Palmas de Gran Canaria), Iñigo Gutierrez, NekaneSainz (Escuela de Ingenieros. Universidad de Navarra. TECNUN), Margarita Marrero,Amaya Goñi (IUMA. Universidad de Las Palmas de Gran Canaria)
111 "Embedded Passive Design for High Speed Circuits ", Genevieve Duchamp, Yves Ousten,Bruno Levrier (Lab. IXL - Univ. Bordeaux 1), Philippe Kertesz (Thales Airborne Systems), Steven Heytens (Rogers NV)
112 "Analysis and Applications of MOS Resistive Cells", M. Teresa Sanz, Santiago Celma, BelénCalvo, Juan Pablo Alegre (Universidad de Zaragoza)
113 "Ladder-type FBAR Filter Synthesis Methodology", Alexandre Shirakawa, Jean-Marie Pham,Pierre Jarry, Eric Kerherve, Elias Hanna (IXL Microelectronics Laboratory)
114 "Study of the Proximity Effect in High Q Inductors with CMOS 0.18 µm Technology", Iosu Cendoya, Nekane Sainz (Tecnun), Jaizki Mendizabal, Roc Berenguer, Unai Alvarado,Andres Garcia-Alonso (CEIT)
115 Session 5b : Low Power / Low Voltage : analog circuits (1)
116 "Wireless Battery Charger Chip for Smart-Card Applications", Franz Xaver Arbinger, Peter Spies, Guenter Rohmer (Fraunhofer Institut Integrierte Schaltungen)
117 "Low-power high-slew-rate rail-to-rail CMOS analog buffer", Ramón G. Carvajal (Universityof Sevilla), Juan M. Carrillo, J. Francisco Duque-Carrillo (University of Extremadura), Antonio Torralba (University of Sevilla)
118 "1.5V Current-Mode CMOS True RMS-DC Converter Based on Class-AB Transconductors",Carlos A. De La Cruz Blas, Antonio Lopez-Martin, Alfonso Carlosena (Public University of Navarra), Jaime Ramirez-Angulo (New Mexico State University)
119 "New Low-voltage High Performance WTA Circuits based on Flipped Voltage Followers",Jaime Ramirez-Angulo, Gladys Omayra Ducoudray (Klipsch School of Electrical and Computer Engineering, New Mexico State University), Ramon Carvajal (University of Sevilla), Antonio Lopez-Martin (University of Navarra)
120 "New Low-Voltage Fully Programmable CMOS Triangle/Trapezoidal Function GeneratorCircuit", Megraj Kachare, Jaime Ramirez-Angulo (Klipsch School of Electrical and ComputerEngineering, New Mexico State University), Antonio Lopez-Martin (Dept. of Electrical andElectronic Engineering, Public University of Navarra, Pamplona (Spain)), Ramon Carvajal(Dpto. de Ingenieria Electronica, Escuela Superior de Ingenieros, Universidad de Sevilla (Spain))
121 "Low-Voltage Micropower Integrated CMOS Log Domain Filter", Antonio J. Lopez-Martin,Carlos A. De La Cruz Blas, Alfonso Carlosena (Public University of Navarra)
XII
122 Session 5c : SOC & Analog Test
123 "An Infrastructure and Application Specific Processor for Testing Analogue and Mixed-SignalSoCs", Francisco Duarte (INESC Porto), Jose Machado da Silva, Jose Alves, Jose Matos(FEUP, INESC Porto)
124 "Test Planning for Mixed-Signal SoCs and Analog BIST: a Case Study ", Luigi Carro(Departamento Eng. Elétrica - Universidade Federal do Rio Grande do Sul), Erika Cota, Marcelo Negreiros (Instituto de Informática - Universidade Federal do Rio Grande do Sul), Marcelo Lubaszewski (Instituto de Microelectrónica de Sevilla - Centro Nacional de Microeléctronica), Antonio Andrade Jr. (Departamento Eng. Elétrica - Universidade Federaldo Rio Grande do Sul)
125 "BIST X-Y Zoning Detector based on Quasi-floating Gate Structure", Ricard Sanahuja, Victor Barcons, Luz Balado, Joan Figueras (Universitat Politècnica de Catalunya (UPC))
126 "An SC Spectrum Analyzer for Testing Analog Circuits", Miguel A. Domínguez, José L. Ausín (University of Extremadura), Guido Torelli (University of Pavia), J. Francisco Duque-Carrillo (University of Extremadura)
127
128 "Hardware Requirements for Testing M-S Circuits based on Multidimensional LissajousCurves", Emili Lupon, Luz Balado, Lucas García, Joan Figueras (Departament d'EnginyeriaElectrònica -Universitat Politècnica de Catalunya)
129 Session 5d : RF Building Blocks
130 " Synchronous Oscillator Locked Loop: A New Delay Locked Loop Using Injection LockedOscillators as Delay Elements.", Franck Badets, Mohamed Benyahia, Didier Belot(STMicroelectronics Central R&D Crolles)
131 "A Fully Integrated Mixer in CMOS 0.35µm Technology for 802.11a WIFI Applications",Roberto Diaz , Ruben Pulido, Amaya Goni-Iturri (Institute for Applied Microelectronics,University of Las Palmas de Gran Canaria, Las Palmas, Spain), Sunil Lalchand Khemchandani, Benito Gonzalez (Institute for Applied Microelectronics and Departamentode Ingenieria Electronica y Automatica, University of Las Palmas de Gran Canaria, Las Palmas, Spain), Javier del Pino (Institute for Applied Microelectronics, University of Las Palmas de Gran Canaria, Las Palmas, Spain)
132 "Effect of Mismatch and Delay on the Quadrature Cross-Coupled RelaxationOscillator/Mixer", Luís Bica Oliveira, Jorge R. Fernandes (IST/INESC-ID)
133 "High-gain LNA in 0.18 µm CMOS Technology for a WLAN Receiver", Iñigo Adin (CEIT), Guillermo Bistué (Tecnun), Carlos Quemada, Hector Solar, Jorge Presa (CEIT), Jon Legarda (Tecnum)
134 "Microwave Low Noise HEMT Gate Mixers ", Faiza Amrouche, Rachid Allam, Jean-MariePaillot (LAII-université de poitiers)
"A Test Methodology to Compute Typical LNA Characterization Parameters", Gabriel Pinho(Faculdade de Engenharia da Universidade do Porto), Jose Machado da Silva, Helio Mendonca, Jose Matos (Faculdade de Engenharia da Universidade do Porto - INESC Porto)
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135 "A 2.45GHz Low Phase-Noise CMOS ", Vincent Cheynet de Baupré, Lakhdar Zaid,Wenceslas Rahajandraibe (L2MP - Polytech), Gilles Bas (STMicroelectronics)
136 Session 6a : New Architectures & Rapid Prototyping and Debugging
137 "100 MHz Floating Point Processing Unit – A Feasibility Study ", Mauro Castelli, Erwin Ofner(Carinthia Tech Institute)
138 "A Linear Sorter Core based on a Programmable Register File", Lluís Ribas, David Castells,Jordi Carrabina (Universitat Autònoma de Barcelona (UAB))
139 "Rapid Prototyping Environment for CMOS Camera IC and Systems Design", Enric Pons(Universitat Autonoma de Barcelona), Jose Luis Merino, Lluis Teres (Centre Nacional de Microelectronica), Jordi Carrabina (Universitat Autonoma de Barcelona)
140 "Advances in Real-time Hardware Debugging using the UNSHADES System", J. Tombs,M.A. Aguirre Echanove, F. Muñoz, V. Baena, A. Torralba (Universidad de Sevilla)
141 Session 6b : Sigma-Delta Modulator Design (1)
142 "A 3-30 MHz Tunable Continuous-Time Bandpass Sigma-Delta A/D Converter for DirectConversion of Radio Signals", David Bisbal, Jacinto San Pablo, Jesús Arias, LuisQuintanilla, José Vicente, Juan Barbolla (Universidad de Valladolid)
143 "A New Method for the High-Level Synthesis of Continuous-Time Cascaded Sigma-DeltaModulators", Ramón Tortosa, José M. de la Rosa, Angel Rodríguez-Vázquez, Francisco V. Fernández (Instituto de Microelectrónica de Sevilla, IMSE-CNM (CSIC))
144 "Modeling All-MOS Log-Domain Sigma-Delta A/D Converters", Xavier Redondo (CentreNacional de Microelectrònica, CSIC), Jofre Pallarès (Barcelona International R&D Core, CNM-Epson), Francisco Serra-Graellls (Centre Nacional de Microelectrònica, CSIC)
145 "A Dual-Band Sigma-Delta Modulator for GSM/WCDMA Receivers", Ana Rusu, Babita Roslind Jose, Mohammed Ismail, Hannu Tenhunen (Royal Institute of Technology (KTH))
146 Session 6c : Analog Test
147 "Probabilistic and Simulation-Based Masked-BIST Implementation" , Fernando Guerreiro(INESC-ID ), José Miguel Fernandes (Schindler, SA), Marcelino Santos, Arlindo Oliveira, Isabel Teixeira, Paulo Teixeira (IST / INESC-ID)
148 "Experimental Analysis of Transient Current Test Based on dIDD Variations in S2I Memory Cells", Yolanda Lechuga, Roman Mozuelos, Miguel Angel Allende, Mar Martinez, SalvadorBracho (Microelectronics Engineering Group (University of Cantabria))
149 "Testing of RF Systems by Zoning the Constellation Diagram", Daniel Arumí Delgado, RosaRodríguez Montañés, Joan Figueras (Universitat Politècnica de Catalunya)
XIV
150 "On the Minimum Number of Measurements for Single Fault Diagnosis in Linear Circuits", Jose Soares Augusto (INESC-ID/FCUL - Physics Dept.)
151 Session 6d : Mixed Signal Circuits for RF Applications
152 "Digitally Programmable UHF Transconductor in Digital CMOS Technologies", AranzazuOtin, Santiago Celma, Concepcion Aldea (Electronic Design Group.University of Zaragoza)
153 "A Mixed-Signal ASIC for FM-DCSK Modulation", Manuel Delgado-Restituto, Antonio J. Acosta, Angel Rodríguez-Vázquez (Instituto de Microelectrónica de Sevilla, IMSE-CNM (CSIC))
154 "A Multi-functional Approach of Frequency Synthesizer dedicated to the next Multi-standardSmart Objects", Christophe Rougier, Jean-Baptiste Begueret, Hervé Lapuyade, Yann Deval (IXL Laboratory), Angelo Malvasi (ACCO)
155 "A Local Oscillator with a Reconfigurable Direct Digital Synthesis System", João Gonçalves,Jorge R. Fernandes (I.S. Técnico/INESC-ID)
156 Session 7a : Analog Design Methods
157 "Optimization Design of Stacking Voltage Triplers for Capacitive Load", Ming Zhang, NicolasLlaser, Dariga Meekhun (University of south paris)
158 "Design Considerations of a Frequency Synthesizer for a Mixed-Signal Built-In-Self-TestApplication", Africa Luque, Diego Vazquez, Adoracion Rueda (Instituto de Microelectronicade Sevilla)
159 "Analog IC Design With A Library Of Parameterized Device Generators", Vincent Bourguet,Laurent de Lamarre, Marie-Minerve Louerat (University of Paris VI, LIP6-ASIM Laboratory)
160 "An Accurate Algorithm for Transistor Sizing in Analog CMOS Design", Pablo Rodiz-Obaya,Juan J. Rodriguez-Andina (Universidad de Vigo), Jaime Ramirez-Angulo (New Mexico State University)
161 "Optimizing SI Class AB Memory Cells", Mourad Fakhfakh, Mourad Loulou, Nouri Masmoudi(LETI-ENIS Sfax Tunisia)
162 "The Importance of Microwave Approach for High Frequency MOS Analog Designers", GillesPetit, Richard Kielbasa (Service des Mesures, Supélec), Vincent Petit (Thales AirborneSystems)
163 Session 7b : New Synchronisation schemes and Asynchronous Circuits
164 "Secured Structures for Secured Asynchronous QDI Circuits" , Alin Razafindraibe, Philippe Maurine, Michel Robert (LIRMM), Fraidy Bouesse, Bertrand Folco, Marc Renaudin (TIMA)
165 "Definition of P/N Width Ratio for CMOS Standard Cell Library ", Alexandre Verle, Philippe Maurine, Nadine Azemard, Daniel Auvergne (LIRMM)
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166 "DPA on Quasi Delay Insensitive Asynchronous Circuits: Concrete Results", Fraidy Bouesse, Marc Renaudin (TIMA Laboratory ), Bruno Robisson, Edith Beigne (CEA-Grenoble), Pierre-Yvan Liardet, Solenn Prevosto (STMicroelectronics, ZI Rousset)
167 "Four Phase Alternating Latches Clocking Scheme for CMOS Sequential Circuits", David Guerrero, Manuel Jesús Bellido, Jorge Juan Chico, Alejandro Millán, Paulino Ruiz de Clavijo, Enrique Ostua (Instituto de Microelectrónica de Sevilla-Centro Nacional de Microelectrónica/Departamento de Tecnología Electrónica-Universidad de Sevilla)
168 "A Memoryless Clock Domain Adaptation Unit IP", Roberto Esper-Chaín, Félix Tobajas,Francisco González, Rubén Arteaga, Roberto Sarmiento (Instituto Universitario de Microelectrónica Aplicada)
169 "Synchronization of Sequential Circuits using the Asynchronous Wave PipeliningTechnique", Stephan Hermanns, Sorin Alexander Huss (Integrated Circuits and Systems, Darmstadt University of Technology)
170 Session 7c : Failure Analysis & Reliabiliy
171 "Flash Memory Cell: Threshold Voltage Sensibility to Geometry", Bertrand Saillet(STMicroelectronics), Jean-Michel Portal (L2MP-Polytech), Didier Née (STMicroelectronics)
172 "Modeling the Influence of Time Skew on Crosstalk Induced Delay in Submicron CMOSTechnologies", Jose L Rossello, Jaume Segura (Universitat de les Illes Balears)
173 "RC on-chip Interconnect Performance Revisited", Philippe Maurine, Nadine Azemard,Daniel Auvergne (LIRMM)
174 "Scalable Substrate Modeling based on 3D Physical Simulation", Sebastien Fregonese(IXL), Didier Celi (ST MICROELECTRONICS), Thomas Zimmer, Cristell Maneux, Pierre Yvan Sulima (IXL)
175 "Extraction of MOSFET Parameters Using Fourier-Space Techniques ", Rodrigo Picos(University of Balearic Islands), Miquel Roca (University of Balearic Islands), BenjaminIñiguez (Universitat Rovira i Virgili), Eugenio Garcia-Moreno (University of Balearic Islands)
176 "Impact of Deterministic Within-Die Variation on the Circuit Performance in NanoscaleSemiconductor Manufacturing", Munkang Choi, Seyed-Abdollah Aftabjahani, Cheng Jia,Linda Milor (Georgia Institute of Technology)
177 Session 7d : Low Power / Low Voltage : analog circuits (2)
178 "A Low-Dropout Voltage Regulator for Biomedical Integrated Systems", Rui Martins,Francisco Vaz (University of Aveiro/IEETA)
179 "A Low-voltage, Low Power Wide-linear Range Subthreshold OTA ", Aimad El mourabit,Guo-Neng Lu, Patrick Pittet (LENAC, Université Claude Bernard - Lyon1)
180 "A Low Voltage I/O Interface for High Speed Buses in GaAs Technology", Roberto Esper-Chaín, Félix Tobajas, Roberto Sarmiento (Instituto de Microelectrónica Aplicada)
XVI
181 "New Low Voltage Class-AB CMOS Unity Gain Buffer and Current Mirror", Antonio Torralba, Ramón G. Carvajal, Mariano Jiménez, Fernando Muñoz (Universidad de Sevilla (SPAIN)), Jaime Ramírez-Angulo (New Mexico State University, USA)
182 "New Low-voltage Class AB/AB CMOS Op-Amp with Rail-to-Rail Input/Output Swing", Milind-Subhash Sawant, Shanta Thoutam, Jaime Ramirez-Angulo (New Mexico State University), Antonio Lopez-Martin (Universidad Publica de Navarra), Ramon G. Carvajal (Escuela Superior de Ingenieros Universidad de Sevilla)
183 "A New Family of Low-Voltage Power-Efficient Class AB CMOS OTAs", Sushmita Baswa (New Mexico State University), Antonio J. Lopez-Martin (Public University of Navarra), Jaime Ramirez-Angulo (New Mexico State University), Ramon G. Carvajal (University of Sevilla)
184 Session 8a : Sigma-Delta Modulator Design (2)
185 "Continuous-Time Sigma-Delta Modulator with Exponential Feedback for Reduced JitterSensitivity ", Jacinto San Pablo, David Bisbal, Luis Quintanilla, Jesus Arias, LourdesEnriquez, Juan Barbolla (Universidad de Valladolid)
186 "Implementation of an RTZ code for feedback DAC on a Sigma-Delta Modulator", Jofre Pallarès, Xavier Redondo (Institut de Microelectrònica de Barcelona-CNM, Spain), Justo Sabadell (Barcelona Branch Office, Epson Europe Electronics, GmbH), Francesc Serra-Graells (Institut de Microelectrònica de Barcelona-CNM, Spain)
187 "Excess-Loop delay reduction on Low-OSR High-Speed Multi-bit Continuous-Time Sigma-Delta Modulators", Susana Paton (Universidad Carlos III), Thomas Poetscher, Antonio Di Giandomenico, Klaus Kolhaupt (Infineon Technologies), Luis Hernandez (Universidad CarlosIII), Andreas Wiesbauer (Infineon Technologies)
188 "Discrete Invariant Set Algorithm for Sigma Delta Modulators Dynamics Analysis", David Camarero de la Rosa, Van-Tam Nguyen, Jean-François Naviner, Patrick Loumeau (EcoleNationale Supérieure des Télécommunications.)
189 Session 8b : Digital Test
190 "Optimization of Digitally Coded Test Vectors for Mixed-Signal Components", AhcèneBounceur, Salvador Mir, Emmanuel Simeu (TIMA Laboratory)
191 "Improving the Efficiency of Arithmetic BIST by Combining Targeted and General PurposePatterns", Salvador Manich, Lucas García, Luz Balado, Josep Rius, Rosa Rodriguez, JoanFigueras (DEE-UPC)
192 "Automatic Verification of RT-Level Microprocessor Cores Using Behavioral Specifications: aCase Study", Ernesto Sanchez, Matteo Sonza Reorda, Giovanni Squillero (Politecnico di Torino), Raoul Velazco (TIMA-CMP Laboratory)
193 "Solving the State Justification Problem using MILP for RTL Specifications", H. Navarro, Juan A. Montiel-Nelson, J. Sosa, José C. García (IUMA, Institute for Applied Microelectronics)
XVII
194 Session 8c : IP-based design
195 "Hiding Technique for Intellectual Property Protection on FPGAs", Luis Parrilla, EncarnacionCastillo, Antonio Garcia, Antonio Lloris (Universidad de Granada)
196 "Rapid Integration of IPs in System on Chips", Salim Ouadjaout (M3Systems), DominiqueHouzet (IETR INSA Rennes)
197 "Block Constraints Budgeting in Timing-Driven Hierarchical Flow", Olivier Omedes(CADENCE DESIGN SYSTEMS), Michel Robert (LIRMM UMR CNRS Montpellier II), Mohamed Ramdani (ESEO Angers)
198 "Flexible HW/SW Implementation of MPEG Systems using FPGA Platforms", Oscar Navas, Antoni Portero, David Novo, Jordi Carrabina, Jordi Escrig, Martí Bonamusa (UniversitatAutònoma Barcelona)
199 Session 8d : Low Power Digital Design
200 "Analytical Estimation of Node Activity in Ripple Carry binary Adders", Antonio Calomarde,Antonio Rubio (Universitat Politècnica de Catalunya)
201 "Increase in Energy Consumption due to Multiple Transitions in Coupled Lines", EugeniIsern, Miquel Roca, Francesc Moll (Universitat Politecnica Catalunya UPC)
202 "Power Characterization of RAMs. An Experimental Approach.", Javier Rellán, José L. Ayala, Marisa López-Vallejo (Departamento de Ingeniería Electrónica (UniversidadPolitécnica de Madrid))
203 "Optimization Protocol based on Performance Metric", Xavier Michel, Alexandre Verle,Nadine Azemard, Philippe Maurine, Daniel Auvergne (LIRMM)
204 Session 9a : Logic and Architectural Synthesis
205 "Assertion Checking of Cyclic Behavioral Descriptions", Iñigo Ugarte, Pablo Sanchez (University of Cantabria)
206 "Behavioural Synthesis for Low Power Applying Operation Transformations", Maria CarmenMolina, Rafael Ruiz-Sautua, Jose Manuel Mendias, Roman Hermida (UniversidadComplutense de Madrid)
207 "Clock Cycle Length Minimization by Arrival Time Aware Scheduling", Rafael Ruiz-Sautua,Maria Carmen Molina, Jose Manuel Mendias (Universidad Complutense de Madrid)
XVIII
208 Session 9b : Communications Systems
209 "An Efficient Priority Queuing System for High Speed Network Processors with QoSSupport", Félix B. Tobajas, Valentín De Armas (Instituto Universitario de MicroelectrónicaAplicada (IUMA) / Departamento de Ingeniería Electrónica y Automática (DIEA)), NéstorCruz(IUMA), Roberto Esper-Chaín (IUMA / DIEA), Rubén Arteaga (IUMA), RobertoSarmiento (IUMA / DIEA)
210 "Optimizations in DVB-RCS Turbo Decoder based on Trellis Structure", Jesus M. Perez Llano, Victor Fernandez Solorzano (Cantabria University)
211 "Implementation of a 2.5Gbps ATM over SDH Transceiver with Add/Drop on a Virtex-II", Rubén Arteaga Mesa, Roberto Esper-Chaín Falcón, Oscar Tubío Araújo, Félix TobajasGuerrero, Valentín De Armas Sosa, Roberto Sarmiento Rodríguez (Instituto Universitario deMicroelectrónica Aplicada (IUMA))
212 "ITU-Compliant Macrocells for Dual Tone Multiple Frequency Transmission and Reception",Arturo Purroy, Isidro Urriza (Universidad de Zaragoza)
213 "Evaluation of a PHM Scheduler Implementation", Francisco Javier González-Castaño(Departamento de Ingeniería Telemática, Universidad de Vigo), Enrique Soto-Campos(Departamento de Tecnología Electrónica, Universidad de Vigo), Rafael Asorey-Cacheda(Departamento de Ingeniería Telemática, Universidad de Vigo), Cristina López-Bravo(Departamento de Tecnologías de la Información y las Comunicaciones, Universidad Politécnica de Cartagena), José Fariña-Rodríguez, Juan José Rodríguez-Andina(Departamento de Tecnología Electrónica, Universidad de Vigo)
214 Session 9c : System Level Design
215 "XML Specification and Tools for Automatic SoC Generation", Màrius Montón, Oriol Font, Jaume Joven (Universitat Autonoma de Barcelona), Pere Garcia (EPSON ElectronicsEurope), Lluís Terés (Centro Nacional de Microelectronica), Jordi Carrabina (Universitat Autonoma de Barcelona)
216 "Generic Programming with abstract parametrized components", Fernando Rincón, JesúsBarba, Juan Carlos López, Juan Pablo Rozas (University of Castilla-La Mancha)
217 "System Level Design using SystemC: a Case Study of Block Turbo Decoder", Erwan Piriou,Christophe Jego, Patrick Adde, Michel Jezequel (ENST-Bretagne)
218 "Object-Oriented Hardware/Software Co-Simulation Using SystemC", Ana Mª Cardells,Juan-José Noguera (Hewlett Packard BPO Spain), Lluís Terés (Centre Nacional de Microelectrònica)
219 "Comparing Design Flows for Structural System Level Specifications facing FPGAPlatforms", David Castells, Marius Monton, Ramon Pla, David Novo, Antoni Portero, OscarNavas (Universitat Autonoma Barcelona)
220 Session 9d : Noise in Electronics
221 "Minimum Noise Figure Comparison of Y-Parameter Based Bipolar Noise Models.", Juan Carlos Milena, Manuel Sanchez, Juan Miguel Lopez (Universitat Politècnica de Catalunya),Antonio J. Garcia (Universidad de Santiago de Compostela)
XIX
XX
222 "Spectral Characterization of the Digital Noise", Miguel Angel Méndez Villegas, José Luis González Jiménez, Diego Mateo Peña, José Antonio Rubio Solá (Universitat Politècnica de Catalunya)
223 "On the Relation between Digital Circuitry Characteristics and Power Supply Noise Spectrum in Mixed-Signal CMOS IC", Miguel Ángel Méndez, José Luis González, Enrique Barajas, Diego Mateo, Antonio Rubio (Electronic Engineering Department, Universitat Politècnica de Catalunya)
Conference Committees Steering Committee
Daniel Auvergne LIRMM, FSalvador Bracho del Pino U. de Cantabria, E
Rafael Burriel Lluna CeDInt, U. Politécnica Madrid. E Fulvio Corno Politecnico Torino, I
Joan Figueras Pàmies U. Politécnica Catalunya, E José Epifanio da Franca Inst. Superior Técnico, P
Leopoldo García Franquelo U. de Sevilla, EEugenio García Moreno U. Illes Balears, E
Miguel A. Hernández y Coll Siemens A.G. Munich, D José Luis Huertas Díaz CNM Sevilla, E
Juan Carlos López López U. Castilla-La Mancha, E Antonio Núñez Ordóñez U. Las Palmas G. Canaria, E
Emilio Olías Ruiz U. Carlos III, EMichel Renovell LIRMM, F
Armando Roy Yarza U. de Zaragoza, EAntonio Rubio Solá U. Politécnica de Catalunya, E
José A.R. Silva Matos U. Porto, PAntonio J. Torralba Silgado U. de Sevilla, E
Javier Uceda Antolín U. Politécnica Madrid, E
General ChairPascal Fouillat ENSEIRB
Programme Co-ChairsMaria Luisa López Vallejo U.Politécnica de Madrid
Jean Tomas U. Bordeaux 1
Local Organizing Committee Stéphane Azzopardi Valérie Cauhapé
Dominique Dallet Yann DevalRégis Devreese Geneviève DuchampIsabelle Dufour Eric Kerhervé
Hervé Lapuyade Nathalie MalbertNicolas Moll Sylvie Renaud
Sylvain Saïghi Angélique TételinPatrick Villesuzanne Jean-Michel Vinassa
Local SecretariatValérie Cauhapé Laboratoire IXL
Université Bordeaux 1 351 Cours de la Libération
33405 Talence Cedex - FRANCETel: +33 (0) 540 002 807Fax: +33 (0) 556 371 545
Registration and Hotel accomodationDominique Aurieres
SUD CONGRES CONSEIL - DCIS'04166 cours du Maréchal Galliéni - 33400
Talence – FRANCEFax : +33 (0) 556 249 948
XXI
Reviewers
Abouchi,N.Acosta Jiménez,A.J.Aguiar,R.L.Aguirre Echanove,M.A.Alarcón,E.Alcubilla,R.Alexandre,A.AlexandresFernández,S.Alves,J.C.Amendola,G.Aragonés,X.Arapoyanni,A.Aubepart,F.Aubry,J.-F.Augusto,J.Auvergne,D.Ayala,J.L.Azcondo,F.J.Badets,F.Ballester Merelo,F.J.Barthelemy,H.Bausells,J.Begueret,J.B.Belhaire,E.Bellido Diaz,M.J.Bota,S.Bourdel,S.Bracho,S.Burriel,R.Campo,E.Canas Ferreira,J.Capraro,S.Carmona,R.Carrera Usiabaga,A.Celma,S.Charlot,B.Chatelon,J.P.Crand,S.Dallet,D.Dejous,C.Del Rio Fernandez,R.Deltimple,N.Deval,Y.Dilillo,L.Dualibe,C.Duchamp,G.Dufour,I.
Erwin,O.Farina Rodriguez,J.Fernandez,A.Ferreiros,J.Ferrer,C.Figueras,J.Fischer,V.Fouillat,P.Garcia Franquelo,L.Garcia Moreno,E.Garda,P.Girard,P.Granado,B.Hebrard,L.Hermida,R.Hernandez,A.Herve,Y.Houzet,D.Isern,E.Izpura,I.Jacquemod,G.Kerherve,E.Landrault,C.Lapuyade,H.Levant,J.-L.Leveugle,R.Levi,H.Lewis,D.Lewis,N.Linan Cembrano,G.Lopez Nozal,L.A.Lopez Vallejo,M.Lopez,C.Lopez,J.C.Lopez-Villegas,J.M.Lorenz,M.G.Louerat,M.-M.Luxey,C.Machado da Silva,J.Madrenas,J.Mancini,S.Maneux,C.Manich,S.Marc,F.Martin,J.L.Martinez Salamero,L.Martinez,M.Mengibar,L.
Meresse,A.Mieyeville,F.Mir,S.Molina,M.C.Montiel-Nelson,J.A.Moreno Arostegui,J.M.Moya,F.Moya,J.M.Navarro,D.Naviner,J.-F.Nebel,W.Nouet,P.O'Connor,I.Olías Ruiz,E.Oliver,J.Ortiz-Conde,A.Ousten,Y.Pérez Verdú,B.Petit,G.Petrashin,P.Pinna,A.Pissaloux,E.Psychalinos,C.Quero,J.Ramdani,M.Rebiere,D.Renaud,S.Renovell,M.Ribas,L.Ribeiro Alves,G.Rincon,F.Rius Vázquez,J.Robert,M.Roca,M.Rodríguez Andina,J.J.Rodríguez,R.Romain,O.Roy,A.Rubio,A.Rueda Rueda,A.Samitier,J.Sanchez Espeso,P.P.Sandoval Hernández,F.Santos,D.M.Santos,H.Santos,M.Sauerer,J.Silva,M.
XXII
Sinclair,D.Stechele,W.Taris,T.Texeira,I.Toledo,L.Tomas,J.Torres,L.
Trullemans,A.-M.Tsiatouhas,Y.Vaz,J.Vázquez García de la Vega,D.Velazco,R.Verdier,F.
Vergos,H.T.Vidal,F.Vieira dos Santos,J.M.Villar,E.Vinassa,J.Zimmer,T.
XXIII
Plenary Sessions
Software Defined Radio : Theory and Applications
Ernesto Perea
STMicroelectronics
A brief overview of Software Defined Radio (SDR) principle yields the required
characteristics for some of the key building blocks. Although the analog-to-digital converter appears to be a severe bottleneck as expected, the huge bit stream the system has to deal with generates the strongest constraints on the Digital Signal Processor. It is demonstrated that a sampled-analog signal processing approach can solve this problem – and initiates others.
Space Electronics : a Challenging World for Designers
Christian Poivey & Kenneth LaBel
Goddard Space Flight Center, NASA
Christian Poivey will address in his talk the concern of Radiation effects for the design of space electronics systems. He will describe first the radiation environment and how this environment affects electronics parts and embedded systems. A special focus will be given on CMOS devices. Then, examples of radiation effects on spacecraft will be presented. The talk will end with a short description of hardening by design methods for CMOS electronics devices.
- 3 -DCIS 2004
Panel Discussion
Panel Discussion
The Bologna Process : Return on Experiment
Moderator:
Prof. Yves Danto (U Bordeaux 1)
Participants:
Prof. Olivier Bonnaud (U. Rennes 1)
Prof. Fausto Fantini (U. de Modena)
Prof. López Barrio (U. Madrid)
Prof. José Silva Matos (U. Porto)
Prof. João Paulo Teixeira (U. Lisboa)
- 5 -DCIS 2004
Exhibits
MicrowindAn introduction to nano-scale CMOS cell design
Etienne SICARDProfessor, INSA
135 av de Rangueil, 31077 Toulouse France
Sonia BENDHIASenior Lecturer, INSA
135 av de Rangueil, 31077 Toulouse France
Abstract: Microwind is a friendly windows-based tool for designing and simulating microelectronic CMOScells at layout level. The tool features full editing facilities, various views (MOS characteristics, 2D crosssection, 3D process viewer), and a high performance built-in analog simulator. Microwind aims at illustrating the technology scale down, the major improvements allowed by nano-scale technologies, as well as main substrate options (buried layer, SOI, RF). The n-channel and p-channel MOS devices, simple/double/triple oxide,simple/double-gate, are illustrated and simulated based on BSIM4 models. Basic cells such as Inverters, logicgates, complex gates, arithmetic blocs, latches can be designed, simulated and optimized in a very efficient way with Microwind. A specific effort has been dedicated to the handling of static, dynamic, non-volatile andmagnetic memories. Furthermore, radio-frequency analog cells, such as mixers, voltage-controlled oscillators,fast phase-lock-loops and power amplifiers are also illustrated by Microwind. Finally, input/output interfacingprinciples, electrostatic discharge protections, pad structure, and package are also covered through numerousexamples. Technologies ranging from 1µm down to 65nm are supported.The tool runs on Windows 98, 2000, NT, and XP. Microwind is used in more than 500 Universities around the world and in industry training centers. The tool has proven very efficient in the illustration of CMOS technology and design principles, either for teachers during their lecture or for students realizing integrated logic or analogfunctions as practical training.
Input clock at 2.44GHz
Precharge of Vc to VDD/2
VCO output reaches 2.44GHz
Vc fluctuation and stabilization
Phase detector stabilized here
Figure 1: Main layout design window Figure 2: Analog simulation (PLL)
Figure 3: 3D view of the process (90nm) Figure 4: Tutorial on MOS device (BSIM4)
Free copies of the complete package+ manual will be available at DCIS'04.
- 7 -DCIS 2004
IC-EmitComparing simulated/measured Parasitic Emission of Integrated Circuits
Etienne SICARDINSA-Lesia
Toulouse, [email protected]
Amaury SOUBEYRANEads-CCR
Suresnes, [email protected]
Abstract: IC-Emit is a Windows-based environment for the simulation of parasitic emission of integrated circuits. The tool consists of a dedicated schematic editor, an IBIS translator, a core activity evaluator, an analog SPICE simulator and a dedicated post-processor. The IBIS translator gives information about the input/output characteristics and the package and supply model. The core activity evaluator translates the integrated circuitspecification into a current source which aims at modeling the core switching noise and on-chip decoupling. The analog simulation is performed by WinSpice, and a post processing features an immediate comparison of predicted and measured spectrum in frequency domain. IC-Emit handles a set of standards for integrated circuit modeling, emission modeling and test setups. IC-Emit can be downloaded from www.ic-emc.org.The tool has been used to modelize successfully the parasitic emission of 16-bit, 32-bit microcontrollers, Xilinx programmable devices as well as dedicated ASICs, within the range 1MHz-2GHz.
The freeware runs on Windows 95,98, NT, XP.
Figure 1: Schematic editor window Figure 2: Core activity evaluator
Figure 3:IBIS loader (Xlinx VirtexII Bga 896) Figure 4: Simulated/measured emission (16bit µc)
Free copies of the package and the manual will be available at DCIS'04.IC-emit has been developped within MEDEA+ "Mesdie" project A-509
- 8 -DCIS 2004
Session 1a
CAD Tools and Optimisation Algorithms
Wednesday nov. 24 9h00 10h00, Lacanau Room
Chairs
Lluis Ribas (U. Autonòma de Catalunya) Eugeni Isern (U. de les Illes Balears)
BNSAT: Representing Boolean Functions in a Non-Canonical Form
J. Saiz, J. Cortadella*, L. Ribas, J. Carrabina Computer Science Department, Universitat Autònoma de Barcelona, Bellaterra, Spain
Joaquin.Saiz Lluis.Ribas [email protected]*Software Department, Univ. Politècnica de Catalunya, Barcelona, Spain, [email protected]
ODERN SAT solvers, which implement improved variants of the Davis-Putnam algorithm,
can determine the satisfiability of large CNF formulae in a few seconds. This fact has
favoured the development of non-canonical methods of representing Boolean functions. In this paper,
we introduce BNSAT, a new package that implements a non-canonical representation of Boolean
functions.
The central idea behind BNSAT is representing a Boolean function F as a composition of
small functions fi. The main data structure employed in BNSAT is a cyclic directed graph with specific
features, resembling a Boolean network. Each non-terminal vertex has an n-variable function f
associated to it. Functions fi are represented by means of BDDs, and BDD variables are shared among
several vertices. As a consequence, a BDD node can be used in the representation of the functions fi of
different vertices. Such reusing of BDD nodes involves a great saving of memory.
Two parameters, the maximum number of fanin nodes and the maximum number of BDD
nodes, control the size of the functions fi. BNSAT package can compute the usual Boolean operations.
The most intuitive way of doing a binary operation among two Boolean functions F1 and F2 is by
operating the BDDs of the vertices that represent those functions. However, limits imposed by the
aforementioned parameters can be exceeded. To avoid this problem, one of the operands (or both) can
be replaced by a BDD variable. It gives rise to four different methods of implementing a binary
operation.
Different strategies have been studied in order to determine the satisfiability of a Boolean
function F represented with BNSAT. By the time being, the strategy that best performs consists of a
specific method of translation into CNF formulae in conjunction with the use of a SAT solver, Zchaff.
The translation into CNF format is based on the use of ESPRESSO.
We have tested BNSAT on some common combinational circuit benchmarks in BLIF format.
It can be seen that the combination of BNSAT and Zchaff outperforms the combination of Zchaff and
a direct BLIF-CNF translator in most cases. In the short term, we are going to implement
Quantification Boolean Formulae, which are widely used in formal verification algorithms. Some of
these algorithms (for instance, reachability analysis) will be implemented in order to explore the
efficiency of BNSAT in the field of formal verification.
M
- 10 -DCIS 2004
SUSANA: a MOS-Mixed-Circuit Simulator Using Logic/ELogicAlgorithms implemented in Python
Tiago Carrisosa (*), Tiago Felix (*), Miguel Jeronimo (*), J. Soares Augusto (*,**)
(*) INESC-ID, R. Alves Redol, 9, 1000-029 Lisboa, Portugal
(**) Physics Dept, Fac. Ciencias da Univ. de Lisboa
In this paper we describe the simulator SUSANA (”Alternative Numerical Algorithms-based Simulator”),
based on the ELogic1 simulation approach and implemented in Python/wxPython.
ELogic, an event-driven simulation algorithm, traditionally used in digital MOS circuits simulation, is also
suitable for simulating analogue and mixed circuits.
SUSANA was applied to large digital ISCAS85 benchmark circuits. Several improvements have been added
to standard ELogic, such as the implementation of a logic simulator to obtain initial conditions before starting
ELogic simulation of digital circuits.
The precision of simulation can be controlled by the user through the number of discrete states (voltages)
allowed for the circuit nodes. The smaller the number of states, the faster the simulation but, also, in this case
the simulation error becomes larger. ∆V , the voltage difference between adjacent states, controls their number.
The use of a very high level programming language (Python) permitted the rapid development, test and
debugging of a quite complex circuit simulator and of the associated visual input and data analysis components.
Examples, results and a description of the simulation environment are presented in the full paper.
The efficiency of SUSANA when compared to Spice, despite being written in Python, is clearly shown in
table 1. In the simulation of the ALU shown in fig. 1 a speed-up of 12X was observed.
SIMULATOR TIME (s) REL. SPEED
Spice 4557 1
SUSANA (∆V =0.25 V) 380 12
Table 1: ALU simulation run times in Spice and in SUSANA
The simulation of pulse propagation in a chain of 1000 inverters has shown a speed-up of 168X due, in part,
to the use of a digital simulator to initialize correctly the ’digital’ values.
Figure 1: Simulation of ALU C3540 with SUSANA (∆V = 0.25 V).
1R. Saleh, S.-J. Jou and A. R. Newton, Mixed-Mode Simulation and Analog Multilevel Simulation, Boston, Massachusetts:
Kluwer Academic Publishers, 1994.
- 11 -DCIS 2004
A Distributed Enhanced Genetic Algorithm Kernel Applied to a Circuit/Level Optimization E-Design
Environment
1,2M. Barros, 2G. Neves, 1J. Guilherme and 2N. Horta
1IPT – Inst. Pol. de Tomar Qt. do Contador – Est. de Serra
2300-313 Tomar, Portugal
2IST/IT - Centre for MicrosystemsAv. Rovisco Pais, 1
1049-001 Lisboa, Portugal
Emails: [email protected], [email protected], [email protected], [email protected]
THIS paper presents a distributing implementation of a circuit/system-level optimization E-Design
environment (fig.1) based on an enhanced modified genetic algorithm kernel. First, we discuss the
main features of the optimization kernel such as automatic search space decomposition, premature
convergence prevention procedures and the ability to optimize a broad range of circuits based on
either an equation-based approach or an simulation-based approach, using Spice-like simulators. Then,
a simple, inexpensive and efficient distributed processing method applied to the serial genetic
algorithm is described. Finally, the achieved increase on optimization efficiency, compared to the
standard genetic algorithm implementation, as well as the validity of the proposed approach, is
demonstrated by a multi-objective, multi-constraint optimization of some well known circuits.
um
Figure 1. E-Design Front-End.
- 12 -DCIS 2004
A CAD Tool for the Design of RTD Programmable
Gates based on MOBILE1
Héctor Pettenghi, María J. Avedillo, and José M. Quintana
Instituto de Microelectrónica de Sevilla, CNM, Sevilla, SPAIN E-mail: hector, avedillo, [email protected]
Resonant Tunnelling Diodes (RTDs) exhibit a negative differential resistance (NDR) region in their cur-
rent-voltage characteristics which can be exploited to significantly increase the functionality implemented by
a single gate in comparison to MOS and bipolar technologies, thus reducing circuit complexity. Because of
these attractive features they are receiving much attention as device elements for circuit applications. How-
ever there is a wide gap between research on the device development and automatic tools to design circuits
using them, which can limit the success of this emergent technology. This paper presents a CAD tool for the
design of complex programmable logic blocks (able to implement a set of functions) using RTDs. Starting
from a functional specification, it generates a sized netlist implementing it. The derived circuits exploit the
MOBILE operating principle but increase the logic complexity which can be implemented with a single
gate by rising the number of negative differential resistance devices connected in series, and by the simul-
taneous implementation of functions with such structure. The tool is based on the maximization of the
number of functions which are simultaneously realized (minimization of number of control variables), and
in the formulation of the design problem as a mixed integer linear problem (MILP) with a suitable cost
function which allows minimizing the circuit complexity in terms of device counts. From the solution, the
sized circuit and the control combination for selecting each function are derived.
Figure 1(a), despicts the circuit derived for a 2 input programmable gate that implements the functions
NAND, OR and EXOR. Simulation results in the figure 1(b) show correct operation.
The proposed tool can be useful in translating the attractive features of RTDs to the circuit level.
1. This effort was partially supported by the EU QUDOS project IST 2001-32358.
vbias
x1
x2
y2
y1
y2
y1
C=0
C=1
Figure 1.- A 2-input programmable gate designed with proposed tool.x2x1
1.2
y1
C
1.30.2 0.2 0.2
y2
1.8Vbias
0.60.6
y2 1=
y1 x1 x2⊕=
y1 x1 x2+=
y2 x1x2=
- 13 -DCIS 2004
Session 1b
Data Converter DesignWednesday nov. 24 9h00 10h00, Bordeaux Room
Chairs
Dominique Dallet (E.N.S.E.I.R.Bordeaux) Josep Samitier (U. de Barcelona)
A New Capacitor-Ratio and Offset Independent Amplifier for Pipelined A/D Converters
F. Muñoz, R.G. Carvajal, A. Torralba, B. Palomo
Departamento de Ingeniería Electrónica, Universidad de Sevilla
HE mismatch of ratio capacitors used in the residue amplifier of the first pipelined stages limits the
resolution in high-resolution pipelined Analog-to-Digital Converters (ADC). In this paper, a new
residue amplifier which is inherently insensitive to capacitor mismatch and amplifier offset is
presented. Using a four-phase switched capacitor circuit, the proposed technique (shown in figure 1)
senses and compensates the mismatch capacitor error.
T
Although other ratio-independent residue amplifiers have been proposed in the literature, the
technique proposed here is, in the authors’ knowledge, the only one which allows an operational
amplifier to be shared between two successive pipelined stages, providing, in addition, cancellation of
the amplifier offset. Simulation results show the potentiality of the proposed technique for the design
of very low-power high-resolution pipelined converters.
- A1
+Vout
Vin
- A2
+
C2
C1
C3
Figure 1. Proposed ratio-independent gain-of-two circuit
- 15 -DCIS 2004
Simulation-based High-level Synthesis of Pipeline Analog-to-Digital Converters
Jesús Ruiz-Amaya, José M. de la Rosa and Manuel Delgado-Restituto
Instituto de Microelectrónica de Sevilla IMSE-CNM (CSIC) Ed. CNM-CICA, Av. Reina Mercedes s/n, 41012 Sevilla, SPAIN.
E-mail: ruiz|jrosa|[email protected]*
HIS paper presents a toolbox for the simulation, optimization and high-level synthesis of pipeline
Analog-to-Digital Converters (ADCs) in MATLAB. The embedded simulator uses SIMULINK
C-coded S-functions to model all required subcircuits including their main error mechanisms. This
approach allows to drastically speed up the simulation CPU-time up to 2 orders of magnitude as
compared with previous approaches – based on the use of SIMULINK elementary blocks. Moreover,
S-functions are more suitable for implementing a more detailed description of the circuit. For all
subcircuits, the accuracy of the behavioural models has been verified by electrical simulation using
HSPICE.
For synthesis purposes, the simulator is used for performance evaluation and combined with an
hybrid optimizer for design parameter selection. The optimizer combines adaptive statistical
optimization algorithm inspired in simulated annealing with a design-oriented formulation of the cost
function. It has been integrated in the MATLAB/SIMULINK platform by using the MATLAB engine
library, so that the optimization core runs in background while MATLAB acts as a computation
engine.
The implementation on the MATLAB platform brings numerous advantages in terms of signal
processing, high flexibility for tool expansion and simulation with other electronic subsystems.
Additionally, the presented toolbox comprises a friendly graphical user interface to allow the designer
to browse through all steps of the simulation, synthesis and post-processing of results. In order to
illustrate the capabilities of the toolbox, a 0.13 m CMOS 12-bit@80MS/s A/D interface for power
line communications is synthesized and high-level sized. Different experiments show the effectiveness
of the proposed methodology.
* This work has been supported by the MEDEA+ (A110 MIDAS) Project.
T
- 16 -DCIS 2004
Digital Background Technique for Gain ErrorCorrection in Pipeline ADCs
Antonio J. Ginés, Eduardo J. Peralías and Adoración RuedaInstituto de Microelectónica de Sevilla (España), Centro Nacional de Microelectrónica
IGH-speed high resolution for communications needs good performance of the analogue blocks in data
converters as well as self-correction/self-calibrating techniques. In the particular case of Pipeline
ADCs, correction techniques can improve the linearity of sub-ADCs dealing with the transition errors, but for
resolution greater than 10 bits, a calibration technique is still necessary. Moreover, even for lower resolutions
calibration can relax the analogue block specification, and therefore, should be considered as an additional
design variable.
There exist foreground calibration techniques that need the interruption of the normal converter operation
to start a calibration cycle. Normally, the error measurements are obtained just after power is turned on. Thus,
any miscalibration, environmental change such as temperature, power supply or component aging cannot be
overcome if the system works continuously. However, background calibration allows that, performing error
measurements during the ADC operation.
This paper presents a new digital technique for background calibration of gain errors in Pipeline ADCs.
The proposed algorithm estimates and corrects both the MDAC gain error of the stage under calibration
(SUC) and the global gain error associated to the least significant stages. This process is performed without
interruption of the conversion and without reduction of the dynamic rate. The proposed system (Fig. 1a) uses
a stage with two input-output characteristics depending on the value of a digital pseudorandom noise signal
N[i] to modulate the output residue of the SUC and to estimate the calibration code by an adaptive averaging
process. The proposed method introduces no significant modifications in the analogue blocks of the Pipeline
ADCs making this technique a very promising alternative for the background calibration of the non-linearity
associated to the gain errors due to the capacitor mismatches and limited OPAMP gain. Simulation results
(Fig. 1b) have probed the stability of the algorithm and the tracking capability for fast gain error changes con-
sidering errors in both the sub-ADC of the SUC and the back-end stages.
c1n
ADC
MDACsub
SUCy2n
x
Fig. 1. a) Schematic of the two-residue stage ; b) Simulation results
++
subDAC
G1
R
-R
ADC2
Z2nr1
r2
RNG N[i]
rT r1 r2 1–+= bits
DIGITAL CORRECTION LOGICDIGITAL CALIBRATION LOGIC
Zcal
rT - bit ADC pipeline
c1n 2– r1 1– 2r1 1– 1–,[ ]∈
a)
0 5 10 15
0.85
0.9
0.95
1
normalised time0 5 10 15
4
6
8
10
12ENOB
Calibrated
Ideal
Theoretical
Calibrated
Calibration Code
gain error 20%=
gain error 18%=
b)
normalised time
H
- 17 -DCIS 2004
Mismatch Properties of MOS and Resistors Calibrated Ladder Structures
Rafael Serrano-Gotarredona, Teresa Serrano-Gotarredona, and Bernabé Linares-BarrancoInstituto de Microelectrónica de Sevilla (IMSE-CNM-CSIC), Ed. CICA, Av. Reina Mercedes
s/n, 41012 Sevilla, SPAIN. Phone: 34-95-5056666, Fax: 34-95-5056686, E-mail:[email protected].
THE mismatch behavior of MOS and resistor based calibrated ladder structures, used in arrays of DACs, is studied theoretically and experimentally. It is found that the calibrated DAC worst case
output current standard deviation is approximately 1/3 that of its individual components. MOSexperimental measurements illustrate the discussed mismatch behavior. Directions on how to designladder DACs for a target precision are provided.
- 18 -DCIS 2004
Session 1c
SiGe Designs Wednesday nov. 24 9h00 10h00, St Emilion Room
Chairs
Jean-Baptiste Bégueret (U. Bordeaux 1) Jean-Michel Fournier (E.N.S.E.R.Grenoble)
A multi-standard SiGe Power Amplifier for GSM900/DCS/PCS/WCDMA applications
Laurent Leyssenne, Jean-Marie Pham, Pierre Jarry, Eric Kerhervé, Daniel Saias*, Didier Belot*
IXL laboratory - UMR 5818 - ENSEIRB- University of Bordeaux - France
*ST Microelectronics - Crolles - France
Email-Addresses: [email protected] - [email protected]
N original multistandard integrated power amplifier is presented for communication systems. It consists
of two sub-amplifiers in parallel, each one being devoted to a specific frequency range, either the 900MHz
bandwidth (GSM), or the 1700/2000MHz bandwidth (DCS/PCS/WCDMA). These circuits are to be
implemented with a SiGe BiCMOS technology. Part of the challenge lies in the integration of both amplifiers on
the same chip and in the reconfigurability of the PCS/DCS/WCDMA amplifier in terms of linearity, power
added efficiency, and output power, via control bits. The architectures of both amplifiers are respectively
described in figure1. This work was done in the framework of a complete transceiver design. That’s why some
extra functionalities such as sleep-mode and bypass mode had to be integrated for convenience purpose with the
upstream frequency synthesizer.
The simulations were carried out in the Cadence environment, with the SpectreRF simulator, and their results
are as follows:
The GSM power amplifier is able to provide a 26.5 dBm output power and a 46% PAE with a 5dBm
input power.
The linear output power and the PAE of the “2GHz” power amplifier were respectively simulated to be
32dBm and 39% with a 5dBm input power in the UMTS mode. In DCS/PCS mode, the maximum output
power is 33dBm and the amplifier features a PAE above 30% for input power values down to –5dBm.
Bypass+
Out+Out-
Bypass-
In+
In-
Switch
Out+Out-
In+In-
DriverStage
Out+
Out-In+In-
Powerstage
Control Logic+PTAT Bias
UMTS mode
BypPass mode
50 ohm - Input50 ohm - Output
Chip
3.3V2.5V
DCS/PCS/WCDMA PA
Sleep mode
Fig 1: Block diagrams of both the single-ended GSM switching-mode amplifier, and the reconfigurable DCS/PCS/WCDMA power amplifier
A
Driverstage
Activestage
Load network Load
Input
Power
control
DCpowersupply
GSM PA
- 20 -DCIS 2004
A SiGe BiCMOS, Low Noise and Wide Band AmplifierWorking at 77 K
D. Prêle(1), G. Klisnick(1), G. Sou(1), M. Redon(1), A. Kreisler(2), C. Boulanger(3)
(1)Laboratoire des Instruments et Systèmes d’Ile de France UPMC-P6, 4 place Jussieu, 75252 Paris cedex 05, France
Email: [email protected]
(2)Laboratoire de Génie Electronique de Paris UMR 8507 CNRS, Supélec-P6/P11, 11 rue Joliot-Curie, 91192 Gif-sur-Yvette cedex, France
Email: [email protected]
(3)Centre National d’Etudes Spatiales 18 avenue Edouard Belin, 31055 Toulouse cedex, France
Email: [email protected]
Aspecific readout circuit operating at cryogenic temperature, has been investigated to process the
low-level signals delivered by high-Tc superconducting (YBaCuO Tc~85 K) hot electron
bolometers. An ASIC has been designed, including a low noise and wide band (quasi DC to 1 GHz)
amplifier, operating from room temperature down to 77K. This amplifier has been successfully tested
at liquid nitrogen temperature (Fig. 1). The vicinity between sensor and processing electronics allows
to reduce parasitic noises due to connecting leads and improves the compactness of the overall
detector. This experiment shows that it is possible to realise, with SiGe BiCMOS technology, an ASIC
designed for processing, in a cryogenic environment, the signal delivered by a YBaCuO hot electron
bolometer on a large frequency scale.
Figure 1. Cryogenic amplifier on the PCB and its cryostat.
- 21 -DCIS 2004
A SiGe Power Amplifier with Dynamic Bias for Efficient Power Control in UMTS/W-CDMA Applications
Nathalie Deltimple1, Eric Kerhervé1, Didier Belot2, Yann Deval1 and Pierre Jarry1
1 IXL Laboratory, CNRS UMR 5818, CNRS FR 2648, ENSEIRB – Bordeaux1 University, 351 cours de la Libération, 33405 Talence Cedex, France,
2 ST Microelectronics, 850 rue Jean Monnet, 38926 Crolles Cedex, France e-mail: [email protected]
OWER AMPLIFIERS (PAs) are the most power consuming components in portable equipment so
achieving high power added efficiency (PAE) PAs are deeply expected. Moreover, in WCDMA
systems, where a non-constante envelop modulation is used, the handset rarely transmits the signal at
maximum power, so it is important to reduce the power consumption at low transmitted powers.
An integrated two-stage power amplifier operating at 1.95 GHz frequency range is proposed. The
PA uses 2.5 V supply voltage and was designed using 0.25µm SiGe BiCMOS technology from ST
Microelectronics. The linear gain is 24 dB and the output 1 dB compression point (CP1) is 26.2 dBm.
The amplifier achieves a maximum PAE of 54%. In order to fulfill UMTS/W-CDMA requirements,
especially on linearity, the output power is 24 dBm in a linear Class-A operation, with a 32.4% PAE.
By acting on the two-stage bias circuits, the amplifier is able to shift dynamically the CP1 at constant
power gain according to input power level. Thanks to this, greater PAE is achieved at low input power
level. For instance, if the PA is backed-off by 6 dB and 11.4 dB from its CP1, PAE is equal to 15.6%
and 5.1% respectively. In order to enhance PAE, the driver stage bias circuit is used to shift the CP1 to
lower level, as shown in Figure 1, whereas the power stage bias circuit realizes the gain compensation,
then PAE reached is 27.1% and 13% respectively. The circuit designed with a SiGe quarter micron
technology from STMicroelectronics is still in progress, the layout is depicted in Figure 2. With a CP1
dynamic controlled, this PA paving the way to reconfigurable PA well suited to multi-mode multi-
band transceivers.
Pin (dBm)
P out
(dBm
)
15
Ibiasd=500µAIbiasd=100µAIbiasd=1.4mA
Pin (dBm)
P out
(dBm
)
15
Ibiasd=500µAIbiasd=100µAIbiasd=1.4mA
Figure 1: Output power as a function of the input
signal level for different driver stage bias conditions
Figure 2: Layout of the integrated UMTS/WCDMA
Power Amplifier
P
RF driver transistor
(20 unit cells)
460
µm
620 µm
RF output transistor
(100 unit cells)
- 22 -DCIS 2004
A 5 GHz SiGe VCO for WLAN Using Optimized Spiral Inductors
A. Goni-Iturri, S. L. Khemchandani, J. del Pino, A. Hernandez Institute for Applied Microelectronics of Las Palmas de Gran Canaria University, Spain.
HE low fabrication cost and high packing density makes Silicon the most suitable material to
choose in many RF IC applications. The devices requirements cannot be fulfilled in many cases
without the use of on-chip inductors. However, standard integrated inductors suffer from their poor
quality factor due to the low resistivity silicon substrate.
In this work, silicon integrated standard spiral inductors are studied and some guidelines to
optimized the performance are deduced. A high-quality factor inductor library on a 0.35 µm SiGe
technology at 5 GHz has been designed using electromagnetic simulations. The inductors, designed
with no changes in the process technology or post-processing techniques, reach values up to 10 nH.
As an application, a completely integrated LC voltage controlled oscillator (VCO) according to the
IEEE 802.11a WLAN standard has been designed. The achieved phase noise is -113 dBc/Hz at 1 MHz
offset, and the power consumption is 116 mW. The total VCO area, shown in Figure 1, is 0.424 mm2.
This work demonstrates the feasibility of a low cost silicon technology for the design of 5 GHz band
circuits.
Figure 1. VCO layout
T
- 23 -DCIS 2004
Session 1d
Built In Self TestWednesday nov. 24 9h00 10h00, Auditorium
Chairs
Michel Renovell (L.I.R.M.Montpellier) Joao Paulo Teixeira (INESC-Lisboa)
Robustness Improvement of a Ratiometric Built-In Current Sensor
Mikaël Cimino, Magali De Matos, Hervé Lapuyade, Jean-Baptiste Bégueret and Yann DevalLaboratoire IXL, Bordeaux, France, [email protected]
N this paper we present a new built-in current sensor (BICS) dedicated to monitor the current of
analog and mixed-signal building blocks. His principle is the same as the initial ratiometric BICS
presented by Yvan Maidon and al. in the ninety’s. This initial BICS was first designed to operate
under a 3.3 Volt power supply, as the CMOS technology used to implement the circuit was a 0.6 µm
one. His output range presented a good linearity but an important technology dispersion.
The new version of the BICS (figure1) uses a design methodology that allows to dramatically reduce
the dispersion (from 70% to 8.5% of the output range). We have first adapted the initial version of the
BICS to a 130 nm VLSI CMOS technology, and have substituted the classical current mirrors by low-
voltage bootstrap cascode ones. This design approach allows a 1.2 Volt power supply and reduces the
channel length modulation effect. At last, we added to the BICS degenerative resistors that prevent the
circuit from thermal burst and improve its robustness.
The new BICS here presented appears to be robust enough to be implemented in mass production
mixed-signal integrated circuits such as System on Chip (SOC) solutions, in which testability is of
major importance.
IDD10 100 Imeas
4/ 1
8/ 2
40/ 1
40/ 1
4/ 1
80/ 2
10/ 0.4
25/ 0.4
40/ 0.540/ 0.5
8/ 280/ 2
80/ 2
40/ 1
100 1005000500500
5/ 1 4/ 2
Imeas
IDD10 100 Imeas
4/ 1
8/ 2
40/ 1
40/ 1
4/ 1
80/ 2
10/ 0.4
25/ 0.4
40/ 0.540/ 0.5
8/ 280/ 2
80/ 2
40/ 1
100 1005000500500
5/ 1 4/ 2
Imeas
Figure 1. The robust built-in current sensor
I
- 25 -DCIS 2004
A Non-Intrusive Built-In Sensor for Transient Current Testing of Digital VLSI
Circuits
B.Alorda, V. Canals and J. Segura Univ. de les Illes Balears, Dept. Fisica, Cra. Valldemossa, km. 7.5, 07071 Palma de Mallorca,
SpainFax: +34 –971 173 426. Tel: +34 –971 172 506. e-mail: [email protected]
W E propose and evaluate a non-intrusive built-in monitor to facilitate the implementation of
transient current based testing of digital CMOS VLSI circuits. The monitor measures the
transient current idd(t) by sensing the voltage drop at an inductance coupled to the magnetic field
produced by the power supply transient current. Designed in 0.18 m CMOS technology, the sensor
proposed has two blocks. The transducer circuit senses the transient current and provides a voltage
waveform, while a second module amplifies the voltage waveform and computes the transient current
waveform Idd(t). Simulation results, using an elaborated CUT model, demonstrate the performance of
the new transceiver element.
- 26 -DCIS 2004
Built-In Current Sensor using Floating-Gate MOS Transistors for Low-Voltage Applications
A.A. Hatzopoulos(1), S. Siskos(2)
Aristotle University of Thessaloniki54124 Thessaloniki- GREECE
(1) Dept. of Electrical and Computer Eng., Electronics Lab., [email protected](currently Visiting Professor at the Katholieke Universiteit Leuven, Belgium)
(2) Dept. of Physics, Electronics Lab., [email protected]
In recent years Floating Gate MOS Transistors (FGT) have found many applications. In case wherethe input terminal is divided in two parts, the FGT can be used as a variable threshold transistor, when thefirst input is used as the signal input of the device and the second is used to control the threshold voltage.
Supply current testing, known as IDDQ testing in CMOS digital circuits, has been recognized for over25 years now as an advantageous method supplementary to the conventional logic testing. It can reveal defects that are missed by logic testers. Various designs have been proposed in the last decade, especiallyfor built-in current testing circuits. A major problem with all built-in current sensors (BICS) is their influence to the normal operation and performance of the CUT. The voltage drop across the currentsensing device is a considerable drawback of the BICS. In this work the application of floating gate transistors in the design of a BICS is proposed. The important benefit from this application is that the voltage drop across the sensing device can be reduced to almost zero value, while preserving adequatelinearity for the current monitoring. This linearity makes the proposed BICS also appropriate for analog and mixed-signal circuit testing.
The proposed BICS structure is given in fig. 1. For two input gates it is shown that for the FGTs it
is: 222112 TGSGSD VVwVwKI . With proper selection of the coupling ratio w2 and using a
corresponding bias voltage VGS2, we can have: 022 TGS VVw , which results in: 2112 GSD VwKI . Since
the value of VDS1 is kept quite low for the range of the ID current under consideration, the voltagedegradation of the CUT supply will be minimal, making this structure suitable for low voltage built-in current sensing applications. The mirrored current in FGT2 can be converted to a voltage by the use of a loading transistor. This voltage output, followed by an appropriate buffer or a latch, may be directly used as a fault indicating flag.The mirrored current can be downscaled for power saving by scaling the sizes of the floating-gatetransistors. The proposed FGT-BICS structure of fig. 1 has been simulated, utilizing various circuits as aCUT. The relation between Vout of the BICS and the supply current of a simple opamp circuit in a voltage inverting configuration with a 10k load as a CUT is plotted in fig. 2, showing very goodlinearity.
VDDVDDLoad
CUT Out
FGT2FGT1 VbiasVbias
(or VSS)
Fig. 1. The proposed FGT-BICS structure with 2-inputfloating-gate transistors.
Fig. 2. BICS voltage output versus supply current of aninverting opamp configuration as the a CUT.
- 27 -DCIS 2004
Experimental Evaluation of a Built-in Current Sensor for Analog Circuits
R. Mozuelos, Y. Lechuga, M.A. Allende, M. Martínez, and S. Bracho Microelectronics Engineering Group, University of Cantabria
Avda. Los Castros s/n, 39005, Santander, Spain roman, yolanda, allende, martinez, [email protected]
T HIS paper presents the experimental characterization of a built-in current sensor (BICS) for
analog circuits. The BICS gives greater specific weight to the higher frequency components of the
current waveform. Thus, an inductive rather than a resistive load has been used to carry out the
conversion of the sampled current to voltage. The circuit has been fabricated with the Austria Micro
System (AMS) 0.6 micron technology.
The test approach relies on obtaining a copy of the supply current by means of the integration of
additional transistors within the current mirrors of the CUT. In this way, the sensor overcomes the
drawback of impacting the effective voltage supply seen by the CUT and consequently degrading the
circuit performance if the sensor were placed in series with the supply/ground node.
The proposed BICS gives an output that reflects the dynamic power supply consumption of the
CUT. This signal has been digitized by a simple window comparator made of logical gates. The key
parameter is the width of the pulse at the sensor output. Thus, a low cost counter or an integrator can
easily do the signal post-processing and the result will be compared with either the one obtained from
simulation or the one obtained from a golden circuit.
Finally, the sensor has been coupled to a transconductance amplifier in order to experimentally
validate the structural test approach. Together with the fault free circuit, three parametric faults were
implemented. The discrimination between them can be easily done by means of the measured value of
the pulse width at the BICS output.
Figure 1. BICS die photograph
- 28 -DCIS 2004
Session 2a
High Level Modeling Wednesday nov. 24 10h30 11h45, Lacanau Room
Chairs
Hervé Lévi (U. Bordeaux 1) Francisco Moya (U. de Castilla-La Mancha)
Compact Modeling of a Magnetic Tunnel Junction Using VHDL-AMS
Jean-Baptiste Kammerer, Luc Hébrard, Michel Hehn, Francis Braun, Patrick Alnot and Alain Schuhl
C URRENTLY, the lack of compact magnetic tunnel junction (MTJ) model is a truly limiting factor
for the design of spintronics circuits. In this paper, we present a compact MTJ model written in
VHDL-AMS. This behavioral model is based on the Stoner-Wohlfarth model and takes most of the
important phenomena such as magnetic coupling, capacitance, and magnetizations dependent
conductance into account. The method employed to model a two layers magnetic tunnel junction is
detailed. Applications of this model such as the simulation of the operation of a MRAM cell and of a
magnetometer are also presented.
- 30 -DCIS 2004
Analogue-Synthesis Tool Developmentfor Switched-Current Systems using VHDL-AMS
Nesrine KSENTINI1,3, Ahmed FAKHFAKH1, Mourad LOULOU1, Nouri MASMOUDI1
Yannick HERVE2, Jean-Jacques CHARLOT3
1 LETI-ENIS, Sfax, Tunisia, [email protected] CNRS-PHASE, Strasbourg, France
3 ENST-COMELEC, Paris, France
T HIS contribution presents a methodology, based on VHDL-AMS modeling, for synthesis and
optimization of systems designed with the switchedcurrent technique (SI). This methodology has
been implemented in Simplorer Software environment and allows a reduction of simulation runtime
and a characterization of SI systems at a high level of the hierarchical design methodology.
- 31 -DCIS 2004
Modeling and simulation of phototransistors using VHDL-AMS
A. Alexandre, A. Pinna, B. Granado, P. Garda Laboratoire des Instruments et Systèmes d’Ile de France, Case 252
Université Pierre et Marie Curie, 4 place Jussieu, F-75252 Paris Cedex 05e-mail : [email protected] , [email protected], [email protected], [email protected]
T O develop Systems On Chip for imaging, models of photodetectors for APS cells are needed.
The mainly used photodetectors are photodiodes but it is possible to realize APS cells with
phototransistors. Next complete models of vertical and lateral phototransistors are presented. They are
based on a physical approach which leads to an electrical model. These models were implemented by
using VHDL-AMS language. The simulations of these structures give the spectral response of these
components and are in good agreement with the usual results.
- 32 -DCIS 2004
Final User oriented SoC modeling Sébastien SNAIDERO, Yannick HERVE
[email protected], [email protected] ERM-PHASE, ENSPS, Parc d’Innovation, BP 10413, F-67400 ILLKIRCH
VVHDL-AMS and other Hardware Description Languages (HDLs) are no longer used only by
modeling specialists. There are thus people who are simple users of models designed by others
and do not know a word about languages. The new simulation software meets the demand with their
new graphical user interface (GUI). They allow the development of simple and fast models easy to use
for industries as well as in depth models for physical studies and laboratories.
To meet the demand of both this new type of users and the developers, the new software needs
some qualities that are hard to gather. For the user side: intuition, simplicity and convenience must
hide the difficulties inherent in the complexity and varieties of languages without impoverish the
power of the HDLs. For developers, a powerful model creation tool that brings a substantial
implementation of the various languages it accepts is required. It must of course allow interfacing
models of these different languages. As developers remain the only ones able to create models from
scratch, they must watch over using datasheets parameters for their models to guarantee their widest
possible use.
The developers on the one side and the users on the other side both require tools adapted to their
needs. The current effort of simulation software corporations is to build extremely powerful and
convenient tools that work for both users and developers.
As the abstraction level of our designs is increasing up to describe whole systems, it is important
to notice that the tools have to and will still have to advance the same way to provide user-friendly
graphical representations of the objects we wish to design.
A complete channel of the module
- 33 -DCIS 2004
Design and Simulation of Mixed-Mode Optical Systems for PSD Applications
Ricardo Doldán, Eduardo Peralías, Alberto Yúfera and Adoración Rueda Instituto de Microelectrónica de Sevilla (IMSE), Centro Nacional de Microelectrónica
(CNM), Av. Reina Mercedes s/n. Edificio CICA. 41012. Sevilla. SPAIN. emails: rdoldan, peralias, yufera, [email protected]
T HIS work describes an optical Position Sensing Detection (PSD) algorithm. A mixed-signal
model of a photodetector cell for electrical simulation has been developed, including the complete
dynamic model for a photodiode. It is shown how standard simulators employed in electrical
environments can be adapted to describe devices included in optical based system. This enables it to
perform complex system characterizations including optical and electrical parts using the same
environment (Spectre), and to extend the mixed-mode simulation concept to a wider field than non-
electrical systems. A system simulation application for Position Sensing Detection (PSD) with a
resolution in the micrometer range is reported along the paper.
- 34 -DCIS 2004
Session 2b
Biometric and Robotic Applications Wednesday nov. 24 10h30 11h45, St Emilion Room
Chairs
Rachid Bouchakour (E.P.U. de Marseille) Jacques Pistre (E.N.S.E.I.R.Bordeaux)
DSP-based Fuzzy Controllers: Application to Parking an Autonomous Robot
Iluminada Baturone1, Francisco J. Moreno-Velo1, Santiago Sánchez-Solano1, Víctor Blanco2,and Joaquín Ferruz2
1 Instituto de Microelectrónica de Sevilla, Centro Nacional de Microelectrónica (IMSE-CNM), Sevilla, SPAIN, email: lumi, velo, [email protected]
2 Departamento de Ingeniería de Sistemas y Automática, Universidad de Sevilla, Sevilla, SPAIN, email: vmblanco, [email protected]
UZZY controllers are used in many applications because of their rapid design by translating
heuristic knowledge, robustness against perturbations, and smoothness in the control action.
However, their direct implementation requires parallel processing and special operators (such as
fuzzification or defuzzification) which are not available at standard digital signal processors (DSPs).
The novel idea followed in this paper is to translate the fuzzy rule bases of a fuzzy controller into non
fuzzy ones that can be implemented easily by using the relational and logical operators, the standard
if-then conditional statements, and the addition and multiplication operators available at a DSP. This is
done by using hierarchical structures and adequate membership functions, connective operators, and
inference methods. The parking problem of an autonomous robot (Figure 1) is described to illustrate
this design process. Experimental results (Figure 2) show the efficiency of the designed fuzzy
controller embedded into a stand-alone card based on a fixed-point DSP from Texas Instruments.
Figure 1. The autonomous robot Romeo 4R. Figure 2. Example of experimental results.
F
x (m)
y (m) initial configuration:x= -2.5 my= 3.0 m
= -117º
x (m)
y (m) initial configuration:x= -2.5 my= 3.0 m
= -117º
- 36 -DCIS 2004
Coprocessor of the Ridge Line Following Fingerprint Algorithm
E. Cantó, N. Canyellas, M. López, M. Fons, F. Fons
Universitat Rovira i Virgili (URV),Department of Electronic, Electrical and Automatic Control Engineering Av. Paisos Catalans 26, 43007 Tarragona(phone: 977558522; fax: 977559605; e-mail: [email protected]).
Fingerprint-based automatic recognition systems are rapidly growing on a wide shell of applications. Most of biometrics authentication systems are implemented on high performance computer based
platforms executing a set of complex algorithms implemented on software. Those solutions cannot be applied to low-cost embedded systems, based in microprocessors without floating-point arithmetic unit. The use of fingerprint biometrics coprocessors is still a young field. A great majority ofcommercial fingerprint OEM modules are based on embedded high performance 32-bit processors or DSPsIn this article we present a biometrics coprocessor to speed up the ride line following minutiaeextraction algorithm. It covers the minutiae extraction stage, the one with higher computationalrequirements. In our work we use the Maio-Maltoni ridge line following algorithm1 because it permitsminutiae extraction directly from the gray-scale fingerprint image, it is computationally less expensivethan others, and it can be rewritten to be implemented without floating point operations. In order to develop an efficient hardware implementation of the coprocessor, in terms of low-cost and high-speed,floating-point computations used on the algorithm have been substituted by fixed-point computations,among other substituted complex functions. It has also been adopted a pipelined scheme to reduce the critical path-delay and to execute several steps in parallel, to increase the clock frequency and throughput.The execution of the steps performed by the coprocessor running at 50MHz, is 14.4 s, while the timedevoted by an ARM7TDMI processor at the same clock speed to execute the same computation tasks, was 211 s averaged. The overall execution time of the algorithm running in the ARM with the coprocessor is reduced from 700ms to 215ms, that is a reduction of about 70%.
1Maio, D.; Maltoni, D. “Direct Gray-Scale Minutiae Detection In Fingerprints” IEEE Transactions on Pattern Analysis and Machine Intelligence, vol 19, No. 1. January 1997
- 37 -DCIS 2004
Iris Biometrics Verifiers for Low Cost Identification Tokens
Judith Liu-Jimenez a, Raul Sanchez-Reillo a, Carmen Sanchez-Avila b, Luis Entrena aa Dpto de Tecnologia Electronica, Grupo de Microelectronica, Universidad Carlos III de
Madrid, c/ Butarque 15, 28911 Leganes (Madrid), Spain, Email: jliu, rsreillo, [email protected]
b Dpto. de Matematica Aplicada a las Tecnologias de la Informacion, E.T.S.I. Telecomunicacion, Universidad Politecnica de Madrid, Ciudad Universitaria, 28040 Madrid,
Spain, Email: [email protected]
T HE use of biometrics is increasing everyday, as security is becoming one of the most important
concerns in Information society. In Biometrics, one of the most promising techniques is iris
recognition, which presents lower error rates than other biometrics techniques. One of the processes in
a biometric system is the verification or matching between data obtained and a template previously
stored. This process can be done on centralized systems, such as a central database, or in a distributed
way, using identification tokens. When developing new identification tokens, computational cost and
processing time should be reduced, to provide cheaper devices, which could allow a viable solution in
a commercial system. The authors, in this paper, develop different implementations of low cost
biometric verifiers, to be included in identification tokens. The biometric technique chosen for that
issue has been iris recognition, and therefore, the verification technique has been based in Hamming
Distance.
- 38 -DCIS 2004
Fingerprint Matching Acceleration in Smart Cards1
Luis Entrena, Raúl Sánchez-Reillo, Almudena Lindoso, Judith Liu Departamento de Tecnología Electrónica
Universidad Carlos III de Madrid entrena, rsreillo, alindoso, [email protected]
T HE increasing demand for pervasive security poses a challenge in achieving robust
authentication at very low cost. Biometrics is the only way to perform a real user authentication,
but has a high computational cost. In this work we study the integration of fingerprint biometrics in
smart cards. Since commercial smart cards use low performance microprocessors, fingerprint
verification may take up to several seconds, which is unacceptable for practical applications. On the
other hand, simply moving to a more powerful processor will result in an important cost penalty and
will not solve the performance bottleneck.
In order to speed-up fingerprint verification, we identify the most critical operations of an efficient
fingerprint matching algorithm. These include the computation of euclidean distance, radial angle and
element matching. Then, we propose an extended instruction set that can be implemented with low
hardware overhead. With this extension, the total time required to complete fingerprint matching is
reduced more than 60% and accounts to less than a second for medium size minutia sets. An improved
instruction for element matching, termed vector matching, is also proposed that provides larger speed-
up. Final results using vector matching allow to perform fingerprint verification in a fraction of a
second for large minutiae sets.
1 This work was supported in part by the Ministerio de Ciencia y Tecnología (Spain) under Project TIC2003-01793
- 39 -DCIS 2004
Hardware implementation of the Bresenham line generation algorithm applied to µ-robot movement
R. Casanova, A. Diéguez, J. Lacort J. SamitierDepartament d’Electrònica, SIC, Universitat de Barcelona, C/Martí Franquès 1, E-08028,
Barcelona, Spain.Email: [email protected],
A digital waveform generator for the movement control of an autonomous microrobot is
presented. The circuit is able to generate trapezoidal and sawtooth signals with programmable
amplitude, period and phase. These control waveforms are used to actuate over a bimorphic
locomotion unit. As the robot has to be capable to operate with nanometric resolution, waveforms
must be generated with great precision. Waveforms are generated by using the Bresenham algorithm
in order to deal with integer operations. The circuit has been designed with the 0.35µm C35b4
technology of AMS.
- 40 -DCIS 2004
Session 2c
Industrial Applications Wednesday nov. 24 10h30 11h45, Auditorium
Chairs
Franck Badets (STMicroelectronics)Daniel Auvergne (L.I.R.M.Montpel
Dual-port serial arbiter with GSM modules for simultaneous local/remote
control of RS232-based devices
Eloi Ramon1, and Lluís Ribas2, Member, IEEE1Electronics, and 2Computer Science Departments,
Universitat Autònoma de Barcelona (UAB), Cerdanyola, Spain, Eloi.Ramon, [email protected]
S ERIAL communications are extensively used within the electronics industry due to its relative
simplicity and low hardware overhead. One of the most popular serial communications standard in
use is certainly the RS232. There are many devices that can be controlled by RS232 ports at home and,
especially, in industrial applications. Remote control and/or monitoring of such devices enables users
to access these devices from anywhere. Particularly, providers of such devices or related services can
remotely monitor their functionality and yield, and take actions accordingly.
SMS (Short Message Service) was introduced as a GSM (Global System for Mobile
Communications) service in 1992. In the last years, SMS have been widely used in remote control due
to its ubiquity, area coverage, cost and ease of use. Despite the increase in GSM/GPRS modules in the
market, most industrial applications are still using SMS as a communication protocol.
In this paper we present a module to both extend the serial communications port for other devices
or protocol adapters and to allow remote control by SMS messaging.
In order not to interfere with local applications, it is interesting that RS232 device ports are kept
connected to local application hosts. Consequently, there should be a module that, apart from having 2
serial ports, is capable of transmitting information wirelessly. Unfortunately, the RS232 serial protocol
is a point-to-point communication one, thus the introduction of a third-party port necessarily requires
the participation of an arbiter that resolves the conflicts and sets the appropriate connections.
The arbiter has been implemented on a SonyEricsson GR47 module, as well as its application-
specific queue reading/writing functions.
The application presented has been used in an industrial appliance designed for MAM Electrónica
to remotely control UPS systems in order to monitor charger status, battery status and condition, utility
status, et cetera.
- 42 -DCIS 2004
Signal Processing Unit for River Tugboat Telemetry System
Mauricio Pardo1, Víctor Manotas1, Juan Carlos Niebles2, Javier Páez1, David Angulo2,Humberto Campanella1, 3
1 Universidad del Norte, Barranquilla, Colombia, e-mail: [email protected] Flota Fluvial Carbonera Ltda, Barranquilla, Colombia
3 Centro Nacional de Microelectrónica (IMB-CNM), Barcelona, Spain [email protected]
HIS article describes the design and implementation of a complete signal acquisition system
intended for use in telemetry systems for river tugboats. Specifically, the design aspects of the
signal processing unit are covered. A complete prototype was built, installed and put into operation in
a tugboat of the fleet of a fluvial transportation company in Northern Colombia (Flota Fluvial
Carbonera Ltda). System functionality was mainly synthesized on an Altera’s FPGA, taking great
advantage on rapid prototyping. Field measurements are reported, making detail in the fuel volume
calculation, which demonstrated to be more accurate than previous method.
As an industrial application, this system represents an innovation for the company’s operation.
Specifically, a method to estimate fuel level and volume is proposed. This method does not require
installation of neither 3D angular sensors nor complex calculation to compensate measurements due to
swell or navigation conditions. A single sensor was installed in each one of the fuel tanks achieving
less difference with real dumped fuel than previous method.
The telemetry system is composed by hardware and software components. The hardware component
is an acquisition and processing unit. The second component is a network management application,
totally tailored in C++ for this telemetry application and conformed by three main software modules.
Processed data is transmitted to the company’s network operation center using a satellite link.
Figure 1. Stand-alone enclosure with the acquisition and signal processing hardware
T
Sensor connection (from machine room)
SignalProcessing
Serialcommunication port
- 43 -DCIS 2004
A Sensorless Electronically Controlled Horn for Automobiles
M. César Rodríguez, César Sanz Universidad Politécnica de Madrid, Spain (e mail: [email protected])
Jacinto M. Acero, Fernando Nozal Robert Bosch España S.A.
T HIS paper describes a new kind of sensorless electronically controlled horn for automobiles.
The main benefits of this new horn over the classical electromechanical counterparts are: a much
longer lifetime, a lower level of generated electromagnetic interference, a better behavior against aging
and stress, the avoidance of the adjusting operation during manufacturing and its multifunction
capabilities. To fulfil all these targets we have substituted the electromechanical breaker of previous
designs by a solid state switch controlled by a microprocessor on board the horn. In order to detect the
resonance condition of the horn we use a novel technique based on the analysis of the current across
the coil (as seen in the figure), avoiding the need for any sound-level, position or motion sensor.
The lifetime of these new horns is, at least, forty times that of previous breaker horns. The level of
generated EMI for these new horns is much lower than previous electromechanical designs. These
new horns can fulfil the requirements of both 95/54 EC directive and CISPR 25 standard, whereas the
breaker horns cannot.
As the horn is now self-adjusting, its behavior against aging and stress is better, as it can
compensate for these factors dynamically. This self-adjusting operation also allowed the avoidance of
the expensive trimming operation during fabrication.
We have also developed a multifunction horn capable of producing different kinds of sound at the
command of the vehicle’s electronic control unit. Both kinds of horns were assembled using the very
same machinery as previous breaker horns and serve as perfect spare parts for them.
ADC
DigitalController
PulseGenerator
RSense
ControlSwitch
PowerSwitch
MagneticAssembly
Figure 1. Proposed sensorless control circuitry
- 44 -DCIS 2004
Design of Low-Power CMOS Read-Out ICs for LargeArrays Cryogenic Infra-Red Sensors
B. Misischi1, F. Serra-Graells1, E. Casanueva2, C. Méndez2 and L. Terés1
[email protected], [email protected], [email protected],[email protected] [email protected]
1 Institut de Microelectrònica de Barcelona, CNM, CSIC (Spain)2 Indra Sistemas S.A. (Spain)
THIS paper describes a complete design methodology for a low power cryogenic design read-out
integrated circuit (ROIC) of large arrays of infra-red (IR) detectors. The presented methodology
includes IR sensor modeling, MOSFET modeling at cryogenic temperature, circuit design, physical
verification strategies and the system-on-chip realization. Also, novel low-power and compact CMOS
circuits are proposed to implement all the required basic building blocks, from the active pixel sensor
(APS) to the composition of the output video signal. The resulting high performance 500×12 array
and 60ns/pixel system-on-chip, capable of capturing high-resolution and real-time infra-red images, like
640×500@100fps, has been designed for a standard 0.35µm CMOS technology from AMS.
Vpixel
+
-
Cint/CDS
reset
reset
reset
Iqwip
Ctest
reset
test
column-bus
select
+
-
Vrefo
Vrefi
Vcol
CB
init+reset
CA
APS
init
Figure 1: Simplified schematic (left) and layout including bumping pads (right) of the APS cell, and landscapeview of the complete ROIC layout including bonding pads (bottom).
- 45 -DCIS 2004
A Dynamic Current Mode Logic to Counteract Power Analysis Attacks
F. Mace, F.-X. Standaert, I. Hassoune, J.-D. Legat, J.-J. QuisquaterUCL Crypto Group, Microelectronics Laboratory (DICE),
Université Catholique de Louvain (UCL), Belgium mace,standaert,hassoune,legat,[email protected]
S INCE their publication in 1998, power analysis attacks have attracted significant attention
within the cryptographic community. So far, they have been successfully applied to different kinds
of implementations (eg: smart cards, ASICs, FPGAs) of cryptographic algorithms. To protect such
devices against power analysis attacks, it has been proposed to use a dynamic and differential logic
style for which the power consumption does not depend on the data handled. In this paper, we suggest
to use the Dynamic Current Mode Logic to counteract power analysis. The resulting circuits exhibit
similar resistance to the previously published proposals but significantly reduce the power delay
product. We also demonstrate that certain criteria previously used to evaluate the resistance against
power analysis have no cryptographic relevance.
- 46 -DCIS 2004
Session 2d
Data Converter Test Wednesday nov. 24 10h30 11h45, Bordeaux Room
Chairs
François Marc (U. Bordeaux 1) Joan Figueras (U. Politècnica de Catalunya)
Digital Diagnosis of Settling Error in Modulators
Gildas Leger, Adoración Rueda Instituto de Microelectrónica de Sevilla (IMSE-CNM), Universidad de Sevilla
Edificio CICA, c/ Tarifa s/n, 41012-SEVILLA, SPAIN. email: [email protected], tlf: 34 95 505 66 66
F IRST and second order sigma-delta modulators are commonly used in cascaded structures to
achieve high resolution analog-to-digital converters. While these modulators are gaining more and
more resolution, they become harder to test. Embedded test solutions and Built-In Self-Test (BIST)
techniques are faced to important issues to ensure the test stimulus precision or the test data
acquisition. This makes functional test a tricky path to follow.
For almost all systems, and in particular for modulators, a behavioural model where the principal
no-idealities are quantified is usually used to settle high-level design specifications and realize high-
level simulations. Designers know that to reach a given precision they have to guarantee a number of
parameters like amplifier DC gain, amplifier slew-rate and bandwidth, capacitor switching noise level,
integrator output range, etc. Hence, from a test viewpoint it can be assumed that the principal causes of
unexpected performance decrease should be related to these high-level design specifications. In other
words, if a modulator is not performing as expected it is likely that some high-level parameter have
been brought out of specification. It is thus of utmost interest to diagnose and measure these
parametric faults.
This paper presents a simple and fully digital test technique able to evaluate amplifiers settling error in
second and first order modulators. These settling errors are related to amplifier gain-bandwidth
product and slew-rate, which are part of the above mentioned set of high level design specifications.
Actually, they are known to be a source of non-linearity. The realistic simulations presented in this
paper exhibit good matching with theory and show very promising results as the integrator settling
error can be determined with good precision. It is also shown that the settling error can directly be
related to a precision loss, which enables a functional interpretation of the test signature.
- 48 -DCIS 2004
Digital Sigma Delta Oscillator : DesignConsiderations
Maher Jridi∗, Chiheb Rebai†, Dominique Dallet∗, Sylvain Engels‡, Laurent Dugoujon§∗Laboratoire IXL UMR CNRS 5818 ENSEIRB Univerist´e Bordeaux 1
351 cours de la lib´eration, 33405 Talence CEDEX FranceTelephone: +33 5 4000 6540, Email:[jridi,dallet]@ixl.fr†Institut Superieur des arts du multim´edia de Manouba
Campus universitaire Manouba, 2010 TunisieTelephone: +216 71 602 050, Email : [email protected]
‡STMicroelectronics - 850, rue Jean Monnet, F-38926 Crolles CedexTelephone: +33 4 76 92 50 26, Email : [email protected]
§STMicroelectronics - 12, rue Jules Horowitz -BP217, 38019 GrenobleTelephone: +33 4 76 58 62 54, Email : [email protected]
This work is deployed in the context of Analog to Digital test Converters where a precise sinusoidalsources is needed. A promising solution to this challenge, a digital Lossless Discrete Integrator (LDI)resonator combined with a 1-bit Delta Sigma modulator and simulation results has been yet presented. Thispresent article proposes a new methodology enablingus to go towards ASIC using the same parametersof FPGA implementation. As summarized later, the digital source is build using digital hardware: digitalregisters, adders, multiplexers and shifters. The work presented will be divided into three parts: this paperfirst reviews the fundamentals of Delta-Sigma modulator based signal generation (Section II): first ofall we talk about the digital resonator and its drawbacks. Then we detail the principle of Sigma Deltaattenuator where we emulate the multiplier operator by Sigma Delta modulator and a multiplexer. Thisoscillator has to generate a precise 1-bit signal to test the ADC. To perform the SNR, we need to increasethe modulator order. A problem of stability can appears if the modulator quantifier is on only one bit.Schreier seems found an empirical methodology to resolve this. SectionIII briefly outlines the materialFPGA implementation. The simulation results will be presented to validate our preferences to the selectedLDI structure and the simulation parameters. The difficulties of this part consist on the data flow wherewe used a fixed-point precision. Section IV describesthe circuit. The layout schema will be shown in theend of this paper.
- 49 -DCIS 2004
Optimal implementation of linear and adaptive filter bank for ADC characterization
1 F. Missaoui, 2 D. Dallet, 1 C. Rebai, 1 A. Ghazel
1 – MEDIATRON Laboratory – Ecole Supérieure de Communication (SUP’COM) 2088 Cité Technologique des Communication, Tunis, TUNISIE
2 – Laboratoire IXL – UMR CNRS 5818 – ENSEIRB – Univeristé Bordeaux 1, 351 cours de la libération, 33405 Talence CEDEX – France, tel : 33 5 4000 2632, [email protected]
N this paper we present a linear and an adaptive filter bank employed to decompose the
signal in its main spectral components in the field of ADC characterization. Both structures
are based on band pass LDI filter (Lossless Digital Integrator) which is known for its efficient
implementation. From real acquisitions, we show the efficiency of linear structure to obtain an
estimation of the main spectral parameters. Nevertheless, if the input frequency is not the
expected one, we have to use an adaptive structure to track the fundamental component. In
this way, we can estimate the spectral parameters related to the ADC performances. These
two architectures were simulated in floating point precision. This paper shows the design
consideration to take into account for the implantation in fixed point precision: the data flow
for the linear structure and the latency problem for the adaptive one.
u(n)
Hu (q)
H (q,n)
e(n)
v(n)
y'(n)
y(n)1z−
k1
1z−
k2
k1
-
1/2XinXout
Figure. Adaptive structure for ADC spectral parameters estimation.
I
- 50 -DCIS 2004
Selection of test techniques for high-resolution modulators
Oscar Guerra*, Sara Escalera*, Jose M. de la Rosa*, Eric Compaigne+, Christophe Galliard+ and Angel Rodríguez-Vázquez*
(guerra, escalera, jrosa, [email protected]; eco, cga@dolphin,fr)*Instituto de Microelectrónica de Sevilla (IMSE-CNM-CSIC)
Av. Reina Mercedes, 41012-Sevilla, SPAIN. Tlfno.: 955056666 Fax:955056686+Dolphin Integration B.P. 65 ZIRST F38242 Meylan, France
The objective of this paper is to select the best test techniques for their application to high-resolution modulators.As the selection process is quite complex, a decision matrix has been created in order evaluate the efficiency of thedifferent techniques in term of their cost. This cost could then be used as a reference to determine the efficiency of thedifferent test techniques compared to the standard methodologies.The decision matrix presented in the chart below is a tool that has been created for the TAMES-2 project in order toevaluate the efficiency of the different test techniques in term of cost. For each test technique, it is required to fill in the decision matrix with all the parameter values. Then, a cost is auto-matically computed. The resulting "Weighted cost" allows to determine if the technique is cost effective or not.For the case of the sensor interface under study1, the key point of test is to cover both static and dynamic measure-ments. Concerning dynamic requirements, the following measurements have to be performed:
• Signal over Noise Ratio: the estimated test time for this FFT is 0.3 s. • Total Harmonic Distortion: the estimated test time for this FFT ranges is 0.6 s.• Efficient number of bits: is obtained automaticaly from the FFT that is used to test the SNR or the THD. Thus, no
additional time (unless some negligible computational time) has to be added to the total test time.Then, for the case of static measurements, the following test measurements have to be performed:
• Integral non linearity, differential non linearity: the computational time for this test is 65s using the standard histo-gram test.
• Static gain and and offset can be calculated from the data achieved for the INL with an almost negligible post-processing time. Thus, no additional time is devoted to this task.
After a detailed study of all the potential test techniques capable of dealing with high-resolution converters, the cho-sen test techniques for the Sensor interface have been the following:
FFT-INL. In order to test the static behavior of the A/D converter, the FFT coefficients resulting after the applicationof the FFT test can be used to generate the coefficients of a polynomial that is the "smoothed" version of the DNL.
Hierarchical-based. This technique has been fully explained in other paper2. The basic idea here is to test theimpact that the extra test circuitry and some specific reconfiguration scheemes have on the circuit performance.
Wavelet-based. This technique is based on the application of the wavelet transform to the output of the converterwhen it is excited using a sine wave. Information about the instantaneous ENOB, SNR and INL can be obtained usingless samples than those required for the standard Histogram technique.
In the paper, these techniques will be compared to two reference test methods, the standard FFT to test the dynamiccharacteristics of the converter and the standard histogram test to measure its static characteristics.
1.J. M. García-González, S. Escalera, J. M. de la Rosa, F. Medeiro, O. Guerra, B. Pérez-Verdú and A. Rodríguez-Vázquez,“Design and Implementation of a 0.35 m CMOS Programmable-gain 2-1 Cascade Modulator for Automotive Sensors”, Pro-ceedings XIX Design of Circuits and Integrated Circuits Conference, pp. 114-119, 2003.
2.O. Guerra, J. Ruiz, J. M. de la Rosa, F. Medeiro and A. Rodríguez-Vázquez, “A decomposition methodology to test high-reso-lution modulators” Proc. 9th International Mixed-Signal Testing Workshop, pp. 65-70, 2003
- 51 -DCIS 2004
Guidelines for the Design of a Sine-Wave Analyzer for BIST Applications
Manuel J. Barragán, Diego Vázquez and Adoración Rueda Instituto de Microelectrónica de Sevilla – Centro Nacional de Microelectrónica (IMSE-CNM)
Universidad de Sevilla Avda Reina Mercedes s/n, Edif. CICA/CNM, 41012 Sevilla (SPAIN)
e-mail: manuelj, dgarcia, [email protected]
HIS paper presents some guidelines for the design of an on-chip analyzer for extracting, in the
digital domain, the main characteristic parameters of an analog sine-wave signal. The analyzer,
reported elsewhere1, is based on a double-modulation, square-wave and sigma-delta, altogether with a
simple digital processing algorithm. We discuss the specifications required for the analog part of the
analyzer and describe an area-efficient implementation of the digital part. In addition, we show
simulations results which demonstrate the validity of the proposed guidelines, while the simplicity and
the robustness of the circuitry make it very suitable for BIST applications.
1D. Vazquez, G. Huertas, G. Leger, A. Rueda, and J. L. Huertas, “A Method for Parameter Extraction of Analog Sine-Wave Signals for Mixed-Signal Built-In-Self-Test Applications”, IEEE Design and Test in Europe (DATE04), Paris, France, Feb. 2004.
T
- 52 -DCIS 2004
Session 3a
Digital Signal Processing in FPGA Platforms
Wednesday nov. 24 14h15 15h45, Lacanau Room
Chairs
Lionel Torres (L.I.R.M.Montpellier) Lluis Teres (IMB-CNM)
Implementing the FFT Algorithm on FPGA Platforms: A Comparative Study of Parallel Architectures*
M. A. Sánchez1, M. Garrido1, M. López-Vallejo1 and J. Grajal2
1Dept. Ingeniería Electrónica, 2 Dept. SSR, ETSIT, UPM, Madrid, Spain,1masanchez, mariog, [email protected] [email protected]
N this paper we present an in depth analysis of the implementation of different FFT architectures
in FPGA platforms. The target applications are radar processing systems and wideband digital
receivers, what enforces hard constraints in processing speed. Thus, parallel pipelined architectures of
the FFT have to be used. In particular, feedback and feedforward architectures are analized in detail,
studying the variations of results with a set of key design parameters: radix, word length, number of
points or the effect of truncation. Additionally, the impact due to the implementation in programmable
devices will be considered when designing and analyzing the different architectures.
I
Two alternative structures have been studied: feedback and feedforward architectures. They provide
very diverse results in terms of area and performance, what results in different applications of the
proposed architectures. In this way, feedback structures can be used for long N-point FFTs, because of
their small area, while feedforward architectures are better suited for applications with hard real-time
constraints due to their better speed.
Figure 1 depicts the results of area and speed obtained for different implementations of feedback
(FB) and feedforward (FF) implementations.
0
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FF SPEED (MSsec /10)FB SPEED (MSsec /10)BRAM (1/1000)SLICES
Figure 1. Area and perfomance results of feedback (FB) and feedforward (FF) architectures for 16, 64, 256 and
1024 points
* This work was supported by the Spanish Ministry of Science and Technology under contract TIC2003-07036.
- 54 -DCIS 2004
An FPGA landmine detection system based on infrared images
F.Pardo*, M. Balsi**, P. López*** and D. Cabello* *Dept. Electrónica y Computación. Univ. de Santiago de Compostela, Spain.
**Dipartimento di Ingegneria Elettronica. Università La Sapienza. Italy.***Fraunhofer Instituf für Integrierte Schaltungen. Erlangen. [email protected] [email protected] [email protected] [email protected]
H umanitarian deminining has become an important issue in regions where an army conflict has
occurred. The detection of small plastic mines can not be done using classical detection
techniques, such as metal detectors, because their metal content is null or very low. The use of infrared
images of the soil is an efficient technique to detect this kind of mines1. This approach is based on
thermal modeling of the heat transfer processes in the soil and at the soil-air interface. This is used to
characterize the soil thermal response to a given stimulus, also known as the thermal signature.
Perturbations on the expected signature constitute reliable indicatives of the presence of mines, due to
the different thermal properties of the soil and the mine. The detection of the mines is divided into two
steps; in the first step a comparison between the real data and the data obtained in the simulation of the
thermal model is made. In the simulation process we first assume that there are no mines present. The
differences between the real data and the simulated data give us indicatives of the presence of
unexpected objects. The second step is an inverse problem, in which the thermal model must be run
for multiple soil configurations, representing different possible depths of burial and different types of
targets (mine, stone ...). The nearest configuration to the real data gives us the estimated position and
the nature of the unexpected targets. Several soil configurations (nature and position of the object)
must be run in order to detect with high precision the position and nature of the unexpected patterns.
This detection scheme is a very long time-consuming process on a personal computer. In this work
an architecture of a system which simulates the thermal model is projected onto an FPGA in order to
reduce the computing time. The system is formed by four memory banks, a processing element, a unit
that generate the addresses that must be loaded/uploaded from the memory and an element that
generates the required control signals. The pipelined structure of the design lets to update several
nodes in parallel. In the current implementation a reduction factor in the computing time of 15 is
achieved.
1 P. López. “Detection of Ladmines from Measured Infrared Images using Thermal Modeling of the soil”. PhD,Univ. Santiago de Compstela, Spain, April 2003.
- 55 -DCIS 2004
Implementation of Optimized FFT on Stratix DSPDevelopment Board
Fabienne NOUVEL, IETR/INSA-20 avenue des Buttes de Coesmes-35043 RENNES. France, [email protected]
This paper deals with the comparison of two FFT/IFFT cores implementations on the Altera Stratix
Component.
As FPGAs are particularly well suited to high-speed and regular functions, they can perform DSP
functions, answering both the need for flexibility and high performances. The FFT, FIR, DCT, … DSP
functions are iterative and need high level pipeline, space and time parallelism. However, FPGA
architecture must be optimized in order to increase the performances of the cores, including MAC (
Multiply Accumulate) hard blocks. Using wired blocks, the DSP functions run faster.
In the first part of this paper, the studied DSP board is presented, by mean of the Stratix component,
connected to SRAM, converters DAC and ADC.
The FFT core is studied in order to perform a high speed Multi-Carrier modulation
/demodulation (MC). In fact, this one is easily carried out in the digital domain by performing
IFFT and FFT operations. In the receiver, after direct FFT, the received sequence is
"equalized" in the frequency domain. Nowadays, MC combined with spread spectrum is
undoubtedly a high potential candidate for the air interface of the 4G cellular networks.
The two IFFT/FFT IP cores are presented and compared : the Altera FFT MegaCore function is a
parameterizeable IP core. It uses an in-place mixed radix 4 and 2 decimation in frequency
architecture, and implements any transform length that is a power of 2. The Jaguar II is a
variable FFT/IFFT core up to 1024 points. Available as a soft-core, it is parameterized to
allow up to 32-bits of resolution (32 Inphase / 32 Quadrature).
The two cores have been implemented, taking into account the specific architecture of the Sratix
component. The results show that the Altera FFT core runs faster than the Jaguar core and used less
resources (both DSP blocks and memories). Neither, the Jaguar requires less cycles than Altera.
Indeed, a 1024 points FFT with a 45 MHz system clock is performed in only 29 s with the Jaguar
core, compared with the 130 s with Altera, which means a ratio of 4. This lowest clock system will
result in a lowest power consumption.
- 56 -DCIS 2004
Comparison of Two Implementations of Scalable Montgomery Coprocessor Embedded in
Reconfigurable Hardware
Miloš Drutarovský(1), Viktor Fischer(2) and Martin Simka(1),(1) Department of Electronics and Multimedia Communications, Technical University of
Kosice, Park Komenského 13, 04120, Kosice, Slovak Republic, Milos.Drutarovsky,[email protected]
(2) Laboratoire Traitement du Signal et Instrumentation, Unité Mixte de Recherche CNRS 5516, Université Jean Monnet, 10, rue Barrouin, 42000 Saint-Etienne, France,
T HIS paper presents a comparison of two possible approaches for the efficient implementation of
a scalable1 Montgomery Modular Multiplication (MM) coprocessor on modern Field
Programmable Logic Devices (FPLDs). The first implementation uses a data path based on
traditionally used redundant Carry-Save Adders (CSA), the second one exploits standard Carry-
Propagate Adder (CPA) with fast carry chain logic not yet used in fully scalable designs. Both
implementations use large embedded memory blocks available in recent FPLDs. Speed and logic
requirements comparisons are performed on the optimized designs. The issues of targeting a design
specifically for a FPLD are considered taking into account the underlying architecture imposed by the
target FPLD technology.
It is shown that carry-save adder is not an optimal building block for constrained scalable MM
coprocessor in modern Altera FPLDs. The proposed implementation method can also be applied for
FPLDs from other vendors since it uses building blocks generally available in modern FPLDs – high-
speed dual-port embedded memories and fast carry-propagated logic.
FA FA FA
FA FA FA
Yw-1(j) Mw-1
(j) Yw-2(j) Mw-2
(j) Y0(j) M0
(j)
1Sw-1(j)
2Sw-1(j)
1Sw-2(j)
2Sw-2(j)
1S0(j)
2S0(j)
qi
t
xi
1Sw-1(j-1)
2Sw-1(j-1)
1Sw-2(j-1)
2Sw-2(j-1)
1S0(j-1)
2S0(j-1)
∆
∆
∆
FA FA FA
FA FA FA
Yw-1(j) Mw-1
(j) Yw-2(j) Mw-2
(j) Y0(j) M0
(j)
Sw-1(j) Sw-2
(j) S0(j)
qi
xi
Sw-1(j-1) Sw-1
(j-1) Sw-1(j-1)
cin1
cin2
cout1
cout2
∆
Figure 1. Block diagrams of analyzed CSA and CPA based processing elements
1 A.F. Tenca, C.K. Koc, “A scalable architecture for modular multiplication based on Montgomery’salgorithm”. IEEE Transactions on Computers , vol. 52, no. 9, pp. 1215-1221, Sept. 2003.
- 57 -DCIS 2004
An implementation of a Parallel Architecture for the Self-Sorting FFT Algorithm applied to IEEE 802.11a
Ainhoa Cortés* , Igone Vélez* , Pilar Calvo* , Juan F. Sevillano † and Andoni Irizar †. * CEIT Research Center, Department of Electronics and Communications, Spain.
† Universidad de Navarra, Department of Electrical and Electronic Engineering, Spain.
N this paper we present an implementation of a parallel architecture for the Self-Sorting (SS) Fast
Fourier Transform Algorithm that optimizes the processing rate for the IEEE 802.11a standard. Two
structures have been developed in the radix-2 Butterfly to improve the architecture. In order to analyze the
dependence of the FFT on the bit-width of the input data and of the twiddle factors, the SNR of our module
has been studied.
The resulting design is parameterizable, regular and modular, presenting constant geometry. The total
processing time required is NrQnN rlog)()2( for a number of points N=rn , where r is the radix and n
represents the number of the stages to process the FFT, computed using Q=ru processors.
The SS algorithm was implemented on a processor column (PEs). The data flow between PEs, by using
eight processors in parallel to execute a FFT-radix 2, is shown in figure 1. In table I we compare the
processing time of our design with other architectures for different clock frequencies. As IEEE 802.11a
needs 4 s as processing time, the Parallel Architecture presented here fulfils the timing specifications.
Table I. Comparison with other architectures
CLK (MHz) Processing time ( s)
Fast-64 1 50 2.82 64-Xilinx 2 50 3.84 Cobra 3 40 5.55 64-Point 4 40 3.2 64-Point 100 1.3 Parallel SS 50 2.22 Parallel SS 40 2.775 Parallel SS 100 1.11
Figure 1. Parallel Architecture
1 L. Fanucci, M. Forliti, and P. Terreni, “Fast: FFT ASIC automated synthesis”, INTEGRATION, the VLSI journal, vol. 33, pp. 230-234, 2000. 2 “Xilinx Product Specification: High-Performance 64-point complex FFT/IFFT V1.0.5”. 3 T. Chen and L. Zhu, “COBRA: A 100- MOPS single-chip programmable and expandable FFT”, IEEE Transactions Very Large Scale Integration (VLSI) Systems, vol. 7, nº 2, pp. 174-182, 1999. 4 Tiong Jiu Ding, John V. McCanny and Yi Hu, “Rapid Design of Application Specific FFT Cores”, IEEE Trans. on Signal Processing, vol. 47, nº 5, pp. 1371-1381, May 1999.
I
PE0
PE7
PE6
PE5
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TOP_FIFO_INPUT TOP_FIFO_OUT
CONTROLSTAGE
ROM
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CONTROLSTAGE
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ROM
- 58 -DCIS 2004
Optimized FPGA implementation of trigonometric functions with large input argument
Javier Hormigo, Manuel Sanchez, Mario A. Gonzalez, Gerardo Bandera, and Julio VillalbaDept. of Computer Architecture, University of Malaga, Spain, [email protected]
T RIGONOMETRIC function evaluation is widely used many current scientific applications such as
digital signal processing, image processing, simulation of physic phenomena, etc. An initial range
reduction is required to perform forward trigonometric functions when the input angle is too large.
The most usual method for range reduction involves two consecutive multiplications. The first one
allows obtaining a scaled version of the reduced input angle, and the second one calculates the correct
value for the reduced input argument.
The CORDIC algorithm is a well-known method for computing trigonometric functions. For the
sine and cosine computation, a vector (1, 1/k) is rotated over the input angle, using iterative rotations
over a fix set of given elementary angles, which are stored in a lookup table.
In this paper, a new range reduction technique which is optimized for the CORDIC algorithm is
proposed. To directly operate over the scaled version of the reduced input angle, the elementary angles
are scaled by the same factor, before store them in the lookup table. Thus, the computation of the
second multiplication is avoided.
The designs based on our proposal require a classical CORDIC module where the table contains
scaled elementary angles. Two basic implementation alternatives are considered: word-serial and
pipeline implementation. Both alternatives have been implemented in FPGA to verify the
improvement obtained with our proposal. For the word-serial implementation, the experimental results
show a speedup of about 32% with the similar hardware cost. For the pipeline case, the classic
approach requires about 32% more CLBs with similar cycle time and large latency.
- 59 -DCIS 2004
Session 3b
Sensors and Smart Objects Wednesday nov. 24 14h15 15h45, St Emilion Room
Chairs
Isabelle Dufour (CNRS-IXL) Salvador Manich (U. Politècnica de Catalunya)
CMOS Buried Double Junction Active Pixel Sensor For High-Sensitivity Low-Resolution Linear Arrays
P. Pittet, G. Carrillo, G.N. Lu, L. Hannati LENAC, Université Lyon 1, Villeurbanne, France, [email protected]
N this paper, we present the study and design of a CMOS active pixel sensor (APS) for a high-
detectivity, low-resolution linear array, which is intended to be used as part of biochemical
microanalysis systems for imaging and spectrophotometric purposes. The proposed CMOS APS
implements a large buried double p-n junction photodetector (BDJ) and charge sensitive regulated
cascode amplifiers. One benefit of using a BDJ photodetector rather than a simple photodiode is that
the former has two junctions for collecting carriers, thus providing higher sensitive response. Another
advantage of employing a BDJ detector is that it can be used as a wavelength-sensitive device, which
may be helpful for selectivity achievements in biochemical analysis applications.
The detector of the APS has an area of 100 µm x 300 µm. To deal with its inherent junction
capacitances related to its size, we propose a pixel circuitry integrating charge sensitive regulated
cascode amplifiers. This allows the use of integration capacitors much lower than the detector’s
parasitic capacitances, thus achieving much higher conversion ratio (160nV/e-) compared to
conventional architectures.
Time domain analysis and simulations are performed for dominant noise source identification and
quantification. At a supply voltage down to 3V, the proposed APS has a dynamic range larger than
60dB. For an integration time of 200ms, the detectivity of the proposed APS is evaluated to be 3.9 1012
cm Hz-½ W-1 for the well channel and 2.3 1012 cm Hz-½ W-1 for the diffusion channel.
Figure 1 a) schematic and b) layout of the 0.8-µm CMOS BDJ APS
I
a) b)
Vdd
Reset
Cint
Vdd
Cint
T'1
T'2
hν
T'3
T2
Readout
DiffusionChannel Output
WellChannel Output
T1
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T4
T5
- 61 -DCIS 2004
An Experience on Wireless Networks for Industrial Applications
Emili Lupon and Gabriel Torrens Departament d'Enginyeria Electrònica, Universitat Politècnica de Catalunya
Av. Diagonal 647, planta 9, E-08028 Barcelona (Spain) Phone: (+34)934016601, Fax: (+34)934017785, e-mail: [email protected]
HIS paper presents a first prototype of a system monitoring a set of remote temperatures. The most
important requirement of the system is that the temperature sensors must be watertight modules,
without any incoming or outgoing wire, neither for information transmission nor for power supplying.
This implies the use of intelligent battery powered temperature sensors, wireless connected to a central
monitoring unit. A reliable RF link on the 433 MHz ISM band has been selected for data transmission
because this type of wireless link is the best suited for the envisaged industrial applications of the
system, as temperature monitoring of cold-storage rooms, where walls must be traversed.
The temperature monitoring system is integrated by up to 255 intelligent temperature sensors and a
master unit. Each temperature sensor is wireless connected to the master unit through an RF link, and
all the RF links are operated at the same frequency, thus requiring only a transceiver in the master.
This architecture implies some design considerations, as an addressing scheme for the nodes in the
system and a transmission policy to avoid collisions (only one node may be transmitting at a time). A
policy based on time multiplexing was selected, thus requiring the existence of a "global" clock from
where to build quasi-synchronous time multiplexing in every node. This "global" clock is obtained
through a synchronization mechanism based on periodical transmission of synchronization frames.
The intelligent temperature sensors are implemented by connecting an accurate temperature sensor
(DS18B20 from Dallas-Maxim) and an RF transceiver (nRF401 from Nordic) to a microcontroller
( PD78F9076 from NEC). For improving the debugging performances, the sensor node has been
completed with an external user interface that includes micro-switches, LED's and an LCD. A simple
master unit has been considered, with its architecture similar to the architecture of the sensor nodes.
A low power design has been considered for the temperature sensor in order to improve its
autonomy. This low power design significantly affects the temperature acquisition process and the
communication mechanism established between the central unit and the remote sensors, as both the
RF emitter and receiver are very consuming subsystems. Sensor autonomy of 8 months has been
achieved for the first prototype when using three AA battery cells for powering it. The system is able
to work at distances greater than 100 m in an open area using cheap loop antennas. Improvements of
the system, as considering the use of the new nRF9E5 device from Nordic, are under study.
T
- 62 -DCIS 2004
Sigma Delta Based Parametrable Sensor Interface A Design Methodology Y. Fellah, N. Abouchi, T. Tixier, A. Aubert, L. Labrak. CPE LYON / INSA - LPM UMR INSA / CNRS C 5511Bâtiment Blaise Pascal - 7, avenue Jean Capelle 69621 VILLEURBANNE
Signal conditioning and analog signal conversion constitute essential links in data processing sequence. Each sensor needs its own interface. However, a same interface,
provided that it is parametrable, can be used to implement various kind of sensors. This article describes a methodology used to design this interface. Our approach is illustrated by an exampleof a Sigma Delta modulator implementing switched capacitors. CMOS standard technology was chosen because it is well adapted to switched capacitor technique. Moreover, Sigma Deltamodulator allows small signal conversion without initial amplification. Finally, direct integration of sensor in the modulator architecture is possible.
The interface that we present uses individually parametrable analog and digital blocks whose assembly constitutes itself a block able to interface a given sensor.The concept is validated on a second order Sigma Delta converter. The adaptation of the first stage integrator enables to associate it with various kind of sensors. The architecture of thesecond stage remains the same.
The methodology is partly illustrated on a capacitive sensor. This sensor replaces thesampling capacitor in a Sigma Delta DT modulator.
- 63 -DCIS 2004
OVE wave chemical sensors are very sensitive Surface Acoustic Waves sensors, their high
sensitivity is due to the confinement of the acoustic energy in a guiding layer nearby the
surface of the sensing area. We present an electronic system devoted to chemical and biological
sensing applications using Love waves delay lines. In an oscillator configuration, detection of target
compounds corresponds to a frequency shift. This oscillation frequency has to be the least noisy as
possible in order to reduce the detection limit. The oscillator loop is composed of Love waves delay
line and electronic feedback to satisfy the Barkhausen conditions in phase and gain. The role of the
electronic feedback is essentially to equilibrate the insertion losses of the delay line estimated to up to
40 dB for gaseous or liquid detection; a change of the wave phase velocity in the delay line due to
sensing implies a variation of the phase condition in the oscillation loop and so an oscillation
frequency shift. The electronic feedback is composed of amplifiers, attenuator, filter and coupler to
sample the oscillation frequency. Noise figure of each component, and especially amplifiers, has been
studied to improve the stability of the oscillator. The transmission of the signal has also been studied
to avoid signal reflection. That’s why the characteristic impedance of the transmission lines have been
matched to 50 and an upper and a lower ground plan have been introduced in the design of the
electronic card. “Short term stability” has been monitored in order to evaluate the stability of the
oscillator. This so-achieved oscillator (Figure 1) results in a short-term stability lower than 1 Hz/s at a
110 MHz working frequency with phase noise of –100 dBc/Hz @ 1kHz. Such stability enables very
low concentration detection of target compound, lower than 1 ppm for gaseous detection.
Figure 1 : Picture of the realized electronic feedback
An improved Love wave oscillator for low concentration chemical sensing application
Nicolas Moll1, Corinne Déjous1, Dominique Rebière1, Jacques Pistré1, Roger Planade2
1 Laboratoire IXL, ENSEIRB, CNRS UMR5818, Univ. Bordeaux 1, 351 cours de la libération, F-33405 Talence, France 2 Centre d’Etudes du Bouchet, (DCE/DGA), 91710 Vert Le Petit, France
L
- 64 -DCIS 2004
S. Maëstre, P. Magnan
CIMI - SUPAERO, Toulouse, France, [email protected]
The hot-carrier effects and electroluminescence are well-known phenomena in CMOS
technology, many devices are not subject to luminescence from transistors but CMOS imagers are
sensitive to it. The requirements of both scaling down the pixel size of CMOS Image Sensor and
maximizing its fill factor are commonly fulfilled by making use of deep submicron process and close-
to-minimum size geometries for in-pixel transistors.
In this context, we analyze the degradation of pixel induced by hot carriers generation of the
in-pixel source follower transistor and show that it is associated with electroluminescence. We present
the different kind of parasitic effects in the CMOS imagers. These effects have been observed in
several process generations and more particularly in a 0.25µm technology with various operating
conditions. In the CMOS imagers, the hot carriers (H-C) generation can occur in the source follower
transistor Msf when this one operates in saturation, i.e. during the selection of the pixel. In this
condition, a secondary impact ionization induces minority carriers that flow in the substrate and can be
collected by the photosensitive areas. Physical 2D simulations (ISE-TCAD environment) of the pixel
behavior allow to location clearly the H-C generation and to demonstrate that impact ionization
induced carriers is measured as an excess current in darkness. Comparison of analog simulations of the
substrate current from Msf and measured photodiode current demonstrates a strong correlation
between the excess dark current of the photosensitive area and the H-C generation revealed by the
substrate current.
Furthermore, this impact ionization phenomenon allows a photon generation whose light
intensity increases as gate voltage Vin decreases. The correlation between the H-C creation (shown by
the substrate current of Msf) and the electroluminescence is presented in a 0.25µm.
Figure. 1 shows the electroluminescence phenomenon from a pixel sub-array using a 0.35µm
process and show that only the selected rows emit light.
Light emission
One pixel
Figure. 1 Electroluminescence from the source follower transistors
when reading successively three different rows. (0.35 process, Ibias=13µA).
Parasitic hot electron effectsin active pixel sensors
- 65 -DCIS 2004
Ring-Oscillator Based Temperature Sensor for Deep Submicron CMOS Technologies
S.A. Bota, V. Canals, J.L. Rosselló, J.A. Segura Grup de Tecnologia Electrònica. Universitat de les Illes Balears. Campus UIB. 07122 Palma.
Spain. Tel. +34971173233. email [email protected].
WITH the increasing power density in deep submicron integrated circuits, the occurrence of
failures due to overheating has considerably increased. In this paper, a simple and efficient built-
in temperature sensor for the on-line thermal monitoring of standard-cell based VLSI circuits is
presented. The proposed smart temperature sensor is based on a ring-oscillator. It has been found that
the oscillation period can reach a linear dependence with temperature using an adequate ratioed
inverter (fig 1). This implies that to obtain an adequate non-linearity error, the ring-oscillator must be
optimised at transistor level; thus involving transistor sizes different to those to the inverters of the
target standard-cell library.
To produce an output signal with a period proportional to the temperature, we have replaced the
inverters of the oscillator by more complex inverting gates. Simulation results obtained in a 0.18mm
CMOS technology show that the non-linearity error of the sensor can be reduced when an adequate set
of standard logic gates is used (fig. 2).
2,8E-10
3,0E-10
3,2E-10
3,4E-10
3,6E-10
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-50 -25 0 25 50 75 100 125 150
Temperature (ºC)
Peri
od (s
)
rat = 1.00rat = 1.75rat = 2.00rat = 2.25rat = 3.00rat = 4.00
Figure 1. Oscillation period versus temperature fordifferent Wp/Wn ratios.
-0,5
-0,4
-0,3
-0,2
-0,1
0,0
0,1
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-50 -25 0 25 50 75 100 125 150
Temperature (ºC)
Erro
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2in+3na25inv2in+3na43in+2no23na3+2no25no25na2
Figure 2. Non-linearity error for different ring-oscillatorconfigurations.
- 66 -DCIS 2004
Session 3c
Bio-inspired Circuits Wednesday nov. 24 14h15 15h45, Auditorium
Chairs
Sylvie Renaud (E.N.S.E.I.R.Bordeaux) Hervé Barthélémy (E.P.U. de Marseille)
Experiments on electrical MFHN neurons
S. Binczak, S. Jacquir, O. Tarlet and J.M. Bilbault Laboratoire LE2I, CNRS UMR 5158, Aile des sciences de l'ingénieur, Université de
Bourgogne, BP 47870, Dijon Cedex, France
We present an electronical analog circuit modelling a FitzHugh-Nagumo neuron with a modified
excitability. To characterize this basic cell, the bifurcation curves between stability with excitation
threshold, bistability and oscillations are investigated. An electrical circuit is then proposed to realize
an unidirectional coupling between two cells, mimicking a chemical synaptic coupling.
In this master-slave configuration, we show experimentally that the coupling strengh and the master
interspike period control the dynamic of the slave neuron, leading to period doubling, chaotic behavior
and synchronization.
The architecture of the neural network is then described allowing small assemblies studies.
- 68 -DCIS 2004
A Mixed Neuromorphic ASIC for Computational Neurosciences
Sylvain SAÏGHI, Jean TOMAS, Yannick BORNAT, Sylvie RENAUD
IXL Laboratory , C.N.R.S. FR 2648 – UMR 5818 C.N.R.S., ENSEIRB – Université Bordeaux 1 351 Cours de la Libération, 33405 Talence Cedex, France (e-mail: [email protected]).
This paper presents an original mixed IC, designed for the development of computational
neuroscience hardware/software tools. This ASIC integrates configurable neuromorphic
functionalities, and computes in real-time the electrical activity of various neural elements,
described by conductance-based models. The neurons structure and models parameters are on-
chip programmable. We present here the key issues of the design, the ASIC main
characteristics, and how it will be integrated in the complete simulation system. Different
neural behaviors (spiking, bursting) are then programmed on the chip. Results are evaluated
and compared to those obtained by software simulation tools.
Index Terms : Mixed Integrated Circuit, Silicon neurons, Neuromorphic engineering, BiCMOS
- 69 -DCIS 2004
Mixed-Mode Class AB Neuron Building Blocks: Analysis and Real Application
G. Zatorre Navarro, N. Medrano Marqués, S. Celma Pueyo Electronic Design Group, University of Zaragoza – Spain
E-mail: gzatorre, nmedrano, [email protected]
RTIFICIAL Neural Networks are computing tools consisting of small processing elements, called
artificial neurons, highly interconnected and arranged in layers. Input-output function carried out
by these systems is learned by means of a training process, adjusting the system free parameters that
connect inputs from a neuron layer with the preceding neuron layer outputs. Previous works have
presented the use of mixed-mode electronic blocks in artificial neurons implementation, showing
promising results applied to real problems. In this work, design and simulation of some electronic
basic blocks to build a class AB mixed-signal current mode artificial neural network are shown. The
main electronic blocks are a four-quadrant mixed-mode multiplier based on an R-2R ladder (Fig. 1)
and a current mode non-linear transfer function (Fig. 2). Resulting practical models are used to
designing a Multilayer Perceptron, and are applied to a sensor linearization problem. In order to
minimize effects due to mismatching and offsets in the proposed blocks along the training process, a
perturbative algorithm is selected to match the suitable weights. The performance achieved with the
neural model in four different sensor samples show a sensor linear range extension (an error lower
than 1 degree) of 50% or more.
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Fig. 1. 3–bit four quadrant mixed A/D multiplier. Fig. 2. Class AB activation function.
A
- 70 -DCIS 2004
A Discrete-Time Cellular Neural Network Architecturefor a Pixel-Level Snake On-chip Implementation
V.M. Brea, D.L. Vilariño, D. Cabello Department of Electronics and Computer Science
University of Santiago de CompostelaSantiago de Compostela, Spain
Phone:+34981563100, Ext. 13572. Fax:+34981528012 Email : [email protected]
I N this paper, we approach the hardware-level design for the on-chip implementation of an active-
contour-based technique, an improved Pixel-Level Snake (PLS) version. Such a new pixel-level
snake technique counts on features from both, parametric and implicit models. This leads to a better
performance on contour detection with low computational cost. The computation time is also
decreased to reach video rate processing with an on-chip implementation by means of an SIMD
architecture with a direct correspondence between pixel and processing element, namely Discrete-
Time Cellular Neural Networks (DTCNN).
The synergy of PLS and CNN, either Continuous-Time (CT) or Discrete-Time (DT) is a promising
tool for real time processing. This has been proven by running the PLS technique reported here onto a
general purpose CTCNN chip, the ACE4K1. The design of a specific CMOS DTCNN chip has also
been successful for the original PLS2. This paper addresses the new DTCNN PLS architecture along
with the new DTCNN PLS cell for a future specific CMOS on-chip implementation.
1 D.L. Vilariño, Cs. Rekeczkey, Implementation of a Pixel-Level Snake Algorithm on a CNNUM-based ChipSet Architecture, IEEE Transactions on Circuits and Systems-I, Volume 51, Issue 5, pp. 885-891, May 2004.2 V.M. Brea, D.L. Vilariño, A. Paasio, D. Cabello, Design of the Processing Core of a Mixed-Signal CMOSDTCNN Chip for Pixel-Level Snakes, IEEE Transactions on Circuits and Systems-I, Volume 51, Issue 5, pp.997-1013, May 2004
- 71 -DCIS 2004
Charge-Packet Driven Mismatch-Calibrated Integrate-and-Fire Neuron for Address-Event-Representation
Rafael Serrano-Gotarredona, Bernabé Linares-Barranco, and Teresa Serrano-Gotarredona Instituto de Microelectrónica de Sevilla, Ed. CICA, Av. Reina Mercedes s/n, 41012 Sevilla,
SPAIN. Phone: 95-505-6666, Fax: 94-505-6686, E-mail: [email protected]
W E present the design and experimental measurements of an integrate-and-fire pixel for
Address-Event-Representation (AER) transceiver chips such that (a) input events can be
weighted according to a digital word, (b) this weight includes a sign bit, (c) the incoming event is
accompanied by a sign bit, and (d) the pixel can be calibrated to compensate for mismatch in large
arrays of these pixels. A prototype has been fabricated in the AMS 0.35µm CMOS process, whose
experimental measurement results are provided.
- 72 -DCIS 2004
Digital Implementation of a Simplicial Cellular Neural Network
Pablo Echevarría, M. Victoria Martínez, José M. Tarela and Inés del Campo Department of Electricity and Electronics, University of the Basque Country, Spain,
T HIS paper presents a high-performance fully digital implementation of cells of the recently
introduced simplicial cellular neural network (CNN). The simplicial CNN exhibits a higher
functional capacity than the standard CNN, while keeping the complexity within acceptable limits.
The theory of canonical piecewise-linear (PWL) representation underlying the simplicial CNN makes
the structure particularly advantageous when the CNN needs to be trained from examples. This work
presents a digital implementation on a FPGA platform of the simplest two-dimensional configuration
of a cell in a simplicial CNN, i.e. a 3x3 neighborhood architecture. The use of reconfigurable devices
to implement emulated digital CNNs provides more flexibility than the VLSI designs because
different architectures can be used on the same FPGA device.
The design proposed here and shown in Fig. 1 implements the interconnection behavior of a
simplicial cell, and has been compiled for an Altera FPGA of the family Stratix, the EP1S80F1508C7
(Stratix I). The result is an occupation of less than 1% of the logic elements, memory bits and DSP-
blocks, and a maximum performance frequency of 171 MHz.
The design is intended to be a part in emulated digital simplicial CNNs, which could offer better
characteristics than software versions for the synthesis of PWL models used in many applications of
nonlinear systems. The results of this work are being compared with software and digital
implementations of other techniques oriented to efficiently represent PWL components.
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- 73 -DCIS 2004
Session 3d
Power Electronics : Devices and Systems
Wednesday nov. 24 14h15 15h45, Bordeaux Room
Chairs
Rafael Burriel (CeDint, U. Politécnica de Madrid) Francisco J. Azcondo (U. de Cantabria)
Optimization of a high voltage p-channel transistor fabricated using a standard CMOS
processA. Pérez-Tomás, X. Jordá, P. Godignon, M. Vellvehí and J. Millán
Centre Nacional de Microelectrònica (CNM-CSIC). Campus UAB, 08193, Barcelona. Spain
Author coordinates: Tel. (34) 93 594 77 00, Fax. (34) 93 580 14 96, e-mail: [email protected]
HIS paper has been focused on the optimization of a simple high voltage extended drain p-
channel structure (ED-pMOSFET), fabricated using an standard low cost twin-tub 2.5 m CMOS
technology. There is a strong interest to monolithically integrate CMOS and high-voltage power
devices within the same process. An existing low-voltage technology should desirably be used for the
fabrication of Power Integrated Circuits (PICs), in order to reduce the development efforts and time.
One of the main advantages of this approach is that all the existing standard cells and libraries can be
used in designing the low voltage parts of the circuit. CMOS technologies can be n-well, p-well or
twin-tub. The fabrication of high voltage pMOSFET devices fully CMOS compatibles with
nMOSFET devices in a twin-tub process could become a difficult or even impossible task.
Only one process step and one mask level has to be added to the standard CMOS process to
implement the ED-pMOSFET. This structure has been optimized attending the channel length LCH, the
extended drain length LED, and the doping level of the extend drain NED. Other parameters (i.e. n-well
doping level) are fixed by the CMOS process. The simulation results (TMA-MEDICI) have been
verified by experimental implementation. ED-pMOSFET transistors with low specific on-resistance
(Active area) Ron=6.0m ·cm2 (@Vg=-5V) and breakdown voltage of 36V have been implemented.
These results evidence that, with an adequate optimization, this device is competitive with most of
previous p-channel devices reported, using more expensive and sophisticated technologies and
processes. The ED-pMOSFET is one of the simplest and fully CMOS compatible of the designs
usually proposed. Along with an n-channel LDMOS (completely integrated within the standard
CMOS process), the ED-pMOSFET composes a CMOS based technology (50V/1A) suitable for
power integrated circuits.
T
- 75 -DCIS 2004
Digital Phase-Shifting for Multiphase Converters
A. de Castro, P. Zumel, O. García, T. Riesgo Universidad Politécnica de Madrid, E.T.S.I.I. División de Ingeniería Electrónica
Madrid/Spain. e-mail: [email protected]
ULTIPHASE converters are becoming especially important among switching power converters.
This is due to their advantages, such as higher current capability, improved dynamic response or
reduced EMI and harmonics, which compensate for their increased components count. Some of the
key applications for multiphase converters are Voltage Regulation Modules (VRM), Dynamic Voltage
Scaling (DVS) or automotive power supplies.
As a drawback, multiphase controllers become more complex than their one-phase counterparts.
However, digital controllers based on custom hardware (FPGAs or ASICs) easily solve the new
problems, such as multiple driving signals generation, current sharing among phases or phase-shifting.
This work especially focuses on the phase-shifting problem (generating the driving signal of each
phase shifted from the previous one), proposing and comparing two different hardware structures for
this task. The first structure is based on additions and comparisons, while the second one uses a shift-
register (see figure). Both methods are explained in detail and compared. The comparison rule that
leads to bigger differences is area. The addition and comparison phase-shifter is appropriate for high
duty cycle resolutions, while the shift-register solution is better for high number of phases. Both
methods have been implemented and tested with a prototype, showing that both allow passive current
sharing (no current loop).
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Figure 1. Addition and comparison phase-shifter (a) and shift-register phase-shifter (b).
M
- 76 -DCIS 2004
Electro-thermal Characterization of Ultracapacitors Used as Power Sources in Hybrid Electric Vehicles
W. Lajnef, O. Briat, S. Azzopardi, E. Woirgard. J.-M. Vinassa Laboratoire IXL CNRS UMR 5818, Université Bordeaux 1
351 Cours de la Libération 33405 Talence Cedex - FRANCE Contact author: e-mail: [email protected]
LTRACAPACITORS are attractive devices for electric storage. Because of their very low serial
resistance, they are able to exchange high levels of instantaneous power. The cell capacitance can
reach five thousand of Farads. They can work at very low temperature and can support a significant
number of charge/discharge cycles. As typical application, they can be used as peak power sources in
Hybrid Electric Vehicles (HEV). The pulsed charge-discharge current mode needed for these
applications, leads us to propose an original approach for electro-thermal characterization of
ultracapacitors.
In a first part, the ultracapacitor electric behavior is investigated and leads to the proposal of a
specific electrical model. It is composed of an access resistor and capacitor, a serial inductor and a non
linear transmission line which is approached by four RC branches. The capacitance of the transmission
line depends on the ultracapacitor voltage. The model parameters are identified using both constant
currents and impedance spectroscopy tests. At last, the model validation is made thanks to a pulsed
current profile.
In the second part, regarding thermal characterization, a simple model based on two thermal time-
constants is presented. The use of high current levels and repetitive charge-discharge profiles that are
energetically neutral has allowed to reach a significant heating of the ultracapacitor. So, the model
parameters have been extracted in order to predict the maximum heating of the device.
At last, the combination of the thermal model and the electrical one allows us to compare simulation
and experimental results. The good matching between these data leads to consider that Joule losses are
the only heating source. So, the proposed electro-thermal characterization is validated.
U
- 77 -DCIS 2004
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- 78 -DCIS 2004
Specific Drivers and Integrated 20V Regulated Charge-Pump for an Autonomous MicroRobot: MiCRoN
A.Saiz Vela, P. Miribel-Català, J.Brufau, R.Casanova, M.Puig-Vidal, J. Samitier.Electronics Department. Instrumentation & Communication Systems Lab. Universidad de
Barcelona. C/ Martí i Franques,1. Barcelona 08028. (SPAIN). asaiz,[email protected]
HIS paper deals with the design of the driving and power supply system for a microrobotic
application where power consumption is a critical issue.
In this paper is presented the evolution of the MINIMAN-V microrobot1 where the main limitation
was the autonomy. Here are presented the drivers and the solution to step-up the standard 3.3V from a
battery needed to bias and actuate on the piezoelectric structures that have been developed by some
partners in the frame of the MICRON project (IST-2001-33567) where our work is being developed.
A first prototype for the electronics has been developed, based on SMD components, where the
power consumption and area are two main key points. Regarding the drivers, based on the LM7301
operationa amplifier, they present an important dc power dissipation on board, around 40mW for each
driver. This amplifier is for general purposes, with 4MHz of GBW, and it is not present any type of
power down control. Our High Voltage Operational Amplifiers (HVOA), have been designed to
present stability for a fixed gain, not for the general case of the unity frequency (fu). So we can adjust
the power dissipation to lower values, and also a power down control has been designed.
Regarding the signals needed for two types of piezoelectric actuators, drivers based on the use of high
voltage operational amplifier have been investigated, with voltage bias up to 20V. Class A and class
AB operational amplifiers in a control loop configuration have been designed to meet the both
requirements of both actuator types. In the case of classe A op amp, a specific design for one actuator
has been made. For both piezoelectric actuators a class AB operational amplifier has been designed,
with power consumption lower than a defined threshold for the total power dissipation of 15mW for
each output. Concerning the integrated power supply, a two phase voltage doubler charge pump has
been adopted to rise the output voltage up to 20V.
1J. López-Sanchez, U. Simu, M. Puig-Vidal, S. Johansson, P. Miribel-Català, E. Montané, S.A. Bota, J. Samitier. "A miniature robot driven
by Smart Power Integrated Circuits". IEEE/RSJ Intl. Conference on Intelligent Robots and Systems, pp. 1954-1959, Lausanne (CHE -SUIZA), 2002.
T
- 79 -DCIS 2004
AC current and DC voltage sensorless control of bidirectional boost-buck converter
Y. Touzani, J.P. Toumazet, P. Le Bars, A. Laurent and F. Gary Laboratoire d’Electrotechnique de Montluçon, avenue Aristide Briand
BP 2235, 03101 Montluçon Cedex, France E-mail: [email protected]
HE power converter technologies for motor drive are very popular in various industrial
applications such as robots, machine tools etc. Those dc or ac drives need a step up / down dc bus
voltage and regeneration capability. In this type of converter, for normal operation, three kinds of
sensors for detecting ac current, dc voltage and load current are basically required. A dc voltage sensor
is demanded for the dc voltage feedback control. The load current sensor is needed to improve
dynamic response in the dc voltage control. The two line current sensors are required for the input
current control and insure power factor control operation.
This paper presents a novel control scheme of bidirectional three-phase PWM boost-buck rectifiers
eliminating both the ac input current and dc output voltage sensors (Figure 1). The purposes of the
proposed control scheme are :
- Reducing the number of sensors, which minimizes the cost of system.
- Improving the reliability by getting rid of input currents disturbance influence.
- obtaining a completely programmable control scheme, which facilitates its evolution without
modifying hardware.
Figure 1. Bidirectional three-phase boost-buck converter without ac-current sensors nor dc-voltage sensor
The dc output voltage is estimated by measuring load and coupling inductance currents. The ac
currents are reconstructed from switching states of the PWM and from measured voltage and coupling
circuit current. The estimators are developed with MATLAB / SIMULINK (Power System Blockset)
software and converter operations are validated for a 5,5 kVA model. The implementation is currently
carried out on a TMS320LF2407A DSP, in order to realize a prototype of this converter.
T
- 80 -DCIS 2004
Session 4a
Image Processing Wednesday nov. 24 16h15 17h45, Lacanau Room
Chairs
Yannick Berthoumieu (E.N.S.E.I.R.Bordeaux)José Luis Martín (U. del País Vasco / Euskal Herriko U.)
Antoni Portero*, Pol Marchal**, J.I.Gomez***, L.Pinuel***, Francky Catthoor**, Jordi Carrabina*
*Dept. Informàtica, UAB, Bellaterra, Spain antoni.portero,[email protected]
**IMEC xvz, Leuven, Belgium marchal,[email protected]
***DACYA, UCM, Madrid, Spain jigomez,[email protected]
eterogeneous multi-processor platforms are an interesting option to satisfy the
computational performance of dynamic multi-media applications at a reasonable energy
cost for embedded portable systems. In this paper is shown different inter frame compression
algorithms for different GOPs (Group of Pictures) and different trade offs in terms of power
consumption/time execution, Image Quality/Bit rate. This study gives information about this
multi-dimension optimization objective. Taking into account packet and transmit data through
a wireless channel. The objective is to adapt this amount of compressed data to transmit it
through the channel bandwidth optimizing system performance.
This paper focuses in the study of different scheduling for a GOP (Group of Pictures) that has to
be compressed with real time requirements. Instead of working having in mind worst case
(WCET). In a multi processor platform different scheduling permit to obtain different working
points (power consumed, executed time). Hence, we would decide what scheduling (working
point) we need in each case depending on the power consumption and time to compress
information.
For example, information is transmitted by a noisy channel and with a short bandwidth. We can
decide to compress more and the time consumed to compress data is higher and also power
consumption maintaining image quality more or less constant (PSNR), or if transmission channel
is wider then we can relax compression. Hence, it adapts performance to channel maintaining
the PSNR.
A study of Trade offs in Inter-frame Compression MPEG-4 for a Multiprocessor platform
H
- 82 -DCIS 2004
Adviser Coprocessor for Image Compression on FPGA
Antonio Guzmán DIET, ESCET
Rey Juan Carlos University Móstoles, Madrid, Spain aguzmanQescet.urjc.es
Marta Beltrán DIET, ESCET
Rey Juan Carlos University Móstoles, Madrid, Spain [email protected]
The image activity measure (IAM) gives a measure of how busy or complicated is an image in terms of edges, contours or textures. It has been demonstrated that there is a strong relationship between the IAM and the reconstruction error values after decompressing an image. This paper pro- poses a coprocessor to perform images Discrete Wavelet Transform (DWT) and to measure their activity from this transform. With the IAM value, the proposed coprocessor is able to advise the user or the host system what kind of compression algorithm is more suitable for each image with the desired compression ratio and reconstruction error. Furthermore, this coprocessor allows to choose the compression algorithm that obtains a greater compression ratio with less reconstruction error.
The adviser coprocessor is implemented on a Field Programmable Gate Array (FPGA) because these reconfigurable platforms provide flexible and high performance solutions at a relatively low cost.
The main contribution of this paper is the design of an adviser coprocessor capable of predicting the reconstruction error of an image when it is compressed with a certain wavelet family to a fixed compression ratio, that is, it can advise what kind of compression technique is more suitable for user requirements. In addition, the implemented prototype allows to evaluate the convenience of sacrificing performance (area and operation frequency) in favour of accuracy, using floating point arithmetic instead of fixed point. The presented design has been developed with a high level hardware description language, HandelC, because it allows the developer to obtain general and parametrized prototypes not biased by architectural choices, without spending a considerable time on the hardware signal semantics.
- 83 -DCIS 2004
Power-Aware Tuning of Dynamic Memory Management for Embedded Real-Time Multimedia
Applications
David Atienza1, Stylianos Mamagkakis2, Miguel Peon1,Francky Catthoor3, Jose M. Mendias1, Dimitrios Soudris2
1DACYA/UCM, Avda. Complutense s/n, 28040, Madrid, Spain.E-Mail:datienza, mendias, [email protected]
2VLSI Center-Demokritus Univ., Thrace, 67100 Xanthi, Greece.E-Mail:smamagka, [email protected]
3IMEC, Kapeldreef 75, 3001 Heverlee, Belgium. E-Mail:[email protected]
I N the near future, portable embedded devices must run multimedia applications with enormous
computational requirements at low energy consumption. These applications demand extensive
memory footprint and must rely on dynamic memory due to the unpredictability of input data (e.g. 3D
streams features) and system behavior (e.g. variable number of applications running concurrently).
Within this context, the dynamic memory subsystem is one of the main sources of power consumption
and embedded systems have very limited batteries to provide efficient general-purpose dynamic
memory management. As a result, consistent design methodologies that can tackle efficiently the
customization of dynamic memory managers according to the complex dynamic behavior of these new
applications for low power embedded systems are in great need.
Nowadays, the complex engineering process of partially customizing dynamic memory managers
for the specific platform features (and the range of dynamic applications that will run on it) can
usually take several months. The reason is that everything is based on manual profiling and testing
according to the inspiration of each developer and his programming style to apply convenient
transformation in his own code. Moreover, if the purpose of the customized dynamic memory manager
(e.g. maximizing performance) is slightly changed (e.g. power reduction is also added as system
constraint in the new design), the new custom dynamic memory manager needs to be redesigned from
the beginning to respect the new requirements. In this paper, we present a new system-level design
approach that is able to obtain a detailed view of the dynamic behavior, i.e. (de)allocation pattern, of
new multimedia applications and optimize it using a step-wise refinement flow to reduce the power
consumption of dynamic memory managers in such embedded systems. In our approach, the dynamic
memory managers are built in a systematic way making use of our own C++ library, which simplifies
enormously the profiling and implementation effort for the designer. The experimental results in real-
life case studies show that our approach improves power consumption up to 89% over current state-of-
the-art dynamic memory managers for complex applications.
- 84 -DCIS 2004
An IIR Based 2D Adaptive and Predictive Cache forImage Processing
Stéphane Mancini and Nicolas EvenoLaboratoire des Images et des Signaux, France, Grenoble
stephane.mancini,[email protected]
I MAGE processing for tasks such as computer vision would benefit of data cache which fits the
bidimensional nature of processed data to overtake SDRAM-like memory access bottleneck, as
illustrated figure 1. Therefore we propose a 2D adaptive and predictive cache (2D-AP cache) relying
on a statistical analysis of 2D coordinate accesses: position and geometry of the 2D cached zone are
determined by parameters such as mean and pseudo-standard deviation (PSD) of the issued coordinates,
considered as a signal. Furthermore a predictive mechanism tries to predict the next image zone that
would be used to download it from memory on time. Results show us that this strategy is very efficient
compared to standards TM32 and PowerPC cache: 2D-AP cache is 50% faster and the memory cache
size is reduced by a factor 4 to 40.
The mean of 2D coordinates issued by the processing unit is used to track the path followed by
the algorithm. That for, a guard region is defined around the current cached zone center and a cache
displacement is performed when the computed mean gets out of this region. The current cached zone
size, guard zone size and displacement speed are computed from the PSD and evolve as the followed
path, assuming a first order constant speed, as shown figure 2. To get high clock frequency and reduce
complexity, the mean and PSD are computed with first order IIR filters which coefficients are power
of 2. Doing so, the cache architecture is made of adders and simple shifters. An interresting result is
that the cache control and performance are independent from the cached memory size.
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Fig. 1: The 2D-AP Cache concept Fig. 2: Path tracking along time
- 85 -DCIS 2004
Real Time Smart Pixels Processing Array forMobile Multimedia Applications
S. López, R. Calzada, A. Tejera, J.F. López and R. SarmientoInstitute for Applied Microelectronics (IUMA)
Department of Electronic Engineering and Control (DIEA)University of Las Palmas de Gran Canaria, Spain, E-35017
OBILE multimedia communications are expected to achieve an unprecedented
growth and worldwide success in the next couple of years, with a potential market
composed by millions of users around the world. Due to its mobile nature, good visual and
voice qualities at high compression ratios as well as reduced area/power dissipation are key
factors for commercial products.
MIn this paper, a Smart Pixels Array designed to perform efficiently key video coding
operations is presented. In particular, the design is capable to compute the Discrete Wavelet
Transform (DWT), Zerotree Entropy (ZTE) coding and Frame Differencing (FD) over SQCIF
video frames (128×96 pixels). The array is composed by a 128×96 bidimensional network of
interconnected smart pixels processors working in a massively parallel fashion, allowing the
operation at very low clock frequencies and hence reducing its power dissipation. Each of
these smart pixel cells present a power dissipation as low as 4.15 µW@128 kHz in a square
area of 110×110 µm2 while the whole array presents a power dissipation of 57.3 mW@128
kHz in a area of 166.24 mm2 using a 0.25 µm CMOS technology. These characteristics make
the designed Smart Pixel Array a highly suitable option for next generation mobile
multimedia devices.
Figure 1. Smart Pixel layout
- 86 -DCIS 2004
Adaptation of Altera Stratix DSP Board for real-time stereoscopic image processing
Pavol Pavelka, Vincent Bertheas, Viktor Fisher, Virginie Fresse.
Laboratoire Traitement du Signal et Instrumentation- 10 rue Barrouin, 42 000 Saint Etienne, France.
Particle Image Velocimetry (PIV) is a method of imaging and analyzing flow fields in Fluid
Mechanics. In PIV technique, motion vectors are extracted by analyzing the displacement of some
particles added in the liquid or air flow. The retained and more appropriate method for our
applications consists in image acquisitions by means of two separated cameras and an FPGA based-
architecture. Separated video cameras are positioned in a stereoscopic position and measurement
consists in comparing two images of particles in real-time. The major difficulty of such system lies in
the camera synchronization as incorrect results can be obtained from a lack of precision in the data
acquisition.
Essence of this work is the design of generic cores and the adaptation of a Stratix DSP board for stereo
applications. Based on our existing PIV system, the principle for generic core designs is presented.
Great flexibility and accessibility are required for the true potential generic aspect of cores. They can
be parameterized to fit to any FPGA device and its embedded platform (including additional devices
such as memories, A/D converters..) and eventually coupled architectures to and ensure an immediate
reuse. Following this novel approach, cores dedicated to video acquisition are camera-independent
including the synchronization of both video signals. Real-time memorization cores targeting buffer
and external memories are developed and a set of parameterized cores are available to cover any
external or peripheral components. The PIV algorithm is implemented on a Stratix DSP board coupled
to an expansion board. Observations are made that the synchronization problem is solved with internal
synchronization mechanisms. Time for image processing development is estimated to few hours,
efforts must only be concentrated on the image processing unit. Number of LEs for synchronization
mechanisms is a small percentage of FPGA area and QoR of the implemented algorithm are satisfying
for real-time video applications.
- 87 -DCIS 2004
Session 4b
Embedded Design & System On Chip
Wednesday nov. 24 16h15 17h45, Auditorium
Chairs
Emilio Olías (U. Carlos III) Michel Robert (L.I.R.M.Montpellier)
Unai Bidarte, Armando Astarloa, José Luis Martín, Jaime Jiménez, and Carlos CuadradoUniversidad del Pais Vasco, E.T.S. de Ingeniería de Bilbao, Departamento de Electrónica y
Telecomunicaciones, Urquijo s/n 48013 Bilbao
EVERAL control applications require very high speed data exchange between data source and
sink elements: industrial machinery like filling or milling machines, polyphonic audio, three
dimensional images, video servers, PC equipment like plotters and printers, etc. After dealing for quite
a long time with such applications, it was considered that much work could be reused, and a generic
and reusable core-based architecture for circuits that require high bandwidth data transfers was
designed in order to reduce the SoC design cycle time as much as possible. The first application in
which the generic architecture was used consisted in a high bandwidth router. The selected technology
was Spartan II from Xilinx. In Spartan II architecture, horizontal routing resources are provided for
on-chip three-state buses. Four partitionable bus lines are provided per Configurable Logic Block
(CLB) row, permitting multiple buses within a row.
The bitrate, defined as the amount of information per second exchanged between two cores, is the
key parameter in the architecture, but the amount of resources needed to comply with the bitrate
specification must also be taken into account. When the application implementation moment arrived
an important decision was whether to use internal three-state buffers or multiplexor logic.
The main drawback of three-state buffers is that many integrated circuits do not have any internal
three-state routing resources available to them, so multiplexor logic interconnections are more portable
than three-state logic designs. Other integrated circuits are very restrictive in terms of location or
quantity of these interconnects. The second one is that it is inherently slower than direct
interconnections, and usually slower than multiplexed constructions. That is because there are always
minimum timing parameters that must be met to turn buffers on-and-off. The main disadvantage of the
multiplexor logic interconnection is that it requires a larger number of routed interconnects and logic
gates (which are not required with the three-state bus approach).
This paper contains a quantitative comparison between three-state and multiplexor logic design
alternatives. We have observed that the resource usage, mainly the number of LUTs, can be
significantly reduced using three-state buffers, without an important drop in the maximum clock
frequency. On the other hand, synthesis and place and route tools achieve good optimization results
and do not show any problem to manage internal three-state buffers.
On the Performance of Three-State and Multiplexor LogicInterconnection for Shared Bus SoC Design
S
- 89 -DCIS 2004
Reconfiguration Control for DynamicallyReconfigurable System
YNAMICALLY Reconfigurable Systems (DRS), those where the hardware can be
changed at runtime, have the potential to enhance hardware flexibility to a degree similar
to that of software. At the same time, they may lead to better performance and a smaller
system size. This happens because it allows that parts of the system not needed in some time
interval be removed from the hardware, to make room for another part of the system, required
at that same time interval.
On the other hand, potential drawbacks of using Run-Time Reconfiguration (RTR)
techniques are the performance penalty induced by long reconfiguration times and the area
overhead to implement the hardware responsible for controlling the reconfiguration process.
Moreover, the deployment of DRSs requires extensive support that is not yet available. This
support is composed by tools to enable the use of RTR techniques and infrastructure to
implement DRSs. A framework for the design, verification and implementation of DRS
named PaDReH has been proposed by the authors as one step forward to reduce this lack of
support.
One of the main problems for enabling reconfigurable systems is the unavailability of a
module to control the hardware reconfiguration process. This configuration controller
commands which reconfigurable IP core(s) must be inserted on the reconfigurable device at
any moment, and which must be removed. The main contribution of this paper is the
proposition of a configuration controller totally built in hardware. This is different from
previous approaches, where software implementations dominate. The proposed controller has
been designed, validated and prototyped successfully in VirtexII Xilinx FPGAs.
Ewerson Carvalho, Ney Calazans, Fernando Moraes and Daniel Mesquita*
Faculdade de Informática – PUCRS – Porto Alegre – Brazilecarvalho, calazans, [email protected]
*LIRMM, Université Montpellier II – Montpellier – [email protected]
D
- 90 -DCIS 2004
A SoC-based Architecture coupled with a CMOS Image Sensor for measurements by Image Processing
Lionel Lelong, Guy Motyl, Gérard Jacquet and Nathalie Bochard. Laboratoire Traitement du Signal et Instrumentation – UMR CNRS 5516
Université Jean MONNET, Bâtiment F 10 rue BARROUIN 42000 SAINT-ETIENNE, FRANCE
e-mail : lionel.lelong, motyl, jacquet, [email protected]
Physical parameters measurement and control by image processing imply real time operations
on high data flow (high resolution image). Real time constraint of many image-processing applications
can be met with specific embedded systems eclipsing the current computer system inefficiency. In this
paper we present an architecture based on System on Chip approach to realise real time measurement
by image processing on flow visualisation applications. The used technique is called Particles Images
Velocimetry (PIV), ensuring a velocity measurement in flows in a non-intrusive way. PIV techniques
ensure extraction a motion vector field from a flow seeded of fine particles. The system consists of
two main devices that are a CMOS image sensor and an FPGA. Processing design is implemented
inside the FPGA, mixing ring and bus communication for data and control flows. Our architecture is
based on a “Dominant input data flow” model. This model presents a large bandwidth of input data
flow and a reduced output data flow coupled with command flow (Fig.1). An embedded processor
(NIOS) controls the entire process; acquisition logic and processing elements are all written in VHDL
language. The implementation of processing design with one processing module (PM) takes up 51 %
of Logical Elements and 87 % of embedded RAM (Quartus II results). Our system runs at a frequency
of 50 MHz, but it can run at a maximum of 61 MHz. The system flexibility realises a compromise
between speed and resolution without any material changes. With the retained FPGA, external RAM
memory is needed. For future implementation, memory banks will all be hosted inside the FPGA. Real
time measurement on high-speed flow reaches a minimum of 100 images/sec using adapted
correlation algorithms.
PM
PM
PM
PM
AcquisitionModule
Static RAM
CMOSImageSensor
Data flow
Command and result flow
PCControlModule
MemoryModule
FPGA
Figure 1: Processing design inside our System on-Chip
- 91 -DCIS 2004
Simulation of a NoC-based Heterogeneous System Using Ns-2
Maria Bolado, Pablo Sánchez Microelectronics Engineering Group, University of Cantabria, Santander, Spain,
mbolado,[email protected]
A CCORDING as the integration capability of silicon increases, the number of resources that can be
implemented on the same chip increases as well. However, new technological problems appear,
and these must be overcome. The Network on Chip paradigm arises to give solutions to such
problems. It is necessary to have appropriate simulation tools for the investigation of NoCs. Since the
paradigm of NoC is still a recent idea, there is not a wide offer of simulation tools yet. Reusing
existing tools originally aimed to the simulation of General Computer Networks is an interesting
alternative. In this work we have developed a component for the simulator Ns-2 (Network Simulator)
which allows the simulation of the communications in a NoC-based heterogeneous system.
We present a simple model of a NoC-based heterogeneous system in which a set of tasks is executed
concurrently by different computation nodes (which represent resources of the NoC). Each
computation node sends data requests to other computation nodes. The time between two successive
data requests in the same computation node is known as Tproc. When a computation node receives a
data request, it sends a response packet with load data after a time Tresp.
Using this model as a basis, we develop a component for Ns-2, named NocprocAgent. The agents in
Ns-2 represent active nodes that generate traffic in the network. The special characteristics of
NocprocAgent, which make it different from other agents, are:
1) The destination of the requests is randomly selected.
2) The requests are sent at random instants of time (Tproc.).
3) The responses are sent at random times after the request is received (Tresp).
Using the developed component, we showed its application in real experiments by simulating a 5×5
mesh (25 computation nodes). Varying the mean and the standard deviation of Tproc and Tresp, and the
buffer size of the links we obtained the amount of dropped packets in the network, which is a
representative magnitude of the behaviour of the network.
We performed several simulations, varying the different parameters, and found that both the link
buffer size and Tproc had a great impact on the NoC performance.
With this work we show that it is possible to integrate new components that adapt a General
Computer Simulator to Networks on Chip, with a relatively low effort.
- 92 -DCIS 2004
A Context-Switch Based Checkpoint and Rollback Scheme
Portolan Michele, Régis LeveugleTIMA Laboratory 46, Avenue Félix Viallet - 38031 Grenoble Cedex – FRANCE
Email: [email protected]; [email protected]
THIS article presents a new proposal for realising a Checkpoint and Rollback (C&R) scheme in
single-processor SoCs, based on the non-orthodox reuse of the context-switch (CS) function, heart
of all multitasking systems. The idea comes from a comparison between the two methods: both are
based upon the restoring of a previously saved state (the checkpoint for C&R, the context for CS).
Anyway CS does not save memory state, so a successful recovery is subjected to certain restrictions
that are pointed out and analysed in the article. In particular three validity criteria are derived and
detailed, presenting possible application conditions.
For the experimental part two practical implementation possibilities are analysed: the direct reuse of
context-switch function and the use of non-local jumps ANSI C functions “setjmp” and “longjmp”.
Preliminary evaluation measures, reported in Table 1, showed that the latter would yield better
performances and be easier to apply as a general method, so it has been chosen to be implemented in a
prototype.
The experimental platform, based on a SPARC V8 VHDL-IP processor, called Leon and developed
by Gaisler Research, turning eCos, the embedded operating system form Red Hat. Overhead
measures, reported in Table 2, have been done on TSIM, the time-accurate Leon simulator, always
from Gailser. A real prototype using a complex development board based on a Xilinx Virtex-II Pro
FPGA is under development.
The results are encouraging, confirming the good estimations. Good expectations are also arisen from
the prototype, whose realisation is proceeding swiftly and will enable complete and accurate measures
in real environments once completed.
Overheads in normal operation(without faults)
Implementation Runtime(%)
C-Switch(%)
Memory(bytes)
Rollback timeafter fault
detection (µs)Without
long jumps 0 0 80 59.375With
long jumps 0.006 6,25 128 3.125
T (µs)Runtime
(%)C-Switch
(%)Memory(bytes)Implementation
withlong jumps 10 0.02 20 128
Table 1 Summary of overhead evaluations (with a scheduler allocating time slots of 50 ms per thread)
Table 2 Overhead Measures
- 93 -DCIS 2004
Realistic 3D information is needed in several application fields. In robotics for example, an
autonomous robot roving in an unknown field must, to situate itself, discover the morphology of his
path, identify and estimate objects shape and dimension in his surrounding in manner to have the best
trajectory.
In medicine, the observation of the human body interior with tools providing additional information
about 3D shape will help physicians to sharpen their diagnostics.
The paper describes the development of an original integrated 3D image sensor. It is a chip scale
component based on active stereovision to reconstruct a realistic 3D representation of an object or a
scene. It has wireless communication ability to transmit its results and to be reconfigured “Over The
Air”. It is dedicated to be used in hazardous and unreachable spots.
The novel approach here is the miniaturization and the chip level integration of several parts, such
as a light emitting component, an image sensor, a HF circuit and a standard digital signal processing
circuit which are not technologically compatible, in the same package.
Cyclope: An Integrated Real-time 3D Image Sensor T. Graba, B. Granado, O. Romain, T. Ea, A. Pinna and P.Garda
Laboratoire des Instruments et Systèmes d’Ile de France (LISIF), Université Pierre et Marie Curie (UPMC)
4 Place Jussieu, BC 252, 75252 Paris Cedex 05, France Institut Supérieur d’Electronique de Paris (ISEP)
28 Rue Notre Dame des Champs, 75006 Paris tarik.graba | bertrand.granado | olivier.romain | andrea.pinna | [email protected]
- 94 -DCIS 2004
Session 4c
Analog Cmos Design Wednesday nov. 24 16h15 17h45, St Emilion Room
Chairs
Alain Fabre (E.N.S.E.I.R.Bordeaux)Jose M. de la Rosa (IMSE-CNM)
High sensitivity and Wide Bandwidth CMOS Transimpedance Amplifier for Optical Receiver Circuit
M. B. Guermaz 1, L. Bouzerara 1, H. Escid 2, and M. T. Belaroussi1
1 Centre de Développement des Technologies Avancées, Microelectronics and Nanotechnologies Division, Cité 20Août 1956, BP.17, Baba Hassen, 16303, Algiers,
ALGERIA 2 Université des Science et de Technologies Houari Boumediene, Systems Engineering
Laboratory, BP.32, Bab Ezzouar 16111, Algiers, ALGERIA [email protected], [email protected]
his paper describes and analyzes a low noise and high bandwidth transimpedance
amplifier featuring a large dynamic range. The designed amplifier is configured on three
identical stages that use an active load compensated by an active resistor to improve the
stability performance of the amplifier. This topology displays a transimpedance gain of
Ωk150 , which is necessary to obtain a high sensitivity of –32dBm. This structure operates at
5V power supply voltage, exhibits a gain bandwidth product of ΩTHz18 and a low noise
level of about HzpA94,0 . This transimpedance amplifier can reach a transmission speed of
sMb240 for a photocurrent of Aµ5,0 . A transmission speed of sMb622 can be achieved by
using a connection with optical fiber containing four channels and this for a photocurrent of
Aµ5,9 . The predicted performances are verified by simulations using PSPICE tool with
0.8µm CMOS AMS parameters. The stability problems that occurred in such a kind of
amplifier, has been solved by using an active resistor at the level of the active load of a stage.
The compensation technique has improved the phase margin of the designed amplifier.
The proposed transimpedance topology presents good performances in terms of noise and
bandwidth features.
The obtained performances fulfill the expected specifications such as a considerable gain, a
very high gain bandwidth product, a good dynamic range and more particularly a very low
noise level at the input, required in most of the communication standards. The main advantage
of the designed architecture resides at the level of its very low noise, thus giving a better
sensitivity in reception combined with a large bandwidth, which makes more feasible to
achieve a higher transmission speed.
T
- 96 -DCIS 2004
1.5V Square-Root Domain Magnitude Locked Loop
Carlos A. De La Cruz-Blas, Antonio López-Martín, and Alfonso Carlosena
Dept. of Electrical and Electronic Eng., Public University of Navarra
Campus Arrosadía, E--31006 Pamplona (Spain)
e-mail: carlos.aristoteles[antonio.lopez][carlosen]@unavarra.es
I N this paper, a current-mode magnitude locked loop based on CMOS companding techniques is
presented. The nonlinear transconductors that form the companding systems are based on the
nonlinear behavior of class-AB transconductors. This novel approach is an alternative one respect to
the conventional technique based on single MOS translinear loops, leading to more compact and
simpler implementations. The circuits are able to operate with very low voltage supply (as low as
V_GS+2V_DSsat). Both numeric and measurement results are provided to demonstrate the circuits and
the technique proposed.
- 97 -DCIS 2004
High-Speed High-precision Analog Rank Order Filter with O(n) complexity in
CMOS TechnologyR. G. Carvajal, J. Ramirez-Angulo, G.O.Ducoudray, and A. López-Martin
Klipsch School of Electrical Eng., New Mexico State University, Las Cruces NM (USA)
A new scheme for analog rank order filtering based on analog buffers is presented. This scheme is
characterized by high-speed, high-precision and simple circuit architectures. The overall architecture
exhibits linear complexity with number of inputs (O(n)) at the rate of one buffer per input. Rank is
easily programmable with the tail current source for all rank order values from the Max to the Min
case and its precision does not depend on the accuracy of the current copy. Simulation as well as
experimental results are presented that verify functionality and accuracy of the proposed circuit.
IbIb
V1
MN5
M11 M12
M13 M14
M15
MN3
MN2
MN4
Vout
CL
I L
Voltage buffer 1
(N-k+1)*Ibias
Ibias Ibias
MN1VN
Voltage buffer N
Figure : Proposed high-speed high-precision analog rank order filter
a) b)Figure 8. Experimental transient response of the rank order filter with four inputs, k varied from k=4 to k=3, and buffers withhigh open-loop gain: a) k=4, b) k=3,
- 98 -DCIS 2004
N this paper simulations and design considerations of a seventh order low-pass elliptic filter are
presented. The filter has the option to provide high frequency boost to correct the possible at-
tenuation in the communication channel. It has a cutoff frequency of 34 MHz with a ripple in the pass-
band less than 1 dB and an attenuation in the rejection band up to 65 dB (without boosting). It is able
to provide a boost of 12 dB. Its noise is below Hz
nV56 and has a IM3 of 60 dB for two signals of
amplitudes A=70mV and frequencies 30-31 MHz . It has been simulated in a 0.18 m process and
consume 485 mW for a 1.8 V power supply.
The filter uses the Gm-C technique. To make the transconductor we have used a technique based on
source degeneration to improve the linearity. To evaluate the characteristics of the filter at the system
level we have used a program which allows the estimation of characteristics like noise and distortion
in a fast way. To evaluate the distortion a method based in the Volterra series has been used. This fact
allows the evaluation of the distortion in all the interesting frequencies, in a short time, with no need of
transient analysis. Figure 1 shows the conceptual schematic of the filter where GB are extra
transconductor which allows the correction of the attenuation provided by the channel at high
frecuencies.
Figure1 : Conceptual schematic of the filter
A Seventh Order Elliptic CMOS Continuous Time Gm-C Filter for PLC applications
Juan F. Fernández-Bootello, Manuel Delgado-Restituto, and Ángel Rodríguez-Vázquez
Instituto de Microelectrónica de Sevilla, Centro Nacional de Microelectrónica Avda. Reina Mercedes s/n, Campus Universidad de Sevilla, E-41012 Sevilla (Spain)
Emails: bootello, mandel, [email protected]; phone +34 955056666
I
- 99 -DCIS 2004
Tunable Gm-C Biquadratic Filter Operating in Moderate Inversion
Jaime Ramirez-Angulo1, Chandrika Durbha1, Antonio J. López-Martín1,2, Ramón G. Carvajal1,3
1 Klipsch School of Electrical Eng., New Mexico State University, Las Cruces, NM, USA. 2 Dept. of Electrical and Electronic Eng., Public University of Navarra, Pamplona, Spain
3 Escuela Superior de Ingenieros, Universidad de Sevilla, Spain (e-mail: [email protected])
An integrated tunable Tow-Thomas Gm-C biquadratic filter is presented, with independently
adjustable frequency, gain, and quality factor in both low-pass and band-pass responses. The
transconductors employed operate in moderate inversion region, leading to an excellent tradeoff
between bandwidth and power dissipation, wide adjustment range of filter parameters, large dynamic
range, and low die area.
The filter has been fabricated in a 0.5- m CMOS technology. Measurement results of the
transconductor and the complete filter are presented. They are in good agreement with theoretical and
simulation results, and demonstrate that operation in moderate inversion region can lead to circuits
with very high linearity and tuning range. The authors believe this is the first time it is recognized that
operation in moderate inversion results in very low distortion levels. This opens potentially many
other applications of moderate inversion, of which the proposed filter is just one.
Figure 1. Measured frequency tuning of LPFoutput
Figure 2. Measured Q tuning of BPF output
- 100 -DCIS 2004
Fully-Differential CMOS Current Conveyor Operating in Moderate Inversion
Antonio J. López-Martín1,2, Jaime Ramirez-Angulo2, Chandrika Durbha2, Ramón G. Carvajal2,3
1 Dept. of Electrical and Electronic Eng., Public University of Navarra, Pamplona, Spain 2 Klipsch School of Electrical Eng., New Mexico State University, Las Cruces, NM, USA.
3 Escuela Superior de Ingenieros, Universidad de Sevilla, Spain (e-mail: [email protected])
Anovel fully-differential CMOS second-generation current conveyor (CCII) topology is presented.
The circuit operates in moderate inversion region, and features high linearity over a wide input
range. Current gain can be tuned in a wide range. These features are essential to extend the utilization
of CCII-based circuits to high-performance VLSI applications. The circuit also features very low input
impedance at the X terminal and low die area. It can be applied as a fully differential universal active
block in several circuit topologies like filters and oscillators.
The proposed circuit, shown in Figure 1, has been implemented in a 0.5- m CMOS technology and
their main performance characteristics have been measured. When the circuit is employed as a
transconductor, measurements show a Total Harmonic Distortion of -66.5 dB with differential input
swings equal to 77% of the 1.3-V supply voltage, transconductance tuning in two decades, and 1.7
mW of static power consumption.
IDIR
VY+
IINV
VCN
IB
M6A
M4A
M3AM2A
M1A
M5A
I1
IZ+
IX+
IDIRIINV
VCNM6B
M4B
M3B M2B
M1B
I2
Iout VY-M5B
IX-
IBIZ-
VX+VX-
Figure 1. FDCCII circuit
FDCCII
Y
X
+-
+-
+-
IIN-
IIN+
VCN
IOUT-
IOUT+
FDCCII
Y
X
+-
+-
+-
Vid
IOUT-
IOUT+
FDCCII
Y
X
+-
+-
+- VOUT-
VOUT+
R
RIIN-
IIN+
VCN
FDCCII
Y
X
+-
+-
+- VOUT-
VOUT+
R
R
R
(a) (b)
(c) (d)
Z Z
ZZ
R
+
_
Vid
+
_
Figure 2. Application examples(a) Current amplifier (b) Transconductor(c) Transresistor (d) Voltage amplifier
- 101 -DCIS 2004
Session 4d
Radiation Effects and EMC Wednesday nov. 24 16h15 17h45, Bordeaux Room
Chairs
Olivier Bonnaud (U. Rennes 1) Francesc Moll (U. Politècnica de Catalunya)
Analysis of Transient Fault Emulation Techniques in Platform FPGAs
AULT Tolerance (FT) has been a traditional requirement for safety-critical applications working in
harsh environments. Very deep submicron and nanometer technologies have increased notably
integrated circuits (ICs) sensitiveness to radiation. Soft errors are currently appearing into ICs working
at earth surface. Therefore, hardened circuits are currently required in many applications where Fault
Tolerance (FT) was not a requirement in the very near past. During the hardening process of a circuit,
fault tolerance evaluation is a key factor. In this sense, the use of platform FPGAs for the emulation of
single-event upset effects (SEU) is gaining attention in order to speed up the fault tolerance evaluation
process.
In this work, two techniques for the evaluation of FT with respect to SEU effects are described and
compared: a Fault-Mask-based and a Scan-Path-based architectures.
Both proposals make profit of the hardware resources, executing most of the tasks in the FPGA
instead of in the host, in order to minimise the bottleneck times in the communication between
software and hardware.
The main difference between both approaches is the fault injection strategy. The first one includes a
fault mask chain that is applied on the circuit when injection time is reached while the second solution
uses scan-path techniques to download the state of the circuit at the fault injection time.
Both techniques are analyzed and compared with respect to area overhead and execution time required
by the emulation process for a benchmark circuit. Experiments performed show that emulation
techniques can obtain the results in seconds while similar experiments can take hours using simulation
based techniques. Furthermore, Mask Scan Injection Technique is better in terms of fault emulation
time, when testbench cycle number is smaller than the number of flip-flops in the circuit (data-path
applications). On the other hand, State Scan Injection Technique is interesting when evaluating the
fault tolerance of circuits with short number of flip-flops and large number of test-bench cycles
(control applications).
Results obtained prove that this system is a cost-effective solution for transient fault evaluation.
M. Portela-García, C. López-Ongil, M. García-Valderas, L. Entrena-Arrontes
Microelectronics Group. Electronic Technology Department University Carlos III of Madrid, Spain.
mportela, celia, mgvalder, [email protected]
F
- 103 -DCIS 2004
Analysis of Input and Feedback Capacitances Effect on Low Noise Preamplifier Performance for X-rays Silicon
Strip Detectors
T. Noulis*, S. Siskos*, G. Sarrabayrouse** *Electronics Laboratory of Physics Department, Aristotle University of Thessaloniki, Greece
**LAAS-CNRS, Toulouse, France [email protected], [email protected], [email protected]
AN analysis of a charge sensitive preamplifier (CSA) noise behavior of a low energy X-rays
silicon strip detector for space applications is presented. Design criteria of CSA noise
optimization are examined in relation to total input stage capacitance and specifically to detector and
feedback capacitance and parasitic capacitances of the input and reset MOSFET. A differentiation of
the total output noise, charge–discharge time and gain associated with detector capacitance is
demonstrated. The effect of input and reset MOS parasitic capacitances on stability and noise
contribution is also examined in a CSA configuration with no feedback capacitance. The preamplifier
was designed in 0.8µm DMILL process and analysis is supported by simulation results.
The output signal of the CSA with zero feedback capacitance is shown in Fig.1. Preamplifier
operates, but not properly, since an instability of the output discharge voltage level and a strong
tendency of the amplifier to oscillate are observed. The design of a low noise preamplifier with no
feedback capacitance can be achieved by optimizing the dimensions of input and reset MOS (Fig.2).
Increase of the input MOS M1 dimensions results higher Cgd, which is considered to be parallel to
feedback path. Decrease of reset MOS implies a reduction of its gate and substrate parasitic
capacitances, and therefore attenuation of oscillation tendency.
Preliminary results of a work that aims at the design of a low noise preamplifier with no feedback
capacitance are presented.
Fig.1. CSA output signal for zero Cf and Cd=2 pF. Fig.2. CSA output signal for large input MOS and small reset MOS (Cf=0pF and Cd=2p).
- 104 -DCIS 2004
A hardware approach for SEU immunity verification using Xilinx FPGA’s
M. Aguirre1, J.N. Tombs1, F. Muñoz1, V. Baena1, A. Torralba1, L.G.Franquelo1, A. Fernández-León2,F. Tortosa-López2 and D. González-Gutiérrez2
1Escuela Superior de Ingenieros Universidad de Sevilla. Camino de los Descubrimientos s/n 41092 Sevilla (SPAIN)aguirre,jon,fmunoz,baena,[email protected]
2Data Systems Division. ESTEC/TOS-ED European Space Agency.
Noordwijk (THE NETHERLANDS)
ummary. Radiation impact on microelectronic structures can produce a variety of effects, of
which many can lead to incorrect values being stored in the memory cells of a digital design
during execution time. When radiation produces unexpected bit-flips they are classified as
single event upsets (SEUs) and critical. Designs must mitigate their effect through the use of special
cell libraries at the physical level, redundant logic design and voting logic in the memory cells and by
the design of robust deadlock-free state machines at the architectural level. The verification of SEU
tolerance in a VLSI design netlist is currently an expensive, difficult, and time consuming task. This
paper presents FT-UNSHADES, a custom circuit emulator that permits the insertion, motorization and
analysis of bit-flips in digital designs. The system requires simple, full automatic and non-intrusive
preparations to the design to be tested, whilst the use of state of the art FPGAs permits that the circuit
emulation is performed at full hardware speed in a highly controlled manner. The SEU insertion
strategy allows selective provocation of bit-flips in any desired flip-flop at any desired time during a
given test, and allows a detailed analysis of the fault-tolerance properties against soft errors of the
circuit itself. The current system design can analyse large designs of up to almost 3 million system
gates whilst inserting over 80K SEUs per hour in a test of 2 million test vectors.
S
FT-UNSHADES system is going to be implemented in the design flow of the microelectronic
section of the European Space Agency as a test platform that should reduce the design-fabrication-test
cycles of space application VLSI designs.
FT-UNSHADES System
- 105 -DCIS 2004
PACE electronics is exposed to heavy ions and other ionizing particles that can produce high
densities of electron-hole pairs in a semiconductor. The flow of electrons taking place inside the
devices may lead to temporary voltage spikes at internal circuit nodes termed Single-Event
Transients (SETs). Simulating this environment is mandatory for testing ICs and essential in order to
evaluate, understand, and mitigate their sensitivity. Evaluating effects of SETs in Analog-to-Digital
Converters (ADC) and their impact at system level is a complex task due to the mixed signal nature of
these architectures and the various errors signatures that may be encountered [1]. An original
methodology is presented for characterizing SEE impact on ADCs performance parameters, based on
the use of pulsed laser system (Fig. 1). The spatial and temporal resolution of the pulsed laser beam
(Fig. 2) is used for identifying the SEU mechanisms in two half-flash ADCs.
_________________________________
[1] W.F. Heidergott, R. Ladbury, P.W. Marshall, S. Buchner, A.B. Campbell, R.A. Reed, J. Hockmuth, N. Kha, C. Hammond, C. Seidleck,A. Assad, “Complex SEU Signatures in High-Speed Analog-to-Digital, Conversion”, IEEE Trans. Nucl. Sci., vol. 48, no. 6, pp. 1828-1832, 2001.
Radiation Hardness Assessment of an ADC for Space Application using a Laser Test Equipment.
V. Pouget, P. Fouillat, D. Lewis, F. Darracq
IXL, UMR CNRS 5818, Université Bordeaux 1, 33405 Talence, France [email protected]
S
scan window L=30ns L=380ns L=480ns
Fig. 1 : Experimental set-up for ICs testing with a pulsed laser.
Fig. 2 : Scan window on two comparators of an ADC and corresponding laser-induced error maps for three laser pulse delays.
Pump laser 10W 532nm
Pulsed laser Ti:Sa
Pulsediagnosis
Pulse picker
100x CCD
WhiteLight
3axis
1.3 mlaser diode
Powermeter
DUT
Oscilloscope
Function generators
Power supplies
4 axis controller
GPIB
RS232
Video board
DAQ board
Parallel portTestboard
Lock-in amplifier
Pulse energycontrol
Pattern generator
Si
Sr
StSf
PC
- 106 -DCIS 2004
Exploitation of the ICEM Model for Jitter Analysis in an Integrated PLL
Jean-Luc Levant, Mohamed Ramdani, Richard Perdriau and M’Hamed Drissi
T HIS paper deals with the use of ICEM (Integrated Circuit Electromagnetic Model) in the EMC
performance assessment of a high-density integrated circuit. This model has already been used for
IC modeling with good accuracy. This work demonstrates how the analysis of IC performance issues,
such as PLL jitter or ADC resolution loss, is made possible thanks to this approach. In particular, the
problem of PLL jitter in a high-density ASIC is solved thanks to this methodology.
- 107 -DCIS 2004
An IP-Based Chip-Level EMC Modeling and Prediction Methodology
Richard Perdriau*, Mohamed Ramdani* and Jean-Luc Levant**
*ESEO - 4, rue Merlet-de-la-Boulaye - BP 30926 - 49009 Angers Cedex 01 - France**ATMEL - La Chantrerie - Route de Gachet - 44300 Nantes - France
Corresponding author : Richard PerdriauTel. (33/0) 2 41 86 67 03 - E-mail : [email protected]
Electromagnetic compatibility (EMC) compliance is a topical demand in any electronic system, often represent-ing a substantial part of the design effort. The sooner EMC rules are taken into account in the design phase, thesooner the target product can be released on the market, lowering engineering costs at the same time. However,with ever increasing complexity in integrated circuits, and thus ever higher emission and susceptibility levels, itcan be seen that reducing these levels in the IC design phase itself then makes it much easier to ensure system-levelEMC compliance.For that purpose, the recent ICEM (Integrated Circuit Electromagnetic Model) proposal, under the IEC (Inter-national Electrotechnical Commission) 62014-3 reference, allows the designer to predict conducted and radiatedemission of an integrated circuit within its environment (PCB with ground planes, decoupling networks, connec-tors). The IC part of this model contains passive elements (package, bonding, metal and MOS capacitances) aswell as an equivalent current generator representing internal activity.This current generator can be obtained by simulating the whole transistor netlist of the IC; however, this methodleads to huge simulation times, making it unusable in EMC expertise, in which the influence of "tunable" parame-ters (package, decoupling capacitors) on emission levels has to be asserted as fast as possible. Moreover, since thedesign of complex ICs is based mostly on reusable blocks (for example microprocessor cores and memory blocks),either proprietary or from third-party intellectual property (IP), a similar ICEM-based reuse methodology for EMCprototyping can be proposed.The VHDL-AMS language, enabling the description of event-driven, mixed-signal behavioral models, turns to bewell suited to this methodology, thanks to its upward compatibility with "digital" VHDL and its standardization.For this purpose, each digital block is associated with a VHDL-AMS block using the same inputs; the VHDL-AMSmodel computes the dynamic current generated on the power supply rail as a function of these inputs, allowing toevaluate activity-dependent emission. These models can then be assembled in order to obtain the whole dynamicsupply current of the integrated circuit, along with functional simulations. Full-chip EMC virtual prototyping isthus made possible, including not only building blocks, but also inputs and outputs (I/Os).For evaluation needs, VHDL-AMS ICEM models have been written for an 8-bit ATMEL microcontroller, includingthe core, the embedded SRAM block and the I/O buffers. Except the buffers, the corresponding models relyon event-driven, piece-wise linear (PWL) approximations of supply current waveforms obtained from transistor-level electrical simulations. For example, SRAM dynamic supply currents can be easily modeled as a function ofread/write modes, addressing and rise/fall times (control signals). Moreover, VHDL-AMS allows I/O buffers to bedescribed without revealing technological data, while including their mutual influence with the IC core; this fillsin the gap of the IBIS (Input/output Buffer Information Specification) and IMIC (Input/output interface Modelfor Integrated Circuits) models.Comparisons between simulations and measurements are really promising and demonstrate the validity of thisapproach for virtual EMC prototyping.
- 108 -DCIS 2004
Session 5a
Devices for High Frequency Circuits Thursday nov. 25 8h30 10h00, Sauternes Room
Chairs
Joaquin Portilla (U. del País Vasco / Euskal Herriko U.) Jean-Marie Paillot (U. de Poitiers)
Integrated MOS Varactors in Accumulation Mode for RF Applications
B. Gonzalez(1), J. Garcia(1), I. Gutierrez(2), N. Sainz(2), M. Marrero-Martin(1), A. Goni-Iturri(1),and A. Hernandez(1)
(1) Instituto Universitario de Microelectrónica Aplicada (IUMA), and Departamento Ing. Electrónica y Automática. Universidad de Las Palmas G. C. Spain, [email protected](2) Escuela de Ingenieros. Universidad de Navarra. TECNUN Spain, [email protected]
N this work, integrated MOS varactors for RF applications in accumulation mode have been
designed, fabricated and measured. The resulting varactor is achieved by changing the operation
mode from depletion to accumulation, whereby the capacitance rises from a minimum value to a
maximum value.
All the integrated varactors have been fabricated in the AMS SiGe 0.8 µm standard process
technology, which are surrounded by measurement structures, the guard ring, in order to use the
Cascade ACP40 GSG microprobes (Figure 1). Their characterization has been carried out with a
measurement system based on the HP8719ES Vector Network Analyzer.
We report studies of the capacitance’s scalability against the geometry of the structure (gate length
and gate width) and, more specifically, against the number and the arrangement of the basic cells. The
impact of metallization is also considered. All our proposed structures present a wide tuning range
over the 35%.
We demonstrate the capacitance of accumulation-mode MOS varactors for RF applications is
scalable with the area occupied. Thus, an integrated varactor library based on these devices can be
easily implemented. Tuning ranges higher than 57% have been obtained with short gate voltage
variations, 2 V, keeping the quality factor over 10. Finally, the performance of the varactors
implemented can be optimized increasing the gate length, keeping the gate width constant.
Figure 1. Microphotograph of the integrated
MOS varactor M1, with guard-rings
I
- 110 -DCIS 2004
Embedded Passive Design for High Speed Circuits
Geneviève DUCHAMP1, Yves OUSTEN1, Bruno LEVRIER1, Philippe KERTESZ2,Steven HEYTENS3
1IXL UMR 5818 - 351 cours de la Libération - 33405 Talence Cedex- 2 THALES Airborne Systems - Centre Nungesser - 2, avenue Gay-Lussac- 78851 Elancourt
3 Rogers NV –Afrikalaan 188 , 9000 Gent – Belgique [email protected] - [email protected]
[email protected] - [email protected]
urrently, the majority of passive devices employed in electronic systems are discrete
components. Small discrete devices dominate the area of PCB mounted process in a typical
electronic product. For example, a cellular phone may consist of only about 20 integrated
circuits (IC) compared to 300-400 passive [1]. Thus, passive components have substantial
influence on system cost, size, and reliability. In order to meet the next generation electronic
packages (smaller, faster, cheaper, and more reliable), alternatives to discrete passives are
necessary. In this framework the present paper proposes a guideline to design embedded
capacitors particularly for high frequency applications and some examples are proposed
A design rule is evidenced: for a squared electrode structure (either simple or multi layer) and
a given technology related to the layer thickness h, the best compromise between the capacitor
value C and the maximum operating frequency range f is given by :
h16fC
202
⋅≤ c.. ε
Figure 1. One layer high frequency ceramic capacitor (magnification x10)
Maximum operating frequency value of 3.4 GHz [1] Yang Rao C.P. Wong, “Electrical and Mechanical Modeling of Embedded Capacitors” , IMAPS 99
C
- 111 -DCIS 2004
Analysis and Applications of MOS Resistive Cells
M.T. Sanz, S. Celma, B. Calvo and J.P. Alegre Grupo de Diseño Electrónico, Facultad de Ciencias, Universidad de Zaragoza, Spain.
e-mails: materesa, scelma, becalvo@ unizar.es
T HE aim of this work is to carry out a comparative study of two ideally linear configurations: a
MOS Resistive Circuit (MRC), widely employed in the literature, and a MOS Current Divider
(MCD) based on a linear current division principle. The suitability of some MOS models for distortion
simulation purposes is discussed. The general charge-sheet model is used to perform simulations in the
MATLAB environment. The harmonic distortion terms are calculated by means of the FFT and the
influence of mismatches on the linearity of both cells is estimated. Approximate strong models do not
perform well in distortion analysis, so it is necessary to resort to the general charge-sheet model.
Simulations carried out in the MATLAB environment show that harmonic distortion components in
the MOS Resistive Circuit (MRC) are far from being cancelled out, whereas the MOS Current Divider
(MCD) is ideally completely linear. It is also shown that, although somewhat more sensitive to
mismatching, the MCD has lower distortion levels and thus is more suitable for high-linearity
applications. As an application of MOS resistive cells, Programmable Gain Amplifiers (PGAs) based
on the MRC and the MCD are introduced and their performance in terms of linearity and accuracy is
studied and compared. Figure 1 shows the variation of HD2 and HD3 with the digital word A(3) for a
500mVp-p output signal in both PGAs. An outstanding improvement in linearity is achieved when
employing the MCD. Further simulations show the influence of mismatching on linearity in both
systems.
0 1 2 3 4 5 6 7-160
-140
-120
-100
-80
-60
-40
digital word
harm
onic
com
pone
nt(d
B)
Fig. 10. Variation of HD2 (--) and HD3 (-) with the digital word forthe (*) MRC-PGA and the (o) MCD-PGA for a 500mVp-p output signal.
- 112 -DCIS 2004
Ladder-type FBAR Filter Synthesis Methodology
A. Shirakawa, J-M. Pham, P. Jarry, E. Kerherve, E. Hanna
IXL Microelectronics Laboratory – UMR 5818 CNRS – Bordeaux I University, 351, Cours de la Libération – 33405 Talence Cedex – France
Phone : +33 (0) 540 002 611 / Fax : +33(0) 556 371 545 / e-mail : [email protected]
THE RF filters and duplexers, essential elements in the radio front-ends, are traditionally based on
dielectric electromagnetic resonators (ceramic materials) or surface acoustic wave (SAW) devices.
SAW filters present good selectivity and very small size, but are limited in frequency (up to 3 GHz)
and have high insertion loss at high power levels (over 1 W). The dielectric filters are capable to
handle high power levels and present good selectivity. However, their application in mobile front-ends
is still limited by their large dimensions.
Film Bulk Acoustic Resonator (FBAR) filters, based on the Bulk Acoustic Wave (BAW)
technology are expected to replace the traditional RF filters technologies. The FBAR filters offer
unique advantages since they present good selectivity (Q up to 1000), can handle high power levels up
to several Watts (up to 3 W) and have reduced dimensions1. Besides, FBAR devices can be
manufactured based on the same VLSI-CMOS basis and be directly integrated above RF active
circuits, making possible the design of the full RF integrated front-end at very competitive costs
(above-IC technology).
This work proposes a synthesis methodology based on the inline bi-mode technique to design ladder
type FBAR filters. Until now, the design methodologies proposed were based on ladder crystal filters.
They are focused on placing the transmission zeroes and need an intensive optimization work to place
the poles of the desired transfer function. The methodology proposed permits not only the placement
of the transmission zeroes, but also the determination of the exact location of transmission poles.
Finally, a design example of a third degree filter for the 2 GHz frequency band is addressed.
1 R. Aigner et al. “Bulk-Acoustic-Wave Filters: Performance Optimization and Volume Manufacturing”. IEEEMTT-S 2003. pp. 2001-2004.
- 113 -DCIS 2004
Study of the Proximity Effect in High Q Inductors with CMOS 0.18 m Technology
I. Cendoya*, N. Sainz*, J. Mendizabal**, R. Berenguer**, U. Alvarado**, A. García-Alonso**
* Escuela Superior de Ingenieros de San Sebastián (TECNUN) – Universidad de Navarra, Spain. ([email protected])
** Centro de Estudios e Investigaciones Técnicas de Guipúzcoa (CEIT), Spain. ([email protected])
HE objective of this study is to acquire high quality inductors designed in CMOS 0.18 m. One
of the pernicious effects for the quality factor of an inductor is the proximity effect. Proximity
effect is due to the magnetic field generated by the own inductor and induces parasitic currents in the
tracks. As a consequence of this study some rules are reported. High quality factors are obtained
(between 8.1 and 12.6)
Focusing in the problem of the proximity effect, some simple design rules to obtain good Q inductors
have been reported. These rules take into account the space between tracks and the center hollow of
the inductor:
1) In relation with the center hollow of an inductor, it has been proved that the actual important
parameter is the ratio between external and internal radius. An inductor with a ratio of 1.75
will be nearly maximized in quality, occupying less area than a bigger one with a little
better Q.
2) The second conclusion is possible to reach is that the improvement of the Q is joined to
minimize space between tracks.
T
- 114 -DCIS 2004
Session 5b
Low Power / Low Voltage : analog circuits (1)
Thursday nov. 25 8h30 10h00, Pyla Room
Chairs
Thomas Zimmer (U. Bordeaux 1) Leopoldo García Franquelo (U. de Sevilla)
Wireless Battery Charger Chip for Smart-Card
Applications
Franz Xaver Arbinger, Peter Spies, Günter Rohmer.
Power Efficient Systems Department, Fraunhofer Institut Integrierte Schaltungen, Erlangen,
Germany, [email protected]
A chip for inductive battery charging is presented, which needs no external components except an
antenna to capture the energy from an electromagnetic field.
The integrated system blocks are a front-end to limit and rectify the induced alternating voltage and a
charge regulator with three control loops for the current, the voltage and the temperature. The external
antenna forms a resonance circuit with the on-chip capacitor. The resonance frequency of the front end
is 13.56 MHz, so it is compatible to the well known smart-card standard. In the electromagnetic field
of commercial reader systems the chip produces an output current to charge a lithium battery with the
mandatory constant-current-constant-voltage (cccv) charge profile.
This architecture is implemented to charge lithium cells at a current of 4 mA up to a cell voltage of 4.2
V. The target application are high-end smart-cards with secondary batteries.
The chip, fabricated in a 0.8 µm BICMOS-technology, includes two contacts for the antenna and two
for the battery. The chip size is 1.5 mm x 2.5 mm. Also present are additional pads for testing the chip
and for using a dc-voltage source for charging. The measurements of the chip show good results and
the whole function has been evaluated by charging the lithium accumulators. The operating current of
the IC is approximately 1 mA.
- 116 -DCIS 2004
Low-Power High-Slew-Rate Rail-to-Rail CMOS Analog Buffer
N n this paper a low-power rail-to-rail CMOS analog buffer is presented. The circuit is based on a
class AB input stage made up of two complementary differential pairs, while a simple additional
circuit allows rail-to-rail operation at the output terminal. Besides, the input capacitance of the circuit
can be reduced by scaling the size of the input devices, decreasing loading effects on the nodes to be
tested or on preceding stages. The class AB capability of the proposed circuit combines a low static
power consumption in quiescent operating conditions and a high drive capability in the dynamic
operation, resulting very suitable for applications with large capacitive loads.
The buffer has been designed in a 0.35- m CMOS technology to operate with a ±1.5 V dual supply.
Simulated results are provided in order to demonstrate the proper operation of the proposed circuit. A
rail-to-rail signal swing is achieved and a THD lower than –44 dB is obtained for a 2.4-Vpp 100-kHz
input sinewave signal, whereas the input capacitance is lower than 32 fF.
Ramón G. Carvajal (1), Juan M. Carrillo (2), J. Francisco Duque-Carrillo (2),and Antonio Torralba (1)
(1) Department of Electronic Engineering, School of Engineering, University of Sevilla Camino de los Descubrimientos, 41092 Sevilla, Spain
E-mail: carvajal,[email protected]
(2) Department of Electronics and Elec. Eng., University of Extremadura Avda. de Elvas s/n, 06071 Badajoz, Spain
E-mail: jmcarcal,[email protected]
I
- 117 -DCIS 2004
1.5V Current-Mode CMOS True RMS-DC Converter Based on Class-AB Transconductors
Carlos A. De La Cruz-Blas, Antonio López-Martín, Alfonso Carlosena, and Jaime Ramírez-Angulo1
Dept. of Electrical and Electronic Eng., Public University of Navarra
Campus Arrosadía, E-31006 Pamplona (Spain)
1 Klipsch School of Electrical and Computer Engineering, New
Mexico State University
( Las Cruces, New Mexico), USA
e-mail: carlos.aristoteles[antonio.lopez][carlosen]@unavarra.es
I N this paper, a current-mode CMOS RMS-DC converter is presented. The basic building blocks
are based on a novel approach to design current--mode computational cells. In such an approach, the
large signal behavior V vs I of class-AB transconductors is conveniently exploited leading to a very
regular and compact implementation. A proper biasing scheme in such transconductors allows a very
low voltage operation with supply voltage as low as V_GS+2V_DSsat. Measurement results from a
practical prototype are presented in order to demonstrate the technique here proposed.
- 118 -DCIS 2004
New low-voltage high performance WTA circuits based on flipped voltage followers
J. Ramírez-Angulo1, G. Ducoudray-Acevedo1, R. G. Carvajal2 and A. Lopez-Martin3
1Klipsch School of Electrical and Computer Engineering, New Mexico State University, 2Escuela Superior de Ingenieros, Universidad de Sevilla, 3 Universidad Publica de Navarra
A new low-voltage CMOS WTA circuit is presented. The proposed circuit exhibits linear complexity
with the number of inputs and it is based on a modified version of the common source scheme. In this
case each input follower is enhanced by local shunt feedback to increase its gain and to reduce its
output impedance. Simulations demonstrate the potential of the circuit to operate at very high speed,
with high precision and with a supply voltage close to a transistor’s threshold voltage. Experimental
verification of the circuit using a 0.5 m CMOS technology is also provided.
V1M A1Vout
(a)c
MB1
MC1
V2M A2
MB2
MC2
VnMAn
MBn
MCnMAout
MBout
Ib
MB
(b)
MD1
I1
MD2
I2
MDn
In MA1
Iout
c
MB1
MC1
M A2
MB2
MC2
VcmMAn
MBn
MCn
MAout
Fig. 1 Proposed circuits : a)Voltage mode FVF based MIN circuit (b) Current-mode version
- 119 -DCIS 2004
New Low-Voltage Fully Programmable CMOS Triangle/Trapezoidal Function
Generator Circuit. Meghraj Kachare1,Jaime Ramírez-Angulo1, Antonio J. López-Martín2, and Ramón G. Carvajal3
1 Klipsch School of Electrical and Computer Eng., New Mexico State University, Las Cruces, NM (USA)
2 Dept. of Electrical and Electronic Engineering, Public University of Navarra, Pamplona (Spain) 3 Dpto. de Ingenieria Electronica, Escuela Superior de Ingenieros, Universidad de Sevilla (Spain)
A versatile low-voltage CMOS circuit with a triangular/trapezoidal transconductance characteristic
and independently programmable height, slope, and horizontal position is presented. Simulation
results using Cadence DFW-II that verify the functionality of the circuit with 1.5V supplies are
presented. A chip prototype has been fabricated in a 0.5 µm technology and experimentally verified.
The circuit can find utilization for the implementation of membership functions in analog and mixed-
signal neuro-fuzzy systems, for piecewise linear approximation and for the implementation of high
resolution, high speed folding A to D converters.
- 120 -DCIS 2004
Low-Voltage Micropower Integrated CMOS Log Domain Filter
Antonio J. López-Martín, Carlos A. De La Cruz Blas and Alfonso CarlosenaDept. of Electrical and Electronic Engineering, Public University of Navarra, Pamplona, Spain
(e-mail: [email protected])
A first-order balanced log-domain low-pass filter (Figure 1) operating in class AB is presented. It
is based on the utilization of floating-gate MOS (FGMOS) transistors biased in weak inversion to
implement the required nonlinear internal processing. A simple circuit topology is obtained due to the
use of FGMOS devices, without the penalty in terms of post-fabrication removal of initial charge often
encountered in former FGMOS circuits.
A prototype fabricated in a 0.8- m CMOS technology (Figure 2) and using a single supply voltage
of 1.2 V achieves three decades of frequency tuning for a static power dissipation of less than 5 W,
and occupies an active area of 0.1 mm2. The 1%-THD dynamic range at 1 kHz is 75 dB. To the
authors’ knowledge this is the first FGMOS log-domain filter reported operating in class AB, which
leads to a dynamic range significantly larger than in former FGMOS filter topologies. The filter is
readily cascadable, leading to higher-order topologies.
Iin+
IB2C
Iout+ Iout-
IB1
M5A M6A
M7A M8A
M1A M2A M3A M4AM9A M10A M11A
Iin-
IB2C
IB1
M5BM6B
M7BM8B
M1BM2BM3B
M4BM9BM10BM11B
Vc+ Vc-
Figure 1. Balanced FGMOS filter proposed
500 m
Figure 2. Microphotograph of the filter
- 121 -DCIS 2004
Session 5c
SOC & Analog Test Thursday nov. 25 8h30 10h00, Margaux Room
Chairs
Hervé Lapuyade (U. Bordeaux 1) José Luis Huertas (IMSE-CNM)
An Infrastructure and Application Specific Processorfor Testing Analogue and Mixed-Signal SoCs
Francisco X. Duarte, José Machado da Silva, José C. Alves, and José S. MatosUniversidade do Porto, Faculdade de Engenharia, and INESC Porto,
Rª Dr. Roberto Frias, 4200-465, Porto, Portugal, fduarte, jms, jca, [email protected]
CONVENTIONAL test approaches are unable to cope with the test requirements of tens or even
hundreds of cores, such as digital and analogue I/O interfaces, complex communication sub-
systems (including optical and radio-frequency circuits), power management, and multiple processors,
deeply embedded in complex systems. The IEEE P1500 Standard for Embedded Core Test 1
infrastructure is currently the main standard proposal for modular embedded design for testability
approach for digital and memory cores. However, the test of analogue and mixed-signal cores has not
been addressed.
This paper presents an infrastructure and methodology for testing analogue and mixed-signal
cores embedded in systems-on-chip. The solution proposed resorts to the reconfigurable block of the
system (e.g., field programmable gate arrays), which can be reused to implement an application
specific instruction-set processor to control and schedule on-chip test operations. This processor’s
architecture can be adjusted according to test needs and the space available for implementation.
Furthermore, it allows reducing the number of signals to be sourced by the tester, and the extension of
data to be transferred between the tester and the core under test, as well as test time by performing on-
chip pre-processing operations. The reconfigurable block can also be used to implement wrapper’s
digital cells that provide routing of system and test signals, as well as the implementation of the digital
test infrastructure.
A demonstration prototype is described which implements the testing of an ADC (analogue to
digital converter). The test processor was designed to accomplish a specific test method that allows
reducing the relevant ADC response data to only a few bytes. Besides the conventional operations
(load/store, arithmetic/logic operations, …) and loop control, the instruction-set comprises test specific
instructions to handle the IEEE 1149.1/4 port that controls the ADC wrapper, to handle the ADC's
specific control signals, and to accumulate ADC’s output data samples. A first-order Sigma-Delta
modulator is embedded in the processor to provide analogue test stimuli. After the test signatures are
uploaded to the tester, a polynomial-fitting algorithm is then used to compute the harmonics’
coefficients that characterize ADC’s non linearity.
1 IEEE P1500 Web Site, http://grouper.ieee.org/groups/1500
- 123 -DCIS 2004
Test Planning for Mixed-Signal SoCs and Analog BIST:a Case Study
Antonio Andrade, Jr.1, Érika Cota2, Marcelo Negreiros2, Luigi Carro1, Marcelo Lubaszewski3
Electrical Engineering Dept., Univ. Federal do R. Grande do Sul, Porto Alegre, Brazil, andradejr, [email protected]
Informatics Institute, Univ. Federal do R. Grande do Sul, Porto Alegre, Brazil,erika, [email protected]
Instituto de Microelec. de Sevilla, CNM, Sevilla, Spain, [email protected]
ANALOG BIST and SoC testing are two topics that have been extensively, but independently,
studied in the last few years. However, current mixed-signals systems require the combination of
these subjects to generate a cost-effective test planning for the whole SoC. This paper discusses the
impact on the global system testing time of using analog BIST based on digital reuse of available
embedded processors in the system. Some advantages of the proposed technique are the system test
time reduction, due to analog BIST, and obliviation of external mixed-signal test equipment
requirements, since the analog test response is performed by the reused processor.
Experimental results show that, as long as the BIST technique reduces the analog testing time, the
reuse of digital blocks to test analog signals is indeed a very efficient strategy, despite the test
serialization, as depicted in figure 1. In addition, better test results may be achieved if the number of
available reusable processor on-chip increases.Requirements in terms of extra test pins and area
overhead are evaluated in the test planning for mixed-signal SoCs. Power restrictions during test are
also considered, as there is a widespread use of SoCs in portable electronic devices, and battery-life of
such devices is a growing concern in industry.
Figure 1. Test Scheduling of the cores of a SoC (a) without and (b) with analog BIST.
- 124 -DCIS 2004
Sanahuja R., Barcons V., Balado L., Figueras J.Departament d'Enginyeria Electrònica, Universitat Politècnica de Catalunya
Diagonal 647, 08028 Barcelona, Spain, [email protected]
n the X-Y zone testing method the fault detection is based on the X-Y composition of two
signals of the CUT circuit, x(t) and y(t), in a similar way that an oscilloscope in X-Y mode
represents the evolution of the two signals on the plane.
In Figure 1 the X-Y composition curves for a good circuit and a defective one are shown.
To detect defects in the CUT a control line is drown tangentially to the non-defective curve.
When a defect changes the shape of the curve, the control line cuts the curve.
The X-Y zone detector implements the control line and is composed by a block configured
as a weighted adder of the two composed signals plus a reference voltage and a comparator
block. In this paper a Quasi-Floating Gate (QFG) structure (see in Figure 2) is used to design
a X-Y zoning detector for BIST in CMOS technologies.
A QFG-based layout core has been developed using 0.35 m AMS CMOS technology. To
use the detector for multiple test purposes, we assume input capacitors Cx and Cy to be
configurable in order to adjust the slope parameters of the control line.
Simulation results based on Spectre analysis reflect the advantages of using the system,
bringing important clues about future designs based on this method. The presented scheme
based only on capacitors makes the system suitable for been integrated as a configurable
BIST solution on CMOS.
BIST X-Y Zoning Detector Based on Quasi-Floating Gate Structure
I
Fig. 2. QFG X-Y zoning detector based on CMOS
vG -vx
vy1
Cx
Cy1Vn
vout
nmleak
vy2
nmleak
Cy2
Cdummy
vG +
-2 -1.5 -1 -0.5 0 0.5 1 1.5 2-2.5
-2
-1.5
-1
-0.5
0
0.5
1
1.5
2
2.5
Fig. 1. X-Y composition curves for a non-defective and a defective circuit and a control line drawn tangential to the non-defective curve
-2 -1.5 -1 -0.5 0 0.5 1 1.5 2-2.5
-2
-1.5
-1
-0.5
0
0.5
1
1.5
2
2.5
Fig. 1. X-Y composition curves for a non-defective and a defective circuit and a control line drawn tangential to the non-defective curve
- 125 -DCIS 2004
M.A. Domínguez(1), J.L. Ausín(1), G. Torelli(2), and J.F. Duque-Carrillo(1)
(1)Department of Electronics and Electrical Engineering, University of Extremadura,06071 Badajoz, Spain. (e-mail: madominguez, jlausin, [email protected]).
(2) Department of Electronics, University of Pavia, I-27100 Pavia, Italy.(e-mail: [email protected])
HE miniaturization of CMOS technology has enabled complex analog and digital cores to be
integrated onto the same silicon substrate. Those cores that are fully accessible from the external
pins can be tested by using appropriate testing equipment. However, in the case of embedded cores,
the access to internal nodes is not possible. Moreover, analog testing dominates the cost of testing
monolithic analog and mixed-signal circuits and, hence, a considerable interest exists in including
some circuitry on-chip to facilitate industrial testing.
The most basic analog functional measurement setup consists of a signal generator exciting the
circuit-under-test (CUT) with a periodic signal, and an instrument which extracts appropriate
parameters from the output response. Hence, a spectral analysis of the output signals is performed.
In this work, an effective approach to the design of on-chip spectrum analyzers based on switched-
capacitor (SC) techniques, is presented. The proposed spectrum analyzer for measurement of spectral-
based metrics of analog circuits is shown in Fig. 1. A programmable SC biquad is used for the
implementation of sine-wave generator and filter, this ensures the synchronization of the system. High
programmability resolution is obtained by using a non-uniform sampling scheme without modifying
any capacitor value. As a result, capacitor spread and total capacitor area are reduced as compared to
traditional solutions and, hence, test area overhead can be minimized. Frequency response, THD and
SNR can be obtained by the proposed spectrum analyzer. To prove the feasibility of the proposed
approach, the design and the implementation of an SC spectrum analyzer in a 0.35
technology are discussed. The circuit occupies a chip area of 0.17 mm2.
fmaster
VrefVref C U T A D CA D C
fclk
fclk, 2fclk, 3fclk
FrequencySynthetizer
VoltageReference
Sine-waveOscillator
Sample & Hold+ Filter
A/DConverter
S&H
Figure 1. On-chip spectrum analyzer block diagram.
An SC Spectrum Analyzer for Testing Analog Circuits
T
- 126 -DCIS 2004
A Test Methodology to Compute Typical LNACharacterization Parameters
Gabriel A. Pinho, José Machado da Silva, and José S. MatosUniversidade do Porto, Faculdade de Engenharia, and INESC Porto,
Rª Dr. Roberto Frias, 4200-465, Porto, Portugal, gabriel, jms, [email protected]
TESTING embedded radiofrequency circuits has become a bottleneck for manufacturers. This paper
presents built-in test methodologies to compute typical LNA characterization parameters. Three
methods are suggested which allow to calculate LNA’s gain, 1dB compression and third-order
intercept points, harmonic distortion, signal-to-noise ratio and noise figure.
The first method relies on finding the transfer function polynomial that best fits a set of points
obtained from the LNA test operation. A varying amplitude stimulus is applied and the respective
output levels captured. The 1dB compression and third-order intercept points are then calculated after
the polynomial coefficients. It is also possible to obtain harmonic coefficients, using the mathematical
relationship between the polynomial coefficients (ai) and the harmonics coefficients (hi).
The second method allows for obtaining gain, phase, and harmonic distortion after correlation
operations. Gain and phase are obtained correlating the LNA’s output signal y(t) with the input in-
phase xs=Asin( t) and in quadrature xc=Acos( t) signals. The distortion-related parameters can be
obtained correlating y(t) with harmonics of the input test signals. If we consider the LNA response at a
two-tone test signal represented by x(t)=Asin( 1t)+Asin( 2t) and perform its correlation with
xs=Asin( 2t) and xc=Acos( 2t) and later with xs=Asin((2 2- 1)t) and xc=Acos((2 2- 1)t), we can
compute the amplitude relation between the fundamental component at frequency 2 and the third-
order intermodulation distortion component at frequency 2 2- 1.
Correlation is also used to estimate signal to noise ratio. This is done by cross-correlating images of
the signal under evaluation at different times. Calculating both input and output signal-to-noise ratios,
SNRin and SNRout, one can obtain LNA’s noise figure.
The infrastructure proposed to
implement the first method is shown in
Figure 1. A variable local oscillator is used
to generate the appropriate LNA test
stimulus, being a RF amplifier with RSSI
output used to measure the LNA output
power.
LO
LNA Desmod.Mixer
Modula.Mixer
T/R
PA
LO
ADC
ADC
DAC
DAC
DSP
IFAmp
Amp RSSIRF Switch
LO
RF Switch
Figure 1. Circuit infrastructure to implement the first method.
- 127 -DCIS 2004
Hardware Requirements for Testing M-S Circuits based on Multidimensional Lissajous Curves
E. Lupon, L. Balado, L. García, and J. Figueras Departament d'Enginyeria Electrònica, Universitat Politècnica de Catalunya
Av. Diagonal 647, planta 9, E-08028 Barcelona (Spain) Phone: (+34)934017784, Fax: (+34)934017785, e-mail: [email protected]
issajous based Testing (LBT) of analog circuits by monitoring the combined evolution of several
signals has been shown to effectively detect both parametric and catastrophic faults. If the
composition is made with more than two signals with different frequencies, the curve obtained is said
a Lissajous knot.
For test purposes, the basic idea is to partition the space of the observed signals into zones and count
the number of zone to zone crossings of the knot during a test session. The contents of these counters
constitute the digital signature of the analog circuit under test. In this paper, a generalized
multidimensional LBT methodology is proposed. The observation space is partitioned into zones using
planes or hyper-planes (see an example in Fig. 1).
To obtain the signature of a knot, a matrix structure containing a cell for all possible transitions
between zones is used. Each cell of the matrix is a counter of the specific transition assigned to the cell
in a complete cycle of the knot. The values of the counters change due to the defect allowing the
possibility of defining metrics for taking the pass/fail decision.
The paper proposes an architecture to determine the Lissajous point position as time evolves. The
circuit monitorises in which of the half-spaces of the hiper-planes the point is located. Weighted
adders and fast hysteresis comparators determine the zone crossings to be counted. The simulation
results show the effectiveness of the proposed testing applied to a biquad filter CUT for three
dimensional knots.
L
0
1
2
3
5
6
7
4
0
1
2
3
5
6
7
4
0
1
2
3
5
6
7
4
0
1
2
3
5
6
7
4
Fig. 1. Cube with a knot curve and a set of possible control planes that divide the
cube in different zones, one of them showed in the right figure.
- 128 -DCIS 2004
Session 5d
RF Building Blocks Thursday nov. 25 8h30 10h00, Auditorium
Chairs
Eric Kerhervé (E.N.S.E.I.R.Bordeaux) Antonio Torralba (U. de Sevilla)
Synchronous Oscillator Locked Loop: A New Delay Locked Loop Using Injection
Locked Oscillators as Delay Elements.
F. Badets (1), M. Benyahia (2), D.Belot (1)
(1) STMicroelectronics C R&D, Crolles, France (2) STMicroelectronics, CR&D, Rabat, Morocco
I N this paper it is explained how injection locked oscillators could be used as delay elements. An
example of implementation of a new DLL called SOLL (Synchronous Oscillator Locked Loop)
using 100 MHz Injection Locked relaxation oscillator is described.
- 130 -DCIS 2004
R. Diaz, R. Pulido, A. Goni-Iturri, S. L. Khemchandani, B. Gonzalez, J. del Pino Institute for Applied Microelectronics of Las Palmas de Gran Canaria University, Spain.
HIS work presents the design of a fully integrated passive mixer for the IEEE 802.11a wireless
LAN standard using a 0.35 µm CMOS standard technology. An operational amplifier has been
used in order to compensate the mixer attenuation. The average DC output voltage of the operational
amplifier is fixed using a common mode feedback (CMFB) circuit.
All passive devices are integrated on chip, including the impedance matching spiral inductors,
which have been designed by electromagnetic simulations. The circuit layout, shown in Figure 1,
occupies a total area of 0.605 mm2 including the spiral inductors.
The mixer provides 43 dB of conversion gain, 45 dB of single sideband noise figure (NF), a third
order input intercept point (IIP3) of 40 dBm, and a power consumption of 3.4 mW. Therefore this
performance is valid for the 802.11a standard.
This work shows that with proper mixer topology and design techniques it is possible to design a mixer
suitable to be used in the 5 GHz band with a low cost silicon technology.
Figure 1. Photograph of the designed mixer
A Fully Integrated Mixer in CMOS 0.35 m Technology for 802.11a WIFI Applications
T
- 131 -DCIS 2004
Effect of Mismatch and Delay on the Quadrature Cross-Coupled Relaxation Oscillator/Mixer
Luís Bica Oliveira and Jorge R. Fernandes I.S.Técnico/INESC-ID-Lisboa,
R. Alves Redol 9, 1000-029 Lisboa, Portugal. E-mail:luis.b.oliveira;[email protected]
A LTHOUGH study of circuits capable to perform two functions, oscillation and mixing, is still
at an exploratory stage, we provide here some clarification on their performance. A cross-coupled
relaxation oscillator has two outputs with the same frequency which are accurately in quadrature. In
this paper we study how to perform mixing directly in the oscillator, which yields an oscillator/mixer
circuit. We evaluate how the inclusion of mixing in the oscillator affects its performance. This is an
oscillator/mixer high-level study where equations for the duty-cycle, quadrature relation and
oscillation frequency as a function of circuit parameters are obtained. We design a 2.4 GHz CMOS
relaxation oscillator-mixer using AMS 0.35µm technology to confirm the theoretical results by
simulation.
- 132 -DCIS 2004
High-gain LNA in 0.18 µm CMOS Technology for a WLAN receiver
I. Adin*, G. Bistué**, C. Quemada*, H. Solar*, J. Presa*, J. Legarda*
CEIT, Centro de Estudios e Investigaciones de Guipúzcoa, San Sebastián. Web page: www.rf.ceit.es. E-mail: [email protected]
** Centro Tecnológico de la Universidad de Navarra (TECNUN), San Sebastián
Two Low Noise Amplifiers (LNA) for 802.11a WLAN standard have been implemented using UMC
0.18 m 6 metal layers CMOS technology. This first Stage amplifier must fulfil the hard requirements
of the standard, here summarised: the available signal treated varies between -82dBm and -30dBm.
Furthermore it must not reach a Noise Figure as high as 14dB, neither a 1dB compression point of -
20dBm, considering the entire working bandwidth (300MHz) of the 5-6 GHz band.
In this work, two circuits have been implemented. The first design is built in a single-ended
architecture whereas the second is differential, both with inductive degeneration in the input stage and
cascode at the output. These two concepts deserve a brief explanation. The inductive degeneration
consists in an inductor connected to the source terminal of the amplifying transistor: It provides a good
input matching adding low noise. Moreover, the cascode improves the frequency response of the
whole circuit as it mitigates the parasitic miller’s effect. It also augments the reverse isolation.
In order to complement the study, the passives components used have been modelled in a pi-model
and characterised in an accurate on-wafer testing which provides a maximum error of 3% in the
inductance value and in the quality of the inductor.
Finally the complete results are exposed. The reached Noise Figure (NF) values vary from 2.2 to
2.8dB, with a gain as high as 19.6 and 13.9dB respectively. The single-ended LNA provides a high
gain with a low noise figure (next to the minimum NF available with this circuit), nevertheless the
differential one lose precision in all fields because of the noise optimisation, which is the most
important parameter of this kind of design.
In any case, the WLAN 802.11a is easily accomplished which proves the adaptation of the UMC
0.18 m 6 metal layers CMOS technology is adequate for this application.
Figure 1. Part of the layout of the LNA
- 133 -DCIS 2004
Microwave Low Noise HEMT Gate Mixers
F. Amrouche, R. Allam, and J-M. Paillot Laboratoire d’Automatique et Informatique Industrielle (LAII), Poitiers University.
IUT d’Angoulême (GEII)/ 4 Avenue de Varsovie, 16021, Angoulême,[email protected]
S EVERAL authors have already analyzed the noise of the gate mixer in which the noise figure is
varied between 7-10 dB in microwave applications. The noise figure in the mixer is higher than the
noise figure in the low noise amplifier (LNA). This mixer noise figure is masked by amplifier gain in
down-conversion receiver. Decrease of this noise figure means a reduction of amplifier number
cascade stages.
The design philosophy of the HEMT gate mixer is based on the idea to exploite nonlinear
behaviours by maximizing the magnitude of the transconductance (Gm) at fundamental frequency to
allow a high conversion gain.
This paper presents the design, simulation and calculation of two down-conversion mixers for LO-
RF frequencies in the 9.5-11 GHz band. The first mixer is the classic design, which is caracterized by
a high conversion gain. The design of the second mixer aimed at obtaining low noise figure in the
HEMT gate mixer by optimizing the input circuit. The optimisation and the design of the low noise
mixer concept is defined by simulation using harmonic balance tool giving by ADS simulator and
calculation using analytical formula describing the mixer noise figure as function of OL power level.
This analytical calculation can be achieved using supperposition method1 used generally to describe
amplifier noise figure. Noise figure analytical calculation points out the role of this nonlinear element
and enables the optimizing of the mixer noise performance. The aim of this work is to demonstrate the
possibility of decreasing the noise figure in the HEMT gate mixer, based on the optimisation of the
input matching circuit2. The noise figure performances of the mixers are measured and compared with
the calculated and simulated performances. The single side band noise figure results are in good
agreement with the experimental data. The LO, RF and IF frequencies chosen for this test are 9.5, 11
and 1.5 GHz, respectively. It is shown that the noise figure is reduced of 4 dB in the low noise mixer
circuit.
1F. Amrouche, R. Allam, J.M. Paillot, “Simulation and analytical calculation of the noise figre in HEMT gatemixers”, 33rd European Microwave Conference, pp. 351-354, Munich, October 2003.
2F. Amrouche and R. Allam, “Analysis and Design of Microwaves Low Noise Mixers”, IEEE MediterraneanMicrowave Symposium, Abstract 137, Marseille, June 2004.
- 134 -DCIS 2004
A 2.45 GHz Low Phase-Noise CMOS Ring Oscillator
V. Cheynet de Beaupré1, L. Zaid1, W. Rahajandraibe1, G. Bas2
1L2MP-Polytech Marseille – UMR CNRS 6137 - Université de ProvenceIMT-Technopôle de Château Gombert, 13451 Marseille, Cedex 20, France
2STMicroelectronics, Zone Industrielle de Rousset, France E-mail: vincent.cheynet,lakhdar.zaid,[email protected]
THIS work deals with the analysis and design of a 2.45 GHz CMOS ring oscillator (VCO)
with phase-noise lower than -90dBc/Hz at 500kHz frequency offset. The study is placed in
low cost context with a dramatic constraint for the silicon area and power consumption.
Generally speaking, a VCO must be able to work at high frequency with optimal power
consumption, but one of the key specifications is phase-noise minimization. We propose in
this paper the design of CMOS ring oscillator dedicated to radiofrequency systems belonging
to IEEE 802.15 standard applications. Derived from the Yan and Luong cell [1], the proposed
structure can work at a frequency twice as high as the initial circuit while insuring low phase-
noise and low power consumption. Implemented on 0.28µm CMOS technology, the circuit
occupies only 35µm×35µm with a power consumption of 19mW. Detailed study of the
proposed structure is performed. Simulation results obtained with STMicroelectronics 0.28µm
CMOS technology are presented.
1W. S. T. Yan and H. C. Luong, “A 900 MHz Low-Phase-Noise Voltage-Controlled Ring Oscillator ,” IEEE
Trans. On Circuits and Systems-II: Analog and Digital Signal Processing, vol. 48, NO. 2, pp. 216-221, Feb.
2001.
- 135 -DCIS 2004
Session 6a
New Architectures & Rapid Prototyping and Debugging Thursday nov. 25 10h30 11h30, Margaux Room
Chairs
Juan José Rodríguez (U. de Vigo) Angel de Castro (U. Politécnica de Madrid)
100 MHz Floating Point Processing Unit– A Feasibility Study
Mauro Castelli and Erwin OfnerFH Technikum Kärnten, Carinthia Tech Institute, Villach, Austria
THIS paper discusses several architectures for a floating-point processing module capable to
execute 100 million multiply-and-accumulate (MAC) instructions per second. Pipelining and
parallel operation will be considered in order to meet the specified goals.
The feasibility of the processing module (Figure 1) to be implemented in a 0.35µm CMOS fabrication
technology with three metal layers and a supply voltage of 3.3V is studied. The module is to serve as
stand-alone IP or as co-processor to a software programmable DSP core, specially designed for mixed-
signal integrated circuits. Beside the MAC instruction, the module needs to compute ADD, SUB,
Fixed2Float and Float2Fixed instructions. Data are assumed to be 16 bits wide, out of which 12 bits
are used for the mantissa and 4 bits for the exponent. 100 million instructions per second need to be
executed with a clock of 100 MHz supplied to the module, with the speed requirement given by the
DSP core. Low power operation and small silicon area are of secondary interest, therefore the use of
pipelining and parallel logic to increase the operating speed is considered.
The application of both techniques provided an increase in speed, however, also additional hardware
costs. Parallel computing did not meet the 100 MHz specification for the 0.35 m mixed-signal
fabrication technology, whilst the pipelined solution did. On the other hand, the pipelined solution did
show a loss of accuracy for input sequences at ½ Nyquist frequency, which is not visible in the parallel
implementation.
Figure 1. ALU with MAC capability
- 137 -DCIS 2004
A Linear Sorter Core based on a Programmable Register File
Lluís Ribas, David Castells, Jordi CarrabinaComputer Science Department, Universitat Autònoma de Barcelona, Bellaterra, Spain,
Lluis.Ribas, David.Castells, [email protected]
ARDWARE sorters exploit inherent concurrency to improve the performance of sequential,
software-based sorting algorithms. They are usually based on Batcher’s odd-even or bitonic
merging networks to attenuate the area-greedy hardware solutions. These sorting networks require
(n log2 n) processing elements (shown in fig. 1) and, with the appropriate pipeline, can sort n data in a
single clock tick. Unfortunately, their size might not fit in single-chip solutions. For cases where
log2 n >> 1, elements of merging networks can be re-used by re-circulating data through them at the
cost of an area overhead due to additional programmable interconnection networks and corresponding
controllers. Simpler controlling schemes can be applied on linear arrays, thus minimizing area
complexity but with a penalty in time, i.e. the number of clock cycles has a linear complexity rather
than a logarithmic one.
In this paper, a new hardware sorter architecture built on a programmable register file is presented.
It is inspired on insertion sorting and composed of n data-slice cells (shown in fig. 2). Each cell
optionally shifts its contents to the next one, thus the result module is named shifter sorter. As other
approaches based on linear arrays, shifter sorters are easily expandable and require minimal control
schemes. Differently from them, they use simpler processing elements.
Results show drastic area savings with respect to other approaches. On the other hand, though
shifter sorters can operate with much faster clock signals, sorting pipelined networks achieve better
time responses with parallel input data. However, the former exhibit a better area-time performance.
For serialized input data, shifter sorters outperform both in area and time previous approaches.
H
Di+1Di
D
pi
Ri
Riload
ck
Ri D 1
0
Figure 2. Processing element of proposed shifter sorters.
A X
A B
1
0
BY
0
1d
d = 0
d = 1
Figure 1. Common processing element of sorting networks and linear arrays.
- 138 -DCIS 2004
Rapid Prototyping Environment for CMOS Camera IC and Systems Design
E.Pons1, J.L.Merino2, L.Terés2, J.Carrabina1
1Departament d’Informàtica, UAB, Barcelona, Spain, [email protected]; 2Centre Nacional de Microelectrònica – CSIC, Campus UAB, Barcelona, Spain
HIS paper presents a methodology for the design, verification and prototyping of complete systems
and microsystems development, that contain CMOS image sensor arrays together with acquisition,
processing and communication features.
When an optical sensor must be selected for a concrete application, some control and image
processing must be performed to provide the required image format to the application. In order to
adapt the image sensor to the desired application, a rapid prototyping environment has been designed.
It allows the test of commercial and custom sensors, with its own control capabilities, as image format,
frame rate, readout management or sensitivity calibration.
The prototyping environment allows the use of a SW environment to develop the image
processing necessary for the required application (OCR, pattern detection…). Due to the
reconfigurability of the system different communication resources are available, as wireless, bluetooth,
USB, UART…So, SW development could be performed over a user-defined platform.
In conclusion, the prototyping environment permits the selection of the best sensor for the
application and the refinement of the processing application using several SW platforms. It’s a useful
and powerful platform for the initial development of an image sensing based system and also for
testing custom CMOS sensors.
This methodology has been validated for different CMOS sensors including proprietary
developments, research focused sensors and commercial sensors. Management of acquisition,
processing and communication was done using proprietary FPGA platforms. Communication methods
include UART, Ethernet and wireless interfaces.
T
- 139 -DCIS 2004
Advances in real-time hardware debugging using the UNSHADES system
J. Tombs, M.A. Aguirre-Echanove Department of Electronic Engineering, University of Seville.
T HE use of FPGAs in the implementation of rapid prototyping systems can allow the fast
creation of powerful digital hardware emulators. This potential for system debugging with such
systems is usually very limited, and mainly related to the monitoring of the external interfaces. To
improve the analysis capabilities of such systems, some commercial packages have been made which
allow the capture of intern signals, but these systems are limited in both the number of signals and the
number of captures that can be made. This article describes the most recent advances made in a
hardware debugger system known as the UNSHADES system. This system, works by the addition of a
small debug controller to the design to be debugged. This controller provides many new design
analysis features such as single clock stepping, state modification or register inspection. All these
options are provided over the entire design without the need for pre-synthesis signal selection. The
addition of the debug controller can be achieved with only minor design modifications and very little
dedicated FPGA resources. A sample debug controller implemented in a virtex2 FPGA requires the
use of just 3 IO pins and 43 logic slices. Over half of these logic slices are dedicated to an optional
32bit cycle counter.
- 140 -DCIS 2004
Session 6b
Sigma-Delta Modulator Design (1) Thursday nov. 25 10h30 11h30, Sauternes Room
Chairs
Andréas Kaiser (IEMN-ISEN) Patrick Loumeau (ENST)
A 3-30 MHz Tunable Continuous-Time Bandpass Sigma-Delta A/D Converter for Direct Conversion of
Radio Signals
D. Bisbal, J. San Pablo, J. Arias, L. Quintanilla, J. Vicente, and J. Barbolla
Departamento de E. y Electrónica, E. T. S. I. de Telecomunicación, Universidad de Valladolid, 47011-Valladolid, Spain, e-mail: [email protected]
CMOS fourth-order continuous-time bandpass delta-sigma modulator has been designed and
simulated. The outstanding features of the proposed modulator are wide-range tuning capability
and low power consumption. We suggest its use in a radio communications receiver front-end to
perform A/D conversion of the RF signal coming from the receiver antenna before any mixing (Fig.
1a). This way, the entire signal process mixing, filtering and demodulation is carried out by
digital circuitry, thus allowing high performance to be achieved with low power consumption and at
low cost. Receiver analog components are reduced to just a low noise amplifier (LNA), a bandpass
filter and an A/D converter. In this paper, the ADC is designed in order to be integrated in a single-
chip, short-wave radio receiver, implemented in standard triple-metal 0.35 m CMOS technology.
The architecture chosen to implement the bandpass ADC (shown in Fig. 1b) consists of the
cascade of two Gm-C resonators. It allows high-resolution digitalization of narrow-band signals
modulated at fo = gm/2 C when clocked at fs 4fo. Capacitors denoted by C are actually capacitors
arrays which allow the modulator to be coarsely tuned. Fine tuning may also be done by adjusting the
transconductances by means of changing the bias point of transconductors. Of course, the sampling
frequency fs must follow the changes either on C or on transconductances. Tuning is done
automatically by means of a master-slave tuning system which is also presented in the paper.
Simulation results show that the proposed ADC achieves an SNDR higher than 60 dB for 10 kHz-
bandwidth AM/SSB signals modulated on a carrier in all the HF band (3-30 MHz), while consuming
only 6 mW.
Figure 1. Example of an electronic circuit picture
A
LNA BP ADC
DECIMATOR(LPF)
CLOCKSINTHESIZER
(Digital tuner)
TUNINGCIRCUIT
BPF
AGC
fCLKfCLK / 4
Vtune
Analog domain Digital domain
DSP
(DEMOD. &FILTERING)
I
Q
1, 0, -1, 0, ...
0, 1, 0, -1, ...
cos(2 fCLK·t)
sin(2 fCLK·t)
14
DECIMATOR(LPF)
REFCLOCK
LNALNA BP ADC
DECIMATOR(LPF)
DECIMATOR(LPF)
CLOCKSINTHESIZER
(Digital tuner)
TUNINGCIRCUITTUNINGCIRCUIT
BPFBPF
AGC
fCLKfCLK / 4
Vtune
Analog domain Digital domain
DSP
(DEMOD. &FILTERING)
I
Q
1, 0, -1, 0, ...
0, 1, 0, -1, ...
cos(2 fCLK·t)
sin(2 fCLK·t)
1414
DECIMATOR(LPF)
DECIMATOR(LPF)
REFCLOCK
z -1CLK
gm
-gm -gm
gmgm
gma0 gma1 gma2 gma3
Vin
Vout
gmb0 gmb1 gmb2 gmb3
DAC
C CCC
z -1z -1CLK
gm
-gm -gm
gmgm
gma0 gma1 gma2 gma3
Vin
Vout
gmb0 gmb1 gmb2 gmb3
DAC
C CCC
(a) (b)
Fig. 1. (a) Proposed receiver architecture. (b) Single-ended block diagram of the 4th-order bandpass modulator, with loop filter based on thecascade of two Gm-C resonators.
- 142 -DCIS 2004
A New Method for the High-Level Synthesis of Continuous-Time Cascaded Modulators
Ramón Tortosa, José M. de la Rosa, Angel Rodríguez-Vázquez and Francisco V. Fernández
Instituto de Microelectrónica de Sevilla IMSE-CNM (CSIC) Ed. CNM-CICA, Av. Reina Mercedes s/n, 41012 Sevilla, SPAIN.
E-mail: tortosa|jrosa|angel|[email protected]*
ONTINUOUS Time (CT) Sigma-Delta Modulators ( Ms) have demonstrated to be an attractive
solution for the implementation of analog-to-digital interfaces in modern broadband
communication systems. In addition to show an intrinsic antialiasing filtering, CT Ms provide faster
operation with lower power consumption than their Discrete-Time (DT) counterparts. In spite of their
mentioned advantages, CT Ms are more sensitive than DT Ms to some circuit errors, namely:
clock jitter, excess loop delay and technology parameter variations. The latter are especially critical for
the realization of cascaded architectures, what has forced to use single-loop topologies in most
reported prototypes.
However, the need to achieve medium-high resolutions (>12bits) within high signal bandwidths
(>20MHz) while guaranteeing stability, has prompted the interest in proper methods for the synthesis
of cascaded CT Ms. These methods are based on applying a DT-to-CT transformation to an
equivalent DT M that fulfils the required specifications. One of the problems of using such a
transformation is that additional feedforward coefficients are normally needed to achieve an absolute
equivalence. As a consequence, a high number of analog components (transconductors and/or
amplifiers and digital-to-analog converters) have to be included in order to implement all the arising
coefficients.
This paper presents a new methodology for the high-level synthesis of cascaded CT Ms, that
based on dispensing with the DT-to-CT equivalence, allows to efficiently place the zeroes/poles of the
loop-filter transfer function and to reduce the number of analog components. This leads to more
efficient architectures in terms of circuitry complexity, power consumption and robustness with
respect to circuit non-idealities. As an application of the proposed method, several new cascaded CT
Ms have been synthesized and optimized to cope with VDSL specifications, i.e 12-bit resolution
within a 20-MHz signal bandwidth. Behavioural simulations considering most critical error
mechanisms are shown to validate the presented approach.
* This work has been supported by the Spanish CICYT Project TIC2001-0929/ADAVERE.
C
- 143 -DCIS 2004
Modeling All-MOS Log-DomainΣ∆ A/D Converters
Xavier Redondo1, Jofre Pallarès2 and Francisco Serra-Graells1
[email protected], [email protected] [email protected] Institut de Microelectrònica de Barcelona, CNM, CSIC (Spain)
2 Barcelona International R&D Core, Epson Europe Electronics GmbH - CNM
THIS paper presents new high-level modeling techniques to improve the simulation speed of all-MOS
oversamplingΣ∆ A/D converters implemented in the log-domain1. Functional modeling for all
basic building blocks is obtained from the analytical circuit analysis at transistor level using advanced
MOS device models, including thermal noise, moderate inversion, non-linear MOS capacitance and
DAC waveform asymmetry. The resulting behavioral models improve simulation speed by more than
1000 times respect to classic SPICE verification, while preserving device-level accuracy, and also allow
the independent study of circuit non-idealities. A complete design example of a fourth-order single-bit
Σ∆ modulator is given for a digital 0.35µm CMOS technology. In this case, the overall dynamic range
estimation of the ADC can be computed, using the same resources, in 1 hour instead of 18 weeks.
Vk
Vdac Ck
Itunk
Icapk
Vk-1
Inoise for bout=1 bout=0
VinpVinnVcap
Icap
transconductance
1s
integration
MOS cap
bout
Vdac
Vk-1
Qcap
Cmos(Vcap)
Vk
Figure 1: Example of high-level modeling for the integrator basic building block (left) and some comparativeresults from functional and electric simulations (right).
1F. Serra-Graells, “1V All-MOSΣ∆ A/D Converters in the Log-Domain,”Journal of Analog Integrated Circuits and Signal Processing,Kluwer Academic Publishers, Special Issue on ISCAS’02, vol. 35, no. 1, pp. 47–57, Apr 2003.
- 144 -DCIS 2004
A Dual-Band Sigma-Delta Modulator for GSM/WCDMA Receivers
Ana Rusu1,2, Babita Roslind Jose1, Mohammed Ismail1,3 and Hannu Tenhunen1
1 IMIT/LECS, Royal Institute of Technology Stockholm, Sweden 2 Technical University of Cluj-Napoca, Romania
3 Ohio State University, Columbus, USAE-mail: ana, ismail, [email protected]
THIS paper presents a dual-band sigma-delta modulator for GSM/WCDMA receivers. The
modulator makes use of low-distortion sigma-delta modulator architecture to attain high linearity
over a wide bandwidth. The dual-band modulator employs a low-distortion 2nd order single bit sigma-
delta modulator for GSM mode and a low-distortion 4th order modified cascaded modulator with
single-bit in the first stage and 4-bit in the second stage for WCDMA mode. In GSM mode, the second
stage is switched off to reduce the power dissipation.
Our sigma-delta modulator shown in Fig. 1 involves two keys design issues. One is the 2nd order
sigma-delta modulator with feedforward signal path, which has a reduced sensitivity to opamp
nonlinearities. The other key issue is an architectural approach, which combines the merits of modified
cascaded (2-2) architecture and multibit quantization in the last stage to make all quantization noise
sources negligible at low OSR.
The modulator is designed in 0.18um CMOS technology and operates at 1.8 supply voltage.
Simulation results show that, the proposed architecture has good tolerance to circuit nonidealities and
achieves a peak SNDR of 75-dB in WCDMA mode for an OSR of 16 and a peak SNDR of 83-dB in
GSM mode, for an OSR of 160.
Fig. 1. Block Diagram of the Dual-Band Sigma-Delta Modulator
- 145 -DCIS 2004
Session 6c
Analog Test Thursday nov. 25 10h30 11h30, Pyla Room
Chairs
Jean-Louis Carbonero (STMicroelectronics)Salvador Bracho (U. de Cantabria)
Probabilistic and Simulation-Based Masked-BIST Implementation
F. Guerreiro, J.M. Fernandes, M.B. Santos, A.L. Oliveira, I.M. Teixeira, J.P. Teixeira IST / INESC-ID, R. Alves Redol, 9, 1000-029 Lisboa, Portugal - [email protected]
Extended Abstract
Among the high-quality BIST (Built-In Self Test) solutions for digital systems (reseeding, bit flipping, etc), the masked-based (m-BIST) solution has been recently proposed. The key added value of m-BIST is the possibility of defining, at RTL, functionality-dependent partially specified input vectors (referred as masks) which can significantly enhance the accessibility of hard to control and/or observe functionality (referred as dark corners). RTL functional test generation can be reused for production test. The m-BIST approach was initially proposed as a simulation based (S-based) technique, using the VeriDOStool and combinational MUT (Modules Under Test). Using this tool, a low cost RTL fault simulation enables the identification of the dark corners [1]. However, mask generation needs to be performed manually, which limits the practical application of the methodology. More recently, a probabilistic-based (P-based) technique to m-BIST has been introduced [2], for combinational or sequential MUT, using the Ascopa tool. However, mask generation is not a trivial problem for sequential MUT. The purpose of this paper is to propose a more general implementation of the m-BIST approach, selectively using the S and P-based techniques to allow automatic mask generation in a cost-effective way. Mask generation was, so far, completely based on controllability data. Now, both controllability and observability metrics are considered. A limited subset of feedback registers is identified at RTL for partial scan BIST solutions. An ITC’99 benchmark circuit (b13) is used as test vehicle. Results are ascertained by fault simulation finally at structural level, leading to high single line stuck-at fault coverage results. Details are provided in the CD-ROM version of the paper. Here, only some key aspects are mentioned. The S-based approach provides estimations of the controllability and detectability, by applying a random test pattern to a given MUT: the percentage of simulation time that each bit (of a given RTL variable) remains at each logic level, ‘0’ or ‘1’, estimates the controllability, and the percentage of times a given fault is detected estimates its detectability. These testability values, obtained by simulation, strongly depend on the input vectors applied to the MUT. Hence, they can only be analysed as testability metrics if a large set of random vectors is applied. This is the case of a BIST solution, for which a pseudo-random (PR) test pattern generator (TPG) is built-in, and run at speed. Experiments show that S-
based results converge to P–based results, as the number of PR test vectors increase. When sequential MUT are under consideration, both primary and secondary inputs (PI, SI, respectively) need to be taken into account. In particular, state variables (the SI) are expected to bring testability problems, thus leading to the use of DfT (Design for Testability) techniques, like test-per-scan. However, not all SI are expected to be difficult to activate and to detect. The more the masks (forcing few PI and SI bits) are loosely deterministic, the less controllability of SI is required. This can save significant resources in terms of test overhead, test time and test power. In the P-based approach, probabilistic controllability and observability metrics evaluation is performed by the Ascopa tool, which also automatically generates the masks required for each condition activation. Mask generation is carried out early when the dependence graph is built. A mask can be generated for each condition identified in the RTL code. At present, the selection of the masks to use is made manually, based on the testability metrics values generated by Ascopa. The ITC’99 b013 benchmark circuit, described at RTL and at logic level (synthesized with a commercial synthesis tool to a Verilog logic description) is used. After the probabilistic based testability analysis and mask generation with Ascopa, RTL fault simulation with VeriDOS is carried out, using a PR test pattern for the PI (except the Reset and Clock lines), with 5k, 25k and 50k input vectors, in order to evaluate the eventual need of additional masks. Good correlation between probabilistic and simulation testability metrics is observed, and some differences explained in the full paper. Consequently, the S-based and P-based techniques used with m-BIST lead, for this case study, to a cost-effective BIST solution, using 20 masks and partial test-per-scan, and leading to high single stuck-at fault coverage on the structural description of the MUT. Observability problems are also identified and, selectively, a limited subset of secondary outputs is identified, to allow a smart, low-cost TPI (Test Point Insertion). References[1] M.B. Santos, F.M. Gonçalves, I.C. Teixeira and J.P. Teixeira,
"Implicit Functionality and Multiple Branch Coverage (IFMB): a Testability Metric for RT-Level", Proc. of the Int. Test Conf. (ITC), pp. 377-385, 2001.
[2] J.M. Fernandes, M.B. Santos, A. Oliveira, J.P. Teixeira, "A Probabilistic Method for the Computation of Testability of RTL Constructs", Proc. of the Design Automation and Test in Europe (DATE) Conf., pp. 176-181, 2004.
- 147 -DCIS 2004
Experimental Analysis of Transient Current Test Basedon IDD Variations in S2I Memory Cells
Y. Lechuga, R. Mozuelos, M. A. Allende, M. Martínez, S. BrachoMicroelectronics Engineering Group; Electronics Technology, Systems and Automation
Engineering Department; University of Cantabria; Santander; Spainyolanda, roman, allende, martinez, [email protected]
he current variations, IDD, appearing in the memory cells of SI circuits in the presence of faults,
give rise to changes in the overall dynamic supply current, IDDT, which are analyzed in the test
methods based on this IDDT current.
The capability of propagation of the effects of the faults injected inside the circuit can be
considered, and we will call it as fault reflection.
Basing on this fault reflection mechanism, a new test method that directly analyzes the current
variations, IDD, appearing in one of the memory cells that constitute the SI circuit, has been
developed.
This test method has the advantage of avoiding the losses of information appearing in the integration
of the IDDX signal, which can mask the effects of the faults.
We have designed and built, in AMS 0.6 technology, a benchmark circuit, shown in Figure 1,
based on a switched-current algorithmic A/D Converter topology to establish the fault coverage
obtained with this new test method, by real measurements; and conclusions have been extracted.
Figure 1. Benchmark circuit detail
T
- 148 -DCIS 2004
Testing of RF Systems by Zoning the Constellation Diagram
D. Arumí-Delgado, R. Rodríguez-Montañés and J. Figueras Universitat Politècnica de Catalunya, Department of Electronics Engineering
Diagonal, 647, P9, 08028 Barcelona, SPAIN Email: [email protected]
F systems take benefit from the digital modulation/demodulation techniques to transmit data. These
techniques separate the signal into two orthogonal components: I (In-phase) and Q (Quadrature). The
representation of the instantaneous value of the signal in the IQ diagram is called constellation diagram.
Sometimes the receiver is not able to recover the data, and the information obtained by the receiver has some
errors.
This paper presents an error detection method, which takes into consideration the symbols detected close
to the limits of the decision regions in constellation diagrams. This information allows the observation of
defects and mismatches, which are hardly detected by the classical phase division method. The method
consists of a properly division of the IQ diagram. When the symbols are obtained, every symbol is related to
an IQ division by means of a code. Due to this codification, errors in RF systems are detected.
Figure 1 shows the constellation symbols for a QPSK (Quadrature Phase Shift Keying) modulation and its
IQ diagram division when [1,-1] is the expected symbol. The method consists in defining four different
regions for every symbol: the expected division, the error division, the close undefined and the far undefined
divisions. The close undefined divisions enclose symbols which are correct, but are close to being wrongly
demodulated. On the other hand, the far undefined divisions give an idea of errors, which are close to being
correctly demodulated.
An OQPSK (Offset QPSK) modulation system has been implemented with Matlab. The faulty behavior of
the system is simulated by means of a Vt mismatched mixer in the quadrature component of the receiver.
Table I shows the simulation results. The difference between the close undefined and far undefined symbols
number shows the existence of the mismatch, which is more hardly detected with the BER value.
[-1 1]
[-1 -1] [1 -1]
[1 1]
2
6
7
3
108
1412
1513
11 9
5
1
0
4
Correct
Error Close undefined
Far undefined
Figure 1. IQ constellation division
R
TABLE ISIMULATION RESULTS
- Vt Mismatch- MISMATCH SYMBOLS
Vt(RF)(mV)
Vt(LO1) (mV)
Vt(LO2) (mV) Ncu/NT Nfu/NT Ne/NT BER
-- -- -- 1.076·10-1 6.337·10-4 1·10-7 6.338·10-4
-- 5 -- 1.107·10-1 7.091·10-4 1·10-7 7.092·10-4
-- 4 -3 1.137·10-1 7.873·10-4 1·10-7 7.874·10-4
5 -- -- 1.078·10-1 6.337·10-4 1·10-7 6.338·10-4
3 5 -- 1.117·10.-1 7.130·10-4 1·10-7 7.131·10-4
-2 -- 3 1.087·10-1 6.664·10-4 1·10-7 6.665·10-4
-6 -2.5 4 1.128·10-1 7.709·10-4 2·10-7 7.711·10-4
Ncu : Number of close undefined symbols Nfu : Number of far undefined symbols Ne : Number of error symbols BER : Bit Error Rate
- 149 -DCIS 2004
On the Minimum Number of Measurements for Single FaultDiagnosis in Linear Circuits
J. Soares Augusto
INESC-ID, R. Alves Redol, 9, 1000-029 Lisboa, Portugal
Physics Dept, Fac. Ciencias da Univ. de Lisboa
We describe a method for performing single fault diagnosis in linear dynamic circuits. The method is useful
for building compact fault dictionaries. It also demonstrates that the minimum number of diagnosis (circuit)
variables needed for single fault diagnosis at a single frequency is two, and this number is independent of the
circuit complexity. In practical cases this number can be larger due to the relationships between circuit variables.
The faulty circuit equations are obtained with a numerical modification of the LU factors of the nominal circuit
matrix resulting from the use of Modified Nodal Analysis (MNA).
The main result in the paper uses the t vectors, which are unique vectors associated to each circuit component,
that appear in the mathematical development of the faulty circuit equations. It is the (complex) elementwise
ratio between the t vector, and the (complex) vector difference dx between the solution of the nominal (good)
circuit (x) and the faulty circuit solution (x′), which allows for fault diagnosis. This ratio will be the same for
all the elements when the selected t vector corresponds to the correct fault.
The minimum number of variables needed for this task is two. An example of dx and of t vectors is shown in
figure 1. The main result is:
Suppose two circuit variables (voltages or currents), xα and xβ , are chosen as diagnosis variables
and that the corresponding elements in each t vector are tα and tβ . Suppose also that there are Nc
different parameters appearing in the MNA system of equations and that there is a fault in the k-th
parameter. Then, xα and xβ are sufficient as diagnosis variables iff:
∀Nc
k=1
α − α′k
tkα
=β − β′
k
tkβ
∀Nc
k=1,∀Nc
j=1,j =k
α − α′k
tjα
=β − β′
k
tjβ
where α and β are, respectively, the nominal values of the diagnosis variables xα and xβ , and α′k and
β′k are, respectively, the values calculated for xα and xβ when there is a fault in the k-th component.
V3
V1
V2
IV
I3
t1 t2 t3 dx1 dx2 dx3
Figure 1: Scaled pictorial representation of t and dx vectors for 3 faults in a small circuit. The vectors represent
phasors of circuit variables, and those not shown are zero.
- 150 -DCIS 2004
Session 6d
Mixed Signal Circuits for RF Applications
Thursday nov. 25 10h30 11h30, Auditorium
Chairs
Juan Maria Collantes (U. del Pais Vasco / Euskal Herriko U.)José Machado Silva (INESC-Porto)
Digitally Programmable UHF Transconductor in Digital CMOS Technologies
A. Otín, S. Celma, C. Aldea Electronic Design Group. University of Zaragoza. Spain
E-mail:aranotin, scelma, [email protected]
HIS paper presents a new approach for realizing digitally programmable VHF/UHF filters
which are suitable for pure digital CMOS technologies and for hard disk drive (HDD) read
channel applications. The strategy followed is based on a technique that provides a
programmable/tunable transconductance, based on a parallel connection of unit folded-cascode cells,
where the total parasitic capacitances are maintained constant thanks to the specific design of the unit
cell.
A fully-balanced current-mode Gm-C integrator has been implemented (Fig. 1, 2). It is able to
operate over the 30 MHz – 220 MHz range with a phase error of less than 4º and 80 dB of dynamic
range for 1% of total harmonic distortion (THD) over all the programming range. The cell has been
proved in 0.35 m and is aimed to be built in 0.18 m CMOS silicon technology. The transconductor
cell consumes 1.63mW from a power supply of 2V.The simulation results confirm this approach as a
fine choice to achieve filters exhibiting a good trade-off between tuning capability and dynamic range
working in the very high frequency range.
A comparison between several programmable filters implemented with similar technologies is also
included. In this way, certain conclusions can be drawn about the proposed design and the benefits of
this technique and the most relevant characteristics of the programmable filters are summarized. The
most striking quality to point out with a study of this comparison is that the use of current mode
operation combined with the proposed strategy leads to substantially wider dynamic ranges with lower
power consumption.
I-OI+
i
gm gm gm
gm gm gmI-
i I+O
CI
CI
I+i
IBIAS
I-o
VB
MP1
MN1VFN
M1
IBIAS
MP2
MN2M2
IBIAS
MP3
MN3M3
I+o
VFN
I- i
VB
IBIAS IBIASIBIAS
MP4
MN4
M4
MP5
MN5M5
MP6
MN6M6
CI
R
CI
R
Fig. 1. Fully-balanced digitally programmable integrator. Fig. 2. Fully-differential current-mode transconductance cell.
T
- 152 -DCIS 2004
A Mixed-Signal ASIC for FM-DCSK Modulation
M. Delgado-Restituto, A. J. Acosta and A. Rodríguez-VázquezInstituto de Microelectrónica de Sevilla, IMSE-CNM (CSIC),
Ed. CNM-CICA, Av. Reina Mercedes s/n, 41012 Sevilla, SPAIN. Phone: +34955056666, Fax: +34955056686, E-mail: mandel|acojim|[email protected]
THIS paper presents a mixed-signal ASIC for a Frequency-Modulated Differential Chaos Shift
Keying (FM-DCSK) communication system [1][2] which has been implemented in a 2P-3M
0.35µm CMOS technology. The prototype has been provided with several programming capabilities to
serve as an experimental platform for the evaluation of the FM-DCSK modulation scheme. The
operation of the integrated circuit is herein illustrated for a data rate of 500kb/s and a transmission
bandwidth in the range of 17MHz. Based on experimental results, an estimation of the Bit Error Rate
(BER) performance of the modulation scheme in a wireless environment at the 2.4GHz ISM band
under different propagation conditions has been realized. Measured results confirm theoretical
predictions.
- 153 -DCIS 2004
A multi-functional approach of frequency synthesizer dedicated to the next multi-standard smart objects.
Christophe Rougier1, Jean-Baptiste Begueret1, Hervé Lapuyade1,Yann Deval1 and Angelo Malvasi2
1 IXL Laboratory, 33405 Talence, France, e-mail : [email protected] 2ACCO, 21 bis rue d’Hennemont, 78100 St Germain-en-Laye, France, e-mail:[email protected]
With the continuous growth of communication standards, microelectronics designers have to adapt their circuits to fulfill telecommunications market. Nowadays, all the modern transceivers rely on multi-standard frequency synthesizer in order to cover various standards using the same devices. This way, using both the high level integration with recent technologies and these devices, we can provide circuits able to process different standards while consuming reasonable silicon area. Nowadays, the frequency synthesizer is one of the most fundamental cells in a telecommunication transceiver. This building block must synthesize the required periodic signals for both the up-conversion in the transmitter and the down-conversion in the receiver. Although multi-standard frequency synthesizers exist, they are dedicated only to a specific communication link (voice link, or data link, or positioning link). The emergent idea is to create architectures able to manage simultaneously various standards for different communication links on the same chip.
According to Fig. 1, it is obvious that future smart systems ought to process, in the same time, standards as GSM or DCS or PCS (for the voice link) as well as Bluetooth or HiperLAN standards (for the data link) and GPS standard for the positioning link. Consequently, these multi-standard systems should own a complex frequency synthesizer which will be both multi-standard and multi-functional.
So, this paper deals with a new approach of the frequency synthesizer, permitting to provide multiple local oscillators for different communication links. The feasibility of this structure to be both “multi-functional” and “multi-standard” will be demonstrated through behavioral simulation results.
It is a real challenge to manage such systems. Indeed, three different local oscillators will be synthesized on the same silicon substrate. So, parasitic couplings between the three local oscillators synthesized within a single silicon substrate may occur. To ensure a well controlled phase relationship between all local oscillators synthesized, a solution is proposed. Next, the multi-functional Frequency Generation Unit (FGU), depicted on the Fig. 2 able to provide at any time the wanted standard for a given communication link (phone, data or positioning) is presented.
« Positioning »link
« Data »link
« Voice »link
« Positioning »link
« Data »link
« Voice »link
Fig 1. Interaction between the different communications link Fig 2. The Frequency Generation Unit purpose.
Voice link
Data link
Positioning link
Voice
Data
Positioning
Fref
MCUsignal
Standard 1a
LOStandard 2
LOStandard 1
LOStandard 3
Frequency
Generation
Unit
I
Q
I
Q
I
Q
Voice link
Data link
Positioning link
Voice
Data
Positioning
Fref
MCUsignal
Standard 1a
LOStandard 2
LOStandard 1
LOStandard 3
Frequency
Generation
Unit
I
Q
I
Q
I
Q
- 154 -DCIS 2004
A Local Oscillator with a Reconfigurable Direct Digital Synthesis System
João Gonçalves(1,3), Jorge R. Fernandes(2,3)
(1) Deptº Engª Electrotécnica, Escola Superior de Tecnologia, Inst. Politécnico de Castelo Branco, Av. Empresário, 6000-767 Castelo Branco, Portugal
Phone (351)272339356 Fax (351)272339399 e-mail: [email protected];(2) Instituto Superior Técnico / (3) INESC-ID Lisboa, R. Alves Redol, 9, 1000-029
Lisboa, Portugal Phone (351)213100327 Fax (351)213145843 e-mail: [email protected]
I N this paper we present a prototype of a local oscillator based on the use of a Phase-Locked Loop (PLL) for the carrier frequency mixed with a Direct Digital Synthesis (DDS) system for channel
tuning. The system is evaluated with MATLAB, and the DDS is implemented in a FPGA and evaluated. The system is flexible, reconfigurable, and allows several types of digital modulation.
- 155 -DCIS 2004
Session 7a
Analog Design Methods Friday nov. 26 8h30 10h00, Auditorium
Chairs
Serge Verdeyme (U. de Limoges) Geneviève Duchamp (U. Bordeaux 1)
Optimization design of stacking voltage triplers for capacitive load
Ming ZHANG, Nicolas LLASER, Dariga MEEKHUN IEF, AXIS
University of South Paris, 91405 Orsay, France [email protected]
Extended abstract: Application of voltage multipliers can be widely found : EEPROM, low voltage integrated circuits, SoC etc. Compared with Dickson charge pump, which was proposed by Dickson in 1976 and is efficient in charge transfer, stacking charge pump proposed by Zhang in 1999 offers a quicker rise time of output voltage. Twostacking voltage triplers with a capacitive load are studied in this article: simple structure and improvedstructure, shown in figures 1 and 2, respectively. The simple structure’s operation takes place in two stepscorresponding to two phases of clock signal: charging pump capacitors and stacking them. In order to achieve the same output voltage with a minimum die area, an optimization design is developed. With the optimizationdesign, not only the circuit die area is minimized but also the maximum output voltage is augmented comparedwith equal capacitor choice of design. However, the presence of parasitic capacitances still limits the maximumoutput voltage. To reduce the influence of parasitic capacitances and further increase the output voltage, the improved structure is proposed. The improved voltage tripler has a three-phase operation and is controlled bytwo clock signals: charging pump capacitors, pre-charging parasitic capacitances and stacking pump capacitors.Consequently, the loss of output voltage is reduced by a factor of 2. By optimization design, not only themaximum output voltage is further increased but also circuit die area is further minimized, as shown in table I. Adesign guide is also given in this article: the choice between simple structure and improved structure and the choice between optimization design and equal capacitors’ design.
Fig. 1. Voltage tripler with symmetrical charge pumpstructure
Table I DESIGN RESULTS
Design Tot. Cap. Area loss C1=C2 simple impossible - -
C1=C2 improved 151pF 66% 60pFOptim. simple 131pF 44% 40pF
Optim. improved 91pF 0% 0pF
Fig. 2. Improved voltage tripler with symmetrical charge pump structure
M1
C1’
M2M1’
M22’ M2
M2’
C1C2’ M21’
Second
V2’
C2M21
SecondFirst stageM3
Output stageVdd Vdd Vdd Vdd M5
V1’ V1 V2
M6V1 V1’Vout
M4
I2’ I2’
M1
C1’
Vdd M2M1’
M22’ M22
M2’
Vdd
Vdd
C1C2’M21’
I2’
V1
V2’
Vdd
C2
V1’
I2
M21
V2V1’ V1
11’
Vdd
2
V1’
Vdd
2’
VddV1
I3’
I4’
I3
I4
Vdd
M23
M24
M23’
M24’
Vout
M3
M4
M5
M6
First stage PrechargingPrecharging Second stage Output stageSecond stage
Cout
- 157 -DCIS 2004
Design Considerations of a Frequency Synthesizer for a Mixed-Signal Built-In-Self-Test Application
África Luque, Diego Vázquez and Adoración RuedaInstituto de Microelectrónica de Sevilla, Centro Nacional de Microelectrónica (IMSE-CNM-CSIC)
Avda. Reina Mercedes s/n, 41012-Sevillaluque, dgarcia, [email protected]
This paper reports a programmable Phase-Locked Loop (PLL) frequency synthesizer designed for a Mixed-Signal Built-In-Self-Test (BIST) application. This synthesizer
generates the required signals for the characterization of sine-wave signals needed by an approach reported elsewhere. The basic structure of a typical PLL has been modified and adapted for the intended application. The structure and operating modes of the different blocks arepresented together with simulation results.
- 158 -DCIS 2004
Analog IC Design With A Library Of Parameterized Device Generators
Vincent Bourguet, Laurent de Lamarre and Marie-Minerve Louërat
University of Paris VI, LIP6-ASIM Laboratory,4, Place Jussieu, 75252 Paris, France
Email : [email protected]
Here we present the CAIRO+ language that allows the analog designer to create generators of analogfunctions.CAIRO+ is aimed to help the designer to capture his knowledge thus creating a library of analog func-tions. Complex hierarchical analog function generators are designed by using existing generators ofsimpler functions. These generators can be designed to be independent of the fabrication process thusenabling process and specification migration.The CAIRO+ language, composed of C++ predefined functions, is a new answer to the problem of elec-trical and layout co-design of analog circuits. It has inherited ideas from the CAIRO language concerninglayout-aware issues [2, 1], yet it has dramatically enhanced the communication between synthesis andlayout throughout the hierarchy.
Designing a Module Generator
A module is an instance of the hierarchical netlist representing the circuit, it is created by a module gene-rator. A module can instantiate other modules. The module tree is defined where one node correspondsto one module. The leaf cell of the module tree is called a device.
In order to predict parasitics resulting from layout, we have chosen an approach using layout templateswith layout device generators. The description of the relative placement of instances inside a module isdescribed by a container tree. A container is composed of abuted containers placed besides each otherin a specific order. There exists vertical and horizontal containers. The leaf cell of the container treecorresponds to a device.
Devices consist of elementary components such as folded MOS transistors, capacitors and resistancesbut also sets of elementary components that have to be matched (i.e. differential pair, current mirrors,capacitor matrices, matched resistances).
The module tree is used to represent the netlist template of the circuit and the container tree is used torepresent the layout template of the same circuit. In order to design a new module generator, the analogdesigner has to write the four following functions corresponding to the four steps of our design flow :
1.Capture of Netlist and Layout Templates. In this step, functions allow the creation of the netlist andthe relative placement of unsized instances.
2.Design Space Exploration. In this step, specifications are propagated from top to down in the moduletree thanks to dedicated functions, the result is a sized schematic.
3.Shape Function. In this step, the module layout shape function of the sized schematic is computed.The shape function gives all the possible aspect ratio for layouts of a sized schematic. The shape functionof a module is computed in a recursive manner, from bottom to top, based on the container tree.
4.Layout Generation. Finally, given a geometrical constraint, like module height or aspect ratio, thefeasible height of the module is selected, by examining it’s shape function. With a recursive top to bottomapproach, the actual shape of devices is selected. Then the relative placement is performed. Hierarchicalrouting from bottom to top is then performed.
RÉFÉRENCES
[1] Mohamed Dessouky, Marie-Minerve Louerat, and Jacky Porte. Layout-Oriented Synthesis of High Perfor-mance Analog Circuits.Proc. DATE 2000, pages 53–57, 2000.
[2] M. Dessouky.Design for Reuse of Analog Circuits. Ph. D. Thesis, University of Paris VI, 2001.
- 159 -DCIS 2004
An accurate algorithm for transistor sizing in analog CMOS design
Pablo Rodiz-Obaya*, Juan J. Rodríguez-Andina* and Jaime Ramírez-Angulo** * Departamento de Tecnología Electrónica, Universidad de Vigo, Spain
** Electrical and Computer Engineering Department, New Mexico State University, USA [email protected], [email protected], [email protected]
HE required design time needed in VLSI to implement a simple amplifier can be very high if
compared to the speed at which much more complex digital building blocks can be developed. The
problem is the lack of widely available useful automated tools for the design of analog circuits. Due to
the complexity of the complete MOS transistor model there is a problem creating useful parametric
cell libraries for analog design.
The goal of this work is to provide an exact method to calculate transistor sizes while keeping the
formulation of the topology simple. The cells are described mathematically in this method with first
order transistor models and simple circuit equations, but a “parameter update” loop makes the solution
to this formulation accurate by recalculating the first order model parameters with the complete
transistor model. This allows the designer to easily specify different circuits and create scripts to
automatically size the transistors as function of the given requirements while obtaining exact solutions.
The proposed method is based in the convex optimization approach already used in GPCAD1 and
others2, but a way to include complete spice simulations is implemented for the iterative calculation of
first order model parameters, therefore achieving more accurate results. The result is a set of scripts
from which an automatically-sizable cell library consisting of several basic analog building blocks has
been obtained. The development of one of them, the MOS inverter, is explained in detail as an
example. The scripts allow the designer to quickly obtain analog circuit prototypes for mixed VLSI
systems. A complete operational amplifier, briefly explained in the paper, has been developed an
actually fabricated with these scripts to demonstrate that the proposed methodology saves much of the
time required to design simple circuits, and thereby concentrates the work of the designer in higher
level tasks such as system specification or resource distribution.
1 M. Hershenson, S. Boyd and T. Lee, “Optimal Design of a CMOS Op-Amp via Geometric Programming”. In IEEE Trans. Computer-Aided Design of Integrated Circuits and Systems, 20:1-21, January 2001.
2 P. Mandal and V. Visvanathan, “CMOS Op-Amp Sizing Using a Geometric Programming Formulation”. In IEEE Trans. Computer-Aided Design of Integrated Circuits and Systems, 20:22-38, January 2001.
T
- 160 -DCIS 2004
Optimizing SI Class AB Memory Cells M. Fakhfakh, M. Loulou, and N. Masmoudi.
Laboratoire d’Electronique et des Technologies de l’Information.National Engineering School of Sfax, Tunisia.
S WITCHED current class AB memory cells are known to be preferment SI cells. However,
designing such optimal cells is a tedious process which reclines on designer experience. Thus,
automating the transistor sizing process is a very important step towards being able to rapidly design
high performance custom cells.
In our paper, we present an idea to improve SI memory cell’s performances. It is mainly based on a
stochastic exploration of an advisedly determined parameter vectors and on building a heuristic by
which an objective function, composed of a weighted sum of error and performance functions
expressed in reduced unities, is optimised. For this purpose we built mathematical models of the cell
and non_idealities affecting it. We applied the proposed procedure to design optimal S2I grounded
gate class AB memory cells. Figure 1 illustrates the proposed heuristic. The optimisation program,
written in C++ software, allowed us to reach high performances in terms of accuracy, SNR and speed.
The obtained results were proved by SPICE and CADENCE simulations. With use of 0.35µm CMOS
process, the treated cell reaches a dynamic range of 80 dB at 16MHz sampling frequency. For top
priority given to settling time, the application of the proposed heuristic allowed us to get less than
0.5ns as settling time. The proposed optimized cell will be used for designing switched current sigma-
delta converters and programmable filters suited for radio frequency applications.
calculus of theobjective function
random initialisation of the parametres' vectors
correction
verification of thepreliminary conditions
random choice ofthe vector variables
'''optimal''' parameters
simulation
Figure 1. The optimization approach
- 161 -DCIS 2004
The Importance of Microwave Approach for High Frequency MOS Analog Designers
ODAY analog products reach X and upper bands because of the increase in the need for low
cost, high bandwidth multimedia products. In order to satisfy to these new criterion, on chip
passive components and especially inductors are required. Thus integrated circuit designers are facing
new problems that typically belong to radio-frequency (RF) area and they have to consider both
method that was reserved up to now to each frequency domain (low and high) and to choose between
them.
In this paper, we present rapid method to evaluate Q-factor of inductors layouted both with the
traditional analog approach and with coplanar waveguides (CPW). Those method are mainly derived
from works from S.S. Mohan et al.1 and E. Yamashita et al.2 concerning the computation of L values.
Then we use equation RLQ where R is evaluated using metal layer conductivity and dimensions.
Finally, using a particular Silicon-On-Sapphire technology this paper shows that the classical analog
approach of printed square spiral cannot always meets both technical (high quality factors for low
inductance values) and industrial (small die areas) requirements at a frequency of 10GHz. CPW, a
second solution borrowed to the hyperfrequency field, are thus studied and their benefits pointed out.
Issues, solutions and limits presented in this papers can be easily extended to any kind of on-chip
inductors and microstrips just by changing values in given equations. RF analog designers must also
use presented methods to quickly and early choose between microstrips and printed inductors during
the design flow. More precise electromagnetic simulation can then be made for precise layout.
1 S.S. Mohan et al., Simple accurate expression for planar spiral inductances, IEEE JSSC, 1419-1424, Oct., 1999 2 E. Yamashita et al., Analysis of microstrip-like transmission lines by nonuniform discretization of integral equations, IEEE MTT-24, 195-200, 1976
Gilles Petit1, Richard Kielbasa2, Vincent Petit3
1Service des Mesures, Supélec, Gif sur Yvette, France, e-mail : [email protected] 2Service des Mesures, Supélec, Gif sur Yvette, France, e-mail : [email protected]
3Thales Airborne Systems, Élancourt, France, e-mail : [email protected]
T
- 162 -DCIS 2004
Session 7b
New Synchronisation schemes and Asynchronous Circuits
Friday nov. 26 8h30 10h00, Bordeaux Room
Chairs
Jonathan Tombs (U. de Sevilla) Fernando Rincón (U. de Castilla-La Mancha)
Secured structures for secured asynchronous QDI circuits
A. Razafindraibe1, M. Robert1, P. Maurine1
F. Bouesse2, B. Folco2, M. Renaudin2
1LIRMM, 161 rue Ada, 34392 Montpellier, France 2TIMA Laboratory 46, avenue Félix Viallet, 38031, France
This paper aims at introducing a novel design methodology of compact, high performance and secured
dual rail primitives widely used in Quasi-Delay Insensitive circuits (QDI). An example of application
of this design methodology to basic QDI primitives is given on a 130nm process. Performances and
security properties of the resulting cells are then compared, using electrical simulations, to the
implementations proposed in former works.
Self-timed circuits appear to be a promising alternative for cryptology since it is more difficult to
correlate leaking syndromes to data flowing in a secured design in absence of a clock signal. Indeed, it
exhibits significant timing and power consumption variations depending on input applied data.
Moreover, with a standard cell approach, QDI physical implementation of required Boolean functions
is sub optimal.
With our novel design methodology, QDI circuits exhibit better security properties than in former
works. This is explained by the equalization of the number of stages used to implement both rails
without adding any extra gate.
S1
A0
A0
A1
A1B0
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V
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Figure 1. Example of Secured QDI “xor2” cell
*With Centre Microélectronique de Gardanne (CMP) and STMicroelectronics collaboration
- 164 -DCIS 2004
Definition of P/N Width Ratio for CMOS Standard Cell Library
A. Verle, P. Maurine, N. Azémard, D. Auvergne
LIRMM, UMR CNRS/Université de Montpellier II, (C5506),161 rue Ada, 34392 Montpellier, France pmaurine, azemard, [email protected]
Summary
The efficiency of cell-based design synthesis of high performance circuit is strongly dependent on the content of the library. Great effort has been given in the design of libraries, to define the optimal selection of the logic gate drive strength. But few justifications are available to determine the P/N width ratio of each cell. The relative merits of different cell libraries can be evaluated in terms of area/power necessary in achieving a particular delay for implementing a specific circuit. For that important effort has been devoted to supply high-performance standard cell libraries. Work has been devoted to define the optimal content of the library as well as for determining the best selection of drives. A fluid cell approach is emerging, in which a cell generation tool is used to create a discrete library with 10 to 25 drive strengths and 1 to 4 different P/N width transistor ratios. The question arise to know if it is possible to define an optimal value for the gate transistor ratio allowing to implement a CMOS logic circuit with the best delay/power trade-off. Recent work has been published on P/N transistor width selection, based on an asymmetric implementation of the gate rise and fall delays. Considering that, for an array of inverters, the minimum delay can be obtained using asymmetric edges, a first order gate delay model has been used to determine an optimal transistor width ratio for each gate. This has been obtained by minimizing the average of the rising and falling delays to obtain an optimal solution in which the Nand gates are over-sized the and the NOR gates under-sized, with respect to that of an inverter.
In fact on a logical path a separate consideration of the falling and rising edges must be given. In this case it can be shown that, considering only the critical edge, the fastest solution is obtained for an inverter implementation with balanced fall and rise delays. Moreover, for gates great attention must be given to the modeling of the transistor serial array current and to the critical input to be considered. On a non critical path the minimum gate area solution can be obtained with unbalanced edges, resulting from identical equal N and P transistor sizes. We want to demonstrate, here, that on a critical path performance constraint satisfaction may result in transistor over-sizing and extra power consuming, if no care is given to the balancing of the gate rise and fall delays.
In this paper we use an extension of the logical effort model to characterize the dissymmetry of gate delay. The delay model is developed around the logical effort, but with an explicit consideration of the input ramp and Miller effects. This model explicitly captures the sensitivity of the delay to the gate structure and P/N width ratio. We propose a method for determining the P/N transistor width ratio for implementing high performance library cells. We have defined the explicit expression of the P/N width ratio, which is shown to be loading factor and structure dependent. This P/N width ratio is shown to allow a path minimum area implementation under delay constraint. Validations have been obtained, with respect to Spice simulations on a 0.18µm process, by comparing, on different benchmarks, simulated values of the delay using different P/N width ratio strategies. We obtain clear evidence that imposing on a logic path equal rise and fall gate delay, results in a high performance implementation for the best area- delay trade-off.
- 165 -DCIS 2004
One of the main difficult task designing security devices is to protect from the so called sidechannel attacks. These attacks take advantage of the correlation that can exist between internalcomputation and the side channel information provided by the considered device. They are ofparticular concern, since they are not invasive (i.e.: do not attempt to the integrity of the device) andcan be quickly set-up at relatively low cost. Among them, one of the most efficiently methods, is theDifferential Power Analysis (DPA), introduced by Paul Kocher in 1998. This attack exploits theexisting dependence between the data processed and the profile of the current consumed by the chip.
Several hardware counter-measures have been proposed against this attack and the asynchronousdesign technology is presented as a good alternative for reducing the current signature locally. In fact,as it has already been reported, asynchronous circuits are likely to improve chip security. Theproperties of QDI asynchronous circuits are exploited, particularly the use of the 1-of-N encoded datascheme and the four-phase handshake protocol.
This paper presents the first concrete results of Differential Power Analysis applied on securedQuasi Delay Insensitive asynchronous logic. We demonstrated on this work by measuring and byquantifying in real chip the benefits brought by the QDI asynchronous logic. For doing so, threedifferent DES circuits have been designed and fabricated: two in asynchronous technology and one insynchronous to be used as a reference.
The contribution of this paper is focused on two aspects: the definition of a DPA resistancecriterion used to compare different designs, and the evaluation of QDI asynchronous logic as a countermeasure to DPA.
The concrete results presented in this paper demonstrate that QDI asynchronous logic significantlyimprove the DPA resistance. This study also enabled us to identify some limits i.e. residual sources ofleakage, that will be addressed in future works.
DPA on Quasi Delay InsensitiveAsynchronous circuits: Concrete Results*
G.F. Bouesse1, M. Renaudin2, and B. Robisson3, E. Beigne4, P.Y. Liardet5, S. Prevosto6, J. Sonzogni7
1,2 F. Bouesse, and M. Renaudin, TIMA laboratory, 26 Av. Félix Viallet, 38031 Grenoble, France,e-mail : [email protected]
3,4 B. Robisson, and E. Beigne, CEA-Grenoble, 17, rue de Martyrs 38054 Grenoble, France,e-mail: [email protected]
5,6,7 P.Y. Liardet, S. Prevosto, J. Sonzogni. , STMicroelectronics, ZI Rousset 13106 Rousset, France,e-mail: [email protected]
- 166 -DCIS 2004
Four phase alternating latches clocking scheme for CMOS sequential circuits
David G. M. Manuel J. B., Jorge J. Ch., Alejandro M., Paulino R. de Clavijo and Enrique O. A., Departamento de Tecnología Electrónica de la Universidad de Sevilla/ Instituto de
Microelectrónica de Sevilla, Centro Nacional de Microelectrónica, Spain guerre,bellido,jjchico,amillan,pruiz,[email protected]
T HE evolution in the VLSI digital circuits design makes it mandatory to pay special attention to
the clocking scheme used to implement the system and to the clock generation and distribution
over the full system. While the gate size and, as a consequence, the gate delay is getting smaller, the
die size is rising. Since the delay in interconnection lines increases quadratically with the line length, it
becomes longer than gate delay. Because of that the skew increases significantly.
The authors present a generalization of the Parallel Alternating Latches Clocking Scheme that uses
separated clock signals to control the load and output enable operation of the latches. This increases
the operation speed of the system without losing clock-skew tolerance. Also, the fact that the clock
frequency is a half of the data rate reduces the switching noise and the power consumption of the clock
distribution network.
- 167 -DCIS 2004
A Memoryless Clock Domain Adaptation Unit IP
R. Esper-Chaín, F. Tobajas, F. González, R. Arteaga, & R. Sarmiento Instituto de Microelectrónica Aplicada, Departamento de Ingeniería Electrónica y Automática
Universidad de Las Palmas de Gran Canaria, Campus Universitario de Tafira, 35017, Las Palmas, Spain
esper,tobajas,fglez,rarteaga,[email protected]
The upcoming of SoC systems with several cores running with independent clock domains, based on the Globally Asynchronous Locally Synchronous paradigm (GALS), makes the transfer of clock domain crossing data a very common problem. In order to solve this problem several techniques has been used, however most of them are complex or require expensive elements such as DLLs or memories. In this paper a very simple unit is presented to perform this operation. This unit has been demonstrated running on a physical implementation done over a FPGA, with a 100% of efficiency.
- 168 -DCIS 2004
Synchronization of Sequential Circuits using the Asynchronous Wave Pipelining Technique
Stephan Hermanns and Sorin Alexander Huss hermanns, [email protected] University of Technology
Hochschulstrasse 10, 64289 Darmstadt, Germanyphone +49 (0)6151 16 3980, fax +49 (0)6151 16 4810
A synchronous and synchronous circuits have their own advantages and drawbacks. A new
approach towards a controller architecture saving some of the respective strong sides is presented.
Asynchronous Wave Pipelining (AWP) performs the computation of independent state machines.
Next-state and output combinational structure hold several data waves, each representing the
computation of the respective state maschine. Controllers with Asynchronous Wave Pipelines hold the
reduced average latency of Huffman Circuits together with their often reduced power consumption but
without the bounds of state coding of asynchronous circuits. AWPs have no need for global clock
signal distribution nets and storage elements such as synchronous state machines.
Proposed structure of switching circuits
Data paths using Wave Pipelining ask for controllers with an adequate performance. An approach
towards sequential circuits using Asynchronous Wave Pipelining was proposed. Constraints imposed
by a gate structure supporting wave pipelining and three conclusions on state machines using wave
pipelining have been stated. A solution following those statements and thereby solving the problem of
synchronizing waves which are carrying either input data or state information was proposed and tested
for an example AWP containing waves of two independent state machines. The use of the hazard-free
switching technique SRCMOS beside the mechanism of synchronization used within a sequential
circuit results in a race-free implementation of asynchronous state machines. The latter bases on the
fact that a single local anisochronous signal is used to synchronize plesiochronous signals in the
Huffman feedback loop.
- 169 -DCIS 2004
Session 7c
Failure Analysis & Reliabiliy Friday nov. 26 8h30 10h00, St Emilion Room
Chairs
Yves Danto (U. Bordeaux 1) Eugenio García (U. de les Illes Balears)
Flash Memory Cell: Threshold Voltage Sensibility to Geometry
B. Saillet1&2 J.M. Portal1 D. Née2
1 L2MP-Polytech-UMR CNRS 6137 IMT - Technopôle de Château Gombert
13451 Marseille Cedex 20, [email protected]
Tel:(33)-491-054-787
2 ST-Microelectronics ZI de Rousset BP 2
F-13106 Rousset CEDEX, France [email protected]
Tel:(33)-442-688-815
lash memories have become over the last few years very relevant choices for any application
requiring non-volatile semiconductor memory.
FThe objective of this paper is to study the impact of Flash cell geometry on the stored value. To do so,
a Design Of Experiment (DOE) approach has been used, giving the variations of the threshold
voltages of a memory cell in function of geometric parameters. The inputs of this DOE are transient
and static electrical simulations of Flash cell. The simulation model is based on a MOS Model 9
transistor coupled with the charge neutrality expression in the cell. The outputs of the DOE are a set of
equations describing the evolution of the threshold voltage of a virgin, an erased and a written cell.
The threshold voltage VT equation is given in a general way as follows where Xi and Xj are the
geometrical parameters and b0, bi, bij are the model coefficients:
ij
jiij
i
iiii
i
iiT XXbXXbXbbV 0
The sensibility of the threshold voltage to the geometric parameters is then discussed from the results
obtained with this set of equations. As represented Figure 1 for example, this kind of equation permit
to study the sensibility with the interaction all the parameters.
VTvirgin(V)
Wpp (µm) Tono (A
)
VTvirgin(V)
Wpp (µm) Tono (A
)Tox (A)
L(µm)
VTerase(V)
Tox (A)
L(µm)
VTerase(V)
Fig.1.a: VTvirgin variations versus Wpp and Tono Fig.1.b: VTerase variations versus Tox and L
- 171 -DCIS 2004
Modeling the Influence of Time Skew on Crosstalk Induced Delay in Submicron CMOS technologies
José Luis Rosselló and Jaume SeguraPhysics Department, Universitat de les Illes Balears, Palma de Mallorca, Balears (Spain),
email:j.rossello;[email protected]
T HE influence of time skew between adjacent signals on crosstalk delay is a complex non-linear
problem without analytical solution that can only be solved exactly using numerical procedures.
Some previous works on crosstalk delay only take into account the worst case1 since the inclusion of a
non-zero time skew is a much more complex problem. Recent works consider the time skew between
signals2 but the gate model used (the traditional resistive model) is too simplistic and no closed form
expression for crosstalk is provided.
The model presented in this work relates crosstalk delay with time skew using a charge-based
description of the propagation delay of CMOS gates3. This charge-based propagation delay model is
found to be a very useful and accurate tool for the timing description of deep-submicron CMOS ICs.
Crosstalk delay is modeled computing the additional charge that is transferred through the circuit due
to the coupling between gates. This additional charge is translated to an increment in the propagation
delay (that increases as the charge to be transferred increases). The influence of time skew between the
victim and aggressor inputs is traduced to a variation in the additional crosstalk-induced charge (that is
accurately translated to a delay variation of the gates). The model provides an intuitive description of
crosstalk delay showing very good agreement with HSPICE simulations for a 0.18 m technology
1 P.D.Gross, R. Arunachalam, K.Rajagopal and L.T.Pileggi, ''Determination of worst-case aggressor alignment for delay calculation,'' in Proc. Int. Conf. Computer-Aided Design (ICCAD), 1998, pp. 212-219
2 W.Y. Chen, S.K. Gupta, and M.A. Breuer, ''Analytical Models for Crosstalk Excitation and Propagation in VLSI Circuits,'' IEEE Transactions on Computer-Aided Design, Vol 21, no 10, pp. 1117-1131, October 2002
3 J.L. Rosselló and J Segura, "An analytical charge-based compact delay model for submicron CMOS inverters" IEEE Transactions on Circuits and Systems I. Vol 51, no. 7, pp. 1301-1311, July 2004
Figure 1. Example of an electronic circuit picture
- 172 -DCIS 2004
RC on-chip interconnect Performance revisited P. Maurine, N. Azémard, D. Auvergne
LIRMM, UMR CNRS/Université de Montpellier II, (C5506),161 rue Ada, 34392 Montpellier, France pmaurine, azemard, [email protected]
Due to the decrease in transistor size of CMOS circuits and to the line aspect ratio increase, on-chip interconnect resistance must be accounted for during timing analysis. This imposes to separate the path delay into gate and interconnect delay. Usually cells are characterized into tables or with equations that represent delay and output slew (transition time) as a function of load capacitance and input slew. The handling of interconnect resistance as been resolved by determining an effective capacitance loading, defined as the capacitance producing the same gate delay as the considered RC load.
The most widely used interconnect delay metric is the Elmore delay metric applied to a lumped interconnect model. Despite its lack of accuracy, this metric has numerous advantages such as to be expressible as a closed-form equation, to constitute an upper bound of the delay and mostly to be additive: the delay on a path is the sum of the delays at the different nodes of the path. The accuracy of this model is not sufficient for actual processes since it does not capture the delay sensitivity to the transition time of the edge controlling the interconnect line and ignores the gate reduction delay induced by the resistive shielding of downstream capacitance. Extending the Elmore's based work, several authors have proposed metrics based on higher circuit moments. While more accurate these approaches do not conserve the simplicity of the Elmore delay. These metrics, using multiple moments of the transfer function, completely lost the additive property of the simple Elmore delay. As a result they can not be used for optimization such as buffer insertion or wire sizing. The operating mode of an interconnect line is to propagate a signal from the output of a transmitter (line input driver) to the input of the receiver (line output driver). The delay across an interconnect line is obtained as the sum of the contributions of the input driver, the line and the output driver. The important parameters, for the corresponding path, are the total delay value between the transmitter input and the receiver output, together with the transition time value of the signal at the receiver output. The accurate determination of this output transition time is of fundamental importance, it has a non-negligible contribution to the delay of the subsequent gate. These parameter values depend on the design of the drivers, their loading and the structure of the wiring. The primary contribution of this work is to propose a complete modeling of the resistance-capacitance (RC) effect on the delay of an interconnect link between two inverters. Considering a previously developed deep sub-micrometer model, in which the inverter (gate) is considered as a current generator, we have obtained an analytical expression allowing to estimate the line loading effect on the transition time and the delay of the input and output drivers. This results in a simple but accurate closed-form equation for estimating the delay in RC interconnect, while conserving the complexity level and the additive property of the Elmore delay. We establish the limiting conditions in using purely capacitive or RC representation of interconnect wire. Load shielding of the input inverter has been captured with respect to the input drive-line transition time ratio, which can be used as an efficient metric for considering shielding effects. The fundamental idea for developing this model is to define a metric for characterizing the resistive shielding of line capacitance. The main difference with other approaches is to evaluate the delay and the transition time of each driver, considered as a current generator, according to their structure, size and effective load. Validations with respect to transmission line simulations (ELDO) on 130nm process, have demonstrated the potential application of this model in estimating the RC interconnect impact on circuit performance. Applications to be considered in using this analytical model are in driver selection and in line repeater insertion.
- 173 -DCIS 2004
Scalable Substrate Modeling based on3D Physical Simulation
S. Fregonese, D. Celi*, T. Zimmer, C. Maneux, P. Y. SulimaLaboratoire de Microélectronique IXL,UMR 5818, University of Bordeaux 1,
33405 Talence, FRANCEPhone: 33 5 40 00 27 66 Fax: 33 5 40 00 27 66°email:[email protected]
* ST Microelectronics, 850 rue Jean Monnet, BP 16, F-38 926 Crolles Cedex, FRANCE
Abstract—The impact of substrate with deep trenches on dc and ac HBT electricalcharacteristics has been studied using physical simulation. A SPICE geometry scalable model isproposed and compared with physical simulation results. Then, the model is implemented in a scalable HBT model based on HICUM level 0 (L0). The proposed model is applied onmeasurement and shows good agreement with the electrical characteristics as a function of the device emitter geometry.
Fig 1: 3D physical simulation structure
- 174 -DCIS 2004
Rodrigo Picosa, Miquel Rocaa, Benjamín Iñiguezb, Eugeni García-Morenoa
a Grup de Tecnologia Electrònica, Universitat de les Illes Balears, [email protected], +34-971-173-137 b Departament d’Enginyeria Electrònica, Elèctrica i Automàtica, Universitat Rovira i Virgili
irect determination of some important MOSFET parameters from experimental measurements
involves the evaluation of auxiliary functions that include, in many methods, the use of
derivatives. The numerical processes implicated introduce great deals of noise that difficult
determining them. Noise reduction filtering allows cleaning the derivative plots. We have
demonstrated a filtering method, and rules to find the filter parameters have been given. Results
obtained with transistors of submicron technologies show that the method is quite insensitive to the
filter parameters.
This method has been successfully applied to extract two different key parameters (threshold
voltage as depicted in Figure 1, and saturation voltage, as shown in Figure 2) in devices from two
different submicron technologies (ES2 0.7 µm and AMS 0.35 µm). The values extracted with this
method compare well with the ones obtained using other methods not based on derivatives. Our
procedure combines the advantage of being physically-based with the noise immunity of the most
recent extraction methods, not based on derivatives, without requiring costly computational effort.
Ge (
V-1
)
Vds (V)
AMS 0.35mm technologyT=300K W=10mm
Vgs=2.97V
Vgs=1.98V
Vgs=0.99V
Figure 1. The position of the maximum determines
the value of the threshold voltage (VTH). The figure
shows unfiltered (crosses) and filtered (circles)
second derivative of measured IDS vs VGS data.
Figure 2. The position of the maximum of the depicted
function (G function) versus VDS determines the value
of the saturation voltage (VDSAT)
Extraction of MOSFET Parameters Using Fourier-Space Techniques
D
- 175 -DCIS 2004
Impact of Deterministic Within-Die Variationon the Circuit Performance
in Nanoscale Semiconductor Manufacturing
Munkang Choi, Seyed-Abdollah Aftabjahani, Cheng Jia, and Linda Milor School of Electrical and Computer Engineering, Georgia Institute of Technology, Atlanta,
Georgia/U.S.A., [email protected]
As semiconductor technology advances into the nanoscale era and more functional blocks are added
into systems on chip (SoC), the interface between circuit design and manufacturing is becoming
blurred. An increasing number of features, traditionally ignored by designers are influencing both
circuit performance and yield. As a result, design tools need to incorporate new factors. One important
source of circuit performance degradation comes from deterministic within-die variation such as
lithography and Cu interconnect chemical mechanical polishing (CMP), so we have established a
methodology to consider systematic and deterministic variation from the proximity effect, lens
aberrations, flare, and CMP in circuit delay analysis and be able to reduce lithographic correction
work. Our methodology involves labeling the cell name by information for lens aberrations and flare.
Neighborhood information relating to the proximity effect is reflected in HSPICE files of cells by tags
on the transistor names. Cell transistor netlists with the modified gate poly geometries are combined
with interconnect RC netlists with modified metal geometries to generate revised cell HSPICE netlists,
which are used to determine revised delay tables.
The tool has been applied to analyze delays of ISCAS85 benchmark circuits in the presence of
imperfect lithography and CMP variation. First, the relationship among speed, leakage current, the
minimum CD, and lithography imperfections was extracted. After we control leakage current, delay is
affected by lens aberrations and flare, which are less likely to be averaged, but not the other effects.
Second, it is revealed that interconnect systematic variation results in a much smaller impact on delay
than gate poly variation since variations in interconnect resistance and capacitance partially cancel
each other. Third, it is found that delay variation, caused by the under-layer effect, is comparable to
that of the same-layer effect.
- 176 -DCIS 2004
Session 7d
Low Power / Low Voltage : analog circuits (2)
Friday nov. 26 8h30 10h00, Lacanau Room
Chairs
Gilles Jacquemod (U. de Nice) Claude Duvanaud (U. de Poitiers)
A Low-Dropout Voltage Regulator for Biomedical Integrated Systems
Rui E. Martins and F. Vaz University of Aveiro – DET/IEETA, Aveiro, Portugal,
[email protected] ; [email protected]
L INEAR regulators are mandatory in the power management of systems-on-chip for biomedical
applications. Biomedical signals which are used the most, like EEG (electroencephalogram), ECG
(electrocardiogram) or EMG (electromyogram), have voltages in the µV range and require analog
front-ends which are incompatible with the high noise and ripple levels of typical switching power
supplies. This paper presents a 0.8 V capacitor-free CMOS low-dropout regulator (LDO) for
biomedical systems. Such low power supply voltage operation is obtained by using a novel voltage
reference, avoiding transistor stacking and maintaining the circuit as simple as possible (see Fig. 1).
This strategy allows a low static current consumption, easy frequency compensation (as only 2 gain
stages are present), reduced silicon area and zero pin count, given that no external component is
required for correct operation (compensation capacitors have small values). The voltage reference is
based upon1, but the use of a low-voltage current mirror, resulted in a 0.2 V reduction in the minimum
power supply.
The proposed LDO has been implemented in a commercial 0.5 µm CMOS, three-metal, double-poly
technology, occupying 0.127 mm2. The maximum supply current is 7.5 µA, and the line/load
regulation is better than 1 % for an output current of 5 mA.
Figure 1. Schematic of the proposed LDO voltage regulator (Cc1 and Cc2 are integrated compensation capacitors)
1 Eric Vittoz and Jan Fellrath, “CMOS analog integrated circuits based on weak inversion operation,” IEEE J. Solid-State Circuits, vol. 12, pp. 224–231, June. 1977.
- 178 -DCIS 2004
A low-voltage, low power, wide-linear range subthreshold OTA
A. El mourabit, G. N. Lu, P.PittetLaboratoire d’Electronique, Nanotechnologies, Capteurs (LENAC), Université Claude
Bernard, Lyon E-mail : [email protected]
We present a new configuration of CMOS subthreshold operational transconductance amplifier (OTA)
for low-power, low-voltage and low-frequency applications. To overcome the problem of narrow
linear range due to subthreshold operation (less than 15 mV for 1% of transconductance variation in
case of a conventional differential pair), we employ floating gate MOS (FGMOS) transistors1, which
allow low voltage and provide wider range with lower gm. Furthermore, we implement a novel
linearization technique consisting of cancellation of cubic distortion term, which further extends the
linear range. This technique can also be applied to a conventional MOS pair, but with the use of
FGMOS devices, its implementation is facilitated. The proposed linearized OTA with subthreshold
FGMOS transistors is designed in a 0.8-µm CMOS process (AMS). For demonstration a monolithic
Gm-C, continuous-time, low-pass, second-order filter is built by using the proposed linearized OTA.
Simulation results of the filter show achievements of 76-dB linearity for a fully balanced input
dynamic range up to 1 Vpp at 1.5V supply voltage. For a tuning range between 10 and 100 Hz, the
power consumption of the filter remains lower than 2 µW. Second-order effects such as that of
FGMOS's parasitic capacitance are analyzed.
Iout
V+V- C1
C2
Vdd
C2
C1
C2
C1/2
C1/2
I1I2 I3
IbVbias
Vdd
Vp
Vn
M3
M2
M1
M9
M'9
M'7
M7M5M4
M6
M'6
M'8
M8
M0
Fig. 1. Proposed FGMOS linearized OTA
1 P. R. Gray, P. J. Hurst, S. H. Lewis and R. G. Meyer, analysis and design of analog integrated circuits, 4th ed.New York: Wiley, 2001.
- 179 -DCIS 2004
A Low Voltage I/O Interface for High Speed Buses in GaAs Technology
R. Esper-Cha´ýn, F. Tobajas, R. SarmientoInstituto de Microelectr´onica Aplicada, Universidad de Las Palmas de Gran Canaria
Campus Universitario de Tafira, 35017, Las Palmas, Spain fesper,tobajas,[email protected]
T HE achievable off-chip bandwidth is a crucial factor in the performance of digital systems. In
intra-system interfaces where both latency and bandwidth are important, highspeed buses have
been adopted as the most effective solution. In this paper, low-voltage I/O circuits for high-speed
buses in GaAs technology are presented. Operation above 600Mbps was demonstrated from
experimental measurements performed on different test chips.
- 180 -DCIS 2004
New low Voltage Class-AB CMOS Unity Gain Buffer and Current Mirror
A. Torralba, R. G.Carvajal, M.Jiménez, F. Muñoz, J. Ramírez-Angulo
C LASS-AB circuits, which are able to deal with currents which are orders of magnitude larger than their quiescent current, are good candidates for low-power analog design. This paper presents
a new, simple, low-voltage class- AB unity-gain buffer, based on the Flipped Voltage Follower cell. This buffer can be used in many applications, and a new low-voltage class-AB current mirror based on the proposed buffer is also presented. Simulation and experimental results are provided.
- 181 -DCIS 2004
New Low-Voltage Class AB/AB CMOS Op- Amp with
Rail-to-Rail Input/Output Swing Milind-Subhash Sawant1
, Shanta Thoutam1,4 , Jaime Ramírez-Angulo1,Antonio. J. López-Martín2 and Ramon G. Carvajal3
1 Klipsch School of Electrical and Computer Eng., New Mexico State University, Las Cruces, NM
2 Dept. of Electrical and Electronic Engineering, Public University of Navarra, Pamplona(Spain),
3Escuela Superior de Ingenieros, Universidad de Sevilla, (Spain),4 Freescale Semiconductor Inc. (Motorola) Austin, TX
A new low-voltage CMOS class AB/AB fully differential op-amp with rail-to-rail input/output
swing and supply voltage of less than two VGS drops is presented. The scheme is based on
combining floating gate transistors, and class AB input and output stages. The op-amp is characterized
by very low static power consumption and enhanced slew-rate. Moreover the proposed op-amp does
not suffer from typical reliability problems related with initial charge trapped in the floating gates
devices. Simulation and experimental results in 0.5µm CMOS technology verify the scheme operating
with 1.8V single supply and close to rail to rail input and output swing.
- 182 -DCIS 2004
A New Family of Low-Voltage Power-EfficientClass AB CMOS OTAs
Sushmita Baswa1, Antonio J. López-Martín1,2, Jaime Ramirez-Angulo1, Ramón G. Carvajal1,3
1 Klipsch School of Electrical Eng., New Mexico State University, Las Cruces, NM, USA. 2 Dept. of Electrical and Electronic Eng., Public University of Navarra, Pamplona, Spain
3 Escuela Superior de Ingenieros, Universidad de Sevilla, Spain (e-mail: [email protected])
A novel family of low-voltage power-efficient class-AB CMOS Operational Transconductance
Amplifiers (OTAs) is described. It is based on the combination of adaptive biasing techniques and
resistive local common-mode feedback (LCMFB), which provides increased dynamic current boosting
and gain-bandwidth product (GBW). Various adaptive biasing schemes are combined with LCMFB,
leading to different class-AB OTA topologies, shown in Figure 1. A 0.5- m CMOS implementation of
three different OTAs based on this technique shows enhancement factors of slew-rate and GBW of up
to 280 and 3.6 respectively for an 80-pF load compared to a conventional class A OTA with the same
quiescent currents and supply voltage, with little overhead in silicon area, noise, and static power
consumption. The circuits can find application in low-voltage low-power switched-capacitor circuits
and in buffers for testing mixed-signal circuits.
R1VINM VINP
M9
M1
M10 M11 M12
M7 M8
M2R2
VOUT
IBIAS
M4
M5
M3
IBIAS
M6
R1VINM VINP
M9
M1
M10 M11 M12
M7 M8
M2R2
VOUT
IBIAS
M4
M5
M3
IBIAS
M6
R1VIN- VIN+
M9
M1
M10 M11 M12
M7 M8
M2R2
VOUT
IBIAS
M4
M5
M3
IBIAS
M6
M13
R3
-+
R4
VIN- VIN+
M9
M1
M10 M11 M12
M7 M8
M2
VOUT
M5
M3
IBIAS
IBIAS
M13
M14
M6
M4
IBIAS
M15
IBIAS
M16
IBIAS
R1 R2
R3 R4
(a) (b)
(c) (d)
M17 M18 XX
Figure 1. Family of Class AB OTAs
- 183 -DCIS 2004
Session 8a
Sigma-Delta Modulator Design (2) Friday nov. 26 10h30 11h30, Bordeaux Room
Chairs
Ramón Gonzalez-Carvajal (U. de Sevilla) Adoración Rueda (IMSE-CNM)
Continuous-time modulator with exponential feedback for reduced jitter sensitivity
J. San Pablo, D. Bisbal, L. Quintanilla, J. Arias, L. Enríquez, J. Barbolla
Department of E. y Electrónica, E.T. S. I. Telecomunicación, Campus “Miguel Delibes”, Universidad de Valladolid, 47011-Valladolid, Spain, [email protected]
current-mode continuous-time Sigma Delta modulator with reduced jitter sensitivity has been
analysed and designed in a 0.35 m CMOS technology. The complete modulator has been
implemented following a current-mode approach and some characteristics are: integrators blocks
present inherent low input impedance without the need of feedback, good stability and no necessity for
common-mode circuit. To our knowledge, there is not any reference in literature of previous CT
modulators implemented completely in current mode. Timing error associated to clock jitter
introduces noise into the in-band spectrum. The tolerable level of clock jitter decreases with increasing
the oversampling ratio, and eventually jitter noise power will exceed quantization noise power. In
addition to that, clock jitter affects Return to Zero/Half Return to Zero modulators –which show a
reduced sensitivity to loop delay and, thus, are used to implement practical Continuous Time Sigma
Delta modulators- more severely than modulators employing Non Return to Zero feedback. Thus, for
high speed and/or high accuracy converters, this represents a serious challenge to chip designers. Jitter
rejection improvement was achieved by using an exponential decaying feedback current. This
reduction has been achieved using exponential-feedback waveforms generated by a switched
capacitor-based DAC (Fig. 1). Simulations show a substantial improvement on jitter rejection with
respect to the conventional rectangular-feedback DAC. Functional and transistor-level simulations
have been carried out and the corresponding results are presented. A dynamic range of 67 dB
(resolution of 10.8 bits), has been achieved for a second order modulator with an oversampling ratio of
64 and 1 MHz of bandwidth (sampling frequency of 128 MHz). In addition to distortion, quantization
noise, clock jitter, thermal, and flicker noise have also been considered. It consumes a power of 2.9
mW at a supply voltage of 2.5 V.
A
R
rzn rzn rzprzp
R
C C
Vref+ Vref-
Iop Ion
prech prech
Fig. 1. HRZ exponential DAC. Differential exponential feedback waveforms have been included.
- 185 -DCIS 2004
Implementation of an RTZ code for feedback DAC on a D modulator
HIS paper presents a detailed study of the return-to-zero (RTZ) feedback code incidence
in continuous-time modulators, including a high-level modelisation for simulation.
Such results derived from the analytical system analysis can be applied to design continuous-
time modulators with an arbitrary feedback waveform while the behavioural model
explained, is a useful tool to speed up high-level simulations when a RTZ code is used,
increasing about ten times simulation speed. Simulation results on a four-order, log-domain
modulator are presented to demonstrate the validity of the proposed models (Fig. 1).
Fig. 1 PSD obtained from the Matlab model, with RTZ feedback code simulation (left) and with the new proposed equivalent model (right)
Jofre Pallarès1, Xavier Redondo1, Francesc Serra-Graells1, Justo Sabadell2
[email protected], [email protected], [email protected] and [email protected]
1 Institut de Microelectrònica de Barcelona-CNM, Spain2 Barcelona Branch Office, Epson Europe Electronics, GmbH
T
- 186 -DCIS 2004
Excess-Loop delay reduction on Low-OSR High-Speed Multi-bit Continuous-Time Sigma-Delta Modulators
Susana Paton, Thomas Pötscher1, Antonio Di Giandomenico1, Klaus Kolhaupt1,Luis Hernandez, Andreas Wiesbauer1, Martin Clara1, Ramon Frutos
Univerisdad Carlos III de Madrid, Spain 1 Infineon Technologies, Design Center Villach, Austria
HIS paper evaluates two techniques to improve the linearity of the main feedback D/A converter
in Multi-Bit Continuous-Time Sigma-Delta modulators (CT-SDM). A Self-Calibrated Current-
Steering (SCCS) implementation of the D/A converter is compared to the usage of a Data Weighted
Averaging (DWA) algorithm on the selection of uncalibrated D/A-elements1. A test-chip including the
two different solutions is presented and measurement results are compared.
The test chip implements a 4th order 4-bit CT-SDM in 0.13 m. It has an analog bandwidth of 15MHz
and an OSR of 10.
The circuit implementation of the DWA algorithm is composed by a shifting-block and a pointer-
calculator. The SCCS-D/A uses a continuous dynamic background calibration of the current sources2.
Fig. 1 shows the measured output power spectrum of the modulator using both SCCS D/A and DWA
D/A. It is shown that the use of SCCS D/A has better SNR. The DWA introduces some excess loop-
delay that reduces the achievable dynamic range. The latency of the shifting-block is the main
contributor to the added delay.
Figure 1. Measured FFT of the two modulators
1 Steven R. Norsworthy, Richard Schreier, Gabor C. Temes, “Delta-Sigma Data Converters: Theory, Design and Simulation”, Wiley-IEEE Press, 1996 2 D. W. J. Groeneveld, H. J. Schouwenaars, H. A. H. Termeer, C. A. A. Bastiaansen, “A Self-Calibration Technique for Monolithic High-Resolution D/A Converters”, IEEE J. Solid-State Circuits, vol. 24, Dec. 1989
T
- 187 -DCIS 2004
Discrete Invariant Set Algorithm for Sigma Delta Modulators Dynamics Analysis
D. Camarero de la Rosa, V-T. Nguyen, J.F. Naviner, P. LoumeauENST Paris, 46, rue Barrault, 75634 Paris, Cedex 13, France. E-mail : [email protected]
his paper presents a new tool to study Σ∆ (sigma delta) modulator dynamics. We call dynamics
the behavior of the integrator states that constitute these modulators. The invariant set concept has
already been used in order to prove Σ∆ stability1. The algorithm presented here also uses the invariant
set concept, but exploited in a different way in order to predict state bounds as exact as desired.
Studied inputs are constant inputs with additive noise whose maximal amplitude is finite.
The main idea is to approximate the input signal rather than the Σ∆ modulator. So, an input of real
numbers with a limited swing is approximated by a finite set of rational numbers, spaced by a distance
ε. Under an input of this nature, the only possible Σ∆ states are confined into a rational grid if the
integrators gains are not irrational numbers. This property is exploited as described in the paper to find
an invariant set of Σ∆ states associated to the approximated input. When ε→0, the approximated input
becomes closer and closer to the input of real numbers. The observed states bounds also seem to
converge on a limit value: the states bounds really associated to desired input.
The predicted bounds obtained in this way seem to be more accurate than others in the bibliography.
1 R. Schreier, M. Goodson, B. Zhang, “An Algorithm for Computing Convex Positively Invariant Sets for Delta-Sigma Modulators”, IEEE Transactions on Circuits and Systems I: Fundamental Theory and Applications, vol. 44, no. 1, January 1997.
T
- 188 -DCIS 2004
Session 8b
Digital Test Friday nov. 26 10h30 11h30, St Emilion Room
Chairs
Mar Martinez (U. de Cantabria) José Silva Matos (INESC-Porto)
Optimisation of digitally coded test vectors for mixed-signal component
Ahcène Bounceur, Salvador Mir and Emmanuel Simeu
Reliable Mixed-signal Systems Group - TIMA Laboratory 46, Avenue Félix Viallet, Grenoble 38031 Cedex, France
Tel: +33 4 76 57 48 04 – Fax: +33 4 76 47 38 14 Email: [email protected] – [email protected] – [email protected]
HIS article proposes a technique to optimize test patterns targeting Analogue and Mixed-Signal
(AMS) cores in System-On-Chip (SoC) devices. In order to render the test of these cores
compatible with the test of digital ones and with the use of low-cost testers, the analogue test patterns
are digitally coded. They can then be scanned into the chip where they are easily converted into the
required analogue test patterns. A Computer-Aided Test (CAT) tool is used to optimize the digital
coding of the desired analogue test signal. Several multi-objective optimization algorithms are
considered to carry out this task, including Monte-Carlo, W.A.R.G.A (Weighted Average Ranking
Genetic Algorithm) and N.S.G.A (Non-dominated Sorting Genetic Algorithm). The results obtained
with these different algorithms are illustrated and compared by using different analogue test signals.
Figure 1. Graphical interface of the optimization tool
23 24 25 26 27 28 29 30 31 32−7
−6
−5
−4
−3
−2
−1
0
SFDR
AD
ev
(SFDR,ADev) − 3−Tone Signal − (30 sec)
NSGA
Monte−Carlo
Pareto Set
Figure 2. Example of solutions given by N.S.G.A and Monte-Carlo algorithms
T
- 190 -DCIS 2004
Improving the Efficiency of Arithmetic BIST by Combining Targeted and General Purpose Patterns
S. Manich, L. García, L. Balado, J. Rius, R. Rodríguez, J. Figueras Universitat Politècnica de Catalunya
Diagonal, 647, P9, 08028 Barcelona, SPAIN [email protected]
RITHMETIC additive test pattern generators (AdTPGs) are now being proposed as an alternative
to linear feedback shift registers (LFSRs) because of their reduced area overhead impact. AdTPGs
allow existing internal datapaths (Figure 1) to be reused to perform the excitation and observation of
potential faults in the circuit without a penalty in the circuit area. As in the case of LFSRs, the
compactness of the information required in memory and the test generation time needed to achieve a
specified fault coverage level (FC) have a significant impact on the necessary resources and thus on
the quality of the test.
In this paper, a new approach to the preparation of the test is proposed. It has been observed that, if
two different and independent preparation methodologies, TPP (Target Purpose Pattern) and GPP
(General Purpose Pattern), are combined, a better quality of the test is obtained than that achieved by
applying them separately. These two types of strategies have their own advantages, focused on the two
different types of faults. TPP is suited for random pattern resistance (rpr) faults while GPP is suited for
non-rpr faults generating patterns with low memory impact. It has been shown how to combine these
strategies in order to reinforce the efficiency of the complete test. ISCAS benchmark circuits have
been selected to compare the results with other existing methodologies. Experiments show that the
combined strategy, named LUCSAM+, improves previously published results. If the same fault
coverage is assumed, the average reduction in memory is 27% and the average reduction in number of
test vectors is 44%.
A
Increment
Adder
Accumulator
Test Vectors
Triplet 1Triplet 2Triplet 3Triplet 4
Triplet k
MEMORY
···
DATAPATH
Figure 1. - Datapath structures in the IC are reused to implement AdTPGs. The area overhead impact is reduced.
- 191 -DCIS 2004
Automatic Verification of RT-Level Microprocessor Cores Using Behavioral Specifications: a Case Study
E. Sanchez*, M. Sonza Reorda*, G. Squillero*, R. Velazco†
* Politecnico di Torino, Dip. Automatica e Informatica, Torino, Italy, edgar.sanchez, matteo.sonzareorda, [email protected]
† TIMA-CMP Laboratory, Grenoble, France, [email protected]
The massive diffusion of system-on-a-chip (SoC) and custom cores make processor obsolescence an increasingly complex problem for industrial embedded computer users. Obsolescence of electronic components affects many safety critical applications, active years longer than it was originally anticipated. The current industrial practice is to replace some of the obsolete parts with more modern ones, redesign small portions of the SoCs or of the application specific integrated circuits (ASICs) to meet the new requirements. Sometimes, the obsolete processor is re-implemented by means of programmable logic devices such as FPGAs. However, verifying that the new version is equivalent to the original hardware one is a harder task.
The paper tackles this problem, proposing an almost fully automatic methodology to check the correctness of a customized microprocessor core by comparing it to a reference model. A reference model can almost always assumed available: it could be a high-level instruction set simulator (ISS), the FPGA implementation of the original core, or even the original design. The reference model is used as a black box to compare behavior, while the process is driven by the knowledge of the customized version alone. The internal description of the original core is not exploited.
testprogram Comparator
µGPRT-LevelSimulator
testprogram
fitness
BehavioralSimulator
behavior behavior
testprogram
testprogram
criticaltests
Figure 1. Proposed Approach
The proposed methodology exploits an evolutionary technique called µGP for automatically generating assembly programs. These test programs are used two times: first they are simulated by a standard RT-level simulator and by an external tool. The expected behavior is eventually compared with the one extracted from the RT-level simulator. The reference model has very few requirements, it does not even need to be cycle-accurate, and it used as a mere black-box.
The approach was tested against a customized version of an 68hc11, where a limited number of unnecessary functionalities were removed. The Motorola 68hc11 is a striking example of small microprocessor core that is exploited in a wide range of custom applications, and several industries modified the original design, implementing it as an FPGA to add specific features, or remove unneeded ones. The proposed methodology was able to devise a compact set of 24 short test programs uncovering 4 potential problems.
- 192 -DCIS 2004
Solving the State Justification Problem using MILPfor RTL Specifications
H. Navarro, Juan A. Montiel Nelson, J. Sosa & José C. García IUMA, Institute for Applied Microelectronics. Integrated Systems Design Division
Department of Electronic Engineering and AutomationUniversity of Las Palmas de Gran Canaria, Las Palmas, E 35017, Spain.
hnavarro, montiel, jsosa, [email protected]
T HE satisfiability problem (SAT) for RTL descriptions has many direct applications in the
electronic design automation (EDA) arena. The majority of industrial hardware verification tools
uses bit level decision procedures, like SAT or BDD based techniques. Unfortunately, these
approaches are not efficient enough, because they do not inherit the word level information from the
RTL design. Most recent approaches to the SAT problem1 are addressed to RTL designs containing
instances of word level arithmetic blocks and bit level Boolean logic. They transform the whole SAT
problem into a mixed integer linear program (MILP) that must be solved externally by a MILP solver.
A complete solution of the SAT problem for RTL descriptions must also provide support for both,
word-level operators in the data flow, and finite state machines (FSMs) for control flow. As the output
of an FSM depends on input values and its actual state, to satisfy an output condition on a FSM
involves solving a state justification problem, i.e. to find a right sequence of states and input values. In
such problems, there exist an optimal state sequence that requires a minimum number of clock cycles.
This paper presents a new approach that automatically solves in a single step, the optimum
input sequence applicable to a given RTL description to reach a desired state. This is accom-
plished by applying a novel time frame expansion method for state justification that guarantees an
optimized solution and avoids performing time frame expansions iteratively.
Experimental results demonstrate that the proposed methodology can solve any state justification
problem in one step for complex FSMs. For this purpose, a basic processor with a reduced set of
arithmetic and logical instructions was selected as a generic FSM. The state justification problem lies
in choosing the best sequence of instructions to establish a desired value into the accumulator, being
zero its initial value.
1 Z. Zeng, P. Kalla and M. Ciesielski, “LPSAT: A Unified Approach to RTL Satisfiability”, Proc. DATE’01, p-398-402, 2001.
- 193 -DCIS 2004
Session 8c
IP-based design Friday nov. 26 10h30 11h30, Auditorium
Chairs
Patrick Garda (U. Paris 6) Juan Carlos Lopez (U. de Castilla-La Mancha)
Hiding Technique for Intellectual Property Protection on FPGAs
L. Parrilla, E. Castillo, A. García and A. LlorisDepartment of Electronics and Computer Technology, University of Granada,
18071 GRANADA (Spain) lparrilla, ecastillo, agarcia, [email protected]
HE Intellectual Property Protection (IPP) of reusable design modules (IP cores) to be
implemented over FPGAs are becoming a problem with the expansion of this design strategy. In
this paper, a new procedure for hiding a digital sign to provide Intellectual Property Protection (IPP)
of circuits based on the Residue Number System (RNS) to be implemented over FPL devices is
presented. The aim is to protect the author rights in the development and distribution of reusable
modules by means of an electronic signature embedded in the FPGA design. The procedure described,
is oriented to circuits based on the RNS, but can be easily extended to any system to be implemented
on FPGAs.
As an example, a CIC-RNS filter, a 128-bits signature identifying both the origin and the recipient
of the design is embedded. The proposed structure allows the protection of the filter without any
penalties in performance circuit. Design examples were implemented using the Virtex2 family devices
of Xilinx. Table 1 compares CIC filter with the CIC signed filter proposed in this paper, both filters
built in RNS-FPL. The table shows the speed grade, the number of SLICEs, the maximum frequency,
the area increase and the speed reduction for the RNS-based study filters. The analysis of the results
shows that the area increase for the signed filters is 25 SLICEs (5.95%). Nevertheless this increase is a
fixed quantity, and in RNS circuits of major area, as the wavelet transform, the additional percentage
of CLBs to the signature embedding and extraction would suppose a minor percentage that the
obtained in the example. In addition, the table shows that there is no penalizations in performance.
T
TABLE I Summary of simulation results
RNS CIC filter RNS CIC signed filter Speedgrade SLICEs Fmax
(MHz) SLICEs Areaincrease
Fmax (MHz)
Speedreduction
-6 420 106.85 445 5.95% 107.19 -0.32%
-5 420 101.34 445 5.95% 103.50 -2.13%
-4 420 88.63 445 5.95% 91.26 -2.97%
- 195 -DCIS 2004
Rapid Integration of IPs in System On Chips
Salim OUADJAOUT M3Systems Inc
1, rue des Oiseaux 31410 LAVERNOSE, France
Dominique HOUZETI.E.T.R Laboratory, INSA-RENNES
20, Av. Des buttes de coësmes35043 RENNES, France [email protected]
I ncreasing system complexity and tight design deadlines due to time to marketimperatives mean that the designers have to achieve the impossible, on a daily basis.
Today, virtual components can be designed to be portable from an architecture to another and protect the investment of making them reusable. This paper presents our work on a SoCprototyping methodology focusing on the hardware design. This methodology is based on automatic high level modelled IP (Intellectual Property) integration. We also describe our IP interconnection methodology that allows a rapid integration of communication between the functional IPs through a custom SystemC channel library. IP integration is performed by standardized IP interface synthesis (VCI/OCP standard). The interface synthesis allowsheterogeneous IPs to communicate in a plug-and-play fashion in the same system.
- 196 -DCIS 2004
Block constraints budgeting in timing-driven hierarchical flow
Olivier Omedes1, Michel Robert2, Mohamed Ramdani3
1 Cadence Design Systems, Valbonne, France, [email protected] LIRMM, UMR CNRS/Université de Montpellier II, Montpellier, France, [email protected]
3 ESEO, Angers, France, [email protected]
A NY physical synthesis solution has limitations on the size of circuits that can be handled in a
single run. “Divide and conquer” approaches, or hierarchical flows, have been introduced to
overcome these limitations. In these flows, chips are sub-divided into smaller blocks. In order to drive
optimizations of these blocks, blocks IO constraints have to be computed. That’s what a block
constraints budgeting algorithm aims to do.
This paper introduces a new block constraints budgeting that speeds up timing closure in timing-
driven hierarchical flows. Existing block budgeting approaches show two main issues. First, they do
not take into account the design logic flexibility; that is to say optimization possibilities such as
buffering, resizing, restructuring, … Second, they treat separately timing constraints (arrival and
required times) and electrical constraints (drive strength, output load, …) whereas they are closely
linked. In the proposed approach, logic flexibility awareness is obtained using a logical effort
modeling1 and some simple restructuring or rebalancing algorithms. Logical effort theory is also used
to better correlate timing and electrical constraints.
The proposed algorithm, called “Flexibility Aware Budgeting” (FAB) has been compared to some
common used budgeting approaches (IMP_T and CPB)2. Experiments based on commercial EDA
tools and real designs show up to 55 % reduction in hierarchical flow run time and lead to a good flow
timing closure.
Approach CPU gain MEM gain Final Slack CPB 66 % 37 % - 0.98 nsIMP_T 66 % 35 % - 0.40 nsFAB 60 % 31 % - 0.25 ns
Table 1. Run time and memory gains, and final slack at the end of the first hierarchical flow iteration.
1 I. Sutherland, B. Sproull, and D. Harris, “Logical Effort: Designing Fast CMOS Circuits”, MORGANKAUFMANN PUBLISHERS, 1999.2 See full paper for IMP_T and CPB descriptions.
- 197 -DCIS 2004
Flexible HW/SW Implementation of MPEG systems using FPGA platforms
Navas O., Portero A., Escrig J., Bonamusa M., Novo D., Carrabina J.
Microelectronic/Computers Science, Universitat Autonòma de Barcelona, Bellaterra, Catalonia/Spain,oscar.navas, antoni.portero, [email protected],
[email protected], [email protected], [email protected]
his paper presents a new HW/SW co-design methodology for applications dealing with real time
video compression based on the use of reconfigurable platforms and embedded processor cores
(IPs) for SoC design.
In order to select the right solution during systems prototyping, we use Matlab environment with
FPGA toolboxes, in order to model, design and verify systems’ performances. Processors cores (either
soft or hard IPs) execute tasks with real-time constrains like building transport streams, managing QoS
through configuration parameters. Besides, intensive real-time data processing tasks are implemented
using special-purpose HW.
A demonstrator based on real-time MPEG video compression has been designed and validated. It
implements video coding using the standard ISO/IEC 13818-2 | ITU-T H.262H (also know as
“MPEG2 Video”), for the Main Profile at Main Level.
We use as starting point, descriptions of video standard models (i.e. ITU-T H.262H), using the
standard source software (written in C). The main advantage of a system level description using
C/C++ programming language is that allows reusing “good” software code as well as their verification
environments.
The HW/SW partition is done after computational complexity analysis of the system in terms of:
estimated number of operations per second, number of clock cycles. It gives the viability of SW
implementation of the algorithms in the softcore NIOS.
Intensive real-time data processing tasks are implemented using special-purpose HW with the
Matlab environment with FPGA toolboxes.
T
- 198 -DCIS 2004
Session 8d
Low Power Digital Design Friday nov. 26 10h30 11h30, Lacanau Room
Chairs
Richard Grisel (U. de Rouen) Juan A. Montiel Nelson (U. Las Palmas de Gran Canaria)
Analytical Estimation of Node Activity in Ripple Carry Binary Adders
A. Calomarde, A. Rubio Department of Electronic Engineering Universitat Politècnica de Catalunya
THIS paper present a closed-form evaluating the number of transitions in main nodes of a 2-level
ripple carry adder as basic circuit in array multipliers, with random inputs. The different transition
causes are considered and classified and has also been taken into account in the derivation of the
expressions. This fact allows the determination of the energetic weight of each class of transition. If
the correct energy is assigned to each class of transition the expression is accurate to evaluate the
energy consumption of such structures. The effect of spurious caused by unbalanced path delays is
also evaluated in a closed-form under the assumption of a given spurious generation mechanism.
- 200 -DCIS 2004
Increase in Energy Consumption due to Multiple Transitions in Coupled Lines
Eugeni Isern, Miquel Roca, Department de Física, Universitat de les Illes Balears, Palma de Mallorca, SPAIN (e-mail: eugeni.isern, [email protected]).
Francesc Moll, Departament d’Enginyeria Electrònica, Universitat Politècnica de Catalunya, Barcelona, SPAIN, (e-mail: [email protected]).
HE paper is devoted to the study of the dissipated power contribution of crosstalk when two
coupled lines make simultaneous or relatively delayed transitions. The importance of the delay, the
line length and the relative driver strength is analyzed through electrical simulation. A typical coupled
structure in a 0.18µm CMOS technology has been considered (Fig. 1). Electromagnetic (in order to
obtain parasitic interconnect parameters) and electrical (to estimate power dissipation) simulations
with HSPICE have been performed. Opposite direction transitions with zero delay produces the larger
increase in energy, while same direction transitions imply a saving in the energy. As a numerical
example, for a 100µm line length the difference between this maximum and minimum energies is
about 143fJ, which represents a 54% reduction. This fact may be used, for instance, to drive
algorithms for coding line transitions in data buses. From the analysis of the line length dependence, it
can be seen that for short lines (<50µm) intrinsic device contributions (shortcircuit, leakage, etc.)
clearly dominates, whereas for larger lines (>100µm) a region exists where the dissipation is mainly
due to charging and discharging the lines, including the coupling capacitor (Fig. 2). This range of
length lines is quite typical in medium to large sized circuits, and therefore crosstalk contribution must
be seriously considered in power estimation tools if accurate predictions are pursuit.
Vi2
Vi1
V1
V2
ic2
ic1
ip2
in2
ip1
in1
C12
C1
C2
i2
i1
i12VDD1
VDD2
line 2
line 1
Cc2
Cc1
10,0
100,0
1000,0
10000,0
1 10 100 1000Length (um)
Ene
rgy (
fJ)
nmos1 pmos1 nmos2 pmos2 total
Fig. 1. Circuit used to model the structure of two coupled lines with inverter drivers.
Fig. 2. Energy dissipated in the driver transistors, in front of the line length (opposite direction transitions).
T
- 201 -DCIS 2004
Power Characterization of RAMs. An Experimental Approach.
Javier Rellán, José L. Ayala, Marisa López-Vallejo Departamento de Ingeniería Electrónica
Universidad Politécnica de Madrid, Spain jrellan,jayala,[email protected]
T HE work presented in this paper shows a characterization methodology for memory
architectures based on circuit simulations. The proposed approach takes into account second order
effects like the effect of internal capacities, not previously considered in analytical models. It also
provides a set of design guidelines to lead the system designer on the electronic design with low-
power constraints from the very early stages.
The power characterization method we follow is based on the following assumptions: First, the
source of power dissipation in this model is the charging and discharging of capacitative loads caused
by signal transitions. Therefore, only dynamic power consumption is considered, even though this
information could be easily extended with the static power information provided by Spice. Second, the
energy consumption in every transition is proportional to the rising and falling times of the signal.
Thus, the experimental method will concentrate on measuring these times.
Our experimental methodology can be summarized as follows: First, a set of different memories are
generated. In order to evaluate either SRAM and DRAM technologies, both kind of schematic
representations have been created. Next, test vectors are also generated to perform all the interesting
logical transitions in the circuits to be evaluated. Then, the Spice simulator included within the
Cadence environment is used to collect the results of the electrical simulations of such circuits.
Finally, and after the careful analysis of these results, a set guide-lines is established for guiding the
circuit and system designer in the efficient design and use of the memory hierarchy when low-power
constraints are involved.
Finally, this research work provides some important rules to the system designer that have to be
remembered when planning the memory hierarchy: The underlying topology (rows and columns) has a
strong impact on the power consumption of the device. For equal memory size, memory rows
containing more than one data word can reduce energy consumption. Memories that split the data
word between more than one row can show lower energy consumption. However, in those cases, the
energy dissipated in the bus during the extra access has to be evaluated. The memory architecture has
to be considered from the very early design stages to make feasible the energy reductions and the
architecture adaptation.
- 202 -DCIS 2004
HE design of more and more complex, integrated and fast circuits implies to manage trade-off
between speed, power and area. This can be achieved with circuit simulators and critical path
analysis tools to modify iteratively the size of the transistors until complete constraint satisfaction.
More general speed-up techniques involve buffer insertion and logic transformation. If these
techniques may be found efficient for speeding-up combinational paths they may have different
impacts in the resulting power dissipation or area. Gate sizing is area (power) expensive and, due to
the resulting capacitive loading effects, may slow down adjacent upward paths. This implies complex
and iterative timing verifications. Buffer insertion preserves path interaction but is only efficient for
relatively highly loaded nodes. To manage these alternatives it is necessary to evaluate and compare
the performance of the different implementations. Without using any robust indicator, selecting
between all these different techniques for the various gates of a library is NP complex and induces
more iterative attempts which are processing time explosive.
A reasonable selection of speed-up technique must be based on a characterization of the available
speed on a critical path, on the determination of the critical nodes and the characterization of the gate
sensitivity to the sizing or buffering alternatives.
Based on a realistic model for gate timing performance, the main contribution of this paper is to
define different metrics for path characterization, transistor sizing and buffer insertion, to be used as
efficient indicators for characterizing the logic gates in terms of sensitivity to the sizing and buffering
techniques. We propose a method for determining the minimum delay, Tmin, achievable on a path.
Then we define, at gate level, the fan out limit for buffer insertion, Flimit. Flimit is used to determine
the path critical nodes and Tmin, to select between sizing and buffer insertion alternatives. We define
a gate sensitivity factor "a", to distribute the delay constraint, allowing path optimization at provably
minimum area cost. These metrics are used to define a general path optimization protocol that is
implemented in an optimization tool based on an accurate representation of the physical abstraction of
the layout (POPS: Performance Optimization by Path Selection). We have developed this tool to give
facilities in analyzing and optimizing combinatorial circuit paths in sub micron technologies.
Validation on various benchmark circuits demonstrate the validity of the defined boundaries for
selecting between the different optimization alternatives.
Optimization Protocol based on Performance Metrics
X. Michel, A. Verle, N. Azémard, P. Maurine, D. Auvergne LIRMM, UMR CNRS/Université de Montpellier II, (C5506),
161 rue Ada, 34392 Montpellier, France azemard, pmaurine, [email protected]
T
- 203 -DCIS 2004
Session 9a
Logic and Architectural Synthesis Friday nov. 26 14h00 15h15, Bordeaux Room
Chairs
Raoul Velazco (I.N.P.Grenoble)Celia Lopez (U. Carlos III)
Assertion Checking of Cyclic Behavioral Descriptions I. Ugarte, P.Sanchez
Microelectronics Engineering Group. TEISA Department. ETSIIT. University of Cantabria Avda. los Castros s/n. 39005 Santander. Cantabria. Spain
ugarte, sanchez @teisa.unican.es
I N order to confront the verification of more and more complex Systems, several Design-for-
Verification methodologies (DFV) have been proposed. One of them, Assertion-based Verification
(ABV) has recently emerged as the functional verification methodology capable of keeping pace with
increasingly complex systems.
This paper presents a static assertion checking technique for hardware behavioral models, which are
modeled with polynomials. The algorithm generates vectors automatically to detect the violation of the
assertion. If no counter-example is found, the assertion is fulfilled by the description. The technique is
based on a modified Interval Analysis (MODIA) and it reduces the verification effort because there is
no need to explicitly unroll loops.
In order to validate the proposed technique, a set of examples have been proposed. These are
executed by the tool ‘SMV’ and the proposed Assertion Checker. The results are shown in Table I.
The tool ‘SMV’ needs to unroll the loops to handle them, while the proposed tools handle loops
without unrolling.
TABLE I
Comparison with property checkers.
SMV AssertionChecker
Number of Evaluated Iterations
Linear < 1s 1 s 10Nonlinear 4.42 s 1 s 5
Cyclic descriptions
The advantage of this method is the efficiency of handling data-dominated algorithms independently
of the range of the data and it can be directly computed over the Control Data Flow Graph. However,
the main disadvantage is the explosion of the number of paths with the number of ‘if-then-else’
structures.
During cyclic description verification, the algorithm looks for possible input combinations that
violate an assertion taking into account all conditional paths. Thus, the memory consumption grows
when the number of iterations increases. In future work, the depth-first search will be implemented to
solve this problem. Additionally, heuristic metrics based on statistical probabilities will be used to
choose the path with highest probability to reach a violation.
- 205 -DCIS 2004
Behavioural Synthesis for Low Power Applying Operation Transformations
María C. Molina, Rafael Ruiz-Sautua, José M. Mendías, Román Hermida Dpto. Arquitectura de Computadores y Automática
Universidad Complutense de Madrid - Spain, [email protected], [email protected],[email protected],[email protected]
complete high-level synthesis algorithm specially suited to reduce power consumption in data
dominated applications is presented. It performs jointly the scheduling and allocation of
behavioural specifications In addition to classical low power methods, it implements novel design
strategies to reduce the datapath power consumption. These new features comprise the successive
transformation of the specification operations until a circuit implementation with minimum power
consumption in functional and storage units is reached. Circuit power consumption is minimized first
by balancing the power consumed per cycle, and secondly by maximizing the bit-level reuse of
datapath HW resources (making possible the disability of the maximum number of datapath FUs per
cycle). The algorithm performs these transformations trying to balance the power consumed per cycle
(due to the execution of the operations), allowing only the execution of operations over functional
units of their same width. To do so, some specification operations are successively transformed into
sets of narrower ones whose types and widths may be different from those of the original operation. In
consequence, some of the specification operations are executed during a set of non-necessarily
consecutive cycles and over a set of narrower functional units, linked by some glue logic to propagate
partial results and carry signals as necessary. Experimental results on some datapath intensive designs
show significant improvements in both power consumption and area reduction over conventional low-
power scheduling algorithms, as shown in Table I. In general, the power consumption and datapath
area reductions grow with the specification heterogeneity (number of different operation types and
widths present in the specification divided by the number of operations).
A
Table 1. Power consumption estimations and area of some synthesized examples
Specification Features Power Consumption (pF) Area (# equivalent gates) # Ope. # Adds. # Mults. # Widths Latency Synopsys Low Power Ours Synopsys LowPower Ours
10 6 4 3 3 3714 2376 1298 328.75 332.9 248.4 20 12 8 4 5 5120 3190 1903 443.45 461.6 305.34 35 23 12 5 7 7668 4533 2884 645.23 590.67 429.3 50 40 10 7 10 9730 6089 3675 822.3 810.41 467.31 70 55 15 8 12 15101 9834 5693 1244.87 1203.5 598.56 85 60 25 8 20 18249 11845 7084 2003.67 1904.45 730.45 100 80 20 10 30 20395 12201 7893 2690.67 2547.23 984.12
- 206 -DCIS 2004
Clock Cycle Length Minimization by Arrival Time Aware Scheduling
Rafael Ruiz-Sautua, María C. Molina, José M. Mendías Dpto. Arquitectura de Computadores y Automática
Universidad Complutense de Madrid - Spain, [email protected], [email protected], [email protected]
ONVENTIONAL scheduling algorithms usually adjust the clock cycle duration to the execution
time of the longest operations. This results in large slack times wasted in those cycles with faster
operations. To reduce the wasted times multi-cycle and chaining techniques have been employed. The
design technique presented in this paper goes one step further. For a fixed latency, the performance
improvement is achieved by selecting the minimum clock cycle duration, which is independent of the
operation execution times. In order to adjust the arrival times of the results calculated to the cycle
duration, some specification operations are fragmented and every fragment scheduled in a different
cycle (non-necessarily consecutive cycles). Also the result bits of one operation are available in the
cycle they are calculated, to be used by any successor. So the execution of one operation may start
even if its predecessors have not finished yet. Additionally, the regularity of operation chains
scheduled in every cycle allows the design of new operators, which not only simplifies the allocation
and binding phases, but also produces more structured and smaller datapaths. Experimental results
show encouraging improvements in performance, as the cycle duration of the synthesized circuits
shows reductions of up to 85% (70% on average) with slight increments in datapath area. Figure 1
compares the cycle length and area of some circuits synthesized by our algorithm to those proposed by
the force-directed scheduling one with chaining and multi-cycle features built-in.
C
010000200003000040000500006000070000
Are
a
10 15 20 25 30 35
Latency
Ours + Patterns Ours Force-directed
0
5
10
15
20
25
Cyc
le le
ngth
10 15 20 25 30 35
Latency
Our approach Force-directeda)
Figure. 1. a) Cycle lengths of the schedules proposed by our algorithm and the force-directed one for some syntheticcircuits, b) area of the implementations obtained from our schedule and the force-directed one.
b)
- 207 -DCIS 2004
Session 9b
Communications Systems Friday nov. 26 14h00 15h15, Auditorium
Chairs
Armando Roy (U. de Zaragora) Roberto Sarmiento (U. Las Palmas de Gran Canaria)
An Efficient Priority Queuing System for High Speed Network Processors with QoS Support
F. Tobajas, V. De Armas, N. Cruz, R. Esper-Chaín, R. Arteaga and R. SarmientoInstituto Universitario de Microelectrónica Aplicada, IUMA
Dpto. de Ingeniería Electrónica y Automática, University of Las Palmas de Gran Canaria 35017 Las Palmas de Gran Canaria, Spain
HE growth and expansion of Internet allow users the communication, education, work and
entertainment, and it requires an increase in bandwidth demand that will stand in the following
years. The integration of new applications with different requirements imposes on switches and IP
routers, not only to absorb the bandwidth increase, but also to provide differentiated services. To
provide QoS guaranties, priority queues are needed to store flows from different aggregates while a
packet scheduler determines the next packet to be transmitted according to a timestamp value. The
feasibility of priority queues with many thousands of entries has critical implications for high speed
networks and the future internet. In this paper, a high performance sorting architecture designed to
efficiently support any Packet Fair Queuing (PFQ) scheduling algorithm in high speed switches or
routers is presented. Its primary function is to quickly sort the timestamp of the packets being stored in
SRAM or other queue memory according to a predetermined algorithm, controlling how packets are
pulled from queues, locating the lowest timestamp, and placing the packet into the output stream.
While a network processor, or even a RISC control processor could do this function, this kind of sort
is pretty compute-intensive. The fast and scalable priotity queuing system proposed in this paper is
based on a pipelined priority heap and was designed in the form of a core, integratable into an ASIC,
being capable of holding up to 65535 entries that may be distributed evenly across 1, 2, 4, or 8
independent priority queues managed by means of a novel low cost method, and providing the address
with the minimum timestamp at one clock cycle, regardless of the level occupation or the number of
independent priorities supported. The basic operations of the proposed priority queuing system allow
new entries to be inserted or the entry with the minimum timestamp value to be extracted from the
priority queue. Other operations include the ability to read the entry with the minimum timestamp
value without altering the priority queue, or to perform both an extraction and an insertion
simultaneously.
T
The implemented priority queuing system performs scheduling and QoS decisions with a throughput
rate close to 70 million per second that would take hundreds of reads, writes and sorting steps on an
unassisted network processor, significantly boosting performance of priority queue-based scheduling
algorithms.
- 209 -DCIS 2004
OPTIMIZATIONS IN DVB-RCS TURBO DECODER BASED ON TRELLIS STRUCTURE
Jesús M. Pérez Llano, Víctor Fernández Solórzano TEISA Department, University of Cantabria, Cantabria, Spain,
chuchi,[email protected]
S INCE Turbo codes were born in 1993, they have been widely spread due to their spectacular
(close to Shannon theoretical limit) performance. On the other hand, this kind of channel coding
scheme presents big area consumption. In order to solve this area consumption there have been
proposed approximations to original decoding algorithm (MAP) as log-MAP or max-log-MAP.
Several standards, as CDMA, UMTS or DVB-RCS, have adopted this kind of channel coding system,
making these algorithms even more interesting for researchers.
DVB-RCS may well become a global satellite standard that allows all equipment manufacturers to
focus on the same technical solution, thus providing a healthy and open competitive environment,
providing enormous benefits to industry and users alike. In this paper several improvements are
presented for the DVB-RCS max-log MAP VLSI design, but they can be applied to any other max-log
MAP based system. Due to the fact that the proposed architectural optimizations do not alter the
original max-log MAP algorithm, they keep the same BER performance.
First two proposed modifications refer to decoder processors, saving an important percentage of the
area spent in classical designs. In the second part of the paper a new way to face the LLR calculation
is presented, saving more than 30% of the area required in classical implementations by performing
two optimizations that can be applied together or separately. Both modifications save an important
amount of area, and only the second one has a slight penalty in the critical path.
- 210 -DCIS 2004
Implementation of a 2.5Gbps ATM over SDH
Transceiver with Add/Drop on a Virtex-II
R. Arteaga, R. Esper-Chaín, O. Tubío, F. Tobajas, V. de Armas and R. Sarmiento
Instituto Universitario de Microelectrónica Aplicada
Universidad de Las Palmas de Gran Canaria, Campus de Tafira 35017, Las Palmas, Spain
rarteaga,esper,otubio,tobajas,armas,[email protected]
Nowadays worldwide optical communications & networking markets are evolving rapidly offering
new services over existing technologies. To face these challenges, it is necessary to make use of some
standards which provide flexibility to avoid the problems due to these fast changes. SDH, ATM and
Utopia interfaces are well known standards which were developed by ITU and ATM Forum. These
standards are quickly spreading the telecommunication network over the world.
In addition, a rapid prototyping solution is required to develop telecommunication devices in a short
period of time, reducing the time to market. Flexibility, rapid prototyping and low cost are some of the
main characteristics in FPGA. Based on these FPGAs, it is possible to implement complete solutions
for most of the high speed communication & networking devices, easing the performance evaluation
of implemented systems using rapid prototyping methodology.
In this paper, the implementation of a STM16c transceiver with add-drop capability is presented.
This system combines three different protocols such as SDH, ATM and Utopia (Universal Test &
Operations PHY Interface for ATM) with five clock domains. In the evaluation process, a transceiver
was implemented in a Virtex-II 6000 device along with traffic generators & analyzers and some
debugging tools, such as logic analyzer and VGA emulation.
As a conclusion, the design of this STM-16c add-drop ATM transceiver was demonstrated to be
fully functional. Also, the design demonstrates to have a moderate size, so it allows the designers to
integrate the transceiver with another components of line cards and improve the integration levels of
the network systems. No commercial solution for the complete system was found to the best of
authors’ knowledge.
- 211 -DCIS 2004
ITU-Compliant Macrocells for Dual Tone Multiple Frequency Transmission and Reception
Arturo Purroy Isidro Urriza
Department of Electronic and Communications Engineering,University of Zaragoza, c/María de Luna 3, 50018 Zaragoza, Spain.
apurroy, [email protected]
HE telephone network provide fast access to almost anywhere in any industrialized country. An
easy interface must be offered to customers of domotic environments in order to manage the home
automation system, and the phone push-button keyboard may be an efficient solution.
Inside the house, a controller device is needed. This device should be able to decode the DTMF
symbols received from the user and link with the home systems to perform the asked service. To
develop cheap solutions, single chip designs are preferred. We present in this paper the design of a
digital transmitter and receiver of DTMF symbols modeled in a hardware description language
(VHDL), which can be easily added to other designs in a single field programmable gate array
(FPGA). The efficiency and physical size of the final design have been the main goal of the research.
Several chips are available which employ analogic circuitry to generate and decode DTMF
signals. The advantages of a digital system include better accuracy, precision, stability, versatility, and
reprogrammability as well as lower chip count, and thereby reduced board-space requirements.
There have been several investigations dealing with efficient DTMF detectors, trying to simplify
the filter bank. DTMF detectors typically consist of a signal analysis front end followed by a decision
logic back end. The usual approach is to use an adaptable filtering architecture which is able to
implement multiple filters with just the hardware of one. The Goertzel algorithm or the non-uniform
discrete Fourier transform (NDFT) are the basis of some DTMF detectors. The number of products
required, and the long windows of samples make their implementation inefficient.
A very efficient solution was proposed by A.A. Deosthali, S.R. McCaslin and B.L. Evans. Using
adaptive notch filters and sophisticated decision logic, their detector meets the ITU standard when
implemented in a 8-bit microcontroller. Based in this algorithm, our design has rearranged all the
filters and signal processing units to take advantage of a specific hardware design. As the original
implementation used a 8-bit microcontroller, when more precision was needed (16-bit data), the
original algorithm had difficulties to operate. This and other implementation problems have been
solved in our design, which complies with the ITU specifications.
Our design is the first solution published that provides VHDL code encapsulated in digital
macrocells. This goal has added constraints to our design, as the size of a macrocell should be
minimized.
T
- 212 -DCIS 2004
Evaluation of a PHM Scheduler Implementation
F. Javier González-Castaño1, Enrique Soto-Campos2, Rafael Asorey-Cacheda1,Cristina López-Bravo3, José Fariña-Rodríguez2, Juan J. Rodríguez-Andina2
1Dpto. Ingeniería Telemática, 2Dpto. Tecnología Electrónica, Universidad de Vigo, Spain3Dpto. Tecnologías Información y Comunicaciones, Univ. Politécnica de Cartagena, Spain
virtual output-queued (VOQ) switch is a particular case of input-queued switch such that
VOQ(i, j) is the partition of input queue i that stores the packets directed towards output j. In a
previous paper1, a new maximal size matching algorithm for VOQ switches, namely Parallel
Hierarchical Matching (PHM), has been proposed. PHM is a distributed algorithm with the same gate
complexity (O(N2), where N is the switch size in I/O ports) as parallel iterative maximal size matching
algorithms such as iSLIP2 and RDSRR. All these algorithms maximize instantaneous throughput in
high-performance VOQ packet switches such as the Cisco 12000, the Lucent Cajun and the Nortel
Versalar TSR45000. Recently, it has been shown that PHM is competitive with them under hot-spot,
bursty and unbalanced traffic. The results in1 suggest that PHM has both the advantages of previous
sequential hierarchical matching algorithms (low hardware complexity) and parallel iterative maximal
matching algorithms (low number of iterations).
In this paper, a PHM implementation is presented and compared to an efficient parallel iterative
maximal matching algorithm implementation.
A full PHM VOQ controller and the parallel iterative maximal matching arbiter in2 have been
implemented for switch sizes 2×2, 4×4, 8×8 and 16x16, using the AMS Standard Cell Library for the
0.35 m 3.3V CMOS process. Table I shows worst-case decision response times per iteration and the
corresponding maximum work frequencies.
The results obtained clearly show that PHM is competitive with state-of-the-art VOQ schedulers in
terms of delay and speed.
1 Asorey-Cacheda, R., González-Castaño, F.J., López-Bravo, C., Pousada-Carballo, J.M. and Rodríguez-Hernández, P.S. "On the Behavior of PHM Distributed Schedulers for Input Buffered Packet Switches'' IEEE Transactions on Communications, vol. 51, no. 7, July 2003. 2 N. McKeown, “iSLIP: A Scheduling Algorithm for Input-Queued Switches,” IEEE/ACM Trans. Networking, vol.7, no. 2, 1999.
A
TABLE IPARALLEL ITERATIVE ALGORITHM VS. PHM:
ITERATION RESPONSE TIMES
Size IMM PHM 2x2 6.85 ns (146 MHz) 1.54 ns (649 MHz) 4x4 10.04 ns (100 MHz) 1.98 ns (505 MHz) 8x8 14.98 ns (67 MHz) 2.60 ns (385 MHz)
16x16 30.67 ns (33 MHz) 2.32 ns (431 MHz)
- 213 -DCIS 2004
Session 9c
System Level Design Friday nov. 26 14h00 15h15, St Emilion Room
Chairs
Yannick Hervé (U. de Strasbourg) Víctor Fernández (U. de Cantabria)
XML Specification and Tools for Automatic SoC Generation
Màrius Montón1, Oriol Font1, Jaime Joven1, Pere Garcia2, Lluís Terés3, Jordi Carrabina 1
1Universitat Autonoma de Barcelona. 2EPSON Electronics Europe. 3Centro Nacional de Microelectronica.
marius.monton, [email protected]. joseporiol.font, [email protected]. [email protected]. [email protected]
HIS paper presents a methodology for building SoC systems starting from XML specifications
oriented to system on chip platforms with virtual component (IPs) integration.
This methodology includes the implementation of a set of tools, written in Java, to generate the
whole set of Hardware (Verilog) and Software (C) files required to synthesize and simulate an entire
AMBA-based SoC.
This tool can assist architectural exploration during model refinement and HW/SW partitioning, as a
critical step to speed-up the design process for new complex SoC systems.
There are a lot of approaches for building SoCs according to application specific purposes. Our tool
can help designer to build different architectures in a short period of time. This will help architectural
exploration and verification at simulation levels and using prototyping platforms.
Our tool, called UltraWizard, is written in Java and starting with a XML file describing bus
architecture and IP interconnection automatically generates HDL files (Verilog) and SW files (.C) to
synthesize and simulate an entire AMBA SoC.
To show tool benefits, an example has been designed that maps into different architectural solutions
that will be generated by UltraWizard and they have been synthesized and tested to our prototyping
platform.
The implemented test system outputs a sinusoidal waveform through a DAC (with a customizable
FIFO depth). Samples are calculated by an ARM7 processor, and sent in different ways to DAC.
1) A simple AHB+APB architecture. CPU is polling DAC continuously and sends new sample to
DAC when it is ready.
2) AHB+APB+IC: An interrupt controller is added to notify CPU when DAC is ready (i.e. FIFO
empty).
3) AHB+APB+DMA: A DMA is used to decrease CPU load. DMA interrupts processor every
time a middle-buffer is empty and DAC interrupts every time its FIFO is empty.
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- 215 -DCIS 2004
Generic Programming with Abstract Parametrized Components
Fernando Rincón, Jesús Barba, Juan Carlos López, Juan Pablo Rozas Computer Architecture, University of Castilla-La Mancha, Ciudad Real, Spain,
fernando.rincon|jesus.barba|juancarlos.lopez|[email protected]
I t's widely agreed that the best way to increase design productivity is to reuse solutions that worked
in the past. The concept reuse, however, has a different meaning when applied to a software system
or a hardware one. While in the former it is normally understood as the reuse of the model, in the latter
it normally means the reuse of components. Working at the level of components makes it more
difficult to find opportunities where a previous solution can be applied again, since often they are to
specific. This scenery is quite similar to what happened with software development not so long ago.
The improvement in quality and design productivity in the software domain has traditionally been
based in raising the level of abstraction of system models, where problems are considered in their own
domain, without the disturbance of implementation details. One clear example is generic
programming, where abstraction is achieved through the use of parameterizable code which can be
applied to any data structure. It is straightforward to suppose that we could expect the same kind of
benefits if we were able to apply all this mature software technologies to the problem of designing
complex hardware systems.
The aim of this paper is to discuss how one of these software technologies (generic programming)
can be applied into hardware by means of a partial port of the Standard Template Library in C++. The
STL is based in the cooperation of three kinds of elements (containers, iterators and algorithms) whose
main purpose is to decouple algorithms from data structures, and where all elements are able to work
with arbitrary data structures. A discussion about the minimal set of elements and characteristics of
this elements is presented, where some clues about the implementation consequences are depicted.
Finally a very simple example has been used to illustrate the benefits of using generic programming to
model hardware systems.
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System level design using SystemC: a case study of block turbo decoder
Erwan Piriou, Christophe Jégo, Patrick Adde and Michel Jézéquel Electronic department GET/ENST-Bretagne, CNRS TAMCIC, Brest, France,
T he objective of this article is to give the results of a project where a conventional design flow is
replaced by a system level design flow using SystemC language. Traditional methods for
designing hardware circuits for digital communication systems use an RTL specification. However,
they suffer form heavy limitations that prevent them from efficiently addressing the algorithmic
complexity and the high flexibility required by the various application profiles. SystemC 2.0 is a
standard design and verification language that spans from concept to implementation. One of the
primary goals of SystemC is to enable modeling of systems that might be implemented in software,
hardware or some combining of the two.
A System On Chip may be defined as a complex circuit that integrates the major functional
elements of a complete system. Recently, FPGA leading suppliers have developed a new type of
programmable circuit: the System On a Programmable Chip. This technology corresponds to the
integration of the software and hardware resources into the same FPGA. In that case, one or more
processor cores, available in the programmable circuit, carry out the execution of the software
resources. This approach provides the flexibility to integrate memory, processors, peripherals, and
other intellectual properties (IP) into the same chip.
The original flow was successfully applied to the design of a block turbo decoder (128,120,4)2.
Currently the iterative process, known as turbo coding, is the most efficient channel coding technique
for digital communications. Block turbo codes are an alternative solution to convolutional turbo codes.
They are especially attractive for high-speed applications and offer a very good coding gain at high
code rates. The functional blocks and the control unit of the elementary BCH decoder (128, 120,4)
were respectively implemented into hardware modules and into a software module. For this reason,
our architectural solution is composed of a Nios embedded processor. The block turbo decoder
(128,120,4)2 was integrated into an Altera's Nios Development Kit, Stratix Professional Edition which
is based on the Stratix EP1S40 device.
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Object-Oriented Hardware/Software Co-Simulation Using SystemC
HE complexity of digital electronic systems has increased considerably lasts years. Software
development costs will overtake hardware development in future technology processes. In this
paper we address this open issue. The main objective is to propose a new methodology that enables us
to re-use SW code in simulation and prototyping to reduce time-to-market and increase the reliability
of the design (Fig. 1). This new methodology will use SystemC and will be object-oriented. In this
paper we will apply our methodology to the case study of a high-speed LVDS interface DMA
transmitter module developed at Hewlett-Packard BPO Spain.
Figure 1. A) Traditional functional verification methodology. B) Co-simulation HW/SW
Ana M. Cardells Tormo¹, Juan José Noguera Serra¹, Lluís Terés Terés²
ana,[email protected], [email protected], [email protected]
¹R&D Department Hewlett-Packard Barcelona, Spain.
²Microelectronic Department, University Autónoma of Barcelona and Electronic Circuits & Systems Design Department, CNM-CSIC.
T
SystemC testbench
C++ stimulus(BasicTest)
RTL
FM
FM
FPGA / ASICPrototyping
Simulator
C++ newstimulus
CPU (compile)
Verilog stimulusC++
stimulus
RTL
FM
FM
FPGA / ASICPrototyping
Electronic engineer SW engineer
conversion
Testbench
C++ newstimulus
CPU (compile)
- 218 -DCIS 2004
Comparing Design Flows for Structural System Level Specifications facing FPGA Platforms
D. Castells, M. Monton, R. Pla, D. Novo, A. Portero, O. Navas, J. Farré, L. Ribas, J.Carrabina Cephis, Universitat Autònoma de Barcelona, Bellaterra, Spain, [email protected]
S YSTEM level design methodologies introduce new design flows that are complementary to the
ones provided by existing toolsets based on HDLs. Therefore, a miscellaneous of tools and
methodologies are available for the design of complex microelectronic systems driven by different
actors playing on the microelectronic arena.
This paper compares three different system level design methodologies derived from MATLAB,
SystemC and JHDL; together with the classical use of HDL languages (in this case VHDL).
A high-speed sorter, defined at structural level, is used as a common specification to test different
methods. Independent development teams with experience in each tool-set are faced to the same
specification. Some development process indicators are selected in order to be able to compare teams’
and tools’ productivity. Also the obtained performance and area usage characteristics for synthesized
circuits are measured and compared. Results are presented for the different development phases (as
shown in Figure 1): Simple design development, unit simulation test, final design development,
complete simulation test and synthesis and physical verification
Finally obtained results show that, for this particular experiment, MATLAB and JHDL have been
more productive than other methods, especially than SystemC. All methodologies produce circuits
with similar area usage but having different timing characteristics, being VHDL the more efficient.
- 219 -DCIS 2004
Session 9d
Noise in Electronics Friday nov. 26 14h00 15h15, Lacanau Room
Chairs
André Touboul (U. Bordeaux 1) Antonio Rubio (U. Politècnica de Catalunya)
Minimum Noise Figure Comparison ofY-Parameter Based Bipolar Noise Models
Juan Carlos Milena, Manuel Sánchez-Lerma, Juan M. López-González and Antonio J. García-Loureiro*
Universitat Politècnica de Catalunya, *Universidad de Santiago de Compostela.
HIS work is dedicated to compare the state of art of Y-parameter based bipolar noise models
using a SiGe low noise heterojunction bipolar transistors: Spice model, thermodynamic approach
model (TDA), correlated shot-noise model (CSN) and TDA/CSN interpolation model. Minimum noise
figure, Fmin , comparison is done using measured and simulated Y-parameters and both of them are
compared with Fmin measurements in low GHz frequency range (from dc up to 6GHz). Simulated Y-
parameters are extracted from Spice small signal equivalent circuit, using Gummel-Poon expressions.
Measured Y-parameters are obtained from Infineon Technologies. Minimum noise figure’s plots
against frequency and against collector current are shown, see figs. 1 and 2. Using simulated Y-
parameters all models fit very good. However, using measured Y-parameters good agreement is only
obtained at low currents. The discrepancies between measured and modeled Fmin are discussed.
Furthermore, two possible improvements are suggested to solve inaccuracy at large currents.
Figure 1. Comparison of modeled and measured Fmin versus frequency (Ic=20mA, n=0.75 ps) with measured Y-parameters.
Fig. 2. Comparison of modeled and measured Fmin versus collector current (freq=6 GHz, n=0.75 ps) with simulated Y-parameters.
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- 221 -DCIS 2004
Spectral Characterization of the Digital Noise
Miguel Ángel Méndez, José Luis González, Diego Mateo and Antonio Rubio (mamendez, jlgonzalez, [email protected], [email protected])
Electronic Engineering Department, Universitat Politècnica de Catalunya C/ Jordi Girona, 1-3, Campus Nord - C4, 08034 Barcelona, SPAIN
HE noise generated by digital circuits is one of the most limiting factors to implement
mixed-signal integrated circuits. The integration of digital and analog circuits in the same
silicon die is conditioned and limited by the noise levels generated in the digital section in
conjunction with the increasingly demanding performance requirements of the analog and
Radio Frequency (RF) sections.
This work describes the most important spectral characteristics of the digital noise, analyzing
its relation with the current demanded by digital gates and the resonance circuit network
formed by digital circuit parasitics in addition to the package and substrate. First, we analyze
the spectral content and characteristics of the digital switching current waveform, deriving its
influence on the digital noise power spectrum. This current source is modeled in frequency
and time domain by a generalized expression. The second relevant factor analyzed is the
resonant circuit network, which determines a transfer function from that primary noise source
to the node where the magnitude and effects of the digital noise is evaluated. This transfer
function (actually a transimpedance) multiplied by the spectral content of the excitation
source determines the overall characteristics of the digital noise power spectrum. The circuit
transfer function acts as a filter that modifies the characteristic spectrum of the digital
switching current, modifying its frequency content mainly near to the package resonance
frequency. In this work analytical models that allow to predict the main characteristics of the
digital noise are presented. These models are applied to some examples, where a good
agreement is found between the analytical expressions and simulation results.
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- 222 -DCIS 2004
On the Relation between Digital Circuitry Characteristics and Power Supply Noise
Spectrum in Mixed-Signal CMOS IC
Miguel Ángel Méndez, José Luis González, Enrique Barajas, Diego Mateo and Antonio Rubio (mamendez, jlgonzalez, ebarajas, [email protected], [email protected]) Electronic Engineering Department, Universitat Politècnica de Catalunya
C/ Jordi Girona 1-3, Campus Nord - C4, 08034 Barcelona, SPAIN
IGITAL power supply noise is a key issue in the design of mixed-signal and Radio Frequency
(RF) integrated circuits (IC). Implementing low-cost and high performance communication
systems on a single silicon die (System on Chip, SoC) raises concerns about switching noise coupling
from the digital section into the sensitive analog parts of the circuits through the common substrate.
In the present work we investigate the relation between the most relevant parameters of the power
supply noise spectral density (PSD) and some digital circuit characteristics: clock frequency, power
supply voltage, synthesis alternatives considered (what means different circuit topologies), and also
technology used.
To do that we have carried out a statistical analysis over a benchmark circuit (ALU181 from
Motorola), analysis which has been extended by using mathematically described switching current
waveforms.
In order to validate the study different measurements over a test chip (see Figure 1 and 2) built on a
0.35µm CMOS, high resistivity p substrate technology, have been carried out.
0.2 0.4 0.6 0.8 1.0 1.2 1.4−80
−70
−60
−50
−40
Frequency (GHz)
dBV/
Hz
0.2 0.4 0.6 0.8 1.0 1.2 1.4−80
−70
−60
−50
−40
Frequency (GHz)
dBV/
Hz
Bandwidth = 100 KHzfclk
= 10 MHz
(a) Vdd
= 3.3 V
(b) Vdd
= 2.5 V
Figure 1. Digital noise generator block diagram Figure 2. Measured PSD for different power supplies.
From the study done we have concluded some relations between circuital characteristics and power
supply noise PSD, which can be used during the design of mixed signal communication ICs in order to
reduce the generation of digital noise and also its effects on sensitive analog parts (for example, taking
into account the noise PSD during the RF frequency planning).
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- 223 -DCIS 2004
