Lab 5. Audio Amplifier - Appendix

1. The VBE MultiplierA VBE multiplier, sometimes called an amplified diode, (Figure 2) is used to provide a

tunable voltage between the bases of the Darlingtons X1 and X2. The purpose of this voltage is

to bias the bases of the two Darlingtons, keeping them in a “slightly” ON state - a quiescent

current of about 20 mA is desirable. Tuning is obtained through the use of potentiometer XRV1.

The quiescent current minimizes the zero crossing distortion associated with the power stage

emitter followers. Since VBE is a function of temperature, Q3 should be mounted on the same

heat sink as X1 and X2 to minimize voltage drift.

Figure 2. Vbe Multiplier

The value of the bias should be about 4 x VBE ≈ 2.8V, enough to forward bias each of the

four base-emitter junctions in the pair of Darlingtons.

Operation

Assume a current I flows down out of node 2. Assume the gain of Q3 is high, so Ib3 can

be ignored. A current I’ = I - IE flows through R1 and R2.

1

V R 1 I ' R 1

V R 2 I ' R 2 R 2R 1

V B E 3

V 2 , 3 V R 1 V R 2 1 R 2R 1

V B E 3

Thus the voltage drop is a multiple of VBE3, and is not restricted to being an integral

multiple.

By placing a potentiometer at the base of Q3 we can adjust the bias to accommodate

tolerance variations between Darlingtons. A capacitor can be used between nodes 2 and 3 to

bypass ac signals so that the VBE multiplier provides a simple dc bias.

Example 1

Design a VBE multiplier to provide a 3.0 V bias. Assume I = 2.5 mA, ß =

100, and VBE = 0.7 V.

V 1 R 2R 1

V B E 3

R 2R 1

V

V B E 3 1

3 . 0 V0 . 7 V

1 3 . 2 9

Let the resistors have standard values R2 = 3.3 kΩ, R1 = 1.0 kΩ. Or we

could let R2 be a 2.7 kΩ fixed resistor and a 1 kΩ potentiometer in series.

This would allow bias adjustment from 2.59 V to 3.29 V.

Note that since the 2N2222 transistor does not have the ideal VBE of 0.7

V, the voltage drop across the VBE multiplier is not exactly 3.0 V.

Note that in Example 1 the calculation produced the ratio of R2 to R1. How do you

choose unique values? To get a better feel for the range of appropriate values, look at the ac

resistance of the VBE multiplier:

r R 2

1

R 2R 1

r e , w h e r e r e

0 . 0 2 5 VI c

This can be derived from the ac equivalent circuit of the VBE multiplier. To produce a

nearly pure dc voltage drop of a given value you would want R2 small, ß large and Ic large.

Example 2

Calculate the ac resistance of the VBE multiplier in the previous example.

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I E I I ' 2 . 5 m A 0 . 7 V1 . 0 k

1 . 8 m A

r e 0 . 0 2 5 V1 . 8 m A

1 3 . 9

r 3 . 3 k

1 0 0 1

3 . 31. 0

1 3 . 9 3 3 . 0 5 9 . 8 9 2 . 8

If this value of r is too high you could lower R2 (and R1), raise I, or add a

capacitor between nodes 2 and 3 with an impedance much less than r at the

frequencies of interest.

2. Bootstrapping[Reference: Transistor Circuit Techniques - Discrete and Integrated, 3rd ed., G.J. Ritchie,

Chapman and Hall, London, 1993]

Bootstrapping is a method of increasing the open loop [no feedback] gain of an amplifier.

In this amplifier, capacitor C4 is the bootstrap capacitor. The higher the open loop gain, the

more accurately the equation AV = – RB/RA will predict the closed loop [with feedback] gain of

the audio amplifier using this design style. Consider the circuit below, where Q1 is used in a

common emitter amplifier and Q2 is an emitter follower, or common collector (CC), amplifier.

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Figure 3. Bootstrapping Example

If the ac voltage at the collector of Q1 is v, then voltage Av is present at the emitter of Q2

and also at the common terminals of R3 and R4, assuming that the impedance of C4 is very

small at the frequencies of interest.

The gain A of the common emitter amplifier is

A R

r e 2 R, w h e r e R R E | | R 3 , a n d r e 2

0 . 0 2 5 VI c 2

The voltage across resistor R4 is v – Av = (1 – A) v.

So for ac signals, the resistance of R4 appears to be R4’ = R4 / (1 – A), which is much higher

than R4 since A is typically just slightly less than 1 for the emitter follower.

Since R4’ is the collector resistance for the common emitter stage, the gain of the CE

stage will become

A V R 4 ' | | r i n(fo l l o w e r)

r e1, w h e r e r i n( fo l l o w e r) r 1 R R

Without the bootstrapping, the gain would be

A V R 3 R 4 | | r i n ( f o l l o w e r)

r e 1

a value much less than that seen if C4 is used.

Example 3. Bootstrapping

Design a CE/CC amplifier pair. Compare the voltage gains with and

without bootstrapping.

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Assumptions and Design Parameters

Vcc = 12 V

Ic for each transistor is 2.5 mA

ß for each transistor is 100

RL = 1 kΩ

low frequency cutoff ≤ 20 Hz

Use 2N2222 for simulations

See appendix for detailed component calculations and PSpice files

Figure 4. Common emitter, common collector amplifiers without

bootstrapping.

Without bootstrapping (Figure 4), the voltage gain is the product of the

gains of the CE stage and the CC stage:

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A v R C | | R E 2

r e 1

R E 2R E 2 r e 2

, w h e r e r e i V t

I c i

2 5 m VI c i

A v 1 . 6 k | | 2 9 0 k

1 0

2 . 9 k 2 . 9 k 1 0

A v 1 5 9 2 9 0 02 9 1 0

1 5 8 . 6

A v P S p i c e 1 2 9

ß x RE2 represents the input resistance of the common collector amplifier,

and is the load for the common emitter amp.

With bootstrapping (Figure 5), the collector resistor of the common

emitter is split into two pieces and the bootstrap capacitor C4 is added.

The impedance of C4 should be small in the audio frequency band.

Figure 5. Common emitter, common collector amplifiers with

bootstrapping. RC was split in half and capacitor C4 added. The amplitude

of the sine wave input was reduced in anticipation of higher gain.

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The voltage gain of the amplifier pair is Av, and the voltage gain of the

common collector stage is A.

A v R C 2 ' | | r in ( fo llo w e r )

r e1 A

w h e r e r e i V t

I c i

2 5 m VI c i

, A R

R r e 2,

R R E 2 | | R C 1, a n d r in ( fo llo w e r ) R

Plug in the component values and solve for the bootstrap gain:

R R E 2 | | R C 1 2 . 9 k | | 8 0 0 6 2 7

r e 2 2 5 m V

2 . 5 m A 1 0

A R

R r e 2

6 2 7 6 3 7

0 . 9 8 4

R C 2 ' R C 21 A

8 0 0

1 0 . 9 8 4 5 0 k

A v 5 0 k | | 6 2 . 7 k

1 0 0 . 9 8 4

A v 2 7 8 2 0 . 9 8 4 2 7 3 7

A v P S p i c e 1 3 0 0

While we can show a large increase in gain (a factor of nearly 20) from

the no bootstrap case, the calculation and simulation do not match. The

reason for this is not yet known.

Feedback

Your textbook can give you a much more detailed description of feedback in amplifier

design than will be presented here. You are encouraged to read the appropriate sections.

Why use feedback in an amplifier design?

• The gain is stabilized against variations in the parameters of the active devices, so no

tuning is needed regardless of which TIP120 Darlington transistor I choose from the bin,

for example. The gain of the amplifier is also protected from variations in temperature,

supplies, or degradation with age.

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• The input and output impedance may be selectively increased or decreased.

• Non-linear signal distortion is reduced.

• The midband frequency range is increased.

These last two properties make feedback very advantageous for audio circuits.

Possible disadvantages of using negative feedback

• The gain is reduced greatly from the open loop case. If high gain is needed, additional

amplification stages will be needed, with the penalties of size and cost.

• The circuit is more likely to go into oscillation.

There are several types of feedback and several ways they can be implemented. Positive

feedback results when the signal being feed back is in phase with the input signal. Negative

feedback occurs when the signal being feed back is 180 degrees out of phase with the input

signal. In a common emitter or common collector amplifier, feedback is introduced into the base

of the first stage transistor. In a common base design, feedback is introduced into the emitter of

the first stage.

Shunt Feedback vs. Series Feedback

Using an opamp with open loop gain Ao to represent the amplifier, we can illustrate

several feedback options.

Figure 6. Shunt negative feedback (left), and series negative feedback (right). The opamps have

open loop gain Ao.

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For shunt negative feedback,

A v R B

R A

A oR A

R A R B

1 A oR A

R A R B

For series negative feedback,

A v R A R B

R A

A oR A

R A R B

1 A oR A

R A R B

Considering the shunt case, we can simplify the gain equation and remove (or nearly so)

the sensitivity of Av to variations in Ao by making Ao very large. In this case, Av = -RB / RA,

an opamp expression you probably remember from your introductory course and lab on circuits.

The audio amplifier in Figure 1 makes use of shunt feedback. A portion of the output

signal is fed back to the base of Q4 in the CE amplifier. Note that another change is to remove

the top of resistor R5 from the dc supply (compare to figure 4 or 5). The bias network now

depends on the dc level of the Class AB amplifier output (node 1 in figure 1), a value slightly

higher than Vcc/2. Resistor R7 determines the input resistance of the amplifier.

Design

The best plan for getting the most understanding of the sudio amplifier would be to

design, simulate and build the circuit of figure 4, modify it with bootstrapping (figure 5) and

then add feedback. For the sake of time, I am going to go directly to the design of the audio amp

of figure 1.

Design Steps

1) Determine the gain needed. This will help decide if you need a preamp. Typical project

specifications are to produce 4 W into an 8Ω speaker from a 50 mV peak to peak input.

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Av = - Vout / Vin

Find the peak voltage needed at the load and compare it to the peak input voltage. A JFET

preamp stage can produce a gain of 2 to 4.

2) Determine the power supply voltage needed. If a preamp is needed it would also use this

supply voltage. To get good symmetrical swing, we set the dc voltage at node 1 to be Vcc/2.

Look at the voltage drops from the supply to the output.

V c c V C E X 1 s a t V R E 1 V c c

2 V lo a d (1 )

where all the dc voltage drops across the output capacitor. Then

V c c 2 V C E X 1 s a t V R E 1 I lo a d p e a k R lo a d (2 )

Assume:

X1 = TIP120 NPN Darlington

X2 = TIP125 PNP Darlington

Keep the voltage drops across RE1 and RE2 small. These resistors are present for thermal

stability only, and you should not waste power in them. Let 0.5 ≤ VRE1, VRE2 ≤ 1 V.

What is the peak load current? What is the VCE(sat) for the NPN Darlington? The output

power equals the product of the rms current and the load resistance:

P o R l o a d I l o a d r m s2

I l o a d p e a k 2 P o

R l o a d( 3 )

The TIP120 Darlington is rated for a continuous IC current of 5.0 A.

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The saturation value of VCE depends on the collector current. For the TIP120 Darlington (see a

data sheet):

Table 1. (VCE)sat vs Collector Current data for TIP120 NPN Darlington Transistor

IC (VCE)sat

(A) (V)

0.5 0.75

1.0 0.85

3.0 1.15

5.0 1.40

Plug your values of (Iload)peak, VRE1, Rload, and (VCEX1)sat into equation 2, and solve for

Vcc. Round the supply to a common value (5V, 12V, 15V, etc.). Since each Darlington

transistor must be able to withstand the full Vcc voltage, check that the VCEO specification is at

least equal to Vcc plus 10 to 20 % safety margin. In this case, the TIP120 and TIP125 each have

VCEO = 60 V.

RE1 and RE2:

Choose a value between 0.5 and 1.0 V for VRE1.

R E 1 R E 2 V R E 1

I l o a d p e a k

The power dissipation in the emitter resistors will be (Iload)peak2 x RE1, and may be a

watt or more.

R1 and R2:

The VBE Multiplier design must take into account both the VBE drops across the

Darlingtons and the drops across RE1 and RE2. So, V(2,3) = 2.8 V + 2VBE1. Use the design

procedure discussed earlier to find R1 and R2.

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R3 and R4:

The base current required for the Darlingtons depends on their minimum hFE or b.

h F E m in 1 0 0 0 f o r T I P 1 2 0 , T I P 1 2 5

I B X 1 p e a k IB X 2 p e a k I l o a d p e a k

1 0 0 0

The current through Q3 and Q4 should exceed this value by a margin of 2 to 10.

The voltage at node 2 should be V(2) = V(1) + VBE1 + 1.4 V. Then

R3 + R4 = (Vcc - V(2)) / Ic3. Split this value about equally between R3 and R4.

RE4:

In a normal CE amplifier design, we would have several volts dropped across RE4. In

this case, since the collector of Q4 drives a PNP Darlington, we have to keep the drop across

RE4 small, otherwise the voltage swing of the negative phase of the output will be limited

(clipped).

Let V(9) be about 0.5 V. Then RE4 = (0.5 V) / Ic3.

R5 and R6:

The value of R5 effects several circuit properties: base bias of Q4, gain, and input

resistance. The voltage at node Vin and the required base bias of Q4 are: V ( V i n ) I E 4 R E 4 0 . 7 V

I b 4 ( m a x ) I e 4

4

I e 4

1 0 0

The current in the divider consisting of R5 and R6 must exceed Ib4(max) by a factor of 2 to 10.

For a reasonable input resistance, we want R7 on the order of 1 kΩ. For a gain in the 100’s, R5

will have to be on the order of 100 kΩ. If R5 is too large you will not have sufficient base drive

current. If R5 is too small, the input resistance (R7) will fall below acceptable values.

Choose several possible values of R5, and try the following calculations:

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I R 5 V ( 1 ) V ( V i n )

R 5I R 6 ( m i n ) I R 5 I b 4 ( m a x )

R 6 V ( V i n )

I R 6 ( m i n )

R 7 R 5A v

IR5 must be greater than Ib4(max) by a factor of at least 2. If not, adjust R5. If R7 is much

below 1 kΩ, you might want to consider adding the JFET preamp to increase the input resistance

and decrease the gain requirement of this audio amplifier to about 100. An alternative would be

to reduce the collector current through Q4. This reduces the base bias requirement, and allows

larger values of R5 and R7.

References

Gibilisco, S. and N. Sclater, ed., Encyclopedia of Electronics, 2nd ed., TAB Professional

Reference Books, Blue Ridge Summit, PA, 1990.

Ritchie G.J., Transistor Circuit Techniques - Discrete and Integrated, 3rd ed., Chapman and

Hall, London, 1993.

Schubert, T. and E. Kim, Active and Non-Linear Electronics, John Wiley, New York, 1996.

Sedra, A., and K. Smith, Microelectronic Circuits, 3rd ed., Saunders, Philadelphia, 1991.

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Appendix

Amplifier Calculations (Figures 4 and 5)

Ic = 2.5 mA, ß = 100, RL = 1kΩ, Vcc = 12 V.

Let the voltage at the base of Q1, VB, be about Vcc/3 or 4 V for good output swing

Then at the emitter of Q1, VE = 4 - VBE = 3.3 V.

Let IE = IC to simplify calculations.

R E 1 V E

I E 3 . 3 V

2 . 5 m A 1. 3 2 k

Let the current through the divider consisting of R1 and R2 be about ten times the Q1 base

current, or about 0.1IE when ß = 100.

R 1 R 2 1 2 V

0 . 2 5 m A 4 8 k

R 2R 1 R 2

V c c 4 V

R 2 4 V

V c cR 1 R 2

4 V1 2 V

4 8 k 1 6 k

R 1 3 2 k

Let the drop across Q1’s collector resistor be about Vcc/3.

R C V c c V c

I c

4 V2 . 5 m A

1. 6 k

The dc bias at the base of Q2 should be 8 V. The emitter voltage should be 7.3 V.

R E 2 7 . 3 V

2 . 5 m A 2 . 9 k

PSpice Input Files

CE/CC Amplifier - no bootstrapping

* nominal gain = -160

* Spice extraction from McLogic

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C1 1 b1 91uF

C2 e2 Vout 33uF

RC Vcc c1 1.6k

RE1 e1 0 1.32k

RE2 e2 0 2.9k

Vin Vin 0 sin(0V 10mV 1kHz)

Vcc Vcc 0 12V

Q1 c1 b1 e1 Q2N2222

CE e1 0 1000uF

R1 Vcc b1 32k

R2 b1 0 16k

RL Vout 0 1k

Q2 Vcc c1 e2 Q2N2222

Rs Vin 1 100

* Pull the transistor models from the eval library

.lib eval.lib

.probe

* Run simulation for 2 cycles, 200 points minimum

.tran 0.02ms 2ms 0 10us

* Show the output voltage in the dialog box as sim runs

.watch tran V([Vout])

.end

******************************************************************************

*********

Bootstrapped CC/CE amplifier

* Nominal gain = -2700

* Spice extraction from McLogic

C1 1 b1 91uF

C2 e2 Vout 33uF

RC1 Vcc 2 800

15

RC2 2 c1 800

RE1 e1 0 1.32k

Vin Vin 0 sin(0V 1mV 1kHz)

Vcc Vcc 0 12V

Q1 c1 b1 e1 Q2N2222

CE e1 0 1000uF

R1 Vcc b1 32k

R2 b1 0 16k

RL Vout 0 1k

Q2 Vcc c1 e2 Q2N2222

RE2 e2 0 2.9k

Rs Vin 1 100

C4 2 e2 4000uF

* Pull the transistor models from the eval library

.lib eval.lib

.probe

* Run simulation for 2 cycles, 200 points minimum

.tran 0.02ms 2ms 0 10us

* Show the output voltage in the dialog box as sim runs

.watch tran V([Vout])

.end

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