ISSN NO: 0745-6999 JOURNAL OF RESOURCE MANAGEMENT
AND TECHNOLOGY
Vol12, Issue3, 2021
Page No:408
A New Control Design of Soft-Switching (S6) Boost-Flyback PFC Converter 1Doddi Satheesh, 2Er, P.Pedda Reddy,3Dr.K. Chithambaraiah Setty
1M.Tech Student, 2Asst.Professor, 3Associate Professor
Dept of EEE
St John's College Of Eng And Tecnology
Abstract—This thesis presents a S6 PFC converter to
enhance the current shaping performance and reduce
the total harmonic distortion (THD). This
improvement is achieved by the aid of an auxiliary
winding which is used to lower the input current
harmonics and also achieve soft-switching condition.
As a result, the switching losses are reduced and
harmonic content of the input current is improved
noticeably in comparison to the conventional S6 PFC
converter. Also the total number of semiconductor
elements is reduced in the proposed topology which
results in lower cost and higher efficiency. The
operating modes of the proposed converter are
discussed in details and the design procedure is
presented. A 200 kHz prototype of the proposed
converter is implemented and the obtained results are
provided to verify the converter theoretical analysis
and operation.
Keywords—power factor correction, soft switching,
single-stage, AC-DC converter, DC-DC converter
I. INTRODUCTION
IN recent years, power conversion equipment
connected to the grid are constantly increasing. In
order to manage the problems associated with the
harmonic pollution of power conversion equipment
and fulfill harmonic current limits set by standards like
IEC61000-3-2[1], it is imperative to develop power
factor correction (PFC) techniques[2]-[4]. Thus, major
research has been carried out to address the mentioned
issues and develop high performance PFC converters
[2]-[10]. Two-stage cascade PFC converter which
consists of a PFC stage and DC/DC stage is an
approach to achieve a smooth output voltage in
conjunction with a high power factor. By using
separate controllers, these converters can achieve high
performance input current shaping and output voltage
regulation. However, the major drawback of this type
of PFC is its high cost due to high device count (of at
least two switches and a separate controller for each
stage) [3]. In order to overcome this problem, single-
stage PFC converters are developed in which the
current shaping and the DC/DC stages are combined
[3]-[10]. In most single stage structures, the DC/DC
stage switch, together with other elements act also as
the current shaping stage. Single stage PFC converters
are commonly used in low power applications like
LED drivers [2],[6], ballast circuits [7],[8] and battery
chargers [9],[10]. These converters operate under
either continuous conduction mode (CCM) or
discontinuous conduction mode (DCM) [11],[12].
Operating the converter under DCM allows the input
inductor current to depend only on the input voltage
and not on the previous cycle parameters which can
eliminate the current control loop of current shaping
stage and simplify the control circuit and also, make it
possible for the DC/DC stage to achieve fast output
regulation. On the other hand, operating under CCM
condition produces less high order harmonics that
means higher efficiency is possible in CCM [12]. In
the proposed topology, the DCM operation is selected
due to selfpower factor correction characteristic and
also other desired features which are discussed. Soft
switching methods are applied to single stage PFC
converters to improve the efficiency and further
increase the operating switching frequency [13]-[22].
However, soft switching characteristic in these
converters is mostly achieved by using additional
switches and other circuit components which results in
more complicated control scheme and extra cost [14]-
[18]. In addition to these drawbacks, the loss
associated with the newly added circuit is another
concern in such methods. Nevertheless, some of these
PFC converters like boost flyback converters proposed
in [16],[17] are not capable of achieving full soft
switching condition. In other to overcome above
mentioned shortcomings, the idea of single stage-
single switch-soft switching (S6) PFC converters are
developed [13],[19]-[22] which achieve soft switching
condition without any extra switch and only by adding
few passive components and/or extra diodes. An
integrated SEPIC-flyback PFC converter is proposed
in [19] as a LED driver. However the converter only
achieves soft-switching at turn-on. S6 LED drivers
proposed in [20],[21] are capable of achieving low
THD but they have high number of components.
Moreover, they do not achieve soft switching
condition at switch turn off which degrades their
efficiency. In [22] a singlestage isolated power-factor-
corrected power supply (SSIPP) is introduced which
uses a regenerative clamping to reduce the voltage
stress and to recycle the energy trapped in the leakage
inductance. However, same as [20] and [21] it suffers
from high number of semiconductor components and
not achieving soft-switching condition at turn-off. In
[13] a boost flyback S6 PFC converter is proposed in
which the same extra elements used to provide soft
switching at turn off, are employed to replace the
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switch of current shaping stage (Fig.1 (a)). As a result,
a fully soft switched PFC is obtained with no
additional switch, simple control system while no
additional losses are imposed. Also, this converter has
fewer elements as compared to its counterparts, but its
boost inductor charge time depends on the input
voltage value and thus varies with the input voltage
amplitude. To lower the input current THD, it is
preferred that the charge time of boost inductor be
dependent only on the converter duty cycle [11]. To
overcome the drawback of the S6 converter in [13], a
new S6 converter with fewer semiconductor elements
is proposed in this paper. The harmonic content of the
proposed converter fulfill the IEC61000-3-2 class D
harmonic current limits and is noticeably lower than
that of converter in [13]. The proposed converter is
fully soft-switched and also the leakage inductance
energy is recovered to improve the converter
efficiency. In addition, the leakage inductance of the
flyback transformer is employed as the resonant
inductance while no extra switch is used. Furthermore,
the switch zero voltage switching (ZVS) turn-off is
resulted to eliminating the high voltage spike on
switch at turn-off instant which reduces losses,
electromagnetic interference (EMI) and the switch
voltage stress. A prototype of the proposed converter
is implemented to verify the converter theoretical
analysis and operation.
Fig. 1. (a) The S6 PFC converter of [13] (b) Proposed
S6 PFC converter
II. DC to DC (DC to DC converter):
Dc-dc power converters are employed in a
variety of applications, including power supplies for
personal computers, office equipment, spacecraft
power systems, laptop computers, and
telecommunications equipment, as well as dc motor
drives. The input to a dc-dc converter is an unregulated
dc voltage Vg. The converter produces a regulated
output voltage V, having a magnitude (and possibly
polarity) that differs from Vg.
There are three basic types of dc-dc converter
circuits, termed as (I)Buck , (II)Boost and (III)Buck-
boost. In all of these circuits, a power device is used as
a switch. This device earlier used was a thyristor,
which is turned on by a pulse fed at its gate. In all these
circuits, the thyristor is connected in series with load
to a dc supply, or a positive (forward) voltage is
applied between anode and cathode terminals. The
thyristor turns off, when the current decreases below
the holding current, or a reverse (negative) voltage is
applied between anode and cathode terminals. So, a
thyristor is to be force-commutated, for which
additional circuit is to be used. Earlier, dc-dc
converters were called ‘choppers’, where thyristors or
GTOs are used. It may be noted here that buck
converter (dc-dc) is called as ‘step-down chopper’,
whereas boost converter (dc-dc) is a ‘step-up
chopper’. In the case of chopper, no buck-boost type
was used. With the advent of bipolar junction
transistor (BJT), which is termed as self-commutated
device, it is used as a switch, instead of thyristor, in
dc-dc converters. Now-adays, MOSFETs are used as a
switching device in low voltage and high current
applications. It may be noted that, as the turn-on and
turn-off time of MOSFETs are lower as compared to
other switching devices, the frequency used for the dc-
dc converters using it (MOSFET) is high, thus,
reducing the size of filters as stated earlier.
Buck Converter: A buck converter (dc-dc) is shown in Fig..
Only a switch is shown, for which a device as
described earlier belonging to transistor family is used.
Also a diode (termed as free wheeling) is used to allow
the load current to flow through it, when the switch
(i.e., a device) is turned off. The load is inductive (R-
L) one. In some cases, a battery (or back emf) is
connected in series with the load (inductive). Due to
the load inductance, the load current must be allowed
a path, which is provided by the diode; otherwise, i.e.,
in the absence of the above diode, the high induced
emf of the inductance, as the load current tends to
decrease, may cause damage to the switching device.
If the switching device used is a thyristor, this circuit
is called as a step-down chopper, as the output voltage
is normally lower than the input voltage.
Fig.Buck converter
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Boost Converter (dc-dc):
A boost converter (dc-dc) is shown in Fig..
Only a switch is shown, for which a device belonging
to transistor family is generally used. Also, a diode is
used in series with the load. The load is of the same
type as given earlier. The inductance of the load is
small. An inductance, L is assumed in series with the
input supply. The position of the switch and diode in
this circuit may be noted, as compared to their position
in the buck converter. In this case, the output voltage
is higher than the input voltage, as contrasted with the
previous case of buck converter (dc-dc). So, this is
called boost converter (dc-dc), when a self
commutated device is used as a switch. Instead, if
thyristor is used in its place, this is termed as step-up
chopper. The variation (range) of the output voltage
can be easily computed.
Fig: Boost Converter
Buck-Boost Converter
A buck-boost converter (dc-dc) is shown in Fig. Only
a switch is shown, for which a device belonging to
transistor family is generally used. Also, a diode is
used in series with the load. The connection of the
diode may be noted, as compared with its connection
in a boost converter (Fig. 17.2a). The inductor, L is
connected in parallel after the switch and before the
diode. The load is of the same type as given earlier. A
capacitor, C is connected in parallel with the load. The
polarity of the output voltage is opposite to that of
input voltage here.
When the switch, S is put ON, the supply current
flows through the path, S and L, during the time
interval, . The currents through both source and
inductor increase and are same, with being positive.
Fig: Buck-Boost Converter
The polarity of the induced voltage is same as that of
the input voltage.
Then, the switch, S is put OFF. The inductor current
tends to decrease, with the polarity of the induced emf
reversing. is negative now, the polarity of the output
voltage, being opposite to that of the input voltage, .
The path of the current is through L, parallel
combination of load & C, and diode D, during the time
interval, . The output voltage remains nearly constant,
as the capacitor is connected across the load. This
circuit can be termed as a buck-boost converter. Also
it may be called as step-up/down chopper. It may be
noted that the inductor current is assumed to be
continuous.
III.POWER FACTOR
The cosine of angle between voltage and
current in an a.c. circuit is known as power factor. In
an a.c. circuit, there is generally a phase difference φ
between voltage and current. The term cos φ is called
the power factor of the circuit. If the circuit is
inductive, the current lags behind the voltage and the
power factor is referred to as lagging. However, in a
capacitive circuit, current leads the voltage and power
factor is said to be leading. Consider an inductive
circuit taking a lagging current I from supply voltage
V; the angle of lag being φ. The phasor diagram of the
circuit is shown in Fig. 6.1.
The circuit current I can be resolved into two
perpendicular components, namely ;
(a) I cos φ in phase with V
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(b) I sin φ 90o out of phase with V
IV.PROPOSED SYSTEM AND CONTROL
DESIGN
A. Description of the Idea Fig. 1(b) illustrates the
schematic of proposed converter. The converter
consists of a boost inductor (LB), dc-link capacitor
(CB), output capacitor (Co), and resonant capacitor
(Cr), power MOSFET (Sw), input bridge diode
rectifier, high frequency diodes (D1, Do) and a three
windings transformer (T) where Np, Ns, and Na
denote the primary, secondary and auxiliary windings
number of turns respectively. To simplify the
theoretical analysis, it is assumed that all
semiconductors components are ideal, the voltage of
DC-link capacitor is almost constant and also during a
switching period, the input voltage is constant due to
high switching frequency and low line frequency. In
the proposed S6 PFC, the auxiliary winding (Na) is
connected in series with the boost inductor. This
formation allows controlling the boost inductor
voltage and shaping its current which results in
lowering the THD. Also the auxiliary winding diode
in [13] is eliminated in the proposed topology which
improves the efficiency and simplifies the converter.
The input inductor (LB) operates in DCM, similar to
the converter of [13] in order to have intrinsic PFC
characteristic. The input voltage-current characteristic
of a boost converter in DCM mode is as below [11]:
iin(t) = D2Ts 2LB vin(t)Vo Vo − vin(t) = D2Ts 2LB (
1 1 − vin(t) Vo )vin(t) (1) where D is the converter
duty cycle, and vin(t) and iin(t) respectively denote the
input voltage and current. Also Vo is the output
voltage and Ts is the switching period. The relation
between vin and iin depends on duty cycle and vin to
vo ratio. Even if the output voltage is assumed greatly
higher than the peak of the input voltage, the relation
between vin and iin is nearly linear, only if the duty
cycle is fixed and then the converter would have an
inherent PFC property. Thus, if the charge time of the
boost inductor varies with time, then the relation
between vin and iin is not linear enough anymore
which leads to intense amount of harmonic content. It
should be noted that (1) is derived from the inductor
Volt-Second balance. In the converter of [13], the
boost inductor is charged through a circuit which
consists of a rectified input voltage source and Cr.
Thus, the charge time of LB depends on the input
voltage and because the input voltage is time variant,
the charge time of the boost converter varies with time.
This results in deformation of the input current with
increased THD. Also, as soft-switching and even
power factor correction operation of this converter
depends on the input voltage (charging voltage of the
resonant capacitor), the performance would reduce
noticeably by varying input voltage. To solve these
problems, the circuit should be improved in such a way
that the input inductor current is proportional to the
switch on time. To overcome the above mentioned
problem, a new S6 PFC is proposed as illustrated in
Fig. 1(b). In this converter, the auxiliary winding (Na)
is connected in series with the boost inductor (LB) to
form a PFC cell together with the resonant capacitor
(Cr). This special connection of the auxiliary winding
in addition to a proper design can result in the
appropriate control of the input inductor voltage.
Controlling the inductor voltage would lead to
controlling its charge and discharge states. Also due to
presence of bulk capacitor voltage (VCB ) in the
charging loop and its almost constant voltage, the
effect of input voltage variations on converter
performance is reduced. B. Operating Principles In the
proposed converter, both LB and Lm are designed to
operate under DCM condition. This would simplify
the control circuit by eliminating the current shaping
control loop. The proposed converter operating modes
in one switching period are illustrated in Fig. 2 and the
steady state theoretical waveforms are shown in Fig.
3. Stage 1(t0-t1): This mode starts when the switch is
turned on under zero current switching (ZCS) due to
DCM operation of the inductor LB and also the
transformer leakage inductance. Diodes D1 and Do are
reversed biased and LB starts charging. while VCB is
applied to Lm and causes the magnetizing inductance
current (Im) to increase linearly. Concurrently, Cr
which has a negative initial voltage is being charged
by the input inductor current and its voltage increases
until it becomes equal to VCB at the t1. The bulk
capacitor current (ICB ) can be obtained by the
ampere-turns of transformer. The transformer is
modeled with an ideal transformer plus a magnetizing
inductance. ICB = −(Im + (Na/Np)ILB ). (2) The
current of power switch is equal to sum of ICB and LB
current (Isw = ILB − ICB ). As discussed later, LB
operates under DCM condition and due to the
transformer leakage inductance, ICB gradually
increases from zero, thus the switch current slowly
rises from zero which indicates ZCS operation of the
power switch as illustrated in Fig.3.
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Fig. 2. Topological operating modes of proposed
converter: (a)Stage1. (b)Stage 2. (c)Stage 3. (d)Stage
4. (e)Stage 5 (f)Stage 6. (g)Stage 7.
Fig. 3. Steady-state analytical waveforms
Stage 2(t1-t2): This interval begins when D1 starts
conducting and fixes the voltage across LB (VLB ) at
a certain value which can be obtained from the below
equation: VLB = Vin + VNa − VCr = Vin + (Na/Np −
1)VCB . (3) As discussed before, in the proposed
converter, the aim is to keep LB in charging mode till
the switch turns off. To keep LB in charging mode, its
voltage must be positive regardless of the input
voltage value, thus at the worst case (Vin = 0): (Na/Np
− 1)VCB ≥ 0 ⇒ Na ≥ Np. (4) It is not desired to choose
Na > Np because it increases the switch current stress
and the transformer cost. For this reason, choosing Na
= Np is the most suitable case. The power switch
current can be written as below: Isw = (Na/Np)ILB +
ILm. (5) Stage 3(t2-t3): This mode starts when the
switch is turned off under zero voltage switching
(ZVS) due to Cr which is placed in parallel with the
primary winding of transformer (Np) and then slowly
discharges in Lm. This feature is achieved due to the
direct connection of Cr to power switch
and fixed voltage of VCB . In this state, VLB is equal
to: VLB = Vin − VCB + (Na/Np)VCr . (6) Thus, until
VCr reaches Na Np (VCB − Vin), VLB is positive and
the input inductor current is still increasing and then
starts reducing. Due to the large size of Lm and small
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size of Cr, the input inductor charge time in this
interval is very short and thus it can be assumed that
the input inductor current starts reducing after turning
off the switch. As described before, this results in
lower harmonic content of input current. Resonant
capacitor continues its discharge until its voltage
decrease to − Np Na Vo and then Do starts conducting.
Stage 4(t3-t4): At t3, Do starts conducting under ZCS
due to the transformer leakage inductance (Llk) which
begins to resonant with Cr. When the resonant
capacitor current (iCr ) becomes equal to LB current
(ILB ), D1 current reaches zero. Stage 5(t4-t5): After
D1 turns off, Lm starts discharging into the load and
LB is also still discharging into the bulk capacitor.
This state ends when both LB and Lm are totally
discharged. It does not matter which one discharges
first but, here it is assumed that Lm is totally
discharged first. This mode indicates the converter
DCM operation. At this time, Cr is charging due to its
positive current and when its voltage reaches Na Np
(VCB −Vin), D1 starts conducting again and a new
resonance between Cr and Llk would start and it would
end when ICr becomes equal to ILB . However due to
discharge of LB, this resonance is damped faster than
the previous stage and with lower voltage and current.
Not that the resonant energy is basically recovered and
has a minute contribution to the converter losses. The
appearance of the resonance at this mode depends on
many factors such as the input voltage, the values of
LB and Cr, the transformer parameters and the load. If
Lm totally discharges before the voltage of Cr reaches
Na Np (VCB − Vin), the new resonance would not
happen. In order to totally eliminate this resonance, a
diode can be placed in series with the primary winding
as shown in Fig.5. By doing so, after half a resonance
period, the resonant current reaches zero and D2
would turn off and disconnects Cr from the circuit.
Then, D1 continues its conduction until LB is totally
discharged. However, in addition to the extra cost of
the new diode and its conducting loss, the conducting
time of D1 increases which contributes to lowering the
efficiency. Also, as discussed later, it can increase the
switch current stress and thus, in general, placing this
new diode is not recommended. Stage 6(t5-t6): In this
stage LB is still discharging into the Fig. 5. Input
inductance voltage in one switching period bulk
capacitor. This mode ends when LB is totally
discharged which indicates the DCM condition Stage
7(t6-t0): The output capacitor continues to supply the
load in this stage while LB is being charged very
slowly. As can be observed in Fig2.g a loop consists
of LB, Lm, Cr, CB and the input source is formed
which due to the high size of Lm and opposite
direction of VCB with the rectified input voltage, a
very slow charging occurs as illustrated in Fig.3. In
this stage Cr is in series with LB and its voltage is
being increased. As a result, the initial value of VCr at
the beginning of the next stage is higher and reaches
VCB faster which would reduce the time interval of
the first stage. Consequently, softswitching condition
is provided even at low duty cycles and limits the extra
current stress on the switch. Also, the sum of ILB and
ILm which is equal to the switch current is zero
because they are equal with opposite directions, thus
the slow charge of LB in this mode, would not affect
the switch ZCS operation. It should be noted that by
adding the diode D2 to the topology, the input inductor
never charges in this mode, because D2 prevents the
formation of the discussed current loop in stage 7 and
thus the advantages of decreasing the initial voltage of
Cr is not achieved.
Fig. 4. Proposed topology to eliminate the repeatable
resonance between Cr and Llk
V.SIMULATION RESULTS:
Fig: Proposed Simulation Diagram
Fig: Pulse
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Fig: Ilb
Fig: Id1
Fig:Icr
Fig: Vcr
Fig: Vsw
Fig:Isw
Fig: Id0
VI.CONCLUSION
This paper proposes an improved S6 PFC
converter with control in order to improve the input
current THD. The harmonic content of the proposed
converter fulfills class D harmonic current limits of
IEC61000-3-2 standard and is noticeably lower than
the conventional converter. Also soft switching is
provided only by an auxiliary winding and a resonant
capacitor without any extra switch to reduce the total
number of components in comparison with similar soft
switching PFC converters. Low number of
components and soft-switching operation of the
converter has contributed to the converter efficiency
improvement. Also, simple control circuit is required
since the converter operates in DCM. Simulation of
the proposed converter is realized and the performance
of the proposed topology.
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