A New Control Design of Soft-Switching (S6) Boost-Flyback

ISSN NO: 0745-6999 JOURNAL OF RESOURCE MANAGEMENT

AND TECHNOLOGY

Vol12, Issue3, 2021

Page No:408

A New Control Design of Soft-Switching (S6) Boost-Flyback PFC Converter 1Doddi Satheesh, 2Er, P.Pedda Reddy,3Dr.K. Chithambaraiah Setty

1M.Tech Student, 2Asst.Professor, 3Associate Professor

Dept of EEE

St John's College Of Eng And Tecnology

Abstract—This thesis presents a S6 PFC converter to

enhance the current shaping performance and reduce

the total harmonic distortion (THD). This

improvement is achieved by the aid of an auxiliary

winding which is used to lower the input current

harmonics and also achieve soft-switching condition.

As a result, the switching losses are reduced and

harmonic content of the input current is improved

noticeably in comparison to the conventional S6 PFC

converter. Also the total number of semiconductor

elements is reduced in the proposed topology which

results in lower cost and higher efficiency. The

operating modes of the proposed converter are

discussed in details and the design procedure is

presented. A 200 kHz prototype of the proposed

converter is implemented and the obtained results are

provided to verify the converter theoretical analysis

and operation.

Keywords—power factor correction, soft switching,

single-stage, AC-DC converter, DC-DC converter

I. INTRODUCTION

IN recent years, power conversion equipment

connected to the grid are constantly increasing. In

order to manage the problems associated with the

harmonic pollution of power conversion equipment

and fulfill harmonic current limits set by standards like

IEC61000-3-2[1], it is imperative to develop power

factor correction (PFC) techniques[2]-[4]. Thus, major

research has been carried out to address the mentioned

issues and develop high performance PFC converters

[2]-[10]. Two-stage cascade PFC converter which

consists of a PFC stage and DC/DC stage is an

approach to achieve a smooth output voltage in

conjunction with a high power factor. By using

separate controllers, these converters can achieve high

performance input current shaping and output voltage

regulation. However, the major drawback of this type

of PFC is its high cost due to high device count (of at

least two switches and a separate controller for each

stage) [3]. In order to overcome this problem, single-

stage PFC converters are developed in which the

current shaping and the DC/DC stages are combined

[3]-[10]. In most single stage structures, the DC/DC

stage switch, together with other elements act also as

the current shaping stage. Single stage PFC converters

are commonly used in low power applications like

LED drivers [2],[6], ballast circuits [7],[8] and battery

chargers [9],[10]. These converters operate under

either continuous conduction mode (CCM) or

discontinuous conduction mode (DCM) [11],[12].

Operating the converter under DCM allows the input

inductor current to depend only on the input voltage

and not on the previous cycle parameters which can

eliminate the current control loop of current shaping

stage and simplify the control circuit and also, make it

possible for the DC/DC stage to achieve fast output

regulation. On the other hand, operating under CCM

condition produces less high order harmonics that

means higher efficiency is possible in CCM [12]. In

the proposed topology, the DCM operation is selected

due to selfpower factor correction characteristic and

also other desired features which are discussed. Soft

switching methods are applied to single stage PFC

converters to improve the efficiency and further

increase the operating switching frequency [13]-[22].

However, soft switching characteristic in these

converters is mostly achieved by using additional

switches and other circuit components which results in

more complicated control scheme and extra cost [14]-

[18]. In addition to these drawbacks, the loss

associated with the newly added circuit is another

concern in such methods. Nevertheless, some of these

PFC converters like boost flyback converters proposed

in [16],[17] are not capable of achieving full soft

switching condition. In other to overcome above

mentioned shortcomings, the idea of single stage-

single switch-soft switching (S6) PFC converters are

developed [13],[19]-[22] which achieve soft switching

condition without any extra switch and only by adding

few passive components and/or extra diodes. An

integrated SEPIC-flyback PFC converter is proposed

in [19] as a LED driver. However the converter only

achieves soft-switching at turn-on. S6 LED drivers

proposed in [20],[21] are capable of achieving low

THD but they have high number of components.

Moreover, they do not achieve soft switching

condition at switch turn off which degrades their

efficiency. In [22] a singlestage isolated power-factor-

corrected power supply (SSIPP) is introduced which

uses a regenerative clamping to reduce the voltage

stress and to recycle the energy trapped in the leakage

inductance. However, same as [20] and [21] it suffers

from high number of semiconductor components and

not achieving soft-switching condition at turn-off. In

[13] a boost flyback S6 PFC converter is proposed in

which the same extra elements used to provide soft

switching at turn off, are employed to replace the


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switch of current shaping stage (Fig.1 (a)). As a result,

a fully soft switched PFC is obtained with no

additional switch, simple control system while no

additional losses are imposed. Also, this converter has

fewer elements as compared to its counterparts, but its

boost inductor charge time depends on the input

voltage value and thus varies with the input voltage

amplitude. To lower the input current THD, it is

preferred that the charge time of boost inductor be

dependent only on the converter duty cycle [11]. To

overcome the drawback of the S6 converter in [13], a

new S6 converter with fewer semiconductor elements

is proposed in this paper. The harmonic content of the

proposed converter fulfill the IEC61000-3-2 class D

harmonic current limits and is noticeably lower than

that of converter in [13]. The proposed converter is

fully soft-switched and also the leakage inductance

energy is recovered to improve the converter

efficiency. In addition, the leakage inductance of the

flyback transformer is employed as the resonant

inductance while no extra switch is used. Furthermore,

the switch zero voltage switching (ZVS) turn-off is

resulted to eliminating the high voltage spike on

switch at turn-off instant which reduces losses,

electromagnetic interference (EMI) and the switch

voltage stress. A prototype of the proposed converter

is implemented to verify the converter theoretical

analysis and operation.

Fig. 1. (a) The S6 PFC converter of [13] (b) Proposed

S6 PFC converter

II. DC to DC (DC to DC converter):

Dc-dc power converters are employed in a

variety of applications, including power supplies for

personal computers, office equipment, spacecraft

power systems, laptop computers, and

telecommunications equipment, as well as dc motor

drives. The input to a dc-dc converter is an unregulated

dc voltage Vg. The converter produces a regulated

output voltage V, having a magnitude (and possibly

polarity) that differs from Vg.

There are three basic types of dc-dc converter

circuits, termed as (I)Buck , (II)Boost and (III)Buck-

boost. In all of these circuits, a power device is used as

a switch. This device earlier used was a thyristor,

which is turned on by a pulse fed at its gate. In all these

circuits, the thyristor is connected in series with load

to a dc supply, or a positive (forward) voltage is

applied between anode and cathode terminals. The

thyristor turns off, when the current decreases below

the holding current, or a reverse (negative) voltage is

applied between anode and cathode terminals. So, a

thyristor is to be force-commutated, for which

additional circuit is to be used. Earlier, dc-dc

converters were called ‘choppers’, where thyristors or

GTOs are used. It may be noted here that buck

converter (dc-dc) is called as ‘step-down chopper’,

whereas boost converter (dc-dc) is a ‘step-up

chopper’. In the case of chopper, no buck-boost type

was used. With the advent of bipolar junction

transistor (BJT), which is termed as self-commutated

device, it is used as a switch, instead of thyristor, in

dc-dc converters. Now-adays, MOSFETs are used as a

switching device in low voltage and high current

applications. It may be noted that, as the turn-on and

turn-off time of MOSFETs are lower as compared to

other switching devices, the frequency used for the dc-

dc converters using it (MOSFET) is high, thus,

reducing the size of filters as stated earlier.

Buck Converter: A buck converter (dc-dc) is shown in Fig..

Only a switch is shown, for which a device as

described earlier belonging to transistor family is used.

Also a diode (termed as free wheeling) is used to allow

the load current to flow through it, when the switch

(i.e., a device) is turned off. The load is inductive (R-

L) one. In some cases, a battery (or back emf) is

connected in series with the load (inductive). Due to

the load inductance, the load current must be allowed

a path, which is provided by the diode; otherwise, i.e.,

in the absence of the above diode, the high induced

emf of the inductance, as the load current tends to

decrease, may cause damage to the switching device.

If the switching device used is a thyristor, this circuit

is called as a step-down chopper, as the output voltage

is normally lower than the input voltage.

Fig.Buck converter


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Boost Converter (dc-dc):

A boost converter (dc-dc) is shown in Fig..

Only a switch is shown, for which a device belonging

to transistor family is generally used. Also, a diode is

used in series with the load. The load is of the same

type as given earlier. The inductance of the load is

small. An inductance, L is assumed in series with the

input supply. The position of the switch and diode in

this circuit may be noted, as compared to their position

in the buck converter. In this case, the output voltage

is higher than the input voltage, as contrasted with the

previous case of buck converter (dc-dc). So, this is

called boost converter (dc-dc), when a self

commutated device is used as a switch. Instead, if

thyristor is used in its place, this is termed as step-up

chopper. The variation (range) of the output voltage

can be easily computed.

Fig: Boost Converter

Buck-Boost Converter

A buck-boost converter (dc-dc) is shown in Fig. Only

a switch is shown, for which a device belonging to

transistor family is generally used. Also, a diode is

used in series with the load. The connection of the

diode may be noted, as compared with its connection

in a boost converter (Fig. 17.2a). The inductor, L is

connected in parallel after the switch and before the

diode. The load is of the same type as given earlier. A

capacitor, C is connected in parallel with the load. The

polarity of the output voltage is opposite to that of

input voltage here.

When the switch, S is put ON, the supply current

flows through the path, S and L, during the time

interval, . The currents through both source and

inductor increase and are same, with being positive.

Fig: Buck-Boost Converter

The polarity of the induced voltage is same as that of

the input voltage.

Then, the switch, S is put OFF. The inductor current

tends to decrease, with the polarity of the induced emf

reversing. is negative now, the polarity of the output

voltage, being opposite to that of the input voltage, .

The path of the current is through L, parallel

combination of load & C, and diode D, during the time

interval, . The output voltage remains nearly constant,

as the capacitor is connected across the load. This

circuit can be termed as a buck-boost converter. Also

it may be called as step-up/down chopper. It may be

noted that the inductor current is assumed to be

continuous.

III.POWER FACTOR

The cosine of angle between voltage and

current in an a.c. circuit is known as power factor. In

an a.c. circuit, there is generally a phase difference φ

between voltage and current. The term cos φ is called

the power factor of the circuit. If the circuit is

inductive, the current lags behind the voltage and the

power factor is referred to as lagging. However, in a

capacitive circuit, current leads the voltage and power

factor is said to be leading. Consider an inductive

circuit taking a lagging current I from supply voltage

V; the angle of lag being φ. The phasor diagram of the

circuit is shown in Fig. 6.1.

The circuit current I can be resolved into two

perpendicular components, namely ;

(a) I cos φ in phase with V


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(b) I sin φ 90o out of phase with V

IV.PROPOSED SYSTEM AND CONTROL

DESIGN

A. Description of the Idea Fig. 1(b) illustrates the

schematic of proposed converter. The converter

consists of a boost inductor (LB), dc-link capacitor

(CB), output capacitor (Co), and resonant capacitor

(Cr), power MOSFET (Sw), input bridge diode

rectifier, high frequency diodes (D1, Do) and a three

windings transformer (T) where Np, Ns, and Na

denote the primary, secondary and auxiliary windings

number of turns respectively. To simplify the

theoretical analysis, it is assumed that all

semiconductors components are ideal, the voltage of

DC-link capacitor is almost constant and also during a

switching period, the input voltage is constant due to

high switching frequency and low line frequency. In

the proposed S6 PFC, the auxiliary winding (Na) is

connected in series with the boost inductor. This

formation allows controlling the boost inductor

voltage and shaping its current which results in

lowering the THD. Also the auxiliary winding diode

in [13] is eliminated in the proposed topology which

improves the efficiency and simplifies the converter.

The input inductor (LB) operates in DCM, similar to

the converter of [13] in order to have intrinsic PFC

characteristic. The input voltage-current characteristic

of a boost converter in DCM mode is as below [11]:

iin(t) = D2Ts 2LB vin(t)Vo Vo − vin(t) = D2Ts 2LB (

1 1 − vin(t) Vo )vin(t) (1) where D is the converter

duty cycle, and vin(t) and iin(t) respectively denote the

input voltage and current. Also Vo is the output

voltage and Ts is the switching period. The relation

between vin and iin depends on duty cycle and vin to

vo ratio. Even if the output voltage is assumed greatly

higher than the peak of the input voltage, the relation

between vin and iin is nearly linear, only if the duty

cycle is fixed and then the converter would have an

inherent PFC property. Thus, if the charge time of the

boost inductor varies with time, then the relation

between vin and iin is not linear enough anymore

which leads to intense amount of harmonic content. It

should be noted that (1) is derived from the inductor

Volt-Second balance. In the converter of [13], the

boost inductor is charged through a circuit which

consists of a rectified input voltage source and Cr.

Thus, the charge time of LB depends on the input

voltage and because the input voltage is time variant,

the charge time of the boost converter varies with time.

This results in deformation of the input current with

increased THD. Also, as soft-switching and even

power factor correction operation of this converter

depends on the input voltage (charging voltage of the

resonant capacitor), the performance would reduce

noticeably by varying input voltage. To solve these

problems, the circuit should be improved in such a way

that the input inductor current is proportional to the

switch on time. To overcome the above mentioned

problem, a new S6 PFC is proposed as illustrated in

Fig. 1(b). In this converter, the auxiliary winding (Na)

is connected in series with the boost inductor (LB) to

form a PFC cell together with the resonant capacitor

(Cr). This special connection of the auxiliary winding

in addition to a proper design can result in the

appropriate control of the input inductor voltage.

Controlling the inductor voltage would lead to

controlling its charge and discharge states. Also due to

presence of bulk capacitor voltage (VCB ) in the

charging loop and its almost constant voltage, the

effect of input voltage variations on converter

performance is reduced. B. Operating Principles In the

proposed converter, both LB and Lm are designed to

operate under DCM condition. This would simplify

the control circuit by eliminating the current shaping

control loop. The proposed converter operating modes

in one switching period are illustrated in Fig. 2 and the

steady state theoretical waveforms are shown in Fig.

3. Stage 1(t0-t1): This mode starts when the switch is

turned on under zero current switching (ZCS) due to

DCM operation of the inductor LB and also the

transformer leakage inductance. Diodes D1 and Do are

reversed biased and LB starts charging. while VCB is

applied to Lm and causes the magnetizing inductance

current (Im) to increase linearly. Concurrently, Cr

which has a negative initial voltage is being charged

by the input inductor current and its voltage increases

until it becomes equal to VCB at the t1. The bulk

capacitor current (ICB ) can be obtained by the

ampere-turns of transformer. The transformer is

modeled with an ideal transformer plus a magnetizing

inductance. ICB = −(Im + (Na/Np)ILB ). (2) The

current of power switch is equal to sum of ICB and LB

current (Isw = ILB − ICB ). As discussed later, LB

operates under DCM condition and due to the

transformer leakage inductance, ICB gradually

increases from zero, thus the switch current slowly

rises from zero which indicates ZCS operation of the

power switch as illustrated in Fig.3.


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Fig. 2. Topological operating modes of proposed

converter: (a)Stage1. (b)Stage 2. (c)Stage 3. (d)Stage

4. (e)Stage 5 (f)Stage 6. (g)Stage 7.

Fig. 3. Steady-state analytical waveforms

Stage 2(t1-t2): This interval begins when D1 starts

conducting and fixes the voltage across LB (VLB ) at

a certain value which can be obtained from the below

equation: VLB = Vin + VNa − VCr = Vin + (Na/Np −

1)VCB . (3) As discussed before, in the proposed

converter, the aim is to keep LB in charging mode till

the switch turns off. To keep LB in charging mode, its

voltage must be positive regardless of the input

voltage value, thus at the worst case (Vin = 0): (Na/Np

− 1)VCB ≥ 0 ⇒ Na ≥ Np. (4) It is not desired to choose

Na > Np because it increases the switch current stress

and the transformer cost. For this reason, choosing Na

= Np is the most suitable case. The power switch

current can be written as below: Isw = (Na/Np)ILB +

ILm. (5) Stage 3(t2-t3): This mode starts when the

switch is turned off under zero voltage switching

(ZVS) due to Cr which is placed in parallel with the

primary winding of transformer (Np) and then slowly

discharges in Lm. This feature is achieved due to the

direct connection of Cr to power switch

and fixed voltage of VCB . In this state, VLB is equal

to: VLB = Vin − VCB + (Na/Np)VCr . (6) Thus, until

VCr reaches Na Np (VCB − Vin), VLB is positive and

the input inductor current is still increasing and then

starts reducing. Due to the large size of Lm and small


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size of Cr, the input inductor charge time in this

interval is very short and thus it can be assumed that

the input inductor current starts reducing after turning

off the switch. As described before, this results in

lower harmonic content of input current. Resonant

capacitor continues its discharge until its voltage

decrease to − Np Na Vo and then Do starts conducting.

Stage 4(t3-t4): At t3, Do starts conducting under ZCS

due to the transformer leakage inductance (Llk) which

begins to resonant with Cr. When the resonant

capacitor current (iCr ) becomes equal to LB current

(ILB ), D1 current reaches zero. Stage 5(t4-t5): After

D1 turns off, Lm starts discharging into the load and

LB is also still discharging into the bulk capacitor.

This state ends when both LB and Lm are totally

discharged. It does not matter which one discharges

first but, here it is assumed that Lm is totally

discharged first. This mode indicates the converter

DCM operation. At this time, Cr is charging due to its

positive current and when its voltage reaches Na Np

(VCB −Vin), D1 starts conducting again and a new

resonance between Cr and Llk would start and it would

end when ICr becomes equal to ILB . However due to

discharge of LB, this resonance is damped faster than

the previous stage and with lower voltage and current.

Not that the resonant energy is basically recovered and

has a minute contribution to the converter losses. The

appearance of the resonance at this mode depends on

many factors such as the input voltage, the values of

LB and Cr, the transformer parameters and the load. If

Lm totally discharges before the voltage of Cr reaches

Na Np (VCB − Vin), the new resonance would not

happen. In order to totally eliminate this resonance, a

diode can be placed in series with the primary winding

as shown in Fig.5. By doing so, after half a resonance

period, the resonant current reaches zero and D2

would turn off and disconnects Cr from the circuit.

Then, D1 continues its conduction until LB is totally

discharged. However, in addition to the extra cost of

the new diode and its conducting loss, the conducting

time of D1 increases which contributes to lowering the

efficiency. Also, as discussed later, it can increase the

switch current stress and thus, in general, placing this

new diode is not recommended. Stage 6(t5-t6): In this

stage LB is still discharging into the Fig. 5. Input

inductance voltage in one switching period bulk

capacitor. This mode ends when LB is totally

discharged which indicates the DCM condition Stage

7(t6-t0): The output capacitor continues to supply the

load in this stage while LB is being charged very

slowly. As can be observed in Fig2.g a loop consists

of LB, Lm, Cr, CB and the input source is formed

which due to the high size of Lm and opposite

direction of VCB with the rectified input voltage, a

very slow charging occurs as illustrated in Fig.3. In

this stage Cr is in series with LB and its voltage is

being increased. As a result, the initial value of VCr at

the beginning of the next stage is higher and reaches

VCB faster which would reduce the time interval of

the first stage. Consequently, softswitching condition

is provided even at low duty cycles and limits the extra

current stress on the switch. Also, the sum of ILB and

ILm which is equal to the switch current is zero

because they are equal with opposite directions, thus

the slow charge of LB in this mode, would not affect

the switch ZCS operation. It should be noted that by

adding the diode D2 to the topology, the input inductor

never charges in this mode, because D2 prevents the

formation of the discussed current loop in stage 7 and

thus the advantages of decreasing the initial voltage of

Cr is not achieved.

Fig. 4. Proposed topology to eliminate the repeatable

resonance between Cr and Llk

V.SIMULATION RESULTS:

Fig: Proposed Simulation Diagram

Fig: Pulse


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Fig: Ilb

Fig: Id1

Fig:Icr

Fig: Vcr

Fig: Vsw

Fig:Isw

Fig: Id0

VI.CONCLUSION

This paper proposes an improved S6 PFC

converter with control in order to improve the input

current THD. The harmonic content of the proposed

converter fulfills class D harmonic current limits of

IEC61000-3-2 standard and is noticeably lower than

the conventional converter. Also soft switching is

provided only by an auxiliary winding and a resonant

capacitor without any extra switch to reduce the total

number of components in comparison with similar soft

switching PFC converters. Low number of

components and soft-switching operation of the

converter has contributed to the converter efficiency

improvement. Also, simple control circuit is required

since the converter operates in DCM. Simulation of

the proposed converter is realized and the performance

of the proposed topology.

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