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RF and Microwave PowerAmplifier and TransmitterTechnologies Part 2

By Frederick H. Raab, Peter Asbeck, Steve Cripps, Peter B. Kenington,Zoya B. Popovich, Nick Pothecary, John F. Sevic and Nathan O. Sokal

Part 1 of this seriesintroduced basicconcepts, discussedthe characteristics of sig-nals to be amplified, andgave background infor-mation on RF powerdevices. Part 2 reviewsthe basic techniques, rat-ings, and implementation

methods for power amplifiers operating at HFthrough microwave frequencies.

6a. BASIC TECHNIQUES FOR RF POWER AMPLIFICATION

RF power amplifiers are commonly desig-nated as classes A, B, C, D, E, and F [19]. Allbut class A employ various nonlinear, switch-ing, and wave-shaping techniques. Classes ofoperation differ not in only the method ofoperation and efficiency, but also in theirpower-output capability. The power-outputcapability (transistor utilization factor) isdefined as output power per transistor nor-malized for peak drain voltage and current of1 V and 1 A, respectively. The basic topologies(Figures 7, 8 and 9) are single-ended, trans-former-coupled, and complementary. Thedrain voltage and current waveforms of select-ed ideal PAs are shown in Figure 10.

Class AIn class A, the quiescent current is large

enough that the transistor remains at alltimes in the active region and acts as a cur-rent source, controlled by the drive.

Consequently, the drain voltage and currentwaveforms are (ideally) both sinusoidal. Thepower output of an ideal class-A PA is

Po = Vom2 / 2R (5)

where output voltage Vom on load R cannotexceed supply voltage VDD. The DC-powerinput is constant and the efficiency of an idealPA is 50 percent at PEP. Consequently, theinstantaneous efficiency is proportional to thepower output and the average efficiency isinversely proportional to the peak-to-averageratio (e.g., 5 percent for x = 10 dB). The uti-lization factor is 1/8.

For amplification of amplitude-modulatedsignals, the quiescent current can be varied inproportion to the instantaneous signal enve-lope. While the efficiency at PEP isunchanged, the efficiency for lower ampli-

Our multi-part series onpower amplifier tech-

nologies and applicationscontinues with a review of

amplifier configurations,classes of operation,

device characterizationand example applications

This series of articles is an expanded version of the paper, Power Amplifiers and Transmitters for RF andMicrowave by the same authors, which appeared in the the 50th anniversary issue of the IEEE Transactions onMicrowave Theory and Techniques, March 2002. 2002 IEEE. Reprinted with permission.

Figure 7 A single-ended power amplifier.

From May 2003 High Frequency ElectronicsCopyright 2003 Summit Technical Media, LLC

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tudes is considerably improved. In anFET PA, the implementationrequires little more than variation ofthe gate-bias voltage.

The amplification process in classA is inherently linear, hence increas-ing the quiescent current or decreas-ing the signal level monotonicallydecreases IMD and harmonic levels.Since both positive and negativeexcursions of the drive affect thedrain current, it has the highest gainof any PA. The absence of harmonicsin the amplification process allowsclass A to be used at frequencies closeto the maximum capability (fmax) ofthe transistor. However, the efficiencyis low. Class-A PAs are therefore typ-ically used in applications requiringlow power, high linearity, high gain,broadband operation, or high-fre-quency operation.

The efficiency of real class-A PAsis degraded by the on-state resistance

or saturation voltage of the transis-tor. It is also degraded by the pres-ence of load reactance, which inessence requires the PA to generatemore output voltage or current todeliver the same power to the load.

Class BThe gate bias in a class-B PA is

set at the threshold of conduction sothat (ideally) the quiescent drain cur-rent is zero. As a result, the transis-tor is active half of the time and thedrain current is a half sinusoid.Since the amplitude of the drain cur-rent is proportional to drive ampli-tude and the shape of the drain-cur-rent waveform is fixed, class-B pro-vides linear amplification.

The power output of a class-B PAis controlled by the drive level andvaries as given by eq. (5). The DC-input current is, however, proportion-al to the drain current which is in

turn proportional to the RF-outputcurrent. Consequently, the instanta-neous efficiency of a class-B PAvaries with the output voltage andfor an ideal PA reaches /4 (78.5 per-cent) at PEP. For low-level signals,class B is significantly more efficientthan class A, and its average efficien-cy can be several times that of class Aat high peak-to-average ratios (e.g.,28 vs. 5 percent for = 10 dB). Theutilization factor is the same 0.125 ofclass A.

In practice, the quiescent currentis on the order of 10 percent of thepeak drain current and adjusted tominimize crossover distortion causedby transistor nonlinearities at lowoutputs. Class B is generally used ina push-pull configuration so that thetwo drain-currents add together toproduce a sine-wave output. At HFand VHF, the transformer-coupledpush-pull topology (Figure 8) is gen-erally used to allow broadband oper-ation with minimum filtering. Theuse of the complementary topology(Figure 9) has generally been limitedto audio, LF, and MF applications bythe lack of suitable p-channel tran-sistors. However, this topology isattractive for IC implementation andhas recently been investigated forlow-power applications at frequen-cies to 1 GHz [20].

Class CIn the classical (true) class-C PA,

the gate is biased below threshold sothat the transistor is active for lessthan half of the RF cycle (Figure 10).Linearity is lost, but efficiency isincreased. The efficiency can beincreased arbitrarily toward 100 per-cent by decreasing the conductionangle toward zero. Unfortunately,this causes the output power (utiliza-tion factor) to decrease toward zeroand the drive power to increasetoward infinity. A typical compromiseis a conduction angle of 150 and anideal efficiency of 85 percent.

The output filter of a true class-CPA is a parallel-tuned type that

Figure 8 Transformer-coupledpush-pull PA.

Figure 9 Complementary PA. Figure 10 Wavefrorms for ideal PAs.

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bypasses the harmonic componentsof the drain current to ground with-out generating harmonic voltages.When driven into saturation, effi-ciency is stabilized and the outputvoltage locked to supply voltage,allowing linear high-level amplitudemodulation.

Classical class C is widely used inhigh-power vacuum-tube transmit-ters. It is, however, little used insolid-state PAs because it requireslow drain resistances, making imple-mentation of parallel-tuned outputfilters difficult. With BJTs, it is alsodifficult to set up bias and drive toproduce a true class-C collector-cur-rent waveform. The use of a series-tuned output filter results in amixed-mode class-C operation that ismore like mistuned class E than trueclass C.

Class DClass-D PAs use two or more tran-

sistors as switches to generate asquare drain-voltage waveform. Aseries-tuned output filter passes onlythe fundamental-frequency compo-nent to the load, resulting in poweroutputs of (8/2)VDD

2/R and(2/2)VDD

2/R for the transformer-cou-pled and complementary configura-tions, respectively. Current is drawnonly through the transistor that ison, resulting in a 100-percent effi-ciency for an ideal PA. The utilizationfactor (1/2 = 0.159) is the highest ofany PA (27 percent higher than thatof class A or B). A unique aspect ofclass D (with infinitely fast switch-ing) is that efficiency is not degradedby the presence of reactance in theload.

Practical class-D PAs suffer fromlosses due to saturation, switchingspeed, and drain capacitance. Finiteswitching speed causes the transis-tors to be in their active regions whileconducting current. Drain capaci-tances must be charged and dis-charged once per RF cycle. The asso-ciated power loss is proportional toVDD

3/2 [21] and increases directly

with frequency.Class-D PAs with power outputs

of 100 W to 1 kW are readily imple-mented at HF, but are seldom usedabove lower VHF because of lossesassociated with the drain capaci-tance. Recently, however, experimen-tal class-D PAs have been tested withfrequencies of operation as high as 1GHz [22].

Class EClass E employs a single transis-

tor operated as a switch. The drain-voltage waveform is the result of thesum of the DC and RF currentscharging the drain-shunt capaci-tance. In optimum class E, the drainvoltage drops to zero and has zeroslope just as the transistor turns on.The result is an ideal efficiency of 100percent, elimination of the lossesassociated with charging the draincapacitance in class D, reduction ofswitching losses, and good toleranceof component variation.

Optimum class-E operationrequires a drain shunt susceptance0.1836/R and a drain series reac-tance 1.15R and delivers a power out-put of 0.577VDD

2/R for an ideal PA[23]. The utilization factor is 0.098.Variations in load impedance andshunt susceptance cause the PA todeviate from optimum operation [24,25], but the degradations in perfor-mance are generally no worse thanthose for class A and B.

The capability for efficient opera-tion in the presence of significantdrain capacitance makes class E use-ful in a number of applications. Oneexample is high-efficiency HF PAswith power levels to 1 kW based uponlow-cost MOSFETs intended forswitching rather than RF use [26].Another example is the switching-mode operation at frequencies ashigh as K band [27]. The class-DE PA[28] similarly uses dead-spacebetween the times when its two tran-sistors are on to allow the load net-work to charge/discharge the draincapacitances.

Class FClass F boosts both efficiency and

output by using harmonic resonatorsin the output network to shape thedrain waveforms. The voltage wave-form includes one or more odd har-monics and approximates a squarewave, while the current includes evenharmonics and approximates a halfsine wave. Alternately (inverse classF), the voltage can approximate ahalf sine wave and the current asquare wave. As the number of har-monics increases, the efficiency of anideal PA increases from the 50 per-cent (class A) toward unity (class D)and the utilization factor increasesfrom 1/8 (class A) toward 1/2 (classD) [29].

The required harmonics can inprinciple be produced by current-source operation of the transistor.However, in practice the transistor isdriven into saturation during part ofthe RF cycle and the harmonics areproduced by a self-regulating mecha-nism similar to that of saturatingclass C. Use of a harmonic voltagerequires creating a high impedance(3 to 10 times the load impedance) atthe drain, while use of a harmoniccurrent requires a low impedance(1/3 to 1/10 of the load impedance).While class F requires a more com-plex output filter than other PAs, theimpedances must be correct at only afew specific frequencies. Lumped-ele-ment traps are used at lower fre-quencies and transmission lines areused at microwave frequencies.Typically, a shorting stub is placed aquarter or half-wavelength awayfrom the drain. Since the stubs fordifferent harmonics interact and theopen or short must be created at avirtual drain ahead of the draincapacitance and bond-wire induc-tance, implementation of suitablenetworks is a bit of an art.Nonetheless, class-F PAs are success-fully implemented from MF throughKa band.

A variety of modes of operation in-between class C, E, and F are possi-

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ble. The maximum achievable effi-ciency [30] depends upon the numberof harmonics, (0.5, 0.707, 0.8165,0.8656, 0.9045 for 1 through 5 har-monics, respectively). The utilizationfactor depends upon the harmonicimpedances and is highest for idealclass-F operation.

6b. LOAD-PULL CHARACTERIZATION

RF-power transistors are charac-terized by breakdown voltages andsaturated drain currents. The combi-nation of the resultant maximumdrain voltage and maximum draincurrent dictates a range of loadimpedances into which useful powercan be delivered, as well as animpedance for delivery of the maxi-mum power. The load impedance formaximum power results in drainvoltage and current excursions fromnear zero to nearly the maximumrated values.

The load impedances correspond-ing to delivery of a given amount ofRF power with a specified maximumdrain voltage lie along parallel-resis-

tance lines on the Smith chart. Theimpedances for a specified maximumcurrent analogously follow a series-resistance line. For an ideal PA, theresultant constant-power contour isfootball-shaped as shown in Figure11.

In a real PA, the ideal drain isembedded behind the drain capaci-tance and bond-wire/package induc-tance. Transformation of the idealdrain impedance through these ele-ments causes the constant-powercontours to become rotated and dis-torted [31]. With the addition of sec-ond-order effects, the contoursbecome elliptical. A set of power con-tours for a given PA somewhatresembles a set of contours for a con-jugate match. However, a true conju-gate match produces circular con-tours. With a power amplifier, theprocess is more correctly viewed asloading to produce a desired poweroutput. As shown in the example ofFigure 12, the power and efficiencycontours are not necessarily aligned,nor do maximum power and maxi-mum efficiency necessarily occur forthe same load impedance. Sets ofsuch load-pull contours are widelyused to facilitate design trade-offs.

Load-pull analyses are generallyiterative in nature, as changing one

parameter may produce a new set ofcontours. A variety of differentparameters can be plotted during aload-pull analysis, including not onlypower and efficiency, but also distor-tion and stability. Harmonicimpedances as well as driveimpedances are also sometimes var-ied.

A load-pull system consists essen-tially of a test fixture, provided withbiasing capabilities, and a pair of low-loss, accurately resettable tuners,usually of precision mechanical con-struction. A load-pull characteriza-tion procedure consists essentially ofmeasuring the power of a device, to agiven specification (e.g., the 1-dBcompression point) as a function ofimpedance. Data are measured at alarge number of impedances andplotted on a Smith chart. Such plotsare, of course, critically dependent onthe accurate calibration of the tuners,both in terms of impedance and loss-es. Such calibration is, in turn, highlydependent on the repeatability of thetuners.

Precision mechanical tuners, withmicrometer-style adjusters, were thetraditional apparatus for load-pullanalysis. More recently, a new gener-ation of electronic tuners hasemerged that tune through the usevaractors or transmission linesswitched by pin diodes. Such elec-tronic tuners [32] have the advantageof almost perfect repeatability andhigh tuning speed, but have muchhigher losses and require highly com-plex calibration routines. Mechanicaltuners are more difficult to controlusing a computer, and move veryslowly from one impedance setting toanother.

In an active load-pull system, asecond power source, synchronized infrequency and phase with the deviceinput excitation, is coupled into theoutput of the device. By controllingthe amplitude and phase of theinjected signal, a wide range ofimpedances can be simulated at theoutput of the test device [33]. Such a

Figure 11 Contant power contoursand transformation.

Figure 12 Example load-pull con-tours for a 0.5-W, 836 MHz PA.(Courtesy Focus Microwaves anddBm Engineering)

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system eliminates the expensivetuners, but creates a substantial cali-bration challenge of its own. The wideavailability of turn-key load-pull sys-tems has generally reduced the appli-cation of active load-pull to situationswhere mechanical or electronic tun-ing becomes impractical (e.g., mil-limeter-wave frequencies).

6c. STABILITYThe stability of a small-signal RF

amplifier is ensured by deriving a setof S-parameters from using mea-sured data or a linear model, andthen establishing the value of the k-factor stability parameter. If the k-factor is greater than unity, at thefrequency and bias level in question,then expressions for matchingimpedances at input and output canbe evaluated to give a perfect conju-gate match for the device. Amplifierdesign in this context is mainly amatter of designing matching net-works which present the prescribedimpedances over the necessary speci-fied bandwidth. If the k factor is lessthan unity, negative feedback or lossymatching must be employed in orderto maintain an unconditionally stabledesign.

A third case is relevant to PAdesign at higher microwave frequen-cies. There are cases where a devicehas a very high k-factor value, butvery low gain in conjugate matchedcondition. The physical cause of thiscan be traced to a device which hasgain roll-off due to carrier-mobilityeffects, rather than parasitics. Insuch cases, introduction of some posi-tive feedback reduces the k-factorand increases the gain in conjugatelymatched conditions, while maintain-ing unconditional stability. This tech-nique was much used in the early eraof vacuum-tube electronics, especiallyin IF amplifiers.

6d. MICROWAVE IMPLEMENTATIONAt microwave frequencies, lumped

elements (capacitors, inductors)become unsuitable as tuning compo-

nents and are used primarily aschokes and by-passes. Matching, tun-ing, and filtering at microwave fre-quencies are therefore accomplishedwith distributed (transmission-line)networks. Proper operation of poweramplifiers at microwave frequenciesis achieved by providing the requireddrain-load impedance at the funda-mental and a number of harmonicfrequencies.

Class FClass-F operation is specified in

terms of harmonic impedances, so itis relatively easy to see how trans-mission-line networks are used.Methods for using transmission linesin conjunction with lumped-elementtuned circuits appear in the originalpaper by Tyler [34]. In modernmicrowave implementation, however,it is generally necessary to use trans-mission lines exclusively. In addition,the required impedances must beproduced at a virtual ideal drain thatis separated from the output networkby drain capacitance, bond-wire/leadinductance.

Typically, a transmission linebetween the drain and the load pro-vides the fundamental-frequencydrain impedance of the desired value.A stub that is a quarter wavelengthat the harmonic of interest and openat one end provides a short circuit atthe opposite end. The stub is placedalong the main transmission line ateither a quarter or a half wavelengthfrom the drain to create either anopen or a short circuit at the drain[35]. The supply voltage is fed to thedrain through a half-wavelength linebypassed on the power-supply end oralternately by a lumped-elementchoke. When multiple stubs are used,the stub for the highest controlledharmonic is placed nearest the drain.Stubs for lower harmonics are placedprogressively further away and theirlengths and impedances are adjustedto allow for interactions. Typically,open means three to ten times thefundamental-frequency impedance,

and shorted means no more 1/10 to1/3 of the fundamental-frequencyimpedance [FR17].

A wide variety of class-F PAs havebeen implemented at UHF andmicrowave frequencies [36-41].Generally, only one or two harmonicimpedances are controlled. In the X-band PA from [42], for example, theoutput circuit provides a match at thefundamental and a short circuit atthe second harmonic. The third-har-monic impedance is high, but notexplicitly adjusted to be open. The 3-dB bandwidth of such an output net-work is about 20 percent, and the effi-ciency remains within 10 percent ofits maximum value over a bandwidthof approximately 10 to 15 percent.

Dielectric resonators can be usedin lieu of lumped-element traps inclass-F PAs. Power outputs of 40 Whave been obtained at 11 GHz withefficiencies of 77 percent [43].

Class EThe drain-shunt capacitance and

series inductive reactance requiredfor optimum class-E operation resultin a drain impedance of R + j0.725Rat the fundamental frequency,j1.7846R at the second harmonic,and proportionately smaller capaci-tive reactances at higher harmonics.At microwave frequencies, class-Eoperation is approximated by provid-ing the drain with the fundamental-frequency impedance and preferablyone or more of the harmonicimpedances [44].

An example of a microwaveapproximation of class E that pro-vides the correct fundamental andsecond-harmonic impedances [44] isshown in Figure 13. Line l2 is a quar-ter-wavelength long at the secondharmonic so that the open circuit atits end is transformed to a short atplane AA'. Line l1 in combinationwith L and C is designed to be also aquarter wavelength to translate theshort at AA' to an open at the tran-sistor drain. The lines l1 to l4 providethe desired impedance at the funda-

July 2003 31

mental. The implementation using anFLK052 MESFET is shown in Figure14 produces 0.68 W at X band with adrain efficiency of 72 percent andPAE of 60 percent [42].

Methods exist for providing theproper impedances through thefourth harmonic [45]. However, theharmonic impedances are not critical[30], and many variations are there-fore possible. Since the transistoroften has little or no gain at the high-er harmonic frequencies, thoseimpedances often have little or noeffect upon performance. A single-stub match is often sufficient to pro-vide the desired impedance at thefundamental while simultaneouslyproviding an adequately highimpedance at the second harmonic,thus eliminating the need for anextra stub and reducing a portion ofthe losses associated with it. Mostmicrowave class-E amplifiers operatein a suboptimum mode [46].Demonstrated capabilities rangefrom 16 W with 80-percent efficiencyat UHF (LDMOS) to 100 mW with60-percent efficiency at 10 GHz [47],[48], [44], [49], [50], [51]. Optical sam-pling of the waveforms [52] has veri-fied that these PAs do indeed operatein class E.

ComparisonPAs configured for classes A (AB),

E, and F are compared experimental-ly in [50] with the following conclu-sions. Classes AB and F have essen-tially the same saturated output

power, but class F has about 15 per-cent higher efficiency. Class E has thehighest efficiency. Gain compressionoccurs at a lower power level for classE than for class F. For a given effi-ciency, class F produces more power.For the same maximum outputpower, the third order intermodula-tion products are about 10 dB lowerfor class F than for class E. Lower-power PAs implemented with smallerRF power devices tend to be moreefficient than PAs implemented withlarger devices [42].

Millimeter-Wave PAsSolid-state PAs for millimeter-

wave (mm-W) frequencies (30 to 100GHz) are predominantly monolithic.Most Ka-band PAs are based uponpHEMT devices, while most W-bandPAs are based upon InP HEMTs.Some use is also made of HBTs at thelower mm-W frequencies. Class A isused for maximum gain. Typical per-formance characteristics include 4 Wwith 30-percent PAE at Ka band [53],250 mW with 25-percent PAE at Qband [54], and 200 mW with 10-per-cent PAE at W band [55]. Devices foroperation at mm-W are inherentlysmall, so large power outputs areobtained by combining the outputs ofmultiple low-power amplifiers in cor-porate or spatial power combiners.

6e. EXAMPLE APPLICATIONSThe following examples illustrate

the wide variety of power amplifiersin use today:

HF/VHF Single SidebandOne of the first applications of

RF-power transistors was linearamplification of HF single-sidebandsignals. Many PAs developed byHelge Granberg have been widelyadapted for this purpose [56, 57]. The300-W PA for 2 to 30 MHz uses a pairof Motorola MRF422 Si NPN transis-tors in a push-pull configuration. ThePA operates in class AB push-pullfrom a 28-V supply and achieves acollector efficiency of about 45 per-cent (CW) and a two-tone IMD ratioof about 30 dBc. The 1-kW amplifieris based upon a push-pull pair ofMRF154 MOSFETs and operatesfrom a 50-V supply. Over the frequen-cy range of 2 to 50 MHz it achieves adrain efficiency of about 58 percent(CW) with an IMD rating of 30 dBc.

13.56-MHz ISM Power SourcesHigh-power signals at 13.56 MHz

are needed for a wide variety ofIndustrial, Scientific, and Medical(ISM) applications such as plasmageneration, RF heating, and semicon-ductor processing. A 400-W class-EPA uses an International RectifierIRFP450LC MOSFET (normallyused for low-frequency switching-mode DC power supplies) operatesfrom a 120-V supply and achieves adrain efficiency of 86 percent [58, 26].Industrial 13.56-MHz RF power gen-erators using class-E output stageshave been manufactured since 1992by Dressler Hochfrequenztechnik(Stolberg, Germany) and Advanced

Figure 13 Idealized microwave class-E PA circuit. Figure 14 Example X-band class-E PA.

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Energy Industries (Ft. Collins, CO).They typically use RF-powerMOSFETs with 500- to 900-V break-down voltages made by DirectedEnergy or Advanced PowerTechnology and produce output pow-ers of 500 W to with 3 kW with drainefficiencies of about 90 percent. TheAdvanced Energy Industries amplifi-er (Figure 15) uses thick-film-hybridcircuits to reduce size. This allowsplacement inside the clean-roomfacilities of semiconductor-manufac-turing plants, eliminating the needfor long runs of coaxial cable from anRF-power generator installed outsidethe clean-room.

VHF FM Broadcast TransmitterFM-broadcast transmitters (88 to

108 MHz) with power outputs from50 W to 10 kW are manufactured byBroadcast Electronics (Quincy,Illinois). These transmitters use up to32 power-combined PAs based uponMotorola MRF151G MOSFETs. ThePAs operate in class C from a 44-Vsupply and achieve a drain efficiencyof 80 percent. Typically, about 6 per-cent of the output power is dissipatedin the power combiners, harmonic-suppression filter, and lightning-pro-tection circuit.

MF AM Broadcast TransmittersSince the 1980s, AM broadcast

transmitters (530 to 1710 kHz) havebeen made with class-D and -E RF-output stages. Amplitude modulationis produced by varying the supplyvoltage of the RF PA with a high-effi-ciency amplitude modulator.

Transmitters made by Harris(Mason, Ohio) produce peak-envelopeoutput powers of 58, 86, 150, 300, and550 kW (unmodulated carrier powersof 10, 15, 25, 50, and 100 kW). The100-kW transmitter combines theoutput power from 1152 transistors.The output stages can use eitherbipolars or MOSFETs, typically oper-ate in class DE from a 300-V supply,and achieve an efficiency of 98 per-cent. The output section of the Harris3DX50 transmitter is shown inFigure 16.

Transmitters made by BroadcastElectronics (Quincy, IL) use class-ERF-output stages based uponAPT6015LVR MOSFETs operatingfrom 130-V maximum supply volt-ages. They achieve drain efficienciesof about 94 percent with peak-enve-lope output powers from 4.4 to 44 kW.The 44-kW AM-10A transmitter com-bines outputs from 40 individual out-put stages.

900-MHz Cellular-TelephoneHandset

Most 900-MHz CDMA handsetsuse power-amplifier modules fromvendors such as Conexant and RFMicro Devices. These modules typi-cally contain a single GaAs-HBTRFIC that includes a single-endedclass-AB PA. Recently developed PAmodules also include a silicon controlIC that provides the base-bias refer-ence voltage and can be commandedto adjust the output-transistor basebias to optimize efficiency whilemaintaining acceptably low amplifierdistortion. over the full ranges oftemperature and output power. A typ-ical module (Figure 17) produces 28dBm (631 mW) at full output with aPAE of 35 to 50 percent.

Cellular-Telephone BaseStation Transmitter

The Spectrian MCPA 3060 cellu-lar base-station transmitter for 1840-1870 MHz CDMA systems providesup to 60-W output while transmittinga signal that may include as many 9modulated carriers. IMD is mini-mized by linearizing a class-AB mainamplifier with both adaptive predis-tortion and adaptive feed-forwardcancellation. The adaptive control

Figure 15 3-kW high efficiency PA for 13.56 ISM-band operation.(Courtesy Advanced Energy)

Figure 16 Output section of a 50-kW AM broadcast transmitter.(Courtesy Harris)

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system adjusts operation as neededto compensate for changes due totemperature, time, and output power.The required adjustments arederived from continuous measure-ments of the system response to aspread-spectrum pilot test signal.The amplifier consumes a maximumof 810 W from a 27-V supply.

S-Band Hybrid Power ModuleA thick-film-hybrid power-ampli-

fier module made by UltraRF (nowCree Microwave) for 1805 to 1880MHz DCS and 1930-1960 MHz PCSis shown in Figure 18. It uses four140-mm LDMOS FETs operatingfrom a 26-V drain supply. The indi-vidual PAs have 11-dB power gainand are quadrature-combined to pro-duce a 100-W PEP output. The aver-age output power is 40 W for EDGEand 7 W for CDMA, with an ACPR of57 dBc for EDGE and 45 dBc forCDMA. The construction is basedupon 0.02-in. thick film with silvermetalization.

GaAs MMIC Power AmplifierA MMIC PA for use from 8 to 14

GHz is shown in Figure 19. Thisamplifier is fabricated with GaAsHBTs and intended for used inphased-array radar. It produces a 3-W output with a PAE of approxi-mately 40 percent [59].

References19. H. L. Krauss, C. W. Bostian, and

F. H. Raab, Solid State RadioEngineering, New York: Wiley, 1980.

20. R. Gupta and D. J. Allstot, Fullymonolithic CMOS RF power amplifiers:Recent advances, IEEE Communi-cations Mag., vol. 37, no. 4, pp. 94-98,April 1999.

21. F. H. Raab and D. J. Rupp, HFpower amplifier operates in both classB and class D, Proc. RF Expo West 93,San Jose, CA, pp. 114-124, March 17-19,1993.

22. P. Asbeck, J. Mink, T. Itoh, and G.Haddad, Device and circuit approaches

for next-generation wireless communi-cations, Microwave J., vol. 42, no. 2, pp.22-42, Feb. 1999.

23. N. O. Sokal and A. D. Sokal,Class Ea new class of high efficiencytuned single-ended switching poweramplifiers, IEEE J. Solid-StateCircuits, vol. SC-10, no. 3, pp. 168-176,June 1975.

24. F. H. Raab, Effects of circuitvariations on the class E tuned poweramplifier, IEEE J. Solid State Circuits,vol. SC-13, no. 2, pp. 239-247, April1978.

25. F. H. Raab, Effects of VSWRupon the class-E RF-power amplifier,Proc. RF Expo East 88, Philadelphia,PA, pp. 299-309, Oct. 25-27, 1988.

26. J. F. Davis and D. B. Rutledge, Alow-cost class-E power amplifier withsine-wave drive, Int. Microwave Symp.Digest, vol. 2, pp. 1113-1116, Baltimore,MD, June 7-11, 1998.

27. T. B. Mader and Z. B. Popovic,The transmission-line high-efficiencyclass-E amplifier, IEEE Microwaveand Guided Wave Letters, vol. 5, no. 9,pp. 290-292, Sept. 1995.

28. D. C. Hamill, Class DE invert-ers and rectifiers for DC-DC conver-sion, PESC96 Record, vol. 1, pp. 854-860, June 1996.

29. F. H. Raab, Maximum efficiencyand output of class-F power amplifiers,IEEE Trans. Microwave Theory Tech.,vol. 47, no. 6, pp. 1162-1166, June 2001.

30. F. H. Raab, Class-E, -C, and -Fpower amplifiers based upon a finitenumber of harmonics, IEEE Trans.Microwave Theory Tech., vol. 47, no. 8,pp. 1462-1468, Aug. 2001.

31. S. C. Cripps, RF PowerAmplifiers for Wireless Communi-cation, Norwood, MA: Artech, 1999.

32. A load pull system with har-monic tuning, Microwave J., pp. 128-132, March 1986.

33. B. Hughes, A. Ferrero, and A.Cognata, Accurate on-wafer power andharmonic measurements of microwaveamplifiers and devices, IEEE Int.Microwave Symp. Digest, Albuquerque,NM, pp. 1019-1022, June 1-5, 1992.

34. V. J. Tyler, A new high-efficiency

Figure 17 Internal view of a dual-band (GSM/DCS) PA module forcellular telephone handsets.(Courtesy RF Micro Devices)

Figure 18 Thick-film hybrid S-bandPA module. (Courtesy UltraRF)

Figure 19 MMIC PA for X- and K-bands.

Acronyms Used in Part 2BJT Bipolar Junction

TransistorDSP Digital Signal

ProcessorIC Integrated CircuitIMD Intermodulation

DistortionMOSFET Metal Oxide Silicon

FET

36 High Frequency Electronics

High Frequency Design

RF POWER AMPLIFIERS

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Author InformationThe authors of this series of arti-

cles are: Frederick H. Raab (leadauthor), Green Mountain RadioResearch, e-mail: f.raab@ieee.org;Peter Asbeck, University ofCalifornia at San Diego; SteveCripps, Hywave Associates; Peter B.Kenington, Andrew Corporation;Zoya B. Popovic, University ofColorado; Nick Pothecary,Consultant; John F. Sevic, CaliforniaEastern Laboratories; and Nathan O.Sokal, Design Automation. Readersdesiring more information shouldcontact the lead author.

Notes1. In Part 1 of this series (May

2003 issue), the references containedin Table 1 were not numbered cor-rectly. The archived version has beencorrected and may be downloadedfrom: www.highfrequencyelectronics.com click on Archives, selectMay 2003 Vol. 2 No. 3 then clickon the article title.

2. This series has been extendedto five parts, to be published in succe-sive issues through January 2004.