Micro-coaxial Micro-fabricated Feeds for Phased Array Antennas

Zoya Popovic, University of Colorado, Boulder, Colorado, zoya@colorado.edu

Abstract— This paper presents an overview of recent research at the University of Colorado, Boulder, in the area of passive and active antenna arrays. Specifically, micro-coaxial array feed components, fabricated in the Nuvotronics PolyStrataTM process will be discussed as a means to miniaturized phased array. Performance of broadband Wilkinson dividers covering 2-22GHz (11:1 bandwidth) implemented in micro-coaxial lines for active transmit arrays is presented along with assembly methods for standard surface-mount resistors within the micro-coaxial environment. Integration of active devices with micro-coaxial components for 20W transmitters covering the 4-18GHz band is shown, and challenges associated with high power densities and interconnect parasitic are addressed. Ka-band (38 GHz) hybrids resulting in a 4x4 Buttler matrix with patch-like antennas integrated in PolyStrataTMare presented. The high-performance micro-coaxial environment is highlighted with its advantages and potential limitations. Frequency-scanned array designs for W and G-band operation are compared for PolyStrataTM micro-coaxial and waveguide-fed arrays. Finally, several other related approaches, including spatially-fed beam-forming arrays, are overviewed. 1. INTRODUCTION

Recent advances in fabrication techniques for micro-electromechanical systems (MEMS), including surface and bulk micromachining, provide tools that make it possible to produce miniaturized TEM rectangular coaxial lines with heights from 50�m to 800�m [1-4]. The electrical properties of these lines that make them attractive for millimeter-wave and broadband applications include: � Ultra-low dispersion, since the TEM mode is dominant

up to 400-500GHz, depending on the size of the coaxial line and its characteristic impedance [3];

� Extremely low coupling between neighboring lines, lower than -70dB through Ka-band for two lines that share an outer conductor wall [5];

� Wide range of characteristic impedances [6], from about 6-140� for a given process, a 23:1 ratio;

� The coaxial lines have an air dielectric, which minimizes the loss since the capacitance per unit length is minimized. Loss lower than 0.1dB/cm at 38GHz has been measured in copper air-filled lines [7];

� High level of hybrid integration of actives and passives is possible, thus enabling new types of components;

� Micro-coaxial components can be accurately modeled using commercial full-wave finite-element methods such as [8] due to its shielded nature.

A large number of millimeter-wave narrowband components such as resonators, hybrids, and antennae were first demonstrated with this technology, e.g. hybrids [4], [9-12], couplers [13], resonators [14], antennae [15], and filters [16]. More recently, broadband components covering 11:1 bandwidths from 1-22GHz and with power handling in the 20-W range were developed [7,17].

Figure 1 – Photograph of several �-coaxial hybrids

covering frequencies from 6 to 60 GHz and fabricated in the PolystrataTM copper process with 5 layers.

Figure 1 shows an example of densely integrated branch-line, rat-race and coupled-line couplers operating from 6 to 60GHz and fabricated on the same Silicon substrate. Notice that the 6-GHz coupler is only 4.5mm on the side. This small footprint is made possible by meandering the coaxial line since neighboring lines or sections of lines practically do not couple. This example makes it immediately obvious that the microcoaxial technology holds promise for dense, miniature, low loss passive circuits, including antenna array feed networks. Antennas can be fabricated in the same process [15], or hybridly integrated with the feed network. In the next section, the copper-based PolyStataTM fabrication process is discussed briefly. This is followed by a discussion of basic components of feed networks. Section 3 presents broadband dividers/combiners and their integration with power amplifiers for a 4-18GHz transmit array. Section 4 addresses 90-degree couplers and their integration in a Buttler matrix with quasi-patch antennas. Section 5 discusses scaling to W and G-band scanning arrays, while Section 6 presents some related work and future directions of this research.

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2. FABRICATION AND HYBRID INTEGRATION The microwave and millimeter-wave components discussed in this paper are made of sections of �-coaxial cable fabricated using a sequential metal-layer deposition process referred to as the PolyStrataTM technology and developed by Nuvotronics LLC (formerly Rohm and Haas Electronic Materials) [18]. Micro-coaxial components have also been demonstrated in the EFABTM process by Microfabrica for operation in the Ka- and V-bands [4], [16]. Although small, rigid devices with excellent performance have been reported, this process uses nickel, with twice the loss of copper for the same transmission-line cross section. This paper focuses on devices implemented in the copper PolyStrataTM process, shown schematically in Figure 2.

Figure 2 – The PolyStrataTM fabrication process consists of sequential copper deposition in layers that are 50 to

100�m thick. The center conductor is supported by periodic dielectric straps, which take less than 0.5% of the volume.

The process involves sequential deposition of copper layers and photoresist on a silicon wafer. Copper layer thicknesses range from 10 �m to 100 �m, with gap-to-height and width-to-height aspect ratios of 1:1.2 and 1:1.5, respectively. The inner conductor is supported by 100-�m long dielectric straps with periodicity of 700 �m. After the desired layers are deposited, the photoresist filling all space unoccupied by copper and dielectric straps is rinsed away (“released”) through 200 �m × 200 �m release holes. With this method, up to 15 independent layers (strata) can be fabricated in a single run. The components can be left on the substrate, or released with a chemical process and mounted on a different host substrate. Substrates other than Silicon are also possible, e.g. alumina, quartz, etc. Vias (as in microstrip) and bridges (as in CPW) are not required in micro-coaxial circuits because the ground conductor can be brought to any level and shaped to any geometry. One of the limitations of the low-temperature PolyStrataTM process thus far has been the inability to monolithically integrate resistors and capacitors. However, the micro-coaxial line environment lends itself naturally to hybrid integration, since both center and outer conductors can be designed to accommodate series and/or shunt surface

mount devices (SMDs). The assembly structures, referred to as “sockets” are required for integrating lumped components such as blocking capacitors, resistors in Wilkinson dividers, or discrete actives and MMICs. Shunt and series sockets were manufactured in a 5-layer process and SMT components were mounted at the Univ. of Colorado using silver epoxy (Epoxy Technology H20E). Nuvotronics has developed a process for depositing solder which allows for large-scale surface mount integration. Figure 3 shows photographs of hybridly integrated surface mount components in 0402 standard packages [19]. The measured and simulated return loss for a 82-pF blocking capacitor was below -15 dB while |S21| was above -0.4 dB from 4-18 GHz. Active sockets can be designed for flip-chip or chip-and- wire assembly. A 1-10 GHz GaN HEMT distributed amplifier with a 600 �m gate periphery on a 100 �m thick SiC substrate was flip-chip assembled with 50 � micro-coaxial lines [20]. The comparison of the measured small-signal performance of the chip both on wafer and in PolyStrataTM confirms that there is no degradation in performance when the active device is hybridly integrated into the monolithic micro-coaxial environment. This indicates that the parasitic are negligible for a properly designed socket. In both cases, |s11|<-40dB and |s21|<-0.05dB were shown for the socket alone across the 4-18 GHz band, Figure 4.

(a)

(b)

Figure 3 – Standard size surface mount passive components, as well as active MMICs can be hybridly integrated in the process. In the photographs, a 0402

blocking capacitor is mounted in series in a 50-� coaxial line (a), and several lumped components are integrated into

a micro-coaxial bias Tee network with a series 3-D inductor.

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Figure 4 – Measured small-signal performance of the GaN on SiC MMIC with standard on-wafer probes compared to

the performance of the same MMIC flip-chipped in the PolyStrataTM micro-coaxial environment from 1-10GHz.

3. BROADBAND DIVIDERS/COMBINERS

In this section, broadband Wilkinson micro-coaxial power dividers are presented as good candidates for phased array feed networks. Dividers with various input and output characteristic impedances can be designed in the PolyStrataTM process, and several prototypes are demonstrated in the 2-22GHz range and detailed in [17]. The integration of active components within the feed network is useful for phased array antennas, and a compact 20-W broadband (4-18GHz) power amplifier in which GaN MMICs is integrated with a Wilkinson divider is briefly described in [21] and the passive network of the PA is shown in the photograph in Figure 4. The layout includes bias networks for the MMICs.

Figure 5 – PolyStrataTM Wilkinson combiner, impedance

matching networks and bias lines for integration with a GaN MMIC PA with 20W output from 4-18GHz. The MMIC is

fabricated by BAE Systems and is not shown here. The Wilkinson dividers are designed to operate from 2-22GHz. For a single-section two-way divider the bandwidth

is around 3:1 for a VSWR of 2:1. Several methods have been developed to increase the bandwidth of such dividers. The greatest experimentally demonstrated bandwidth with the Wilkinson divider to the best of the authors’ knowledge is 4:1 (3–12 GHz) using a strip-line configuration with four sections [22]. The layout of the divider designed for 50-ohm ports is shown in Figure 6. It consists of 5 sections with one 50-� and four 200-� 0402 resistors mounted in passive series sockets. The lines are meandered to reduce the component footprint and 5 strata are used in this design.

Figure 6 –PolyStrataTM Wilkinson 17-mm long combiner designed for 2-22GHz operation with 5 sections and 0402 hybridly integrated resistors with passive series sockets.

Some relevant features of these dividers for phased array corporate feed networks include:

- Excellent match and low insertion loss, as shows in the measured results in Figure 7. Note that the insertion loss is lower for narrower-band designs as well as for micro-coaxial lines with larger dimensions (more layers) [17];

- Excellent amplitude and phase balance over a broad bandwidth, as shown in the measured data of Fig. 8;

- High isolation, Figure 7, allowing element-to-element isolation in feed networks;

- Small current flow in resistors for 20% load mismatch at the output ports and 1W input power was shown with transient simulations. For example, for impedances of 40 - 60� at ports 2 and 3, a maximal current of 4.7mA flows through the first resistor, so standard 3-W 0402 surface mount resistors can be used with a large margin [17];

- For 11 strata of copper, the possible characteristic impedance range of the lines is increased, allowing a reduced footprint design (smaller number of sections) and lower insertion loss, e.g. 0.7dB at 18GHz [17];

- The dividers can be designed with different output port impedances for a better match to antennas or amplifiers. For examples, in [17], a design of a divider with 12-� input and output ports for better match to GaN power amplifiers is shown.

The components are measured using CPW probes coupled to the micro-coaxial line via specially designed launch structures [9] and micro-coaxial calibration standards [6].

MMIC active socket

Wilkinson divider

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Figure 7 – Measured and simulated return loss, isolation and insertion loss of the Wilkinson PolyStrataTM Wilkinson

5-strata 5-section combiner.

Figure 8 – Measured amplitude and phase balance of the Wilkinson PolyStrataTM Wilkinson combiner.

6. KA-BAND MICRO-COAXIAL BUTTLER

MATRIX Various branch-line couplers have been demonstrated in PolyStrataTM, including stacked Ka-band versions centered at 36 GHz [11]. Two pairs of stacked couplers are shown in Figure 10, with a compact Buttler matrix network in mind. Measured results show the return loss and isolation is better than 20 dB, the output amplitudes are 3.3 dB±0.1 dB, and the output phase differences are 90�±1.0� from 34 to 38GHz. The cross section of the 50-� line is 100�m tall by 139�m wide for the inner conductor and 300�m by 300�m for the outer conductor spacing. The 35-� line is 100�m tall by 265�m wide for the inner conductor and 300 �m tall by 400�m wide for the outer conductor spacing. The wall

thicknesses are 100�m on the sides, and 20�m and 50�m on the bottom and top, respectively.

Figure 9 – Layout of a stacked pair of branch-line couplers in a 10-strata process. The couplers are designed to operate at 36GHz and are stacked on top of each other in order to enable a Buttler matrix with vertical coaxial interconnects

between layers.

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Figure 10 – (a) Simulated and measured S-parameter data

for a length-compensated branch line coupler. (b) The difference in output magnitude and phase.

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P8P4

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Since the couplers need to be stacked in a Butler matrix, the isolation of stacked couplers is relevant, and the simulated results are shown in Figure 11.

Figure 11 – Full-wave simulation of a stacked pair of hybrids. The parameters of interest in this simulation are the coupling between the lower and upper hybrid with Port 1 of the lower hybrid excited (per Figure 9).

Figure 12 – Photograph of a 30-GHz 4-element linear corporate-fed array of air-supported patch antennas fabricated in the PolyStrataTM process. The feed is

meandered between patches on a lower layer (not visible).

Several types of air-supported patch antennas with micro-coaxial feeds have been reported [15], [23-26]. Figure 12 shows a linear array fed by a corporate feed network which is meandered between the antenna elements, thus considerably reducing the overall size of the array. The hybrids and antenna elements can be combined into a monolithic tile phased array as shown in Figure 13. The 36-GHz phased array is fabricated entirely in the PolyStrataTM process. There are three distinct regions (from left to right) in this array: the radiating region with the 4x4 antenna array, a 16x16 Butler matrix with phase compensation network feeding into the antennas, and the region reserved for MMIC integration and active control. This phased array is built from 10 copper strata with thicknesses varying between 50 and 100�m. Dielectric supports are included on two different vertical locations and are placed underneath the central conductor. Low line losses of 0.16dB/cm, coupler output misbalances below 0.1dB and 2º, excellent antenna match, and overall yield in excess of 99% are demonstrated at the operating frequency. This was the first demonstration of a fully integrated antenna array - beamformer built with micron-scale rectangular coaxial lines [27]. 6. W / G-BAND FREQUENCY SCANNING ARRAYS Micro-coaxial components implemented in the PolyStrata process have been demonstrated through V-band, and a WR10 waveguide to coaxial line transition has been shown at W-band [28]. Scaling a phased array to W band and above has not yet been demonstrated in this technology, but if possible, would enable miniaturized, light-weight arrays. This section shows a design of a G-band frequency-scanned arrays [29] currently being fabricated. Two architectures, are introduced to provide a wider scanning angle than conventional structures: (1) an array of slots fed by a micro-coaxial non-dispersive line where the delay is increased by meandering; and (2) a slotted waveguide array designed for the dominant mode at G-band [30]. The dimensions of the PolyStrata G-band waveguide feed lines is 640�m x 1.3mm, and can be fabricated in the same process as micro-coaxial lines.

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Figure 13 – Photograph of a monolithic 16-element antenna array fed with a Butler matrix integrated in PolyStrataTM

micro-coaxial lines. The array is on the left, followed by the feed network which includes sockets for active components.

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The loss of micro-coaxial line and dominant-mode rectangular waveguide can be calculated at G-band assuming both perfectly smooth copper, as well as copper fabricated with an RMS roughness of 0.13�m, as is the largest roughness in the PolyStrataTM process. The predicted losses ag G and W bands are shown Table 1.

Table 1 – Calculated losses at W and G-bands.

The vertical wall roughness is expected to be mostly present, since the horizontal layers are extremely smooth. It can be seen that at G-band the waveguide is expected to have about two times lower loss than micro-coaxial lines. At W-band, the loss of waveguide is even lower comparatively, but the waveguide is too tall to be practical for PolyStrata fabrication especially over a large area as would be needed for an array. Figure 14(a) shows a micro-coaxial fed slot array with 10 double slots. Since the micro-coaxial lines have very low dispersion, for this application their dispersion needs to be increased. Meandering the transmission lines between double radiating slots and introducing double slots increases the scan angle for a given length. The width of the inner conductor under the slots controls the input and output match as well as coupling to each antenna. Using double radiating slots with a 10% difference in length increases the bandwidth. The slot array is matched to the input and output feed coaxial lines characteristic impedance in order to obtain a non-resonant array. Figure 14(b) shows the simulated beam steering of this array in the E-plane, using Ansoft HFSS. A perfect conductor boundary condition was imposed on the sides of the array, while waveports were defined on the input and output. The scanning is 30 degrees from 128–155 GHz, and the gain is within 1.5 dB for all scan angles. However, the pattern degrades at higher frequencies, due to non-uniform excitation of the slots. Because coaxial lines are purely TEM, they can be more easily scaled to different (lower) frequencies, by scaling the slot length and distance. Moreover a circuit model can be used for design, because TEM lines behave as ideal transmission lines over a wide range of frequencies. Figure 15(a) shows a waveguide slotted array with 10 slots. In this case longitudinal slots are used. In order to achieve the greatest possible scan angle and the lowest gain variation over the scanning bandwidth, several parameters, such as the length and width of the slots, the distance between each slot, and the position of the slots from the

center of the waveguide were optimized. A non-standard waveguide was designed to operate near the fundamental mode cut-off frequency. In this way high dispersion over the operating bandwidth is obtained, resulting in a wider scanning angle than conventional slotted waveguide arrays. Figure 15(b) shows the simulated beam steering pattern. Both arrays have 30� scanning over a 26GHz bandwidth at G-band. Dispersive slotted waveguides have lower loss, however the feeding mechanism for the microcoaxial lines is easier to implement and measurements can be performed more directly. Waveguides at lower frequencies, such as W-band, would be more difficult to implement in the sequential copper deposition process due to their increased dimensions. With the PolyStrata process, longer arrays can be fabricated, e.g. 40 mm long lines are demonstrated in e.g. [17], corresponding to a 25 element array at G-band. This approach can also be extended to 2-D arrays [31].

(a)

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Figure 14 – (a) Layout of a frequency scanned 10-element linear slot array fed with a micro-coaxial line. (b) Simulated

radiation pattern as a function of frequency.

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(b) Figure 15 – (a) Layout of a frequency scanned 10-element

linear –band waveguide array. (b) Simulated radiation pattern as a function of frequency.

7. DISCUSSION AND CONCLUSIONS A number of high-quality microwave and millimeter wave components have been demonstrated in the PolyStrataTM process, with excellent agreement between simulations and measured data, allowing for reliable computer-aided design with first-pass success. Figure 16 shows a brief summary of the design flow on the example of a branch-line coupler and quasi-patch antenna, which would be the main components for a Butler-matrix fed phased array. Applications for this technology can benefit from the unique electrical properties of low loss, high isolation, three-dimensional components, large range of characteristic impedances and new types of packaging techniques. For example, the high isolation implies that usual constraints of spacing between transmit and receive portions of a radar or communication system are no longer limited by the transmission line medium. Another advantage of high isolation is that the coaxial lines can be fabricated in multiple interconnected levels with low-loss sharp 90-degree turns, allowing higher density of components as

compared to traditional 2D structures such as microstrip. Such a simple component as a high isolation coaxial crossover, highly challenging to do at millimeter wavelengths, enables new devices such as miniature M x N switch matrices to be realized. In active antenna arrays, e.g. electronically-scanned arrays (ESAs), the unit cell needs to be smaller than a half of a free-space wavelength squared. As the frequency increases towards W-band, the wavelength and thus allowed real-estate per element scales down faster than the size of active elements (MMICs). PolyStrata™ multilayer technology allows design in the third dimension allowing more functionality for a given footprint. An example of a T/R module stack-up for an active array is shown in Figure 17 with a 16 element antenna and all core electrical functionalities included within the profile of 0.64 square centimeters [32]. Such cells could be could be used alone for tiny high-bandwidth data links or combined into large arrays and be digitally steered. Although this paper discusses only two types of phased array, namely Butler matrix fed arrays and frequency-scanned arrays, the micro-coaxial technology can be applied with great advantages to other beam-steering array architectures such as constrained lenses, e.g. described in [34] at Ka-band and in [35] at X-band. A diagram of a planar constrained lens is shown in Figure 18, along with a patch antenna and stripline implementation at Ka-band and measured radiation patterns. The steering and beam quality is limited by the delay line, which in stripline or microstrip has a limited length (true time delay) as well as dispersion. Micro-coaxial true-time delay can enable phase-shifterless beam steering lenses with large scan angles and low levels of aberrations. In summary, this paper presents an overview of work related to phased array and done over the past five years at the University of Colorado in the PolyStrataTM process and in close collaboration with Nuvotronics LLC. Experimental results from 2 to 40GHz are shown, along with designs for ongoing W-band and G-band components for frequency-steered antennas, showing great promise of this technology for reducing the size and weight of future phased arrays.

ACKNOWLEDGMENTS I would like to thank my colleague and collaborator, Prof. Dejan Filipovic at the University of Colorado for the many years of the very hard and extremely rewarding joint work in the area covered by this paper. Without him, there would be nothing to write about. I also thank the team at Nuvotronics LLC, especially Dr. Kenneth Vanhille (formerly at the University of Colorado), Dr. Jean-Marc Rollin and David Sherrer. Many thanks to Drs. Negar Ehsan (now at NASA), Stefania Romisch and Leonardo Ranzani (now at NIST), and Mr. Evan Cullens at the University of Colorado, as well as the team at BAE Systems that led the 3D-MERFS and DMT DARPA programs which enabled this work to get started.

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REFERENCES [1] D. S. Filipovic, Z. Popovic, K. Vanhille, M. Lukic, S. Rondineau, M. Buck, G. Potvin, D. Fontaine, C. Nichols, D. Sherrer, S. Zhou, W. Houck, D. Fleming, E. Daniel, W. Wilkins, V. Sokolov, and J. Evans, “Modeling, design, fabrication, and performance of rectangular �-coaxial lines and components,” in 2006 Proc. IEEE MTT-S Int. Microwave Symp. Dig., San Francisco, California, June 2006, pp. 1393–1396. [2] J. Reid, E. Marsh, R. Webster “Micro-machined Rectangular-Coaxial Transmission Lines,”. IEEE Trans. Microwave Theory Tech., Aug 2006 ,pp. 3433-3442. [3] M. Lukic, S. Rondineau, Z. Popovic, and D. S. Filipovic, “Modeling of realistic rectangular micro-coaxial lines,” Microwave Theory and Techniques, IEEE Transactions on, vol. 54, no. 5, pp. 2068–2076, 2006. [4] R. T. Chen, E. R. Brown, and R. S. Singh, “A compact 30-GHz low loss balanced hybrid coupler fabricated using micromachined integrated coax,” in 2004 Proc. IEEE Radio and Wireless Conf., Atlanta, GA, Sept. 2004, pp. 227–230. [5] Y. Saito and D. Filipovic, “Analysis and design of monolithic rectangular coaxial lines for minimum

coupling,” IEEE Trans. Microwave Theory and Techniques, vol. 55, no. 12, pp. 2521–2530, 2007. [6] N. Ehsan, E. Cullen, K. Vanhille, D. Frey, S. Rondineau, R. Actis, S. Jessup, R. L. Jr., A. Immorlica, D. Nair, D. Filipovi´c, and Z. Popovic, “Micro-coaxial lines for active hybrid-monolithic circuits,” Microwave Symposium, 2009. IEEE/MTT-S International, 2009. [7] D. Filipovic, M. Lukic, Y. Lee, and D. Fontaine, “Monolithic rectangular coaxial lines and resonators with embedded dielectric support,” IEEE Microwave and Wireless Comp. Lett., vol. 18, no. 11, pp. 740–742, 2008. [8] A. Corporation, “High frequency structure simulator.” [Online]. Available: http://ansoft.com/hfss/ [9] K. Vanhille, “Design and characterization of microfabricated three-dimensional millimeter-wave components,” Ph.D. dissertation, Univ. of Colorado, 2007.

Figure 16 – Summary of design flow for a coupler and antenna components of a phased array with a Butler matrix feed. From left to right: CAD models from HFSS follow fabrication design guidelines such as layer aspect ratio and

release hole density. The 3-D designs are converted directly to a layout used for mask fabrication and sequential deposition resulting in Polystrata components. Calibration standards are fabricated on the same wafer and allow

measurements that are in excellent agreement with the design simulations allowing first-pass success.

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Figure 17 – Sketch of multi-layer phased array module with 16 elements, showing the layers of Polystrata interconnects and the packaging possibilities. The dense integration is possible because of the high isolation of micro-

coaxial lines, as well as high-quality vertical interconnects.

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Figure 18 – (a) Constrained lens array architecture. Two antenna arrays are printed on two sides of a substrate and connected with true time delay lines allowing the lens to perform a spatial Fourier transform in a 3-D version of a

Rotman lens. The green and red arrows indicate incident waves for a receive lens and corresponding focal points on the focal surface where the beams are formed without a need for phase shifters. (b) Implementation of a dual simultaneous 28-GHz lens for formation flying satellites [34]. (c) Feed side patch antenna array on TMM6 and layout of delay line

network implemented in strip line and slot-coupled to the feed side and radiating side patches of the lens. The stripline network is limited due to coupling and dispersion. (d) Measured radiation patterns of the lens at 28GHz.

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