Mar. 2005 CWC/AETHERWIRE/CEA-LETI/STM/MERL doc.: IEEE 802. 15-05-xxxx-00-004a Slide 1 Project: IEEE P802.15 Working Group for Wireless Personal Area Networks

Mar. 2005

CWC/AETHERWIRE/CEA-LETI/STM/MERL

doc.: IEEE 802. 15-05-xxxx-00-004a

Slide 1

Project: IEEE P802.15 Working Group for Wireless Personal Area Networks (WPANs)Project: IEEE P802.15 Working Group for Wireless Personal Area Networks (WPANs)

Submission Title: STM_CEA-LETI_CWC_AETHERWIRE_MITSUBISHI 15.4aCFP responseDate Submitted: January 4th, 2005Source: Ian Oppermann (1), Mark Jamtgaard (2), Laurent Ouvry (3), Philippe Rouzet (4), Andreas F. Molisch(5),

Philip Orlik(5), Zafer Sahinoglu (5), Rick Roberts (6), Vern Brethour (7), Adrian Jennings (7) Companies:

(1) CWC-University of Oulu, Tutkijantie 2 E, 90570 Oulu, FINLAND (2) Æther Wire & Location, Inc., 520 E. Weddell Drive, Suite 5, Sunnyvale, CA 94089, USA (3) CEA-LETI, 17 rue des Martyrs 38054, Grenoble Cedex, FRANCE (4) STMicroelectronics, CH-1228, Geneva, Plan-les-Ouates, SWITZERLAND (5) MERL, 201 Broadway, Boston, USA(6) Harris, (7)Time Domain, Hansville, Alabama

Voice: (1) +358 407 076 344, (2) 408 400 0785 (3) +33 4 38 78 93 88, (4) +41 22 929 58 66 (5) +1 617 621 7500E-Mail: (1) [email protected], (2) [email protected](3) [email protected], (4) [email protected], (5)

{molisch, porlik, zafer}@merl.com, [email protected], {vern.brethour, adrian.jennings}@timedomain.com

Abstract: UWB proposal for 802.15.4a alt-PHY

Purpose: Proposal based on UWB impulse radio for the IEEE 802.15.4a CFP

Notice: This document has been prepared to assist the IEEE P802.15. It is offered as a basis for discussion and is not binding on the contributing individual(s) or organization(s). The material in this document is subject to change in form and content after further study. The contributor(s) reserve(s) the right to add, amend or withdraw material contained herein.

Release: The contributors acknowledge and accept that this contribution becomes the property of IEEE and may be made publicly available by P802.15

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Slide 2

List of Authors• CWC – Ian Oppermann, Alberto Rabbachin (1)

• AetherWire – Mark Jamtgaard, Patrick Houghton (2)

• CEA-LETI – Laurent Ouvry, Samuel Dubouloz, Sébastien de Rivaz, Benoit Denis, Michael Pelissier, Manuel Pezzin et al. (3)

• STMicroelectronics – Gian Mario Maggio, Chiara Cattaneo, Philippe Rouzet & al. (4)

• MERL – Andreas F. Molisch, Philip Orlik, Zafer Sahinoglu (5)

• Harris – Rick Roberts (6)

• Time Domain – Vern Brethour, Adrian Jennings (7)

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Slide 3

Outline• Introduction• Background• Transmitted Signal• Receiver Architectures• Bandwidth Usage• Optional Aspects• System performances• Link budget• Framing, throughput• Power Saving • Ranging and Delay Estimation • Feasibility• Conclusions

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Slide 4

Proposal Main Features

1. Impulse-radio based (pulse-shape independent)

2. Support for different receiver architectures (coherent/non-coherent)

3. Flexible modulation format

4. Support for multiple rates

5. Enables accurate ranging/positioning

6. Support for multiple SOP

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Slide 5

Motivation for (2-4):

• Supports homogenous and heterogeneous network architectures

• Different classes of nodes, with different reliability requirements (and cost) must inter-work

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Slide 6

UWB Technology

• Impulse-Radio (IR) based:– Very short pulses Reduced ISI– Robustness against fading– Episodic transmission (for LDR) allowing

long sleep-mode periods and energy saving

• Low-complexity implementation

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Slide 7

Modulation Features

• Simple, scalable modulation format

• Flexibility for system designer

• Modulation compatible with multiple coherent/non-coherent receiver schemes

• Time hopping (TH) to achieve multiple access

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Slide 8

Commonalities with Other Proposals

• Many commonalities exist between the CWC/Aetherwire/LETI/STM/MERL proposal and other proposals, including FT :

– UWB technology– Modulation features– Bandwidth usage– Ranging approach

• Discussions are under way for future collaborations and merging

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Slide 9


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Slide 10

Definitions

Coherent RX: The phase of the received carrier waveform is known, and utilized for demodulation

Differentially-coherent RX: The carrier phase of the previous signaling interval is used as phase reference for demodulation

Non-coherent RX: The phase information (e.g. pulse polarity) is unknown at the receiver -operates as an energy collector

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Slide 11

Pros (+) and cons (-) of RX architectures:

Coherent• + : Sensitivity• + : Use of polarity to carry data• + : Optimal processing gain achievable• - : Complexity of channel estimation and RAKE receiver• - : Longer acquisition time

Differential (or using Transmitted Reference)• + : Gives a reference for faster channel estimation (coherent approach)• + : No channel estimation (non-coherent approach)• - : Asymptotic loss of 3dB for transmitted reference (not for DPSK)

Non-coherent• + : Low complexity• + : Acquisition speed• - : Sensitivity, robustness to SOP and interferers

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Slide 12

Time Hopping Impulse Radio (TH-IR) - Principle

Ts

Tc

Tf

+1

-1

• Each symbol represented by sequence of very short pulses (see also Win & Scholtz 2000)

• Each user uses different sequence (Multiple access capability)

•Bandwidth mostly determined by pulse shape

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Slide 13

Transmitted Reference (TR)

• TR schemes simplify the channel estimation process• Reference waveform available for synchronisation• Potentially more robust (than non-coherent) under

SOP operation• Supports both coherent/differentially-coherent

demodulation

• Implementation challenges:– Analogue: Implementing delay value, – delay mismatch, jitter

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Slide 14

Transmitted Reference

Ts

Tc

Tf

Td

+1

-1

•First pulse serves as template for estimating channel distortions

•Second pulse carries information

•Drawback: Waste of 3dB energy on reference pulses

reference

data

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Slide 15


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Slide 16

Design Parameters (1) • Motivation:

– Flexible waveform – Simple– Compatible with multiple coherent/non-coherent receiver schemes

• Large Bandwidth• (+) Higher transmit power• (+) improved time resolution• (-) Increased design complexity• (-) Less stringent requirements on out of band interference filtering Signal BW of 500 MHz - 2 GHz in Upper bands

Signal BW of 700 MHz in 0 to 960 MHz Lower band (low band)

• Long Pulse Repetition Period• (+) more energy per pulse (easier to detect single pulse)• (+) Lower inter-pulse interference due to channel delay spread• (-) Higher peak voltage requirements at transmitter• (-) Longer acquisition time Frame duration between 40ns (first realization) and 125ns (second realizations). Higher values

for the frame duration have been mentioned. Further discussions are required to fix the values

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Slide 17

• Simple modulation schemes:• BPPM combined with Transmitted Reference

• Channelization :• Coherent schemes: Use of TH codes and polarity codes• Non-coherent schemes: Use of TH codes (polarity codes for spectrum smoothing

only)

• Long TH code length• (+) higher processing gain, robustness to SOP operation• (-) Lower bit-rate• (-) Longer acquisition time, shorter frame size (synch. phase) TH code length 8 or 16

TH code : binary position (delay of 0 or τΔ ), bi-phaseFor first realization, higher-order TH with shorter chip duration (multiples of

2ns) can be used. This is under discussion

Design Parameters (2)

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Slide 18

Transmission• Basic idea: use modulation scheme that allows coherent,

differentially coherent, and incoherent reception

• Combine BPPM with more sophisticated TR scheme– Non-coherent receiver sees BPPM with pulse stream per bit– More sophisticated receiver sees BPPM (1 bit) plus bits carried in

more sophisticated modulation scheme (e.g. extended TR)

• Advantages: – Coherent, differential and non-coherent receiver may coexist– reference can be used for synch and threshold estimation

• Concept can be generalized to N-ary TR system

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Slide 19

Waveform Design

• Coexistence of coherent and non-coherent architectures

• Combine BPPM with BPSK

• Divide each symbol into two 125 ns BPPM slots (250 ns symbol)

• In either slot transmit a signal that can be received with a variety of receivers: differentially coherent or coherent receivers.

• Non-coherent receivers just look for energy in the early or late slots to decode the bit.

• Other receivers understand the fine structure of the signal.

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Slide 20

Waveform Design

• Two possible realizations:

– The whole symbol (consisting of N_f frames) is BPPM-modulated.

– Have a 2-ary time hopping code, so that each frame has BPPM according to TH code

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Slide 21

First Realization

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Slide 22

Second Realization

Ts

« 11 »

« 01 »2-PPM + TR baseM = 2(with two bits/symbol)One bit/symbol also Possible !!!

« 10 »

« 00 »

(coherent decoding possible) 2-PPM + 16 chips 2-ary TH code

This is a time-hopping that can be exploited by non-coherent receiverTime hopping code is (2,2) code of length 8 or 16 Effectively 28 or 216 codes to select for channelization for non-coherent scheme

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Slide 23

Mitigation of peak voltage through multi pulses

Tf=PPI

Tf=PPI

IS « EQUIVALENT » TO

ppV = peak-to-peak voltage

ppV/2

M = 4

M = 1

Tf=PPI

ppV/sqrt(2)

M = 2

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Slide 24

Coexistence of Different Receiver Architectures

• Want waveform that allows TR reception without penalizing coherent reception

• That is achieved by special encoding and waveform shaping within each frame. Does not affect the co-existence of coherent/non-coherent receivers

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Slide 25

Basic Properties

• Use of Doublets with memory from previous bit. (Encoding of reference pulse with previous bit) – Agreed on 20ns separation between pulses

– Extensible to higher order TR for either reducing the penalty in transmitting the reference pulse or increasing the bit rate?

– Also allows the use of multi-DOUBLET

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Slide 26

Ts

Differential Encoding of Bits

+1 -1 +1 -1 +1 -1

-1 -1 +1 +1 -1 -1

0 0 1 1 0 0 1

b0 b4b3b2b1 b5b-1

Tx Bits

Reference Polarity

Note: This slide is meant to describe the encoding of data on the reference pulse and data pulse in the basic modulation format. For simplicity we have omitted the multipulse/multiframes per symbol structure.

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Slide 27

Total Modulation Scheme (First Realization)THE KEY SLIDE OF THE PROPOSAL: this is the modulation format that allows

Coherent, differentially coherent, and non-coherent demodulation at once

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Slide 28

Higher-order modulation

TH PatternTH Code 1,1 1,1 0,1 0,0 1,0 0,1Data 1,1 1,1 1,1 1,1 1,1 0,0

Pulse Shift, polarity invert

τΔ + τdelay

Enhanced Mode 1

D« 1 1 »

« 1 0 »

D D τdelay +τΔ

τΔ τdelay τΔ + τdelayτdelayτΔτΔ + τdelay τΔ + τdelay

Basic Mode (as seen by non-coherent)

« 1 »τdelay +τΔD D D

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Slide 29

Comments on Transmitted Signal

• Frame period for solution 2 is Tframe = (Np * D) + τΔ + τdelay

– Τdelay is some allowance for channel delay spread

• Frame period could be dynamic modified dependant on – the estimated channel delay spread or – ability of receiver to cope with delay spread

• Symbol period is length of the TH code x Tframe

– Upper Band Nominally 250 ns x 16 = 4 µs– Lower Band Nominally 500 ns x 8 = 4 µs

• Realistic Receiver structures exist for multi-pulse TR schemes (see back-up slides)

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Slide 30


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Slide 31

Proposed RAKE -- Coherent Receiver

Rake ReceiverFinger Np

Demultiplexer Rake ReceiverFinger 2

Rake ReceiverFinger 1

Summer

Channel Estimation

Convolutional Decoder Data

Sink

Sequence Detector

• Addition of Sequence Detector – Proposed modulation may be viewed as having memory of length 2• Main component of Rake finger: pulse generator• A/D converter: 3-bit, operating at symbol rate• No adjustable delay elements required

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Slide 32

Proposed Transmitted Reference Receiver – Differentially Coherent

Td

0

MatchedFilter

Convolutional Decoder

•Addition of Matched Filter prior to delay and correlate operations improves output signal to noise ratio and reduces noise-noise cross terms

SNR of decision statistic

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Slide 33

Band MatchedBand Matched

LNALNALNALNA BPFBPF

TrackingTrackingThresholThreshol

ds ds settingsetting

TrackingTrackingThresholThreshol

ds ds settingsetting

DumpDumpLatchLatchDumpDumpLatchLatch

RA

ZR

AZ

DUMPDUMP

ControlledControlledIntegratorIntegrator

ADCADC

BPPM Demodulation BPPM Demodulation branchbranch

RAZRAZ

Differentially-Coherent/Non-Coherent Receiver Architecture Basic Mode and Enhanced Mode 1

x2 x2r(t)

Recyle this branch for Enhanced Data Rate Modes

DelayDelayDelayDelay

TR Demodulation TR Demodulation

branchbranch

DumpDumpLatchLatchDumpDumpLatchLatchADCADC

ControlledControlledIntegratorIntegrator BPPM BPPM

Synch Synch TriggerTrigger

TRTR Energy AnalyzerBlock index for

acquisition reference

Leading-edge refinement search

Range info

Ranging branch

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Slide 34


LNALNALNALNA BPFBPF

De-spreading TH Codes

r(t) TH Sequence Matched

Filter

Bit Demodulation Bit Demodulation


LNALNALNALNA BPFBPFr(t) TH

Sequence Matched Filter

Bit Demodulation Bit Demodulation

Case I - Coherent TH despreading

ADCADC

ADCADCb(t)

soft info

Case II – Non-coherent / differential TH despreading

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Slide 35


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Slide 36

Bandwidth Usage (1/4)

• Flexible use of (multi-)bands

• Signal bandwidth may be 500 MHz to 2 GHz

• Bandwidth may change depending on application and regulatory environment

• Use of TH and/or polarity randomization for spectral smoothing

• Different bandwidth use options being considered

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Slide 37

Bandwidth Usage – 2GHz option (2/4)

0.96 3.1 5.1 6.0 8.0 8.1 10.1 GHz

ISM Band

Upper Band 1

Upper Band 2

Upper Band 3Lower band

ISM Band

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Slide 38

Bandwidth Usage -500 MHz Option (3/4)

0.96 3.1 5.1 6.0 8.0 8.1 10.1 GHz

ISM Band

Lower band

ISM Band

Upper Bands 1 - 4

Upper Bands 5 - 12

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Slide 39

Bandwidth Usage –Variable Option (4/4)

0.96 3.1 5.1 6.0 8.0 8.1 10.1 GHz

ISM Band

Lower band

ISM Band

Upper Bands 1 - 4

Upper Bands 5 - 12

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Slide 40


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Slide 41

Spectral Shaping & Interference Suppression (Optional)

• Basis pulse: use simple pulse shape gaussian, raised cosine, chaotic, etc.

• Drawbacks:– Possible loss of power compared to FCC-

allowed power– Strong radiation at 2.45 and 5.2 GHz

frequency (Hz)

Monocycle, 5th derivative of gaussian pulse

Power spectral density of the monocycle

10

log

10|P

(f)|

2 d

B

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Slide 42

Linear Pulse Combination • Solution: linear combination of delayed, weighted pulses

– Adaptive determination of weight and delay

– Number of pulses and delay range restricted

– Can adjust to interferers at different distances

(required nulldepth) and frequencies

• Weight/delay adaptation in two-step procedure• Initialization as solution to quadratic optimization problem

(closed-form)• Refinement by back-propagating neural network

• Matched filter at receiver good spectrum helps coexistence and interference suppression

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Slide 43

Spectral Shaping & Polarity Scrambling

Td = 10 ns

Td = 20 ns

0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2

x 1010

-220

-210

-200

-190

-180

-170

-160

-150

-140

-130

-120

W/ Polarity ScramblingW/O Polarity Scrambling

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Slide 44

Adaptive Frame Duration

• Advantage of large number of pulses per symbol:

– Smaller peak-to-average ratio

– Increased possible number of SOPs

• Disadvantage:

– Increased inter-frame interference

– In TR: also increased interference from reference pulse to data pulse

• Solution: adaptive frame duration

– Feed back delay spread and interference to transmitter

– Depending on those parameters, TX chooses frame duration

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Slide 45


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Slide 46

PER in 15.4a Channel Model Non-Coherent (Energy Collection) BPPM

Framing format:

PreamblePreamble(32 bits)(32 bits)

SFDSFD(8 bits)(8 bits)

LENLEN(8 bits)(8 bits)

MHR+MSDUMHR+MSDU(240 bits)(240 bits)

CRCCRC(16 bits)(16 bits)

• Simulations over 1000 channel responses • BW = 2GHz – Integration Time = 80ns• Implementation loss + Noise figure margin : 11 dB• Max range is determined from:

Required Eb/N0, Implementation margin Path loss characteristics

X1 (CM8) X2 (CM1) X3 (CM5) X4 (CM9)

Required Eb/N0 19.5 dB 20 dB 21 dB 21.5 dB

Max Range (I) 10.78 m 84.61 m 86.72 m 58.67 m

Max Range (II) 7.33 m 53.25 m 54.15 m 34.72 m

16 17 18 19 20 21 22

10-2

10-1

100

2PPM - PER vs. Eb / N

0 - 15.4a Channel Models

Eb / N

0 [dB]

Pac

ket

Err

or R

ate

X1-CM8: industrial NLOSX2-CM1: Residential LOSX3-CM5: Outdoor LOSX4-CM9: Agricultural Areas

Case I: 250 kbps – PRP 250 ns with 16 pre-integrations = 4 µs

Case II: 250 kbps – PRP 500 ns with 8 post-integrations

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Slide 47

PER/BER in 15.4a Channel Model DBPSK (RAKE)

10 11 12 13 14 15 16 17 18 19 2010

-3

10-2

10-1

100

PE

R

X1X2X3X4

Eb/N0 (dB)

DBPSK – PER vs. Eb/N0 – 15.4a Channel Models

X1 (CM8) X2 (CM1) X3 (CM5) X4 (CM9)

Required Eb/N0 18 dB 17.5 dB 18.5 dB 18.5 dB

Max Range (I) 12.66 m 116.70 m 120.27 m 90.84 m

Max Range (II) 9.18 m 79.34 m 81.23 m 58.67 m

Implementation loss and Noise figure margin : 11 dB

Theoretical BER Curves – Integration Time = 50 ns

0 5 10 1510

-5

10-4

10-3

10-2

10-1

Eb/N0 (dB)

BE

R

BER vs Eb/N0 for binary modulations - 100 equivalent uncorrelated samples

ideal MRC BPAM solutionDifferentially coherent BPAM

Case I: 250 kbps – PRP 250 ns with 16 pre-integration = 4 µs

Case II: 250 kbps – PRP 500 ns with 8 post-integrations

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Slide 48


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Slide 49

Link Budget: Non-Coherent (Energy Collection) BPPM

ParameterMandatory Value

(PRP = 4 µs) Optional Value

(PRP = 500ns - 8 integrations)

Peak Payload bit rate (Rb) 250 kb/s 250 kb/s

Average Tx Power Gain (PT) -10.64 dBm -10.64 dBm

Tx antenna gain (GT) 0 dBi 0 dBi

f’c: (geometric frequency) 3.873 GHz 3.873 GHz

Path Loss @ 1m: L1 = 20log10(4..f’c / c) 44.20 dB 44.20 dB

Path Loss @ d m: L2 = 20log10(d) 29.54 dB @ d = 30 m 29.54 dB @ d = 30 m

Rx Antenna Gain (GR) 0 dBi 0 dBi

Rx Power (PR = PT + GT + GR – L1 – L2) -84.38 dBm -84.38 dBm

Average noise power per bit: N = -174 + 10log10(Rb) -120.02 dBm -120.02 dBm

Rx noise figure (NF) 7 dB 7 dB

Average noise power per bit (PN = N + NF) -113.02 dBm -113.02 dBm

Minimum Eb/N0 (S) 14 dB 17.6 dB

Implementation Loss (I) 5 dB 5 dB

Link Margin (M = PR - PN – S – I) 9.64 dB 6.04 dB

Proposed Min. Rx Sensitivity Level -94.02 dBm -90.42 dBm

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Slide 50

Link Budget: DBPSK (RAKE)

ParameterMandatory Value

(PRP = 4 µs)Optional Value

(PRP = 500ns - 8 integrations)

Peak Payload bit rate (Rb) 250 kb/s 250 kb/s

Average Tx Power Gain (PT) -10.64 dBm -10.64 dBm

Tx antenna gain (GT) 0 dBi 0 dBi

f’c: (geometric frequency) 3.873 GHz 3.873 GHz

Path Loss @ 1m: L1 = 20log10(4..f’c / c) 44.20 dB 44.20 dB

Path Loss @ d m: L2 = 20log10(d) 29.54 dB @ d = 30 m 29.54 dB @ d = 30 m

Rx Antenna Gain (GR) 0 dBi 0 dBi

Rx Power (PR = PT + GT + GR – L1 – L2) -84.38 dBm -84.38 dBm

Average noise power per bit: N = -174 + 10log10(Rb) -120.02 dBm -120.02 dBm

Rx noise figure (NF) 7 dB 7 dB

Average noise power per bit (PN = N + NF) -113.02 dBm -113.02 dBm

Minimum Eb/N0 (S) 13 dB 16 dB

Implementation Loss (I) 5 dB 5 dB

Link Margin (M = PR - PN – S – I) 10.64 dB 7.64 dB

Proposed Min. Rx Sensitivity Level -95.02 dBm -92.02 dBm

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Slide 52

Framing – 802.15.4 Compatible

CAP CFP IPBP

Beacon Interval

BP : Beacon Period

CAP : Contention Access Period

CFP : Contention Free Period

IP : Inactive Period (optional)

Superframe Duration

Beacon slot CAP slot CFP slot

0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15

Preamble SFD PSDU = MPDUPHY layer

PPDU (PHY Protocol Data Unit)

PSDU (PHY Service Data Unit)SHR

Octets 324 1

Frame length

PHR

1

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Slide 53

Throughput

Preamble SFD PSDU = MPDUPHY layer


PSDU (PHY Service Data Unit)SHR

Bytes 324 1

Frame length

PHR

1

Preamble SFD PSDUPHY layer


PSDUSHR

Bytes 54 1

Frame length

PHR

1

Data Frame (32 octet PSDU) ACK Frame (5 octet PSDU)

Tdata TackT_ACK

• Numerical example (high-band)

• Preamble + SFD + PHR = 6 octets

• Tdata = 1.216 ms

• T_ACK = 50 s (turn around time requested by IEEE 802.15.4 is 192s)

• Tack = 0.352 ms

• IFS = 100μs

Throughput = 32 octets/1.718 ms = 149 kb/s Average data-rate at receiver PHY-SAP 250 kb/s (Basic Mode)

IFS

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Slide 55

Saving Power• Power Saving techniques achieved by combining advantages offered at 3 levels:

– Technology (best if CMOS)

– Architecture (flexible schemes provided by the TH+pulse modulation)

– System level (framing, protocol usage)

• Selected techniques used in one existing realization (see proof of concept slides) – Low-duty cycle Episodic transmission/reception

• Scheduled wake-up

• 80s RTOS tick

– Ad-hoc networking using multi-hop• Special rapid acquisition codes / algorithm

• Matchmaking further reduces acquisition time

– Multi-stage time-of-day clock• Synchronous counter / current mode logic for highest speed stages

• Ripple counter / static CMOS for lowest speed stages

– Compute-intensive correlation done in hardware

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Slide 57

Ranging

• Motivation :– Benefit from high time resolution (thanks to signal bandwidth):

• Theoretically: 2GHz provides less than 20cm resolution

• Practically: Impairments, low cost/complexity devices should support ~50cm accuracy with simple detection strategies (better with high resolution techniques)

• Approach :– Use Two Way Ranging between 2 devices with no network

constraint (preferred); no need for time synchronization among nodes

– Use One Way Ranging and TDOA under some network constraints (if supported)

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Slide 58

TOA Delay Estimation - Non-Coherent

• Use bank of integrators to determine coarse synchronisation “uncertainty” region

– Symbol synchronisation “uncertainty” region given by coarse synchronisation ( e.g., 4ns-20ns)

• A refinement search is applied onto the uncertainty region by either– further dividing it into narrower non overlapping regions for non-coherent refinement

(e.g., 1ns –> 4ns) or

– Coherent search with a template correlation

Energy Analyzer

Integrator outputs

Leading Edge Search Refinement

2|)(| RBkTtr)(tr NEEE ...21

},..1,0{ Nk

Performed within the selected uncertainty region

Detects the coarse “uncertainty region”

LEk̂

Range info

nsTns RB 204

TRB: the length of uncertainty region

RBf TTN /

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Slide 59

TOA Delay Estimation - Non-Coherent (cont’d)• The algorithm selects the maximum value integration window index and

then it searches backward to find the first integration value which crosses an adaptively set threshold.

• If there are no values crossing the threshold, the peak position is used for the TOA estimation.

Actual TOA

MES based TOA Estimate

Strongest Path, energy block

Contains leading edge

Threshold based TOA Estimate

Threshold

Searchback window

MES-SB based TOA Estimate

MES: Maximum Energy SearchTC: Threshold comparisonSB: Search Back

0 1 2 N

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Slide 60

ENERGY Spread in CM1

• PDF of TOA estimation errors are illustrated for MES at various EbN0

– CM1, integration interval 4ns, Tf=200ns (results will be updated for Tf=240ns)

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Slide 61

Ranging Simulation Settings

Notations and Terms Definition Value in Simulations

Tf Pulse repetition interval, frame 200ns

Nb Number of blocks within a Tf 50

Nc Number of refinement intervals within a Tf 400

TH{} Time hopping sequence in chips {h1, ...., h5}

POL{} Polarity codes {p1, ..., p5}

N1 Number of frames in the 1st-step 50

N2 Number of frames in the refinement 30

BW Bandwidth 2GHz

C Number of correlators (refinement stage) 10

Note: Results are to be provided when Tf is set to 240ns.

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Slide 62

Ranging Results• IEEE 802.15.4a CM1-Residential LOS

Round Trip ranging error (with no drift compensation)

– ~16cm (0.088ms), no clock drift

– ~17.1cm (1ppm)

– ~20.1cm (4ppm)

– ~26cm (10ppm)

– ~56cm (40ppm)

True Distance (m) One-way ranging error (confidence level)

25m 8cm (97%)

30m 8cm (~90%)

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Slide 63

Two Way Ranging (TWR)

B

BAAAOFAOF f

fTfTT

12

1~ Reply

Main Limitations / Impact of Clock Drift on Perceived Time

Range estimation is affected by :

• Relative clock drift between A and B

• Prescribed response delay

• Clock accuracy in A and B

•Channel response (weak direct path)

Is the frequency offset relative to the nominal ideal frequency 0f0. f

Simple immediate TWR made unusable with reasonnable crystal accuracies. Solution is :

• Performing fine drift estimation/compensation

• Benefiting from cooperative transactions (estimated clock ratios …)

250 kbps, 38 bytes PPDU 500 kbps, 9 bytes PPDU

f/f \ Treply (max error)

1408 s 1226 s 336 s 154 s

4 ppm 1.69 m 1.47 m 0.40 m 0.18 m

25 ppm 10.56 m 9.19 m 2.52 m 1.15 m

40 ppm 16.9 m 14.7 m 4.0 m 1.8 mExample using Imm-ACK SIFS of 15.4 and 15.3

of respectively 192us and 10 usand PPDU size of respectively 38 and 9 bytes

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Slide 64


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Slide 65

Antenna Feasibility Capacitive Dipole and Various Bowtie Antennas

Bowtie antenna

55 mm

40 mm

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Slide 66

“Proof-of-Concept” (1) Non-coherent Transceiver

5 Mbps BPPM 350 ps pulse train

with long scrambling code

UWBANTENNA

DLLCM

Oscilator Generator

x16528 MHz

Flagrst8....

Flagrst1

RX

Controlline

Energycollection

DIGITALCONTROL

LOGIC

DLLPG

-DIGITALLOGIC UWB

PG

TX

Fref33 MHz

BASEBANDDSP

(decision logic)

DETECTIONBIT

DECISION

(digital)

2

( )

LNAVGA

GAIN SELECTION

A/DConverter

SWITCH

/2

Non-coherent, Energy Collection Receiver

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Slide 67

“Proof-of-Concept” (2) Non-coherent Transceiver

UWB Transmitter400 μm x 400 μm

0.35 μm CMOS

UWB TransceiverTest architecture <10 mm2

0.35 μm SiGe Bi-CMOS

UWB-IR BPPM Non-Coherent Transceiver Implementation

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Slide 68

“Proof-of-Concept” (3): Transmitter - Lower Band

UWB Transmitter chip for generating impulse doublets

Del

ayD

elay

Buf

fers

Buf

fers

N-CN-Channel DriversN-Channel Drivers

P-Channel DriversP-Channel Drivers

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Slide 69

“Proof-of-Concept” (4): Receiver - Lower Band

Coherent UWB Receiver withmultiple time integrating correlators

32

Tim

e-I

nte

gra

ting

32

Tim

e-I

nte

gra

ting

Co

rre

lato

rsC

orr

ela

tors

PL

L L

oo

p F

ilte

rP

LL

Lo

op

Filt

er

VG

C A

mp

VG

C A

mp

CodeCodeSequenceSequenceGeneratorsGenerators

LF RTCLF RTC

DA

Cs

DA

Cs

““ Ra

ils”

for

test

ing

R

ails

” fo

r te

stin

g

an

alo

g c

ircu

itsa

na

log

cir

cuitsDA

Cs

DA

Cs

Hig

h-F

requ

ency

Hig

h-F

requ

ency

Rea

l Tim

e C

lock

Rea

l Tim

e C

lock

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Slide 70

“Proof-of-Concept” (5) High Speed Coherent Circuit Elements

3-5 GHz LNAChip and layout

20 GHz digitizer for UWB

20 GHz DLL for UWB

RF front end chips in CMOS 0.13m, 1.2V

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Slide 71


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Slide 72

Conclusions• Proposal based upon UWB impulse radio

– High time resolution suitable for precise ranging using TOA– Modulation:

• Pulse-shape independent• Robust under SOP operation• Facilitates synchronization/tracking• Supports multiple coherent/non-coherent RX architectures

• System tradeoffs– Modulation optimized for several aspects (requirements, performances, flexibility,

technology)– Trade-off complexity/performance RX

• Flexible implementation of the receiver– Coherent, differential, non-coherent (energy collection)– Analogue, digital

• Fits with multiple technologies– Easy implementation in CMOS– Very low power solution (technology, architecture, system level)

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Slide 73

Backup Slides

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Slide 74

BER Performance in AWGN Channel

5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20

10-5

10-4

10-3

10-2

10-1

Eb/N0

BE

R

MRC Solution (coherent)Differential SolutionEnergy Collection solution in OOK

Transmitted Reference (one pulse)

-3 dB : the “reference” is not in the same PRP !

PER = 1% with 32 bytes PSDU acceptable BER 4x10-5 with no channel coding

Nerrorbiterrorpacketp 11

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Slide 75

BER Performance in AWGN Channel

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Slide 76

Antenna Practicality• Bandwidth: 3 GHz-10 GHz

• Form factor

• Omni-directionaly

x

z

7 mm

ground plane Ø 80 mm

antenna hat Ø 24 mm

-26

-24

-22

-20

-18

-16

-14

-12

-10

-8

-6

-4

-2

0

2 3 4 5 6 7 8 9 10 11 12

|S11| (d

B)

frequency (GHz)

M.S.measured

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Slide 77

Mobile (xm,ym)

Anchor 1 (xA1,yA1)

Anchor 2 (xA2,yA2)

Anchor 3 (xA3,yA3)

Positioning from TDOA

3 anchors with known positions (at least) are

required to find a 2D-position from a couple of TDOAs

2222

31

222232

1133

2233

MAMAMAMA

MAMAMAMA

yyxxyyxxd

yyxxyyxxd

3132

~,

~dd

Measurements Estimated Position

MM yx ~,~Specific Positioning Algorithms

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Slide 78

TR BPPM Scheme Comparison

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Slide 79

Assumptions and Notes

• Results are theoretical calculations• Assumes ideal ”impulse” UWB pulses in AWGN

channel• Different TR-BBP options are considered with different

number of pulses per pulse train

• Multipath fading simulations can be performed to back up theory

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Slide 80

Pulse repetition structures TR BPPM with doublets (Scheme 1)

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Slide 81

Pulse repetition structuresTR BPPM single reference (Scheme 2)

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Slide 82

Pulse repetition structuresAuto Correlation BPPM with doublets (Scheme 3)

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Slide 83

Pulse repetition structuresAuto Correlation BPPM single reference (Scheme 4)

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Slide 84

Pulse repetition structuresAuto Correlation BPPM alternate (Scheme 5)

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Slide 85

Parameters

• PPI slot - slot inside each TH chip containing a burst of pulses including reference pulses (ref. slides from Laurent / CEA)

• Np represents the number of pulses in each PPI slot

• The energy E per PPI slot is kept constant• The pulse energy Ep = E/Np

• TW represent the time-bandwidth product

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Slide 86

Ep/N0 degradation versus number of pulses per pulse train

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Slide 87

Ep/N0 degradation versus Time/Bandwidth product

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Slide 88

Ep/N0 degradation versus number of pulses per pulse train

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Slide 89

Conclusions

• Scheme 5 - “AC Alternate” performs better then all the other pulse repetition structures.

• AC generally performs better than TR• “AC alternate” and “AC with doublets” have the advantage of

requiring only a single delay line.

• Scheme 5 - “AC Alternate”, was proposed at Monterey meeting in January.

• Criticism was given based on ”accumulated noise” in noise-cross-noise-cross-noise... Products”.

• Seems to outperform other schemes with simple analysis• Also more readily implementable since fixed delay line can be

used.

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Slide 90

Channel / Delay Estimation Coherent Approach

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Slide 91

Channel / Delay Estimation Coherent Approach

• Swept delay correlator

• Principle: estimating only one channel sample per symbol.

Similar concept as STDCC channel sounder of Cox (1973).

• Sampler, AD converter operating at SYMBOL rate (1.2 Msamples/s)

• Requires longer training sequence

• Two-step procedure for estimating coefficients:

– With lower accuracy: estimate at which taps energy is significant

– With higher accuracy: determine tap weights

• “Silence periods”: for estimation of interference

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Slide 92

Optimal Energy-Threshold Analysis (CM-1)

• Optimal normalized threshold (normalized with respect to the difference between the maximum and minimum energy blocks) changes with Eb/N0 and block size.

• Smaller thresholds are required in general at high Eb/N0, while larger thresholds at lower SNR values

MAEMean Absolute Error in detecting leading energy block with simple threshold crossing (1000 channel realizations)

Eb/N0 : {8 --- 26dB}

Documents

Mar. 2005 CWC/AETHERWIRE/CEA-LETI/STM/MERL doc.: IEEE 802. 15-05-xxxx-00-004a Slide 1 Project: IEEE P802.15 Working Group for Wireless Personal Area Networks