iW1692Low-Power Off-line Digital PWM Controller

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10 Featuresbull Primary-side feedback eliminates opto-isolators and

simplifies design

bull Multi-mode operation for highest overall efficiency

bull Built-incabledropcompensation

bull Very tight output voltage regulation

bull Noexternalloopcompensationcomponentsrequired

bull ComplieswithCECEPAIECnoloadpowerconsumptionand average efficiency regulations

bull Built-inoutputconstant-currentcontrolwithprimary-sidefeedback

bull Lowstart-upcurrent(10microAtypical)

bull Built-insoftstart

bull Built-inshortcircuitprotection

bull AC line underovervoltage and output overvoltage protection

bull 40 kHz PWM switching frequency

bull PFM operation at light load

bull Built-inISENSEpinshortprotection

bull Space-saving SOT-23 package

Figure 201 iW1692 Typical Application Circuit

20 DescriptionThe iW1692 is a high performance ACDC power supply controller which uses digital control technology to build peak current mode PWM flyback power supplies The device provides high efficiency along with a number of key built-in protection features while minimizing the external component count and bill of material cost The iW1692 removes the need for secondary feedback circuitry while achieving excellent line and load regulation It also eliminates the need for loop compensation components while maintaining stability over all operating conditions Pulse-by-pulse waveform analysis allowsforaloopresponsethatismuchfasterthantraditionalsolutions resulting in improved dynamic load response The built-in power limit function enables optimized transformerdesign in universal off-line applications and allows for a wide input voltage range

The low start-up power and PFM operation at light load ensure that the iW1692 is ideal for applications targeting the newest regulatory standards for standby power

30 Applicationsbull Low power ACDC adapterchargers for cell phones

PDAs digital still cameras

bull Standby supplies for televisions DVDs set-top boxes andotherconsumerelectronics

L

NVOUT

RTN

+ ++

+

U1iW1692

VCC

ISENSE

VIN

OUTPUT

GND

VSENSE

4

2

6

5

3

1

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Pin Name Type Pin Description1 VSENSE

Input Voltage sense input from the auxiliary winding

2 GND Ground Ground connection

3 OUTPUT Output Gate drive output for the external power MOSFET switch

4 VCC Input Supply voltage

5 ISENSE Input Primary current sense Used for cycle-by-cycle peak current control and limit

6 VIN Input Senses average rectified input voltage

Parameter Symbol Value UnitsDC supply voltage range (pin 4 ICC = 20mA max) VCC

-03 to 18 V

DC supply current at VCCpin ICC20 mA

Output (pin 3) -03 to 18 V

VSENSEinput(pin1) -03 to 40 V

ISENSEinput(pin5) -03 to 40 V

VIN input (pin 6) -03 to 18 V

Power dissipation at TA le 25degC PD400 mW

Maximumjunctiontemperature TJ(MAX)125 degC

Storage temperature TSTGndash65 to 150 degC

Lead temperature during IR reflow for le 15 seconds TLEAD260 degC

Thermal resistance junction-to-ambient θJA240 degCW

ESD rating per JEDEC JESD22-A114 (HBM) 2000 V

Latch-Up test per JEDEC 78 plusmn100 mA

50 Absolute Maximum RatingsAbsolute maximum ratings are the parametric values or ranges which can cause permanent damage if exceeded For maximum safe operating conditions refer to Electrical Characteristics in Section 60

40 Pinout Description

iW1692

VSENSE

GND

OUTPUT

VIN

ISENSE

VCC

1

2

3

6

5

4

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VCC = 12 V -40degC le TA le 85degC unless otherwise specified (Note 1)

Parameter Symbol Test Conditions Min Typ Max UnitVIN SECTION (Pin 6)

Start-up voltage threshold VINST TA= 25degC positive edge 366 407 448 mV

Start-upcurrent IINST VCC=10V 10 15 microA

Shutdown low voltage threshold VUVDC TA= 25degC negative edge 216 240 264 mV

Shutdown high voltage threshold VOVDC TA= 25degC positive edge 1737 1930 2123 V

Inputimpedance ZIN Afterstart-up 20 kW

VSENSE SECTION (Pin 1)

Input leakage current IBVS VSENSE = 2 V 1 μA

Nominal voltage threshold VSENSE(NOM) TA=25degC negative edge 1507 1538 1569 V

Output OVP threshold VSENSE(MAX) TA=25degC negative edge 1667 1700 1734 V

OUTPUT SECTION (Pin 3)

Output low level ON-resistance RDS(ON)LO ISINK=5mA 45 100 W

Output high level ON-resistance RDS(ON)HI ISOURCE=5mA 65 100 W

Rise time (Note 2) tRTA = 25degC CL = 330 pF10 to 90 40 75 ns

Fall time (Note 2) tFTA = 25degC CL = 330 pF90 to 10 40 75 ns

Output switching frequency (Note 3) fS ILOADgt15ofmaximum 36 40 44 kHz

VCC SECTION (Pin 4)

Maximum operating voltage VCC(MAX) 16 V

Start-upthreshold VCC(ST) VCC rising 110 120 132 V

Undervoltage lockout threshold VCC(UVL) VCC falling 55 60 66 V

Operating current ICCQCL = 330 pF VSENSE = 15 V 25 35 mA

ISENSE SECTION (Pin 5)

Peaklimitthreshold VPEAK 1000 mV

CClimitthreshold VCC-TH 900 mV

60 Electrical Characteristics

NotesNote 1 Adjust VCC above the start-up threshold before setting at 12 VNote 2 These parameters are not 100 tested guaranteed by design and characterizationNote 3 Frequency variation includes plusmn12 dithering for EMI suppression

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70 Typical Performance Characteristics124

123

122

121

120

V CC

Sta

rt-u

p Th

resh

old

(V)

Ambient Temperature (degC)-50 -25 0 25 50 75 100

Figure 703 Start-Up Threshold vs Temperature

2015

2010

2005

2000

1995

Inte

rnal

Ref

eren

ce V

olta

ge (V

)

Ambient Temperature (degC)-50 -25 0 25 50 75 100

VCC = 12 V

Figure 704 Internal Reference vs Temperature

28

26

24

22

20

18

16

V CC

Sup

ply

Cur

rent

(mA

)

Load Capacitance (pF)0 200 400 600 800 1000

VCC = 12 VTA = 25deg

Figure 701 Supply Current vs Load Capacitance

44

42

40

38

36

Switc

hing

Fre

quen

cy (k

Hz)

Ambient Temperature (degC)-50 -25 0 25 50 75 100

VCC = 12 V

Figure 702 Switching Frequency vs Temperature

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80 Functional Block Diagram

90 Theory of Operation

ndash

+

VIN

GND

PORPOR

VCC

2

VSENSE

VFB

VVMS

VIPK

VIN_A

OUTPUT

128 mV

1538 V

ndash

+CMP

ndash

+CMP

ISENSE5

10 V

02 V ~ 09 V

02 V ~ 20 V

IPEAK

ADC

VOCP

ndash

Start-up

DAC

ndash

+

1

6 4

DigitalLogic

Control

GateDriver

3

Figure 801 iW1692 Functional Block Diagram

The iW1692 is a digital controller which uses a new proprietary primary-side control technology to eliminate the opto-isolated feedback and secondary regulation circuits required in traditional designs This results in a low-cost solution for low power ACDC adapters The core PWM processor uses fixed-frequency Discontinuous Conduction Mode (DCM) operation at heavy load and switches to variable frequency operation at light loads to maximize efficiency Furthermore iWattrsquos digital control technology enables fast dynamic response tight output regulation and full featured circuit protection with primary-side control

Referring to the block diagram in Figure 801 the digital logic control generates the switching on-time and off-time information based on the line voltage and the output voltage feedback signal The system loop is internally compensated inside the digital logic control and no external analog components are required for loop compensation The iW1692 uses an advanced digital control algorithm to reduce system design time and improve reliability

Furthermoreaccuratesecondaryconstant-currentoperationis achieved without the need for any secondary-side sense and control circuits

The iW1692 uses PWM mode control at higher output power levels and switches to PFM mode at light load to minimize power dissipation Additional built-in protection features include overvoltage protection (OVP) output short circuit protection (SCP) AC low line brown out over current protection single pin fault protection and ISENSEfaultdetection

iWattrsquos digital control scheme is specifically designed to address the challenges and trade-offs of power conversion design This innovative technology is ideal for balancing new regulatory requirements for green mode operation with more practical design considerations such as lowest possible cost smallest size and high performance output control

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VCC

VCC(ST)

OFF

ENABLE

Start-upSequencing

200micros

VIN VIN Impedance = 20k

VINSWON

Figure 921 Start-up Sequencing Diagram

93 Understanding Primary FeedbackFigure 931 illustrates a simplified flyback converter When the switch Q1 conducts during tONthecurrentigisdirectlydrawn from rectified sinusoid vg The energy Eg isstoredin the primary windingThe rectifying diode D1 is reverse biasedandtheloadcurrentIOissuppliedbythesecondarycapacitorCO When Q1 turns off D1 conducts and the stored energy Eg(t) is delivered to the output

+

vin(t)

TS(t)

IO

VO

NP NS

NAUX

D1

Q1

VAUX

COvg(t)

ig(t)+

ndash

iin(t) id(t)

Figure 931 Simplified Flyback Converter

In order to regulate the output voltage within a tight specification the information about the output voltage and load current needs to be accurately sensed In the DCM flyback converter this information can be read via the auxiliary winding or the primary magnetizing inductance (LM) During the Q1on-timetheloadcurrentissuppliedfromthe output filter capacitor CO The voltage across the primary winding is vg(t) assuming the voltage dropped across Q1iszero The current in Q1rampsuplinearlyatarateof

( ) ( )g g

M

di t v tdt L

=

(91)

91 Pin DetailPin 1 ndash VSENSE

Sense signal input from auxiliary winding This provides the secondary voltage feedback used for output regulation

Pin 2 ndash GND

Analog digital and power ground

Pin 3 ndash OUTPUT

Gate drive signal for the external power MOSFET switch

Pin 4 ndash VCC

Power supply for the controller during normal operation The controller starts up when VCC reaches 12 V (typical) and shuts-downwhentheVCC voltage is below 6 V (typical) A 100 nF decoupling capacitor should be connected between theVCC pin and GND

Pin 5 ndash ISENSE

Primary current sense

Pin 6 ndash VIN

Sense signal input representing the instantaneous rectified line voltage VIN is used for line regulation The internal impedanace is 20 kW and the scale factor is 00043 It also provides input undervoltage and overvoltage protection This pin also provides the supply current to the IC during start-up

92 Start-upPrior to start-up the VIN pin charges up the VCC capacitorthrough the diode between VINandVCC When VCC isfullycharged to a voltage higher than VCC(UVL) threshold then theVIN_SW turns on and the analog-to-digital converter begins to sense the input voltage The iW1692 commences soft-start function as soon as the voltage on VIN pin is above VINST

The iW1692 incorporates an internal soft-start function The soft-start time is set at 30 ms Once the VIN pin voltage has reached its turn-on threshold the iW1692 starts switching but limits the on-time to a percentage of the maximum on-time During the first 1 ms the on-time is limited to 25 During the next 1 ms the on-time is limited to 50 and during the last 1 ms the on-time is limited to 75

IfatanytimetheVCC voltage drops below VCC(UVL) threshold then all the digital logic is fully reset At this time the VIN_SWswitches off so that the VCC capacitor can be charged up again

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Attheendofon-timethecurrenthasrampedupto

( ) ( )( ) g ON

gM

v t t ti t

Ltimes

=

(92)

This current represents a stored energy of

2( )2M

g gLE i t= times

(93)

WhenQ1turnsoffattOig(t)inLM forces a reversal of polarities on all windings Ignoring the communication-time caused by the leakage inductance LK at the instant of turn-off tO theprimarycurrenttransferstothesecondaryatanamplitudeof

( ) ( )Pd g

S

Ni t i tN

= times

(94)

Assuming the secondary winding is master the auxiliary winding is slave

VAUX 0V

VAUX = -VG xNAUX

NP

See equation 95

Figure 932 Auxiliary Voltage Waveforms

The auxiliary voltage is given by

( )AUXAUX O

S

NV V VN

= + ∆

(95)

and reflects the output voltage as shown in Figure 932

The voltage at the load differs from the secondary voltage by a diode drop and IR losses The diode drop is a function of current as are IR losses Thus if the secondary voltage is alwaysreadataconstantsecondarycurrentthedifferencebetween the output voltage and the secondary voltage is a fixed ΔV Furthermore if the voltage can be read when the secondarycurrentissmallΔV is small

The real-time waveform analyzer in the iW1692 reads this information cycle by cycle and then generates a feedback voltage VFB The VFB signal precisely represents the output voltage and is used to regulate the output voltage

94 Understanding CC and CV modeThe constant current mode (CC mode) is useful in battery charging applications During this mode of operation the iW1692 will regulate the output current at a constant maximum level regardless of the output voltage drop while avoiding continuous conduction mode

To achieve this regulation the iW1692 senses the load current indirectly through the primary current The primary currentisdetectedbytheISENSE pin through a resistor from the MOSFET source to ground (RSS) This resistor value is given by

( )( )2

CSS

OUTMAX

N KR

Itimes

=times

(96)

N is the ratio of primary turns to secondary turns of thetransformerandKC is 0264 V

95 Constant Voltage OperationAfter soft-start is completed the digital control block measures the output conditions If the output conditions do not warrant PFMmodeandtheISENSE signal is not consistently over 09 V then the device will operate in constant voltage mode

If no voltage is detected on VSENSE after 20 pulses it is assumed that the auxiliary winding of the transformer is either open or shorted and the iW1692 shuts down

As long as calculated TON for CV is less than the TONinCCthe IC operates in constant voltage mode

96 Constant Current OperationThe iW1692 has been designed to work in constant-current mode for battery charging applications If the output voltage drops but does not go below 20 of the nominal designed value the device operates in this mode

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Out

put V

olta

ge

Output CurrentICC

VNOM

CV mode

CC

mod

e

Figure 961 Modes of operation

97 Variable Frequency ModeThe iW1692 is designed to operate in discontinuous conduction (DCM) mode at a fixed frequency of 40 kHz in both CC and CV modes To avoid operation in continuous conduction (CCM) mode the iW1692 checks for the falling edge of the VSENSE input on every cycle If a falling edge of VSENSE is not detected during the normal 25μs period the switching period is extended until the falling edge VSENSEdoes occur If the switching period reaches 75μs without VSENSE being detected the iW1692 immediately shuts off

98 PFM Mode at Light LoadThe iW1692 operates in a fixed frequency PWM mode whenIOUT is greater than approximately 5 of the specified maximum load current As the output load IOUTisreducedtheon-time tON is decreased At the moment that tONdropsbelow tON_MIN thecontroller transitions toPulseFrequencyModulation (PFM) mode Thereafter the on-time is modulated by the line voltage and the off-time is modulated by the load current The device automatically returns to PWM mode when the load current increases

99 Internal Loop CompensationThe iW1692 incorporates an internal Digital Error Amplifier with no requirement for external loop compensation The loop stability is guaranteed by design to provide at least 45 degrees of phase margin and ndash20dB of gain margin

910 Voltage Protection FunctionsThe iW1692 includes functions that protect against input and output overvoltage

The input voltage is monitored by the VINpinandtheoutputvoltage is monitored by the VSENSE pin If the voltage at these pins exceed their undervoltage or overvoltage thresholds for more than 6 cycles the iW1692 shuts-down immediately However the IC remains biased which discharges the VCCsupply Once VCC drops below the UVLO threshold the controller resets itself and then initiates a new soft-startcycle The controller continues attempting start-up but does not fully start-up until the fault condition is removed

The output voltage can be high enough to damage the output capacitor when the feedback loop is broken The iW1692 uses the primary feedback only with no secondary feedback loop When the VSENSE pin is shorted to GND (by shortingopen sense resistor) The controller will shut off with 6 consecutive pulses after start-up

911 Cable Drop CompensationThe iW1692-30 incorporates an innovative method to compensate for any IR drop in the secondary circuitry including cable and cable connector A 5 W AC adapter with 5 VDC output has 6 deviation at 1 A load current due to the drop across the DC cable without cable compensation The iW1692-30 cancels this error by providing a voltage offset to the feedback signal based on the amount of load current detected The iW1692-30 has 300mV of cable drop compensation at maximum current The iW1692-00 does not include any cable compensation

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100 Design Example101 Design ProcedureThis design example gives the procedure for a flyback converter using iW1692 Refer to figure 1201 for the application circuit The design objectives for this adapter are given in table 101 It meets UL IEC and CEC requirements

Figure 301 Design Flow Chart

Determine Cable Drop Compensation

Determine the Design Specifications(Vout Iout_max Vin_max Vin_min efficiency and ripple)

Determine Turns Ratio

Determine Input Bulk Capacitance

Determine Current Sensing Resistor

Determine Magnetizing Inductance

Determine Primary Turns

Determine Secondary Turns

Can you wind this transformer

Determine Bias Turns

Determine Vsense Turns and Resistors

Determine Output Capacitance

Determine Snubber Network

Determine Ton Delay Compensation

Finish

Yes No

Determine Rvin Resistors

Is the real cable drop compensation value OK Yes

No

Figure 1001 iW1692 Design Flow Chart

Parameter Symbol Range

Input Voltage VIN 85 - 264 VRMS

Frequency fIN 47 - 64 Hz

NoLoadInput PIN 200 mW

Output Voltage VOUT_CABLE 495 - 505 V

Output Current IOUT 1A

Output Ripple VRIPPLE lt100mV

Power Out POUT 5WCEC Efficiency h 65

Table 101 iW1692 Design Specification Table

102 Cable Drop CompensationCable Drop Compensation is an option included in the iW1692-30 This option helps maintain the output voltage at the end of the cable that the power supply is designed for During CV (constant voltage) mode the output current changes as the voltage remains constant This is true for the output voltage at the output of the power supply board however in certain applications the device to be charged is not directly connected to the power supply but ratheris connected via a cable This cable is seen by the power supply as a resistance So as the output current increases the output voltage at the end of the cable begins to drop With the cable compensation option the iW1692 can compensate for the resistance of the cable by incrementally increasing the output voltage seen on the power supply board to cancel out the selected cable resistance

To find the right cable compensation type for a given cable pickthecabledropcompensationnumberthatisclosesttothe voltage drop of the cable under maximum output current Use equation 101 for VOUTwhereVFD is the forward voltage oftheoutputdiode

_ _ _OUT OUT CABLE CABLE DROP COMPENSATION fdV V V V= + +(101)

Using equation 101 we know for this design VOUT is 55 V assuming no cable drop compensation is chosen and the forwarddropontheoutputdiode(VFD) is 500 mV

103 Input SelectionVIN resistors are chosen primarily to scale down the inputvoltage for the IC The scale factor for the input voltage in the IC is 00043 and the internal impedance of this pin is 20 kΩ Therefore the VINresistorsshouldequateto

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20 20 46300043Vin

kR k MW= minus W = W

(102)

From equation 102 ideally RVIN should be 463 MΩ because R1 R10 and R11 add up to approximately 46 MΩ By selecting the value of RVIN the VINTON_MAX_PFM and (VINTON)MAX_LIMITaredetermined

( ) _900 sec00043

2020

IN ON MAX LIMIT

Vin

VV Tk

R k

sdotmicrosdot = times

W + W

(103)

( ) 185 sec0004320

20

IN ON PFM

Vin

VV Tk

R k

sdotmicrosdot = times

W + W (104)

Keep in mind by changing RVIN to be something other than 463 MΩ the minimum and maximum input voltage for start-up will also change

Since the iW1692 uses the exact scaled value of VINforitscalculations C6 should be included to filter out any noise that mayappearontheVIN signal This is especially important for line-in surge conditions

104 Turns RatioThe maximum allowable turns ratio between the primary and secondary winding is determined by the minimum detectable reset time of the transformer during PFM mode

( )_

_

IN ON PFMtr MAX

RESET MIN OUT

V TN

T Vtimes

=times (105)

To avoid continuous conduction the turns ratio must be high enough so that TRESET does not exceed TPERIOD ndash TON ndash TDEAD TPERIOD is given by the PWM switching frequency of 40 kHz TRESET_MAX is given by

_ _RESET MAX PERIOD ON MAX DEADT T T T= minus minus (106)

Thus the minimum turns ratio is given by

( )_

_

IN ON MAXtr MIN

RESET MAX OUT

V TN

T Vtimes

=times

(107)

(VINtimesTON)PFM is limited by the iW1692 to be 185 Vμs and TRESET_MIN is required by the IC to be 23 μs

_185 sec 15

23 sec 55tr MAXVN

Vsdotmicro

= =micro times (108)

The product of VIN and TON is limited by equation 109 for CC limit performance For this example we choose 750 Vμsec

( )700 sec 850 secIN ON MAXV V T Vsdotmicro lt times lt sdotmicro (109)

Assuming TON_MAX is 97micros solving for the minimum turns ratio yields

_

_

25 sec 97 sec 48 sec

105 secRESET MAX

RESET MAX

T

T

= micro minus micro minus micro

= micro (1010)

( )_750 sec 13

105 sec 55tr MINVN

Vsdotmicro

= =micro times (1011)

Pickanumberbetween themaximumandminimumturnsratio in the example the turn ratio is 13 A turn ratio in the range of 11 to 15 is suggested for optimal performance

105 Input Bulk CapacitorThe input bulk capacitance (C1+C2) is chosen to maintain enough input power to sustain constant output power even as the input voltage is dropping In order for this to be true totalinputbulkcapacitancemustbe

_

_2

1 2 2 2_ _

arcsin2

2

(2 )

INDC MIN

INAC MIN

V

VIN

INAC MIN INDC MIN line

OUT OUTIN

PowerSupply

P

C CV V f

V IP

times

times times

π + =

minus times

times=

h

(1012)

VINAC_MIN is the minimum input voltage (rms) to be inputted intothepowersupplyandƒlineisthelowestlinefrequencyforthe power supply (in this case 47 Hz) VINDC_MINiscalculatedbasedonthe(VINtimes TON)MAX product

( )_

_

IN ON MAXINDC MIN

ON MAX

V TV

Ttimes

= (1013)

First we must find TON_MAX to get VINDC_MIN In order for the powersupplytofunctionindiscontinuousconductionmodeTON_MAX should be smaller than the switching period minus the transformer reset time Given that the transformer reset timeis

( )IN ON MAXRESET

tr OUT

V TT

N Vtimes

=sdot (1014)

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Then the maximum on-time must be

( )_

IN ON MAXON MAX PERIOD dead

tr OUT

V TT T T

N Vtimes

= minus minussdot (1015)

TDEAD is about 48 μsec Knowing TPERIOD has to be 25 μsec because of the 40 kHz switching frequency

_750 sec25 sec 48 sec 97 sec13 55ON MAX

VTV

sdotmicro= micro minus minus micro = micro

times (1016)

From this result we can now get VINDC_MIN from equation1013

( )_

_772IN ON MAX

INDC MINON MAX

V TV V

Ttimes

= = (1017)

Substituting VINDC_MIN into equation 1012 we get

( )

( ) ( )

7722 85

1 2 2 2

arcsin2 641

21206

(2 85 772 ) 47

VVW

C C FV V Hz

times times times π + = = micro

times minus times (1018)

Increase the value of C1+C2 to account for efficiency losses For this example 136 microF is chosen

106 Current Sense ResistorThe ISENSEresistordeterminesthemaximumcurrentoutputof the power supply The output current of the power supply isdeterminedby

1_2

RESETOUT tr PRI PK

PERIOD

TI N I

T= times times times

(1019)

When the maximum current output is achieved the voltage seenon the ISENSEpin (VISENSE) should reach its maximum Thus at constant current limit

__

Isense CCPRI PK

Isense

VI

R=

(1020)

Substituting this into equation 1019 gives

_2 OUT Isense PERIOD

Isense CCtr RESET

I R TV

N Ttimes times

= times (1021)

During constant current mode where output current is at its maximum the first term in Equation 1021 is constant Therefore we can call this KC Substituting this back into equation 1021 we get

_PERIOD

Isense CC CRESET

TV K

T= times

(1022)

For iW1692 KC is 0264 V therefore RIsensedependsonthemaximumoutputcurrentby

2tr C

Isense xOUT

N KR

Itimes

= times htimes (1023)

Using this equation and Ntr from section 104

13 0264 87 152 1Isense

VRA

times= times = W

times (1024)

We recommend using plusmn1 tolerance resistors for RIsense

107 Magenitizing InductanceA feature of the iW1692 is the lack of dependence on the magnetizing inductance for the CC curve

Although the constant current limit does not depend on the magnetizing inductance there are still restrictions on the magnetizing inductance The maximum LMislimitedbytheamountofpowerthatneedstocomeoutofthetransformerin order for the power supply to regulate This is given by

( )

( )

2

__

_

402

IN ON MAXM MAX

XFMR MAX

OUT fd OUTXFMR MAX

X

V T kHzL

P

V V IP

times times=

times

minus times=

h

(1025)

The minimum LM is limited by the maximum allowableprimarypeakcurrent(IPRI_PK) 09 V on the ISENSEpinshouldcorrespond to the maximum allowable primary peak current Therefore the maximum primary peak current is

_09

PRI PKIsense

VIR

lt (1026)

Thus LMislimitedby

( )_ 09

IN ON MAXM MIN

Isense

V TL

V Rsdot

= (1027)

There is also a lower limit on ISENSE signal of 02 V This gives a second maximum value on LM compare this with the value obtained from equation 1025 and pick the smaller of the two values

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( )

2_

_ 2

2

02 40XFMR MAX Isense

M MAXP R

LV kHz

times times=

times (1028)

Assuming that the efficiency of the transformer is about 87 wecanobtaintheamountofpowerthatneedstocomeoutofthetransformeras

5 57587 87OUTP W W= =

(1029)

Substituting this into equation 1025 we get

( )2

_750 sec 40

1962 575M MAX

V kHzL mH

Wsdotmicro times

= =times (1030)

To get the minimum value of the primary inductance use the value for RISENSE from equation 1024

_09 615PRI PK

VI Alt =W (1031)

Substituting this primary peak current into equation 1027

_750 sec 12509 15M MIN

VL mHV

sdotmicro= =

W (1032)

Choose a primary inductance somewhere between 191 mH and 142 mH we chose 15 mH

Given a nominal LM we can now find the minimum turns ratio between primary and secondary that ensures the powersupply does not function at variable frequency (VF) mode before a certain desired voltage VOUT_VF Under VF mode theconstantcurrentIOUTmaynotbeasaccurateasinpulsewidth modulation mode

( )

_

__

2OUT VF OUT M PERIOD

xtr VF

OUT VF RESET

V I L T

NV T

times times times times

h=

times

(1033)

and

( )

_

_

2

2

OUT SHUTDOWN OUT M PERIOD

xON

INAC MIN

V I L T

TV

times times times times

h = times

(1034)

108 Primary WindingInordertokeepthetransformerfromsaturationthemaximumflux density must not be exceeded Therefore the minimum primary winding on the transformer must meet

( )IN ON MAXPRI

MAX e

V TN

B Asdot

getimes (1035)

Where BMAX is maximum flux density and Ae is the corearea

Picking (VINtimesTON)MAX to be 750 Vμsec and getting the maximum flux density and core area from the transformer datasheetwecancalculate theminimumnumberof turnsfor the primary winding Substitute BMAX as 320mT and the area of the core to be 192 mm2 we solve equation 1035 to get

2750 sec 1221

320 192PRIVN turns

mT mmsdotmicro

ge =times (1036)

To avoid hitting the maximum flux density pick a value for NPRIto be higher than this In this example 144 turns is picked

109 Secondary WindingFrom the primary winding turns we obtain the secondary winding

PRI

SECtr

NNN

= (1037)

Thus in our example

144 1113SECN turns= =

(1038)

At this point it is advantageous to make sure the primary winding and secondary winding chosen is actually feasible to wind

1010 Bias WindingVCC is the supply to the iW1692 and should be between 12 V and 16 V Capacitor C7 stores the VCC charge during IC operation and the controller checks this voltage and makes sure itrsquos within range The zener Z1 protects the IC from getting a VCC over voltage Thus the number of auxiliary windings needs to ensure that VCC does not exceed 16 V

( )SEC CC fdBIAS

OUT

N V VN

V

times +=

(1039)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

The number of auxiliary windings can be calculated using equation 1039

11 125 2555BIAS

turns VN turnsVtimes

= = (1040)

Here wersquove actually chosen a lower number for the bias winding 22 turns

1011 VSENSE Resistors and WindingThe output voltage regulation is mainly determined by the feedback signal VSENSE

_SENSE OUT PCB SENSEV V K= times (1041)

Where

4

4 3

VsenseSENSE

SEC

NRKR R N

= times+ (1042)

InternallyVSENSE is compared to a reference voltage VFB_MAX WithnocabledropcompensationVFB_MAX is 1538 V however cabledropcompensationactually increases this referenceby a specific amount as a function of the present output current Therefore under perfect regulation conditions

_ _SENSE FB MAX CableDropCompensation actual SENSEV V V K= + sdot(1043)

Substitute this into equation 1043 and solve for KSENSE

_

_ _

FB MAXSENSE

OUT PCB CableDropComp actual

VK

V V=

minus (1044)

Up to this point cable drop compensation voltage is assumed to be independent of the power supply circuitry when inreality this is not completely true

VCableDropComp_actual is affected by output regulation magnetizing inductance and RVIN

220_ 120

_ _2

_

M OUT OUT

Vin sw PowerSupply

L V IkCableDropComp actual k R f

CABLE DROP COMP OUT

FB MAX

V K

V VK

V

sdot sdot sdotWW+ sdoth

= sdot + sdot

times times(1045)

Where K1 is -1155 Vμsec and K2 is 0130 V-1μsec-1 Plug this number into equation 1045 and then find VSENSE winding and resistor values to satisfy equation 1042

For this design example we did not implement cable drop compensationsoVCableDropComp_actual is 0 V The cable drop for a 18 m 22 AWG wire is

1 2 531 18 190mOUT Cable mI R A m mVWtimes = times times times =

(1046)

Substituting this into equation 1044 we get

1538 035 0SENSE

VKV V

= =minus (1047)

Solving for R4 in equation 1042 assuming R3 is 20 kΩ and NVSENSE is 24 turns we get R4 should be around 3 kΩ

Since the iW1692 uses the VSENSE signal to determine the regulation point this signal can not be too noisy Thus C8 is used to help filter the VSENSE signal

1012 Output CapacitorsThe output capacitors are important for controlling the output voltage ripple of the power supply This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus how much charge is lost before the next switching cycle is the ripple on the output voltage Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then

_ _

OUTOUT

OUT RIPPLE PK

QC

V=

(1048)

Since charge is equal to current times time

_

_ _

OUT OFF MAXOUT

OUT RIPPLE PK

I TC

Vtimes

= (1049)

The maximum off-time during maximum output current is 25 μs

Assuming we want to get under 50 mV of ripple on the output we substitute this into equation 1049 to get

1 25 sec 50050OUT

AC FmV

times micro= = micro

(1050)

In this calculations ESR and ESL are ignored the reason this calculation is still valid is because of the second stage LC filter L3 and C11 These two components reduce the ESR and ESL ripple

1013 Snubber NetworkThe snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive The goal is to dissipate the energy from the leakage inductance of the transformer For simplicity and a more conservative design first assume the energy of the

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

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MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

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MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page2 82007

Pin Name Type Pin Description1 VSENSE

Input Voltage sense input from the auxiliary winding

2 GND Ground Ground connection

3 OUTPUT Output Gate drive output for the external power MOSFET switch

4 VCC Input Supply voltage

5 ISENSE Input Primary current sense Used for cycle-by-cycle peak current control and limit

6 VIN Input Senses average rectified input voltage

Parameter Symbol Value UnitsDC supply voltage range (pin 4 ICC = 20mA max) VCC

-03 to 18 V

DC supply current at VCCpin ICC20 mA

Output (pin 3) -03 to 18 V

VSENSEinput(pin1) -03 to 40 V

ISENSEinput(pin5) -03 to 40 V

VIN input (pin 6) -03 to 18 V

Power dissipation at TA le 25degC PD400 mW

Maximumjunctiontemperature TJ(MAX)125 degC

Storage temperature TSTGndash65 to 150 degC

Lead temperature during IR reflow for le 15 seconds TLEAD260 degC

Thermal resistance junction-to-ambient θJA240 degCW

ESD rating per JEDEC JESD22-A114 (HBM) 2000 V

Latch-Up test per JEDEC 78 plusmn100 mA

50 Absolute Maximum RatingsAbsolute maximum ratings are the parametric values or ranges which can cause permanent damage if exceeded For maximum safe operating conditions refer to Electrical Characteristics in Section 60

40 Pinout Description

iW1692

VSENSE

GND

OUTPUT

VIN

ISENSE

VCC

1

2

3

6

5

4

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

VCC = 12 V -40degC le TA le 85degC unless otherwise specified (Note 1)

Parameter Symbol Test Conditions Min Typ Max UnitVIN SECTION (Pin 6)

Start-up voltage threshold VINST TA= 25degC positive edge 366 407 448 mV

Start-upcurrent IINST VCC=10V 10 15 microA

Shutdown low voltage threshold VUVDC TA= 25degC negative edge 216 240 264 mV

Shutdown high voltage threshold VOVDC TA= 25degC positive edge 1737 1930 2123 V

Inputimpedance ZIN Afterstart-up 20 kW

VSENSE SECTION (Pin 1)

Input leakage current IBVS VSENSE = 2 V 1 μA

Nominal voltage threshold VSENSE(NOM) TA=25degC negative edge 1507 1538 1569 V

Output OVP threshold VSENSE(MAX) TA=25degC negative edge 1667 1700 1734 V

OUTPUT SECTION (Pin 3)

Output low level ON-resistance RDS(ON)LO ISINK=5mA 45 100 W

Output high level ON-resistance RDS(ON)HI ISOURCE=5mA 65 100 W

Rise time (Note 2) tRTA = 25degC CL = 330 pF10 to 90 40 75 ns

Fall time (Note 2) tFTA = 25degC CL = 330 pF90 to 10 40 75 ns

Output switching frequency (Note 3) fS ILOADgt15ofmaximum 36 40 44 kHz

VCC SECTION (Pin 4)

Maximum operating voltage VCC(MAX) 16 V

Start-upthreshold VCC(ST) VCC rising 110 120 132 V

Undervoltage lockout threshold VCC(UVL) VCC falling 55 60 66 V

Operating current ICCQCL = 330 pF VSENSE = 15 V 25 35 mA

ISENSE SECTION (Pin 5)

Peaklimitthreshold VPEAK 1000 mV

CClimitthreshold VCC-TH 900 mV

60 Electrical Characteristics

NotesNote 1 Adjust VCC above the start-up threshold before setting at 12 VNote 2 These parameters are not 100 tested guaranteed by design and characterizationNote 3 Frequency variation includes plusmn12 dithering for EMI suppression

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

70 Typical Performance Characteristics124

123

122

121

120

V CC

Sta

rt-u

p Th

resh

old

(V)

Ambient Temperature (degC)-50 -25 0 25 50 75 100

Figure 703 Start-Up Threshold vs Temperature

2015

2010

2005

2000

1995

Inte

rnal

Ref

eren

ce V

olta

ge (V

)

Ambient Temperature (degC)-50 -25 0 25 50 75 100

VCC = 12 V

Figure 704 Internal Reference vs Temperature

28

26

24

22

20

18

16

V CC

Sup

ply

Cur

rent

(mA

)

Load Capacitance (pF)0 200 400 600 800 1000

VCC = 12 VTA = 25deg

Figure 701 Supply Current vs Load Capacitance

44

42

40

38

36

Switc

hing

Fre

quen

cy (k

Hz)

Ambient Temperature (degC)-50 -25 0 25 50 75 100

VCC = 12 V

Figure 702 Switching Frequency vs Temperature

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

80 Functional Block Diagram

90 Theory of Operation

ndash

+

VIN

GND

PORPOR

VCC

2

VSENSE

VFB

VVMS

VIPK

VIN_A

OUTPUT

128 mV

1538 V

ndash

+CMP

ndash

+CMP

ISENSE5

10 V

02 V ~ 09 V

02 V ~ 20 V

IPEAK

ADC

VOCP

ndash

Start-up

DAC

ndash

+

1

6 4

DigitalLogic

Control

GateDriver

3

Figure 801 iW1692 Functional Block Diagram

The iW1692 is a digital controller which uses a new proprietary primary-side control technology to eliminate the opto-isolated feedback and secondary regulation circuits required in traditional designs This results in a low-cost solution for low power ACDC adapters The core PWM processor uses fixed-frequency Discontinuous Conduction Mode (DCM) operation at heavy load and switches to variable frequency operation at light loads to maximize efficiency Furthermore iWattrsquos digital control technology enables fast dynamic response tight output regulation and full featured circuit protection with primary-side control

Referring to the block diagram in Figure 801 the digital logic control generates the switching on-time and off-time information based on the line voltage and the output voltage feedback signal The system loop is internally compensated inside the digital logic control and no external analog components are required for loop compensation The iW1692 uses an advanced digital control algorithm to reduce system design time and improve reliability

Furthermoreaccuratesecondaryconstant-currentoperationis achieved without the need for any secondary-side sense and control circuits

The iW1692 uses PWM mode control at higher output power levels and switches to PFM mode at light load to minimize power dissipation Additional built-in protection features include overvoltage protection (OVP) output short circuit protection (SCP) AC low line brown out over current protection single pin fault protection and ISENSEfaultdetection

iWattrsquos digital control scheme is specifically designed to address the challenges and trade-offs of power conversion design This innovative technology is ideal for balancing new regulatory requirements for green mode operation with more practical design considerations such as lowest possible cost smallest size and high performance output control

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

VCC

VCC(ST)

OFF

ENABLE

Start-upSequencing

200micros

VIN VIN Impedance = 20k

VINSWON

Figure 921 Start-up Sequencing Diagram

93 Understanding Primary FeedbackFigure 931 illustrates a simplified flyback converter When the switch Q1 conducts during tONthecurrentigisdirectlydrawn from rectified sinusoid vg The energy Eg isstoredin the primary windingThe rectifying diode D1 is reverse biasedandtheloadcurrentIOissuppliedbythesecondarycapacitorCO When Q1 turns off D1 conducts and the stored energy Eg(t) is delivered to the output

+

vin(t)

TS(t)

IO

VO

NP NS

NAUX

D1

Q1

VAUX

COvg(t)

ig(t)+

ndash

iin(t) id(t)

Figure 931 Simplified Flyback Converter

In order to regulate the output voltage within a tight specification the information about the output voltage and load current needs to be accurately sensed In the DCM flyback converter this information can be read via the auxiliary winding or the primary magnetizing inductance (LM) During the Q1on-timetheloadcurrentissuppliedfromthe output filter capacitor CO The voltage across the primary winding is vg(t) assuming the voltage dropped across Q1iszero The current in Q1rampsuplinearlyatarateof

( ) ( )g g

M

di t v tdt L

=

(91)

91 Pin DetailPin 1 ndash VSENSE

Sense signal input from auxiliary winding This provides the secondary voltage feedback used for output regulation

Pin 2 ndash GND

Analog digital and power ground

Pin 3 ndash OUTPUT

Gate drive signal for the external power MOSFET switch

Pin 4 ndash VCC

Power supply for the controller during normal operation The controller starts up when VCC reaches 12 V (typical) and shuts-downwhentheVCC voltage is below 6 V (typical) A 100 nF decoupling capacitor should be connected between theVCC pin and GND

Pin 5 ndash ISENSE

Primary current sense

Pin 6 ndash VIN

Sense signal input representing the instantaneous rectified line voltage VIN is used for line regulation The internal impedanace is 20 kW and the scale factor is 00043 It also provides input undervoltage and overvoltage protection This pin also provides the supply current to the IC during start-up

92 Start-upPrior to start-up the VIN pin charges up the VCC capacitorthrough the diode between VINandVCC When VCC isfullycharged to a voltage higher than VCC(UVL) threshold then theVIN_SW turns on and the analog-to-digital converter begins to sense the input voltage The iW1692 commences soft-start function as soon as the voltage on VIN pin is above VINST

The iW1692 incorporates an internal soft-start function The soft-start time is set at 30 ms Once the VIN pin voltage has reached its turn-on threshold the iW1692 starts switching but limits the on-time to a percentage of the maximum on-time During the first 1 ms the on-time is limited to 25 During the next 1 ms the on-time is limited to 50 and during the last 1 ms the on-time is limited to 75

IfatanytimetheVCC voltage drops below VCC(UVL) threshold then all the digital logic is fully reset At this time the VIN_SWswitches off so that the VCC capacitor can be charged up again

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

Attheendofon-timethecurrenthasrampedupto

( ) ( )( ) g ON

gM

v t t ti t

Ltimes

=

(92)

This current represents a stored energy of

2( )2M

g gLE i t= times

(93)

WhenQ1turnsoffattOig(t)inLM forces a reversal of polarities on all windings Ignoring the communication-time caused by the leakage inductance LK at the instant of turn-off tO theprimarycurrenttransferstothesecondaryatanamplitudeof

( ) ( )Pd g

S

Ni t i tN

= times

(94)

Assuming the secondary winding is master the auxiliary winding is slave

VAUX 0V

VAUX = -VG xNAUX

NP

See equation 95

Figure 932 Auxiliary Voltage Waveforms

The auxiliary voltage is given by

( )AUXAUX O

S

NV V VN

= + ∆

(95)

and reflects the output voltage as shown in Figure 932

The voltage at the load differs from the secondary voltage by a diode drop and IR losses The diode drop is a function of current as are IR losses Thus if the secondary voltage is alwaysreadataconstantsecondarycurrentthedifferencebetween the output voltage and the secondary voltage is a fixed ΔV Furthermore if the voltage can be read when the secondarycurrentissmallΔV is small

The real-time waveform analyzer in the iW1692 reads this information cycle by cycle and then generates a feedback voltage VFB The VFB signal precisely represents the output voltage and is used to regulate the output voltage

94 Understanding CC and CV modeThe constant current mode (CC mode) is useful in battery charging applications During this mode of operation the iW1692 will regulate the output current at a constant maximum level regardless of the output voltage drop while avoiding continuous conduction mode

To achieve this regulation the iW1692 senses the load current indirectly through the primary current The primary currentisdetectedbytheISENSE pin through a resistor from the MOSFET source to ground (RSS) This resistor value is given by

( )( )2

CSS

OUTMAX

N KR

Itimes

=times

(96)

N is the ratio of primary turns to secondary turns of thetransformerandKC is 0264 V

95 Constant Voltage OperationAfter soft-start is completed the digital control block measures the output conditions If the output conditions do not warrant PFMmodeandtheISENSE signal is not consistently over 09 V then the device will operate in constant voltage mode

If no voltage is detected on VSENSE after 20 pulses it is assumed that the auxiliary winding of the transformer is either open or shorted and the iW1692 shuts down

As long as calculated TON for CV is less than the TONinCCthe IC operates in constant voltage mode

96 Constant Current OperationThe iW1692 has been designed to work in constant-current mode for battery charging applications If the output voltage drops but does not go below 20 of the nominal designed value the device operates in this mode

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

Out

put V

olta

ge

Output CurrentICC

VNOM

CV mode

CC

mod

e

Figure 961 Modes of operation

97 Variable Frequency ModeThe iW1692 is designed to operate in discontinuous conduction (DCM) mode at a fixed frequency of 40 kHz in both CC and CV modes To avoid operation in continuous conduction (CCM) mode the iW1692 checks for the falling edge of the VSENSE input on every cycle If a falling edge of VSENSE is not detected during the normal 25μs period the switching period is extended until the falling edge VSENSEdoes occur If the switching period reaches 75μs without VSENSE being detected the iW1692 immediately shuts off

98 PFM Mode at Light LoadThe iW1692 operates in a fixed frequency PWM mode whenIOUT is greater than approximately 5 of the specified maximum load current As the output load IOUTisreducedtheon-time tON is decreased At the moment that tONdropsbelow tON_MIN thecontroller transitions toPulseFrequencyModulation (PFM) mode Thereafter the on-time is modulated by the line voltage and the off-time is modulated by the load current The device automatically returns to PWM mode when the load current increases

99 Internal Loop CompensationThe iW1692 incorporates an internal Digital Error Amplifier with no requirement for external loop compensation The loop stability is guaranteed by design to provide at least 45 degrees of phase margin and ndash20dB of gain margin

910 Voltage Protection FunctionsThe iW1692 includes functions that protect against input and output overvoltage

The input voltage is monitored by the VINpinandtheoutputvoltage is monitored by the VSENSE pin If the voltage at these pins exceed their undervoltage or overvoltage thresholds for more than 6 cycles the iW1692 shuts-down immediately However the IC remains biased which discharges the VCCsupply Once VCC drops below the UVLO threshold the controller resets itself and then initiates a new soft-startcycle The controller continues attempting start-up but does not fully start-up until the fault condition is removed

The output voltage can be high enough to damage the output capacitor when the feedback loop is broken The iW1692 uses the primary feedback only with no secondary feedback loop When the VSENSE pin is shorted to GND (by shortingopen sense resistor) The controller will shut off with 6 consecutive pulses after start-up

911 Cable Drop CompensationThe iW1692-30 incorporates an innovative method to compensate for any IR drop in the secondary circuitry including cable and cable connector A 5 W AC adapter with 5 VDC output has 6 deviation at 1 A load current due to the drop across the DC cable without cable compensation The iW1692-30 cancels this error by providing a voltage offset to the feedback signal based on the amount of load current detected The iW1692-30 has 300mV of cable drop compensation at maximum current The iW1692-00 does not include any cable compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

100 Design Example101 Design ProcedureThis design example gives the procedure for a flyback converter using iW1692 Refer to figure 1201 for the application circuit The design objectives for this adapter are given in table 101 It meets UL IEC and CEC requirements

Figure 301 Design Flow Chart

Determine Cable Drop Compensation

Determine the Design Specifications(Vout Iout_max Vin_max Vin_min efficiency and ripple)

Determine Turns Ratio

Determine Input Bulk Capacitance

Determine Current Sensing Resistor

Determine Magnetizing Inductance

Determine Primary Turns

Determine Secondary Turns

Can you wind this transformer

Determine Bias Turns

Determine Vsense Turns and Resistors

Determine Output Capacitance

Determine Snubber Network

Determine Ton Delay Compensation

Finish

Yes No

Determine Rvin Resistors

Is the real cable drop compensation value OK Yes

No

Figure 1001 iW1692 Design Flow Chart

Parameter Symbol Range

Input Voltage VIN 85 - 264 VRMS

Frequency fIN 47 - 64 Hz

NoLoadInput PIN 200 mW

Output Voltage VOUT_CABLE 495 - 505 V

Output Current IOUT 1A

Output Ripple VRIPPLE lt100mV

Power Out POUT 5WCEC Efficiency h 65

Table 101 iW1692 Design Specification Table

102 Cable Drop CompensationCable Drop Compensation is an option included in the iW1692-30 This option helps maintain the output voltage at the end of the cable that the power supply is designed for During CV (constant voltage) mode the output current changes as the voltage remains constant This is true for the output voltage at the output of the power supply board however in certain applications the device to be charged is not directly connected to the power supply but ratheris connected via a cable This cable is seen by the power supply as a resistance So as the output current increases the output voltage at the end of the cable begins to drop With the cable compensation option the iW1692 can compensate for the resistance of the cable by incrementally increasing the output voltage seen on the power supply board to cancel out the selected cable resistance

To find the right cable compensation type for a given cable pickthecabledropcompensationnumberthatisclosesttothe voltage drop of the cable under maximum output current Use equation 101 for VOUTwhereVFD is the forward voltage oftheoutputdiode

_ _ _OUT OUT CABLE CABLE DROP COMPENSATION fdV V V V= + +(101)

Using equation 101 we know for this design VOUT is 55 V assuming no cable drop compensation is chosen and the forwarddropontheoutputdiode(VFD) is 500 mV

103 Input SelectionVIN resistors are chosen primarily to scale down the inputvoltage for the IC The scale factor for the input voltage in the IC is 00043 and the internal impedance of this pin is 20 kΩ Therefore the VINresistorsshouldequateto

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page0 82007

20 20 46300043Vin

kR k MW= minus W = W

(102)

From equation 102 ideally RVIN should be 463 MΩ because R1 R10 and R11 add up to approximately 46 MΩ By selecting the value of RVIN the VINTON_MAX_PFM and (VINTON)MAX_LIMITaredetermined

( ) _900 sec00043

2020

IN ON MAX LIMIT

Vin

VV Tk

R k

sdotmicrosdot = times

W + W

(103)

( ) 185 sec0004320

20

IN ON PFM

Vin

VV Tk

R k

sdotmicrosdot = times

W + W (104)

Keep in mind by changing RVIN to be something other than 463 MΩ the minimum and maximum input voltage for start-up will also change

Since the iW1692 uses the exact scaled value of VINforitscalculations C6 should be included to filter out any noise that mayappearontheVIN signal This is especially important for line-in surge conditions

104 Turns RatioThe maximum allowable turns ratio between the primary and secondary winding is determined by the minimum detectable reset time of the transformer during PFM mode

( )_

_

IN ON PFMtr MAX

RESET MIN OUT

V TN

T Vtimes

=times (105)

To avoid continuous conduction the turns ratio must be high enough so that TRESET does not exceed TPERIOD ndash TON ndash TDEAD TPERIOD is given by the PWM switching frequency of 40 kHz TRESET_MAX is given by

_ _RESET MAX PERIOD ON MAX DEADT T T T= minus minus (106)

Thus the minimum turns ratio is given by

( )_

_

IN ON MAXtr MIN

RESET MAX OUT

V TN

T Vtimes

=times

(107)

(VINtimesTON)PFM is limited by the iW1692 to be 185 Vμs and TRESET_MIN is required by the IC to be 23 μs

_185 sec 15

23 sec 55tr MAXVN

Vsdotmicro

= =micro times (108)

The product of VIN and TON is limited by equation 109 for CC limit performance For this example we choose 750 Vμsec

( )700 sec 850 secIN ON MAXV V T Vsdotmicro lt times lt sdotmicro (109)

Assuming TON_MAX is 97micros solving for the minimum turns ratio yields

_

_

25 sec 97 sec 48 sec

105 secRESET MAX

RESET MAX

T

T

= micro minus micro minus micro

= micro (1010)

( )_750 sec 13

105 sec 55tr MINVN

Vsdotmicro

= =micro times (1011)

Pickanumberbetween themaximumandminimumturnsratio in the example the turn ratio is 13 A turn ratio in the range of 11 to 15 is suggested for optimal performance

105 Input Bulk CapacitorThe input bulk capacitance (C1+C2) is chosen to maintain enough input power to sustain constant output power even as the input voltage is dropping In order for this to be true totalinputbulkcapacitancemustbe

_

_2

1 2 2 2_ _

arcsin2

2

(2 )

INDC MIN

INAC MIN

V

VIN

INAC MIN INDC MIN line

OUT OUTIN

PowerSupply

P

C CV V f

V IP

times

times times

π + =

minus times

times=

h

(1012)

VINAC_MIN is the minimum input voltage (rms) to be inputted intothepowersupplyandƒlineisthelowestlinefrequencyforthe power supply (in this case 47 Hz) VINDC_MINiscalculatedbasedonthe(VINtimes TON)MAX product

( )_

_

IN ON MAXINDC MIN

ON MAX

V TV

Ttimes

= (1013)

First we must find TON_MAX to get VINDC_MIN In order for the powersupplytofunctionindiscontinuousconductionmodeTON_MAX should be smaller than the switching period minus the transformer reset time Given that the transformer reset timeis

( )IN ON MAXRESET

tr OUT

V TT

N Vtimes

=sdot (1014)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

Then the maximum on-time must be

( )_

IN ON MAXON MAX PERIOD dead

tr OUT

V TT T T

N Vtimes

= minus minussdot (1015)

TDEAD is about 48 μsec Knowing TPERIOD has to be 25 μsec because of the 40 kHz switching frequency

_750 sec25 sec 48 sec 97 sec13 55ON MAX

VTV

sdotmicro= micro minus minus micro = micro

times (1016)

From this result we can now get VINDC_MIN from equation1013

( )_

_772IN ON MAX

INDC MINON MAX

V TV V

Ttimes

= = (1017)

Substituting VINDC_MIN into equation 1012 we get

( )

( ) ( )

7722 85

1 2 2 2

arcsin2 641

21206

(2 85 772 ) 47

VVW

C C FV V Hz

times times times π + = = micro

times minus times (1018)

Increase the value of C1+C2 to account for efficiency losses For this example 136 microF is chosen

106 Current Sense ResistorThe ISENSEresistordeterminesthemaximumcurrentoutputof the power supply The output current of the power supply isdeterminedby

1_2

RESETOUT tr PRI PK

PERIOD

TI N I

T= times times times

(1019)

When the maximum current output is achieved the voltage seenon the ISENSEpin (VISENSE) should reach its maximum Thus at constant current limit

__

Isense CCPRI PK

Isense

VI

R=

(1020)

Substituting this into equation 1019 gives

_2 OUT Isense PERIOD

Isense CCtr RESET

I R TV

N Ttimes times

= times (1021)

During constant current mode where output current is at its maximum the first term in Equation 1021 is constant Therefore we can call this KC Substituting this back into equation 1021 we get

_PERIOD

Isense CC CRESET

TV K

T= times

(1022)

For iW1692 KC is 0264 V therefore RIsensedependsonthemaximumoutputcurrentby

2tr C

Isense xOUT

N KR

Itimes

= times htimes (1023)

Using this equation and Ntr from section 104

13 0264 87 152 1Isense

VRA

times= times = W

times (1024)

We recommend using plusmn1 tolerance resistors for RIsense

107 Magenitizing InductanceA feature of the iW1692 is the lack of dependence on the magnetizing inductance for the CC curve

Although the constant current limit does not depend on the magnetizing inductance there are still restrictions on the magnetizing inductance The maximum LMislimitedbytheamountofpowerthatneedstocomeoutofthetransformerin order for the power supply to regulate This is given by

( )

( )

2

__

_

402

IN ON MAXM MAX

XFMR MAX

OUT fd OUTXFMR MAX

X

V T kHzL

P

V V IP

times times=

times

minus times=

h

(1025)

The minimum LM is limited by the maximum allowableprimarypeakcurrent(IPRI_PK) 09 V on the ISENSEpinshouldcorrespond to the maximum allowable primary peak current Therefore the maximum primary peak current is

_09

PRI PKIsense

VIR

lt (1026)

Thus LMislimitedby

( )_ 09

IN ON MAXM MIN

Isense

V TL

V Rsdot

= (1027)

There is also a lower limit on ISENSE signal of 02 V This gives a second maximum value on LM compare this with the value obtained from equation 1025 and pick the smaller of the two values

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page2 82007

( )

2_

_ 2

2

02 40XFMR MAX Isense

M MAXP R

LV kHz

times times=

times (1028)

Assuming that the efficiency of the transformer is about 87 wecanobtaintheamountofpowerthatneedstocomeoutofthetransformeras

5 57587 87OUTP W W= =

(1029)

Substituting this into equation 1025 we get

( )2

_750 sec 40

1962 575M MAX

V kHzL mH

Wsdotmicro times

= =times (1030)

To get the minimum value of the primary inductance use the value for RISENSE from equation 1024

_09 615PRI PK

VI Alt =W (1031)

Substituting this primary peak current into equation 1027

_750 sec 12509 15M MIN

VL mHV

sdotmicro= =

W (1032)

Choose a primary inductance somewhere between 191 mH and 142 mH we chose 15 mH

Given a nominal LM we can now find the minimum turns ratio between primary and secondary that ensures the powersupply does not function at variable frequency (VF) mode before a certain desired voltage VOUT_VF Under VF mode theconstantcurrentIOUTmaynotbeasaccurateasinpulsewidth modulation mode

( )

_

__

2OUT VF OUT M PERIOD

xtr VF

OUT VF RESET

V I L T

NV T

times times times times

h=

times

(1033)

and

( )

_

_

2

2

OUT SHUTDOWN OUT M PERIOD

xON

INAC MIN

V I L T

TV

times times times times

h = times

(1034)

108 Primary WindingInordertokeepthetransformerfromsaturationthemaximumflux density must not be exceeded Therefore the minimum primary winding on the transformer must meet

( )IN ON MAXPRI

MAX e

V TN

B Asdot

getimes (1035)

Where BMAX is maximum flux density and Ae is the corearea

Picking (VINtimesTON)MAX to be 750 Vμsec and getting the maximum flux density and core area from the transformer datasheetwecancalculate theminimumnumberof turnsfor the primary winding Substitute BMAX as 320mT and the area of the core to be 192 mm2 we solve equation 1035 to get

2750 sec 1221

320 192PRIVN turns

mT mmsdotmicro

ge =times (1036)

To avoid hitting the maximum flux density pick a value for NPRIto be higher than this In this example 144 turns is picked

109 Secondary WindingFrom the primary winding turns we obtain the secondary winding

PRI

SECtr

NNN

= (1037)

Thus in our example

144 1113SECN turns= =

(1038)

At this point it is advantageous to make sure the primary winding and secondary winding chosen is actually feasible to wind

1010 Bias WindingVCC is the supply to the iW1692 and should be between 12 V and 16 V Capacitor C7 stores the VCC charge during IC operation and the controller checks this voltage and makes sure itrsquos within range The zener Z1 protects the IC from getting a VCC over voltage Thus the number of auxiliary windings needs to ensure that VCC does not exceed 16 V

( )SEC CC fdBIAS

OUT

N V VN

V

times +=

(1039)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

The number of auxiliary windings can be calculated using equation 1039

11 125 2555BIAS

turns VN turnsVtimes

= = (1040)

Here wersquove actually chosen a lower number for the bias winding 22 turns

1011 VSENSE Resistors and WindingThe output voltage regulation is mainly determined by the feedback signal VSENSE

_SENSE OUT PCB SENSEV V K= times (1041)

Where

4

4 3

VsenseSENSE

SEC

NRKR R N

= times+ (1042)

InternallyVSENSE is compared to a reference voltage VFB_MAX WithnocabledropcompensationVFB_MAX is 1538 V however cabledropcompensationactually increases this referenceby a specific amount as a function of the present output current Therefore under perfect regulation conditions

_ _SENSE FB MAX CableDropCompensation actual SENSEV V V K= + sdot(1043)

Substitute this into equation 1043 and solve for KSENSE

_

_ _

FB MAXSENSE

OUT PCB CableDropComp actual

VK

V V=

minus (1044)

Up to this point cable drop compensation voltage is assumed to be independent of the power supply circuitry when inreality this is not completely true

VCableDropComp_actual is affected by output regulation magnetizing inductance and RVIN

220_ 120

_ _2

_

M OUT OUT

Vin sw PowerSupply

L V IkCableDropComp actual k R f

CABLE DROP COMP OUT

FB MAX

V K

V VK

V

sdot sdot sdotWW+ sdoth

= sdot + sdot

times times(1045)

Where K1 is -1155 Vμsec and K2 is 0130 V-1μsec-1 Plug this number into equation 1045 and then find VSENSE winding and resistor values to satisfy equation 1042

For this design example we did not implement cable drop compensationsoVCableDropComp_actual is 0 V The cable drop for a 18 m 22 AWG wire is

1 2 531 18 190mOUT Cable mI R A m mVWtimes = times times times =

(1046)

Substituting this into equation 1044 we get

1538 035 0SENSE

VKV V

= =minus (1047)

Solving for R4 in equation 1042 assuming R3 is 20 kΩ and NVSENSE is 24 turns we get R4 should be around 3 kΩ

Since the iW1692 uses the VSENSE signal to determine the regulation point this signal can not be too noisy Thus C8 is used to help filter the VSENSE signal

1012 Output CapacitorsThe output capacitors are important for controlling the output voltage ripple of the power supply This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus how much charge is lost before the next switching cycle is the ripple on the output voltage Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then

_ _

OUTOUT

OUT RIPPLE PK

QC

V=

(1048)

Since charge is equal to current times time

_

_ _

OUT OFF MAXOUT

OUT RIPPLE PK

I TC

Vtimes

= (1049)

The maximum off-time during maximum output current is 25 μs

Assuming we want to get under 50 mV of ripple on the output we substitute this into equation 1049 to get

1 25 sec 50050OUT

AC FmV

times micro= = micro

(1050)

In this calculations ESR and ESL are ignored the reason this calculation is still valid is because of the second stage LC filter L3 and C11 These two components reduce the ESR and ESL ripple

1013 Snubber NetworkThe snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive The goal is to dissipate the energy from the leakage inductance of the transformer For simplicity and a more conservative design first assume the energy of the

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

VCC = 12 V -40degC le TA le 85degC unless otherwise specified (Note 1)

Parameter Symbol Test Conditions Min Typ Max UnitVIN SECTION (Pin 6)

Start-up voltage threshold VINST TA= 25degC positive edge 366 407 448 mV

Start-upcurrent IINST VCC=10V 10 15 microA

Shutdown low voltage threshold VUVDC TA= 25degC negative edge 216 240 264 mV

Shutdown high voltage threshold VOVDC TA= 25degC positive edge 1737 1930 2123 V

Inputimpedance ZIN Afterstart-up 20 kW

VSENSE SECTION (Pin 1)

Input leakage current IBVS VSENSE = 2 V 1 μA

Nominal voltage threshold VSENSE(NOM) TA=25degC negative edge 1507 1538 1569 V

Output OVP threshold VSENSE(MAX) TA=25degC negative edge 1667 1700 1734 V

OUTPUT SECTION (Pin 3)

Output low level ON-resistance RDS(ON)LO ISINK=5mA 45 100 W

Output high level ON-resistance RDS(ON)HI ISOURCE=5mA 65 100 W

Rise time (Note 2) tRTA = 25degC CL = 330 pF10 to 90 40 75 ns

Fall time (Note 2) tFTA = 25degC CL = 330 pF90 to 10 40 75 ns

Output switching frequency (Note 3) fS ILOADgt15ofmaximum 36 40 44 kHz

VCC SECTION (Pin 4)

Maximum operating voltage VCC(MAX) 16 V

Start-upthreshold VCC(ST) VCC rising 110 120 132 V

Undervoltage lockout threshold VCC(UVL) VCC falling 55 60 66 V

Operating current ICCQCL = 330 pF VSENSE = 15 V 25 35 mA

ISENSE SECTION (Pin 5)

Peaklimitthreshold VPEAK 1000 mV

CClimitthreshold VCC-TH 900 mV

60 Electrical Characteristics

NotesNote 1 Adjust VCC above the start-up threshold before setting at 12 VNote 2 These parameters are not 100 tested guaranteed by design and characterizationNote 3 Frequency variation includes plusmn12 dithering for EMI suppression

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

70 Typical Performance Characteristics124

123

122

121

120

V CC

Sta

rt-u

p Th

resh

old

(V)

Ambient Temperature (degC)-50 -25 0 25 50 75 100

Figure 703 Start-Up Threshold vs Temperature

2015

2010

2005

2000

1995

Inte

rnal

Ref

eren

ce V

olta

ge (V

)

Ambient Temperature (degC)-50 -25 0 25 50 75 100

VCC = 12 V

Figure 704 Internal Reference vs Temperature

28

26

24

22

20

18

16

V CC

Sup

ply

Cur

rent

(mA

)

Load Capacitance (pF)0 200 400 600 800 1000

VCC = 12 VTA = 25deg

Figure 701 Supply Current vs Load Capacitance

44

42

40

38

36

Switc

hing

Fre

quen

cy (k

Hz)

Ambient Temperature (degC)-50 -25 0 25 50 75 100

VCC = 12 V

Figure 702 Switching Frequency vs Temperature

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

80 Functional Block Diagram

90 Theory of Operation

ndash

+

VIN

GND

PORPOR

VCC

2

VSENSE

VFB

VVMS

VIPK

VIN_A

OUTPUT

128 mV

1538 V

ndash

+CMP

ndash

+CMP

ISENSE5

10 V

02 V ~ 09 V

02 V ~ 20 V

IPEAK

ADC

VOCP

ndash

Start-up

DAC

ndash

+

1

6 4

DigitalLogic

Control

GateDriver

3

Figure 801 iW1692 Functional Block Diagram

The iW1692 is a digital controller which uses a new proprietary primary-side control technology to eliminate the opto-isolated feedback and secondary regulation circuits required in traditional designs This results in a low-cost solution for low power ACDC adapters The core PWM processor uses fixed-frequency Discontinuous Conduction Mode (DCM) operation at heavy load and switches to variable frequency operation at light loads to maximize efficiency Furthermore iWattrsquos digital control technology enables fast dynamic response tight output regulation and full featured circuit protection with primary-side control

Referring to the block diagram in Figure 801 the digital logic control generates the switching on-time and off-time information based on the line voltage and the output voltage feedback signal The system loop is internally compensated inside the digital logic control and no external analog components are required for loop compensation The iW1692 uses an advanced digital control algorithm to reduce system design time and improve reliability

Furthermoreaccuratesecondaryconstant-currentoperationis achieved without the need for any secondary-side sense and control circuits

The iW1692 uses PWM mode control at higher output power levels and switches to PFM mode at light load to minimize power dissipation Additional built-in protection features include overvoltage protection (OVP) output short circuit protection (SCP) AC low line brown out over current protection single pin fault protection and ISENSEfaultdetection

iWattrsquos digital control scheme is specifically designed to address the challenges and trade-offs of power conversion design This innovative technology is ideal for balancing new regulatory requirements for green mode operation with more practical design considerations such as lowest possible cost smallest size and high performance output control

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

VCC

VCC(ST)

OFF

ENABLE

Start-upSequencing

200micros

VIN VIN Impedance = 20k

VINSWON

Figure 921 Start-up Sequencing Diagram

93 Understanding Primary FeedbackFigure 931 illustrates a simplified flyback converter When the switch Q1 conducts during tONthecurrentigisdirectlydrawn from rectified sinusoid vg The energy Eg isstoredin the primary windingThe rectifying diode D1 is reverse biasedandtheloadcurrentIOissuppliedbythesecondarycapacitorCO When Q1 turns off D1 conducts and the stored energy Eg(t) is delivered to the output

+

vin(t)

TS(t)

IO

VO

NP NS

NAUX

D1

Q1

VAUX

COvg(t)

ig(t)+

ndash

iin(t) id(t)

Figure 931 Simplified Flyback Converter

In order to regulate the output voltage within a tight specification the information about the output voltage and load current needs to be accurately sensed In the DCM flyback converter this information can be read via the auxiliary winding or the primary magnetizing inductance (LM) During the Q1on-timetheloadcurrentissuppliedfromthe output filter capacitor CO The voltage across the primary winding is vg(t) assuming the voltage dropped across Q1iszero The current in Q1rampsuplinearlyatarateof

( ) ( )g g

M

di t v tdt L

=

(91)

91 Pin DetailPin 1 ndash VSENSE

Sense signal input from auxiliary winding This provides the secondary voltage feedback used for output regulation

Pin 2 ndash GND

Analog digital and power ground

Pin 3 ndash OUTPUT

Gate drive signal for the external power MOSFET switch

Pin 4 ndash VCC

Power supply for the controller during normal operation The controller starts up when VCC reaches 12 V (typical) and shuts-downwhentheVCC voltage is below 6 V (typical) A 100 nF decoupling capacitor should be connected between theVCC pin and GND

Pin 5 ndash ISENSE

Primary current sense

Pin 6 ndash VIN

Sense signal input representing the instantaneous rectified line voltage VIN is used for line regulation The internal impedanace is 20 kW and the scale factor is 00043 It also provides input undervoltage and overvoltage protection This pin also provides the supply current to the IC during start-up

92 Start-upPrior to start-up the VIN pin charges up the VCC capacitorthrough the diode between VINandVCC When VCC isfullycharged to a voltage higher than VCC(UVL) threshold then theVIN_SW turns on and the analog-to-digital converter begins to sense the input voltage The iW1692 commences soft-start function as soon as the voltage on VIN pin is above VINST

The iW1692 incorporates an internal soft-start function The soft-start time is set at 30 ms Once the VIN pin voltage has reached its turn-on threshold the iW1692 starts switching but limits the on-time to a percentage of the maximum on-time During the first 1 ms the on-time is limited to 25 During the next 1 ms the on-time is limited to 50 and during the last 1 ms the on-time is limited to 75

IfatanytimetheVCC voltage drops below VCC(UVL) threshold then all the digital logic is fully reset At this time the VIN_SWswitches off so that the VCC capacitor can be charged up again

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

Attheendofon-timethecurrenthasrampedupto

( ) ( )( ) g ON

gM

v t t ti t

Ltimes

=

(92)

This current represents a stored energy of

2( )2M

g gLE i t= times

(93)

WhenQ1turnsoffattOig(t)inLM forces a reversal of polarities on all windings Ignoring the communication-time caused by the leakage inductance LK at the instant of turn-off tO theprimarycurrenttransferstothesecondaryatanamplitudeof

( ) ( )Pd g

S

Ni t i tN

= times

(94)

Assuming the secondary winding is master the auxiliary winding is slave

VAUX 0V

VAUX = -VG xNAUX

NP

See equation 95

Figure 932 Auxiliary Voltage Waveforms

The auxiliary voltage is given by

( )AUXAUX O

S

NV V VN

= + ∆

(95)

and reflects the output voltage as shown in Figure 932

The voltage at the load differs from the secondary voltage by a diode drop and IR losses The diode drop is a function of current as are IR losses Thus if the secondary voltage is alwaysreadataconstantsecondarycurrentthedifferencebetween the output voltage and the secondary voltage is a fixed ΔV Furthermore if the voltage can be read when the secondarycurrentissmallΔV is small

The real-time waveform analyzer in the iW1692 reads this information cycle by cycle and then generates a feedback voltage VFB The VFB signal precisely represents the output voltage and is used to regulate the output voltage

94 Understanding CC and CV modeThe constant current mode (CC mode) is useful in battery charging applications During this mode of operation the iW1692 will regulate the output current at a constant maximum level regardless of the output voltage drop while avoiding continuous conduction mode

To achieve this regulation the iW1692 senses the load current indirectly through the primary current The primary currentisdetectedbytheISENSE pin through a resistor from the MOSFET source to ground (RSS) This resistor value is given by

( )( )2

CSS

OUTMAX

N KR

Itimes

=times

(96)

N is the ratio of primary turns to secondary turns of thetransformerandKC is 0264 V

95 Constant Voltage OperationAfter soft-start is completed the digital control block measures the output conditions If the output conditions do not warrant PFMmodeandtheISENSE signal is not consistently over 09 V then the device will operate in constant voltage mode

If no voltage is detected on VSENSE after 20 pulses it is assumed that the auxiliary winding of the transformer is either open or shorted and the iW1692 shuts down

As long as calculated TON for CV is less than the TONinCCthe IC operates in constant voltage mode

96 Constant Current OperationThe iW1692 has been designed to work in constant-current mode for battery charging applications If the output voltage drops but does not go below 20 of the nominal designed value the device operates in this mode

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

Out

put V

olta

ge

Output CurrentICC

VNOM

CV mode

CC

mod

e

Figure 961 Modes of operation

97 Variable Frequency ModeThe iW1692 is designed to operate in discontinuous conduction (DCM) mode at a fixed frequency of 40 kHz in both CC and CV modes To avoid operation in continuous conduction (CCM) mode the iW1692 checks for the falling edge of the VSENSE input on every cycle If a falling edge of VSENSE is not detected during the normal 25μs period the switching period is extended until the falling edge VSENSEdoes occur If the switching period reaches 75μs without VSENSE being detected the iW1692 immediately shuts off

98 PFM Mode at Light LoadThe iW1692 operates in a fixed frequency PWM mode whenIOUT is greater than approximately 5 of the specified maximum load current As the output load IOUTisreducedtheon-time tON is decreased At the moment that tONdropsbelow tON_MIN thecontroller transitions toPulseFrequencyModulation (PFM) mode Thereafter the on-time is modulated by the line voltage and the off-time is modulated by the load current The device automatically returns to PWM mode when the load current increases

99 Internal Loop CompensationThe iW1692 incorporates an internal Digital Error Amplifier with no requirement for external loop compensation The loop stability is guaranteed by design to provide at least 45 degrees of phase margin and ndash20dB of gain margin

910 Voltage Protection FunctionsThe iW1692 includes functions that protect against input and output overvoltage

The input voltage is monitored by the VINpinandtheoutputvoltage is monitored by the VSENSE pin If the voltage at these pins exceed their undervoltage or overvoltage thresholds for more than 6 cycles the iW1692 shuts-down immediately However the IC remains biased which discharges the VCCsupply Once VCC drops below the UVLO threshold the controller resets itself and then initiates a new soft-startcycle The controller continues attempting start-up but does not fully start-up until the fault condition is removed

The output voltage can be high enough to damage the output capacitor when the feedback loop is broken The iW1692 uses the primary feedback only with no secondary feedback loop When the VSENSE pin is shorted to GND (by shortingopen sense resistor) The controller will shut off with 6 consecutive pulses after start-up

911 Cable Drop CompensationThe iW1692-30 incorporates an innovative method to compensate for any IR drop in the secondary circuitry including cable and cable connector A 5 W AC adapter with 5 VDC output has 6 deviation at 1 A load current due to the drop across the DC cable without cable compensation The iW1692-30 cancels this error by providing a voltage offset to the feedback signal based on the amount of load current detected The iW1692-30 has 300mV of cable drop compensation at maximum current The iW1692-00 does not include any cable compensation

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100 Design Example101 Design ProcedureThis design example gives the procedure for a flyback converter using iW1692 Refer to figure 1201 for the application circuit The design objectives for this adapter are given in table 101 It meets UL IEC and CEC requirements

Figure 301 Design Flow Chart

Determine Cable Drop Compensation

Determine the Design Specifications(Vout Iout_max Vin_max Vin_min efficiency and ripple)

Determine Turns Ratio

Determine Input Bulk Capacitance

Determine Current Sensing Resistor

Determine Magnetizing Inductance

Determine Primary Turns

Determine Secondary Turns

Can you wind this transformer

Determine Bias Turns

Determine Vsense Turns and Resistors

Determine Output Capacitance

Determine Snubber Network

Determine Ton Delay Compensation

Finish

Yes No

Determine Rvin Resistors

Is the real cable drop compensation value OK Yes

No

Figure 1001 iW1692 Design Flow Chart

Parameter Symbol Range

Input Voltage VIN 85 - 264 VRMS

Frequency fIN 47 - 64 Hz

NoLoadInput PIN 200 mW

Output Voltage VOUT_CABLE 495 - 505 V

Output Current IOUT 1A

Output Ripple VRIPPLE lt100mV

Power Out POUT 5WCEC Efficiency h 65

Table 101 iW1692 Design Specification Table

102 Cable Drop CompensationCable Drop Compensation is an option included in the iW1692-30 This option helps maintain the output voltage at the end of the cable that the power supply is designed for During CV (constant voltage) mode the output current changes as the voltage remains constant This is true for the output voltage at the output of the power supply board however in certain applications the device to be charged is not directly connected to the power supply but ratheris connected via a cable This cable is seen by the power supply as a resistance So as the output current increases the output voltage at the end of the cable begins to drop With the cable compensation option the iW1692 can compensate for the resistance of the cable by incrementally increasing the output voltage seen on the power supply board to cancel out the selected cable resistance

To find the right cable compensation type for a given cable pickthecabledropcompensationnumberthatisclosesttothe voltage drop of the cable under maximum output current Use equation 101 for VOUTwhereVFD is the forward voltage oftheoutputdiode

_ _ _OUT OUT CABLE CABLE DROP COMPENSATION fdV V V V= + +(101)

Using equation 101 we know for this design VOUT is 55 V assuming no cable drop compensation is chosen and the forwarddropontheoutputdiode(VFD) is 500 mV

103 Input SelectionVIN resistors are chosen primarily to scale down the inputvoltage for the IC The scale factor for the input voltage in the IC is 00043 and the internal impedance of this pin is 20 kΩ Therefore the VINresistorsshouldequateto

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MK-4008-NR Page0 82007

20 20 46300043Vin

kR k MW= minus W = W

(102)

From equation 102 ideally RVIN should be 463 MΩ because R1 R10 and R11 add up to approximately 46 MΩ By selecting the value of RVIN the VINTON_MAX_PFM and (VINTON)MAX_LIMITaredetermined

( ) _900 sec00043

2020

IN ON MAX LIMIT

Vin

VV Tk

R k

sdotmicrosdot = times

W + W

(103)

( ) 185 sec0004320

20

IN ON PFM

Vin

VV Tk

R k

sdotmicrosdot = times

W + W (104)

Keep in mind by changing RVIN to be something other than 463 MΩ the minimum and maximum input voltage for start-up will also change

Since the iW1692 uses the exact scaled value of VINforitscalculations C6 should be included to filter out any noise that mayappearontheVIN signal This is especially important for line-in surge conditions

104 Turns RatioThe maximum allowable turns ratio between the primary and secondary winding is determined by the minimum detectable reset time of the transformer during PFM mode

( )_

_

IN ON PFMtr MAX

RESET MIN OUT

V TN

T Vtimes

=times (105)

To avoid continuous conduction the turns ratio must be high enough so that TRESET does not exceed TPERIOD ndash TON ndash TDEAD TPERIOD is given by the PWM switching frequency of 40 kHz TRESET_MAX is given by

_ _RESET MAX PERIOD ON MAX DEADT T T T= minus minus (106)

Thus the minimum turns ratio is given by

( )_

_

IN ON MAXtr MIN

RESET MAX OUT

V TN

T Vtimes

=times

(107)

(VINtimesTON)PFM is limited by the iW1692 to be 185 Vμs and TRESET_MIN is required by the IC to be 23 μs

_185 sec 15

23 sec 55tr MAXVN

Vsdotmicro

= =micro times (108)

The product of VIN and TON is limited by equation 109 for CC limit performance For this example we choose 750 Vμsec

( )700 sec 850 secIN ON MAXV V T Vsdotmicro lt times lt sdotmicro (109)

Assuming TON_MAX is 97micros solving for the minimum turns ratio yields

_

_

25 sec 97 sec 48 sec

105 secRESET MAX

RESET MAX

T

T

= micro minus micro minus micro

= micro (1010)

( )_750 sec 13

105 sec 55tr MINVN

Vsdotmicro

= =micro times (1011)

Pickanumberbetween themaximumandminimumturnsratio in the example the turn ratio is 13 A turn ratio in the range of 11 to 15 is suggested for optimal performance

105 Input Bulk CapacitorThe input bulk capacitance (C1+C2) is chosen to maintain enough input power to sustain constant output power even as the input voltage is dropping In order for this to be true totalinputbulkcapacitancemustbe

_

_2

1 2 2 2_ _

arcsin2

2

(2 )

INDC MIN

INAC MIN

V

VIN

INAC MIN INDC MIN line

OUT OUTIN

PowerSupply

P

C CV V f

V IP

times

times times

π + =

minus times

times=

h

(1012)

VINAC_MIN is the minimum input voltage (rms) to be inputted intothepowersupplyandƒlineisthelowestlinefrequencyforthe power supply (in this case 47 Hz) VINDC_MINiscalculatedbasedonthe(VINtimes TON)MAX product

( )_

_

IN ON MAXINDC MIN

ON MAX

V TV

Ttimes

= (1013)

First we must find TON_MAX to get VINDC_MIN In order for the powersupplytofunctionindiscontinuousconductionmodeTON_MAX should be smaller than the switching period minus the transformer reset time Given that the transformer reset timeis

( )IN ON MAXRESET

tr OUT

V TT

N Vtimes

=sdot (1014)

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MK-4008-NR Page 82007

Then the maximum on-time must be

( )_

IN ON MAXON MAX PERIOD dead

tr OUT

V TT T T

N Vtimes

= minus minussdot (1015)

TDEAD is about 48 μsec Knowing TPERIOD has to be 25 μsec because of the 40 kHz switching frequency

_750 sec25 sec 48 sec 97 sec13 55ON MAX

VTV

sdotmicro= micro minus minus micro = micro

times (1016)

From this result we can now get VINDC_MIN from equation1013

( )_

_772IN ON MAX

INDC MINON MAX

V TV V

Ttimes

= = (1017)

Substituting VINDC_MIN into equation 1012 we get

( )

( ) ( )

7722 85

1 2 2 2

arcsin2 641

21206

(2 85 772 ) 47

VVW

C C FV V Hz

times times times π + = = micro

times minus times (1018)

Increase the value of C1+C2 to account for efficiency losses For this example 136 microF is chosen

106 Current Sense ResistorThe ISENSEresistordeterminesthemaximumcurrentoutputof the power supply The output current of the power supply isdeterminedby

1_2

RESETOUT tr PRI PK

PERIOD

TI N I

T= times times times

(1019)

When the maximum current output is achieved the voltage seenon the ISENSEpin (VISENSE) should reach its maximum Thus at constant current limit

__

Isense CCPRI PK

Isense

VI

R=

(1020)

Substituting this into equation 1019 gives

_2 OUT Isense PERIOD

Isense CCtr RESET

I R TV

N Ttimes times

= times (1021)

During constant current mode where output current is at its maximum the first term in Equation 1021 is constant Therefore we can call this KC Substituting this back into equation 1021 we get

_PERIOD

Isense CC CRESET

TV K

T= times

(1022)

For iW1692 KC is 0264 V therefore RIsensedependsonthemaximumoutputcurrentby

2tr C

Isense xOUT

N KR

Itimes

= times htimes (1023)

Using this equation and Ntr from section 104

13 0264 87 152 1Isense

VRA

times= times = W

times (1024)

We recommend using plusmn1 tolerance resistors for RIsense

107 Magenitizing InductanceA feature of the iW1692 is the lack of dependence on the magnetizing inductance for the CC curve

Although the constant current limit does not depend on the magnetizing inductance there are still restrictions on the magnetizing inductance The maximum LMislimitedbytheamountofpowerthatneedstocomeoutofthetransformerin order for the power supply to regulate This is given by

( )

( )

2

__

_

402

IN ON MAXM MAX

XFMR MAX

OUT fd OUTXFMR MAX

X

V T kHzL

P

V V IP

times times=

times

minus times=

h

(1025)

The minimum LM is limited by the maximum allowableprimarypeakcurrent(IPRI_PK) 09 V on the ISENSEpinshouldcorrespond to the maximum allowable primary peak current Therefore the maximum primary peak current is

_09

PRI PKIsense

VIR

lt (1026)

Thus LMislimitedby

( )_ 09

IN ON MAXM MIN

Isense

V TL

V Rsdot

= (1027)

There is also a lower limit on ISENSE signal of 02 V This gives a second maximum value on LM compare this with the value obtained from equation 1025 and pick the smaller of the two values

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page2 82007

( )

2_

_ 2

2

02 40XFMR MAX Isense

M MAXP R

LV kHz

times times=

times (1028)

Assuming that the efficiency of the transformer is about 87 wecanobtaintheamountofpowerthatneedstocomeoutofthetransformeras

5 57587 87OUTP W W= =

(1029)

Substituting this into equation 1025 we get

( )2

_750 sec 40

1962 575M MAX

V kHzL mH

Wsdotmicro times

= =times (1030)

To get the minimum value of the primary inductance use the value for RISENSE from equation 1024

_09 615PRI PK

VI Alt =W (1031)

Substituting this primary peak current into equation 1027

_750 sec 12509 15M MIN

VL mHV

sdotmicro= =

W (1032)

Choose a primary inductance somewhere between 191 mH and 142 mH we chose 15 mH

Given a nominal LM we can now find the minimum turns ratio between primary and secondary that ensures the powersupply does not function at variable frequency (VF) mode before a certain desired voltage VOUT_VF Under VF mode theconstantcurrentIOUTmaynotbeasaccurateasinpulsewidth modulation mode

( )

_

__

2OUT VF OUT M PERIOD

xtr VF

OUT VF RESET

V I L T

NV T

times times times times

h=

times

(1033)

and

( )

_

_

2

2

OUT SHUTDOWN OUT M PERIOD

xON

INAC MIN

V I L T

TV

times times times times

h = times

(1034)

108 Primary WindingInordertokeepthetransformerfromsaturationthemaximumflux density must not be exceeded Therefore the minimum primary winding on the transformer must meet

( )IN ON MAXPRI

MAX e

V TN

B Asdot

getimes (1035)

Where BMAX is maximum flux density and Ae is the corearea

Picking (VINtimesTON)MAX to be 750 Vμsec and getting the maximum flux density and core area from the transformer datasheetwecancalculate theminimumnumberof turnsfor the primary winding Substitute BMAX as 320mT and the area of the core to be 192 mm2 we solve equation 1035 to get

2750 sec 1221

320 192PRIVN turns

mT mmsdotmicro

ge =times (1036)

To avoid hitting the maximum flux density pick a value for NPRIto be higher than this In this example 144 turns is picked

109 Secondary WindingFrom the primary winding turns we obtain the secondary winding

PRI

SECtr

NNN

= (1037)

Thus in our example

144 1113SECN turns= =

(1038)

At this point it is advantageous to make sure the primary winding and secondary winding chosen is actually feasible to wind

1010 Bias WindingVCC is the supply to the iW1692 and should be between 12 V and 16 V Capacitor C7 stores the VCC charge during IC operation and the controller checks this voltage and makes sure itrsquos within range The zener Z1 protects the IC from getting a VCC over voltage Thus the number of auxiliary windings needs to ensure that VCC does not exceed 16 V

( )SEC CC fdBIAS

OUT

N V VN

V

times +=

(1039)

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The number of auxiliary windings can be calculated using equation 1039

11 125 2555BIAS

turns VN turnsVtimes

= = (1040)

Here wersquove actually chosen a lower number for the bias winding 22 turns

1011 VSENSE Resistors and WindingThe output voltage regulation is mainly determined by the feedback signal VSENSE

_SENSE OUT PCB SENSEV V K= times (1041)

Where

4

4 3

VsenseSENSE

SEC

NRKR R N

= times+ (1042)

InternallyVSENSE is compared to a reference voltage VFB_MAX WithnocabledropcompensationVFB_MAX is 1538 V however cabledropcompensationactually increases this referenceby a specific amount as a function of the present output current Therefore under perfect regulation conditions

_ _SENSE FB MAX CableDropCompensation actual SENSEV V V K= + sdot(1043)

Substitute this into equation 1043 and solve for KSENSE

_

_ _

FB MAXSENSE

OUT PCB CableDropComp actual

VK

V V=

minus (1044)

Up to this point cable drop compensation voltage is assumed to be independent of the power supply circuitry when inreality this is not completely true

VCableDropComp_actual is affected by output regulation magnetizing inductance and RVIN

220_ 120

_ _2

_

M OUT OUT

Vin sw PowerSupply

L V IkCableDropComp actual k R f

CABLE DROP COMP OUT

FB MAX

V K

V VK

V

sdot sdot sdotWW+ sdoth

= sdot + sdot

times times(1045)

Where K1 is -1155 Vμsec and K2 is 0130 V-1μsec-1 Plug this number into equation 1045 and then find VSENSE winding and resistor values to satisfy equation 1042

For this design example we did not implement cable drop compensationsoVCableDropComp_actual is 0 V The cable drop for a 18 m 22 AWG wire is

1 2 531 18 190mOUT Cable mI R A m mVWtimes = times times times =

(1046)

Substituting this into equation 1044 we get

1538 035 0SENSE

VKV V

= =minus (1047)

Solving for R4 in equation 1042 assuming R3 is 20 kΩ and NVSENSE is 24 turns we get R4 should be around 3 kΩ

Since the iW1692 uses the VSENSE signal to determine the regulation point this signal can not be too noisy Thus C8 is used to help filter the VSENSE signal

1012 Output CapacitorsThe output capacitors are important for controlling the output voltage ripple of the power supply This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus how much charge is lost before the next switching cycle is the ripple on the output voltage Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then

_ _

OUTOUT

OUT RIPPLE PK

QC

V=

(1048)

Since charge is equal to current times time

_

_ _

OUT OFF MAXOUT

OUT RIPPLE PK

I TC

Vtimes

= (1049)

The maximum off-time during maximum output current is 25 μs

Assuming we want to get under 50 mV of ripple on the output we substitute this into equation 1049 to get

1 25 sec 50050OUT

AC FmV

times micro= = micro

(1050)

In this calculations ESR and ESL are ignored the reason this calculation is still valid is because of the second stage LC filter L3 and C11 These two components reduce the ESR and ESL ripple

1013 Snubber NetworkThe snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive The goal is to dissipate the energy from the leakage inductance of the transformer For simplicity and a more conservative design first assume the energy of the

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MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

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MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

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MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

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130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

70 Typical Performance Characteristics124

123

122

121

120

V CC

Sta

rt-u

p Th

resh

old

(V)

Ambient Temperature (degC)-50 -25 0 25 50 75 100

Figure 703 Start-Up Threshold vs Temperature

2015

2010

2005

2000

1995

Inte

rnal

Ref

eren

ce V

olta

ge (V

)

Ambient Temperature (degC)-50 -25 0 25 50 75 100

VCC = 12 V

Figure 704 Internal Reference vs Temperature

28

26

24

22

20

18

16

V CC

Sup

ply

Cur

rent

(mA

)

Load Capacitance (pF)0 200 400 600 800 1000

VCC = 12 VTA = 25deg

Figure 701 Supply Current vs Load Capacitance

44

42

40

38

36

Switc

hing

Fre

quen

cy (k

Hz)

Ambient Temperature (degC)-50 -25 0 25 50 75 100

VCC = 12 V

Figure 702 Switching Frequency vs Temperature

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

80 Functional Block Diagram

90 Theory of Operation

ndash

+

VIN

GND

PORPOR

VCC

2

VSENSE

VFB

VVMS

VIPK

VIN_A

OUTPUT

128 mV

1538 V

ndash

+CMP

ndash

+CMP

ISENSE5

10 V

02 V ~ 09 V

02 V ~ 20 V

IPEAK

ADC

VOCP

ndash

Start-up

DAC

ndash

+

1

6 4

DigitalLogic

Control

GateDriver

3

Figure 801 iW1692 Functional Block Diagram

The iW1692 is a digital controller which uses a new proprietary primary-side control technology to eliminate the opto-isolated feedback and secondary regulation circuits required in traditional designs This results in a low-cost solution for low power ACDC adapters The core PWM processor uses fixed-frequency Discontinuous Conduction Mode (DCM) operation at heavy load and switches to variable frequency operation at light loads to maximize efficiency Furthermore iWattrsquos digital control technology enables fast dynamic response tight output regulation and full featured circuit protection with primary-side control

Referring to the block diagram in Figure 801 the digital logic control generates the switching on-time and off-time information based on the line voltage and the output voltage feedback signal The system loop is internally compensated inside the digital logic control and no external analog components are required for loop compensation The iW1692 uses an advanced digital control algorithm to reduce system design time and improve reliability

Furthermoreaccuratesecondaryconstant-currentoperationis achieved without the need for any secondary-side sense and control circuits

The iW1692 uses PWM mode control at higher output power levels and switches to PFM mode at light load to minimize power dissipation Additional built-in protection features include overvoltage protection (OVP) output short circuit protection (SCP) AC low line brown out over current protection single pin fault protection and ISENSEfaultdetection

iWattrsquos digital control scheme is specifically designed to address the challenges and trade-offs of power conversion design This innovative technology is ideal for balancing new regulatory requirements for green mode operation with more practical design considerations such as lowest possible cost smallest size and high performance output control

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

VCC

VCC(ST)

OFF

ENABLE

Start-upSequencing

200micros

VIN VIN Impedance = 20k

VINSWON

Figure 921 Start-up Sequencing Diagram

93 Understanding Primary FeedbackFigure 931 illustrates a simplified flyback converter When the switch Q1 conducts during tONthecurrentigisdirectlydrawn from rectified sinusoid vg The energy Eg isstoredin the primary windingThe rectifying diode D1 is reverse biasedandtheloadcurrentIOissuppliedbythesecondarycapacitorCO When Q1 turns off D1 conducts and the stored energy Eg(t) is delivered to the output

+

vin(t)

TS(t)

IO

VO

NP NS

NAUX

D1

Q1

VAUX

COvg(t)

ig(t)+

ndash

iin(t) id(t)

Figure 931 Simplified Flyback Converter

In order to regulate the output voltage within a tight specification the information about the output voltage and load current needs to be accurately sensed In the DCM flyback converter this information can be read via the auxiliary winding or the primary magnetizing inductance (LM) During the Q1on-timetheloadcurrentissuppliedfromthe output filter capacitor CO The voltage across the primary winding is vg(t) assuming the voltage dropped across Q1iszero The current in Q1rampsuplinearlyatarateof

( ) ( )g g

M

di t v tdt L

=

(91)

91 Pin DetailPin 1 ndash VSENSE

Sense signal input from auxiliary winding This provides the secondary voltage feedback used for output regulation

Pin 2 ndash GND

Analog digital and power ground

Pin 3 ndash OUTPUT

Gate drive signal for the external power MOSFET switch

Pin 4 ndash VCC

Power supply for the controller during normal operation The controller starts up when VCC reaches 12 V (typical) and shuts-downwhentheVCC voltage is below 6 V (typical) A 100 nF decoupling capacitor should be connected between theVCC pin and GND

Pin 5 ndash ISENSE

Primary current sense

Pin 6 ndash VIN

Sense signal input representing the instantaneous rectified line voltage VIN is used for line regulation The internal impedanace is 20 kW and the scale factor is 00043 It also provides input undervoltage and overvoltage protection This pin also provides the supply current to the IC during start-up

92 Start-upPrior to start-up the VIN pin charges up the VCC capacitorthrough the diode between VINandVCC When VCC isfullycharged to a voltage higher than VCC(UVL) threshold then theVIN_SW turns on and the analog-to-digital converter begins to sense the input voltage The iW1692 commences soft-start function as soon as the voltage on VIN pin is above VINST

The iW1692 incorporates an internal soft-start function The soft-start time is set at 30 ms Once the VIN pin voltage has reached its turn-on threshold the iW1692 starts switching but limits the on-time to a percentage of the maximum on-time During the first 1 ms the on-time is limited to 25 During the next 1 ms the on-time is limited to 50 and during the last 1 ms the on-time is limited to 75

IfatanytimetheVCC voltage drops below VCC(UVL) threshold then all the digital logic is fully reset At this time the VIN_SWswitches off so that the VCC capacitor can be charged up again

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

Attheendofon-timethecurrenthasrampedupto

( ) ( )( ) g ON

gM

v t t ti t

Ltimes

=

(92)

This current represents a stored energy of

2( )2M

g gLE i t= times

(93)

WhenQ1turnsoffattOig(t)inLM forces a reversal of polarities on all windings Ignoring the communication-time caused by the leakage inductance LK at the instant of turn-off tO theprimarycurrenttransferstothesecondaryatanamplitudeof

( ) ( )Pd g

S

Ni t i tN

= times

(94)

Assuming the secondary winding is master the auxiliary winding is slave

VAUX 0V

VAUX = -VG xNAUX

NP

See equation 95

Figure 932 Auxiliary Voltage Waveforms

The auxiliary voltage is given by

( )AUXAUX O

S

NV V VN

= + ∆

(95)

and reflects the output voltage as shown in Figure 932

The voltage at the load differs from the secondary voltage by a diode drop and IR losses The diode drop is a function of current as are IR losses Thus if the secondary voltage is alwaysreadataconstantsecondarycurrentthedifferencebetween the output voltage and the secondary voltage is a fixed ΔV Furthermore if the voltage can be read when the secondarycurrentissmallΔV is small

The real-time waveform analyzer in the iW1692 reads this information cycle by cycle and then generates a feedback voltage VFB The VFB signal precisely represents the output voltage and is used to regulate the output voltage

94 Understanding CC and CV modeThe constant current mode (CC mode) is useful in battery charging applications During this mode of operation the iW1692 will regulate the output current at a constant maximum level regardless of the output voltage drop while avoiding continuous conduction mode

To achieve this regulation the iW1692 senses the load current indirectly through the primary current The primary currentisdetectedbytheISENSE pin through a resistor from the MOSFET source to ground (RSS) This resistor value is given by

( )( )2

CSS

OUTMAX

N KR

Itimes

=times

(96)

N is the ratio of primary turns to secondary turns of thetransformerandKC is 0264 V

95 Constant Voltage OperationAfter soft-start is completed the digital control block measures the output conditions If the output conditions do not warrant PFMmodeandtheISENSE signal is not consistently over 09 V then the device will operate in constant voltage mode

If no voltage is detected on VSENSE after 20 pulses it is assumed that the auxiliary winding of the transformer is either open or shorted and the iW1692 shuts down

As long as calculated TON for CV is less than the TONinCCthe IC operates in constant voltage mode

96 Constant Current OperationThe iW1692 has been designed to work in constant-current mode for battery charging applications If the output voltage drops but does not go below 20 of the nominal designed value the device operates in this mode

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

Out

put V

olta

ge

Output CurrentICC

VNOM

CV mode

CC

mod

e

Figure 961 Modes of operation

97 Variable Frequency ModeThe iW1692 is designed to operate in discontinuous conduction (DCM) mode at a fixed frequency of 40 kHz in both CC and CV modes To avoid operation in continuous conduction (CCM) mode the iW1692 checks for the falling edge of the VSENSE input on every cycle If a falling edge of VSENSE is not detected during the normal 25μs period the switching period is extended until the falling edge VSENSEdoes occur If the switching period reaches 75μs without VSENSE being detected the iW1692 immediately shuts off

98 PFM Mode at Light LoadThe iW1692 operates in a fixed frequency PWM mode whenIOUT is greater than approximately 5 of the specified maximum load current As the output load IOUTisreducedtheon-time tON is decreased At the moment that tONdropsbelow tON_MIN thecontroller transitions toPulseFrequencyModulation (PFM) mode Thereafter the on-time is modulated by the line voltage and the off-time is modulated by the load current The device automatically returns to PWM mode when the load current increases

99 Internal Loop CompensationThe iW1692 incorporates an internal Digital Error Amplifier with no requirement for external loop compensation The loop stability is guaranteed by design to provide at least 45 degrees of phase margin and ndash20dB of gain margin

910 Voltage Protection FunctionsThe iW1692 includes functions that protect against input and output overvoltage

The input voltage is monitored by the VINpinandtheoutputvoltage is monitored by the VSENSE pin If the voltage at these pins exceed their undervoltage or overvoltage thresholds for more than 6 cycles the iW1692 shuts-down immediately However the IC remains biased which discharges the VCCsupply Once VCC drops below the UVLO threshold the controller resets itself and then initiates a new soft-startcycle The controller continues attempting start-up but does not fully start-up until the fault condition is removed

The output voltage can be high enough to damage the output capacitor when the feedback loop is broken The iW1692 uses the primary feedback only with no secondary feedback loop When the VSENSE pin is shorted to GND (by shortingopen sense resistor) The controller will shut off with 6 consecutive pulses after start-up

911 Cable Drop CompensationThe iW1692-30 incorporates an innovative method to compensate for any IR drop in the secondary circuitry including cable and cable connector A 5 W AC adapter with 5 VDC output has 6 deviation at 1 A load current due to the drop across the DC cable without cable compensation The iW1692-30 cancels this error by providing a voltage offset to the feedback signal based on the amount of load current detected The iW1692-30 has 300mV of cable drop compensation at maximum current The iW1692-00 does not include any cable compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

100 Design Example101 Design ProcedureThis design example gives the procedure for a flyback converter using iW1692 Refer to figure 1201 for the application circuit The design objectives for this adapter are given in table 101 It meets UL IEC and CEC requirements

Figure 301 Design Flow Chart

Determine Cable Drop Compensation

Determine the Design Specifications(Vout Iout_max Vin_max Vin_min efficiency and ripple)

Determine Turns Ratio

Determine Input Bulk Capacitance

Determine Current Sensing Resistor

Determine Magnetizing Inductance

Determine Primary Turns

Determine Secondary Turns

Can you wind this transformer

Determine Bias Turns

Determine Vsense Turns and Resistors

Determine Output Capacitance

Determine Snubber Network

Determine Ton Delay Compensation

Finish

Yes No

Determine Rvin Resistors

Is the real cable drop compensation value OK Yes

No

Figure 1001 iW1692 Design Flow Chart

Parameter Symbol Range

Input Voltage VIN 85 - 264 VRMS

Frequency fIN 47 - 64 Hz

NoLoadInput PIN 200 mW

Output Voltage VOUT_CABLE 495 - 505 V

Output Current IOUT 1A

Output Ripple VRIPPLE lt100mV

Power Out POUT 5WCEC Efficiency h 65

Table 101 iW1692 Design Specification Table

102 Cable Drop CompensationCable Drop Compensation is an option included in the iW1692-30 This option helps maintain the output voltage at the end of the cable that the power supply is designed for During CV (constant voltage) mode the output current changes as the voltage remains constant This is true for the output voltage at the output of the power supply board however in certain applications the device to be charged is not directly connected to the power supply but ratheris connected via a cable This cable is seen by the power supply as a resistance So as the output current increases the output voltage at the end of the cable begins to drop With the cable compensation option the iW1692 can compensate for the resistance of the cable by incrementally increasing the output voltage seen on the power supply board to cancel out the selected cable resistance

To find the right cable compensation type for a given cable pickthecabledropcompensationnumberthatisclosesttothe voltage drop of the cable under maximum output current Use equation 101 for VOUTwhereVFD is the forward voltage oftheoutputdiode

_ _ _OUT OUT CABLE CABLE DROP COMPENSATION fdV V V V= + +(101)

Using equation 101 we know for this design VOUT is 55 V assuming no cable drop compensation is chosen and the forwarddropontheoutputdiode(VFD) is 500 mV

103 Input SelectionVIN resistors are chosen primarily to scale down the inputvoltage for the IC The scale factor for the input voltage in the IC is 00043 and the internal impedance of this pin is 20 kΩ Therefore the VINresistorsshouldequateto

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page0 82007

20 20 46300043Vin

kR k MW= minus W = W

(102)

From equation 102 ideally RVIN should be 463 MΩ because R1 R10 and R11 add up to approximately 46 MΩ By selecting the value of RVIN the VINTON_MAX_PFM and (VINTON)MAX_LIMITaredetermined

( ) _900 sec00043

2020

IN ON MAX LIMIT

Vin

VV Tk

R k

sdotmicrosdot = times

W + W

(103)

( ) 185 sec0004320

20

IN ON PFM

Vin

VV Tk

R k

sdotmicrosdot = times

W + W (104)

Keep in mind by changing RVIN to be something other than 463 MΩ the minimum and maximum input voltage for start-up will also change

Since the iW1692 uses the exact scaled value of VINforitscalculations C6 should be included to filter out any noise that mayappearontheVIN signal This is especially important for line-in surge conditions

104 Turns RatioThe maximum allowable turns ratio between the primary and secondary winding is determined by the minimum detectable reset time of the transformer during PFM mode

( )_

_

IN ON PFMtr MAX

RESET MIN OUT

V TN

T Vtimes

=times (105)

To avoid continuous conduction the turns ratio must be high enough so that TRESET does not exceed TPERIOD ndash TON ndash TDEAD TPERIOD is given by the PWM switching frequency of 40 kHz TRESET_MAX is given by

_ _RESET MAX PERIOD ON MAX DEADT T T T= minus minus (106)

Thus the minimum turns ratio is given by

( )_

_

IN ON MAXtr MIN

RESET MAX OUT

V TN

T Vtimes

=times

(107)

(VINtimesTON)PFM is limited by the iW1692 to be 185 Vμs and TRESET_MIN is required by the IC to be 23 μs

_185 sec 15

23 sec 55tr MAXVN

Vsdotmicro

= =micro times (108)

The product of VIN and TON is limited by equation 109 for CC limit performance For this example we choose 750 Vμsec

( )700 sec 850 secIN ON MAXV V T Vsdotmicro lt times lt sdotmicro (109)

Assuming TON_MAX is 97micros solving for the minimum turns ratio yields

_

_

25 sec 97 sec 48 sec

105 secRESET MAX

RESET MAX

T

T

= micro minus micro minus micro

= micro (1010)

( )_750 sec 13

105 sec 55tr MINVN

Vsdotmicro

= =micro times (1011)

Pickanumberbetween themaximumandminimumturnsratio in the example the turn ratio is 13 A turn ratio in the range of 11 to 15 is suggested for optimal performance

105 Input Bulk CapacitorThe input bulk capacitance (C1+C2) is chosen to maintain enough input power to sustain constant output power even as the input voltage is dropping In order for this to be true totalinputbulkcapacitancemustbe

_

_2

1 2 2 2_ _

arcsin2

2

(2 )

INDC MIN

INAC MIN

V

VIN

INAC MIN INDC MIN line

OUT OUTIN

PowerSupply

P

C CV V f

V IP

times

times times

π + =

minus times

times=

h

(1012)

VINAC_MIN is the minimum input voltage (rms) to be inputted intothepowersupplyandƒlineisthelowestlinefrequencyforthe power supply (in this case 47 Hz) VINDC_MINiscalculatedbasedonthe(VINtimes TON)MAX product

( )_

_

IN ON MAXINDC MIN

ON MAX

V TV

Ttimes

= (1013)

First we must find TON_MAX to get VINDC_MIN In order for the powersupplytofunctionindiscontinuousconductionmodeTON_MAX should be smaller than the switching period minus the transformer reset time Given that the transformer reset timeis

( )IN ON MAXRESET

tr OUT

V TT

N Vtimes

=sdot (1014)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

Then the maximum on-time must be

( )_

IN ON MAXON MAX PERIOD dead

tr OUT

V TT T T

N Vtimes

= minus minussdot (1015)

TDEAD is about 48 μsec Knowing TPERIOD has to be 25 μsec because of the 40 kHz switching frequency

_750 sec25 sec 48 sec 97 sec13 55ON MAX

VTV

sdotmicro= micro minus minus micro = micro

times (1016)

From this result we can now get VINDC_MIN from equation1013

( )_

_772IN ON MAX

INDC MINON MAX

V TV V

Ttimes

= = (1017)

Substituting VINDC_MIN into equation 1012 we get

( )

( ) ( )

7722 85

1 2 2 2

arcsin2 641

21206

(2 85 772 ) 47

VVW

C C FV V Hz

times times times π + = = micro

times minus times (1018)

Increase the value of C1+C2 to account for efficiency losses For this example 136 microF is chosen

106 Current Sense ResistorThe ISENSEresistordeterminesthemaximumcurrentoutputof the power supply The output current of the power supply isdeterminedby

1_2

RESETOUT tr PRI PK

PERIOD

TI N I

T= times times times

(1019)

When the maximum current output is achieved the voltage seenon the ISENSEpin (VISENSE) should reach its maximum Thus at constant current limit

__

Isense CCPRI PK

Isense

VI

R=

(1020)

Substituting this into equation 1019 gives

_2 OUT Isense PERIOD

Isense CCtr RESET

I R TV

N Ttimes times

= times (1021)

During constant current mode where output current is at its maximum the first term in Equation 1021 is constant Therefore we can call this KC Substituting this back into equation 1021 we get

_PERIOD

Isense CC CRESET

TV K

T= times

(1022)

For iW1692 KC is 0264 V therefore RIsensedependsonthemaximumoutputcurrentby

2tr C

Isense xOUT

N KR

Itimes

= times htimes (1023)

Using this equation and Ntr from section 104

13 0264 87 152 1Isense

VRA

times= times = W

times (1024)

We recommend using plusmn1 tolerance resistors for RIsense

107 Magenitizing InductanceA feature of the iW1692 is the lack of dependence on the magnetizing inductance for the CC curve

Although the constant current limit does not depend on the magnetizing inductance there are still restrictions on the magnetizing inductance The maximum LMislimitedbytheamountofpowerthatneedstocomeoutofthetransformerin order for the power supply to regulate This is given by

( )

( )

2

__

_

402

IN ON MAXM MAX

XFMR MAX

OUT fd OUTXFMR MAX

X

V T kHzL

P

V V IP

times times=

times

minus times=

h

(1025)

The minimum LM is limited by the maximum allowableprimarypeakcurrent(IPRI_PK) 09 V on the ISENSEpinshouldcorrespond to the maximum allowable primary peak current Therefore the maximum primary peak current is

_09

PRI PKIsense

VIR

lt (1026)

Thus LMislimitedby

( )_ 09

IN ON MAXM MIN

Isense

V TL

V Rsdot

= (1027)

There is also a lower limit on ISENSE signal of 02 V This gives a second maximum value on LM compare this with the value obtained from equation 1025 and pick the smaller of the two values

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page2 82007

( )

2_

_ 2

2

02 40XFMR MAX Isense

M MAXP R

LV kHz

times times=

times (1028)

Assuming that the efficiency of the transformer is about 87 wecanobtaintheamountofpowerthatneedstocomeoutofthetransformeras

5 57587 87OUTP W W= =

(1029)

Substituting this into equation 1025 we get

( )2

_750 sec 40

1962 575M MAX

V kHzL mH

Wsdotmicro times

= =times (1030)

To get the minimum value of the primary inductance use the value for RISENSE from equation 1024

_09 615PRI PK

VI Alt =W (1031)

Substituting this primary peak current into equation 1027

_750 sec 12509 15M MIN

VL mHV

sdotmicro= =

W (1032)

Choose a primary inductance somewhere between 191 mH and 142 mH we chose 15 mH

Given a nominal LM we can now find the minimum turns ratio between primary and secondary that ensures the powersupply does not function at variable frequency (VF) mode before a certain desired voltage VOUT_VF Under VF mode theconstantcurrentIOUTmaynotbeasaccurateasinpulsewidth modulation mode

( )

_

__

2OUT VF OUT M PERIOD

xtr VF

OUT VF RESET

V I L T

NV T

times times times times

h=

times

(1033)

and

( )

_

_

2

2

OUT SHUTDOWN OUT M PERIOD

xON

INAC MIN

V I L T

TV

times times times times

h = times

(1034)

108 Primary WindingInordertokeepthetransformerfromsaturationthemaximumflux density must not be exceeded Therefore the minimum primary winding on the transformer must meet

( )IN ON MAXPRI

MAX e

V TN

B Asdot

getimes (1035)

Where BMAX is maximum flux density and Ae is the corearea

Picking (VINtimesTON)MAX to be 750 Vμsec and getting the maximum flux density and core area from the transformer datasheetwecancalculate theminimumnumberof turnsfor the primary winding Substitute BMAX as 320mT and the area of the core to be 192 mm2 we solve equation 1035 to get

2750 sec 1221

320 192PRIVN turns

mT mmsdotmicro

ge =times (1036)

To avoid hitting the maximum flux density pick a value for NPRIto be higher than this In this example 144 turns is picked

109 Secondary WindingFrom the primary winding turns we obtain the secondary winding

PRI

SECtr

NNN

= (1037)

Thus in our example

144 1113SECN turns= =

(1038)

At this point it is advantageous to make sure the primary winding and secondary winding chosen is actually feasible to wind

1010 Bias WindingVCC is the supply to the iW1692 and should be between 12 V and 16 V Capacitor C7 stores the VCC charge during IC operation and the controller checks this voltage and makes sure itrsquos within range The zener Z1 protects the IC from getting a VCC over voltage Thus the number of auxiliary windings needs to ensure that VCC does not exceed 16 V

( )SEC CC fdBIAS

OUT

N V VN

V

times +=

(1039)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

The number of auxiliary windings can be calculated using equation 1039

11 125 2555BIAS

turns VN turnsVtimes

= = (1040)

Here wersquove actually chosen a lower number for the bias winding 22 turns

1011 VSENSE Resistors and WindingThe output voltage regulation is mainly determined by the feedback signal VSENSE

_SENSE OUT PCB SENSEV V K= times (1041)

Where

4

4 3

VsenseSENSE

SEC

NRKR R N

= times+ (1042)

InternallyVSENSE is compared to a reference voltage VFB_MAX WithnocabledropcompensationVFB_MAX is 1538 V however cabledropcompensationactually increases this referenceby a specific amount as a function of the present output current Therefore under perfect regulation conditions

_ _SENSE FB MAX CableDropCompensation actual SENSEV V V K= + sdot(1043)

Substitute this into equation 1043 and solve for KSENSE

_

_ _

FB MAXSENSE

OUT PCB CableDropComp actual

VK

V V=

minus (1044)

Up to this point cable drop compensation voltage is assumed to be independent of the power supply circuitry when inreality this is not completely true

VCableDropComp_actual is affected by output regulation magnetizing inductance and RVIN

220_ 120

_ _2

_

M OUT OUT

Vin sw PowerSupply

L V IkCableDropComp actual k R f

CABLE DROP COMP OUT

FB MAX

V K

V VK

V

sdot sdot sdotWW+ sdoth

= sdot + sdot

times times(1045)

Where K1 is -1155 Vμsec and K2 is 0130 V-1μsec-1 Plug this number into equation 1045 and then find VSENSE winding and resistor values to satisfy equation 1042

For this design example we did not implement cable drop compensationsoVCableDropComp_actual is 0 V The cable drop for a 18 m 22 AWG wire is

1 2 531 18 190mOUT Cable mI R A m mVWtimes = times times times =

(1046)

Substituting this into equation 1044 we get

1538 035 0SENSE

VKV V

= =minus (1047)

Solving for R4 in equation 1042 assuming R3 is 20 kΩ and NVSENSE is 24 turns we get R4 should be around 3 kΩ

Since the iW1692 uses the VSENSE signal to determine the regulation point this signal can not be too noisy Thus C8 is used to help filter the VSENSE signal

1012 Output CapacitorsThe output capacitors are important for controlling the output voltage ripple of the power supply This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus how much charge is lost before the next switching cycle is the ripple on the output voltage Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then

_ _

OUTOUT

OUT RIPPLE PK

QC

V=

(1048)

Since charge is equal to current times time

_

_ _

OUT OFF MAXOUT

OUT RIPPLE PK

I TC

Vtimes

= (1049)

The maximum off-time during maximum output current is 25 μs

Assuming we want to get under 50 mV of ripple on the output we substitute this into equation 1049 to get

1 25 sec 50050OUT

AC FmV

times micro= = micro

(1050)

In this calculations ESR and ESL are ignored the reason this calculation is still valid is because of the second stage LC filter L3 and C11 These two components reduce the ESR and ESL ripple

1013 Snubber NetworkThe snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive The goal is to dissipate the energy from the leakage inductance of the transformer For simplicity and a more conservative design first assume the energy of the

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

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110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

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MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

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130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

80 Functional Block Diagram

90 Theory of Operation

ndash

+

VIN

GND

PORPOR

VCC

2

VSENSE

VFB

VVMS

VIPK

VIN_A

OUTPUT

128 mV

1538 V

ndash

+CMP

ndash

+CMP

ISENSE5

10 V

02 V ~ 09 V

02 V ~ 20 V

IPEAK

ADC

VOCP

ndash

Start-up

DAC

ndash

+

1

6 4

DigitalLogic

Control

GateDriver

3

Figure 801 iW1692 Functional Block Diagram

The iW1692 is a digital controller which uses a new proprietary primary-side control technology to eliminate the opto-isolated feedback and secondary regulation circuits required in traditional designs This results in a low-cost solution for low power ACDC adapters The core PWM processor uses fixed-frequency Discontinuous Conduction Mode (DCM) operation at heavy load and switches to variable frequency operation at light loads to maximize efficiency Furthermore iWattrsquos digital control technology enables fast dynamic response tight output regulation and full featured circuit protection with primary-side control

Referring to the block diagram in Figure 801 the digital logic control generates the switching on-time and off-time information based on the line voltage and the output voltage feedback signal The system loop is internally compensated inside the digital logic control and no external analog components are required for loop compensation The iW1692 uses an advanced digital control algorithm to reduce system design time and improve reliability

Furthermoreaccuratesecondaryconstant-currentoperationis achieved without the need for any secondary-side sense and control circuits

The iW1692 uses PWM mode control at higher output power levels and switches to PFM mode at light load to minimize power dissipation Additional built-in protection features include overvoltage protection (OVP) output short circuit protection (SCP) AC low line brown out over current protection single pin fault protection and ISENSEfaultdetection

iWattrsquos digital control scheme is specifically designed to address the challenges and trade-offs of power conversion design This innovative technology is ideal for balancing new regulatory requirements for green mode operation with more practical design considerations such as lowest possible cost smallest size and high performance output control

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

VCC

VCC(ST)

OFF

ENABLE

Start-upSequencing

200micros

VIN VIN Impedance = 20k

VINSWON

Figure 921 Start-up Sequencing Diagram

93 Understanding Primary FeedbackFigure 931 illustrates a simplified flyback converter When the switch Q1 conducts during tONthecurrentigisdirectlydrawn from rectified sinusoid vg The energy Eg isstoredin the primary windingThe rectifying diode D1 is reverse biasedandtheloadcurrentIOissuppliedbythesecondarycapacitorCO When Q1 turns off D1 conducts and the stored energy Eg(t) is delivered to the output

+

vin(t)

TS(t)

IO

VO

NP NS

NAUX

D1

Q1

VAUX

COvg(t)

ig(t)+

ndash

iin(t) id(t)

Figure 931 Simplified Flyback Converter

In order to regulate the output voltage within a tight specification the information about the output voltage and load current needs to be accurately sensed In the DCM flyback converter this information can be read via the auxiliary winding or the primary magnetizing inductance (LM) During the Q1on-timetheloadcurrentissuppliedfromthe output filter capacitor CO The voltage across the primary winding is vg(t) assuming the voltage dropped across Q1iszero The current in Q1rampsuplinearlyatarateof

( ) ( )g g

M

di t v tdt L

=

(91)

91 Pin DetailPin 1 ndash VSENSE

Sense signal input from auxiliary winding This provides the secondary voltage feedback used for output regulation

Pin 2 ndash GND

Analog digital and power ground

Pin 3 ndash OUTPUT

Gate drive signal for the external power MOSFET switch

Pin 4 ndash VCC

Power supply for the controller during normal operation The controller starts up when VCC reaches 12 V (typical) and shuts-downwhentheVCC voltage is below 6 V (typical) A 100 nF decoupling capacitor should be connected between theVCC pin and GND

Pin 5 ndash ISENSE

Primary current sense

Pin 6 ndash VIN

Sense signal input representing the instantaneous rectified line voltage VIN is used for line regulation The internal impedanace is 20 kW and the scale factor is 00043 It also provides input undervoltage and overvoltage protection This pin also provides the supply current to the IC during start-up

92 Start-upPrior to start-up the VIN pin charges up the VCC capacitorthrough the diode between VINandVCC When VCC isfullycharged to a voltage higher than VCC(UVL) threshold then theVIN_SW turns on and the analog-to-digital converter begins to sense the input voltage The iW1692 commences soft-start function as soon as the voltage on VIN pin is above VINST

The iW1692 incorporates an internal soft-start function The soft-start time is set at 30 ms Once the VIN pin voltage has reached its turn-on threshold the iW1692 starts switching but limits the on-time to a percentage of the maximum on-time During the first 1 ms the on-time is limited to 25 During the next 1 ms the on-time is limited to 50 and during the last 1 ms the on-time is limited to 75

IfatanytimetheVCC voltage drops below VCC(UVL) threshold then all the digital logic is fully reset At this time the VIN_SWswitches off so that the VCC capacitor can be charged up again

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MK-4008-NR Page7 82007

Attheendofon-timethecurrenthasrampedupto

( ) ( )( ) g ON

gM

v t t ti t

Ltimes

=

(92)

This current represents a stored energy of

2( )2M

g gLE i t= times

(93)

WhenQ1turnsoffattOig(t)inLM forces a reversal of polarities on all windings Ignoring the communication-time caused by the leakage inductance LK at the instant of turn-off tO theprimarycurrenttransferstothesecondaryatanamplitudeof

( ) ( )Pd g

S

Ni t i tN

= times

(94)

Assuming the secondary winding is master the auxiliary winding is slave

VAUX 0V

VAUX = -VG xNAUX

NP

See equation 95

Figure 932 Auxiliary Voltage Waveforms

The auxiliary voltage is given by

( )AUXAUX O

S

NV V VN

= + ∆

(95)

and reflects the output voltage as shown in Figure 932

The voltage at the load differs from the secondary voltage by a diode drop and IR losses The diode drop is a function of current as are IR losses Thus if the secondary voltage is alwaysreadataconstantsecondarycurrentthedifferencebetween the output voltage and the secondary voltage is a fixed ΔV Furthermore if the voltage can be read when the secondarycurrentissmallΔV is small

The real-time waveform analyzer in the iW1692 reads this information cycle by cycle and then generates a feedback voltage VFB The VFB signal precisely represents the output voltage and is used to regulate the output voltage

94 Understanding CC and CV modeThe constant current mode (CC mode) is useful in battery charging applications During this mode of operation the iW1692 will regulate the output current at a constant maximum level regardless of the output voltage drop while avoiding continuous conduction mode

To achieve this regulation the iW1692 senses the load current indirectly through the primary current The primary currentisdetectedbytheISENSE pin through a resistor from the MOSFET source to ground (RSS) This resistor value is given by

( )( )2

CSS

OUTMAX

N KR

Itimes

=times

(96)

N is the ratio of primary turns to secondary turns of thetransformerandKC is 0264 V

95 Constant Voltage OperationAfter soft-start is completed the digital control block measures the output conditions If the output conditions do not warrant PFMmodeandtheISENSE signal is not consistently over 09 V then the device will operate in constant voltage mode

If no voltage is detected on VSENSE after 20 pulses it is assumed that the auxiliary winding of the transformer is either open or shorted and the iW1692 shuts down

As long as calculated TON for CV is less than the TONinCCthe IC operates in constant voltage mode

96 Constant Current OperationThe iW1692 has been designed to work in constant-current mode for battery charging applications If the output voltage drops but does not go below 20 of the nominal designed value the device operates in this mode

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

Out

put V

olta

ge

Output CurrentICC

VNOM

CV mode

CC

mod

e

Figure 961 Modes of operation

97 Variable Frequency ModeThe iW1692 is designed to operate in discontinuous conduction (DCM) mode at a fixed frequency of 40 kHz in both CC and CV modes To avoid operation in continuous conduction (CCM) mode the iW1692 checks for the falling edge of the VSENSE input on every cycle If a falling edge of VSENSE is not detected during the normal 25μs period the switching period is extended until the falling edge VSENSEdoes occur If the switching period reaches 75μs without VSENSE being detected the iW1692 immediately shuts off

98 PFM Mode at Light LoadThe iW1692 operates in a fixed frequency PWM mode whenIOUT is greater than approximately 5 of the specified maximum load current As the output load IOUTisreducedtheon-time tON is decreased At the moment that tONdropsbelow tON_MIN thecontroller transitions toPulseFrequencyModulation (PFM) mode Thereafter the on-time is modulated by the line voltage and the off-time is modulated by the load current The device automatically returns to PWM mode when the load current increases

99 Internal Loop CompensationThe iW1692 incorporates an internal Digital Error Amplifier with no requirement for external loop compensation The loop stability is guaranteed by design to provide at least 45 degrees of phase margin and ndash20dB of gain margin

910 Voltage Protection FunctionsThe iW1692 includes functions that protect against input and output overvoltage

The input voltage is monitored by the VINpinandtheoutputvoltage is monitored by the VSENSE pin If the voltage at these pins exceed their undervoltage or overvoltage thresholds for more than 6 cycles the iW1692 shuts-down immediately However the IC remains biased which discharges the VCCsupply Once VCC drops below the UVLO threshold the controller resets itself and then initiates a new soft-startcycle The controller continues attempting start-up but does not fully start-up until the fault condition is removed

The output voltage can be high enough to damage the output capacitor when the feedback loop is broken The iW1692 uses the primary feedback only with no secondary feedback loop When the VSENSE pin is shorted to GND (by shortingopen sense resistor) The controller will shut off with 6 consecutive pulses after start-up

911 Cable Drop CompensationThe iW1692-30 incorporates an innovative method to compensate for any IR drop in the secondary circuitry including cable and cable connector A 5 W AC adapter with 5 VDC output has 6 deviation at 1 A load current due to the drop across the DC cable without cable compensation The iW1692-30 cancels this error by providing a voltage offset to the feedback signal based on the amount of load current detected The iW1692-30 has 300mV of cable drop compensation at maximum current The iW1692-00 does not include any cable compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

100 Design Example101 Design ProcedureThis design example gives the procedure for a flyback converter using iW1692 Refer to figure 1201 for the application circuit The design objectives for this adapter are given in table 101 It meets UL IEC and CEC requirements

Figure 301 Design Flow Chart

Determine Cable Drop Compensation

Determine the Design Specifications(Vout Iout_max Vin_max Vin_min efficiency and ripple)

Determine Turns Ratio

Determine Input Bulk Capacitance

Determine Current Sensing Resistor

Determine Magnetizing Inductance

Determine Primary Turns

Determine Secondary Turns

Can you wind this transformer

Determine Bias Turns

Determine Vsense Turns and Resistors

Determine Output Capacitance

Determine Snubber Network

Determine Ton Delay Compensation

Finish

Yes No

Determine Rvin Resistors

Is the real cable drop compensation value OK Yes

No

Figure 1001 iW1692 Design Flow Chart

Parameter Symbol Range

Input Voltage VIN 85 - 264 VRMS

Frequency fIN 47 - 64 Hz

NoLoadInput PIN 200 mW

Output Voltage VOUT_CABLE 495 - 505 V

Output Current IOUT 1A

Output Ripple VRIPPLE lt100mV

Power Out POUT 5WCEC Efficiency h 65

Table 101 iW1692 Design Specification Table

102 Cable Drop CompensationCable Drop Compensation is an option included in the iW1692-30 This option helps maintain the output voltage at the end of the cable that the power supply is designed for During CV (constant voltage) mode the output current changes as the voltage remains constant This is true for the output voltage at the output of the power supply board however in certain applications the device to be charged is not directly connected to the power supply but ratheris connected via a cable This cable is seen by the power supply as a resistance So as the output current increases the output voltage at the end of the cable begins to drop With the cable compensation option the iW1692 can compensate for the resistance of the cable by incrementally increasing the output voltage seen on the power supply board to cancel out the selected cable resistance

To find the right cable compensation type for a given cable pickthecabledropcompensationnumberthatisclosesttothe voltage drop of the cable under maximum output current Use equation 101 for VOUTwhereVFD is the forward voltage oftheoutputdiode

_ _ _OUT OUT CABLE CABLE DROP COMPENSATION fdV V V V= + +(101)

Using equation 101 we know for this design VOUT is 55 V assuming no cable drop compensation is chosen and the forwarddropontheoutputdiode(VFD) is 500 mV

103 Input SelectionVIN resistors are chosen primarily to scale down the inputvoltage for the IC The scale factor for the input voltage in the IC is 00043 and the internal impedance of this pin is 20 kΩ Therefore the VINresistorsshouldequateto

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page0 82007

20 20 46300043Vin

kR k MW= minus W = W

(102)

From equation 102 ideally RVIN should be 463 MΩ because R1 R10 and R11 add up to approximately 46 MΩ By selecting the value of RVIN the VINTON_MAX_PFM and (VINTON)MAX_LIMITaredetermined

( ) _900 sec00043

2020

IN ON MAX LIMIT

Vin

VV Tk

R k

sdotmicrosdot = times

W + W

(103)

( ) 185 sec0004320

20

IN ON PFM

Vin

VV Tk

R k

sdotmicrosdot = times

W + W (104)

Keep in mind by changing RVIN to be something other than 463 MΩ the minimum and maximum input voltage for start-up will also change

Since the iW1692 uses the exact scaled value of VINforitscalculations C6 should be included to filter out any noise that mayappearontheVIN signal This is especially important for line-in surge conditions

104 Turns RatioThe maximum allowable turns ratio between the primary and secondary winding is determined by the minimum detectable reset time of the transformer during PFM mode

( )_

_

IN ON PFMtr MAX

RESET MIN OUT

V TN

T Vtimes

=times (105)

To avoid continuous conduction the turns ratio must be high enough so that TRESET does not exceed TPERIOD ndash TON ndash TDEAD TPERIOD is given by the PWM switching frequency of 40 kHz TRESET_MAX is given by

_ _RESET MAX PERIOD ON MAX DEADT T T T= minus minus (106)

Thus the minimum turns ratio is given by

( )_

_

IN ON MAXtr MIN

RESET MAX OUT

V TN

T Vtimes

=times

(107)

(VINtimesTON)PFM is limited by the iW1692 to be 185 Vμs and TRESET_MIN is required by the IC to be 23 μs

_185 sec 15

23 sec 55tr MAXVN

Vsdotmicro

= =micro times (108)

The product of VIN and TON is limited by equation 109 for CC limit performance For this example we choose 750 Vμsec

( )700 sec 850 secIN ON MAXV V T Vsdotmicro lt times lt sdotmicro (109)

Assuming TON_MAX is 97micros solving for the minimum turns ratio yields

_

_

25 sec 97 sec 48 sec

105 secRESET MAX

RESET MAX

T

T

= micro minus micro minus micro

= micro (1010)

( )_750 sec 13

105 sec 55tr MINVN

Vsdotmicro

= =micro times (1011)

Pickanumberbetween themaximumandminimumturnsratio in the example the turn ratio is 13 A turn ratio in the range of 11 to 15 is suggested for optimal performance

105 Input Bulk CapacitorThe input bulk capacitance (C1+C2) is chosen to maintain enough input power to sustain constant output power even as the input voltage is dropping In order for this to be true totalinputbulkcapacitancemustbe

_

_2

1 2 2 2_ _

arcsin2

2

(2 )

INDC MIN

INAC MIN

V

VIN

INAC MIN INDC MIN line

OUT OUTIN

PowerSupply

P

C CV V f

V IP

times

times times

π + =

minus times

times=

h

(1012)

VINAC_MIN is the minimum input voltage (rms) to be inputted intothepowersupplyandƒlineisthelowestlinefrequencyforthe power supply (in this case 47 Hz) VINDC_MINiscalculatedbasedonthe(VINtimes TON)MAX product

( )_

_

IN ON MAXINDC MIN

ON MAX

V TV

Ttimes

= (1013)

First we must find TON_MAX to get VINDC_MIN In order for the powersupplytofunctionindiscontinuousconductionmodeTON_MAX should be smaller than the switching period minus the transformer reset time Given that the transformer reset timeis

( )IN ON MAXRESET

tr OUT

V TT

N Vtimes

=sdot (1014)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

Then the maximum on-time must be

( )_

IN ON MAXON MAX PERIOD dead

tr OUT

V TT T T

N Vtimes

= minus minussdot (1015)

TDEAD is about 48 μsec Knowing TPERIOD has to be 25 μsec because of the 40 kHz switching frequency

_750 sec25 sec 48 sec 97 sec13 55ON MAX

VTV

sdotmicro= micro minus minus micro = micro

times (1016)

From this result we can now get VINDC_MIN from equation1013

( )_

_772IN ON MAX

INDC MINON MAX

V TV V

Ttimes

= = (1017)

Substituting VINDC_MIN into equation 1012 we get

( )

( ) ( )

7722 85

1 2 2 2

arcsin2 641

21206

(2 85 772 ) 47

VVW

C C FV V Hz

times times times π + = = micro

times minus times (1018)

Increase the value of C1+C2 to account for efficiency losses For this example 136 microF is chosen

106 Current Sense ResistorThe ISENSEresistordeterminesthemaximumcurrentoutputof the power supply The output current of the power supply isdeterminedby

1_2

RESETOUT tr PRI PK

PERIOD

TI N I

T= times times times

(1019)

When the maximum current output is achieved the voltage seenon the ISENSEpin (VISENSE) should reach its maximum Thus at constant current limit

__

Isense CCPRI PK

Isense

VI

R=

(1020)

Substituting this into equation 1019 gives

_2 OUT Isense PERIOD

Isense CCtr RESET

I R TV

N Ttimes times

= times (1021)

During constant current mode where output current is at its maximum the first term in Equation 1021 is constant Therefore we can call this KC Substituting this back into equation 1021 we get

_PERIOD

Isense CC CRESET

TV K

T= times

(1022)

For iW1692 KC is 0264 V therefore RIsensedependsonthemaximumoutputcurrentby

2tr C

Isense xOUT

N KR

Itimes

= times htimes (1023)

Using this equation and Ntr from section 104

13 0264 87 152 1Isense

VRA

times= times = W

times (1024)

We recommend using plusmn1 tolerance resistors for RIsense

107 Magenitizing InductanceA feature of the iW1692 is the lack of dependence on the magnetizing inductance for the CC curve

Although the constant current limit does not depend on the magnetizing inductance there are still restrictions on the magnetizing inductance The maximum LMislimitedbytheamountofpowerthatneedstocomeoutofthetransformerin order for the power supply to regulate This is given by

( )

( )

2

__

_

402

IN ON MAXM MAX

XFMR MAX

OUT fd OUTXFMR MAX

X

V T kHzL

P

V V IP

times times=

times

minus times=

h

(1025)

The minimum LM is limited by the maximum allowableprimarypeakcurrent(IPRI_PK) 09 V on the ISENSEpinshouldcorrespond to the maximum allowable primary peak current Therefore the maximum primary peak current is

_09

PRI PKIsense

VIR

lt (1026)

Thus LMislimitedby

( )_ 09

IN ON MAXM MIN

Isense

V TL

V Rsdot

= (1027)

There is also a lower limit on ISENSE signal of 02 V This gives a second maximum value on LM compare this with the value obtained from equation 1025 and pick the smaller of the two values

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page2 82007

( )

2_

_ 2

2

02 40XFMR MAX Isense

M MAXP R

LV kHz

times times=

times (1028)

Assuming that the efficiency of the transformer is about 87 wecanobtaintheamountofpowerthatneedstocomeoutofthetransformeras

5 57587 87OUTP W W= =

(1029)

Substituting this into equation 1025 we get

( )2

_750 sec 40

1962 575M MAX

V kHzL mH

Wsdotmicro times

= =times (1030)

To get the minimum value of the primary inductance use the value for RISENSE from equation 1024

_09 615PRI PK

VI Alt =W (1031)

Substituting this primary peak current into equation 1027

_750 sec 12509 15M MIN

VL mHV

sdotmicro= =

W (1032)

Choose a primary inductance somewhere between 191 mH and 142 mH we chose 15 mH

Given a nominal LM we can now find the minimum turns ratio between primary and secondary that ensures the powersupply does not function at variable frequency (VF) mode before a certain desired voltage VOUT_VF Under VF mode theconstantcurrentIOUTmaynotbeasaccurateasinpulsewidth modulation mode

( )

_

__

2OUT VF OUT M PERIOD

xtr VF

OUT VF RESET

V I L T

NV T

times times times times

h=

times

(1033)

and

( )

_

_

2

2

OUT SHUTDOWN OUT M PERIOD

xON

INAC MIN

V I L T

TV

times times times times

h = times

(1034)

108 Primary WindingInordertokeepthetransformerfromsaturationthemaximumflux density must not be exceeded Therefore the minimum primary winding on the transformer must meet

( )IN ON MAXPRI

MAX e

V TN

B Asdot

getimes (1035)

Where BMAX is maximum flux density and Ae is the corearea

Picking (VINtimesTON)MAX to be 750 Vμsec and getting the maximum flux density and core area from the transformer datasheetwecancalculate theminimumnumberof turnsfor the primary winding Substitute BMAX as 320mT and the area of the core to be 192 mm2 we solve equation 1035 to get

2750 sec 1221

320 192PRIVN turns

mT mmsdotmicro

ge =times (1036)

To avoid hitting the maximum flux density pick a value for NPRIto be higher than this In this example 144 turns is picked

109 Secondary WindingFrom the primary winding turns we obtain the secondary winding

PRI

SECtr

NNN

= (1037)

Thus in our example

144 1113SECN turns= =

(1038)

At this point it is advantageous to make sure the primary winding and secondary winding chosen is actually feasible to wind

1010 Bias WindingVCC is the supply to the iW1692 and should be between 12 V and 16 V Capacitor C7 stores the VCC charge during IC operation and the controller checks this voltage and makes sure itrsquos within range The zener Z1 protects the IC from getting a VCC over voltage Thus the number of auxiliary windings needs to ensure that VCC does not exceed 16 V

( )SEC CC fdBIAS

OUT

N V VN

V

times +=

(1039)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

The number of auxiliary windings can be calculated using equation 1039

11 125 2555BIAS

turns VN turnsVtimes

= = (1040)

Here wersquove actually chosen a lower number for the bias winding 22 turns

1011 VSENSE Resistors and WindingThe output voltage regulation is mainly determined by the feedback signal VSENSE

_SENSE OUT PCB SENSEV V K= times (1041)

Where

4

4 3

VsenseSENSE

SEC

NRKR R N

= times+ (1042)

InternallyVSENSE is compared to a reference voltage VFB_MAX WithnocabledropcompensationVFB_MAX is 1538 V however cabledropcompensationactually increases this referenceby a specific amount as a function of the present output current Therefore under perfect regulation conditions

_ _SENSE FB MAX CableDropCompensation actual SENSEV V V K= + sdot(1043)

Substitute this into equation 1043 and solve for KSENSE

_

_ _

FB MAXSENSE

OUT PCB CableDropComp actual

VK

V V=

minus (1044)

Up to this point cable drop compensation voltage is assumed to be independent of the power supply circuitry when inreality this is not completely true

VCableDropComp_actual is affected by output regulation magnetizing inductance and RVIN

220_ 120

_ _2

_

M OUT OUT

Vin sw PowerSupply

L V IkCableDropComp actual k R f

CABLE DROP COMP OUT

FB MAX

V K

V VK

V

sdot sdot sdotWW+ sdoth

= sdot + sdot

times times(1045)

Where K1 is -1155 Vμsec and K2 is 0130 V-1μsec-1 Plug this number into equation 1045 and then find VSENSE winding and resistor values to satisfy equation 1042

For this design example we did not implement cable drop compensationsoVCableDropComp_actual is 0 V The cable drop for a 18 m 22 AWG wire is

1 2 531 18 190mOUT Cable mI R A m mVWtimes = times times times =

(1046)

Substituting this into equation 1044 we get

1538 035 0SENSE

VKV V

= =minus (1047)

Solving for R4 in equation 1042 assuming R3 is 20 kΩ and NVSENSE is 24 turns we get R4 should be around 3 kΩ

Since the iW1692 uses the VSENSE signal to determine the regulation point this signal can not be too noisy Thus C8 is used to help filter the VSENSE signal

1012 Output CapacitorsThe output capacitors are important for controlling the output voltage ripple of the power supply This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus how much charge is lost before the next switching cycle is the ripple on the output voltage Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then

_ _

OUTOUT

OUT RIPPLE PK

QC

V=

(1048)

Since charge is equal to current times time

_

_ _

OUT OFF MAXOUT

OUT RIPPLE PK

I TC

Vtimes

= (1049)

The maximum off-time during maximum output current is 25 μs

Assuming we want to get under 50 mV of ripple on the output we substitute this into equation 1049 to get

1 25 sec 50050OUT

AC FmV

times micro= = micro

(1050)

In this calculations ESR and ESL are ignored the reason this calculation is still valid is because of the second stage LC filter L3 and C11 These two components reduce the ESR and ESL ripple

1013 Snubber NetworkThe snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive The goal is to dissipate the energy from the leakage inductance of the transformer For simplicity and a more conservative design first assume the energy of the

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

VCC

VCC(ST)

OFF

ENABLE

Start-upSequencing

200micros

VIN VIN Impedance = 20k

VINSWON

Figure 921 Start-up Sequencing Diagram

93 Understanding Primary FeedbackFigure 931 illustrates a simplified flyback converter When the switch Q1 conducts during tONthecurrentigisdirectlydrawn from rectified sinusoid vg The energy Eg isstoredin the primary windingThe rectifying diode D1 is reverse biasedandtheloadcurrentIOissuppliedbythesecondarycapacitorCO When Q1 turns off D1 conducts and the stored energy Eg(t) is delivered to the output

+

vin(t)

TS(t)

IO

VO

NP NS

NAUX

D1

Q1

VAUX

COvg(t)

ig(t)+

ndash

iin(t) id(t)

Figure 931 Simplified Flyback Converter

In order to regulate the output voltage within a tight specification the information about the output voltage and load current needs to be accurately sensed In the DCM flyback converter this information can be read via the auxiliary winding or the primary magnetizing inductance (LM) During the Q1on-timetheloadcurrentissuppliedfromthe output filter capacitor CO The voltage across the primary winding is vg(t) assuming the voltage dropped across Q1iszero The current in Q1rampsuplinearlyatarateof

( ) ( )g g

M

di t v tdt L

=

(91)

91 Pin DetailPin 1 ndash VSENSE

Sense signal input from auxiliary winding This provides the secondary voltage feedback used for output regulation

Pin 2 ndash GND

Analog digital and power ground

Pin 3 ndash OUTPUT

Gate drive signal for the external power MOSFET switch

Pin 4 ndash VCC

Power supply for the controller during normal operation The controller starts up when VCC reaches 12 V (typical) and shuts-downwhentheVCC voltage is below 6 V (typical) A 100 nF decoupling capacitor should be connected between theVCC pin and GND

Pin 5 ndash ISENSE

Primary current sense

Pin 6 ndash VIN

Sense signal input representing the instantaneous rectified line voltage VIN is used for line regulation The internal impedanace is 20 kW and the scale factor is 00043 It also provides input undervoltage and overvoltage protection This pin also provides the supply current to the IC during start-up

92 Start-upPrior to start-up the VIN pin charges up the VCC capacitorthrough the diode between VINandVCC When VCC isfullycharged to a voltage higher than VCC(UVL) threshold then theVIN_SW turns on and the analog-to-digital converter begins to sense the input voltage The iW1692 commences soft-start function as soon as the voltage on VIN pin is above VINST

The iW1692 incorporates an internal soft-start function The soft-start time is set at 30 ms Once the VIN pin voltage has reached its turn-on threshold the iW1692 starts switching but limits the on-time to a percentage of the maximum on-time During the first 1 ms the on-time is limited to 25 During the next 1 ms the on-time is limited to 50 and during the last 1 ms the on-time is limited to 75

IfatanytimetheVCC voltage drops below VCC(UVL) threshold then all the digital logic is fully reset At this time the VIN_SWswitches off so that the VCC capacitor can be charged up again

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

Attheendofon-timethecurrenthasrampedupto

( ) ( )( ) g ON

gM

v t t ti t

Ltimes

=

(92)

This current represents a stored energy of

2( )2M

g gLE i t= times

(93)

WhenQ1turnsoffattOig(t)inLM forces a reversal of polarities on all windings Ignoring the communication-time caused by the leakage inductance LK at the instant of turn-off tO theprimarycurrenttransferstothesecondaryatanamplitudeof

( ) ( )Pd g

S

Ni t i tN

= times

(94)

Assuming the secondary winding is master the auxiliary winding is slave

VAUX 0V

VAUX = -VG xNAUX

NP

See equation 95

Figure 932 Auxiliary Voltage Waveforms

The auxiliary voltage is given by

( )AUXAUX O

S

NV V VN

= + ∆

(95)

and reflects the output voltage as shown in Figure 932

The voltage at the load differs from the secondary voltage by a diode drop and IR losses The diode drop is a function of current as are IR losses Thus if the secondary voltage is alwaysreadataconstantsecondarycurrentthedifferencebetween the output voltage and the secondary voltage is a fixed ΔV Furthermore if the voltage can be read when the secondarycurrentissmallΔV is small

The real-time waveform analyzer in the iW1692 reads this information cycle by cycle and then generates a feedback voltage VFB The VFB signal precisely represents the output voltage and is used to regulate the output voltage

94 Understanding CC and CV modeThe constant current mode (CC mode) is useful in battery charging applications During this mode of operation the iW1692 will regulate the output current at a constant maximum level regardless of the output voltage drop while avoiding continuous conduction mode

To achieve this regulation the iW1692 senses the load current indirectly through the primary current The primary currentisdetectedbytheISENSE pin through a resistor from the MOSFET source to ground (RSS) This resistor value is given by

( )( )2

CSS

OUTMAX

N KR

Itimes

=times

(96)

N is the ratio of primary turns to secondary turns of thetransformerandKC is 0264 V

95 Constant Voltage OperationAfter soft-start is completed the digital control block measures the output conditions If the output conditions do not warrant PFMmodeandtheISENSE signal is not consistently over 09 V then the device will operate in constant voltage mode

If no voltage is detected on VSENSE after 20 pulses it is assumed that the auxiliary winding of the transformer is either open or shorted and the iW1692 shuts down

As long as calculated TON for CV is less than the TONinCCthe IC operates in constant voltage mode

96 Constant Current OperationThe iW1692 has been designed to work in constant-current mode for battery charging applications If the output voltage drops but does not go below 20 of the nominal designed value the device operates in this mode

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

Out

put V

olta

ge

Output CurrentICC

VNOM

CV mode

CC

mod

e

Figure 961 Modes of operation

97 Variable Frequency ModeThe iW1692 is designed to operate in discontinuous conduction (DCM) mode at a fixed frequency of 40 kHz in both CC and CV modes To avoid operation in continuous conduction (CCM) mode the iW1692 checks for the falling edge of the VSENSE input on every cycle If a falling edge of VSENSE is not detected during the normal 25μs period the switching period is extended until the falling edge VSENSEdoes occur If the switching period reaches 75μs without VSENSE being detected the iW1692 immediately shuts off

98 PFM Mode at Light LoadThe iW1692 operates in a fixed frequency PWM mode whenIOUT is greater than approximately 5 of the specified maximum load current As the output load IOUTisreducedtheon-time tON is decreased At the moment that tONdropsbelow tON_MIN thecontroller transitions toPulseFrequencyModulation (PFM) mode Thereafter the on-time is modulated by the line voltage and the off-time is modulated by the load current The device automatically returns to PWM mode when the load current increases

99 Internal Loop CompensationThe iW1692 incorporates an internal Digital Error Amplifier with no requirement for external loop compensation The loop stability is guaranteed by design to provide at least 45 degrees of phase margin and ndash20dB of gain margin

910 Voltage Protection FunctionsThe iW1692 includes functions that protect against input and output overvoltage

The input voltage is monitored by the VINpinandtheoutputvoltage is monitored by the VSENSE pin If the voltage at these pins exceed their undervoltage or overvoltage thresholds for more than 6 cycles the iW1692 shuts-down immediately However the IC remains biased which discharges the VCCsupply Once VCC drops below the UVLO threshold the controller resets itself and then initiates a new soft-startcycle The controller continues attempting start-up but does not fully start-up until the fault condition is removed

The output voltage can be high enough to damage the output capacitor when the feedback loop is broken The iW1692 uses the primary feedback only with no secondary feedback loop When the VSENSE pin is shorted to GND (by shortingopen sense resistor) The controller will shut off with 6 consecutive pulses after start-up

911 Cable Drop CompensationThe iW1692-30 incorporates an innovative method to compensate for any IR drop in the secondary circuitry including cable and cable connector A 5 W AC adapter with 5 VDC output has 6 deviation at 1 A load current due to the drop across the DC cable without cable compensation The iW1692-30 cancels this error by providing a voltage offset to the feedback signal based on the amount of load current detected The iW1692-30 has 300mV of cable drop compensation at maximum current The iW1692-00 does not include any cable compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

100 Design Example101 Design ProcedureThis design example gives the procedure for a flyback converter using iW1692 Refer to figure 1201 for the application circuit The design objectives for this adapter are given in table 101 It meets UL IEC and CEC requirements

Figure 301 Design Flow Chart

Determine Cable Drop Compensation

Determine the Design Specifications(Vout Iout_max Vin_max Vin_min efficiency and ripple)

Determine Turns Ratio

Determine Input Bulk Capacitance

Determine Current Sensing Resistor

Determine Magnetizing Inductance

Determine Primary Turns

Determine Secondary Turns

Can you wind this transformer

Determine Bias Turns

Determine Vsense Turns and Resistors

Determine Output Capacitance

Determine Snubber Network

Determine Ton Delay Compensation

Finish

Yes No

Determine Rvin Resistors

Is the real cable drop compensation value OK Yes

No

Figure 1001 iW1692 Design Flow Chart

Parameter Symbol Range

Input Voltage VIN 85 - 264 VRMS

Frequency fIN 47 - 64 Hz

NoLoadInput PIN 200 mW

Output Voltage VOUT_CABLE 495 - 505 V

Output Current IOUT 1A

Output Ripple VRIPPLE lt100mV

Power Out POUT 5WCEC Efficiency h 65

Table 101 iW1692 Design Specification Table

102 Cable Drop CompensationCable Drop Compensation is an option included in the iW1692-30 This option helps maintain the output voltage at the end of the cable that the power supply is designed for During CV (constant voltage) mode the output current changes as the voltage remains constant This is true for the output voltage at the output of the power supply board however in certain applications the device to be charged is not directly connected to the power supply but ratheris connected via a cable This cable is seen by the power supply as a resistance So as the output current increases the output voltage at the end of the cable begins to drop With the cable compensation option the iW1692 can compensate for the resistance of the cable by incrementally increasing the output voltage seen on the power supply board to cancel out the selected cable resistance

To find the right cable compensation type for a given cable pickthecabledropcompensationnumberthatisclosesttothe voltage drop of the cable under maximum output current Use equation 101 for VOUTwhereVFD is the forward voltage oftheoutputdiode

_ _ _OUT OUT CABLE CABLE DROP COMPENSATION fdV V V V= + +(101)

Using equation 101 we know for this design VOUT is 55 V assuming no cable drop compensation is chosen and the forwarddropontheoutputdiode(VFD) is 500 mV

103 Input SelectionVIN resistors are chosen primarily to scale down the inputvoltage for the IC The scale factor for the input voltage in the IC is 00043 and the internal impedance of this pin is 20 kΩ Therefore the VINresistorsshouldequateto

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page0 82007

20 20 46300043Vin

kR k MW= minus W = W

(102)

From equation 102 ideally RVIN should be 463 MΩ because R1 R10 and R11 add up to approximately 46 MΩ By selecting the value of RVIN the VINTON_MAX_PFM and (VINTON)MAX_LIMITaredetermined

( ) _900 sec00043

2020

IN ON MAX LIMIT

Vin

VV Tk

R k

sdotmicrosdot = times

W + W

(103)

( ) 185 sec0004320

20

IN ON PFM

Vin

VV Tk

R k

sdotmicrosdot = times

W + W (104)

Keep in mind by changing RVIN to be something other than 463 MΩ the minimum and maximum input voltage for start-up will also change

Since the iW1692 uses the exact scaled value of VINforitscalculations C6 should be included to filter out any noise that mayappearontheVIN signal This is especially important for line-in surge conditions

104 Turns RatioThe maximum allowable turns ratio between the primary and secondary winding is determined by the minimum detectable reset time of the transformer during PFM mode

( )_

_

IN ON PFMtr MAX

RESET MIN OUT

V TN

T Vtimes

=times (105)

To avoid continuous conduction the turns ratio must be high enough so that TRESET does not exceed TPERIOD ndash TON ndash TDEAD TPERIOD is given by the PWM switching frequency of 40 kHz TRESET_MAX is given by

_ _RESET MAX PERIOD ON MAX DEADT T T T= minus minus (106)

Thus the minimum turns ratio is given by

( )_

_

IN ON MAXtr MIN

RESET MAX OUT

V TN

T Vtimes

=times

(107)

(VINtimesTON)PFM is limited by the iW1692 to be 185 Vμs and TRESET_MIN is required by the IC to be 23 μs

_185 sec 15

23 sec 55tr MAXVN

Vsdotmicro

= =micro times (108)

The product of VIN and TON is limited by equation 109 for CC limit performance For this example we choose 750 Vμsec

( )700 sec 850 secIN ON MAXV V T Vsdotmicro lt times lt sdotmicro (109)

Assuming TON_MAX is 97micros solving for the minimum turns ratio yields

_

_

25 sec 97 sec 48 sec

105 secRESET MAX

RESET MAX

T

T

= micro minus micro minus micro

= micro (1010)

( )_750 sec 13

105 sec 55tr MINVN

Vsdotmicro

= =micro times (1011)

Pickanumberbetween themaximumandminimumturnsratio in the example the turn ratio is 13 A turn ratio in the range of 11 to 15 is suggested for optimal performance

105 Input Bulk CapacitorThe input bulk capacitance (C1+C2) is chosen to maintain enough input power to sustain constant output power even as the input voltage is dropping In order for this to be true totalinputbulkcapacitancemustbe

_

_2

1 2 2 2_ _

arcsin2

2

(2 )

INDC MIN

INAC MIN

V

VIN

INAC MIN INDC MIN line

OUT OUTIN

PowerSupply

P

C CV V f

V IP

times

times times

π + =

minus times

times=

h

(1012)

VINAC_MIN is the minimum input voltage (rms) to be inputted intothepowersupplyandƒlineisthelowestlinefrequencyforthe power supply (in this case 47 Hz) VINDC_MINiscalculatedbasedonthe(VINtimes TON)MAX product

( )_

_

IN ON MAXINDC MIN

ON MAX

V TV

Ttimes

= (1013)

First we must find TON_MAX to get VINDC_MIN In order for the powersupplytofunctionindiscontinuousconductionmodeTON_MAX should be smaller than the switching period minus the transformer reset time Given that the transformer reset timeis

( )IN ON MAXRESET

tr OUT

V TT

N Vtimes

=sdot (1014)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

Then the maximum on-time must be

( )_

IN ON MAXON MAX PERIOD dead

tr OUT

V TT T T

N Vtimes

= minus minussdot (1015)

TDEAD is about 48 μsec Knowing TPERIOD has to be 25 μsec because of the 40 kHz switching frequency

_750 sec25 sec 48 sec 97 sec13 55ON MAX

VTV

sdotmicro= micro minus minus micro = micro

times (1016)

From this result we can now get VINDC_MIN from equation1013

( )_

_772IN ON MAX

INDC MINON MAX

V TV V

Ttimes

= = (1017)

Substituting VINDC_MIN into equation 1012 we get

( )

( ) ( )

7722 85

1 2 2 2

arcsin2 641

21206

(2 85 772 ) 47

VVW

C C FV V Hz

times times times π + = = micro

times minus times (1018)

Increase the value of C1+C2 to account for efficiency losses For this example 136 microF is chosen

106 Current Sense ResistorThe ISENSEresistordeterminesthemaximumcurrentoutputof the power supply The output current of the power supply isdeterminedby

1_2

RESETOUT tr PRI PK

PERIOD

TI N I

T= times times times

(1019)

When the maximum current output is achieved the voltage seenon the ISENSEpin (VISENSE) should reach its maximum Thus at constant current limit

__

Isense CCPRI PK

Isense

VI

R=

(1020)

Substituting this into equation 1019 gives

_2 OUT Isense PERIOD

Isense CCtr RESET

I R TV

N Ttimes times

= times (1021)

During constant current mode where output current is at its maximum the first term in Equation 1021 is constant Therefore we can call this KC Substituting this back into equation 1021 we get

_PERIOD

Isense CC CRESET

TV K

T= times

(1022)

For iW1692 KC is 0264 V therefore RIsensedependsonthemaximumoutputcurrentby

2tr C

Isense xOUT

N KR

Itimes

= times htimes (1023)

Using this equation and Ntr from section 104

13 0264 87 152 1Isense

VRA

times= times = W

times (1024)

We recommend using plusmn1 tolerance resistors for RIsense

107 Magenitizing InductanceA feature of the iW1692 is the lack of dependence on the magnetizing inductance for the CC curve

Although the constant current limit does not depend on the magnetizing inductance there are still restrictions on the magnetizing inductance The maximum LMislimitedbytheamountofpowerthatneedstocomeoutofthetransformerin order for the power supply to regulate This is given by

( )

( )

2

__

_

402

IN ON MAXM MAX

XFMR MAX

OUT fd OUTXFMR MAX

X

V T kHzL

P

V V IP

times times=

times

minus times=

h

(1025)

The minimum LM is limited by the maximum allowableprimarypeakcurrent(IPRI_PK) 09 V on the ISENSEpinshouldcorrespond to the maximum allowable primary peak current Therefore the maximum primary peak current is

_09

PRI PKIsense

VIR

lt (1026)

Thus LMislimitedby

( )_ 09

IN ON MAXM MIN

Isense

V TL

V Rsdot

= (1027)

There is also a lower limit on ISENSE signal of 02 V This gives a second maximum value on LM compare this with the value obtained from equation 1025 and pick the smaller of the two values

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page2 82007

( )

2_

_ 2

2

02 40XFMR MAX Isense

M MAXP R

LV kHz

times times=

times (1028)

Assuming that the efficiency of the transformer is about 87 wecanobtaintheamountofpowerthatneedstocomeoutofthetransformeras

5 57587 87OUTP W W= =

(1029)

Substituting this into equation 1025 we get

( )2

_750 sec 40

1962 575M MAX

V kHzL mH

Wsdotmicro times

= =times (1030)

To get the minimum value of the primary inductance use the value for RISENSE from equation 1024

_09 615PRI PK

VI Alt =W (1031)

Substituting this primary peak current into equation 1027

_750 sec 12509 15M MIN

VL mHV

sdotmicro= =

W (1032)

Choose a primary inductance somewhere between 191 mH and 142 mH we chose 15 mH

Given a nominal LM we can now find the minimum turns ratio between primary and secondary that ensures the powersupply does not function at variable frequency (VF) mode before a certain desired voltage VOUT_VF Under VF mode theconstantcurrentIOUTmaynotbeasaccurateasinpulsewidth modulation mode

( )

_

__

2OUT VF OUT M PERIOD

xtr VF

OUT VF RESET

V I L T

NV T

times times times times

h=

times

(1033)

and

( )

_

_

2

2

OUT SHUTDOWN OUT M PERIOD

xON

INAC MIN

V I L T

TV

times times times times

h = times

(1034)

108 Primary WindingInordertokeepthetransformerfromsaturationthemaximumflux density must not be exceeded Therefore the minimum primary winding on the transformer must meet

( )IN ON MAXPRI

MAX e

V TN

B Asdot

getimes (1035)

Where BMAX is maximum flux density and Ae is the corearea

Picking (VINtimesTON)MAX to be 750 Vμsec and getting the maximum flux density and core area from the transformer datasheetwecancalculate theminimumnumberof turnsfor the primary winding Substitute BMAX as 320mT and the area of the core to be 192 mm2 we solve equation 1035 to get

2750 sec 1221

320 192PRIVN turns

mT mmsdotmicro

ge =times (1036)

To avoid hitting the maximum flux density pick a value for NPRIto be higher than this In this example 144 turns is picked

109 Secondary WindingFrom the primary winding turns we obtain the secondary winding

PRI

SECtr

NNN

= (1037)

Thus in our example

144 1113SECN turns= =

(1038)

At this point it is advantageous to make sure the primary winding and secondary winding chosen is actually feasible to wind

1010 Bias WindingVCC is the supply to the iW1692 and should be between 12 V and 16 V Capacitor C7 stores the VCC charge during IC operation and the controller checks this voltage and makes sure itrsquos within range The zener Z1 protects the IC from getting a VCC over voltage Thus the number of auxiliary windings needs to ensure that VCC does not exceed 16 V

( )SEC CC fdBIAS

OUT

N V VN

V

times +=

(1039)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

The number of auxiliary windings can be calculated using equation 1039

11 125 2555BIAS

turns VN turnsVtimes

= = (1040)

Here wersquove actually chosen a lower number for the bias winding 22 turns

1011 VSENSE Resistors and WindingThe output voltage regulation is mainly determined by the feedback signal VSENSE

_SENSE OUT PCB SENSEV V K= times (1041)

Where

4

4 3

VsenseSENSE

SEC

NRKR R N

= times+ (1042)

InternallyVSENSE is compared to a reference voltage VFB_MAX WithnocabledropcompensationVFB_MAX is 1538 V however cabledropcompensationactually increases this referenceby a specific amount as a function of the present output current Therefore under perfect regulation conditions

_ _SENSE FB MAX CableDropCompensation actual SENSEV V V K= + sdot(1043)

Substitute this into equation 1043 and solve for KSENSE

_

_ _

FB MAXSENSE

OUT PCB CableDropComp actual

VK

V V=

minus (1044)

Up to this point cable drop compensation voltage is assumed to be independent of the power supply circuitry when inreality this is not completely true

VCableDropComp_actual is affected by output regulation magnetizing inductance and RVIN

220_ 120

_ _2

_

M OUT OUT

Vin sw PowerSupply

L V IkCableDropComp actual k R f

CABLE DROP COMP OUT

FB MAX

V K

V VK

V

sdot sdot sdotWW+ sdoth

= sdot + sdot

times times(1045)

Where K1 is -1155 Vμsec and K2 is 0130 V-1μsec-1 Plug this number into equation 1045 and then find VSENSE winding and resistor values to satisfy equation 1042

For this design example we did not implement cable drop compensationsoVCableDropComp_actual is 0 V The cable drop for a 18 m 22 AWG wire is

1 2 531 18 190mOUT Cable mI R A m mVWtimes = times times times =

(1046)

Substituting this into equation 1044 we get

1538 035 0SENSE

VKV V

= =minus (1047)

Solving for R4 in equation 1042 assuming R3 is 20 kΩ and NVSENSE is 24 turns we get R4 should be around 3 kΩ

Since the iW1692 uses the VSENSE signal to determine the regulation point this signal can not be too noisy Thus C8 is used to help filter the VSENSE signal

1012 Output CapacitorsThe output capacitors are important for controlling the output voltage ripple of the power supply This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus how much charge is lost before the next switching cycle is the ripple on the output voltage Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then

_ _

OUTOUT

OUT RIPPLE PK

QC

V=

(1048)

Since charge is equal to current times time

_

_ _

OUT OFF MAXOUT

OUT RIPPLE PK

I TC

Vtimes

= (1049)

The maximum off-time during maximum output current is 25 μs

Assuming we want to get under 50 mV of ripple on the output we substitute this into equation 1049 to get

1 25 sec 50050OUT

AC FmV

times micro= = micro

(1050)

In this calculations ESR and ESL are ignored the reason this calculation is still valid is because of the second stage LC filter L3 and C11 These two components reduce the ESR and ESL ripple

1013 Snubber NetworkThe snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive The goal is to dissipate the energy from the leakage inductance of the transformer For simplicity and a more conservative design first assume the energy of the

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

Attheendofon-timethecurrenthasrampedupto

( ) ( )( ) g ON

gM

v t t ti t

Ltimes

=

(92)

This current represents a stored energy of

2( )2M

g gLE i t= times

(93)

WhenQ1turnsoffattOig(t)inLM forces a reversal of polarities on all windings Ignoring the communication-time caused by the leakage inductance LK at the instant of turn-off tO theprimarycurrenttransferstothesecondaryatanamplitudeof

( ) ( )Pd g

S

Ni t i tN

= times

(94)

Assuming the secondary winding is master the auxiliary winding is slave

VAUX 0V

VAUX = -VG xNAUX

NP

See equation 95

Figure 932 Auxiliary Voltage Waveforms

The auxiliary voltage is given by

( )AUXAUX O

S

NV V VN

= + ∆

(95)

and reflects the output voltage as shown in Figure 932

The voltage at the load differs from the secondary voltage by a diode drop and IR losses The diode drop is a function of current as are IR losses Thus if the secondary voltage is alwaysreadataconstantsecondarycurrentthedifferencebetween the output voltage and the secondary voltage is a fixed ΔV Furthermore if the voltage can be read when the secondarycurrentissmallΔV is small

The real-time waveform analyzer in the iW1692 reads this information cycle by cycle and then generates a feedback voltage VFB The VFB signal precisely represents the output voltage and is used to regulate the output voltage

94 Understanding CC and CV modeThe constant current mode (CC mode) is useful in battery charging applications During this mode of operation the iW1692 will regulate the output current at a constant maximum level regardless of the output voltage drop while avoiding continuous conduction mode

To achieve this regulation the iW1692 senses the load current indirectly through the primary current The primary currentisdetectedbytheISENSE pin through a resistor from the MOSFET source to ground (RSS) This resistor value is given by

( )( )2

CSS

OUTMAX

N KR

Itimes

=times

(96)

N is the ratio of primary turns to secondary turns of thetransformerandKC is 0264 V

95 Constant Voltage OperationAfter soft-start is completed the digital control block measures the output conditions If the output conditions do not warrant PFMmodeandtheISENSE signal is not consistently over 09 V then the device will operate in constant voltage mode

If no voltage is detected on VSENSE after 20 pulses it is assumed that the auxiliary winding of the transformer is either open or shorted and the iW1692 shuts down

As long as calculated TON for CV is less than the TONinCCthe IC operates in constant voltage mode

96 Constant Current OperationThe iW1692 has been designed to work in constant-current mode for battery charging applications If the output voltage drops but does not go below 20 of the nominal designed value the device operates in this mode

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

Out

put V

olta

ge

Output CurrentICC

VNOM

CV mode

CC

mod

e

Figure 961 Modes of operation

97 Variable Frequency ModeThe iW1692 is designed to operate in discontinuous conduction (DCM) mode at a fixed frequency of 40 kHz in both CC and CV modes To avoid operation in continuous conduction (CCM) mode the iW1692 checks for the falling edge of the VSENSE input on every cycle If a falling edge of VSENSE is not detected during the normal 25μs period the switching period is extended until the falling edge VSENSEdoes occur If the switching period reaches 75μs without VSENSE being detected the iW1692 immediately shuts off

98 PFM Mode at Light LoadThe iW1692 operates in a fixed frequency PWM mode whenIOUT is greater than approximately 5 of the specified maximum load current As the output load IOUTisreducedtheon-time tON is decreased At the moment that tONdropsbelow tON_MIN thecontroller transitions toPulseFrequencyModulation (PFM) mode Thereafter the on-time is modulated by the line voltage and the off-time is modulated by the load current The device automatically returns to PWM mode when the load current increases

99 Internal Loop CompensationThe iW1692 incorporates an internal Digital Error Amplifier with no requirement for external loop compensation The loop stability is guaranteed by design to provide at least 45 degrees of phase margin and ndash20dB of gain margin

910 Voltage Protection FunctionsThe iW1692 includes functions that protect against input and output overvoltage

The input voltage is monitored by the VINpinandtheoutputvoltage is monitored by the VSENSE pin If the voltage at these pins exceed their undervoltage or overvoltage thresholds for more than 6 cycles the iW1692 shuts-down immediately However the IC remains biased which discharges the VCCsupply Once VCC drops below the UVLO threshold the controller resets itself and then initiates a new soft-startcycle The controller continues attempting start-up but does not fully start-up until the fault condition is removed

The output voltage can be high enough to damage the output capacitor when the feedback loop is broken The iW1692 uses the primary feedback only with no secondary feedback loop When the VSENSE pin is shorted to GND (by shortingopen sense resistor) The controller will shut off with 6 consecutive pulses after start-up

911 Cable Drop CompensationThe iW1692-30 incorporates an innovative method to compensate for any IR drop in the secondary circuitry including cable and cable connector A 5 W AC adapter with 5 VDC output has 6 deviation at 1 A load current due to the drop across the DC cable without cable compensation The iW1692-30 cancels this error by providing a voltage offset to the feedback signal based on the amount of load current detected The iW1692-30 has 300mV of cable drop compensation at maximum current The iW1692-00 does not include any cable compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

100 Design Example101 Design ProcedureThis design example gives the procedure for a flyback converter using iW1692 Refer to figure 1201 for the application circuit The design objectives for this adapter are given in table 101 It meets UL IEC and CEC requirements

Figure 301 Design Flow Chart

Determine Cable Drop Compensation

Determine the Design Specifications(Vout Iout_max Vin_max Vin_min efficiency and ripple)

Determine Turns Ratio

Determine Input Bulk Capacitance

Determine Current Sensing Resistor

Determine Magnetizing Inductance

Determine Primary Turns

Determine Secondary Turns

Can you wind this transformer

Determine Bias Turns

Determine Vsense Turns and Resistors

Determine Output Capacitance

Determine Snubber Network

Determine Ton Delay Compensation

Finish

Yes No

Determine Rvin Resistors

Is the real cable drop compensation value OK Yes

No

Figure 1001 iW1692 Design Flow Chart

Parameter Symbol Range

Input Voltage VIN 85 - 264 VRMS

Frequency fIN 47 - 64 Hz

NoLoadInput PIN 200 mW

Output Voltage VOUT_CABLE 495 - 505 V

Output Current IOUT 1A

Output Ripple VRIPPLE lt100mV

Power Out POUT 5WCEC Efficiency h 65

Table 101 iW1692 Design Specification Table

102 Cable Drop CompensationCable Drop Compensation is an option included in the iW1692-30 This option helps maintain the output voltage at the end of the cable that the power supply is designed for During CV (constant voltage) mode the output current changes as the voltage remains constant This is true for the output voltage at the output of the power supply board however in certain applications the device to be charged is not directly connected to the power supply but ratheris connected via a cable This cable is seen by the power supply as a resistance So as the output current increases the output voltage at the end of the cable begins to drop With the cable compensation option the iW1692 can compensate for the resistance of the cable by incrementally increasing the output voltage seen on the power supply board to cancel out the selected cable resistance

To find the right cable compensation type for a given cable pickthecabledropcompensationnumberthatisclosesttothe voltage drop of the cable under maximum output current Use equation 101 for VOUTwhereVFD is the forward voltage oftheoutputdiode

_ _ _OUT OUT CABLE CABLE DROP COMPENSATION fdV V V V= + +(101)

Using equation 101 we know for this design VOUT is 55 V assuming no cable drop compensation is chosen and the forwarddropontheoutputdiode(VFD) is 500 mV

103 Input SelectionVIN resistors are chosen primarily to scale down the inputvoltage for the IC The scale factor for the input voltage in the IC is 00043 and the internal impedance of this pin is 20 kΩ Therefore the VINresistorsshouldequateto

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page0 82007

20 20 46300043Vin

kR k MW= minus W = W

(102)

From equation 102 ideally RVIN should be 463 MΩ because R1 R10 and R11 add up to approximately 46 MΩ By selecting the value of RVIN the VINTON_MAX_PFM and (VINTON)MAX_LIMITaredetermined

( ) _900 sec00043

2020

IN ON MAX LIMIT

Vin

VV Tk

R k

sdotmicrosdot = times

W + W

(103)

( ) 185 sec0004320

20

IN ON PFM

Vin

VV Tk

R k

sdotmicrosdot = times

W + W (104)

Keep in mind by changing RVIN to be something other than 463 MΩ the minimum and maximum input voltage for start-up will also change

Since the iW1692 uses the exact scaled value of VINforitscalculations C6 should be included to filter out any noise that mayappearontheVIN signal This is especially important for line-in surge conditions

104 Turns RatioThe maximum allowable turns ratio between the primary and secondary winding is determined by the minimum detectable reset time of the transformer during PFM mode

( )_

_

IN ON PFMtr MAX

RESET MIN OUT

V TN

T Vtimes

=times (105)

To avoid continuous conduction the turns ratio must be high enough so that TRESET does not exceed TPERIOD ndash TON ndash TDEAD TPERIOD is given by the PWM switching frequency of 40 kHz TRESET_MAX is given by

_ _RESET MAX PERIOD ON MAX DEADT T T T= minus minus (106)

Thus the minimum turns ratio is given by

( )_

_

IN ON MAXtr MIN

RESET MAX OUT

V TN

T Vtimes

=times

(107)

(VINtimesTON)PFM is limited by the iW1692 to be 185 Vμs and TRESET_MIN is required by the IC to be 23 μs

_185 sec 15

23 sec 55tr MAXVN

Vsdotmicro

= =micro times (108)

The product of VIN and TON is limited by equation 109 for CC limit performance For this example we choose 750 Vμsec

( )700 sec 850 secIN ON MAXV V T Vsdotmicro lt times lt sdotmicro (109)

Assuming TON_MAX is 97micros solving for the minimum turns ratio yields

_

_

25 sec 97 sec 48 sec

105 secRESET MAX

RESET MAX

T

T

= micro minus micro minus micro

= micro (1010)

( )_750 sec 13

105 sec 55tr MINVN

Vsdotmicro

= =micro times (1011)

Pickanumberbetween themaximumandminimumturnsratio in the example the turn ratio is 13 A turn ratio in the range of 11 to 15 is suggested for optimal performance

105 Input Bulk CapacitorThe input bulk capacitance (C1+C2) is chosen to maintain enough input power to sustain constant output power even as the input voltage is dropping In order for this to be true totalinputbulkcapacitancemustbe

_

_2

1 2 2 2_ _

arcsin2

2

(2 )

INDC MIN

INAC MIN

V

VIN

INAC MIN INDC MIN line

OUT OUTIN

PowerSupply

P

C CV V f

V IP

times

times times

π + =

minus times

times=

h

(1012)

VINAC_MIN is the minimum input voltage (rms) to be inputted intothepowersupplyandƒlineisthelowestlinefrequencyforthe power supply (in this case 47 Hz) VINDC_MINiscalculatedbasedonthe(VINtimes TON)MAX product

( )_

_

IN ON MAXINDC MIN

ON MAX

V TV

Ttimes

= (1013)

First we must find TON_MAX to get VINDC_MIN In order for the powersupplytofunctionindiscontinuousconductionmodeTON_MAX should be smaller than the switching period minus the transformer reset time Given that the transformer reset timeis

( )IN ON MAXRESET

tr OUT

V TT

N Vtimes

=sdot (1014)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

Then the maximum on-time must be

( )_

IN ON MAXON MAX PERIOD dead

tr OUT

V TT T T

N Vtimes

= minus minussdot (1015)

TDEAD is about 48 μsec Knowing TPERIOD has to be 25 μsec because of the 40 kHz switching frequency

_750 sec25 sec 48 sec 97 sec13 55ON MAX

VTV

sdotmicro= micro minus minus micro = micro

times (1016)

From this result we can now get VINDC_MIN from equation1013

( )_

_772IN ON MAX

INDC MINON MAX

V TV V

Ttimes

= = (1017)

Substituting VINDC_MIN into equation 1012 we get

( )

( ) ( )

7722 85

1 2 2 2

arcsin2 641

21206

(2 85 772 ) 47

VVW

C C FV V Hz

times times times π + = = micro

times minus times (1018)

Increase the value of C1+C2 to account for efficiency losses For this example 136 microF is chosen

106 Current Sense ResistorThe ISENSEresistordeterminesthemaximumcurrentoutputof the power supply The output current of the power supply isdeterminedby

1_2

RESETOUT tr PRI PK

PERIOD

TI N I

T= times times times

(1019)

When the maximum current output is achieved the voltage seenon the ISENSEpin (VISENSE) should reach its maximum Thus at constant current limit

__

Isense CCPRI PK

Isense

VI

R=

(1020)

Substituting this into equation 1019 gives

_2 OUT Isense PERIOD

Isense CCtr RESET

I R TV

N Ttimes times

= times (1021)

During constant current mode where output current is at its maximum the first term in Equation 1021 is constant Therefore we can call this KC Substituting this back into equation 1021 we get

_PERIOD

Isense CC CRESET

TV K

T= times

(1022)

For iW1692 KC is 0264 V therefore RIsensedependsonthemaximumoutputcurrentby

2tr C

Isense xOUT

N KR

Itimes

= times htimes (1023)

Using this equation and Ntr from section 104

13 0264 87 152 1Isense

VRA

times= times = W

times (1024)

We recommend using plusmn1 tolerance resistors for RIsense

107 Magenitizing InductanceA feature of the iW1692 is the lack of dependence on the magnetizing inductance for the CC curve

Although the constant current limit does not depend on the magnetizing inductance there are still restrictions on the magnetizing inductance The maximum LMislimitedbytheamountofpowerthatneedstocomeoutofthetransformerin order for the power supply to regulate This is given by

( )

( )

2

__

_

402

IN ON MAXM MAX

XFMR MAX

OUT fd OUTXFMR MAX

X

V T kHzL

P

V V IP

times times=

times

minus times=

h

(1025)

The minimum LM is limited by the maximum allowableprimarypeakcurrent(IPRI_PK) 09 V on the ISENSEpinshouldcorrespond to the maximum allowable primary peak current Therefore the maximum primary peak current is

_09

PRI PKIsense

VIR

lt (1026)

Thus LMislimitedby

( )_ 09

IN ON MAXM MIN

Isense

V TL

V Rsdot

= (1027)

There is also a lower limit on ISENSE signal of 02 V This gives a second maximum value on LM compare this with the value obtained from equation 1025 and pick the smaller of the two values

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page2 82007

( )

2_

_ 2

2

02 40XFMR MAX Isense

M MAXP R

LV kHz

times times=

times (1028)

Assuming that the efficiency of the transformer is about 87 wecanobtaintheamountofpowerthatneedstocomeoutofthetransformeras

5 57587 87OUTP W W= =

(1029)

Substituting this into equation 1025 we get

( )2

_750 sec 40

1962 575M MAX

V kHzL mH

Wsdotmicro times

= =times (1030)

To get the minimum value of the primary inductance use the value for RISENSE from equation 1024

_09 615PRI PK

VI Alt =W (1031)

Substituting this primary peak current into equation 1027

_750 sec 12509 15M MIN

VL mHV

sdotmicro= =

W (1032)

Choose a primary inductance somewhere between 191 mH and 142 mH we chose 15 mH

Given a nominal LM we can now find the minimum turns ratio between primary and secondary that ensures the powersupply does not function at variable frequency (VF) mode before a certain desired voltage VOUT_VF Under VF mode theconstantcurrentIOUTmaynotbeasaccurateasinpulsewidth modulation mode

( )

_

__

2OUT VF OUT M PERIOD

xtr VF

OUT VF RESET

V I L T

NV T

times times times times

h=

times

(1033)

and

( )

_

_

2

2

OUT SHUTDOWN OUT M PERIOD

xON

INAC MIN

V I L T

TV

times times times times

h = times

(1034)

108 Primary WindingInordertokeepthetransformerfromsaturationthemaximumflux density must not be exceeded Therefore the minimum primary winding on the transformer must meet

( )IN ON MAXPRI

MAX e

V TN

B Asdot

getimes (1035)

Where BMAX is maximum flux density and Ae is the corearea

Picking (VINtimesTON)MAX to be 750 Vμsec and getting the maximum flux density and core area from the transformer datasheetwecancalculate theminimumnumberof turnsfor the primary winding Substitute BMAX as 320mT and the area of the core to be 192 mm2 we solve equation 1035 to get

2750 sec 1221

320 192PRIVN turns

mT mmsdotmicro

ge =times (1036)

To avoid hitting the maximum flux density pick a value for NPRIto be higher than this In this example 144 turns is picked

109 Secondary WindingFrom the primary winding turns we obtain the secondary winding

PRI

SECtr

NNN

= (1037)

Thus in our example

144 1113SECN turns= =

(1038)

At this point it is advantageous to make sure the primary winding and secondary winding chosen is actually feasible to wind

1010 Bias WindingVCC is the supply to the iW1692 and should be between 12 V and 16 V Capacitor C7 stores the VCC charge during IC operation and the controller checks this voltage and makes sure itrsquos within range The zener Z1 protects the IC from getting a VCC over voltage Thus the number of auxiliary windings needs to ensure that VCC does not exceed 16 V

( )SEC CC fdBIAS

OUT

N V VN

V

times +=

(1039)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

The number of auxiliary windings can be calculated using equation 1039

11 125 2555BIAS

turns VN turnsVtimes

= = (1040)

Here wersquove actually chosen a lower number for the bias winding 22 turns

1011 VSENSE Resistors and WindingThe output voltage regulation is mainly determined by the feedback signal VSENSE

_SENSE OUT PCB SENSEV V K= times (1041)

Where

4

4 3

VsenseSENSE

SEC

NRKR R N

= times+ (1042)

InternallyVSENSE is compared to a reference voltage VFB_MAX WithnocabledropcompensationVFB_MAX is 1538 V however cabledropcompensationactually increases this referenceby a specific amount as a function of the present output current Therefore under perfect regulation conditions

_ _SENSE FB MAX CableDropCompensation actual SENSEV V V K= + sdot(1043)

Substitute this into equation 1043 and solve for KSENSE

_

_ _

FB MAXSENSE

OUT PCB CableDropComp actual

VK

V V=

minus (1044)

Up to this point cable drop compensation voltage is assumed to be independent of the power supply circuitry when inreality this is not completely true

VCableDropComp_actual is affected by output regulation magnetizing inductance and RVIN

220_ 120

_ _2

_

M OUT OUT

Vin sw PowerSupply

L V IkCableDropComp actual k R f

CABLE DROP COMP OUT

FB MAX

V K

V VK

V

sdot sdot sdotWW+ sdoth

= sdot + sdot

times times(1045)

Where K1 is -1155 Vμsec and K2 is 0130 V-1μsec-1 Plug this number into equation 1045 and then find VSENSE winding and resistor values to satisfy equation 1042

For this design example we did not implement cable drop compensationsoVCableDropComp_actual is 0 V The cable drop for a 18 m 22 AWG wire is

1 2 531 18 190mOUT Cable mI R A m mVWtimes = times times times =

(1046)

Substituting this into equation 1044 we get

1538 035 0SENSE

VKV V

= =minus (1047)

Solving for R4 in equation 1042 assuming R3 is 20 kΩ and NVSENSE is 24 turns we get R4 should be around 3 kΩ

Since the iW1692 uses the VSENSE signal to determine the regulation point this signal can not be too noisy Thus C8 is used to help filter the VSENSE signal

1012 Output CapacitorsThe output capacitors are important for controlling the output voltage ripple of the power supply This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus how much charge is lost before the next switching cycle is the ripple on the output voltage Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then

_ _

OUTOUT

OUT RIPPLE PK

QC

V=

(1048)

Since charge is equal to current times time

_

_ _

OUT OFF MAXOUT

OUT RIPPLE PK

I TC

Vtimes

= (1049)

The maximum off-time during maximum output current is 25 μs

Assuming we want to get under 50 mV of ripple on the output we substitute this into equation 1049 to get

1 25 sec 50050OUT

AC FmV

times micro= = micro

(1050)

In this calculations ESR and ESL are ignored the reason this calculation is still valid is because of the second stage LC filter L3 and C11 These two components reduce the ESR and ESL ripple

1013 Snubber NetworkThe snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive The goal is to dissipate the energy from the leakage inductance of the transformer For simplicity and a more conservative design first assume the energy of the

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

Out

put V

olta

ge

Output CurrentICC

VNOM

CV mode

CC

mod

e

Figure 961 Modes of operation

97 Variable Frequency ModeThe iW1692 is designed to operate in discontinuous conduction (DCM) mode at a fixed frequency of 40 kHz in both CC and CV modes To avoid operation in continuous conduction (CCM) mode the iW1692 checks for the falling edge of the VSENSE input on every cycle If a falling edge of VSENSE is not detected during the normal 25μs period the switching period is extended until the falling edge VSENSEdoes occur If the switching period reaches 75μs without VSENSE being detected the iW1692 immediately shuts off

98 PFM Mode at Light LoadThe iW1692 operates in a fixed frequency PWM mode whenIOUT is greater than approximately 5 of the specified maximum load current As the output load IOUTisreducedtheon-time tON is decreased At the moment that tONdropsbelow tON_MIN thecontroller transitions toPulseFrequencyModulation (PFM) mode Thereafter the on-time is modulated by the line voltage and the off-time is modulated by the load current The device automatically returns to PWM mode when the load current increases

99 Internal Loop CompensationThe iW1692 incorporates an internal Digital Error Amplifier with no requirement for external loop compensation The loop stability is guaranteed by design to provide at least 45 degrees of phase margin and ndash20dB of gain margin

910 Voltage Protection FunctionsThe iW1692 includes functions that protect against input and output overvoltage

The input voltage is monitored by the VINpinandtheoutputvoltage is monitored by the VSENSE pin If the voltage at these pins exceed their undervoltage or overvoltage thresholds for more than 6 cycles the iW1692 shuts-down immediately However the IC remains biased which discharges the VCCsupply Once VCC drops below the UVLO threshold the controller resets itself and then initiates a new soft-startcycle The controller continues attempting start-up but does not fully start-up until the fault condition is removed

The output voltage can be high enough to damage the output capacitor when the feedback loop is broken The iW1692 uses the primary feedback only with no secondary feedback loop When the VSENSE pin is shorted to GND (by shortingopen sense resistor) The controller will shut off with 6 consecutive pulses after start-up

911 Cable Drop CompensationThe iW1692-30 incorporates an innovative method to compensate for any IR drop in the secondary circuitry including cable and cable connector A 5 W AC adapter with 5 VDC output has 6 deviation at 1 A load current due to the drop across the DC cable without cable compensation The iW1692-30 cancels this error by providing a voltage offset to the feedback signal based on the amount of load current detected The iW1692-30 has 300mV of cable drop compensation at maximum current The iW1692-00 does not include any cable compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

100 Design Example101 Design ProcedureThis design example gives the procedure for a flyback converter using iW1692 Refer to figure 1201 for the application circuit The design objectives for this adapter are given in table 101 It meets UL IEC and CEC requirements

Figure 301 Design Flow Chart

Determine Cable Drop Compensation

Determine the Design Specifications(Vout Iout_max Vin_max Vin_min efficiency and ripple)

Determine Turns Ratio

Determine Input Bulk Capacitance

Determine Current Sensing Resistor

Determine Magnetizing Inductance

Determine Primary Turns

Determine Secondary Turns

Can you wind this transformer

Determine Bias Turns

Determine Vsense Turns and Resistors

Determine Output Capacitance

Determine Snubber Network

Determine Ton Delay Compensation

Finish

Yes No

Determine Rvin Resistors

Is the real cable drop compensation value OK Yes

No

Figure 1001 iW1692 Design Flow Chart

Parameter Symbol Range

Input Voltage VIN 85 - 264 VRMS

Frequency fIN 47 - 64 Hz

NoLoadInput PIN 200 mW

Output Voltage VOUT_CABLE 495 - 505 V

Output Current IOUT 1A

Output Ripple VRIPPLE lt100mV

Power Out POUT 5WCEC Efficiency h 65

Table 101 iW1692 Design Specification Table

102 Cable Drop CompensationCable Drop Compensation is an option included in the iW1692-30 This option helps maintain the output voltage at the end of the cable that the power supply is designed for During CV (constant voltage) mode the output current changes as the voltage remains constant This is true for the output voltage at the output of the power supply board however in certain applications the device to be charged is not directly connected to the power supply but ratheris connected via a cable This cable is seen by the power supply as a resistance So as the output current increases the output voltage at the end of the cable begins to drop With the cable compensation option the iW1692 can compensate for the resistance of the cable by incrementally increasing the output voltage seen on the power supply board to cancel out the selected cable resistance

To find the right cable compensation type for a given cable pickthecabledropcompensationnumberthatisclosesttothe voltage drop of the cable under maximum output current Use equation 101 for VOUTwhereVFD is the forward voltage oftheoutputdiode

_ _ _OUT OUT CABLE CABLE DROP COMPENSATION fdV V V V= + +(101)

Using equation 101 we know for this design VOUT is 55 V assuming no cable drop compensation is chosen and the forwarddropontheoutputdiode(VFD) is 500 mV

103 Input SelectionVIN resistors are chosen primarily to scale down the inputvoltage for the IC The scale factor for the input voltage in the IC is 00043 and the internal impedance of this pin is 20 kΩ Therefore the VINresistorsshouldequateto

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page0 82007

20 20 46300043Vin

kR k MW= minus W = W

(102)

From equation 102 ideally RVIN should be 463 MΩ because R1 R10 and R11 add up to approximately 46 MΩ By selecting the value of RVIN the VINTON_MAX_PFM and (VINTON)MAX_LIMITaredetermined

( ) _900 sec00043

2020

IN ON MAX LIMIT

Vin

VV Tk

R k

sdotmicrosdot = times

W + W

(103)

( ) 185 sec0004320

20

IN ON PFM

Vin

VV Tk

R k

sdotmicrosdot = times

W + W (104)

Keep in mind by changing RVIN to be something other than 463 MΩ the minimum and maximum input voltage for start-up will also change

Since the iW1692 uses the exact scaled value of VINforitscalculations C6 should be included to filter out any noise that mayappearontheVIN signal This is especially important for line-in surge conditions

104 Turns RatioThe maximum allowable turns ratio between the primary and secondary winding is determined by the minimum detectable reset time of the transformer during PFM mode

( )_

_

IN ON PFMtr MAX

RESET MIN OUT

V TN

T Vtimes

=times (105)

To avoid continuous conduction the turns ratio must be high enough so that TRESET does not exceed TPERIOD ndash TON ndash TDEAD TPERIOD is given by the PWM switching frequency of 40 kHz TRESET_MAX is given by

_ _RESET MAX PERIOD ON MAX DEADT T T T= minus minus (106)

Thus the minimum turns ratio is given by

( )_

_

IN ON MAXtr MIN

RESET MAX OUT

V TN

T Vtimes

=times

(107)

(VINtimesTON)PFM is limited by the iW1692 to be 185 Vμs and TRESET_MIN is required by the IC to be 23 μs

_185 sec 15

23 sec 55tr MAXVN

Vsdotmicro

= =micro times (108)

The product of VIN and TON is limited by equation 109 for CC limit performance For this example we choose 750 Vμsec

( )700 sec 850 secIN ON MAXV V T Vsdotmicro lt times lt sdotmicro (109)

Assuming TON_MAX is 97micros solving for the minimum turns ratio yields

_

_

25 sec 97 sec 48 sec

105 secRESET MAX

RESET MAX

T

T

= micro minus micro minus micro

= micro (1010)

( )_750 sec 13

105 sec 55tr MINVN

Vsdotmicro

= =micro times (1011)

Pickanumberbetween themaximumandminimumturnsratio in the example the turn ratio is 13 A turn ratio in the range of 11 to 15 is suggested for optimal performance

105 Input Bulk CapacitorThe input bulk capacitance (C1+C2) is chosen to maintain enough input power to sustain constant output power even as the input voltage is dropping In order for this to be true totalinputbulkcapacitancemustbe

_

_2

1 2 2 2_ _

arcsin2

2

(2 )

INDC MIN

INAC MIN

V

VIN

INAC MIN INDC MIN line

OUT OUTIN

PowerSupply

P

C CV V f

V IP

times

times times

π + =

minus times

times=

h

(1012)

VINAC_MIN is the minimum input voltage (rms) to be inputted intothepowersupplyandƒlineisthelowestlinefrequencyforthe power supply (in this case 47 Hz) VINDC_MINiscalculatedbasedonthe(VINtimes TON)MAX product

( )_

_

IN ON MAXINDC MIN

ON MAX

V TV

Ttimes

= (1013)

First we must find TON_MAX to get VINDC_MIN In order for the powersupplytofunctionindiscontinuousconductionmodeTON_MAX should be smaller than the switching period minus the transformer reset time Given that the transformer reset timeis

( )IN ON MAXRESET

tr OUT

V TT

N Vtimes

=sdot (1014)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

Then the maximum on-time must be

( )_

IN ON MAXON MAX PERIOD dead

tr OUT

V TT T T

N Vtimes

= minus minussdot (1015)

TDEAD is about 48 μsec Knowing TPERIOD has to be 25 μsec because of the 40 kHz switching frequency

_750 sec25 sec 48 sec 97 sec13 55ON MAX

VTV

sdotmicro= micro minus minus micro = micro

times (1016)

From this result we can now get VINDC_MIN from equation1013

( )_

_772IN ON MAX

INDC MINON MAX

V TV V

Ttimes

= = (1017)

Substituting VINDC_MIN into equation 1012 we get

( )

( ) ( )

7722 85

1 2 2 2

arcsin2 641

21206

(2 85 772 ) 47

VVW

C C FV V Hz

times times times π + = = micro

times minus times (1018)

Increase the value of C1+C2 to account for efficiency losses For this example 136 microF is chosen

106 Current Sense ResistorThe ISENSEresistordeterminesthemaximumcurrentoutputof the power supply The output current of the power supply isdeterminedby

1_2

RESETOUT tr PRI PK

PERIOD

TI N I

T= times times times

(1019)

When the maximum current output is achieved the voltage seenon the ISENSEpin (VISENSE) should reach its maximum Thus at constant current limit

__

Isense CCPRI PK

Isense

VI

R=

(1020)

Substituting this into equation 1019 gives

_2 OUT Isense PERIOD

Isense CCtr RESET

I R TV

N Ttimes times

= times (1021)

During constant current mode where output current is at its maximum the first term in Equation 1021 is constant Therefore we can call this KC Substituting this back into equation 1021 we get

_PERIOD

Isense CC CRESET

TV K

T= times

(1022)

For iW1692 KC is 0264 V therefore RIsensedependsonthemaximumoutputcurrentby

2tr C

Isense xOUT

N KR

Itimes

= times htimes (1023)

Using this equation and Ntr from section 104

13 0264 87 152 1Isense

VRA

times= times = W

times (1024)

We recommend using plusmn1 tolerance resistors for RIsense

107 Magenitizing InductanceA feature of the iW1692 is the lack of dependence on the magnetizing inductance for the CC curve

Although the constant current limit does not depend on the magnetizing inductance there are still restrictions on the magnetizing inductance The maximum LMislimitedbytheamountofpowerthatneedstocomeoutofthetransformerin order for the power supply to regulate This is given by

( )

( )

2

__

_

402

IN ON MAXM MAX

XFMR MAX

OUT fd OUTXFMR MAX

X

V T kHzL

P

V V IP

times times=

times

minus times=

h

(1025)

The minimum LM is limited by the maximum allowableprimarypeakcurrent(IPRI_PK) 09 V on the ISENSEpinshouldcorrespond to the maximum allowable primary peak current Therefore the maximum primary peak current is

_09

PRI PKIsense

VIR

lt (1026)

Thus LMislimitedby

( )_ 09

IN ON MAXM MIN

Isense

V TL

V Rsdot

= (1027)

There is also a lower limit on ISENSE signal of 02 V This gives a second maximum value on LM compare this with the value obtained from equation 1025 and pick the smaller of the two values

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page2 82007

( )

2_

_ 2

2

02 40XFMR MAX Isense

M MAXP R

LV kHz

times times=

times (1028)

Assuming that the efficiency of the transformer is about 87 wecanobtaintheamountofpowerthatneedstocomeoutofthetransformeras

5 57587 87OUTP W W= =

(1029)

Substituting this into equation 1025 we get

( )2

_750 sec 40

1962 575M MAX

V kHzL mH

Wsdotmicro times

= =times (1030)

To get the minimum value of the primary inductance use the value for RISENSE from equation 1024

_09 615PRI PK

VI Alt =W (1031)

Substituting this primary peak current into equation 1027

_750 sec 12509 15M MIN

VL mHV

sdotmicro= =

W (1032)

Choose a primary inductance somewhere between 191 mH and 142 mH we chose 15 mH

Given a nominal LM we can now find the minimum turns ratio between primary and secondary that ensures the powersupply does not function at variable frequency (VF) mode before a certain desired voltage VOUT_VF Under VF mode theconstantcurrentIOUTmaynotbeasaccurateasinpulsewidth modulation mode

( )

_

__

2OUT VF OUT M PERIOD

xtr VF

OUT VF RESET

V I L T

NV T

times times times times

h=

times

(1033)

and

( )

_

_

2

2

OUT SHUTDOWN OUT M PERIOD

xON

INAC MIN

V I L T

TV

times times times times

h = times

(1034)

108 Primary WindingInordertokeepthetransformerfromsaturationthemaximumflux density must not be exceeded Therefore the minimum primary winding on the transformer must meet

( )IN ON MAXPRI

MAX e

V TN

B Asdot

getimes (1035)

Where BMAX is maximum flux density and Ae is the corearea

Picking (VINtimesTON)MAX to be 750 Vμsec and getting the maximum flux density and core area from the transformer datasheetwecancalculate theminimumnumberof turnsfor the primary winding Substitute BMAX as 320mT and the area of the core to be 192 mm2 we solve equation 1035 to get

2750 sec 1221

320 192PRIVN turns

mT mmsdotmicro

ge =times (1036)

To avoid hitting the maximum flux density pick a value for NPRIto be higher than this In this example 144 turns is picked

109 Secondary WindingFrom the primary winding turns we obtain the secondary winding

PRI

SECtr

NNN

= (1037)

Thus in our example

144 1113SECN turns= =

(1038)

At this point it is advantageous to make sure the primary winding and secondary winding chosen is actually feasible to wind

1010 Bias WindingVCC is the supply to the iW1692 and should be between 12 V and 16 V Capacitor C7 stores the VCC charge during IC operation and the controller checks this voltage and makes sure itrsquos within range The zener Z1 protects the IC from getting a VCC over voltage Thus the number of auxiliary windings needs to ensure that VCC does not exceed 16 V

( )SEC CC fdBIAS

OUT

N V VN

V

times +=

(1039)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

The number of auxiliary windings can be calculated using equation 1039

11 125 2555BIAS

turns VN turnsVtimes

= = (1040)

Here wersquove actually chosen a lower number for the bias winding 22 turns

1011 VSENSE Resistors and WindingThe output voltage regulation is mainly determined by the feedback signal VSENSE

_SENSE OUT PCB SENSEV V K= times (1041)

Where

4

4 3

VsenseSENSE

SEC

NRKR R N

= times+ (1042)

InternallyVSENSE is compared to a reference voltage VFB_MAX WithnocabledropcompensationVFB_MAX is 1538 V however cabledropcompensationactually increases this referenceby a specific amount as a function of the present output current Therefore under perfect regulation conditions

_ _SENSE FB MAX CableDropCompensation actual SENSEV V V K= + sdot(1043)

Substitute this into equation 1043 and solve for KSENSE

_

_ _

FB MAXSENSE

OUT PCB CableDropComp actual

VK

V V=

minus (1044)

Up to this point cable drop compensation voltage is assumed to be independent of the power supply circuitry when inreality this is not completely true

VCableDropComp_actual is affected by output regulation magnetizing inductance and RVIN

220_ 120

_ _2

_

M OUT OUT

Vin sw PowerSupply

L V IkCableDropComp actual k R f

CABLE DROP COMP OUT

FB MAX

V K

V VK

V

sdot sdot sdotWW+ sdoth

= sdot + sdot

times times(1045)

Where K1 is -1155 Vμsec and K2 is 0130 V-1μsec-1 Plug this number into equation 1045 and then find VSENSE winding and resistor values to satisfy equation 1042

For this design example we did not implement cable drop compensationsoVCableDropComp_actual is 0 V The cable drop for a 18 m 22 AWG wire is

1 2 531 18 190mOUT Cable mI R A m mVWtimes = times times times =

(1046)

Substituting this into equation 1044 we get

1538 035 0SENSE

VKV V

= =minus (1047)

Solving for R4 in equation 1042 assuming R3 is 20 kΩ and NVSENSE is 24 turns we get R4 should be around 3 kΩ

Since the iW1692 uses the VSENSE signal to determine the regulation point this signal can not be too noisy Thus C8 is used to help filter the VSENSE signal

1012 Output CapacitorsThe output capacitors are important for controlling the output voltage ripple of the power supply This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus how much charge is lost before the next switching cycle is the ripple on the output voltage Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then

_ _

OUTOUT

OUT RIPPLE PK

QC

V=

(1048)

Since charge is equal to current times time

_

_ _

OUT OFF MAXOUT

OUT RIPPLE PK

I TC

Vtimes

= (1049)

The maximum off-time during maximum output current is 25 μs

Assuming we want to get under 50 mV of ripple on the output we substitute this into equation 1049 to get

1 25 sec 50050OUT

AC FmV

times micro= = micro

(1050)

In this calculations ESR and ESL are ignored the reason this calculation is still valid is because of the second stage LC filter L3 and C11 These two components reduce the ESR and ESL ripple

1013 Snubber NetworkThe snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive The goal is to dissipate the energy from the leakage inductance of the transformer For simplicity and a more conservative design first assume the energy of the

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

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MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

100 Design Example101 Design ProcedureThis design example gives the procedure for a flyback converter using iW1692 Refer to figure 1201 for the application circuit The design objectives for this adapter are given in table 101 It meets UL IEC and CEC requirements

Figure 301 Design Flow Chart

Determine Cable Drop Compensation

Determine the Design Specifications(Vout Iout_max Vin_max Vin_min efficiency and ripple)

Determine Turns Ratio

Determine Input Bulk Capacitance

Determine Current Sensing Resistor

Determine Magnetizing Inductance

Determine Primary Turns

Determine Secondary Turns

Can you wind this transformer

Determine Bias Turns

Determine Vsense Turns and Resistors

Determine Output Capacitance

Determine Snubber Network

Determine Ton Delay Compensation

Finish

Yes No

Determine Rvin Resistors

Is the real cable drop compensation value OK Yes

No

Figure 1001 iW1692 Design Flow Chart

Parameter Symbol Range

Input Voltage VIN 85 - 264 VRMS

Frequency fIN 47 - 64 Hz

NoLoadInput PIN 200 mW

Output Voltage VOUT_CABLE 495 - 505 V

Output Current IOUT 1A

Output Ripple VRIPPLE lt100mV

Power Out POUT 5WCEC Efficiency h 65

Table 101 iW1692 Design Specification Table

102 Cable Drop CompensationCable Drop Compensation is an option included in the iW1692-30 This option helps maintain the output voltage at the end of the cable that the power supply is designed for During CV (constant voltage) mode the output current changes as the voltage remains constant This is true for the output voltage at the output of the power supply board however in certain applications the device to be charged is not directly connected to the power supply but ratheris connected via a cable This cable is seen by the power supply as a resistance So as the output current increases the output voltage at the end of the cable begins to drop With the cable compensation option the iW1692 can compensate for the resistance of the cable by incrementally increasing the output voltage seen on the power supply board to cancel out the selected cable resistance

To find the right cable compensation type for a given cable pickthecabledropcompensationnumberthatisclosesttothe voltage drop of the cable under maximum output current Use equation 101 for VOUTwhereVFD is the forward voltage oftheoutputdiode

_ _ _OUT OUT CABLE CABLE DROP COMPENSATION fdV V V V= + +(101)

Using equation 101 we know for this design VOUT is 55 V assuming no cable drop compensation is chosen and the forwarddropontheoutputdiode(VFD) is 500 mV

103 Input SelectionVIN resistors are chosen primarily to scale down the inputvoltage for the IC The scale factor for the input voltage in the IC is 00043 and the internal impedance of this pin is 20 kΩ Therefore the VINresistorsshouldequateto

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page0 82007

20 20 46300043Vin

kR k MW= minus W = W

(102)

From equation 102 ideally RVIN should be 463 MΩ because R1 R10 and R11 add up to approximately 46 MΩ By selecting the value of RVIN the VINTON_MAX_PFM and (VINTON)MAX_LIMITaredetermined

( ) _900 sec00043

2020

IN ON MAX LIMIT

Vin

VV Tk

R k

sdotmicrosdot = times

W + W

(103)

( ) 185 sec0004320

20

IN ON PFM

Vin

VV Tk

R k

sdotmicrosdot = times

W + W (104)

Keep in mind by changing RVIN to be something other than 463 MΩ the minimum and maximum input voltage for start-up will also change

Since the iW1692 uses the exact scaled value of VINforitscalculations C6 should be included to filter out any noise that mayappearontheVIN signal This is especially important for line-in surge conditions

104 Turns RatioThe maximum allowable turns ratio between the primary and secondary winding is determined by the minimum detectable reset time of the transformer during PFM mode

( )_

_

IN ON PFMtr MAX

RESET MIN OUT

V TN

T Vtimes

=times (105)

To avoid continuous conduction the turns ratio must be high enough so that TRESET does not exceed TPERIOD ndash TON ndash TDEAD TPERIOD is given by the PWM switching frequency of 40 kHz TRESET_MAX is given by

_ _RESET MAX PERIOD ON MAX DEADT T T T= minus minus (106)

Thus the minimum turns ratio is given by

( )_

_

IN ON MAXtr MIN

RESET MAX OUT

V TN

T Vtimes

=times

(107)

(VINtimesTON)PFM is limited by the iW1692 to be 185 Vμs and TRESET_MIN is required by the IC to be 23 μs

_185 sec 15

23 sec 55tr MAXVN

Vsdotmicro

= =micro times (108)

The product of VIN and TON is limited by equation 109 for CC limit performance For this example we choose 750 Vμsec

( )700 sec 850 secIN ON MAXV V T Vsdotmicro lt times lt sdotmicro (109)

Assuming TON_MAX is 97micros solving for the minimum turns ratio yields

_

_

25 sec 97 sec 48 sec

105 secRESET MAX

RESET MAX

T

T

= micro minus micro minus micro

= micro (1010)

( )_750 sec 13

105 sec 55tr MINVN

Vsdotmicro

= =micro times (1011)

Pickanumberbetween themaximumandminimumturnsratio in the example the turn ratio is 13 A turn ratio in the range of 11 to 15 is suggested for optimal performance

105 Input Bulk CapacitorThe input bulk capacitance (C1+C2) is chosen to maintain enough input power to sustain constant output power even as the input voltage is dropping In order for this to be true totalinputbulkcapacitancemustbe

_

_2

1 2 2 2_ _

arcsin2

2

(2 )

INDC MIN

INAC MIN

V

VIN

INAC MIN INDC MIN line

OUT OUTIN

PowerSupply

P

C CV V f

V IP

times

times times

π + =

minus times

times=

h

(1012)

VINAC_MIN is the minimum input voltage (rms) to be inputted intothepowersupplyandƒlineisthelowestlinefrequencyforthe power supply (in this case 47 Hz) VINDC_MINiscalculatedbasedonthe(VINtimes TON)MAX product

( )_

_

IN ON MAXINDC MIN

ON MAX

V TV

Ttimes

= (1013)

First we must find TON_MAX to get VINDC_MIN In order for the powersupplytofunctionindiscontinuousconductionmodeTON_MAX should be smaller than the switching period minus the transformer reset time Given that the transformer reset timeis

( )IN ON MAXRESET

tr OUT

V TT

N Vtimes

=sdot (1014)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

Then the maximum on-time must be

( )_

IN ON MAXON MAX PERIOD dead

tr OUT

V TT T T

N Vtimes

= minus minussdot (1015)

TDEAD is about 48 μsec Knowing TPERIOD has to be 25 μsec because of the 40 kHz switching frequency

_750 sec25 sec 48 sec 97 sec13 55ON MAX

VTV

sdotmicro= micro minus minus micro = micro

times (1016)

From this result we can now get VINDC_MIN from equation1013

( )_

_772IN ON MAX

INDC MINON MAX

V TV V

Ttimes

= = (1017)

Substituting VINDC_MIN into equation 1012 we get

( )

( ) ( )

7722 85

1 2 2 2

arcsin2 641

21206

(2 85 772 ) 47

VVW

C C FV V Hz

times times times π + = = micro

times minus times (1018)

Increase the value of C1+C2 to account for efficiency losses For this example 136 microF is chosen

106 Current Sense ResistorThe ISENSEresistordeterminesthemaximumcurrentoutputof the power supply The output current of the power supply isdeterminedby

1_2

RESETOUT tr PRI PK

PERIOD

TI N I

T= times times times

(1019)

When the maximum current output is achieved the voltage seenon the ISENSEpin (VISENSE) should reach its maximum Thus at constant current limit

__

Isense CCPRI PK

Isense

VI

R=

(1020)

Substituting this into equation 1019 gives

_2 OUT Isense PERIOD

Isense CCtr RESET

I R TV

N Ttimes times

= times (1021)

During constant current mode where output current is at its maximum the first term in Equation 1021 is constant Therefore we can call this KC Substituting this back into equation 1021 we get

_PERIOD

Isense CC CRESET

TV K

T= times

(1022)

For iW1692 KC is 0264 V therefore RIsensedependsonthemaximumoutputcurrentby

2tr C

Isense xOUT

N KR

Itimes

= times htimes (1023)

Using this equation and Ntr from section 104

13 0264 87 152 1Isense

VRA

times= times = W

times (1024)

We recommend using plusmn1 tolerance resistors for RIsense

107 Magenitizing InductanceA feature of the iW1692 is the lack of dependence on the magnetizing inductance for the CC curve

Although the constant current limit does not depend on the magnetizing inductance there are still restrictions on the magnetizing inductance The maximum LMislimitedbytheamountofpowerthatneedstocomeoutofthetransformerin order for the power supply to regulate This is given by

( )

( )

2

__

_

402

IN ON MAXM MAX

XFMR MAX

OUT fd OUTXFMR MAX

X

V T kHzL

P

V V IP

times times=

times

minus times=

h

(1025)

The minimum LM is limited by the maximum allowableprimarypeakcurrent(IPRI_PK) 09 V on the ISENSEpinshouldcorrespond to the maximum allowable primary peak current Therefore the maximum primary peak current is

_09

PRI PKIsense

VIR

lt (1026)

Thus LMislimitedby

( )_ 09

IN ON MAXM MIN

Isense

V TL

V Rsdot

= (1027)

There is also a lower limit on ISENSE signal of 02 V This gives a second maximum value on LM compare this with the value obtained from equation 1025 and pick the smaller of the two values

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page2 82007

( )

2_

_ 2

2

02 40XFMR MAX Isense

M MAXP R

LV kHz

times times=

times (1028)

Assuming that the efficiency of the transformer is about 87 wecanobtaintheamountofpowerthatneedstocomeoutofthetransformeras

5 57587 87OUTP W W= =

(1029)

Substituting this into equation 1025 we get

( )2

_750 sec 40

1962 575M MAX

V kHzL mH

Wsdotmicro times

= =times (1030)

To get the minimum value of the primary inductance use the value for RISENSE from equation 1024

_09 615PRI PK

VI Alt =W (1031)

Substituting this primary peak current into equation 1027

_750 sec 12509 15M MIN

VL mHV

sdotmicro= =

W (1032)

Choose a primary inductance somewhere between 191 mH and 142 mH we chose 15 mH

Given a nominal LM we can now find the minimum turns ratio between primary and secondary that ensures the powersupply does not function at variable frequency (VF) mode before a certain desired voltage VOUT_VF Under VF mode theconstantcurrentIOUTmaynotbeasaccurateasinpulsewidth modulation mode

( )

_

__

2OUT VF OUT M PERIOD

xtr VF

OUT VF RESET

V I L T

NV T

times times times times

h=

times

(1033)

and

( )

_

_

2

2

OUT SHUTDOWN OUT M PERIOD

xON

INAC MIN

V I L T

TV

times times times times

h = times

(1034)

108 Primary WindingInordertokeepthetransformerfromsaturationthemaximumflux density must not be exceeded Therefore the minimum primary winding on the transformer must meet

( )IN ON MAXPRI

MAX e

V TN

B Asdot

getimes (1035)

Where BMAX is maximum flux density and Ae is the corearea

Picking (VINtimesTON)MAX to be 750 Vμsec and getting the maximum flux density and core area from the transformer datasheetwecancalculate theminimumnumberof turnsfor the primary winding Substitute BMAX as 320mT and the area of the core to be 192 mm2 we solve equation 1035 to get

2750 sec 1221

320 192PRIVN turns

mT mmsdotmicro

ge =times (1036)

To avoid hitting the maximum flux density pick a value for NPRIto be higher than this In this example 144 turns is picked

109 Secondary WindingFrom the primary winding turns we obtain the secondary winding

PRI

SECtr

NNN

= (1037)

Thus in our example

144 1113SECN turns= =

(1038)

At this point it is advantageous to make sure the primary winding and secondary winding chosen is actually feasible to wind

1010 Bias WindingVCC is the supply to the iW1692 and should be between 12 V and 16 V Capacitor C7 stores the VCC charge during IC operation and the controller checks this voltage and makes sure itrsquos within range The zener Z1 protects the IC from getting a VCC over voltage Thus the number of auxiliary windings needs to ensure that VCC does not exceed 16 V

( )SEC CC fdBIAS

OUT

N V VN

V

times +=

(1039)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

The number of auxiliary windings can be calculated using equation 1039

11 125 2555BIAS

turns VN turnsVtimes

= = (1040)

Here wersquove actually chosen a lower number for the bias winding 22 turns

1011 VSENSE Resistors and WindingThe output voltage regulation is mainly determined by the feedback signal VSENSE

_SENSE OUT PCB SENSEV V K= times (1041)

Where

4

4 3

VsenseSENSE

SEC

NRKR R N

= times+ (1042)

InternallyVSENSE is compared to a reference voltage VFB_MAX WithnocabledropcompensationVFB_MAX is 1538 V however cabledropcompensationactually increases this referenceby a specific amount as a function of the present output current Therefore under perfect regulation conditions

_ _SENSE FB MAX CableDropCompensation actual SENSEV V V K= + sdot(1043)

Substitute this into equation 1043 and solve for KSENSE

_

_ _

FB MAXSENSE

OUT PCB CableDropComp actual

VK

V V=

minus (1044)

Up to this point cable drop compensation voltage is assumed to be independent of the power supply circuitry when inreality this is not completely true

VCableDropComp_actual is affected by output regulation magnetizing inductance and RVIN

220_ 120

_ _2

_

M OUT OUT

Vin sw PowerSupply

L V IkCableDropComp actual k R f

CABLE DROP COMP OUT

FB MAX

V K

V VK

V

sdot sdot sdotWW+ sdoth

= sdot + sdot

times times(1045)

Where K1 is -1155 Vμsec and K2 is 0130 V-1μsec-1 Plug this number into equation 1045 and then find VSENSE winding and resistor values to satisfy equation 1042

For this design example we did not implement cable drop compensationsoVCableDropComp_actual is 0 V The cable drop for a 18 m 22 AWG wire is

1 2 531 18 190mOUT Cable mI R A m mVWtimes = times times times =

(1046)

Substituting this into equation 1044 we get

1538 035 0SENSE

VKV V

= =minus (1047)

Solving for R4 in equation 1042 assuming R3 is 20 kΩ and NVSENSE is 24 turns we get R4 should be around 3 kΩ

Since the iW1692 uses the VSENSE signal to determine the regulation point this signal can not be too noisy Thus C8 is used to help filter the VSENSE signal

1012 Output CapacitorsThe output capacitors are important for controlling the output voltage ripple of the power supply This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus how much charge is lost before the next switching cycle is the ripple on the output voltage Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then

_ _

OUTOUT

OUT RIPPLE PK

QC

V=

(1048)

Since charge is equal to current times time

_

_ _

OUT OFF MAXOUT

OUT RIPPLE PK

I TC

Vtimes

= (1049)

The maximum off-time during maximum output current is 25 μs

Assuming we want to get under 50 mV of ripple on the output we substitute this into equation 1049 to get

1 25 sec 50050OUT

AC FmV

times micro= = micro

(1050)

In this calculations ESR and ESL are ignored the reason this calculation is still valid is because of the second stage LC filter L3 and C11 These two components reduce the ESR and ESL ripple

1013 Snubber NetworkThe snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive The goal is to dissipate the energy from the leakage inductance of the transformer For simplicity and a more conservative design first assume the energy of the

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page0 82007

20 20 46300043Vin

kR k MW= minus W = W

(102)

From equation 102 ideally RVIN should be 463 MΩ because R1 R10 and R11 add up to approximately 46 MΩ By selecting the value of RVIN the VINTON_MAX_PFM and (VINTON)MAX_LIMITaredetermined

( ) _900 sec00043

2020

IN ON MAX LIMIT

Vin

VV Tk

R k

sdotmicrosdot = times

W + W

(103)

( ) 185 sec0004320

20

IN ON PFM

Vin

VV Tk

R k

sdotmicrosdot = times

W + W (104)

Keep in mind by changing RVIN to be something other than 463 MΩ the minimum and maximum input voltage for start-up will also change

Since the iW1692 uses the exact scaled value of VINforitscalculations C6 should be included to filter out any noise that mayappearontheVIN signal This is especially important for line-in surge conditions

104 Turns RatioThe maximum allowable turns ratio between the primary and secondary winding is determined by the minimum detectable reset time of the transformer during PFM mode

( )_

_

IN ON PFMtr MAX

RESET MIN OUT

V TN

T Vtimes

=times (105)

To avoid continuous conduction the turns ratio must be high enough so that TRESET does not exceed TPERIOD ndash TON ndash TDEAD TPERIOD is given by the PWM switching frequency of 40 kHz TRESET_MAX is given by

_ _RESET MAX PERIOD ON MAX DEADT T T T= minus minus (106)

Thus the minimum turns ratio is given by

( )_

_

IN ON MAXtr MIN

RESET MAX OUT

V TN

T Vtimes

=times

(107)

(VINtimesTON)PFM is limited by the iW1692 to be 185 Vμs and TRESET_MIN is required by the IC to be 23 μs

_185 sec 15

23 sec 55tr MAXVN

Vsdotmicro

= =micro times (108)

The product of VIN and TON is limited by equation 109 for CC limit performance For this example we choose 750 Vμsec

( )700 sec 850 secIN ON MAXV V T Vsdotmicro lt times lt sdotmicro (109)

Assuming TON_MAX is 97micros solving for the minimum turns ratio yields

_

_

25 sec 97 sec 48 sec

105 secRESET MAX

RESET MAX

T

T

= micro minus micro minus micro

= micro (1010)

( )_750 sec 13

105 sec 55tr MINVN

Vsdotmicro

= =micro times (1011)

Pickanumberbetween themaximumandminimumturnsratio in the example the turn ratio is 13 A turn ratio in the range of 11 to 15 is suggested for optimal performance

105 Input Bulk CapacitorThe input bulk capacitance (C1+C2) is chosen to maintain enough input power to sustain constant output power even as the input voltage is dropping In order for this to be true totalinputbulkcapacitancemustbe

_

_2

1 2 2 2_ _

arcsin2

2

(2 )

INDC MIN

INAC MIN

V

VIN

INAC MIN INDC MIN line

OUT OUTIN

PowerSupply

P

C CV V f

V IP

times

times times

π + =

minus times

times=

h

(1012)

VINAC_MIN is the minimum input voltage (rms) to be inputted intothepowersupplyandƒlineisthelowestlinefrequencyforthe power supply (in this case 47 Hz) VINDC_MINiscalculatedbasedonthe(VINtimes TON)MAX product

( )_

_

IN ON MAXINDC MIN

ON MAX

V TV

Ttimes

= (1013)

First we must find TON_MAX to get VINDC_MIN In order for the powersupplytofunctionindiscontinuousconductionmodeTON_MAX should be smaller than the switching period minus the transformer reset time Given that the transformer reset timeis

( )IN ON MAXRESET

tr OUT

V TT

N Vtimes

=sdot (1014)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

Then the maximum on-time must be

( )_

IN ON MAXON MAX PERIOD dead

tr OUT

V TT T T

N Vtimes

= minus minussdot (1015)

TDEAD is about 48 μsec Knowing TPERIOD has to be 25 μsec because of the 40 kHz switching frequency

_750 sec25 sec 48 sec 97 sec13 55ON MAX

VTV

sdotmicro= micro minus minus micro = micro

times (1016)

From this result we can now get VINDC_MIN from equation1013

( )_

_772IN ON MAX

INDC MINON MAX

V TV V

Ttimes

= = (1017)

Substituting VINDC_MIN into equation 1012 we get

( )

( ) ( )

7722 85

1 2 2 2

arcsin2 641

21206

(2 85 772 ) 47

VVW

C C FV V Hz

times times times π + = = micro

times minus times (1018)

Increase the value of C1+C2 to account for efficiency losses For this example 136 microF is chosen

106 Current Sense ResistorThe ISENSEresistordeterminesthemaximumcurrentoutputof the power supply The output current of the power supply isdeterminedby

1_2

RESETOUT tr PRI PK

PERIOD

TI N I

T= times times times

(1019)

When the maximum current output is achieved the voltage seenon the ISENSEpin (VISENSE) should reach its maximum Thus at constant current limit

__

Isense CCPRI PK

Isense

VI

R=

(1020)

Substituting this into equation 1019 gives

_2 OUT Isense PERIOD

Isense CCtr RESET

I R TV

N Ttimes times

= times (1021)

During constant current mode where output current is at its maximum the first term in Equation 1021 is constant Therefore we can call this KC Substituting this back into equation 1021 we get

_PERIOD

Isense CC CRESET

TV K

T= times

(1022)

For iW1692 KC is 0264 V therefore RIsensedependsonthemaximumoutputcurrentby

2tr C

Isense xOUT

N KR

Itimes

= times htimes (1023)

Using this equation and Ntr from section 104

13 0264 87 152 1Isense

VRA

times= times = W

times (1024)

We recommend using plusmn1 tolerance resistors for RIsense

107 Magenitizing InductanceA feature of the iW1692 is the lack of dependence on the magnetizing inductance for the CC curve

Although the constant current limit does not depend on the magnetizing inductance there are still restrictions on the magnetizing inductance The maximum LMislimitedbytheamountofpowerthatneedstocomeoutofthetransformerin order for the power supply to regulate This is given by

( )

( )

2

__

_

402

IN ON MAXM MAX

XFMR MAX

OUT fd OUTXFMR MAX

X

V T kHzL

P

V V IP

times times=

times

minus times=

h

(1025)

The minimum LM is limited by the maximum allowableprimarypeakcurrent(IPRI_PK) 09 V on the ISENSEpinshouldcorrespond to the maximum allowable primary peak current Therefore the maximum primary peak current is

_09

PRI PKIsense

VIR

lt (1026)

Thus LMislimitedby

( )_ 09

IN ON MAXM MIN

Isense

V TL

V Rsdot

= (1027)

There is also a lower limit on ISENSE signal of 02 V This gives a second maximum value on LM compare this with the value obtained from equation 1025 and pick the smaller of the two values

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page2 82007

( )

2_

_ 2

2

02 40XFMR MAX Isense

M MAXP R

LV kHz

times times=

times (1028)

Assuming that the efficiency of the transformer is about 87 wecanobtaintheamountofpowerthatneedstocomeoutofthetransformeras

5 57587 87OUTP W W= =

(1029)

Substituting this into equation 1025 we get

( )2

_750 sec 40

1962 575M MAX

V kHzL mH

Wsdotmicro times

= =times (1030)

To get the minimum value of the primary inductance use the value for RISENSE from equation 1024

_09 615PRI PK

VI Alt =W (1031)

Substituting this primary peak current into equation 1027

_750 sec 12509 15M MIN

VL mHV

sdotmicro= =

W (1032)

Choose a primary inductance somewhere between 191 mH and 142 mH we chose 15 mH

Given a nominal LM we can now find the minimum turns ratio between primary and secondary that ensures the powersupply does not function at variable frequency (VF) mode before a certain desired voltage VOUT_VF Under VF mode theconstantcurrentIOUTmaynotbeasaccurateasinpulsewidth modulation mode

( )

_

__

2OUT VF OUT M PERIOD

xtr VF

OUT VF RESET

V I L T

NV T

times times times times

h=

times

(1033)

and

( )

_

_

2

2

OUT SHUTDOWN OUT M PERIOD

xON

INAC MIN

V I L T

TV

times times times times

h = times

(1034)

108 Primary WindingInordertokeepthetransformerfromsaturationthemaximumflux density must not be exceeded Therefore the minimum primary winding on the transformer must meet

( )IN ON MAXPRI

MAX e

V TN

B Asdot

getimes (1035)

Where BMAX is maximum flux density and Ae is the corearea

Picking (VINtimesTON)MAX to be 750 Vμsec and getting the maximum flux density and core area from the transformer datasheetwecancalculate theminimumnumberof turnsfor the primary winding Substitute BMAX as 320mT and the area of the core to be 192 mm2 we solve equation 1035 to get

2750 sec 1221

320 192PRIVN turns

mT mmsdotmicro

ge =times (1036)

To avoid hitting the maximum flux density pick a value for NPRIto be higher than this In this example 144 turns is picked

109 Secondary WindingFrom the primary winding turns we obtain the secondary winding

PRI

SECtr

NNN

= (1037)

Thus in our example

144 1113SECN turns= =

(1038)

At this point it is advantageous to make sure the primary winding and secondary winding chosen is actually feasible to wind

1010 Bias WindingVCC is the supply to the iW1692 and should be between 12 V and 16 V Capacitor C7 stores the VCC charge during IC operation and the controller checks this voltage and makes sure itrsquos within range The zener Z1 protects the IC from getting a VCC over voltage Thus the number of auxiliary windings needs to ensure that VCC does not exceed 16 V

( )SEC CC fdBIAS

OUT

N V VN

V

times +=

(1039)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

The number of auxiliary windings can be calculated using equation 1039

11 125 2555BIAS

turns VN turnsVtimes

= = (1040)

Here wersquove actually chosen a lower number for the bias winding 22 turns

1011 VSENSE Resistors and WindingThe output voltage regulation is mainly determined by the feedback signal VSENSE

_SENSE OUT PCB SENSEV V K= times (1041)

Where

4

4 3

VsenseSENSE

SEC

NRKR R N

= times+ (1042)

InternallyVSENSE is compared to a reference voltage VFB_MAX WithnocabledropcompensationVFB_MAX is 1538 V however cabledropcompensationactually increases this referenceby a specific amount as a function of the present output current Therefore under perfect regulation conditions

_ _SENSE FB MAX CableDropCompensation actual SENSEV V V K= + sdot(1043)

Substitute this into equation 1043 and solve for KSENSE

_

_ _

FB MAXSENSE

OUT PCB CableDropComp actual

VK

V V=

minus (1044)

Up to this point cable drop compensation voltage is assumed to be independent of the power supply circuitry when inreality this is not completely true

VCableDropComp_actual is affected by output regulation magnetizing inductance and RVIN

220_ 120

_ _2

_

M OUT OUT

Vin sw PowerSupply

L V IkCableDropComp actual k R f

CABLE DROP COMP OUT

FB MAX

V K

V VK

V

sdot sdot sdotWW+ sdoth

= sdot + sdot

times times(1045)

Where K1 is -1155 Vμsec and K2 is 0130 V-1μsec-1 Plug this number into equation 1045 and then find VSENSE winding and resistor values to satisfy equation 1042

For this design example we did not implement cable drop compensationsoVCableDropComp_actual is 0 V The cable drop for a 18 m 22 AWG wire is

1 2 531 18 190mOUT Cable mI R A m mVWtimes = times times times =

(1046)

Substituting this into equation 1044 we get

1538 035 0SENSE

VKV V

= =minus (1047)

Solving for R4 in equation 1042 assuming R3 is 20 kΩ and NVSENSE is 24 turns we get R4 should be around 3 kΩ

Since the iW1692 uses the VSENSE signal to determine the regulation point this signal can not be too noisy Thus C8 is used to help filter the VSENSE signal

1012 Output CapacitorsThe output capacitors are important for controlling the output voltage ripple of the power supply This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus how much charge is lost before the next switching cycle is the ripple on the output voltage Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then

_ _

OUTOUT

OUT RIPPLE PK

QC

V=

(1048)

Since charge is equal to current times time

_

_ _

OUT OFF MAXOUT

OUT RIPPLE PK

I TC

Vtimes

= (1049)

The maximum off-time during maximum output current is 25 μs

Assuming we want to get under 50 mV of ripple on the output we substitute this into equation 1049 to get

1 25 sec 50050OUT

AC FmV

times micro= = micro

(1050)

In this calculations ESR and ESL are ignored the reason this calculation is still valid is because of the second stage LC filter L3 and C11 These two components reduce the ESR and ESL ripple

1013 Snubber NetworkThe snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive The goal is to dissipate the energy from the leakage inductance of the transformer For simplicity and a more conservative design first assume the energy of the

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

Then the maximum on-time must be

( )_

IN ON MAXON MAX PERIOD dead

tr OUT

V TT T T

N Vtimes

= minus minussdot (1015)

TDEAD is about 48 μsec Knowing TPERIOD has to be 25 μsec because of the 40 kHz switching frequency

_750 sec25 sec 48 sec 97 sec13 55ON MAX

VTV

sdotmicro= micro minus minus micro = micro

times (1016)

From this result we can now get VINDC_MIN from equation1013

( )_

_772IN ON MAX

INDC MINON MAX

V TV V

Ttimes

= = (1017)

Substituting VINDC_MIN into equation 1012 we get

( )

( ) ( )

7722 85

1 2 2 2

arcsin2 641

21206

(2 85 772 ) 47

VVW

C C FV V Hz

times times times π + = = micro

times minus times (1018)

Increase the value of C1+C2 to account for efficiency losses For this example 136 microF is chosen

106 Current Sense ResistorThe ISENSEresistordeterminesthemaximumcurrentoutputof the power supply The output current of the power supply isdeterminedby

1_2

RESETOUT tr PRI PK

PERIOD

TI N I

T= times times times

(1019)

When the maximum current output is achieved the voltage seenon the ISENSEpin (VISENSE) should reach its maximum Thus at constant current limit

__

Isense CCPRI PK

Isense

VI

R=

(1020)

Substituting this into equation 1019 gives

_2 OUT Isense PERIOD

Isense CCtr RESET

I R TV

N Ttimes times

= times (1021)

During constant current mode where output current is at its maximum the first term in Equation 1021 is constant Therefore we can call this KC Substituting this back into equation 1021 we get

_PERIOD

Isense CC CRESET

TV K

T= times

(1022)

For iW1692 KC is 0264 V therefore RIsensedependsonthemaximumoutputcurrentby

2tr C

Isense xOUT

N KR

Itimes

= times htimes (1023)

Using this equation and Ntr from section 104

13 0264 87 152 1Isense

VRA

times= times = W

times (1024)

We recommend using plusmn1 tolerance resistors for RIsense

107 Magenitizing InductanceA feature of the iW1692 is the lack of dependence on the magnetizing inductance for the CC curve

Although the constant current limit does not depend on the magnetizing inductance there are still restrictions on the magnetizing inductance The maximum LMislimitedbytheamountofpowerthatneedstocomeoutofthetransformerin order for the power supply to regulate This is given by

( )

( )

2

__

_

402

IN ON MAXM MAX

XFMR MAX

OUT fd OUTXFMR MAX

X

V T kHzL

P

V V IP

times times=

times

minus times=

h

(1025)

The minimum LM is limited by the maximum allowableprimarypeakcurrent(IPRI_PK) 09 V on the ISENSEpinshouldcorrespond to the maximum allowable primary peak current Therefore the maximum primary peak current is

_09

PRI PKIsense

VIR

lt (1026)

Thus LMislimitedby

( )_ 09

IN ON MAXM MIN

Isense

V TL

V Rsdot

= (1027)

There is also a lower limit on ISENSE signal of 02 V This gives a second maximum value on LM compare this with the value obtained from equation 1025 and pick the smaller of the two values

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page2 82007

( )

2_

_ 2

2

02 40XFMR MAX Isense

M MAXP R

LV kHz

times times=

times (1028)

Assuming that the efficiency of the transformer is about 87 wecanobtaintheamountofpowerthatneedstocomeoutofthetransformeras

5 57587 87OUTP W W= =

(1029)

Substituting this into equation 1025 we get

( )2

_750 sec 40

1962 575M MAX

V kHzL mH

Wsdotmicro times

= =times (1030)

To get the minimum value of the primary inductance use the value for RISENSE from equation 1024

_09 615PRI PK

VI Alt =W (1031)

Substituting this primary peak current into equation 1027

_750 sec 12509 15M MIN

VL mHV

sdotmicro= =

W (1032)

Choose a primary inductance somewhere between 191 mH and 142 mH we chose 15 mH

Given a nominal LM we can now find the minimum turns ratio between primary and secondary that ensures the powersupply does not function at variable frequency (VF) mode before a certain desired voltage VOUT_VF Under VF mode theconstantcurrentIOUTmaynotbeasaccurateasinpulsewidth modulation mode

( )

_

__

2OUT VF OUT M PERIOD

xtr VF

OUT VF RESET

V I L T

NV T

times times times times

h=

times

(1033)

and

( )

_

_

2

2

OUT SHUTDOWN OUT M PERIOD

xON

INAC MIN

V I L T

TV

times times times times

h = times

(1034)

108 Primary WindingInordertokeepthetransformerfromsaturationthemaximumflux density must not be exceeded Therefore the minimum primary winding on the transformer must meet

( )IN ON MAXPRI

MAX e

V TN

B Asdot

getimes (1035)

Where BMAX is maximum flux density and Ae is the corearea

Picking (VINtimesTON)MAX to be 750 Vμsec and getting the maximum flux density and core area from the transformer datasheetwecancalculate theminimumnumberof turnsfor the primary winding Substitute BMAX as 320mT and the area of the core to be 192 mm2 we solve equation 1035 to get

2750 sec 1221

320 192PRIVN turns

mT mmsdotmicro

ge =times (1036)

To avoid hitting the maximum flux density pick a value for NPRIto be higher than this In this example 144 turns is picked

109 Secondary WindingFrom the primary winding turns we obtain the secondary winding

PRI

SECtr

NNN

= (1037)

Thus in our example

144 1113SECN turns= =

(1038)

At this point it is advantageous to make sure the primary winding and secondary winding chosen is actually feasible to wind

1010 Bias WindingVCC is the supply to the iW1692 and should be between 12 V and 16 V Capacitor C7 stores the VCC charge during IC operation and the controller checks this voltage and makes sure itrsquos within range The zener Z1 protects the IC from getting a VCC over voltage Thus the number of auxiliary windings needs to ensure that VCC does not exceed 16 V

( )SEC CC fdBIAS

OUT

N V VN

V

times +=

(1039)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

The number of auxiliary windings can be calculated using equation 1039

11 125 2555BIAS

turns VN turnsVtimes

= = (1040)

Here wersquove actually chosen a lower number for the bias winding 22 turns

1011 VSENSE Resistors and WindingThe output voltage regulation is mainly determined by the feedback signal VSENSE

_SENSE OUT PCB SENSEV V K= times (1041)

Where

4

4 3

VsenseSENSE

SEC

NRKR R N

= times+ (1042)

InternallyVSENSE is compared to a reference voltage VFB_MAX WithnocabledropcompensationVFB_MAX is 1538 V however cabledropcompensationactually increases this referenceby a specific amount as a function of the present output current Therefore under perfect regulation conditions

_ _SENSE FB MAX CableDropCompensation actual SENSEV V V K= + sdot(1043)

Substitute this into equation 1043 and solve for KSENSE

_

_ _

FB MAXSENSE

OUT PCB CableDropComp actual

VK

V V=

minus (1044)

Up to this point cable drop compensation voltage is assumed to be independent of the power supply circuitry when inreality this is not completely true

VCableDropComp_actual is affected by output regulation magnetizing inductance and RVIN

220_ 120

_ _2

_

M OUT OUT

Vin sw PowerSupply

L V IkCableDropComp actual k R f

CABLE DROP COMP OUT

FB MAX

V K

V VK

V

sdot sdot sdotWW+ sdoth

= sdot + sdot

times times(1045)

Where K1 is -1155 Vμsec and K2 is 0130 V-1μsec-1 Plug this number into equation 1045 and then find VSENSE winding and resistor values to satisfy equation 1042

For this design example we did not implement cable drop compensationsoVCableDropComp_actual is 0 V The cable drop for a 18 m 22 AWG wire is

1 2 531 18 190mOUT Cable mI R A m mVWtimes = times times times =

(1046)

Substituting this into equation 1044 we get

1538 035 0SENSE

VKV V

= =minus (1047)

Solving for R4 in equation 1042 assuming R3 is 20 kΩ and NVSENSE is 24 turns we get R4 should be around 3 kΩ

Since the iW1692 uses the VSENSE signal to determine the regulation point this signal can not be too noisy Thus C8 is used to help filter the VSENSE signal

1012 Output CapacitorsThe output capacitors are important for controlling the output voltage ripple of the power supply This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus how much charge is lost before the next switching cycle is the ripple on the output voltage Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then

_ _

OUTOUT

OUT RIPPLE PK

QC

V=

(1048)

Since charge is equal to current times time

_

_ _

OUT OFF MAXOUT

OUT RIPPLE PK

I TC

Vtimes

= (1049)

The maximum off-time during maximum output current is 25 μs

Assuming we want to get under 50 mV of ripple on the output we substitute this into equation 1049 to get

1 25 sec 50050OUT

AC FmV

times micro= = micro

(1050)

In this calculations ESR and ESL are ignored the reason this calculation is still valid is because of the second stage LC filter L3 and C11 These two components reduce the ESR and ESL ripple

1013 Snubber NetworkThe snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive The goal is to dissipate the energy from the leakage inductance of the transformer For simplicity and a more conservative design first assume the energy of the

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page2 82007

( )

2_

_ 2

2

02 40XFMR MAX Isense

M MAXP R

LV kHz

times times=

times (1028)

Assuming that the efficiency of the transformer is about 87 wecanobtaintheamountofpowerthatneedstocomeoutofthetransformeras

5 57587 87OUTP W W= =

(1029)

Substituting this into equation 1025 we get

( )2

_750 sec 40

1962 575M MAX

V kHzL mH

Wsdotmicro times

= =times (1030)

To get the minimum value of the primary inductance use the value for RISENSE from equation 1024

_09 615PRI PK

VI Alt =W (1031)

Substituting this primary peak current into equation 1027

_750 sec 12509 15M MIN

VL mHV

sdotmicro= =

W (1032)

Choose a primary inductance somewhere between 191 mH and 142 mH we chose 15 mH

Given a nominal LM we can now find the minimum turns ratio between primary and secondary that ensures the powersupply does not function at variable frequency (VF) mode before a certain desired voltage VOUT_VF Under VF mode theconstantcurrentIOUTmaynotbeasaccurateasinpulsewidth modulation mode

( )

_

__

2OUT VF OUT M PERIOD

xtr VF

OUT VF RESET

V I L T

NV T

times times times times

h=

times

(1033)

and

( )

_

_

2

2

OUT SHUTDOWN OUT M PERIOD

xON

INAC MIN

V I L T

TV

times times times times

h = times

(1034)

108 Primary WindingInordertokeepthetransformerfromsaturationthemaximumflux density must not be exceeded Therefore the minimum primary winding on the transformer must meet

( )IN ON MAXPRI

MAX e

V TN

B Asdot

getimes (1035)

Where BMAX is maximum flux density and Ae is the corearea

Picking (VINtimesTON)MAX to be 750 Vμsec and getting the maximum flux density and core area from the transformer datasheetwecancalculate theminimumnumberof turnsfor the primary winding Substitute BMAX as 320mT and the area of the core to be 192 mm2 we solve equation 1035 to get

2750 sec 1221

320 192PRIVN turns

mT mmsdotmicro

ge =times (1036)

To avoid hitting the maximum flux density pick a value for NPRIto be higher than this In this example 144 turns is picked

109 Secondary WindingFrom the primary winding turns we obtain the secondary winding

PRI

SECtr

NNN

= (1037)

Thus in our example

144 1113SECN turns= =

(1038)

At this point it is advantageous to make sure the primary winding and secondary winding chosen is actually feasible to wind

1010 Bias WindingVCC is the supply to the iW1692 and should be between 12 V and 16 V Capacitor C7 stores the VCC charge during IC operation and the controller checks this voltage and makes sure itrsquos within range The zener Z1 protects the IC from getting a VCC over voltage Thus the number of auxiliary windings needs to ensure that VCC does not exceed 16 V

( )SEC CC fdBIAS

OUT

N V VN

V

times +=

(1039)

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

The number of auxiliary windings can be calculated using equation 1039

11 125 2555BIAS

turns VN turnsVtimes

= = (1040)

Here wersquove actually chosen a lower number for the bias winding 22 turns

1011 VSENSE Resistors and WindingThe output voltage regulation is mainly determined by the feedback signal VSENSE

_SENSE OUT PCB SENSEV V K= times (1041)

Where

4

4 3

VsenseSENSE

SEC

NRKR R N

= times+ (1042)

InternallyVSENSE is compared to a reference voltage VFB_MAX WithnocabledropcompensationVFB_MAX is 1538 V however cabledropcompensationactually increases this referenceby a specific amount as a function of the present output current Therefore under perfect regulation conditions

_ _SENSE FB MAX CableDropCompensation actual SENSEV V V K= + sdot(1043)

Substitute this into equation 1043 and solve for KSENSE

_

_ _

FB MAXSENSE

OUT PCB CableDropComp actual

VK

V V=

minus (1044)

Up to this point cable drop compensation voltage is assumed to be independent of the power supply circuitry when inreality this is not completely true

VCableDropComp_actual is affected by output regulation magnetizing inductance and RVIN

220_ 120

_ _2

_

M OUT OUT

Vin sw PowerSupply

L V IkCableDropComp actual k R f

CABLE DROP COMP OUT

FB MAX

V K

V VK

V

sdot sdot sdotWW+ sdoth

= sdot + sdot

times times(1045)

Where K1 is -1155 Vμsec and K2 is 0130 V-1μsec-1 Plug this number into equation 1045 and then find VSENSE winding and resistor values to satisfy equation 1042

For this design example we did not implement cable drop compensationsoVCableDropComp_actual is 0 V The cable drop for a 18 m 22 AWG wire is

1 2 531 18 190mOUT Cable mI R A m mVWtimes = times times times =

(1046)

Substituting this into equation 1044 we get

1538 035 0SENSE

VKV V

= =minus (1047)

Solving for R4 in equation 1042 assuming R3 is 20 kΩ and NVSENSE is 24 turns we get R4 should be around 3 kΩ

Since the iW1692 uses the VSENSE signal to determine the regulation point this signal can not be too noisy Thus C8 is used to help filter the VSENSE signal

1012 Output CapacitorsThe output capacitors are important for controlling the output voltage ripple of the power supply This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus how much charge is lost before the next switching cycle is the ripple on the output voltage Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then

_ _

OUTOUT

OUT RIPPLE PK

QC

V=

(1048)

Since charge is equal to current times time

_

_ _

OUT OFF MAXOUT

OUT RIPPLE PK

I TC

Vtimes

= (1049)

The maximum off-time during maximum output current is 25 μs

Assuming we want to get under 50 mV of ripple on the output we substitute this into equation 1049 to get

1 25 sec 50050OUT

AC FmV

times micro= = micro

(1050)

In this calculations ESR and ESL are ignored the reason this calculation is still valid is because of the second stage LC filter L3 and C11 These two components reduce the ESR and ESL ripple

1013 Snubber NetworkThe snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive The goal is to dissipate the energy from the leakage inductance of the transformer For simplicity and a more conservative design first assume the energy of the

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

The number of auxiliary windings can be calculated using equation 1039

11 125 2555BIAS

turns VN turnsVtimes

= = (1040)

Here wersquove actually chosen a lower number for the bias winding 22 turns

1011 VSENSE Resistors and WindingThe output voltage regulation is mainly determined by the feedback signal VSENSE

_SENSE OUT PCB SENSEV V K= times (1041)

Where

4

4 3

VsenseSENSE

SEC

NRKR R N

= times+ (1042)

InternallyVSENSE is compared to a reference voltage VFB_MAX WithnocabledropcompensationVFB_MAX is 1538 V however cabledropcompensationactually increases this referenceby a specific amount as a function of the present output current Therefore under perfect regulation conditions

_ _SENSE FB MAX CableDropCompensation actual SENSEV V V K= + sdot(1043)

Substitute this into equation 1043 and solve for KSENSE

_

_ _

FB MAXSENSE

OUT PCB CableDropComp actual

VK

V V=

minus (1044)

Up to this point cable drop compensation voltage is assumed to be independent of the power supply circuitry when inreality this is not completely true

VCableDropComp_actual is affected by output regulation magnetizing inductance and RVIN

220_ 120

_ _2

_

M OUT OUT

Vin sw PowerSupply

L V IkCableDropComp actual k R f

CABLE DROP COMP OUT

FB MAX

V K

V VK

V

sdot sdot sdotWW+ sdoth

= sdot + sdot

times times(1045)

Where K1 is -1155 Vμsec and K2 is 0130 V-1μsec-1 Plug this number into equation 1045 and then find VSENSE winding and resistor values to satisfy equation 1042

For this design example we did not implement cable drop compensationsoVCableDropComp_actual is 0 V The cable drop for a 18 m 22 AWG wire is

1 2 531 18 190mOUT Cable mI R A m mVWtimes = times times times =

(1046)

Substituting this into equation 1044 we get

1538 035 0SENSE

VKV V

= =minus (1047)

Solving for R4 in equation 1042 assuming R3 is 20 kΩ and NVSENSE is 24 turns we get R4 should be around 3 kΩ

Since the iW1692 uses the VSENSE signal to determine the regulation point this signal can not be too noisy Thus C8 is used to help filter the VSENSE signal

1012 Output CapacitorsThe output capacitors are important for controlling the output voltage ripple of the power supply This is because the amount of charge stored on a capacitor is related to the voltage seen across the capacitor thus how much charge is lost before the next switching cycle is the ripple on the output voltage Assuming an ideal capacitor where ESR (equivalent series resistance) and ESL (equivalent series inductance) are negligible then

_ _

OUTOUT

OUT RIPPLE PK

QC

V=

(1048)

Since charge is equal to current times time

_

_ _

OUT OFF MAXOUT

OUT RIPPLE PK

I TC

Vtimes

= (1049)

The maximum off-time during maximum output current is 25 μs

Assuming we want to get under 50 mV of ripple on the output we substitute this into equation 1049 to get

1 25 sec 50050OUT

AC FmV

times micro= = micro

(1050)

In this calculations ESR and ESL are ignored the reason this calculation is still valid is because of the second stage LC filter L3 and C11 These two components reduce the ESR and ESL ripple

1013 Snubber NetworkThe snubber network is implemented to reduce the voltage stress on the MOSFET immediately following the turn off of the gate drive The goal is to dissipate the energy from the leakage inductance of the transformer For simplicity and a more conservative design first assume the energy of the

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page4 82007

leakage inductance is only dissipated through the snubber Thus

2 2 21 1_ 32 2lk pri pk pk valL I C V V times times = times times minus (1051)

LLK canbemeasured from the transformer IPRI_PK is 09 V divided by RISENSEandVPKisthepeakVDS of the MOSFET Choose C3 keeping in mind that the larger the value of C3 you choose the lower the voltage stress is that is applied to the MOSFET However capacitors are more expensive the larger their capacitance Choose C3 based on these two criteriaandselectVPKandVVAL Now a resistor needs to be selected to dissipate VPK to VVAL during the on-time of the gate driver The dissipation of this resistor is given by

5 3

TperiodR Cval

pk

Ve

V

minussdot=

(1052)

Using equation 1052 solve for R5 This will give a conservative estimate of what C3 and R5 should be

Included in the snubber network is also a resistor (R6) in series with the diode (D6) D6 directs the current to C3 when the MOSFET is turned off however there is some reverse current that goes through the diode immediately after the MOSFET is turned back on This reverse current occurs because there is a short period of time when thediode still conducts after switching from forward biased to reverse biased This conduction will distort the falling edge of theVSENSE curve and affect the operation of the IC So the resistor R6 is there to diminish the reverse current that goes through D6 immediately after the MOSFET is turned on

1014 TON Delay FilteriW1692 also contains a feature that allows for adjustment to match high line and low line constant current curves The mismatch in high line and low line curves is due to the delay from the IC propagation delay driver turn on delay and the MOSFET actually turn on delay The driver turn on delay is further increased by R9 this is to lessen the amount of stress on the MOSFET during turn on To adjust for these delays the iW1692 factors these delays into its calculations and slightly over compensates for them to provide flexibility R15 and C5 provide extra delay in the circuit to tweak the compensation To determine values R15 and C5 follow these steps

1 Measure the difference between high line and low line constant current limit without R15 and C5

2 Find the curve that best matches this difference from Figure 1107

3 Find the LM that matches the power supply Match the tRC

4 Find R15 and C5 from equation 1053

15 5RC R Ct = times (1053)

We observe that the difference between high line and low line constant current limit is 20 mA Matching the primary inductance 2 mH and the curve we find tRC to be 44times10-8s We then pick R15 to be 1 kΩ and substitute into equation 1053

8544 10 sec 1k Cminussdot = Wtimes (1054)

Solving for C5 in equation 1054 we get 44 pF The result should be a match between high line and low line constant current curves See figure 1107 for details

1015 PCB LayoutIn the iW1692 there are two signals that are important to control output performance these are the ISENSE signal and the VSENSE signal The ISS resistor should be close to thesource of the MOSFET to avoid any trace resistance from contaminating the ISENSE signal Also the ISENSE signal should beplacedclosetotheISENSE pin The VSENSE signal should be placed close to the transformer to improve the quality of the sensing signal

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

110 Design Example Performance Characteristics

Figure 1104 VSENSE Short Before Start-up (no load)

Figure 1105 ISENSE Short at 90 VAC

Figure 1106 output Short Fault (50 load)

74

70

66

64

60

56

52

Effic

ienc

y (

)

Output Current (mA)0 250 500 750 1000

90 VAC264 VAC

Figure 1101 VSENSE Efficiency at 90 VAC and 264 VAC

Figure 1102 Regulation without Cable Drop Compensation

Figure 1103 Regulation with Cable Drop Compensation

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page 82007

120 Application Circuit

N RTN

L

T1-B

T1-A

5V1A

Q1

D1 - D4 IN4007

+ +

++

U1iW1692

4

2

6

5

3

1

R147 kΩ R11

22 MΩ

R16100 Ω

R9100 Ω

R151 kΩ

R1356 kΩ

R1815 Ω

R10243 MΩ

L11 mH

L31microH

R14560Ω

R5100 kΩ

F110 Ω

R6150Ω

R121Ω

R320kΩ

R430 kΩ

C168 microF400 V

C268 microF400 V

D5FR102

D6

Z115V

C31 nF500 V

C10650 microF10 V

C11 330 microF10V

C547 pF

C6470 pF

C7470 nF

C868 pF

C947microF

OUTPUT

GND

ISENSE

VIN

VCC

VSENSE

Figure 1201 Typical Application Circuit

Figure 1107 TON Compensation Chart

110 Design Example Performance Characteristics

180

150

120

90

60

30

0

(R15

x C

5) τ

RC

(ns)

Magnetizing Inductance LM (mH)0 075 150 225 30

50 mA40 mA30 mA20 mA10 mA

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page7 82007

130 Physical Dimensions

Figure 1301 Physical dimensions 6-lead SOT-23 package

140 Ordering Information

1 3

46 5

2

6-Lead Small Outline Transistor Package

D

Compliant to JEDEC Standard MO-178AB Controlling dimensions are in millimeters

SEATINGPLANE

A1

COPLANARITY010

E1

B

e

e1

A A2

C

L α

Sym

bol

Millimeters

A1

MIN MAX

000 015A - 145

B 030 050C 008 022D 280 300E

E1e 095 BSC

e1 190 BSC

280 BSC 165 BSC

L 030 0600deg 8degα

A2 090 130

E

290 BSC

Part Number Mark Option Package Operating Temp Range Description

iW1692-00 Cxxx Cable Drop Compensation 0 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

iW1692-30 Dxxx Cable Drop Compensation 300 mV SOT23-6L -40degC le TA le 85degC -Tape amp Reel1

Note 1 Tape amp Reel packing quantity is 3000 units

Note 2 In the mark column ldquoxxxrdquo represents the lot ID code Refer to ILG-005 device marking specification for more detailed information

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt

iW1692Low-Power Off-line Digital PWM Controller

MK-4008-NR Page8 82007

iWatt Inc is a fabless semiconductor company that develops intelligent power management ICs for computer communication and consumer markets The companyrsquos patented pulseTraintrade technology the industryrsquos first truly digital approach to power system regulation is revolutionizing power supply design

Trademark Informationcopy 2007 iWatt Inc All rights reserved iWatt the iW light bulb and pulseTrain are trademarks of iWatt Inc All other trademarks and registered trademarks are the property of their respective companies

Contact Information

WebhttpwwwiwattcomE-mailinfoiwattcomPhone 408-374-4200Fax 408-341-0455iWatt Inc101 Albright WayLos Gatos CA 95032-1827

DisclaimeriWatt reserves the right to make changes to its products and to discontinue products without notice The applications information schematic diagrams and other reference information included herein is provided as a design aid only and are therefore provided as-is iWatt makes no warranties with respect to this information and disclaims any implied warranties of merchantability or non-infringement of third-party intellectual property rights

Certain applications using semiconductor products may involve potential risks of death personal injury or severe property or environmental damage (ldquoCritical Applicationsrdquo)

IWATT SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED INTENDED AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS

Inclusion of iWatt products in critical applications is understood to be fully at the risk of the customer Questions concerning potential risk applications should be directed to iWatt Inc

iWatt semiconductors are typically used in power supplies in which high voltages are present during operation High-voltage safety precautions should be observed in design and operation to minimize the chance of injury

About iWatt