IEEE Transactions on Power Electronics Volume 12 Issue 6 1997 [Doi 10.1109_63.641500] Yim-Shu Lee; Bo-Tao Lin -- Adding Active Clamping and Soft Switching to Boost-flyback Single-stage

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 12, NO. 6, NOVEMBER 1997 1017

Adding Active Clamping and Soft Switchingto Boost-Flyback Single-Stage IsolatedPower-Factor-Corrected Power Supplies

Yim-Shu Lee and Bo-Tao Lin

Abstract—Single-stage isolated power-factor-corrected powersupplies (SSIPP’s) have attractive features of fast regulation andsingle control loop. However, SSIPP circuits also have highervoltage stress and heavier loss (when compared with a normaldc–dc converter), which severely limit their practical applications.In this paper, we propose to add active clamping to SSIPPto recycle the energy trapped in the leakage inductance of thetransformer (to keep the switch voltage stress low) and to achievesoft switching of the electronic devices (to further reduce the loss).This arrangement significantly improves the viability of SSIPP forpractical applications. The auxiliary active-clamping transistor inthe proposed circuit uses the same control/driver circuit as themain switching transistor. Simulations and experimental worksverifying the operation of the converter are reported.

Index Terms— AC–DC converter, active clamping, power-factor correction, soft switching.

I. INTRODUCTION

OWING TO THE growing concern about the harmonicpollution of power distribution systems and the adoption

of standards such as IEC 1000-3-2, there is a need to reduce theharmonic contents of the ac line currents of power supplies. Tomeet this requirement, it is customary to add a power-factor-correction (PFC) preregulator to the dc–dc converter of theswitching power supply [4]–[8]. Although relatively simple,mature, and viable for a wide power range of applications,this is not an optimized design. In order to reduce the overallsize and cost, researchers attempted to integrate the functionsof power-factor correction and isolated dc–dc conversion intoa single power stage.

References [9]–[12] proposed to use parallel power process-ing circuits to achieve the functions of PFC and fast regulationof output voltage. Parallel power processing circuits havehigh conversion efficiency. But they need complex convertertopologies and sophisticated control techniques. On the otherhand, [13]–[16] proposed to use series power processingcircuits, which use conventional converters and control tech-niques. Compared with the parallel processing scheme, theseries processing scheme has a much simpler circuit structure,but lower conversion efficiency. Therefore, in order to compete

Manuscript received May 20, 1996; revised April 22, 1997. This workwas supported by the Industry Department of the Hong Kong Governmentand Hong Kong Polytechnic University. Recommended by Associate Editor,R. L. Steigerwald.

The authors are with the Department of Electronic Engineering, Hong KongPolytechnic University, Kowloon, Hong Kong.

Publisher Item Identifier S 0885-8993(97)08087-3.

with the parallel scheme, the conversion efficiency of the seriesscheme needs to be improved.

Among the circuits using the series scheme, the familyof single-stage isolated power-factor-corrected power supplies(SSIPP’s) proposed in [16] is particularly attractive due toits fast regulation and single pulse-width modulated (PWM)control loop. An example of the Boost-flyback SSIPP isshown in Fig. 1. It is believed that such supplies will findwide applications in low-cost low-power consumer products.However, SSIPP circuits suffer from disadvantages of highervoltage stress and heavier loss when compared with ordinarydc–dc converters. The problem becomes even worse at highswitching frequencies because the unavoidable leakage induc-tance of the power transformer will then produce large voltagespikes during switching. Although a dissipative snubber circuitmay be used to reduce the voltage spike, it further increasesthe loss of the circuit. These unfavorable factors severely limitthe practical applications of SSIPP.

In this paper, we propose an active-clamping circuit [1]–[3]for the Boost-flyback SSIPP to solve the problems men-tioned above. The active-clamping circuit limits the voltagespikes, recycles the energy trapped in the leakage inductance,and provides a mechanism for achieving soft switching ofthe electronic switches to reduce the switching loss. In thisway, the viability of SSIPP for practical applications willbe significantly improved. In our design, the auxiliary switchuses the same control/driver circuit as the main switch. Theadditional cost is therefore minimal. In Section II, details of thecircuit operation are presented. Design criteria are developedin Section III. Simulations and experimental results are givenin Section IV. The conclusions are summarized in Section V.

II. PRINCIPLE OF OPERATION

A. Basic Circuit Topology

The proposed Boost-flyback SSIPP with active clampingand soft switching is shown in Fig. 2. The circuit can beunderstood as a cascaded connection of a Boost converterfollowed by a flyback converter, as identified by the dotted-line boxes. The two converters (called cells) share the samemain switch S1. (S2 is an auxiliary switch only.) Nodes Aand B serve as the output terminals of the Boost cell and, atthe same time, the input terminals of the flyback cell. TheBoost cell is designed to operate in discontinuous current

0885–8993/97$10.00 1997 IEEE

1018 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 12, NO. 6, NOVEMBER 1997

Fig. 1. Boost-flyback SSIPP.

Fig. 2. Boost-flyback SSIPP with active clamping and soft switching.

mode (DCM), acting as an inherent power-factor corrector.The flyback cell is designed to function as a postregulator,which can operate in either continuous current mode (CCM)or DCM [2], [3]. The diode D1 is added to prevent the primarycurrent of transformer T from circulating through D2.represents the sum of the parasitic capacitances of S1 andS2 and any external capacitance. represents the sum of thetransformer leakage inductance and any external inductance.

and form a series resonant circuit, which makes softswitching possible. and S2 form an active clamping circuit,limiting the resonant voltage across the main switch.is anenergy-storage capacitor [13], [16].

B. Operation of the Boost Cell

Referring to Fig. 2, we assume that is a constant dcvoltage source within each switching cycle. The filteringcapacitors and are functionally also constant voltagesources within a switching cycle. Since is very large, theconverter can be decoupled into two almost independent cells,as shown in Fig. 3, i.e., a Boost cell in DCM followed by aflyback cell in either CCM or DCM. It is also assumed thatthe clamping capacitor is much larger than the resonantcapacitor .

The Boost cell shown in Fig. 3(a) consists of input inductor, diodes D1 and D2, main switch S1, energy-storage capacitor

, and is the equivalent loading resistance ofthe Boost cell. The Boost cell, operating in DCM, has threeswitching stages as shown in Fig. 4, where the thick linesindicate the conducting paths. The operation of the Boost cellcan be explained as follows.

1) [Fig. 4(a)]: The main switch S1 is turned on at(the beginning of a switching cycle), and the diode

D2 is cut off (reversely biased). The input inductorislinearly charged from zero by the input voltage source

2) [Fig. 4(b)]: S1 is turned off at , and D2 startsto conduct. is charged by (the current throughuntil becomes zero at

3) [Fig. 4(c)]: D2 is naturally cut off at , andstays idle.

When S1 is turned on again at the beginning of the nextcycle, the operations described above will repeat once more.

C. Operation of the Flyback Cell

We now study the actively clamped/soft-switched flybackcell shown in Fig. 3(b), where the flyback transformerisrepresented by an equivalent circuit showing the magnetizingand leakage inductances. Here, it is assumed that the flybackcell is operating in CCM. Referring to the six topological states

LEE AND LIN: ADDING ACTIVE CLAMPING AND SOFT SWITCHING TO POWER SUPPLIES 1019

(a)

(b)

Fig. 3. Decomposition of converter into (a) Boost cell and (b) flyback cellwith active clamping and soft switching.

of the converter cell shown in Fig. 5, the detailed operation ofthe circuit can be explained as follows.

1) [Fig. 5(a)]: At S1 is turned on while S2is off. The output rectifier D3 is reversely biased. Themagnetizing inductance together with the reso-nant inductance is linearly charged up. The current

increases linearly.2) [Fig. 5(b)]: At S1 is turned off. is charged

by the magnetizing current shouldactually be charged in a resonant manner. However,since the charging time is short compared with theoscillating period increasesalmost linearly. Also, since the rise of is relativeslow, the turn-off loss of S1 is small (soft turn-off).

3) [Fig. 5(c)]: At increases to the valueof (the voltage across , and the body diode of S2starts to conduct. fixes the voltage across andto Since is much larger than nearly all ofthe magnetizing current is now diverted tothe diode to charge up Consequently, increasesaccording to

(1)

4) [Fig. 5(d)]: At has increased to the pointwhere the secondary transformer voltage is sufficientto forward bias D3. The transformer primary voltageis then clamped by the large output capacitancetoapproximately , and and begin to resonate.However, since is very large, is discharged almostlinearly through While is flowing in the positivedirection (through the body diode of S2), S2 is turned on

(soft turn-on). A little later becomes negative (becauseis larger than

5) [Fig. 5(e)]: At S2 is turned off, effectivelyremoving from the circuit. A new resonant circuit isformed by and The relative slow change ofensures a soft turn-off of S2. The transformer primaryvoltage remains at as is discharged.

6) [Fig. 5(f)]: At the body diode of S1 startsto conduct. (In order to achieve a soft turn-on, S1 isturned on while the body diode of S1 is conducting.)When is equal to at , the secondary currentdecreases to zero and D3 becomes reversely biased. Themagnetizing and resonant inductances and thenbegin to be linearly charged up again, starting anotherswitching cycle.

D. Interactions Between the Boost Cell and the Flyback Cell

The explanation given above is useful for the understandingof the operation of each converter cell individually. However,when the two cells are combined together to form a single-stage converter, there will be interactions.

Fig. 6 shows the actual waveforms of the complete Boost-flyback SSIPP. Regarding these waveforms, the followingshould be noted.

1) Due to the conduction of the body diode of the MOSFETS1 even before it is turned on by the gate drive, theeffective duty cycle of the main switch (S1 together withits body diode) is larger than that of the gate drive.

2) In explaining the operation of the flyback cell, it is as-sumed in Fig. 5(b) that when S1 is turned off, capacitor

is charged only by In fact, if we refer backto Fig. 2, we can see that is actually first chargedby both and when is smaller than[during as shown in Fig. 6(e)]. It is onlywhen that is charged by alone.This accounts for the break point of at , asshown in Fig. 6(d).

3) When falls below at [Fig. 6(d)], willalso be able to flow. However, since the initial value of

is small, it can hardly affect the waveform ofwithin the duration

4) , where isswitching period It is during time intervalsto and to that interactions between these twoconverter cells occur. However, since these two intervalsare relatively short with respect to the switching cycle,the SSIPP can still be approximately considered as twoindividual converter cells when performing dc analysisand developing design criteria.

III. D ESIGN CRITERIA

Since the proposed converter can be approximately decou-pled into two converter cells, as shown in Fig. 3, the design ofthe converter can also be decomposed into the design of twoindependent cells. The voltage and equivalent resistance

shown in Fig. 3 are two important parameters that mustfirst be determined.


(a) (b)

(c)

Fig. 4. Topological states of Boost cell in DCM.

(a) (b)

(c) (d)

(e) (f)

Fig. 5. Topological states of flyback cell with active clamping and soft switching.

A. Determination of and

Assume that the effective duty cycle of the Boost cell andthat of the flyback cell are approximately the same and equal to

It is required that the Boost cell [Fig. 3(a)]

always remains in DCM operation. Referring to Fig. 6, we

define as Assuming that is constant within


Fig. 6. Steady-state waveforms of the converter.

Fig. 7. Voltage ratioM as a function of duty cycleD:

one switching cycle, the input inductor current is given by

(2.1)

(2.2)(2.3)

where

(3)

Since the transformer inductance is quite large, weassume that the average value of denoted as is

Fig. 8. Voltage ratio(nVo=Vg) as a function of duty cycleD:

Fig. 9. Relationship between� and duty cycleD:

constant. So, the energy flowing into the flyback cell in onecycle is

(4)

and the output energy of the flyback cell in one cycle is

(5)

where is the current passing through It should be notedthat the approximation made in (5) is acceptable because thetransition intervals to and to are practically quiteshort (as can be seen in the measured waveforms in Fig. 12).Since the average value of is zero, (5) becomes

(6)

Letting we have

(7)


Equation (7) gives the steady-state relationship between theinput and output voltages of the flyback cell. (It is the sameas that of a flyback converter without active clamping.)

Now, we assume that is an ac input followed by a bridgerectifier so that the input voltage to the Boost cell is given by

(8)

where is the period of ac line voltage. The energyabsorbed from the ac line in a half cycle can bededuced from (2) and (8) (by assuming a constantwithineach switching cycle)

(9)

The energy delivered to the loadduring the same half cycleis

(10)

Equating (9) with (10) gives

(11)

or

(12)

where

(13.1)

(13.2)

(13.3)

Note that cannot be solved analytically and is load depen-dent. However, based on (12), design curves can be generatedfor design purposes.

A set of design curves showing the relationship betweenduty cycle and the voltage ratio for different values of

is shown in Fig. 7. It can be seen that for a given valueof a larger gives a lower voltage ratio (and thereforea lower This is similar to that of a conventional Boost-flyback SSIPP employing a flyback converter cell in CCMoperation [16].

From Fig. 7 and (7), we can obtain a set of design curvesshowing the relationship between duty cycleand the input-output voltage gain for different values of asshown in Fig. 8. It can be seen that for a giventhe duty cycle decreases with an increase of loadingresistance

On the other hand, , the equivalent loading resistanceas seen by the Boost cell, can be easily found as

(14)

B. Design of the Boost Cell

It is critical to select a proper input inductor to ensurethat the Boost cell operates in DCM even when the loadingcurrent is maximum. Based on the design criteria developedin [17], we can find that must satisfy the condition of

(15)

where can be determined from Fig. 7. Combining (13.2),(14), and (15) gives

(16)

The that satisfies the above inequality for any given dutycycle is given in the unshaded area in Fig. 9.

It should be noted that although it is desirable to have alarge in order to reduce the voltage stress, it is not practicalto have because it will then imply a duty cycle lessthan 0.23 (as shown in Fig. 9).

C. Design of the Flyback Cell

Since the flyback cell functions as a dc–dc regulator, itsdesign is similar to that of an actively clamped flyback dc–dcconverter [2], except the selection of switch S1.

1) Selection of Main Switch S1:The maximum voltageacross the main switch S1 is equal to Applying the volt-second balance law to the power transformer T, we obtain theaverage voltage across the clamping capacitor

(17)

The maximum current through S1, which occurs at around, is given by (referring to Fig. 2 for the circuit and

Fig. 6 for the waveform)

Magnetizing current of transformer T

Peak current in

Therefore

(18)

2) Selection of Clamping Capacitor : The resonant fre-quency formed by and should be sufficiently low sothat is kept almost constant and is discharged almostlinearly between and [as shown in Fig. 6(e)]. Thus, thefollowing relationship should be guaranteed:

(19)


(a)

(b)

Fig. 10. Experimental circuit and gate-drive waveforms.

3) Selection of Transformer Inductance : The presenceof the active-clamping circuit does not significantly alterthe waveform of the primary current of the flyback cell(as compared with a standard flyback converter in CCMoperation). Therefore, the inductance can be found fromthe general inequality expression

(20)

4) Time Delay Between S1 and S2 Gate Signals:In orderto achieve soft turn-on, the switch S1 must be turned onbefore becomes positive [during as shownin Fig. 5(f)]. The delay time between the turn-off of S2 andthe turn-on of S1 is therefore critical. The optimum value ofthis delay is one quarter of the resonant period formed of

and

(21)

where is defined in Fig. 6(a). The delay time betweenthe turn-off of S1 and the turn-on of S2 [shown as TD2 inFig. 6(a)] is not so crucial because the time interval for whichpositive flows through the body diode of S2, as shown inFig. 5(d), is quite long. Usually it is chosen that is a littlelarger than

5) Value of : In order to achieve soft turn-on forS1, there must be sufficient energy stored in the resonantinductor to completely discharge the resonant capacitor

This requires

or (22)


Fig. 11. Simulated waveforms of Boost-flyback SSIPP with active clamping and soft switching.

where

(23)

A slightly larger may be selected to ensure soft switchingcondition.

D. Design Example

Due to the nonlinear relationships among and thedesign of the SSIPP is not straightforward and may require anumber of iterations. The objective of such iterations is toensure the following.

1) The Boost cell remains in DCM operation even whenthe loading current is maximum and the input voltageis minimum.

2) The voltage is not excessively large, even when theloading current is minimum and the input voltage ismaximum.

Based on the design curves shown in Figs. 7–9, we shallillustrate the design procedure with a simplified example.Suppose the SSIPP is to be operated from a 110-Vrms 50-Hz ac mains and the output voltage required is 28-V dc for aloading resistance that may vary between 10–30

1) First Iteration: It is first assumed that the transformerhas a turns ratio of three. (This is estimated from the inputto output voltage ratio.) Therefore

It is also assumed that (A large value of helps toreduce the voltage stress.) From Fig. 8, it is found that for

we have However,


Fig. 12. Measured waveforms ofvGS1; iL; and iD3: Scales:vGS1 (10V/div); iL (2 A/div); iD3 (5 A/div). Time scale: 2�s/div:

Fig. 9 shows that the point lies in theshaded area, implying a CCM operation of the Boost cell,which is not allowed.

2) Second Iteration:We now reduce the value of from 1to 0.7. From Fig. 8, it is found that for

is 0.24. From Fig. 9, we also find that the pointis within the unshaded area, implying

a DCM operation of the Boost cell. This is an acceptableoperating point. By assuming kHz, the value of

can be found from (13.2)

H

Next, we estimate the voltage and current stresses of thecircuit. For , implying , we find fromFig. 7 that if will be 1.7. Thus, from (13.3),

The corresponding voltage stress of S1, foundfrom (17), is 349 V. The maximum current through S1 underthis condition, found from (18), is 4.9 A. When ,implying we estimate from Fig. 8 that is equal to0.17 (found from the interception of an interpolatedcurve with the line). From Fig. 7, we findthat for , is 2.7, implying a of373 V. The corresponding voltage stress of S1 is 457 V. It isassumed that the results of the second iteration are acceptable.

It should be noted that when the loading resistance is furtherincreased above the value of , the voltage will increasefurther. This is an inherent problem of all SSIPP. Possiblemethods to solve this problem are as follows.

1) Use frequency modulation [14] (in addition to duty-cyclemodulation) to increase the switching frequency of theconverter when reaches an upper limit.

2) Design the flyback cell in such a way that it will enterDCM operation when is above a certain value [15],[16].

Fig. 13. Measured waveforms ofvGS1; vCr ; and iLr : Scales:vGS1 (10V/div); vCr (100 V/div); iLr (2 A/div). Time scale: 2�s/div:

Fig. 14. Measured input line voltage (CH1) and line current (CH2) wave-forms. Scales: CH1 (50 V/div); CH2 (1 A/div). Time scale: 5 ms/div.

Both methods are helpful to reduce any further increaseof

IV. SIMULATIONS AND EXPERIMENTAL RESULTS

Based on the design criteria given above, a circuit, as shownin Fig. 10(a), with the following parameters (similar to those ofthe design example in Section III), was designed, simulated,constructed, and tested:

1) Input voltage V AC;2) Output voltage V dc;3) Output power W;4) H;5) F;6) F;7) pF;8) H;


Fig. 15. Efficiency comparison of Boost-flyback SSIPP with (a) active clamping and (b) passive snubber.

9) F;10) Switches S1 and S2: IRF840;11) Diodes D1 and D2: MUR1560;12) Diode D3: MBR10100;13) Transformer turns ratio ;14) Transformer magnetization inductance H;15) Switching frequency kHz.

A. Simulations

Fig. 11 shows the simulated waveforms of the converterwith an input voltage of 110 V dc. As indicated by thewaveform of the Boost cell operated in DCM. When S1was off, the voltage across it (namely, was clampedat , which was nearly constant. It was also obvious thatthe rising and falling time intervals of (namely, thesoft-switching time intervals) were short compared with theswitching period, resulting in an almost square waveNote that each time after S1 was turned on, the output rectifiercurrent decreased linearly with a finite slope so that thereverse recovery problem of the output rectifier D3 was trivial.

B. Experimental Results

The experimental circuit shown in Fig. 10(a) was built toverify the theory and simulations described above. (The circuitparameters were all the same as those for simulation.) Thecapacitance consists of only the output capacitances of S1and S2, and no external capacitor is added.consists ofthe leakage inductance of the power transformerand anexternal inductor of 20 H

It should be interesting to note that only a single con-trol/driver circuit was required to drive the main switch S1 andthe auxiliary switch S2. [The predicted gate-drive waveformsare shown in Fig. 10(b).] The delay times and of thedriving signals were adjusted by and (together withthe input capacitances of the MOSFET’s). Thus, the control

circuit [within the dashed box in Fig. 10(a)] could be thesame as that of an SSIPP without active clamping. In theexperimental circuit, a UC3842 was used to generate the PWMcontrol signal and a UC3709 was used as the driver.

Before putting the circuit to an ac line test, the dc operationof the circuit was first studied. The input voltage () usedwas 110-V dc, and the output voltage was regulated at 28-Vdc. The output power was 80 W. Fig. 12 shows the measuredwaveforms of the gate-drive voltage of S1 the inputinductor current and the output rectifier current Fig. 13shows the measured waveforms of the drain-to-source voltageof S1 (namely, and the resonant inductor current ,as defined in Fig. 10(a). These waveforms agreed well withtheoretical prediction (Fig. 6) and simulation (Fig. 11). Thewaveform of the input inductor current , shown in Fig. 12,indicated that the Boost cell operated in DCM. The waveformof the output rectifier current , also shown in Fig. 12,demonstrated that the rectifier should have no reverse recoveryproblem. From Fig. 13, it can be seen that the drain-to-source voltage of S1 fell to zero before its gate-drivevoltage rose above the threshold voltage of the MOSFET, thusacquiring soft turn-on.

The PFC performance of the circuit was then tested for anac input ( ) of 110 Vrms (50 Hz) and a dc output of 28 V(regulated). The input voltage and input current waveforms foran output power of 80 W are shown in Fig. 14. The measuredpower factor and the total harmonic distortion (THD) of theline current were 0.98% and 13%, respectively. The measuredefficiency was 86.5%.

For the purpose of comparison, a circuit without activeclamping, but with a passive snubber, was also constructed andevaluated. In the circuit with the passive snubber, the externalpart of inductor [shown in Fig. 10(a)] was removed, and thesnubber components were so selected that the voltage stressof switch S1 was 370 V at full load (80 W) (compared with340 V for the circuit with active clamping, as indicated by


the waveform in Fig. 13). The conversion efficiencies ofthe two circuits were then compared and plotted in Fig. 15. Itis obvious that the circuit with active clamping has a higherconversion efficiency (higher by 3%–4%).

V. CONCLUSIONS

The operation, characteristics, and design considerations ofBoost-flyback SSIPP with active clamping and soft switchinghave been studied. The incorporation of the active-clampingcircuit provides a mechanism for clamping the switch voltagestress, recycling the energy trapped in the transformer leakageinductance, and achieving soft switching of active switches andrectifier. The proposed circuit uses a common control/drivercircuit for both the main and auxiliary switches. Simulationand experimental results show excellent agreement with thetheoretical predictions. It is expected that similar clampingcircuits may also be added to other SSIPP to improve theirperformance.

ACKNOWLEDGMENT

The authors wish to thank Y. L. Cheng for constructingthe hardware and carrying out the experimental work for thisproject.

REFERENCES

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[3] K. Yoshida et al., “Zero voltage switching approach for flyback con-verter,” in Proc. IEEE INTELEC’92, pp. 324–329.

[4] K. H. Liu and Y. L. Lin, “Current waveform distortion in power factorcorrection circuits employing discontinuous-mode Boost converters,” inProc. IEEE PESC’89 Rec., pp. 825–829.

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[6] P. Gil et al., “Analysis of the zero-current-switched quasiresonantconverters used as power factor preregulators,” inProc. IEEE PESC’92Rec., pp. 1052–1060.

[7] D. Y. Huh et al., “New group of ZVS PWM converters operable onconstant frequency and its application to power factor correction circuit,”in Proc. IEEE PESC’92 Rec., pp. 1441–1446.

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[9] Y. Jiang et al., “Single-stage single-phase parallel power factor correc-tion scheme,” inProc. IEEE PESC’94 Rec., pp. 1145–1151.

[10] M. H. Kherulawaet al., “A fast-response high power factor converterwith a single power stage,” inProc. IEEE PESC’91 Rec., pp. 769–779.

[11] J. P. Gegner, “High power factor ac-to-dc converter using a reactiveshunt regulator,” inProc. IEEE PESC’94 Rec., pp. 349–355.

[12] R. Srinivasanet al., “Single-phase parallel power processing schemewith power factor control,”Int. J. Electron., vol. 80, no. 2, pp. 291–306,1996.

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[14] M. M. Jovanovicet al., “Reduction of voltage stress in integrated high-quality rectifier-regulators by variable-frequency control,” inProc. IEEEAPEC’94, pp. 569–575.

[15] M. J. Willers et al., “A BIFRED converter with a wide load range,” inProc. IEEE IECON, 1994, pp. 226–231.

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[17] D. Simonettiet al., “Analysis of the conduction boundary of a Boost PFPfed by universal input,” inProc. IEEE PESC’96 Rec., pp. 1204–1208.

Yim-Shu Lee, for a photograph and biography, see this issue, p. 992.

Bo-Tao Lin, for a photograph and biography, see this issue, p. 992.

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IEEE Transactions on Power Electronics Volume 12 Issue 6 1997 [Doi 10.1109_63.641500] Yim-Shu Lee; Bo-Tao Lin -- Adding Active Clamping and Soft Switching to Boost-flyback Single-stage