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8/18/2019 An1683x Flyback Design
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Power Conversion
Version 1.2 , March 2000
Application Note
AN-SMPS-1683X-1
CoolSET™
TDA16831...-34 for OFF-Line Switch Mode Power Supplies
Author: Harald Zöllinger
Published by Infineon Technologies AG
http://www.infineon.com
N e v e r s t o p t h i n k i n g
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Contents:
Circuit description ....................................................................................................... 2
Operating Principles.................................................................................................... 2
Circuit diagram:........................................................................................................ 4
Design procedure for fixed frequency Flyback Converter with TDA16831...-34
operating in discontinuous current mode. ................................................................... 5
Define input parameters:.......................................................................................... 5
Input Diode Bridge: .................................................................................................. 5
Determine Input capacitor:....................................................................................... 5
Transformer Design: ................................................................................................ 7
Winding design: ....................................................................................................... 8
Output Rectifier:..................................................................................................... 10
Output Capacitor:................................................................................................... 10
Output Filter: .......................................................................................................... 11
VCC-Supply: .......................................................................................................... 11
Calculation of snubber network:............................................................................. 12
Calculation of losses: ............................................................................................. 13
Voltage regulation loop: ......................................................................................... 14
Regulation loop:..................................................................................................... 15
Transfer characteristics of regulation loop elements:............................................. 15
Transformer Construction ...................................................................................... 20
Layout Recommendation:...................................................................................... 21
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Circuit description
The TDA 1683X is a current mode pulse width modulator with an integrated CoolMOS Transistor. Itmeets the need for minimum external control circuitry for a flyback application.Current mode control means that the current through the MOS transistor and flyback transformer iscompared with a feedback signal derived from the output voltage of the flyback application. The resultof that comparision determines the on time of the MOS transistor.To minimize external circuitry the current sense circuitry is integrated within the CoolSET controller.The oscillator resistor and capacitor which determine the switching frequency are also integrated,reducing the external connections. Special efforts have been made to compensate temperaturedependancy and to minimize tolerances of the passive components.
Operating PrinciplesThe TDA1683X is designed for a current mode flyback configuration in discontinous current mode.The control circuit has a fixed frequency, and the duty cycle of integrated Cool-Mos switch iscontrolled to maintain a constant output voltage.The diagram below (Fig. 1) shows the input voltage and the primary and secondary transformercurrent.When the Cool-Mos transistor is turned on, the start of all windings on the transformer will go positive.The rectifier diode on the secondary side will be reverse-biased and will not conduct. Therefore nocurrent will flow in the secondary while the Mosfet is turned on. During this phase energy is beingstored in the primary winding inductance and the transformer may be treated as a simple seriesinductor. The diagram shows that there will be a linear increase of primary current (Ipri) while theprimary Cool-MOS switch is on.When the Cool-MOS transistor is turned off, the voltage will reverse on all windings (flyback action)until clamped by the secondary side widing through the secondary rectifier diode. Now the secondaryrectifier diode will conduct, and the magnetizing energy in the core will now transfer to the outputduring the reset interval.This current will decrease from it’s peak value to zero, as shown in the diagram (Isec). In this period thecomplete stored energy in the primary inductance will be transferred to the secondary (neglectinglosses), before the next store cycle starts. The secondary voltage is “reflected” back through thetransformer turns ratio to the primary winding and added to the input voltage (VIN+VR). Additionaltransient voltage may appear on the primary winding due to energy stored in uncoupled “leakage”inductance in the primary winding which isn’t clamped by the secondary side winding.If the flyback current does not reach zero before the next “on” -cycle the converter is operating incontinous current mode. When this system reverts to the continous operation, the transfer function ischanged to a two pole system with low output impedance and additional design rules becomeimportant.
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Voltage and Current waveforms in discontinous mode operation:
VIN = VINMIN VIN > VINMIN
Fig. 1
VINMIN + VR
VINMIN
0 T
IPEAK IPRI
VIN + VR
VIN
0
IPEAK IPRI
ISEC
IPEAKISEC
IPEAK
Light load full load
tOFF tON TOFF T tON
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Circuit diagram:
Fig. 2
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Design procedure for fixed frequency Flyback Converter withTDA16831...-34 operating in discontinuous current mode.
Procedure Example
Define input parameters:
Minimal AC input voltage : VacminMaximal AC input voltage : VacmaxLine frequency facMax. Output power: POmaxMin. Output power: POminOutput voltage: VOUTOutput ripple voltage: VOrippleReflection voltage: VREstimated efficiency: ηDC ripple voltage: Vripple
Auxiliary Voltage Vaux.Optocoupler Gain: GCUsed CoolSET
85V270V50Hz40W 1W12V0,05V
100V0,820V12V1TDA16834 for 40W @ 25°C
There are no special requirements imposed on theinput rectifier and storage capacitor in the flybackconverter. The components will be selected to meet
the power rating and hold-up requirements.Maximum input power:
η
OUT
MAX
PP = (Eq 1)
Input Diode Bridge:
ϕ cosmin ⋅=
ac
MAX PRMS
V
P I (Eq 2)
2max ⋅= acdcinpk V V (Eq 3)
Determine Input capacitor:
Minimum peak input voltage at ”no load” condition
2minmin ⋅= ac pk dc V V (Eq 4)
W W
P MAX 508,0
40==
AV
W I PRMS 98,0
6,085
50=
⋅=
V V V dcinpk 3822270 =⋅=
V V V pk dc 120285min =⋅=
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ripple pk dcdc V V V −= minmin (Eq 5)
Calculating discharging time at each half line cycle:
+⋅=90
arcsin
15 min
min
pk dc
dc
D
V
V
msT (Eq 6)
Required energy at discharging time:
D MAX IN T PW ⋅= (Eq 7)
Calculating input capacitor value:
2min
2min
2
dc pk dc
IN IN
V V
W C
−
⋅= (Eq 8)
Alternative a rule of thumb on choosing CIN
Input voltage CIN115V 2µF/W230V 1µF/W85V ...270V 2 ...3µF/W....................
IN
IN
pk dcdcC
W V V
⋅−=
22minmin (Eq 9)
Select a capacitor out of Siemens/Epcos Databookof Aluminium Electrolytic Capacitors.
The following types are preferred:
For 85°C Applications:Series B43303-........ 2000h lifetime B43501-........ 10000h lifetime
For 105°C Applications:Series B43504-........ 3000h lifetime B43505-........ 5000h lifetime
we choose a ripple voltage of 20V
V V V V dc 10020120min =−=
msV
V
msT D 1,890
120
100arcsin
15 =
+•=
WsmsW W IN 41,01,850 =⋅=
F V V
WsC IN µ 186
1000014400
41,0222
=−
⋅=
F F W µ µ 150350 =⋅
We choose 180µF 400V
V F
WsV V dc 2,99
180
41,0214400 2min =
⋅−=
µ
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Transformer Design:
Calculation of peak current on primary inductance:
maxmin
2
DV
P I
dc
MAX LPK ⋅
⋅= (Eq 10)
3
max D I I LPK LRMS ⋅= (Eq 11)
Calculating of primary inductance within limit of
maximum Duty-Cycle :
f I
V D L
LPK
dcP ⋅
⋅= minmax (Eq 12)
Select core type and inductance factor (AL) fromSiemens/Epcos ferrite Databook or CD-ROMPassive Components.
Fix maximum flux density:Bmax ≈ 0,2T ...0,3T for ferrite cores depending on corematerial.
We choose 0,2T for material N27
The primary turns can be calculated as:
L
PP
A
L N = (Eq 13)
Number of secondary turns can be calculated as:
( )
R
FDIODE OUT P
V
V V N Ns
+⋅= (Eq 14)
Note the internally limited Duty Cycle!!
Dmax = 0,5 (see datasheet TDA16834)
AV
W I LPK 14,2
47,099
502=
⋅⋅
=
A A I LRMS 85,03
5,014,2 =⋅=
H kHz A
V LP µ 217
10014,2
9947,0=
⋅⋅
=
Selected core: E 32/16/9Material = N27AL = 244 nH
s = 0,5 mmAe = 83 mm
2
AN = 108,5 mm2
lN = 64,4 mm
weight ≈ 30gPV = 190mW/g (200mT, 100kHz, 100°C)
85,29244
217==
nH
H N P
µ turns
we choose Np = 30 turns
( )81,3
100
7,01230=
+⋅=
V
V V Ns
we choose Ns = 4 turns
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Number of auxiliary turns can be calculated as:
( )
R
FDIODE auxP
aux V
V V N
N
+⋅
= (Eq 15)
Postcalculation of primary inductance, primary peakcurrent, max. flux density and gap:
lPP A N L ⋅= 2 (Eq 16)
f Lp
DV I
dc Lpk
⋅
⋅= maxmin (Eq 17)
eP
LPK P
A N
I L B
⋅⋅
=max (Eq 18)
P
eP
L
A N s
⋅⋅⋅⋅=
− 27104 π
(Eq 19)
Winding design:
(see also page 20Transformer Construction)
The primary winding of 30 turns has to be split into15+15 turns in order to get best coupling betweenprimary and secondary winding.
The effective bobbin width and winding cross sectioncan be calculated:
M BW BW e ⋅−= 2 (Eq 20)
BW
BW A A
e N Ne
⋅= (Eq 21)
Calculate copper section for primary and secondarywinding:
The winding cross section AN has to be splitted intothe number of windings.Primary winding 0,5Secondary winding 0,45Auxiliary winding 0,05
( )81,3100
7,01230
=+⋅
= V V V
N aux
we choose Naux = 4 turns
H nH LP µ 220244302 =⋅=
AkHz H
V I Lpk 12,2
100220
47,099=
⋅⋅
= µ
mT mm
A H B 187
8330
12,2220
2max =
⋅
⋅=
µ
mmmH
mms 43,0
22,0
8330104 227=
⋅⋅⋅⋅=
−π
From bobbin datasheet E32/16/9: BW = 20,1mm
Margin determined: M = 4mm
mmmmmm BW e 1,12421,20 =⋅−=
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Copper space factor f Cu :0,2 ....0,4
BW N
BW f A A
P
eCu N P ⋅
⋅⋅⋅=
5,0 (Eq 22)
( )( )( )d AWG log28277,197,9 ⋅−⋅= (Eq 23)
BW N
BW f A A
s
eCu N s ⋅
⋅⋅⋅=
45,0 (Eq 24)
BW N
BW f A A
aux
eCu N aux ⋅
⋅⋅⋅=
05,0 (Eq 25)
With the effective bobbin width we check the numberof turns per layer:
P
Pd
BWe N = (Eq 26)
We calculate the available area for each winding:
Used for calculation: f Cu =0,3
22
31,01,2030
1,123,05,1085,0mm
mm AP =⋅
⋅⋅⋅=
⇒ diameter dp ≈ 0,64mm ⇒ 22 AWG
22
20,21,204
1,123,05,10845,0mm
mm As =
⋅
⋅⋅⋅=
⇒ diameter ds 2 x 0,8mm ⇒ 2 x 20 AWG
22
24,01,204
1,123,05,10805,0mm
mm Aaux =⋅
⋅⋅⋅=
⇒ diameter da ≈ 0,64mm ⇒ 22 AWG
Primary:
1764,0
1,12==
mm
mm N P turns per layer
⇒ 2 layer needed
Secondary:
48,02
1,12=
⋅=
mm
mm N S turns per layer
Aux.:
Can be neglected !
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Output Rectifier:
The output rectifier diodes in flyback converters aresubject to a large peak and rms current stress. Thevalues depend on the load, leakage inductance,operating mode and output capacitor ESR.
Calculation of the maximum reverse voltage:
⋅+=
P
S dcinpk OUT RDIODE
N
N V V V (Eq 27)
Calculation of the maximum current:
S
P LPK SPK
N
N I I = (Eq 28)
max31 D I I SPK SRMS ⋅⋅= (Eq 29)
V V V V RDIODE 9,6230
438212 =
⋅+=
A A I SPK 9,154
30
81,2 ==
A A I SRMS 7,647,0319,15 =⋅⋅=
Output Capacitor:
Output capacitors are highly stressed in flybackconverters. Normally the capacitor will be selected for3 major parameters: capacitance value, low ESRand ripple current rating.
Max. voltage overshoot: ∆VOUT
Number of clock periods: ncp
f V
I C
OUT
OUTMAX
OUT ⋅∆
⋅=
cpn (Eq 30)
Select a capacitor out of SIEMENS/Epcos Databookof Aluminium Electrolytic Capacitors.
The following types are preferred:
For 85°C Applications:Series B41826-........ 4000h lifetime
For 105°C Applications:Series B41856-........ 2000h lifetime
To calculate the output capacitor, it is necessary to fixthe maximum voltage overshoot in case of switchingoff @ maximum load condition.After switching off the load, the regulation loopneeds about 5...10 periods of internal clock to reduce
the duty cycle.
V V OUT 5,0=∆
ncp = 5
F kHzV
AC OUT µ 333
1005,0
533,3=
⋅⋅
=
We select 470µF 25V:
B41826-A5477-M
ESR ≈ Zmax = 0,06Ω @ 100kHzIacR = 2,2A
ISRMS = 6,7A ⇒ 3 capacitor in parallel needed!
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Output Filter:
The output filter consists of one capacitor and oneinductor in a L-C filter topology.
Zero frequency of output capacitor and associatedESR:
OUT ESR
ZCOUT C R
f ⋅⋅⋅
=π 2
1 (Eq 31)
Calculating the needed inductance for substitute thezero of the output capacitor:
ZCOUT
ESROUT
f
R L
⋅⋅=
π 2 (Eq 32)
VCC-Supply:
Start-up Resistor:
ICCLmax = max. Quiescent Current
Il = VCC-Capacitor Load-Current
CVCC = Value of VCC-Capacitor
lCCL
dcstart
I I
V R
+=
max
min (Eq 33)
Start-up Time:
l
CCH VCC start
I
V C t
⋅= (Eq 34)
Internal Zener Diode:Depending on the transformer construction and loadcondition the auxiliary supply voltage varies within an
operating range. If VCC exceeds VZ (16V), theinternal zener diode conducts. In this case we haveto observe the internal power dissipation limits oruse an external zener diode on VCC pin.
kHzF
f ZCOUT 6,547006,02
1=
⋅Ω⋅⋅=
µ π
H kHz
LOUT µ π
56,036,52
06,0=
⋅⋅⋅Ω
=
ICCLmax = 80µA
Il = 40µA
CVCC = 22µF
Ω=+
= k A
V Rstart 827
4080
99
µ
R6 = R7 =1/2 Rstart = 413,5kΩ
Choose: 410kΩ
s A
V F t start 6,6
40
1222=
⋅=
µ
µ
Before the IC can be plugged into the applicationboard, the VCC capacitor has always to bedischarged!
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Calculation of snubber network:
Rdc BRDSS snub V V V V −−= max (Eq 35)
For calculating the snubber network it is neccesary toknow the leakage inductance. Most common way isto have the value of the leakage inductance inpercent of the primary inductance. If it is known thatthe transformer construction is very consistent,measuring the primary leakage inductance byshorting the secondary windings will give an exactnumber, assuming the availability of a good LCRanalyser.
% x Lp L LK ⋅=
( ) snubsnub R LK LPK
snubV V V
L I C
⋅+⋅
=2
(Eq 36)
( )
f I L
V V V R
LPK LK
Rsnub
snub R
⋅⋅⋅
−+=
2
22
5,0 (Eq 37)
V V V V V snub 118100382600 =−−=
In our example we choose 5% of primary inductancefor leakage inductance.
H H L LK µ µ 11%5220 =⋅=
( ) nF
V V V
H AC snub 9,1
118118100
1112,2 2=
⋅+⋅
= µ
≈ 2,2nF
( )Ω=
⋅⋅⋅
−+= k
kHz A H
V V V Rsnub 1,15
10012,2115,0
1001001182
22
µ ≈15k
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Calculation of losses:
Input diode bridge:
2⋅⋅= F PRMS DIN V I P (Eq 38)
Calculation of copper resistance:
P
P N PCu
A
p N l R 100
⋅⋅= (Eq 39)
Calculating of copper loss:
Cu MAX LPK Cu R D I P ⋅⋅⋅= 312 (Eq 40)
Output rectifier diode:
FDIODE SPK DDIODE V D
I P ⋅−
⋅=3
1 max (Eq 41)
MOSFET :TDA16834
COSS ≈ 40pFRDSON = 1,6Ω (@ 150°C)
Switching losses:
f V C P dcOSS SON ⋅⋅⋅= 2
min21 (Eq 42)
( ) r LPK RdcSOFF t f I V V P ⋅⋅⋅+⋅= min61 (Eq 43)
Conduction losses:
max2
31 D I RP LPK DSON D ⋅⋅⋅= (Eq 44)
W V AP DIN 96,12122,1 =⋅⋅=
Copper resistivity p 100 at 100°C = 0,0172Ωmm2 /m
Ω=Ω⋅⋅
= mmm
mmmmm RPCu 3,116
33,0
/ 2,17300644,02
2
Ω=Ω⋅⋅
= mmm
mmmmm RSCu 9,4
04,1
/ 2,1740644,02
2
mW m APPCu 2,823,1163147,049,4
2 =Ω⋅⋅⋅=
mW m APSCu 1949,43147,08,252
2 =Ω⋅⋅⋅=
∑ =+= mW mW mW PCu 2801942,82
W V P DDIODE 36,57,0
3
47,019,15 =⋅
−⋅=
mW kHzV pF PSON 25100994021 2 =⋅⋅⋅=
( ) mW nskHz AV V PSOFF 2103010012,21009961 =⋅⋅⋅+⋅=
W AP D 13,147,05,46,131 2 =⋅⋅Ω⋅=
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Voltage regulation loop:
Reference: TL431VREF =2,5V
IKAmin=1mA
Optocoupler: SFH617-3Gc = 1 ...2 ≡ CTR 100% ...200%VFD = 1,2VIFmax =10mA (maximum current limit)
Primary side:
Feedback voltage:Values from TDA16831...34 datasheet
Vrefint = 5.5V typ.
VFBmax = 4,8V
RFB = 3,7k typ.
FB
ref
FB R
V
I int
max = (Eq 45)
FB
FBref
FB R
V V I
maxint
min
−= (Eq 46)
Secondary side:
−= 121
REF
OUT
V
V R R (Eq 47)
the value of R2 can be fixed at 4,7k
( )
max3
)(
F
REF FDOUT
I
V V V R
+−≥ (Eq 48)
min
min3
4KA
FBFD
I
Gc
I RV
R
⋅+
≤ (Eq 49)
Fig. 3
Fig. 4
mAk
V I FB 5,17,3
5,5max =Ω=
mAk
V V I FB 19,0
7,3
8,45,5min =Ω
−=
k
V
V k R 86,171
5,2
127,41 =
−⋅=
( )k
mA
V V V R 83,0
10
)5,22,1(123 =
+−≥ ≈ 910R
k mA
mA RV
R 4,11
1
2,09102,1
4 =
⋅+
≤ ≈ 1,2k
FB3,7k
5,5V
VFB
R3 R1R4
R5
R2
C2
C1
TL431
Vout
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Regulation loop:
Fig. 5
Transfer characteristics of regulation loop elements:
3
73
R
k GK
C FB
⋅= Feedback
GC= Optocoupler gain
Vout
Vref
R R
RK VD =+
=21
2 Voltage Divider
( )
⋅
+⋅+
⋅⋅+⋅
⋅⋅⋅
⋅=
5
5
21
1
2
1)(
C R R
p
C R p R
R
f L
Z pF
ESR L
ESR L
L
P
PWM
PWR
η Power stage
ZPWM = Transimpedance ∆VFB / ∆ID
9
29
9
1
1)(
C L pC R p
C R p pF
ESR
ESR LC
⋅⋅+⋅⋅+
⋅⋅+= Output filter
( )
)251(121
21
2151)(
C R pC R R
R R p
C C R p pFr
⋅⋅+⋅⋅+⋅
⋅
+⋅⋅+= Regulator
_
+
KFBKVD
FPWR(p) FLC(p)
Fr(p)
Vout
Vref
VIN
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Zero’s and Poles of the transfer characteristics:
Poles of powerstage @ min. and max. load:
Ω=== 6,340
12 2
max
2
W
V
P
V R
O
OUT LH Ω=== 144
1
12 2
min
2
W
V
P
V R
O
OUT LL
5
1
C R foh
LH ⋅⋅=π
HzF
foh 7,6214106,3
1=
⋅Ω⋅=
µ π
5
1
C R fol
LL ⋅⋅=π
HzF
fol 57,11410144
1=
⋅Ω⋅=
µ π
The gain of the optocoupler stage KFB and the voltage divider KVD we use as a constant.
3
73
R
k GK
C FB
⋅= KFB = 6,6 ⇒ GFB = 16,4db
Vout
Vref
R R
RK VD =+
=21
2 KVD = 0,208 ⇒ GVD = -13,6db
With adjustment of the transfer characteristics of the regulator we want to have equal gain within the
operating range and to compensate the pole fo of the powerstage FPWR(ω).
Because of the compensation of the output capacitors zero (see page 10 Eq31, Eq32) we neglect this zeroand the LC-Filter pole.So the transfer characteristics of the power stage is reduced to a single pole response.
In order to calculate the gain of the open loop we have to select the crossover frequency.
We calculate the gain of the Power-Stage with max. output power at the selected crossover frequencyfg = 3kHz:
ZPWM of TDA16834 =1,3 V/A
( )
−⋅+
⋅+−⋅⋅=
−−21 16,011
1T
t
T
t
ON
ON
PWM ON PWM
ON ON
eeT T t t
Z t Z (formula according data sheet page 12)
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with this formula we calculate ZPWM @ max. duty cycle:
( ) A
V eensnss
s A
V t Z ns
s
ns
s
ON PWM 8,116,08508507,47,4
13,1 200
7,4
850
7,4
=
−⋅+
⋅+−⋅⋅=
−− µ µ
µ µ
Gain @ crossover frequency:
+
⋅⋅⋅⋅
⋅=2
1
1
2
1)(
fo
fg
f L R
Z fgF
p L
PWM
PWR
η
065,0
7,62
30001
1
2
8,01002206,3
8,1
1)3(
2=
+
⋅⋅⋅⋅
⋅= kHzuH R
kHzF PWR
⇒ GPWR(3kHz) = -23,7db
Transfer characteristics:
Fig. 6
1 10 100 1 103
1 104
1 105
50
0
5050
50
G PWR( )ω
Gr( )ω
G FB
G MOD
0
.1 1051 ω ( )i
.2 π
GPWR(ω)
Gain[db]
GVD
Gr(ω)
GFB
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At the crossover frequency we calculate for the open loop gain:
Gol(ω) = Gs (ω) + Gr (ω) = 0.
With the equations of the transfer characteristics we calculate the gain of the regulation loop @ fg.
The gain of the regulation loop we calculate:
Gs = GFB + GPWR + GVD = 16,4db – 23,7db – 13,6db
Gs = -20,9db
We calculate the separate components of the regulator:
Gs (ω) + Gr (ω) = 0 ⇒ Gr = 0 – (-20,9db) = 20,9db
( )
)251(121
21
)2151)(
C R pC R R
R R p
C C R p pFr
⋅⋅+⋅⋅+⋅
⋅
+⋅⋅+=
( )
21
215log20
R R
R R RGr
⋅+⋅
⋅= ⇒ 21
21105 20
R R
R R R
Gr
+⋅
⋅=
k k R 7,4172,3105 209,20
=⋅= ≈ 43k
252
1
C R fp
⋅⋅⋅=
π ⇒
fg RC
⋅⋅⋅⋅=
252
12
π fp = 2*fg
pF kHzk
C 6176432
12 =
⋅⋅⋅=
π ≈ 680pF
In order to have enough phase margin @ low load condition we select the zero frequency of compensationnetwork at the middle between min. and max. load pole of power stage.
oh
ol
f
f
ohom f f log5,0
10⋅
⋅= Hz Hz f om 92,9107,62 7,62
57,1log5,0
=⋅=⋅
( )2152
1
C C R fz
+⋅⋅⋅=
π ⇒ 2
52
11 C
fom RC −
⋅⋅⋅=
π
nF pF Hzk
C 38468092,9432
11 =−
⋅⋅⋅=
π ≈ 390nF
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Open Loop Gain
Fig. 7
Open Loop Phase
Fig. 8
1 10 100 1 103
1 104
1 105
50
0
50
70
60
Gr( )ω
Gs( )ω
G( )ω
0
.1 1051 ω ( )i
.2 π
1 10 100 1 103
1 104
1 105
180
142
104
66
28
1010
180
φr( )ω
φs( )ω
φ( )ω
0
.1 1051 ω ( )i
.2 π
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Transformer Construction
The winding topology has a considerable influence on the performance and relaibility of thetransformer.To reduce leakage inductance and proximity to acceptable limits, the use of a sandwich construction isrecommended.In order to meet international safety requirements a transformer for off-line power supply must haveadequate insulation between primary and secondary winding.This can be achived by using a margin wound construction or using triple insulated wire for thesecondary winding.The creepage distance for universal input voltage range is typically 8mm. This sets a minimum marginwidth as a half of the creepage distance to 4mm. Additional the neccesary insulation between primaryand secondary winding is provided using three layers of basic insulation tape.
Example of winding topology for margin wound transformers:
Fig. 9
Example of winding topology with triple insulated wire for secondary winding:
Fig. 10
BW* : value from bobbin datasheet
Primarysecond half
BW*
Primaryfirst half
Auxiliary
SecondaryTriple InsulatedWire
Primarysecond half
Auxiliary
Secondary
margin margin
Triple insulation
Creepagedistance
BW*
BWe
Primaryfirst half
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Layout Recommendation:
Fig. 11
In order to avoid crosstalk between Power- and Signal-Path on the board we have to use careregarding the track layout when designing the PCB.
The Power-Path (see Fig. 11) has to be as short as possible and separated from the VCC-Path andthe Feedback-Path. All GND-Paths have to be connected together at pin 8 (star ground) (1 and 14 atG-type) of TDA16831...34.
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References
[1] Keith Billings, Switch Mode Power Supply Handbook
[2] Ralph E. Tarter, Solid-State Power Conversion Handbook
[3] R. D. Middlebrook and Slobodan Cuk, Advances in Switched-Mode PowerConversion
[4] Herfurth Michael, Ansteuerschaltungen für getaktete Stromversorgungen mitErstellung eines linearisierten Signalflußplans zur Dimensionierung der Regelung
[5] Herfurth Michael, Topologie, Übertragungsverhalten und Dimensionierung häufig eingesetzter Regelverstärker
[6] TDA16831 –4Off-line SMPS Controller with 600V CoolMOS on BoardDatasheet, Infineon Technologies
Revision HistoryApplication Note AN-SMPS-1683X-1Actual Release: V1.2 Date:13.03.2000 Previous Release: V1.1Page ofactualRel.
Page ofprev. Rel.
Subjects changed since last release
24 21 Formatting
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For questions on technology, delivery and prices please contact the InfineonTechnologies Offices in Germany or the Infineon Technologies Companies and
Representatives worldwide: see the address list on the last page or our webpage at
http://www.infineon.com
CoolMOS™ and CoolSET™ are trademarks of Infineon Technologies AG.
Edition 2000-03--03Published by Infineon Technologies AG,St.-Martin-Strasse 53,D-81541 München
© Infineon Technologies AG 2000.All Rights Reserved.
Attention please!
The information herein is given to describe certain components and shall not be considered as warranted characteristics.Terms of delivery and rights to technical change reserved.
We hereby disclaim any and all warranties, including but not limited to warranties of non-infringement, regarding circuits, descriptions and chartsstated herein.
Infineon Technologies is an approved CECC manufacturer.
Information
For further information on technology, delivery terms and conditions and prices please contact your nearest Infineon Technologies Office inGermany or our Infineon Technologies Representatives worldwide (see address list).
Warnings
Due to technical requirements components may contain dangerous substances. For information on the types in question please contact your
nearest Infineon Technologies Office.
Infineon Technologies Components may only be used in life-support devices or systems with the express written approval of InfineonTechnologies, if a failure of such components can reasonably be expected to cause the failure of that life-support device or system, or to affect thesafety or effectiveness of that device or system. Life support devices or systems are intended to be implanted in the human body, or to supportand/or maintain and sustain and/or protect human life. If they fail, it is reasonable to assume that the health of the user or other persons may be
endangered.
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24 f 24 AN SMPS 1683X 1
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