851

Comparison of Medium Voltage IGBT-based 3L-ANPC-VSCs

José A. Sayago*, Steffen Bernet* and Thomas Brückner** * Dresden University of Technology, Dept. of Electrical Engineering, Dresden, Germany

** Converteam GmbH, Berlin, Germany

Abstract — Recently the 3L-ANPC-VSC was introduced to overcome the unequal loss distribution and the resulting unsymmetrical semiconductor junction temperature distribution as main structural drawback of the 3L-NPC-VSC. This paper compares conventional 3L-NPC-VSCs with 3L-ANPC-VSCs using 3.3kV, 4.5kV, and 6.5kV IGBT modules for industrial Medium Voltage Drives for rated converter output voltages of VLL = 2.3kV, 3.3kV, and 4.16kV in a switching frequency range of fs = 300Hz to 1050Hz. Maximum achievable converter power, converter efficiency, switch utilization, active silicon area and material costs are determined and compared for given IGBT and diode modules. The comparison is the basis for a derivation of advantageous operating conditions of 3L-NPC-VSCs and 3L-ANPC-VSCs.

I. INTRODUCTIONToday the Three-Level Neutral Point Clamped Voltage

Source Converter (3L-NPC-VSC) is widely used in medium voltage drives (MVD) for industry, marine, mining, and traction applications [1], [2]. Its clearly superior output voltage quality is the NPC converter’s most distinct advantage over the two-level converter. One substantial disadvantage which had not yet been overcome was the unequal distribution of semiconductor losses among the devices in the conventional 3L-NPC-VSC. This unequal loss distribution yields an unequal semiconductor junction temperature distribution [4], [7].

To overcome the uneven loss distribution and to mitigate the resulting drawbacks, the Three-Level VSC with active NPC switches – named Active Neutral Point Clamped VSC (3L-ANPC-VSC) – was introduced in [4]. The topology is shown in Figure 1.

Figure 1. Circuit configuration of the 3L-ANPC-VSC

The 3L-ANPC-VSC is a derivative of the 3L-NPC-VSC, with two additional active switches named Tx5 and Tx6 in antiparallel to the NPC diodes. These additional active NPC switches enable new switch states and new commutations compared to the conventional 3L-NPC-

VSC. They allow specific utilization of the upper and lower path of the neutral tap and a deliberate distribution of conduction and switching losses. Thus, a significant increase of the output power or switching frequency can be achieved. Recent publications have proven the effectiveness of the 3L-ANPC-VSC for MV high-power applications including its modulation and operation [2], [3], [4], [6], [7]. A first commercial prototype was recently announced by ABB [5]. However, the inclusion of its two active NPC switches per phase leg together with their heatsinks and gate-drive units (GDUs) plus power supplies, leads to an increase of the total material costs in relation to the conventional 3L-NPC-VSC. This additional cost must be considered when assessing the overall performance, i.e. power and efficiency of MV converter topologies. This paper provides a comprehensive comparison of the ANPC converter with the conventional NPC converter under this perspective.

II. DATA OF POWER SEMICONDUCTORS, HEATSINKS AND GATE-DRIVE UNITS

Table I gives the most important data for the design and comparison of 3L-ANPC-VSCs including the installed switch power, active silicon area, and individual and total costs for semiconductors, heatsinks and gate-drive units. As given in Table I, semiconductor devices of the voltage classes 3.3kV, 4.5kV, and 6.5kV need to be applied in the 3L-NPC-VSC with output voltages of VLL = 2.3kV, 3.3kV, and 4.16kV, respectively. IGBT modules of these ratings are available from various manufacturers. For this benchmark, modules from Infineon Technologies [8], [9] (3.3kV and 6.5kV modules) and from Mitsubishi Electric [10] (4.5kV modules) have been selected. Please note that all considered IGBT modules are applied at high volume in medium voltage industrial and traction converters.

To enable a fair comparison of 3L-ANPC-VSCs for converter voltages of VLL = 2.3kV, 3.3kV and 4.16kV, 3.3kV, 4.5kV and 6.5kV IGBT modules with identical footprints (190 140 mm2), identical heatsinks (Eupec KW51 with Rthh-a = 6K/kW, vw = 6.2 liter/min) and comparable reliability and lifetime requirements have been assumed [11]. It should be noted, that all considered 3L-ANPC-VSCs are based on the state-of-the-art PEBBs (e.g. [12]).

Thus, the expense for the converter construction is almost constant at varying converter voltages. In contrast, the different switch utilizations will cause differences of the costs of power semiconductor devices, heatsinks and gate-drive units. It can be taken from Table I that both the installed switch power and the active silicon area of the 3L-ANPC-VSCs are increased by 29% due to the addition of two extra IGBT modules.

Vdc

2

Vdc

2

T11

T12

T13

T14

T15

T16

T21

T22

T23

T24

T25

T26

T31

T32

T33

T34

T35

T36va,b,c iph

D11

D12

D13

D14

D21

D22

D23

D24

D31

D32

D33

D34

D15

D16

D25

D26

D35

D36

+

0 NP

(a,b,c)

978-1-4244-1668-4/08/$25.00 ©2008 IEEE

852

TABLE I.POWER SEMICONDUCTOR DESIGN FOR 2.3kV, 3.3kV AND 4.16kV 3L-NPC-VSCS AND 3L-ANPC-VSCS

The two additional active NPC switches lead to an increase of PEBB material costs by 27% to 30%.

Obviously one core question remains to be answered within this paper: at what converter voltage and switching frequency do IGBT-based 3L-ANPC-VSCs pay off by converting energy more efficiently at the lowest material costs compared to conventional 3L-NPC-VSCs?

III. MODULATION SCHEMESSwitching losses at medium-voltage levels are

relatively high, as are the costs of the semiconductors. Hence, all of the commercial converters mentioned in the section II are operated at rather low switching frequencies of up to about fs = 1kHz. Where required, an improved output voltage quality can be achieved by the means of filters. Beside closed-loop direct torque control strategies, field-oriented controllers with an open-loop pulse-width modulator are commonly used. Since in this case the modulator purely functions as an actuator to realize the voltage command, it can be analyzed independently. The control does not need to be observed for the comparison, which can be focused on losses and harmonic performance of the modulation schemes.

For the range of switching frequencies up to 1 kHz different modulation schemes proved to be especially advantageous. A general rule of thumb says that pulse-width modulation (PWM) schemes achieve good results for frequency ratios where fs/f1 = 15 and greater. Assuming a fundamental target frequency of f1 = 50 Hz, PWM could be used for switching frequencies of fs = 750Hz and above. Among the different versions of PWM for the

3L-NPC-VSC, schemes that use phase-disposed carriers (PD modulation) respectively the nearest three vectors (NTV modulation) deliver the best harmonic performance [13].

These equivalent schemes can be implemented either as carrier-based modulation or as space-vector modulation (SVM). Furthermore, a third-harmonic common mode offset should be added in the carrier-based modulation or the active vectors should be exactly centered within the switching period in an SVM implementation. Figure 2(b) shows the reference waveform using 3L-SVM in which exact centering of the middle vectors is achieved. However, this scheme is not the most adequate for M 0.5775 with respect to the power loss distribution, as will be detailed later in this section.

For frequency ratios below fs/f1 = 15 regular PWM delivers only suboptimal solutions. Selective Harmonic Elimination (SHE PWM) schemes always being synchronised to the fundamental frequency provide a better performance. As the name implies, their switching angles are calculated to eliminate certain harmonic components. They are determined by solving (preferably numerically) a set of nonlinear and transcendental equations [14], [15], [16]:

for odd values of h,

1ph ph,h

h

ˆV ( t ) V sin( h t ) (1)

Voltage level of the converter (Semiconductor device)

2.3kV (FZ1200R33KF2C)

3.3kV (CM900HB90H)

4.16kV (FZ600R65KF1)

Nominal dc-link voltage Vdc,n 3400V 4854V 6118V Commutation voltage Vcom 1700V 2427V 3059V

Maximum device voltage VDRM 3300V 4500V 6500V Rated device dc voltage Vcom@100FIT 1800V 2250V 3600V

Ratio Vcom / Vcom@100FIT 0.94 1.08 0.85 Rated IGBT current IC,n at jmax =125°C 1200A 900A 600A

IGBT module footprint area 190 140 [mm2] 190 140 [mm2] 190 140 [mm2]Rth-jc [K/kW] (IGBT) 8.5 10 11 Rth-jc [K/kW] (Diode) 17 20 21

Rth-ch [K/kW] per module 4 7 6

Three-Level Converter Topology NPC ANPC NPC ANPC NPC ANPC

Installed switch power Ss= VDRM · IC,n·NIGBT+0.5 · VDRM · IC,n·NDIODE

83.16 MVA

(100%)

106.92MVA(129%)

85.05 MVA

(100%)

109.35MVA (129%)

81.90 MVA

(100%)

105.3MVA(129%)

Converter active silicon area ASi932 [cm2](100%)

1199 [cm2](129%)

920 [cm2](100%)

1183 [cm2](129%)

940 [cm2](100%)

1210 [cm2](129%)

IGBT module costs (NPC: 12 modules; ANPC: 18 modules)

8.400€ 12.600€ 11.880€ 17.820€ 12.000€ 18.000€

NPC diode-module costs for 3L-NPC-VSCs (3 modules), (2 diode/module)

1.200€ - 1.869€ - 2.280€ -

Gate-drive unit costs (NPC: 12 units; ANPC: 18 units)

1.512€ 2.268€ 1.620€ 2.430€ 1.890€ 2.835€

Heatsink costs (NPC: 15 units; ANPC: 18 units)

3.675€ 4.410€ 3.675€ 4.410€ 3.675€ 4.410€

Expense of semiconductor devices, GDUs, and heatsinks

14.787€ (100%)

19.278€ (130%)

19.044€ (100%)

24.660€ (129%)

19.845€ (100%)

25.245€ (127%)

853

1

1

4 1 12

Ndc,n k

ph,h kk

VV̂ ( ) cos( h )

h(2)

1ph,h

dc,n

V̂M=

V

2

(3)

1 0ph,h kV̂ ( ) (4)

21 2 N... (5)

where

ph,hV̂ peak voltage of the harmonic component Vph instantaneous phase voltage h harmonic order (e.g., 5th, 7th …11th )

N number of switching angles per quarter-fundamental period

k switching angles

From the Fourier Expansion of the phase to neutral-point voltage (1) and (2) and taking the proper switching angles (5), the selected harmonic components (4) can be eliminated from the output voltage. From equations (2) and (4), it is revealed that for the elimination of N-1 harmonics, a transcendental system with N equations needs to be solved. In any case, N-1 equations are set to zero (i.e., eliminated harmonics), as the fundamental harmonic component is controlled by the modulation depth (M) defined by (3).

Most popular are the SHE schemes featuring three or five pulses per half-fundamental period. Since triple-harmonic components do not need to be included in the set of eliminated harmonics (they cancel from the line-to-line output voltage due to the 120° phase shift) the 5th and 7th or the 5th, 7th, 11th, and 13th harmonics can be eliminated with these two schemes. Thus, despite their low switching frequencies they offer a surprisingly good harmonic performance.

The switched waveforms and switching angles (solutions) varying as a function of the modulation depth are given in Figure 2(c)-(f) for both schemes. According to their effective switching frequencies of fs = 300Hz and 500Hz (for a f1 = 50Hz output), the three-angle and five-angle schemes are referred to as SHE300 and SHE500, respectively.

Figure 3(a)-(c) shows the harmonic spectra of the line-to-line voltage for SHE300 and SHE500 in comparison with 3L-SVM at a comparable fs = 450Hz. The figure clearly visualizes the benefit of harmonic elimination. The weighted total harmonic distortion (WTHDN) of the 3L-SVM lies between the two SHE schemes, corresponding to its switching frequency. But WTHDN is not the only criterion. The distinct elimination of low-order harmonics provides a larger gap between the fundamental and the first harmonic present and, thus, enables a simpler low pass sine filter design. In comparison, 3L-SVM produces a broad spectrum with

small but not negligible 5th and 7th harmonics being present that are more costly to filter.

The PWM scheme applied to the 3L-NPC-VSC, also depends on the modulation depth. In the case of very low modulation depths the fundamental output frequency is typically low as well. Synchronous schemes as SHE PWM can therefore not be used. Another important criterion becoming especially critical in this situation is the distribution of semiconductor losses.1 This issue has been discussed in [11]. Two-level SVM (2L-SVM) has been proven to be favourable with respect to the loss distribution at small modulation depths in [17].

This scheme can be realized as carrier-based modulation by displacing the reference voltages into a single carrier band by a common dc offset as depicted in Figure 2(a). The common mode offset is alternated between the carrier bands in order to balance the power losses among devices as well as to control the neutral-point potential [17]. In contrast to 3L-SVM this modulation scheme can only be utilized for M 0.5775. Versions of it are practically being used in the commercially available 3L-NPC-VSCs.

Figure-3(d) shows the WTHDN for all considered modulation schemes as a function of the modulation depth. For the SHE schemes and 3L-SVM at high modulation depth M = 1.15, the criterion has already been found to be roughly inversely proportional to their switching frequencies (see Figure 3(a)-(c)). At medium modulation depths 0.3 < M < 0.7 the 3L-SVM displays a lower WTHDN than SHE500. For small M < 0.3 the total harmonic distortion of SHE500, 3L-SVM, and 2L-SVM is almost identical.

From the above discussion it becomes clear that for larger modulation depths 3L-SVM and the SHE schemes are the preferable schemes for fs > 500Hz and fs 500Hz, respectively. 2L-SVM is always required for low modulation depths. During the evaluation of the converters for the full operating area (sections IV and V) 3L-SVM and SHE schemes are therefore always used in conjunction with 2L-SVM: 3L-SVM and SHE for M > 0.3, 2L-SVM for M 0.3.

1 The power loss distribution amongst the semiconductors within the 3L-NPC-VSC is inherently unequal. The maximum phase current is limited by the most stressed devices only and the usable output power of the entire converter is reduced [7], [11].

854

0 2 3 4

0

1

-1[ ]t

(a)

0 2 3 4

0

1

-1[ ]t

(b)Modulating signal

Signal is swapped between both carrier bands

Modulating signal

voff-3L

voff-2L

refe

renc

e vo

ltage

-1

0

1

0 /2 3 /2 2[ ]t

132

-1

0

1

0 /2 3 /2 2[ ]t

1

2 3

4

5

0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.90

102030405060708090

[deg]

switc

hing

ang

le

1

23

0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.150

102030405060708090

[deg]

1

2

3

4

5

(c)

(e)

(d)

(f)

phas

e vo

ltage

2L-SVM 3L-SVM

SHE300 SHE500

modulation depth modulation depth1 11.15[M] [M]

Figure 2. Modulation schemes for the 3L-NPC-VSC (a) normalized reference waveform for two-level SVM with centered middle vectors for fs = 450Hz, fs/f1 = 9, M = 0.5775 (b) same for three-level SVM with centered middle vectors for fs = 450Hz, fs/f1 = 9, M= 0.5775 (c) switching pattern for SHE300 (fs = 300Hz, fs/f1 = 6, M = 1.15) (d) switching pattern for SHE500 (fs = 500Hz, fs/f1 =

10, M = 1.15) (e) switching angles versus modulation depth for SHE300 (f) switching angles versus modulation depth for SHE500

855

TABLE II.WORST-CASE OPERATING POINTS OF MV 3L-ANPC-VSCS

IV. MAXIMUM PHASE CURRENT AND CONVERTER POWER

Up to now there has been no detailed comparison between three-level NPC and ANPC converters at voltage levels of VLL = 2.3kV, 3.3kV, and 4.16kV for switching frequencies below fs = 1050Hz. The calculation presented here is based on an iterative simulation procedure developed earlier [18]. Loss data of the IGBT and diode modules of Table I are taken from [11].

Table II summarizes the worst-case operating points of three-level ANPC converters including their maximum phase current, output power, and power ratio between ANPC and NPC converter topologies.

To guarantee full four-quadrant operation, the lowest phase currents among the four worst-case operating points of each converter are considered for the comparisons. As seen in Table II, the limiting points are always to be found at the upper boundary of modulation depth M = 1.15, some at inverter, some at rectifier operation. This can be explained by the fact that at many operating points, the maximum phase currents for inverter and rectifier operation are very close.

Figure-4 compares three-level NPC and ANPC converters in terms of converter power. For this comparison, the converter power of 3L-NPC-VSCs is assigned to 100%. As could be expected, 3L-ANPC-VSCs enable an increase of the output power for each considered switching frequency and system voltage. One interesting result observed in Figure 4(a)-(d) is that the increase of the maximum converter output power, by the application of the loss-balancing scheme in ANPC converters, increases with rising switching frequency and rated converter voltage. For the 3.3kV ANPC converter the gain is especially high. A power increase by 77% is achieved for fs = 1050Hz (Figure 4(d)). Figure 4(e) summarizes the power ratings of all considered converters. As expected, the maximum output power decreases as the switching frequency increases. However, with the application of the ANPC loss-balancing scheme, the 3L-ANPC-VSC enables a lower reduction of the output power at increasing switching frequency. More specifically, when the switching frequency increases from fs = 300Hz to 1050Hz, the 3L-ANPC-VSCs exhibit converter power deratings of 20%, 38%, and 44%

MV system voltage (line-to-line output voltage)Operating points

2.3kV 3.3kV 4.16kV

a j maxC C=37 , 67 a j maxC C=37 , 67 a j maxC C=37 , 67effective

switchingfrequency

MPWM

schemepf

iphSc

MVA

c( ANPC )

c( NPC )

S

S

%

pf

iphSc

MVA

c( ANPC )

c( NPC )

S

S

%

pf

iphSc

MVA

c( ANPC )

c( NPC )

S

S

%

0.05 2L-SVM 1/-1 1100 4.38 118 1/-1 820 4.69 114 1/-1 600 4.32 130 300Hz

1.15 SHE 1 870 3.47 101 -1 630 3.60 113 1 450 3.24 115 0.05 2L-SVM 1/-1 1070 4.26 122 1/-1 750 4.29 125 1/-1 500 3.60 128

500Hz 1.15 SHE 1 840 3.35 108 -1 600 3.43 125 1 400 2.88 125 0.05 2L-SVM 1/-1 1070 4.26 130 1/-1 670 3.83 134 1/-1 430 3.10 132

750Hz 1.15 3L-SVM 1 760 3.03 113 1 480 2.74 155 1 305 2.20 145 0.05 2L-SVM 1/-1 990 3.94 130 1/-1 550 3.14 138 1/-1 350 2.52 130

1050Hz 1.15 3L-SVM 1 700 2.79 117 1 390 2.23 177 1 250 1.80 152

Figure 3. Normalized harmonic spectra of the line-to-line output voltage for the 3L-NPC-VSC (a) SHE300 (fs = 300Hz, M = 1.15) (b) conventional 3L-SVM (fs = 450Hz, M = 1.15) (c) SHE500 (fs = 500Hz, M = 1.15) (d) harmonic performance of 3L-SVM and

SHE PWM schemes as a function of the modulation depth

1 2L-SVM-450Hz 3 SHE5004 SHE3002 3L-SVM-450Hz

PWM:

0 5 10 15 20 25 30 35 40

0 5 10 15 20 25 30 35 4010-2

10-1

10 5 10 15 20 25 30 35 40

(a)

(b)

(c)

NWTHD.

%215

NWTHD.

%172

NWTHD%

1.50

10-2

10-1

1

10-2

10-1

1

1

h

h

f

f

,

, 1

ˆˆ

LL h

LL h

V

V

0

1

2

3

NWTHD

%

0 0.2 0.4 0.6 0.8 1 1.1[M]

12

34

(d)

856

at VLL = 2.3kV, 3.3kV, and 4.16kV respectively, compared to reductions of 30%, 61%, and 58% for conventional 3L-NPC-VSCs.

V. COMPARISON OF EFFICIENCY, SWITCH UTILIZATION AND MATERIAL COSTS

In terms of efficiency, it is clear that an increase of the switching frequency leads to higher switching losses which in turn reduce the maximum converter power. By its principle of operation, the ANPC loss-balancing scheme leads to a more equal distribution of semiconductor losses and junction temperatures but not to a reduction of losses and an improvement of efficiency. The slight differences in figure 5(a) are caused by the different operating currents of 3L-NPC-VSCs and

3L-ANPC-VSCs at a constant maximum device junction temperature of jmax = 67°C. The most efficient operation of 3L-ANPC-VSCs is achieved at a voltage level of VLL = 3.3kV for fs 500Hz.

Figure-5(b) presents the switch utilization of the converter topologies. As expected, the switch utilization decreases with increasing switching frequency. Three particularities can be observed:

1) The 2.3kV NPC converter enables the highest switch utilization for all switching frequencies.

2) The ANPC converter enables a substantial increase of the switch utilization at converter voltages of 3.3kV and 4.16kV for switching frequencies of fs

750Hz.

0

1

2

3

2.3kV-NPC 2.3kV-ANPC 3.3kV-NPC 3.3kV-ANPC 4.16kV-NPC 4.16kV-ANPC

100%91%

78%70%

100%97%87%

80%

100%

86%

55%

39%

100%95%

76%

62%

100%

82%

54%42%

100%

89%

68%

56%

500HzSHE 750Hz3L-SVM 1050Hz3L-SVM300HzSHE

[ ]cS MVA

0

1

2

3

2.3kV 3.3kV 4.16kV

[ ]cS MVA113%

100% 155%

100%100%

145%

750Hz

0

1

2

3

2.3kV 3.3kV 4.16kV

[ ]cS MVA 500Hz100%

100%100%

108% 125%

125%

0

1

2

3

2.3kV 3.3kV 4.16kV

[ ]cS MVA 300Hz100%100%

100%

101% 113%115%

3L-NPC 3L-ANPC

0

1

2

3

2.3kV 3.3kV 4.16kV

[ ]cS MVA

117%100%

177%

100% 100%

152%

1050Hz

(a) (b)

(c) (d)

(e)

Figure 4. Maximum output power of 3L-ANPC-VSCs (a) SHE300; (b) SHE500; (c) fs = 750Hz (3L-SVM); (d) fs =1050Hz (3L-SVM); (e) same as a function of the switching frequency (at jmax = 67°C, a = 37°C, VLL = 2.3kV

(FZ1200R33KF2C), VLL = 3.3kV (CM900HB90H), VLL = 4.16kV (FZ600R65KF1))

857

3) The increase of the switch utilization (by about 20% (fs = 750Hz) and 38% (fs = 1050Hz)) by the use of the ANPC converter has a maximum at 3.3kV converter voltage.

The ratio of the semiconductor active silicon area per rated converter power is inversely correlated to the switch utilization as shown in Figure-5(c). The 2.3kV NPC converter enables the lowest expense of silicon. The ANPC converter reduces the silicon area per MVA by 17% and 12% at fs = 750Hz and by 27% and 15% at fs =1050Hz for converter voltages of 3.3kV and 4.16kV, respectively.

Figure-6 shows the material costs of semiconductors, heatsinks and gate-drive units per converter power. These costs typically contribute about 30%-40% to the total power part material costs. For a converter voltage of VLL

= 2.3kV the NPC converter is clearly the most economical solution. The cost advantage is between 29% (fs = 300Hz) and 12 % (fs = 1050Hz).

In contrast, 3L-ANPC-VSCs are most cost efficient at 3.3kV and 4.16kV for switching frequencies of fs

750Hz. For both converter voltages the break even of the 3L-ANPC-VSC is close to a switching frequency of fs =500Hz.

99

99.2

99.4

99.6

99.8

100

2.3kV-NPC 2.3kV-ANPC 3.3kV-NPC 3.3kV-ANPC 4.16kV-NPC 4.16kV-ANPC

[%]

0

200

400

600

800

2.3kV-NPC 2.3kV-ANPC 3.3kV-NPC 3.3kV-ANPC 4.16kV-NPC 4.16kV-ANPC

500HzSHE 750Hz3L-SVM 1050Hz3L-SVM300HzSHE

2cm

MVA

100%110%

128%144%

127%132%146%

158%

106%124%

191%

269%

121%127%

159%

195%

123%

150%

229%

291%

137%155%

202%

247%

(-17%)

(-27%)(-12%)

(-15%)

0

10

20

30

40

2.3kV-NPC 2.3kV-ANPC 3.3kV-NPC 3.3kV-ANPC 4.16kV-NPC 4.16kV-ANPC

1000c

s

S

S

100%

91%

78%70%

79% 76%69%

63%

91%

78%

50%

36%

80%75%

61%

49%

83%

68%

45%

35%

75%66%

51%41%

(+18%)

(+13%)(+38%)

(+20%)

Figure 5. (a) Efficiency of 3L-NPC-VSCs and 3L-ANPC-VSCs at jmax = 67°C, a = 37°C (Phase currents according to Table II, VLL = 2.3kV (FZ1200R33KF2C), VLL = 3.3kV (CM900HB90H), VLL = 4.16kV (FZ600R65KF1)) (b) Switch utilization of 3L-

NPC-VSCs and 3L-ANPC-VSCs, according to Table I and II (c) Active silicon area per MVA of 3L-NPC-VSCs and 3L-ANPC-VSCs, according to Table I and II

(a)

(b)

(c)

858

VI. CONCLUSIONSIn this paper, the 3L-ANPC-VSC has been compared

to 3L-NPC-VSCs at different voltage levels. Depending on the converter voltage and switching frequency, the 3L-ANPC-VSC enables an increase of the maximum converter power of up to 77%. This enormous power increase of the 3L-ANPC-VSC is attractive for applications where the required converter power cannot be reached with PEBBs on the basis of the largest IGBT and diode modules of a certain voltage class.

The situation is different where an increase of the output power is not required. In this case, the 3L-ANPC-VSC has to compete against the 3L-NPC-VSC of the next higher current rating. Cost indicators, e.g., the switch utilization ratio, provide a measure for its effectiveness.

At the 2.3kV voltage level, the 3L-NPC-VSC is the more cost-effective solution. At the 3.3kV and 4.16kV voltage levels and switching frequencies fs > 500Hz, however, the ANPC converter provides better cost-per-power ratio than the NPC converter. This result proves the 3L-ANPC-VSC to be a very attractive topology beyond the results of previous investigations.

REFERENCES[1] S. Bernet, “Recent Developments of High Power Converters for

Industry and Traction Applications,” IEEE Trans.Power Electron.,2000, vol. 15, no.6, pp. 1102-1117.

[2] S. Bernet, “State of the Art and Developments of Medium Voltage Converters – An Overview,” Przeglad Elektrotechniczny (Electrical Review), vol. 82, no. 5, May 2006, pp. 1–10.

[3] T. Brückner, S. Bernet, and P. K. Steimer, “The active NPC converter for medium-voltage applications,” in Conf. Rec. IEEE-IAS Annu. Meeting, Hong Kong, China, 2005, pp. 84–91.

[4] T. Brückner and S. Bernet, “Loss balancing in three-level voltage source inverters applying active NPC switches,” in Proc. IEEE-PESC, Vancouver, Canada, 2001, pp. 1135–1140.

[5] P. K. Steimer, O. Apeldoorn, B. Ødegård, S. Bernet, and T. Brückner, “Very high power IGCT PEBB technology,” in Proc. IEEE-PESC, Recife, Brazil, 2005, pp. 1–7.

[6] T. Brückner, The active NPC Converter for Medium-Voltage Drives. Aachen: Verlag Shaker, 2006.

[7] T. Brückner, S. Bernet, H. Güldner, “The Active NPC Converter and its Loss-balancing Control,” IEEE Trans. Ind. Electron., vol. 52, no. 3, June 2005, pp. 855-868.

[8] Datasheet FZ1200R33KF2C, Tech. Information IGBT-Modules, Infineon GmbH, Warstein, Germany, 2003.

[9] Datasheet FZ600R65KF1, Tech. Information IGBT-Modules, Infineon GmbH, Warstein, Germany, 2002.

[10] Datasheet CM900HB90H, Tech. Information IGBT-Modules, Powerex Corp, Pennsylvania, USA, 2005.

[11] J. A. Sayago, T. Brückner, and S. Bernet, “How to Select the System Voltage of MV drives A comparison of Semiconductor Expenses,” accepted for publication in IEEE Trans. Ind. Electron.Mar, 2008.

[12] SIEMENS: Sinamics MV Medium voltage drives, catalog D 12, Automation and Drives, Siemens AG, Nuremberg, Germany, 2006; information available online http://www.automation.siemens.com

[13] D. G. Holmes, T. A. Lipo, Pulse with Modulation for Power Converters. New York: Wiley, 2003.

[14] H. S. Patel, R. G. Hoft, “Generalized Techniques of Harmonic Elimination and Voltage Control in Thyristor Inverter: Part I-Harmonic Elimination,” IEEE Trans. Ind. Applicat., Vol. 9, No.3, May/Jun., 1973, pp.310-317.

[15] F. Jenni, D. Wüest, Steuerverfahren für selbstgeführte Stromrichter. Stuttgart: B. G. Teubner, 1995.

[16] S. Sirisukprasert, J.S. Lai, T.H. Liu, “Optimum harmonic reduction with a wide range of modulation indexes for multilevel converters,” IEEE on Ind. Appl. Conference, vol.4, pp.2094-2099 2000.

[17] T. Brückner and D. G. Holmes, “Optimal pulse-width modulation for three-level inverters,” IEEE Trans. Power Electron., vol. 20, no. 1, pp. 82-89, Jan. 2005.

[18] T. Matsuo, S. Bernet, R. S. Colby, and T. A. Lipo, “Modeling and simulation of matrix converter / induction motor drive,” Mathematics and Computers in Simulation, vol. 46, pp. 175–195, 1998.

[ ]sf Hz0

2

4

6

€k

MVA 2.3kV

100%

129%

300SHE

100%

121%

500SHE

100%112%

1050SVM750SVM

100%115%

0

4

8

12

3.3kV

100%115%

€k

MVA

300SHE [ ]sf Hz

100%104%

500SHE

100%

1050SVM

73%100%84%

750SVM

0

4

8

12

16

€k

MVA

100%110%

4.16kV

300SHE [ ]sf Hz

100%102%

500SHE

88%

750SVM

100% 84%

100%

1050SVM

IGBT-module

NPC diode-module

Heat-sink module

Gate-drive unit

a)

b)

c)

NPCANPC

NPCANPC

NPCANPC

Figure-6. Costs of semiconductors, gate-drive units and heatsinks per output power of 3L-NPC-VSCs and 3L-NPC-VSCs, according

to Table I and II (at jmax = 67°C, a = 37°C, VLL = 2.3kV (FZ1200R33KF2C), VLL = 3.3kV (CM900HB90H), VLL = 4.16kV

(FZ600R65KF1))