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Self Tuning Pick-ups for Inductive Power Transfer

Grant A. Covic Senior Member IEEE, J.T. Boys, A.M.W. Tam Student Member IEEE and J. C.-H. Peng The University of Auckland, Department of Electrical and Computer Engineering

Private Bag 92019, Auckland 1142, New Zealand

Abstract—Inductive power transfer systems are now commonly used in ultra clean or dirty industrial applications to deliver power to both moving and stationary loads without contact. Each load requires a power pick-up and regulator that is coupled to a track, which carries a resonant current at VLF frequencies (typically 10-50kHz). Modern pick-ups are tuned for resonance at the track frequency and their power delivery increasingly depends on accurate tuning of the resonant circuit. In this paper a new self tuning power regulator for inductive power transfer systems is proposed and a control strategy described. The system uses a binary weighted series of capacitors that can be switched via simple relays across a parallel resonant pick-up circuit. There are no additional switching losses during tuning, detection and operation so that the technique is applicable to high power pick-ups. The tuning circuit is evaluated under simulation and then made to operate on a practical 500W pick-up regulator used in materials handling applications. In both simulation and experiment it is shown to successfully detect the tuning state of the system and correct for it during operation.

I. INTRODUCTION

Inductive power transfer (IPT) systems are now commonly used in industry where power is required to be delivered to one or more loads without physical contact. Such systems have low maintenance and high reliability and the ability to operate in ultra clean or ultra dirty systems and remain unaffected by dirt, or moisture. Medium to high power applications of this technology include continuous power transfer to people movers, contact-less battery charging of electric vehicles [1-5] and materials handling systems [6]. Low power systems comprise compact electronic devices such as medical implants [7] and sensors [8-10].

A typical IPT system comprises a resonant power converter operating at VLF frequencies that regulates a current (I1) in a track loop as shown in Fig.1. Fixed frequency supplies are preferred particularly in systems with more than one pick-up as this guarantees the frequency irrespective of the tuning of the supply and simplifies the design of all pick-ups. In each pick-up an inductor (L2), comprising a magnetic core with high frequency winding is magnetically coupled to the track, and tuned for resonance at the track frequency using compensation capacitors. This tuning enables the output power to be increased in proportion to the circuit Q.

Modern pick-up regulators tend to operate with circuit Q’s between 5-10 because of increased power demands and a desire to minimize the weight of the magnetic material. The power delivery increasingly depends on accurate tuning of the circuit capacitors as the bandwidth of the system reduces with Q. Additionally such systems

are finding application in difficult to reach areas such as those subject to extreme weather cycles, underwater, and clean rooms. In such applications there is a need to minimize maintenance.

3Input

Power Supply + Output Compensation

track conductor inductance = L1

I1

Switched- Mode

Controller

L2

DC power

Pickup Compen -sation

Pickup Inductance

Figure 1. A typical IPT System

A tuned pick-up is most efficient and can deliver the greatest amount of power when it is perfectly tuned. As such, in this work a method of detecting the state of the tuned circuit and of correcting any mistuning using switched capacitors is proposed. If the mistuning is severe a service call can be flagged. The method proposed herein is simplistic in concept, has minimal distortion and loss, and can be applied to high power pick-ups. The tuning detection and control is slow, but is present to correct for the degradation of capacitors over time due to aging and thermal variations, and therefore does not need to be fast.

Adding variable tuning elements into an IPT pick-up controller is not a new idea. Various authors have suggested using either switched capacitors or inductors to deliberately detune a pick-up and thereby regulate power in single pick-up applications [2,8-11]. In IPT systems with multiple pick-ups however, this severely loads the supply by reflecting unwanted reactive loads onto the track. Furthermore such methods assume that the system’s ideal tuning point does not change due to capacitor degradation and do not attempt to detect the tuning point. Moreover all of the above systems attempt detuning during operation, but most require precise switching of active devices which under operation introduce significant harmonics in the tuning circuit due to the non-linear control of the switched inductors or capacitors that must be controlled at the track frequency (typically 10kHz or greater). Consequently few practical circuits have been realized at even modest power levels due to control demands and inherent operational losses.

This paper begins with an overview of a traditional pick-up IPT regulator including the necessary compensation and the effect of the switching regulator on a tuned and un-tuned circuit. This is followed by a description of a simple modified switched capacitor method that can be used to detect mistuning and tune the system with little or no distortion at high power.

978-1-4244-1668-4/08/$25.00 ©2008 IEEE

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Experimental and simulated measurements are presented verifying the techniques validity on a pick-up controller rated at 500W which is used for materials handling applications.

II. THE PARALLEL TUNED PICK-UP CONTROLLER

A pick-up coil can be either series or parallel compensated using a capacitor C2, although parallel tuned pick-ups (Fig. 2) are more common due to their inherent current limiting capabilities. The pick-up magnetic coupling (M) between L2 and the track depends on its physical position. The voltage induced in the pick-up coil at track frequency ( ) is 1MIjVoc [5]. This combined with the short circuit current of the coil ( 2LjVI ocsc ), both of which can be easily measured, allows the uncompensated VA (Su) of the pick-up winding to be determined as:

2

22

1 LMIIVS scocu (1)

The AC tuning causes a resonant current to flow in the tuned L2C2 circuit which is boosted by Q, while the voltage (Vc) across the capacitor is similarly boosted by the circuit Q. Consequently the output of a parallel tuned circuit presents a boosted voltage to the rectifier and load (shown in Fig. 2), while the resonant current circulates in the L2C2 loop. For the circuit of Fig. 2 (assuming switch Sis not operating) then this boosted voltage is described by (2). Assuming the rectifier diodes are held in continuous conduction during operation [12] then the relationship between the rms value of the AC resonant voltage (Vc) and the average DC output voltage can also be described (here VF represents the forward voltage drop of a diode) as:

Foocc VVQVV 322

(2)

If the pick-up coupling (M) to the track is constant (as is common in materials handling applications where the pick-up is attached to a trolley which moves in a constrained fashion along a monorail), and the track current and frequency are assumed to be regulated to be constant by the IPT system supply, then Voc is essentially constant. Consequently if the output voltage is also controlled constant by a regulator then the maximum Q that is possible during circuit operation is also a constant given by:

oc

Fo

VVVQ 3

22max (3)

If the diode losses are ignored and noting that there must be power conservation across the rectifier while the diodes are in continuous operation, then the fundamental component of the rms current at the input of the rectifier will also be related to the average current in the DC inductor (IL) by the ratio: 22 .

LR

Figure 2. A parallel tuned pick-up with boost regulator

The maximum output power of the circuit is therefore given by (4), so that the uncompensated power is effectively boosted by the circuit Q during operation to drive the load.

max(max) QSP uo (4)

A typical IPT pick-up has to deliver the output power regardless of tuning, and consequently if an output power is required which is less than maximum, a switching regulator is required to lower Q to a suitable value by adjusting the duty cycle D of switch S in Fig. 2.

The effect of the switching regulator on the circuit operation can be investigated using state space analysis techniques noting that the second half of the circuit is essentially a current sourced boost regulator. As stated the operating Q is directly controlled by varying the duty cycle in proportion to )1( D . As such the tuned voltage (Vc), and output power (Po) under regulation also vary in proportion to )1( D and are given by:

DVQVV oocc 122

(5)

DVILM

DPP ooo 122

1 12

(max) (6)

With parallel tuning the resonant circuit at the input of Fig. 2 acts as if it is driven by a current source (Isc) so that the input voltage can be converted to a Norton equivalent as shown in the transformations from Fig 3(a) to Fig. 3(b). If the circuit is tuned perfectly and has no loss, then Iscalso flows into the rectifier. In practice tuning errors act to reduce the driving current from this ideal, as shown in Fig. 3 where a portion of current is required to feed the mistuned reactance ( 2C ). Pick-up inductor losses due to rL also reduce the available power to the load (shown explicitly in Fig 3b where the equivalent Norton resistance (rx) shunts the ideal capacitor). rx can be shown to be given by:

Lx rC

Lr2

2 (7)

If the pick-up inductor is wound using Litz then these losses are normally low and can therefore be ignored in further analysis, otherwise they should be included.

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2Lrxr

(a) (b)

Figure 3. Creating a Norton equivalent of the mistuned circuit

The DC inductor (LDC in Fig. 2) is chosen to keep the diodes in continuous conduction and limit current through the regulator switch when it is turned on. If this value is large it will reflect some small inductive impedance back on to the resonant circuit causing some detuning, but this can be determined and compensated at design.

Consequently the output regulator circuit can be considered as an equivalent lumped load (Req) across the AC circuit (Fig 3). In practice Req includes the losses in the regulator circuit, but these are normally small for medium to high power pick-ups. The equivalent load across the resonant tank can be expressed in terms of the actual output load resistance via state space analysis as shown in (8).

22

18

DRR Leq (8)

Any mistuning which results from degradation of capacitors over time has the most undesirable effect on the pick-up, as it permanently reduces the power capability of the pick-up during operation. As such it is the object of this paper to quantify these effects and find a means of compensation.

The output voltage regulator automatically compensates for the reduction in average current through the DC inductor (IL) at light loads by reducing the duty cycle to maintain the output load current constant following (9). Consequently for a known output loading condition, changes in duty cycle can be used as a direct measure of mistuning.

)1( DII Lo (9)

Po and Vo are maintained with the regulator action, although additional reactive currents will continue to be present during operation. Notably Req, Q and Vc alsoincrease because of this controlled decrease in D.

At the point at which the switch is fully off, the regulator can no longer compensate for the loss of current and the circuit will be operating at Qmax. Any further mistuning will cause Po to drop, with a corresponding reduction in Vo. The diodes of Fig. 2 will remain under continuous conduction for all but the lightest loads. As such Vc will now be directly coupled via a simple filter to the output voltage, and therefore will also decrease following (2).

III. A VARIABLE TUNED PICK-UP REGULATOR

A. Steady State Analysis

The circuit of Fig. 2 was investigated by simulation and experimentally on a 500W pick-up regulator used for materials handling applications with a selection of capacitor values for C2. The actual parameters of both circuits are given in Table 1. In both cases the output voltage was held constant at 300V irrespective of the load and the tuning, simply by adjusting the duty cycle of the switch.

The results of the simulations with varying C2 are shown in Fig. 4. As expected (as discussed in the previous section), with ideal tuning, )1( D and Vc are minimized for all loading conditions, but increase with mistuning. When operating at the extremes (with 230LR ), compensation is no longer possible and VVc 336 asexpected from (2). Vo and Vc both fall if the mistuning increases beyond this point, as shown when 210LRand )(22 1.01 idealCC .

The experimental results for Vc and )1( D with varying C2 were similar to that simulated in Fig 4 although as shown in Fig. 5 only the latter plot is presented for comparison as the focus here is on the use of duty cycle to both detect mistuning and control the power flow.

Notably there are small differences between the simulated and practical circuits which are reflected in the simulated versus measured results. In the steady state simulations a larger DC inductor and much smaller DC capacitor were used than actually employed in the practical circuit. Practical diode and switch models were used but these were not actually those of the final circuit devices, and losses were ignored in the modeled inductors/capacitors. The consequence of using a larger inductor and much smaller capacitor is that the driving current is maintained closer to that ideally expected from a state space averaging analysis, and the time taken to reach steady state is smaller than in practice. The final constructed circuit used a pick-up inductor wound using Litz with low winding resistance. Wima capacitors with low ESR resistance were used for tuning, an International Rectifier IGBT (IRGP20B60PD) switch and several (HFA16PB120) ultra-soft recovery diodes were used for the rectifier and output stage. The switch controller is here regulated at around 32kHz.

TABLE I. CIRCUIT PARAMETERS

Track Current 60A Nominal Track Frequency (fo) 10kHz

Open Circuit Voltage (Voc) 55.2V Short Circuit Current (Isc) 1.51A

Pick-up Self Inductance (L2) 576μH Pick-up Ideal C2 439nF

Simulated DC-Inductors (LDC) 18mH/3mH Actual DC Inductor (LDC) 3mH

Simulated DC Capacitor (CDC) 100uF Actual DC Capacitor (CDC) 1530uF

Controller Output Voltage (Vo) 300V Qmax (equation (3)) 6.2

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220

240

260

280

300

320

340

394 404 419 431 441 451 460 471 484C2 (nF)

Vc (

V)R=210 R=230 R=250 R=270

(a)

65

70

75

80

85

90

95

100

394.6 404.5 419.4 431 441 451 459.5 470.5 483.6

C2 (nF)

1-D

(%)

R=210 R=230 R=250 R=270

(b)

Figure 4. Pick-up output controlled at 300V at various RL with(a) Vc (rms) vs. C2 (b) (1 D) vs. C2

65

70

75

80

85

90

95

100

394.6 404.5 419.4 431 441 451 459.5 470.5 483.6

C2 (nF)

1-D

(%)

R=210 R=230 R=250 R=270

Figure 5. Experimental variation of (1 D) vs. C2

B. Transisent Analysis Several transient simulations were undertaken to verify

the expected results and then compared with the practical circuit when operating at various loads. Fig. 6 and 7 show the positive half of the resonant voltage under start up conditions with the output capacitor initially discharged. In Fig 6(a) and 7(a) the resonant circuit is tuned at its ideal value for the track frequency (using 439nF) while in Fig. 6(b) and 7(b) C2 is deliberately mistuned by 5% (417nF). In both cases the output load is held constant at 250(360W) and a DC inductor of 3mH is used. In the experimental measures of Fig. 7 the controlled output voltage is also shown. Notably the switch remains off during the initial transient. Regulation only begins when the output voltage reaches 300V (equivalent to Vcreaching a peak of around 475V as defined by (2), taking into account the voltage drops of the diodes ~ 1V each).

(a)

(b)

Figure 6. Simulated start up transients of Vc for (a) C2 = 439nF (b) C2 = 417nF

(a)

(b)

Figure 7. Experimental start up transients of both Vo (upper trace) and Vc (lower trace) for (a) C2 = 439nF (b) C2 = 417nF

After reaching this peak the resonant voltage reduces (with time constant related to the output filter and the magnitude of the driving current) to the value given by Fig. 4(a) (within the resolution of the feedback controller) as a result of the regulator action. Notably the time to reach 300V is greater as the level of detuning increases because of the effective reduction in the source current. If the detuning is too high then the desired 300V level may never be reached, the duty cycle will stay at zero and the

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magnitude of Vc will depend on value of Vo achieved as given by (2).

C. A Practical Self-Tuning Regulator In the experimental controller (Fig. 8) a microcontroller

(not shown) is used to measure and regulate Vo by controlling D. The switching frequency of the IGBT is at a 32kHz rate, while the output voltage is regulated over a 20V hysteresis range of 290-310V (nominally ~ 300V). A second microcontroller was added (as shown explicitly in Fig. 8) that measures Vo and Io and the filtered (averaged) value of D. It has the ability to send shutdown and start-up commands to the controlling microprocessor within the boost converter and to control a separate tuning board.

The tuned circuit of the pick-up regulator is initially deliberately mistuned with a C2=417.7nF. An additional tuning board is added in parallel (as shown), comprising 4 additional binary weighted capacitors to allow 16 (approximately equal) tuning steps over a suitable tuning range (C=5.02nF, 2C=10.98nF, 4C=21.90nF, 8C=47.03nF). In order to facilitate tuning with minimal loss, tuning capacitors are only switched in and out of the circuit when the main circuit switch (S) in Fig. 2 is turned on and the resonant voltage across C2 has had sufficient time to completely collapse. This allows low cost relays to be used to switch the various capacitors in and out of the circuit without loss, but lengthens the tuning process due to their slow operating time. Under these starting conditions, the microprocessor does not know the ideal tuning point and therefore begins a search until it detects that the duty cycle is maximized.

Fig. 9 shows the experimental operation at initialization with a known constant output load of 360W. The controller then sequentially adds capacitance while regulating the voltage near 300V until the correct tuning point (4C) is found. Notably the steady state Vc varies during each tuning test but this is not measured – rather the average D is measured only when Vo is controlled. Prior to each test, the regulator is shut off and Vc quickly collapses. A new capacitance is then switched in, and after a suitable wait time that allows for the relay to stabilize, the circuit decoupling switch (S) is then tuned off and the resonance builds up in the newly tuned tank. After the initial overshoot which charges the depleted output capacitor, the regulator operates to control the output voltage and Vc settles to steady-state at which point the filtered D is measured. This process is repeated until the desired operating point is determined (the point at which maximizes the duty cycle (minimizes )1( D ). The microprocessor ensures that the load remains stable during the measurements otherwise a particular measurement is discarded.

Figure 8. Schematic of the automatic tuning circuit

Figure 9. Experimental variation of Vc during start-up tuning

In a materials handling plant, the workload for each powered vehicle is normally cyclic. At the beginning of each cycle when a vehicle is waiting to begin operation, it is often stationary and the pick-ups have but nominal load or are unloaded. At this time the pick-up could be tested to determine if the circuit is operating on- or off-tune. Consequently a suitable load can be applied and tuning tests undertaken to evaluate the state of the system. For tuning purposes it is desirable to have as much discrimination as possible in the feedback signals. As shown in Fig.’s 4(b) and 5, the variation in )1( D is greatest at higher power demands providing the output power can be regulated over a sufficiently wide tuning range. The simplest solution is therefore to ensure the load remains essentially constant during the test by applying a known load. In the experimental system of Table 1 any load between 300-450W was found to be suitable.

In practice only three tuning points are required to determine the correct tuning point of the system or to determine how far off tune the system and whether a service call is required. The controller simply keeps track of the ideal tuned capacitance and tests this point along with two others determined by adding and subtracting a fixed amount of capacitance either side. A parabola can be fitted to the data and the closest minimum tuning point determined. A range of tests were taken at various loading conditions each following the above procedure. Experimentally it was found that either 3C or 4C was chosen by the microcontroller which is to be expected when noting that with but 1% shift in the resonant capacitance the variation in duty cycle will be small (as shown in Fig. 5). In practice the presence of LDC will reflect a slight inductive reactance on the resonant circuit, so that the ideal capacitor may be slightly smaller than 439nF. Fig. 10 shows two such tests during normal operation with a constant load. If a larger duty cycle is found the system continues to search until a new maximum is found. If no new maximum is discovered but a new capacitor value is found with an identical duty cycle to that which the system was originally operating with, this new operating point is selected, otherwise the original operating point is kept. In Fig 10(a) a load of ~370W is used and the system was originally at 3C. After testing both 4C and 5C, 4C is selected as the tuned capacitor. In Fig. 10(b) a load of ~390W is used. Here the system begins at 4C and after testing 5C the system moves to test 3C and finds it has essentially the same maximum duty cycle as that originally used and therefore it is selected.

1C 2C 3C 4C 5C 4C0CTuning Process

In Tune Off Tune

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(a)

(b)

Figure 10. The variation of Vc during stationary tuning tests with (a) a load of ~370W (b) a load of ~390W

In the above work a practical 500W pick-up was used that is deliberately switched at high frequency. The consequence of this is that the regulator operates as a pure current sourced boost converter forcing the operating Vc to vary with mis-tuning and load. However many high-power pick-ups operate with slow switching regulators (typically only a few hundred Hz) [6], so that the shorting switch is off or on for tens of cycles of the operating resonant track frequency. Consequently the output regulator appears as simple filter when the switch is off forcing the operational Q Qmax as given by (3), and a Vcgiven by (2). When switch S is closed, Vc will completely collapse and the operating Q 0. The time averaged Q andvoltage Vc will however be identical to that indicated by (5) and (6) since D will continue to change as discussed under load and de-tuning. As such all of the above control remains valid even for these slow switching regulators and the controller will continue to work just as effectively independent of the controller employed.

IV. CONCLUSIONS

A low loss, zero distortion, tuning system was implemented with controller on a practical 500W pick-up and found to successfully detect and correct the tuning of the circuit in situ. Simulations and experimental measurements under steady state and transient conditions show the effectiveness and robustness of this simple technique. To the best of the authors’ knowledge, this is the first such system applicable to real medium and high power IPT pick-ups.

REFERENCES

[1] G. A. Covic, G. A. J. Elliott, O. H. Stielau, R. M. Green, and J. T. Boys, "The design of a contact-less energy transfer system for a people mover system," International conference on Power System Technology, 2000. Proceedings. Powercon 2000, 2, Perth Australia, 4-7 Dec., 2000, pp.79-84.

[2] E.H. Lechner, D.M. Empey and S.E. Schladover “Testing of a roadway powered electric vehicle prototype” Proc. of the 10th Int. Electric Vehicle Symp., 3-5 Dec. Hong Kong, 1990, pp. 959–973

[3] C.-S. Wang, O. H. Stielau, and G. A. Covic, "Design considerations for a contactless electric vehicle battery charger," IEEE Trans. Ind. Electronics, vol. 52, pp. 1308-1314, October 2005.

[4] C. Byungcho, N. Jaehyun, C. Honnyong, A. Taeyoung, and C. Seungwon, "Design and implementation of low-profile contactless battery charger using planar printed circuit board windings as energy transfer device," IEEE Trans. Ind. Electronics, vol. 51, pp. 140-147, February 2004.

[5] H. Sakamoto, K. Harada, S. Washimiya, K. Takehara, Y. Matsuo, and F. Nakao, "Large air-gap coupler for inductive charger [for electric vehicles]," IEEE Trans. Magnetics, vol. 35, pp. 3526-3528, September 1999.

[6] J. T. Boys, G. A. Covic, and A. W. Green, "Stability and control of inductively coupled power transfer systems," IEE Proc. EPA, vol. 147, pp. 37-43, January 2000.

[7] W. Guoxing, L. Wentai, M. Sivaprakasam, and G. A. Kendir, "Design and analysis of an adaptive transcutaneous power telemetry for biomedical implants," IEEE Trans. Circuits and Systems I: Regular Papers, vol. 52, pp. 2109-2117, October 2005.

[8] A. P. Hu and S. Hussmann, "Improved power flow control for contactless moving sensor applications," IEEE Trans. Power Electronics Letters, vol. 2, pp. 135-138, December 2004.

[9] P. Si, A.P. Hu, S., Malpas and D. Budgett “Switching frequency analysis of dynamically detuned ICPT power pick-ups” International Conference on Power System Technology, 2006. PowerCon 2006. Oct. 2006 pp. 1-8

[10] J.U. Hsu, A.P. Hu, P. Si and A. Swain “Power flow control of a 3-D wireless power pick-up” 2nd IEEE Conference on Industrial Electronics and Applications, 2007. ICIEA 2007. 23-25 May 2007 pp.2172 – 2177

[11] J.E.I. James, J.T. Boys, G.A. Covic “A Variable Inductor based tuning method for ICPT Pick-ups” 7th International Power Engineering Conf., IPEC2005, Singapore, 28 Nov.-1 Dec., 2005, pp. 1142-1146

[12] J.T. Boys, G.A. Covic and Y Xu “DC analysis technique for inductive power transfer pick-ups” IEEE Trans., Power Electronics Letters, 1 no. 2, pp. 51-53, June 2003

4C 5C 4C3C3 step tuning process

New tuned point chosen Begin re-tune

5C 3C4C3 step tuning process

New tuned point chosen Begin re-tune