HIGH-E B M S · 2016-07-19 · would like to thank my friends, Hazem Soliman, Ahmed Mohamed, Mohamed Ramadan, Essam El Sahwi, and Sherif Helmy. My deepest gratitude goes to my parents,

ASSISTING CONVERTER BASED TOPOLOGIES FOR LOW-VOLUME

HIGH-EFFICIENCY BATTERY MANAGEMENT SYSTEMS

by

Mahmoud Fawzy Aziz Shousha

A thesis submitted in conformity with the requirements for the degree of Doctor of Philosophy

Graduate Department of Electrical and Computer Engineering

University of Toronto

© Copyright by Mahmoud Shousha 2016

ii

Assisting Converter Based Topologies for Low-Volume High-Efficiency

Battery Management Systems

Mahmoud Fawzy Aziz Shousha

Doctor of Philosophy

Graduate Department of Electrical and Computer Engineering

University of Toronto

2016

Abstract

Conventional battery management systems (BMSs) for applications that require voltage stepping-

up, such as automotive and portable systems, usually consist of a battery pack, a battery charger,

and a step-up dc-dc converter.

In those applications, a balancing circuit which compensates for different states of charge (SOC)

of individual cells is a highly desirable feature. The SOC difference occurs due to aging and

differences in the manufacturing process. Through cell balancing the effective capacity and

lifetime of the battery pack can be significantly increased. Even though the benefits of cell

balancing circuits have been recognized, their use is relatively sparse in the targeted applications,

due to overly large extra cost, weight, and volume they add to the system.

In this work, a novel battery management architecture (BMA) that integrates voltage stepping-up

and cells balancing functionalities in a single converter topology is introduced, offering a solution

iii

for a cost-effective implementation of the balancing feature. Moreover, the new architecture has a

smaller overall size and better power processing efficiency than the conventional solutions not

incorporating the balancing feature.

In the new BMA, named the assisting converter architecture, the output of the converter is placed

on top of the battery pack and is therefore processing only a portion of the output power.

Two assisting converters utilizing the new BMA are introduced in this dissertation namely, the

assisting dual active bridge (DAB) converter and the assisting flyback converter. The assisting

DAB converter represents the battery power management system for automotive applications and

the assisting flyback converter is designed for portable electronics applications.

iv

Acknowledgements

I would like to express my sincere thanks, appreciation, and gratitude to my advisor, Professor

Aleksandar Prodić, for his precious advice, support, and continuing encouragement. In Professor

Prodić I saw a practical example in conveying a continuous spirit of adventure and excitement for

research and teaching. Without his genuine ideas and positive feedback, this dissertation would

not have been possible. I have totally enjoyed working in Laboratory for Power Management and

Integrated SMPS under his supervision.

I would like to thank my dissertation committee members, Prof. Reza Iravani and Prof. Josh Taylor

for their insight and their valuable and thoughtful feedback. Also, I would like to thank Prof. Bruno

Allard and Prof. Zeb Tate for their insightful comments and questions in the final examination.

I would like to also thank our industry partners John Milios and Victor Marten for introducing an

exciting project to us and giving us the opportunity to work on it.

I take this opportunity to record my grateful regards to all my friends and colleagues in the

Laboratory of Power Management and Integrated SMPS, Amir Parayandeh, Aleksandar Radić,

SM Ahsanuzzaman, Behzad Mahdavikhah, Conny Huerta Oliviares, Adrian Straka, Tim McRae,

Parth Jane, Amr Amin, Nenad Vukadinović, Maryam Amouzandeh, and Shadi Dashmiz. Also, I

would like to thank my friends, Hazem Soliman, Ahmed Mohamed, Mohamed Ramadan, Essam

El Sahwi, and Sherif Helmy.

My deepest gratitude goes to my parents, my wife, my newborn son, my sisters, and my brother-

in-law for their support and understanding. My father’s personality and attitudes always inspired

me trying to be a better student.

v

Finally, I would like to gratefully acknowledge the financial support of the Department of

Electrical and Computer Engineering at the University of Toronto.

vi

Dedication

To Sarah and Ali…

vii

Table of Contents

Acknowledgement ……………………………………………………………………………….iv

Dedication………………………………………………………………………………………...vi

Table of Contents………………………………………………………………………………...vii

List of Figures………………………………………………………………...…………………...x

List of Tables…………………………………………………………………………………..xviii

List of Acronyms………………………………………………………………………………...xix

Chapter 1 Introduction………………………………………………………………………......1

1.1 Motivation…………………………………………………………………………………..1

1.2 Thesis Objectives…………………………………………………………………………...5

1.3 Thesis Organization and Contributions…………………………………………………….6

1.4 Thesis Outline………………………………………………………………………………9

1.5 References…………………………………………………………………………………..9

Chapter 2 Background and Previous Art………………………………………………………12

2.1 Passive Balancing Circuits………………………………………………………………...13

2.2 Active balancing Circuits………………………………………………………………….14

2.2.1 Switch-Capacitor Based Topologies………………………………………………….15

2.2.2 Inductor-Based Topologies…………………………………………………………...18

2.3 Conclusions…………………………………………………………………………..........26

2.4 References…………………………………………………………………………………26

Chapter 3 Assisting Converter Based Integrated Battery Management System for Electromobility

Applications……………………………………………………………………………………...32

3.1 Introduction………………………………………………………………………………..33

3.2 Principle of Operation……………………………………………………………………..35

viii

3.2.1 Step up function and system level efficiency………………………………………… 36

3.2.2 Cell balancing during movement, plug-in, and standstill modes of operation ……… .38

3.3 Assisting Converter Based on Isolated Dual Active Bridge Topology……………….......40

3.4 Practical Implementation……………………………………………………………….....44

3.4.1 Controller……………………………………………………………………………..44

3.4.2 Snubber Capacitors and EMI Minimization…………………………………………...57

3.4.3 Input Filter for Reducing Battery Current Ripple……………………………………..59

3.4.4 Efficiency Improvement at Light-to-Medium Loads………………………................60

3.4.5 Gate Driving…………………………………………………………………………..60

3.4.6 Volume of Passive Components and Silicon Area Comparison……………………...62

3.5 Experimental Results……………………………………………………………………...65

3.6 Conclusions………………………………………………………………………………..71

3.7 References…………………………………………………………………………………72

Chapter 4 High Power Density Assisting Step-Up Converter with Integrated Battery Balancing

Feature…………….……………………………………………………………………………...79

4.1 Introduction………………………………………………………………………………..80

4.2 Principle of Operation……………………………………………………………………..82

4.2.1 Assisting Conversion…………………………………………………………………82

4.2.2 Cells Balancing……………………………………………………………………….83

4.3 Practical Implementation………………………………………………………………….83

4.3.1 Controller……………………………………………………………………………..87

4.3.2 Comparison of Passive Components Volume………………………………………...88

4.3.3 Silicon Area Comparison……………………………………………………………..91

4.4 Experimental Results……………………………………………………………………...93

4.5 Conclusions………………………………………………………………………………..97

ix

4.6 References…………………………………………………………………………………97

Chapter 5 Other Possible Applications for Assisting Converters……………………………102

5.1 Introduction………………………………………………………………………………103

5.2 Uninterruptible Power Supplies and Data Centers………………………………………103

5.3 Low-to-Medium Scale Grid Storage Systems and Smart Homes………………….……105

5.4 Datacom and Wireless Communication systems……………………………………........107

5.5 Conclusions………………………………………………………………………………108

5.6 References……………………..…………………………………………………………108

Chapter 6 Conclusions and Future Work…………………………………………………….111

6.1 Conclusions………………………………………………………………………………112

6.2 Future Work……………………………………………………………………………...113

6.3 References………………………………………………………………………………..114

Appendix A Calculation of Power Transfer Equations for Square Wave Ac Voltages ………115

Appendix B Cost Analysis and Suitability for a Large Number-of-Cells System...................118

B.1 References……………………………………………………………………………….122

x

List of Figures

Figure 1.1. Typical battery power management system…………………………………………..2

Figure 1.2. Conventional battery power management system of a portable device………………3

Figure 1.3. The proposed battery management architecture based on the assisting converter……5

Figure 1.4. Assisting converter based on multi-module dual-active bridge (DAB) converter……7

Figure 1.5. Assisting converter based on two-module Flyback converter………………………..8

Figure 2.1. Shunt resistor passive balancing circuit……………………………………………..13

Figure 2.2. Shunt mosfets passive balancing circuits……………………………………………14

Figure 2.3. Single-tiered switch-capacitor topology……………………………………………..15

Figure 2.4. Double-tiered switch-capacitor topology……………………………………………16

Figure 2.5. Single switched capacitor topology………………………………………………….17

Figure 2.6. Buck-boost based active balancing circuit…………………………………………..19

Figure 2.7. Cuk based active balancing circuit…………………………………………………..20

Figure 2.8. Buck-boost + Cuk based active balancing circuit…………………………………...20

Figure 2.9. Multi-winding flyback based active balancing circuit………………………………21

Figure 2.10. Two-winding flyback based active balancing circuit………………………………22

xi

Figure 2.11. Two-stage flyback based active balancing circuit………………………………….24

Figure 2.12. Multi-winding forward transformer based active balancing circuit………………..25

Figure 3.1. (a) Conventional battery power management system for automotive applications (top);

(b) assisting converter based architecture (bottom)………………………………………………34

Figure 3.2. Multi-input assisting converter………………………………………………………37

Figure 3.3. Operation during movements………………………………………………………...39

Figure 3.4. Plug-in mode of operation……………………………………………………………39

Figure 3.5. Operation during standstill…………………………………………………………...40

Figure 3.6. Assisting converter based on multi-phase dual-active bridge (DAB) converter…….41

Figure 3.7. Alternative implementation based on multi-winding transformer…………………..42

Figure 3.8. A dual-active bridge (DAB) converter module……………………………………...43

Figure 3.9. Key voltage waveforms of a DAB module.................................................................43

Figure 3.10. The block diagram of the used controller……………………………………...…...44

Figure 3.11. Gate driving sequence of primary side modules (Fig.3.6) for the simultaneous cell

balancing (left) and one-by-one cell balancing (right) for motoring mode. Top four waveforms:

gate drive signals of Qk1 transistors (Fig.3.4) of the four cells on the primary side; Bottom

waveforms: gate drive signals of the transistor Q1 on the secondary side……………………....46

xii

Figure.3.12. Simulation results for ac wave forms for primary and secondary sides during

simultaneous balancing during motoring mode of operation when cells are not balanced…..........47

Figure.3.13. Simulation results for converter operation when all cells are balanced……………48

Figure.3.14. Simulation results for converter operation during simultaneous balancing for battery

cells………………………………………………………………………………………………48

Figure 3.15. An example for energy transfer between battery cells during plug-in and standstill

modes of operation for simultaneous cell balancing (left) where energy is transferred from the top

two cells to the bottom two cells and one-by-one balancing (right) where energy is transferred

from the top three cells to the bottom cell………………………………………………………...49

Figure 3.16. Gate driving sequence of primary side modules (Fig.3.6) for the simultaneous cell

balancing (left) and one-by-one cell balancing (right) for regenerative braking. Top four

waveforms: gate drive signals of Qk1 transistors (Fig.3.4) of the four cells on the primary side;

Bottom waveforms: gate drive signals of the transistor Q1 on the secondary side……………......50

Figure.3.17. Simulation results for converter operation during simultaneous balancing in

regenerative braking mode……………………………………………………………………….50

Figure.3.18. Simulation results for converter operation during one-by-one balancing in motoring

mode……………………………………………………………………………………………...51

Figure.3.19. Simulation results for converter operation during transferring energy between battery

cells………………………………………………………………………………………………52

xiii

Figure.3.20. Simulation results for converter operation during one-by-one balancing in

regenerative braking mode……………………………………………………………………….52

Figure.3.21. Simulation results of transient response of the assisting DAB converter……………53

Figure 3.22. Secondary side phase shift modulator with integrated non-overlapping time

generator…………………………………………………………………………………………55

Figure 3.23. Key waveforms of the secondary side phase shift modulator with integrated dead time

block………………………………………………………………………………………...........55

Figure 3.24. K-output primary side phase shift modulator with merged dead-time generator…..56

Figure 3.25. Resonant charging/discharging of the parasitic capacitances……………………….57

Figure 3.26. Ringing and overvoltage across mosfets…………………………………………...58

Figure 3.27: Input current waveform of the DAB converter…………………………………….59

Figure 3.28. DAB with added snubber and filtering capacitors…………………………………59

Figure 3.29. Gate driving scheme, (a) gate driving scheme for cell 4 primary side module, (b) gate

driving schemer for cell 3 primary side module, (c) gate driving scheme for cell 2 primary side

module, (d) gate driving scheme for cell 1 primary side module, (e) gate driving scheme for

secondary side module…………………………………………………………………………...62

Figure 3.30. Normalized comparison between conventional and assisting DAB converters……..64

Figure 3.31. ZVS in primary side modules……………………………………………………….66

xiv

Figure 3.32. ZVS in secondary side module……………………………………………………66

Figure 3.33. Assisting DAB converter operation when the four cells are balanced, iP1, iP2, iP3, and

iP4 are the primary side currents, all current channels are 5A/Div., Vo is the output voltage

(20V/Div.)………………………………………………………………………………………..67

Figure 3.34. Assisting DAB converter operation for simultaneous balancing when

Vcell4>Vcell3>Vcell2>Vcell1, iP1, iP2, iP3, and iP4 are the primary side currents, all current channels are

5A/Div., Vo is the output voltage (20V/Div.)……………………………………………………..67

Figure 3.35. Assisting DAB converter operation for one-by-one cell balancing when top cell is

overcharged, iP1, iP2, iP3, and iP4 are the primary side currents, all current channels are 5A/Div., Vo

is the output voltage (20V/Div.)…………………………………………………………………68

Figure 3.36. Assisting DAB converter operation for energy transfer between the cells, Icell1, Icell2,

Icell3, and Icell4 are the cells currents, all channels are 2A/Div……………………………………68

Figure 3.37. Assisting DAB converter operation during regenerative braking emulation when the

four cells are balanced, Icell1, Icell2, Icell3, and Icell4 are the cells currents, all channels are 5A/Div….69

Figure 3.38. Efficiency curves…………………………………………………………………...70

Figure 4.1. Conventional battery power management system of a portable device……………..80

Figure 4.2. Assisting flyback based architecture ………………………………………………..81

Figure 4.3. Practical implementation of the assisting flyback and its digital controller…………84

Figure 4.4. Gating sequences for different scenarios…………………………………………….84

xv

Figure 4.5. Simulation results for operation of the converter under balanced condition and heavy

load……………………………………………………………………………………………….85

Figure 4.6. Simulation results for operation of the converter under unbalanced condition and heavy

load……………………………………………………………………………………………….85

Figure 4.7. Simulation results for operation of the converter under balanced condition and light

load……………………………………………………………………………………………….86

Figure 4.8. Simulation results for operation of the converter under unbalanced condition and light

load……………………………………………………………………………………………….86

Figure 4.9. Simulation results for transient response of the assisting flyback converter during

heavy-to-light and light-to-heavy loads………………………………………………………….87

Figure 4.10. Assisting flyback converter waveforms after adding decoupling capacitors………..89

Figure 4.11. (a) Input voltage drop calculation for assisting flyback converter, (b) input voltage

drop calculation for conventional boost converter………………………………………………..90

Figure 4.12. Normalized values for passive components and silicon area comparisons………...92

Figure 4.13. Circuits’ components contribution to the overall volume of both converters……..92

Figure 4.14. Assisting converter operation when the two cells are balanced at heavy load, flyback

currents, Vout is the output voltage measurement (10V/Div.). ifb1 is the bottom module flyback

current (2A/Div.). ifb2 is the top module flyback current (2A/Div.)……………………………...94

xvi

Figure 4.15. Assisting converter operation when the top cell is overcharged at heavy load, flyback

currents. Vout is the output voltage measurement (10V/Div.). ifb1 is the bottom module flyback


Figure 4.16. Assisting converter operation when the two cells are balanced at light load, flyback



Figure 4.17. Assisting converter operation when the top cell is overcharged at light load, flyback



Figure 4.18. Power processing efficiency curves comparison…………………………………...96

Figure 5.1. Ac system based data center………………………………………………………...104

Figure 5.2. Dc system based data center………………………………………………………...105

Figure 5.3. Low-to-medium scale grid storage system and smart home architecture……………106

Figure 5.4. Communication systems power architecture……………………………………….107

Figure A.1. DAB main waveforms……………………………………………………………..116

Figure B.1. Losses per primary side switch…………………………………………………….119

Figure B.2. Losses per secondary side switch……………………………………………….......119

Figure B.3. Losses per inductor…………………………………………………………………120

xvii

Figure B.4. Losses per transformer……………………………………………………………...120

Figure B.5. Losses per output capacitor…………………………………………………………120

Figure B.6. System losses……………………………………………………………………….120

Figure B.7. System’s efficiency………………………………………………………………...121

Figure B.8. System’s cost……………………………………………………………………….121

xviii

List of Tables

Table 2.1. Comparison between switch-capacitor based and inductor based active balancing

circuit…………………………………………………………………………………………….25

Table 3.1. Experimental setup components and parameters..........................................................71

Table.4.1. Experimental setup parameters………………………………………………………..96

xix

List of Acronyms

A Ampere AC Alternating Current ADC Analog-to-Digital Converter AH Ampere Hour BMS Battery Management System C Cent DAB Dual Active Bridge DC Direct Current Div. Division DPWM Digital Pulse Width Modulation EIS Electro-Impedance Spectroscopy EMI Electro-Magnetic Interference EV Electric Vehicle FPGA Field Programmable Gate Arrays

xx

GSM Global System for Mobile communications HEV Hybrid Electric Vehicle HV High Voltage Hz Hertz IC Integrated Circuit ICT Information and Communication Technology LCD Liquid Crystal Display LDO Low-dropout regulator LV Low Voltage Mosfet Metal Oxide Semiconductor Field Effect Transistor PC Personal Computer PID Proportional Integral Derivative PoL Point of Load PSM Phase Shift Modulator PV Photovoltaics PWM Pulse Width Modulation

xxi

RCD Resistance Capacitor Diode RF Radio Frequency RMS Root Mean Square SC Switch-Capacitor Sel. Selector SMPS Switch-Mode Power Supply SOC State of Charge SOH State of Health UPS Uninterruptible Power Supply V Volt W Watt ZVS Zero Voltage Switching

1

Chapter 1

Introduction

1.1 Motivation

In the recent years, battery-powered devices have been extensively used in numerous applications

ranging from low-power devices, such as smartphones, laptops, and tablet computers, consuming

up to few hundred watts, to medium-to-high power applications such as electric vehicles (EVs),

2

hybrid electric vehicles (HEVs), data centers, grid storage, and smart homes, consuming power up

to several gigawatts.

In portable and automotive systems, the battery management system (BMS) usually consists of the

following main blocks, shown in Fig.1.1: a number of series battery cells (a battery pack), a step-

up dc-dc converter [1]-[2], a battery charger, and, in very rare cases, a balancing circuit.

Step-updc-dc Converter

Balancing circuit

-

+

Vbatt

+

Vout

-

CoutLoad

BMS

ChargerAc voltage

Figure 1.1. Typical battery power management system.

Step-up dc-dc converters are needed since many functional blocks of these systems require a well-

regulated dc voltage that is higher than that of the battery pack.

The importance of the balancing circuits can be appreciated knowing that the individual cells of a

battery pack usually have different states of charge (SOC). The SOC differences happens due to

various reasons, such as manufacturing variations in physical volume, variations in internal

impedance of each cell, differences in self-discharge rates, aging effects, and thermal gradient

across the battery pack [3]. The SOC imbalance results in degradation in battery lifetime and

underutilization of the battery pack, as if one cell in the pack is close to being over-discharged or

3

over-charged, the discharging/charging process is stopped immediately, regardless the SOC of the

rest of the cells. This is done for safety reasons which are especially critical with Li-Ion batteries,

which are the most commonly used nowadays, and their overcharge can cause an explosion of the

battery pack.

The balancing circuits can be divided into two main categories. The first one is the passive

balancing circuits which revolve around dissipating the excessive energy through resistors. The

disadvantages of these circuits are low efficiency and the related heating issues. The second

category comprises of the active balancing circuits, which are far more efficient. The active

balancing circuits are transferring the excessive energy from the overcharged cells to the

undercharged ones through auxiliary dc-dc converters [4]-[11]. Even though the active cell

balancing can increase the battery lifetime by a factor of three, their use is relatively sparse, due to

the overly large extra cost and weight they add to the system.

Dc-dcbus

converter

+PoL

PoL

LDO

Loads

Cbus

Vout

+

_

+Vbatt

Balancingcircuit

Step-up dc-dc converter

Vcell1

Vcell2

-

Vbus V1

V2

Vn

+

_

_

BMS

Figure 1.2. Conventional battery power management system of a portable device.

4

Two widely used battery management systems are of interest in this work. The first one is the

battery management system for automotive applications shown, in Fig.1.1. The second one is the

battery management system for low-power portable electronics, such as smartphones and tablet

PCs, shown in Fig.1.2. These two systems are selected since it is well known that for these two

applications volume, cost, efficiency, and weight are crucial requirements. High power density,

i.e. low volume/weight combined with high power processing efficiency, is crucial in both of these

applications since it translates into a longer battery lifetime/ range and lower overall

volume/weight of the device. Both of these are of critical importance for the targeted applications.

In a longer term, the improved power density also results in cost reduction which is an important

factor in widening the market of these systems. It should be noted that improving the efficiency

and reducing the volume need to be done simultaneously, since reducing the volume without

improving the efficiency results in higher heat density and increased cooling system requirements,

which can nullify all advantages obtained through volume reduction.

Although these design requirements are essential for these systems, the conventional battery

management systems in the targeted applications still take a significant amount of the overall

device volume and weight and usually do not utilize cell balancing. The large volume is usually

caused by the use of relatively low power density boost converters in these applications, which

occupy a significant portion of the total volume and weight of the two battery management systems

of the interests. In automotive systems they often provide the full motor power [1] and in portable

systems, they are usually used to supply the back panel lighting, which consumes up to 60% of the

total power in some devices such as Samsung Galaxy tab [12]-[13]. Also, the existing solutions to

provide cells balancing add a significant extra cost, volume, weight, or degrade system efficiency,

as explained earlier. Therefore the manufacturers of these systems are usually left with two bitter

5

options of either eliminating the balancing circuits and sacrificing the battery lifetime or using the

existing solutions and increasing the volume, weight, and cost of the devices.

1.2 Thesis Objectives

The main goal of this thesis is to introduce a new battery management architecture that provides a

balancing feature, reduces overall volume and the cost of the system, and improves the system

efficiency at the same time. These advantages are achieved by integrating the voltage step-up, and

balancing functions inside a single converter topology, which processes only a portion of the

output power unlike the existing solutions. The introduced architecture, named assisting converter

architecture [14], is shown in Fig.1.3. It operates at such that the stepped-up output voltage is

formed as a sum of the battery pack voltage Vbatt and the output voltage of a bi-directional multi-

input single output converter, Vcf, where the inputs of the converter are connected to the battery

Step-updc-dc converter

&Balancing circuit

-

+

Vbatt

+

Vout

-

Cf

Load

Vcf

+

-

Iout

Figure 1.3. The proposed battery management architecture based on the assisting converter.

6

pack. In other words, instead of providing the entire output voltage and power, the converter in

this configuration is just assisting the battery, by providing a portion of power. This portion of

power is proportional to the difference between the output and the battery pack voltages.

1.3 Thesis Organization and Contributions

Two types of converters utilizing the assisting architecture are developed in this thesis, namely, a

multi-phase assisting dual active bridge (DAB) converter for automotive applications and a two-

phase assisting bi-directional flyback converter for portable applications.

In the first part of the thesis, assisting DAB converter for automotive applications, shown in

Fig.1.4, is described. Also, a complementary controller that performs cells balancing and regulates

the output voltage at the same time is presented. This DAB combines the balancing and step-up

functions in one stage and results in a smaller volume than the conventional boost-based or DAB

solutions. The advantages over the conventional solutions are obtained because the DAB assisting

converter processes only a 66% of the rated output power of the conventional solutions, allowing

for cost-effective implementation. Furthermore, the assisting DAB converter has better power

processing efficiency, compared to the conventional boost and DAB converters providing the same

amount of power at the output. The assisting DAB converter has the peak power processing

efficiency of 95.4% with almost flat efficiency curve, which is comparable or even better than the

best state-of the-art solutions [15].

In the second part of the thesis, the assisting converter of Fig.1.5, based on flyback converter is

described. This converter is designed for portable applications supplied by two or more battery

cells. The controller of this topology is implemented based on the pulse width modulation (PWM).

7

This converter also shares the properties of general assisting architectures, combining the

balancing and step-up functions in one stage. In comparison with commonly used boost that has

approximately the same power processing efficiency curve with a peak efficiency of 93.4%, the

assisting flyback processing 45% of the output power and has about 23% smaller overall volume.

Figure 1.4. Assisting converter based on multi-module dual-active bridge (DAB) converter.

8

To summarize, the main contributions of this thesis is a new power management architecture, and

two converter topologies utilizing the architecture, namely:

A dual-active bridge based assisting converter for automotive applications that combines

the balancing and step-up functions with reduced volume and better power processing

efficiency than the conventional solution; and its accompanying digital controller.

A flyback based assisting converter for low power applications that also combines the

balancing and step-up functions and has a smaller volume and approximately the same

power processing efficiency compared to the conventional solution.

Figure 1.5. Assisting converter based on two-module Flyback converter.

9

1.4 Thesis Outline

The thesis is organized as follows:

Chapter 2 reviews the main challenges regarding the balancing circuits and emphasizes on the

drawbacks that make the use of balancing circuits is relatively sparse.

Chapter 3 shows the use of the assisting conversion concept in the battery management system for

automotive applications. The principle of the operation of the converter is described and main

implementation issues for the introduced converter are addressed. This is followed by the

experimental results that confirm the advantages of the converter.

Chapter 4 describes the assisting flyback step-up dc-dc converter for battery management system

of low power applications. A practical implementation and experimental results are also shown.

Chapter 5 investigates other possible applications utilizing the assisting concept. Three possible

applications are studied. As the first possible application, the converters for uninterruptible power

supplies (UPSs) and data centers are investigated. Then, possibilities of using the assisting

architecture in the converters for grid storage systems for smart homes is briefly reviewed, and

finally a potential use of the introduced architecture in converters for datacom and wireless

communication systems is addressed.

Finally, this thesis is concluded in chapter 6, summarizing the main contributions and possible

directions for future research.

1.5 References

[1] A. Emadi, "Advanced Electric Drive Vehicles," Florida, CRC Press, 2014.

[2] "TI Tablet Solutions," Datasheet, Texas Instrument, 2013, available http://www.ti.com.

10

[3] C. Jian, N. Schofield, and A. Emadi, "Battery Balancing Methods: A Comprehensive

Review, " in Proc. IEEE Vehicle Power and Propulsion Conf., 2008; pp. 1–6.

[4] A. Baughman and M. Ferdowsi, "Double-tiered capacitive shuttling method for balancing

series-connected batteries," in Proc. IEEE Vehicle Power and Propulsion Conf., 2005,

pp.109-113.

[5] R. Fukui and H. Koizumi, "Double-tiered switched capacitor battery charge equalizer with

chain structure," in Proc. 39th Annu. IEEE Ind. Electron. Society Conf., 2013, pp.6715-

6720.

[6] C. Karnjanapiboon, Y. Rungruengphalanggul, and I. Boonyaroonate, "The low stress

voltage balance charging circuit for series connected batteries based on buck-boost

topology," in Proc. IEEE Circuits and Syst. Symp., 2003, pp.III-284,III-287.

[7] N. H. Kutkut, "A Modular Nondissipative Current Diverter for EV

Battery Charge Equalization," in Proc. 13th Annu. IEEE Appl. Power Electron.

Conf. Expo., 1998, pp. 686-690.

[8] X. Lu, W. Qian, F. Z. Peng, "Modularized Buck-Boost + Cuk Converter for High Voltage

Series Connected Battery Cells," in Proc. 27th Annu. IEEE Appl. Power Electron. Conf.

Expo., 2012, pp.2272-2278.

[9] "LTC3300-1 - High Efficiency Bidirectional Multicell Battery Balancer," Datasheet,

Linear technology, available www.linear.com.

[10] C. Bonfiglio and W. Roessler, "A Cost Optimized Battery Management System with

Active Cell Balancing for Lithium Ion Battery Stacks," in Proc. IEEE Vehicle Power and

Propulsion Conf., 2009, pp.304,309.

11

[11] S. Li, C. Mi, and M. Zhang, "A high efficiency low cost direct battery balancing circuit

using a multi-winding transformer with reduced switch count," in Proc. 27th Annu. IEEE

Appl. Power Electron. Conf. Expo., 2012, pp.2128-2133.

[12] R. M. Soneira, "Tablet Display Technology Shoot-Out," [online]. Available:

http://www.displaymate.com/Tablet_ShootOut_2.htm.

[13] D. Schmidt, "Samsung Galaxy Tab S 10.5 Tablet Review," [online]. Available

http://www.notebookcheck.net/Samsung-Galaxy-Tab-S-10-5-Tablet

Review.124253.0.html.

[14] A. Prodic, M. Shousha, V. Marten, and J. Milios, "Assisting Converter," US Patent

US8779700, July 2014.

[15] F. Krismer and J. W. Kolar, "Accurate power loss model derivation of a high-current dual

active bridge converter for an automotive application," IEEE Trans. Ind. Electron., vol.57,

no.3, pp.881-891, March 2010.

12

Chapter 2

Background and Previous Art

This chapter presents an overview of the previous work related to cells balancing circuits. At first

the passive balancing circuits, which evolve around dissipating energy from excessively charged

cells in shunt resistors or transistors, and their shortcomings of low efficiency and high temperature

13

rise are addressed in the first section. Second the active balancing circuits, which are way more

efficient than the passive balancing circuits, are discussed. The active balancing circuits can be

divided into two main categories; the first category is the switch-cap based topologies and the

second category is the inductor-based switch-mode power supply topologies. The different

topologies and their main advantages/disadvantages are addressed in the following sections

revealing the areas for further improvement which is the main condense of this thesis.

2.1 Passive Balancing circuits The basic idea of the passive balancing circuits [1]-[5] is to connect a resistor in parallel with each

cell to dissipate excessive energy of overcharged cells during charging or discharging processes

as shown in Fig. 2.1. During discharging process, the shunt resistor(s) is connected to overcharged

cell(s) during operation such that these cells supply more current than the rest of the cells which

leads eventually to cells balancing. During charging process, all switches are turned on at the same

time such that overly charged cells receive less current. This happens since these cells have higher

voltages than the rest of the cells and hence provide higher currents through the shunt resistors

which results eventually in less current into overcharged cells during charging.

Figure 2.1. Shunt resistor passive balancing circuit.

14

In order to add more control over the shunt current, a modification to shunt resistor passive

balancing circuits is done in [5]. The modification is to connect mosfets that operate in the linear

region in parallel with each cell such that the on-resistance of the mosfets, and hence the shunt

current, is proportional to the SOC of the associated cells as shown in Fig.2.2. The main drawbacks

of the passive balancing circuits generally are the low operating efficiency and high heat generation

which makes the use of these circuits unappealing in the targeted battery management systems.

2.2 Active Balancing Circuits The idea of the active balancing circuits [6]-[41] is based on transferring the energy of overcharged

cells to those with less charge rather than dissipating this energy in shunt elements. Since the

excessive energy is not dissipated, the active balancing circuits are way more efficient than the

passive balancing circuits. The active balancing circuits can be divided into two main categories;

the first category is the switch-capacitor based topologies and the second category is the inductor-

based switch-mode power supplies topologies. The different topologies in each category are

discussed in the next subsections.

Figure 2.2. Shunt mosfets passive balancing circuits.

15

2.2.1 Switch-Capacitor Based Topologies Switch-capacitor (SC) balancing circuits [6]-[17] utilize a network of switches and capacitors only

to transfer the excessive energy between battery cells. These balancing circuits operate by

capacitive energy transfer between the circuits’ ports rather than inductive energy transfer used in

inductor-based switch-mode power supply topologies. The next subsections describe the different

SC balancing circuits and address the advantages and disadvantages of these topologies.

2.2.1.1 Single-Tiered Switch-Capacitor

Single-tiered switch-capacitor topology [6]-[10] is the basic and the most straight forward switch-

capacitor topology. It requires N-1 capacitors and 2N bi-directional switches to balance N cells, an

example of 3 cells is shown in Fig.2.3. The balancing algorithm is simple as all switches are

controlled by the same signal such that the excessive energy is transferred among all cells

sequentially. For instance, when S1, S2, and S3 are in the upper positions, C1 and C2 are connected

in parallel with E1 and E2 and hence the capacitors are charged or discharged to have the same

Figure 2.3. Single-tiered switch-capacitor topology.

16

voltages of E1 and E2. After that, S1, S2, and S3 are in the lower position and hence connecting C1

and C2 to E2 and E3 respectively and the same process mentioned earlier happens. After few cycles

of this sequence, battery cells are balanced. The main advantages of this topology are simple

control algorithm, high power density [18], high efficiency at low-power applications up to few

watts [19], the topology can be used during charging or discharging processes, minimum

electromagnetic interference (EMI) since there are no magnetic components, and it is easy to

integrate the whole balancing circuit on chip. The main disadvantages are relatively long balancing

time since energy needs to shuttle through all battery cells, high inrush current at start-up, high

pulsating currents during normal operations which have a negative impact on battery lifetime, and

inherent energy loss associated with charge transfer between the capacitors [20].

2.2.1.2 Double-Tiered Switch-Capacitor

Double-tiered switch-capacitor topology [11]-[17] is a modified version of the single tiered switch-

capacitor topology. The only difference is that it uses two capacitor stages or tiers to provide more

current paths among the battery cells and hence reduce the impedance of the current path to speed

up cells balancing. It needs N capacitors and 2N bi-directional switches to balance N cells, an

Figure 2.4. Double-tiered switch-capacitor topology.

17

example of 3 cells is given in Fig. 2.4. This topology has the same advantages of the single tiered

switch-capacitor topology. Moreover, it reduces the balancing time to more than half [21] because

of the presence of the second capacitor tier. But still it has the same disadvantages of the single-

tiered switch-capacitor topology which are high in-rush current at start-up, high pulsating currents

during normal operation, the need to shuttle energy along the battery pack which limits balancing

speed and system efficiency, and inherent energy loss for complete charge/discharge cycle.

2.2.1.3 Single Switched Capacitor

Single switched capacitor topology [8]-[10] is another form of the basic single-tiered switch-

capacitor topology but it utilizes only one capacitor and N+5 bi-directional switches to balance N

cells which makes it more cost efficient and higher power dense than the two previously described

topologies. Figure 2.5 shows an example of a 3-cells single switched capacitor balancing circuit.

If the same simple control algorithm of the two previously mentioned topologies is employed,

Figure 2.5. Single switched capacitor topology.

18

balancing speed is limited to 1/N of that of the single-tiered switch-capacitor topology when using

the same capacitance value. However, different control algorithms can be used to speed up cells

balancing. One of these algorithms is to connect any two cells in parallel using the corresponding

switches which is known as cell to cell balancing method [8], but still the balancing speed becomes

low when both cells have close or equal voltages. The main disadvantages are since more switches

are used, higher switching and conduction losses are present in addition to the same disadvantages

associated with any switch-capacitor topology which makes the inductor-based SMPS based

topologies are more attractive for the targeted applications.

2.2.2 Inductor-Based Topologies

Inductor-based active balancing circuits operate based on inductive energy transfer as opposed to

capacitive energy transfer used by their switch-capacitor based counterparts. The inductor-based

balancing circuits can be divided into two main subcategories; the first subcategory is the non-

isolated dc-dc converters and the second subcategory is the isolated dc-dc converters. The

operation, advantages, and disadvantage of different topologies in each subcategory are described

in the following subsections.

2.2.2.1 Non-Isolated Dc-Dc Converters

The most commonly used non-isolated converters for cells balancing are buck-boost [22]-[26] and

Cuk [27]-[32] converters or a combination of them [33] as shown in Figs. 2.6, 2.7, and 2.8

19

respectively. The buck-boost balancing circuit is used to transfer energy indirectly between any

neighboring cells by taking the excessive energy of an overcharged cell, storing it in the

corresponding power transfer inductor (Li) of a module and then releasing it to the neighboring

undercharged cell. It has a modular design and requires N-1 inductors and 2N-2 switches to balance

N cells, an example of 3 cells is shown in Fig.2.6.

In Cuk converter based balancing circuit, energy is transferred directly between any two

neighboring cells using the Cuk converter formed by the capacitor (Ci) and inductors (Li) and (Li+1)

in the corresponding module. It has also a modular design and requires N-1 modules to balance N

cells. Each module has two inductors, two switches, and one capacitor, an example of 3 cells is

shown in Fig.2.7.

A hybrid between buck-boost and Cuk converters is proposed in [33] to reduce the number of

inductors and active components per cell without increasing the voltage stress across

semiconductor devices. This circuit has a modular structure as well and requires N/2 inductors,

N/2-1 capacitors, and N switches to balance N cells, an example of 4 cells is shown in Fig.2.8. The

balancing is done using a simple PWM control with a fixed duty cycle of 50% [33]. The top two

L1

Q1

Q2Q3

Q4

L2

E1

E2

E3Module

Q1

Q2

Ts

Figure 2.6. Buck-boost based active balancing circuit.

20

battery cells, E1 and E2, are balanced using the buck-boost converter formed by Q1, Q2, and L1

while the bottom two cells, E3 and E4, are balanced by the buck-boost converter formed by Q3, Q4,

and L2. In addition, the middle cells E2 and E3 are balanced using the cuk converter formed by Q2,

Q3, C1, L1, and L2. The gating pulses for the case when E1>E2 and E3>E4 are shown in Fig.2.8. The

main advantages of the non-isolated inductor-based balancing circuits are small components count

and they operate with fairly simple PWM controllers [34]. On the other hand, these topologies

L1

Q1

Q2Q3

Q4

L3

E1

E2

E3

C1

C2

Module

L2

L4

Q1

Q2

Ts

Figure 2.7. Cuk based active balancing circuit.

L1

Q1

Q2

Q3

Q4

E1

E2

E3

C1

E4

L2

Module

t

t

t

t

Q1

Q2

Q3

Q4

Ts

Figure 2.8. Buck-boost + Cuk based active balancing circuit.

21

require fairly large passive components which add to the volume, weight, and cost of the battery

management system. Also, energy cannot be transferred between any two arbitrary cells directly

which limits balancing speed and system efficiency.

2.2.2.2 Isolated Dc-Dc Converters

The most commonly used isolated dc-dc converter for cells balancing is the flyback converter and

its different derivations [35]-[40]. The system presented in [35]-[38] utilizes a multi-winding

transformer and two switches (or one switch) per cell on the primary winding to transfer the energy

from an overcharged cell to the whole battery pack or to transfer the energy from the battery pack

to any undercharged cell(s). The mosfets in the primary sides are used to charge the magnetizing

E1

EN

E2

E3

1:1:…:n

Q11

Q21

Q31

QN1

D11

D21

D31

DN1

Q1

Lm

Rs1 Cs1

Ds1

Ds2

Rs2 Cs2

Ds3

Rs3Cs3

DsN

RsNCsN

Snubber circuit

Module

Figure 2.9. Multi-winding flyback based active balancing circuit.

22

branch inductance (Lm) in the first part of the cycle and the mosfet in the secondary side is used to

release the energy stored in Lm to the battery pack. Also, it can be controlled such that the secondary

side mosfet is used to charge Lm and the primary side mosfets are used to release the energy stored

in Lm to undercharged cells. The series diodes (D1, D2... DN1) are used to prevent cross conduction

between primary modules. Figure 2.9 shows the multi-winding flyback balancing circuit.

A similar approach is presented in [39]. The main difference is this system trades off the number

of switches with the number of transformer windings to reduce the volume and cost of the

balancing circuit. Fig 2.10 shows the two-winding flyback balancing circuit. The main advantages

E1

EN

E2

1:n

Q11

Q12

Q21

Q22

QN1

QN2

D11

D21

DN1

Q1

Lm

Module

Figure 2.10. Two-winding flyback based active balancing circuit.

23

of the two previously described solutions are simple control algorithm, small components count,

and flexible energy transfer. The main drawbacks are the low efficiency of the flyback converter

especially at higher power levels and transformer bulky size. These systems operate at relatively

low efficiency due to the presence of snubber circuits needed to limit voltage overshoot across

semiconductor devices by dissipating the energy stored in the leakage inductance of the flyback

transformer into the snubber resistance (Rs). The bulky size comes as a result of the storage

requirements of the flyback transformer.

A two-stage solution based on the flyback converter is presented in [40] to reduce voltage stress

across switches and hence overcome the consequent problems of high switching losses, high cost,

high on-resistance, and large silicon area. In the two previously described solutions, the secondary

side is connected to the battery back which makes the used semiconductor devices experience high

voltage stresses especially in applications that require a large number of cells such as automotive

applications. This can be solved by connecting the secondary side of the flyback transformer to an

intermediate capacitor C1 that serves as an input to the second stage. The first stage runs at a fixed

duty cycle and the capacitor voltage is controlled at the desired level defined by the blocking

voltage of the used semiconductor devices. The second stage is used to transfer the energy stored

in the intermediate capacitor to the battery pack. The main drawback of this solution is lower

system efficiency due to using two cascaded power stages. For example, if each stage runs at an

efficiency of a 90% the complete system has an overall efficiency of an 81%. Also, this work did

not address the problem of cross conduction between the modules and did not take into account

the bi-directional nature of the flyback converter which opens a possibility to transfer the energy

from the battery pack to any desired cell through the corresponding module. In addition, the

topology uses a large number of flyback bulky size transformers which reflects on the overall

24

volume and weight of the balancing circuit. Figure 2.11 shows the two stage flyback balancing

circuit.

Another approach based on connecting any unbalanced cells through a multi-winding forward

transformer rather than using a flyback converter is introduced in [41]. The balancing is done by

turning on and off all mosfets at the same time such that when battery cells have voltage

differences, the current flows from overcharged cell(s) to undercharged cell(s) and hence balancing

them. The main advantages of this system are it does not have the common problems associated

with the flyback converter such as bulky size due to its transformer storage requirements and

snubber circuits associated with the leakage inductance of the flyback transformer in addition to

simple control and inherent zero voltage switching. The main disadvantage is the balancing current

E1

EN

E2

E3

1:n1

Q1

Q2

Q3

QN

Lm

D1

D2

1:n1

1:n1

1:n1

D3

DN

C1

Q1CLm

D1C

1:n2

Module

Figure 2.11. Two-stage flyback based active balancing circuit.

25

is controlled by the voltage difference between cells and the parasitic resistance of the conduction

path which means the balancing becomes too slow for the energy transfer between cells having

similar output voltages. The multi-winding forward transformer based balancing circuit is shown

in Fig.2.12.

Table 2.1 shows a comparison between switch-capacitor based and inductor based active balancing

circuits summarizing key figures of merits for both approaches.

Table 2.1. Comparison between switch-capacitor based and inductor based active balancing circuit

Switch-Cap. based Inductor based Ability to operate during charging/discharging

Yes Yes

Cost Less expensive Relatively expensive Volume Smaller Relatively big Weight Lighter Relatively heavy

E1

Q11

1:….:1

Module

C11 C21

L11

Lm

E2

Q21C21 C21

L21

EN

QN1CN1 CN2

LN1

Figure 2.12. Multi-winding forward transformer based active balancing circuit.

26

Efficiency Low for high power applications and long-

stack batteries

Relatively higher

Flexibility to balance two arbitrary battery cells directly

Not flexible since energy need to shuttle along the

battery pack

Not flexible for non-isolated and flexible for isolated

topologies Suitability for long-stack batteries

Not suitable Suitable

2.3 Conclusions State of the art balancing circuits are investigated and reviewed in this chapter. From the conducted

literature review, it’s quite obvious that the state of the art balancing circuits either operate with

lossy resistive components or are implemented as auxiliary circuits that add to the cost, volume,

and weight of the system, in case of inductive based balancing circuits, or lacks balancing

efficiency and flexibility, in case of switch-cap based balancing circuit, which makes the use of

balancing circuits is rare in battery management systems regardless their importance.

2.4 References

[1] “bq76PL536,” Datasheet, Texas Instrument, [online]. Available: www.ti.com, Dec. 2009.

[2] “MAX11068”, Datasheet, Maxim Integrated, [online]. Available:

www.maxinintegrated.com, June 2010.

[3] W. C. Lee, D. Drury, and P. Mellor, "Comparison of passive cell balancing and active cell

balancing for automotive batteries," in Proc. IEEE Vehicle Power and Propulsion Conf.,

2011, pp.1-7.

[4] I. Aizpuru, U. Iraola, J. M. Canales, M. Echeverria, and I. Gil, "Passive balancing design

for Li-ion battery packs based on single cell experimental tests for a CCCV charging

mode," in Proc. Int. Clean Electrical Power Conf., 2013, pp.93-98.

27

[5] M. Uno, "Interactive charging performance of a series connected battery with shunting

equalizers," in Proc. 31st Annu. IEEE Telecom.Energy Conf., 2009, pp.1-4.

[6] G. A. Kobzev, "Switched-capacitor systems for battery equalization," in Proc. Int.

Scientific and Practical Conference of Students, Post-graduates and Young Scientists,

2000, pp.57-59.

[7] M. Daowd, N. Omar, P. van den Bossche, and J. van Mierlo, "Passive and Active Battery

Balancing Comparison based on MATLAB Simulation, " in Proc. IEEE Vehicle Power

and Propulsion Conf., 2011, pp. 1–7.

[8] C. Jian, N. Schofield, and A. Emadi, "Battery Balancing Methods: A Comprehensive

Review, " in Proc. IEEE Vehicle Power and Propulsion Conf., 2008, pp. 1–6.

[9] Z-G. Kong, C-B. Zhu, R-G. Lu, and S-K. Cheng, "Comparison and Evaluation of

ChargeEqualization Technique for Series Connected Batteries," in Proc .37th Annu. IEEE

Power Electron. Spec. Conf., 2006, pp. 1–6.

[10] S. W. Moore amd P. J. Schneider, "A Review of Cell Equalization Methods for Lithium

Ion and Lithium Polymer Battery Systems," in Proc. World Congress Society of Automatic

Engineers, 2001.

[11] M-Y. Kim, C-H. Kim, J-H. Kim, and G-W. Moon, "A Chain Structure of Switched

Capacitor for Improved Cell Balancing Speed of Lithium-Ion Batteries," IEEE Trans. Ind.

Electron., vol.61, no.8, pp.3989-3999, Aug. 2014.

[12] A. Baughman and M. Ferdowsi, "Double-tiered capacitive shuttling method for balancing

series-connected batteries," in Proc. IEEE Vehicle Power and Propulsion Conf., 2005,

pp.109-113.

28

[13] R. Fukui and H. Koizumi, "Double-tiered switched capacitor battery charge equalizer with

chain structure," in Proc. 39th Annu. IEEE Ind. Electron. Society Conf., 2013, pp.6715-

6720.

[14] C. Pascual and P. T. Krein, “Switched capacitor system for automatic series battery

equalization,” in Proc. 12th Annu. IEEE Appl. Power Electron. Conf. expo., 1997, pp. 848-

854.

[15] C. Pascual and P. Krein, "Switched capacitor system for automatic series battery

equalization," Modern Techniques and Technology, 2000, pp. 57-59.

[16] A. Baughman and M. Ferdowsi, "Analysis of the Double-Tiered Three-Battery Switched

Capacitor Battery Balancing System," in Proc. IEEE Vehicle Power and Propulsion Conf.,

2006, pp.1-6.

[17] A. C. Baughman and M. Ferdowsi, "Double-Tiered Switched-Capacitor Battery Charge

Equalization Technique," IEEE Trans. Ind. Electron., vol.55, no.6, pp.2277-2285, June

2008.

[18] S. R. Sanders, E. Alon, H-P. Le, M. D. Seeman, M. John, V. W. Ng, "The Road to Fully

Integrated DC–DC Conversion via the Switched-Capacitor Approach," IEEE Trans. Power

Electron., vol.28, no.9, pp.4146-4155, Sept. 2013.

[19] G. Zhu and A. Ioinovici, "Switched-capacitor power supplies: DC voltage ratio, efficiency,

ripple, regulation," in Proc. IEEE Circuits and Syst. symp., 1996, pp.553-556.

[20] S. Ben-Yaakov, “Switched capacitor converters,” in Proc. IEEE Appl. Power Electron.

Conf. and Expo., 2009, available www.ee.bgu.ac.il/∼pel/seminars/APEC09.pdf.

29

[21] M. Daowd, M. Antoine, N. Omar, P. van den Bossche,

and J. van Mierlo, “Single Switched Capacitor Battery Balancing

System Enhancements,“ Energies, Vol.6, pp.2149-2174,2013.

[22] C. Karnjanapiboon, Y. Rungruengphalanggul, and I. Boonyaroonate, "The low stress

voltage balance charging circuit for series connected batteries based on buck-boost

topology," in Proc. IEEE Circuits and Syst. Symp., pp.III-284,III-287, 2003.

[23] J. Xu, S. Li, C. Mi, Z. Chen, and B. Cao, "SOC Based Battery Cell Balancing with a Novel

Topology and Reduced Component Count," Energies, Vol.6, 2726-2740, 2013.

[24] Yevgen Barsukov, "Battery Power Managemnt for Portable Devices," Artech House

Publishers, 2013.

[25] J. F. Reynaud, C. E. Carrejo, O. Gantet, P. Aloïsi, B. Estibals, and C. Alonso, "Active

Balancing Circuit for Advanced Lithium-Ion Batteries Used in Photovoltaic Application,"

in Proc. Renewable Energies and Power Quality Conf., 2010.

[26] N. H. Kutkut, "A Modular Nondissipative Current Diverter for EV

Battery Charge Equalization," in Proc. 13th Annu. IEEE Appl. Power Electron.

Conf. Expo., 1998, pp. 686-690.

[27] Y-S. Lee and M-W. Cheng, "Intelligent Control Battery Equalization for Series Connected

Lithium-Ion Battery Strings," IEEE Trans. Ind. Electron., vol. 52, pp. 1297-1307, 2005.

[28] Y-S. Lee, C-Y. Duh, G-T. Chen, and S-C. Yang, "Battery Equalization Using Bi-

directional Cuk Converter in DCVM Operation," in Proc. 36th Annu. IEEE Power Electron.

Spec. Conf., 2005, pp. 765-771.

30

[29] Y-S. Lee and J-Y. Duh, "Fuzzy-Controlled Individual-Cell Equaliser Using Discontinuous

Inductor Current-Mode Cuk Convertor for Lithium-Ion Chemistries," IEE Electric Power

Applications Proc., vol. 152, pp. 1271-1282, 2005.

[30] Y-S. Lee, C-E. Tsai, Y-P. Ko, and M-W. Cheng, "Charge Equalization Using Quasi-

Resonant Converters in Battery String for Medical Power Operated Vehicle Application,"

in Proc. Power Electron. Conf., 2010, pp. 2722-2728.

[31] S-M. Wang, M-W. Cheng, Y-S. Lee, R-H. Chen, and W-T. Sie, "Intelligent Charged

System for Lithium-Ion Battery Strings," in Proc. 31st Annu. IEEE Int. Telecommun.

Energy Conf., 2009, pp. 1-6.

[32] M-W. Cheng, S-M. Wang, Y-S. Lee, and S-H. Hsiao, "Fuzzy Controlled Fast Charging

System for Lithium-Ion Batteries," in Proc. Power Electron. and Drive Syst. Conf., 2009,

pp. 1498-1503.

[33] X. Lu, W. Qian, F. Z. Peng, "Modularized Buck-Boost + Cuk Converter for High Voltage

Series Connected Battery Cells," in Proc. 27th Annu. IEEE Appl. Power Electron. Conf.

Expo., 2012, pp.2272-2278.

[34] A.V. Peterchev, X. Jinwen, and S.R. Sanders, "Architecture and IC Implementation of a

digital VRM controller," IEEE Trans. Power Electron., vol. 18, pp. 356–364, Jan 2003.

[35] "LTC3300-1 - High Efficiency Bidirectional Multicell Battery Balancer," Datasheet,

Linear technology, [online]. Available: www.linear.com.

[36] Y. C. Hsieh, C. S. Moo, W. Y. Ou-Yang , "A Bi-directional Charge Equalization Circuit

for Series-connected Batteries," in Proc. Power Electron. and Drives Syst. Conf., 2005,

pp.1578-1583.

31

[37] Y. Zeng, Z. He, M. Gao, and H. Pan, "An Active Energy balancing System for Lithium-

Ion Battery Pack," in Proc. Computer Science and Electron. Engineering Conf., 2013, pp

118-121.

[38] C. Bonfiglio and W. Roessler, "A Cost Optimized Battery Management System with

Active Cell Balancing for Lithium Ion Battery Stacks," in Proc. IEEE Vehicle Power and

Propulsion Conf., 2009, pp.304,309.

[39] J-W. Shin, G-S. Seo, C-Y. Chun, and B-H. Cho, "Selective Flyback Balancing Circuit with

Improved Balancing Speed for Series Connected Lithium-Ion Batteries," in Proc. Int.

Power Electron. Conf., 2010, pp.1180-1184.

[40] H-S. Park, C-E. Kim, G-W. Moon, J-H. Lee, and J. K. Oh, "Two-Stage Cell Balancing

Scheme for Hybrid Electric Vehicle Lithium-Ion Battery Strings," in Proc. 38th Annu. IEEE

Power Electron. Spec. Conf., 2007, pp.273-279.




32

Chapter 3

Assisting Converter Based Integrated Battery

Management System for Electromobility Applications

This chapter introduces a high power density battery management system for electromobility

applications that integrates voltage step-up and cell balancing functions inside a single converter

topology. The introduced system is based on assisting conversion concept, implemented with

multiple dual active bridge (DAB) converter modules. The assisting conversion, where the

33

converter is only processing a portion of the output power proportional to the voltage difference

between the battery pack and the output of the converter, allows for the use of a smaller converter

than those of conventional systems. Also, at the system level, this type of conversion results in a

higher power processing efficiency. The modular structure of the converter provides multiple

connections to battery cells and allows on-line cell balancing, both during charging and

discharging of the battery pack. The operation of the system is regulated by a practical digital

controller that performs cell balancing and, at the same time, regulates the output voltage. The

principle of operation and the practical implementation aspects of the assisting DAB converter are

explained and addressed in the following sections.

3.1 Introduction A general power management system for electromobility applications, such as electric vehicles

(EV) [1], hybrid electric vehicles (HEV) [2], small task-oriented vehicles (STOV), E-bikes, or

electric scooters [3]-[4], is shown in Fig.1.a. It consists of a battery pack, a step-up stage (usually

a bidirectional boost-based converter), an on-board or an off-board battery charger, and a motor

drive providing power for an electric motor. Usually, all the blocks are implemented separately.

In some, relatively rare cases, power management systems also include an additional cell balancing

circuit [5]-[19], which compensates for different states of charges (SOC) of individual cells,

occurring due to the variations in the manufacturing process, aging, and other external influences

[20]. In most frequently used architectures, having long strings of cells connected in series, for

safety and cell protection reasons, the charging process stops when the cell with the highest SOC

is charged, even though other cells have not been fully charged. Similarly, the battery pack is

disconnected when the cell with the lowest SOC reaches a low threshold value, even though the

34

other cells are still able to provide power. The balancing circuits [5]-[19] minimize the SOC

differences between the cells and, in that way, significantly extend the effective use of the battery

pack.

Generally, the balancing circuits can be divided into passive and active balancing systems. In

passive systems cells are balanced by dissipating energy from excessively charged cells, through

resistors [5]-[6]. The active balancing systems are far more efficient. In these systems, the energy

of over-charged cells is transferred to those with less charge using dc-dc converters [7]-[19].

Vba

tt

Vou

t

Figure 3.1. (a) Conventional battery power management system for automotive applications (top); (b) assisting converter based architecture (bottom).

35

Even though the benefits of the active cell balancing are known, their use in the targeted

electromobiliy applications is relatively sparse. This is mostly due to an overly large extra

weight/volume and cost they add to the system, which in the targeted mobility applications are of

a key importance and carefully controlled [20].

The primary objective of this chapter is to introduce the battery management architecture of Fig.3.1

(b) that integrates voltage step-up and active balancing functions inside a single converter stage.

At the same time, the entire new architecture has a smaller volume and better power processing

efficiency than a single step-up module of the conventional solutions. By combining functions and

reducing the volume, the introduced system can potentially compensates for the extra cost of the

balancing circuit, both in terms of the price and weight, allowing a wider adoption of active

balancing in the targeted electromobility applications. Also, a correlated potential benefit is an

extension of the range, due to improved power processing efficiency and reduction of the overall

weight of the system. The previously mentioned advantages are obtained by combining assisting

power conversion [21]-[23], where a relatively small step-up converter only processes a portion of

the output power proportional to the voltage difference between the outputs of the converter and

the battery pack, and implementation based on low-voltage dual active bridge (DAB) [24]-[26]

modules.

3.2 Principle of Operation

The architecture of Fig.3.1 (b) has multiple inputs and a single output. The inputs are connected

to the battery cells and the output, i.e. the output capacitor Cf, is placed on top of the battery pack,

forming the total output voltage Vout. Therefore, the converter only provides a portion of the

power, proportional to the difference between the total output voltage and that of the battery pack

36

Vbatt. This reduction in the power processing requirements allows a smaller converter to be used

and also results in systel-level power processing efficiency improvements, as explained in the

next subsection.

Various forms of partial power processing have been proposed for a number of applications [27]-

[33]. Examples include, converters processing difference in power between photovoltaic (PV) cells

[27]–[31], isolated converters for data centers [32], and a converter for a fully-electric airplane

[33]. In all of these systems, the converters processing only a portion of the output power have

demonstrated significant improvements in both power processing efficiency and volume

reduction.

In the new architecture of Fig.3.1 (b) the concept of partial power processing is also utilized and

the functions of the converter are integrated, to include both step-up and the cell-balancing features

without any additional hardware. Also, a single controller, providing regulation of both functions

is introduced, presenting a more reliable and cost-effective solution than two dedicated controllers

usually used in conventional balancing solutions (Fig.3.1 (a)).

3.2.1 Step up function and system level efficiency

The operation of the assisting dual active bridge architecture can be described through block

diagrams of Figs.3.1 (b) and 3.2. As mentioned earlier, it operates such that the stepped-up output

voltage is formed as a sum of the battery pack voltage Vbatt and the output voltage of a bi-

directional multi-input single output converter stage, Vcf, where the inputs of the converter are

connected to the battery cells. In other words, instead of providing the entire output voltage and

power, the converter in this configuration is just assisting the battery by providing a portion of

37

the power. This portion of the output power, Passisting, is proportional to the difference between

the desired output and battery pack voltages, as shown with the following equation:

VVV

PPout

battoutoutassisting

- , Eq.3.1

+-

Vassisting

Vout

+

-Vcelln

+

-

Vcell1-

+

Iout

Passisting n

Passisting1

Vcell2-

+

Passisting2+

-

Vbatt

Figure 3.2. Multi-input assisting converter.

where, referring to Fig.3.2, Passisting = Passisting1 + Passisging2+ … + Passistingn is the total power

delivered by all converter modules, Pout is the output power delivered to the load, Vbatt is the

battery pack voltage, and Vout is the output voltage of the converter.

This reduction in the power processing requirements allows a smaller converter to be used,

compared to conventional solutions of Fig.1.a, and also makes achieving a high power processing

efficiency at the system level easier. The efficiency improvements can be explained by looking

at the general equation for the system efficiency,

system Pout

Pout Plosses, Eq.3.2

38

and by combining (3.1) and (3.2)

11

1

1

assistingout

battout

system

V

VV

, Eq.3.3

where Plosses are power losses of the assisting converter, ηassisting is the efficiency of the assisting

converter, and it is assumed that the direct energy transfer from the battery is lossless. These

equations show that the system-level efficiency is significantly higher than that of the assisting

converter itself. For example if the converter processes a 20% of the output power and has 80%

efficiency, the overall system efficiency of a 95.2% is achieved (90.9% if a 40% of Pout is

processed). It means that, in order to achieve targeted overall system efficiency, the assisting

converter can be designed with much less stringent power processing requirements than the

conventional solutions, further reducing the overall system cost and complexity.

3.2.2 Cell balancing during movement, plug-in, and standstill modes of operation Two modes of operation take place during the vehicle/scooter movements, the first mode of

operation is motor mode of operation of the electric machine, i.e. motoring mode, and the second

mode of operation is generator mode of operation of the electric machine, i.e. regenerative braking.

In both modes, the battery charger is disconnected, as shown in Fig.3.3. During motoring mode,

the assisting converter is controlled to provide a regulated dc voltage, Vout., for the motor drive

while implementing battery cell balancing at the same time. The balancing is done by controlling

cells currents such that the cells with higher SOC provide more current than the others. During

regenerative braking, the motor drive operates as a rectifier with output voltage, Vreg. Since, as

indicated with Fig.3.2, the converter is bidirectional, during this mode the converter operates as a

39

step down stage and the braking energy is recycled back to battery pack. At the same time the cell

balancing is performed in a similar manner as in the previous case.

During the plug-in mode, shown in Fig.3.4, the vehicle/scooter cannot be driven, because of the

charging interlock [34]. In this mode the motor drive does not draw any current and the system

can be described with the diagram shown in Fig.3.4. Now, the assisting converter operates as an

Figure 3.3. Operation during movements.

Figure 3.4. Plug-in mode of operation.

40

active balancing circuit that transfers energy between cells, to achieve cell balancing. During this

mode the output voltage is still regulated in order to allow for bidirectional energy transfer.

At standstill, i.e. during parking or in traffic, the motor drive does not draw current, i.e.

considered to be disconnected, and the battery charger is not connected as well. During this mode

of operation, cell balancing is done, similar to plug-in mode, by transferring energy between cells

while maintaining the desired output voltage, as shown in Fig.3.5.

3.3 Assisting Converter Based on Isolated Dual Active Bridge Topology An implementation of the assisting converter based on a multi-phase isolated dual active bridge

converter is shown in Fig.3.6. In addition to providing bi-directional energy flow, the dual active

bridge (DAB) [24]-[26] has a number of other features that make it attractive for the targeted

applications. Those include high power processing efficiency (achieved through inherent zero

voltage or current switching) and much smaller inductor volume comparing to the conventional

hard switching and resonant topologies. The small inductance value opens a possibility for

Figure 3.5. Operation during standstill.

41

elimination of a discrete inductor through the utilization of the transformer leakage inductance.

The system of Fig.3.6 consists of a number of transformers whose primary windings are

connected to the individual battery cells and the secondary windings linked to the output

capacitor, through small inductors. An implementation based on a multi-tap transformer is also

possible as shown in Fig.3.7. A digital controller implementing phase-shift modulation regulates

the operation of this converter. The phase shift control provides both the output voltage regulation

and cell balancing through the regulation of the currents of the individual cells during motoring

Figure 3.6. Assisting converter based on multi-phase dual-active bridge (DAB) converter.

42

and regenerative braking modes or by transferring energy between battery cells during plug-in

and standstill modes of operation. The details of the digital controller are discussed in the next

section. It should be noted that the power transfer equation from only one cell to the output is

described by Eq.3.4. The derivation of the power transfer equation is done in appendix A.

Lπω

φπφVVnP

kkcfcell

k

)-(= , Eq.3.4

Figure 3.7. Alternative implementation based on multi-winding transformer.

43

where Vcell is the voltage of the battery cell, n the turns ratio of the transformer, Vcf is the voltage

of the floating capacitor (Figs. 3.1 (b), 3.6, and 3.7), = 2fsw where fsw is the switching

frequency of the converter, and φk is the phase shift (delay) between the voltages v1(t) and v2(t),

shown in Figs.3.8 and 3.9.

Figure 3.8. A dual-active bridge (DAB) converter module.

Figure 3.9. Key voltage waveforms of a DAB module.

44

3.4 Practical Implementation 3.4.1 Controller

The main goal of the controller of Fig.3.6 is to maintain the output voltage at the desired value

while providing cells balancing during the previously described modes of operation. The control,

whose more detailed diagram is shown in Fig.3.10, is performed through phase shift modulation,

where the angle on the secondary side is used for the output voltage regulation and set of the

angles on the primary (relative phase shifts) for cell balancing.

Mode select input, shown in Fig.3.6, is used to differentiate between the regenerative braking

mode and the rest of the modes of operation. For all modes of operation except for regenerative

braking, the controller regulates the output voltage at the desired value while controlling cells

currents during motoring mode and transferring energy between cells during standstill and plug-

Voltage Loop

PID Comp.

Vref [n]

Hvout [n]

+-

Secondary side phase

shift modulator

Relativephase shift calculator

Primary side phase

shift modulator

Power stage

Voltage attenuator

HADC

e[n]Secondary

pulses

Primary pulses

v[n]

vout(t)

Controller

SOC

r1[n] to r4[n]

Hvout(t)

Mode select

reg1[n] to reg4[n]

Figure 3.10. The block diagram of the used controller.

45

in modes. For these modes of operation, voltage loop is implemented in a digital fashion. The

attenuated output voltage Hvout(t) is converted into its digital equivalent with an analog-to-digital

converter (ADC) Hvout[n]. This value is then compared to the desired reference Vref[n] and the

resulting error e[n] is passed to voltage loop PID compensator. The compensator calculates a

value v[n], which is the input for the secondary side phase shift modulator. Based on this input,

the secondary side phase shifter adjusts the phase shift between the secondary side switches and

one of the set of primary side switches, i.e. reference set, such that the desired output voltage is

obtained.

The relative phase shifts between the DAB modules on the primary sides are adjusted based on

the cells state of charge (SOC). The calculation of the relative phase shifts between primary side

modules is performed by the primary side phase shift calculator, which sends two different

control signals sets, r1[n] to r4[n] and reg1[n] to reg4[n]. During all modes of operation except

for regenerative braking, only the first control signals set, r1[n] to r4[n], are used to generate

primary side gating pulses from primary side phase shift modulator.

During regenerative braking, the second set of control signals, reg1[n] to reg4[n], are used by the

primary side phase shift modulator and the secondary side phase shift modulator assigns a

predefined phase shift, rather than using the output of the PID compensator which takes place in

the rest of the modes, to the secondary side module such that secondary side gating pulses lead

primary side pulses and hence braking energy can be recycled back to battery cells.

46

With the respect of the methods for calculating the relative phase shifts for cell balancing, two

different methods have been investigated, named simultaneous and one-by-one cell balancing.

Key waveforms of the both schemes for a 4-cell battery system are shown in Figs.3.11, 3.12, and

3.13. In this case, the primary signals of the DAB module connected to the cell 4 are used as a 0

angle reference for setting up the phase shift of the voltage loop, i.e. for the anglev.

3.4.1.1 Simultaneous balancing

In simultaneous balancing, the relative phase shifts between the primary sides of the DAB

modules are assigned based on their SOC. Ideally, the relative angle for each of the modules

would be calculated from Eq.(4), based on the desired output voltage, required output power, and

SOC information.

Vcell1<Vcell2<Vcell3<Vcell4

Figure 3.11. Gate driving sequence of primary side modules (Fig.3.6) for the simultaneous cell balancing (left) and one-by-one cell balancing (right) for motoring mode. Top four

waveforms: gate drive signals of Qk1 transistors (Fig.3.4) of the four cells on the primary side; Bottom waveforms: gate drive signals of the transistor Q1 on the secondary side.

47

This control strategy would provide simultaneous charging/discharging of all cells that is in perfect

accordance with their SOC. However, from the practical point of view, the direct implementation

of such a method would be quite challenging. The challenges are related to high computational

requirements, especially when large numbers of battery cells are used. Hence, to reduce the

hardware complexity, a simplified balancing method is applied. In this implementation, one of 4

pre-defined discrete values of the relative phase shifts is assigned to each cell, depending on its

state of the charge. The gate driving signals of the transistors from Fig.6 for this modulation

scheme are shown in Fig.3.11. In this case it is assumed that cell 1 has the lowest SOC and,

therefore, Vcell1 is the lowest. Also, it is assumed that Vcell2 < Vcell3 < Vcell4. It can be seen that with

the respect of the signal on the relative phase shifts are set such that the cell with the lowest SOC

has the lowest angle and, according to Eq. (4), delivers the smallest amount of energy.

Figure 3.11 shows cell balancing scheme during motoring mode of operation. It should be noted

that during this mode of operation the lowest SOC cells are disconnected at medium-to-light loads

as explained later which results in secondary side phase shift value, v[n], is larger than the relative

phase shifts of the active phases which means energy transfer between battery cells is not used

during this mode of operation. Figs. 3.12-3.14 show simulation results for this mode of operation.

Figure.3.12. Simulation results for ac wave forms for primary and secondary sides during simultaneous balancing during motoring mode of operation when cells are not balanced.

48

During charging and standstill, the same primary gate driving signals, from Fig.3.11, are used. The

PID compensator naturally sets the secondary side phase shift at a value located between the

primary side relative phase shifts to keep that output voltage in regulation. That is done to allow

for bidirectional energy transfer and not to exceed the voltage rating of the secondary side devices.

Figure.3.13. Simulation results for converter operation when all cells are balanced.

Figure.3.14. Simulation results for converter operation during simultaneous balancing

for battery cells.

49

Under these conditions, energy is transferred between cells since secondary side phase shift is

located between the primary side relative phase shifts, r1 to r4, and primary modules with

overcharged cells are assigned gating pulses that lead the gating pulses of primary side modules

with undercharged cells. An example of gating pulses during these modes of operation where

energy is transferred from the top two cells to the bottom two cells is shown in Fig. 3.15.

During regenerative braking, the primary side phase shift modulator assigns different relative

phase shift set, reg1 to reg4 as shown in Fig. 3.16, to primary side modules such that secondary

side pulses lead the primary side pulses, and hence recycling energy back to battery cells. In this

case cell balancing is done, very similar to motoring mode, by making the cell with the lowest

SOC has the lowest angle and hence receives the smallest amount of energy. Figure 3.17 shows

simulation results for this mode of operation.

Q11

Q21

Q31

Q41

Q1

φr

φv

Ɵ π 2π 0

Set by the PID

Figure 3.15. An example for energy transfer between battery cells during plug-in and standstill modes of operation for simultaneous cell balancing (left) where energy is transferred from the

top two cells to the bottom two cells and one-by-one balancing (right) where energy is transferred from the top three cells to the bottom cell.

50

Figure 3.16. Gate driving sequence of primary side modules (Fig.3.6) for the simultaneous cell balancing (left) and one-by-one cell balancing (right) for regenerative braking. Top four waveforms: gate drive signals of Qk1 transistors (Fig.3.4) of the four cells on the primary side; Bottom waveforms: gate drive signals of the transistor Q1 on

the secondary side.

Figure.3.17. Simulation results for converter operation during simultaneous balancing

in regenerative braking mode.

51

3.4.1.2 One-by-one balancing

The simultaneous cell balancing method takes a lot of time for balancing the cells with similar

SOC and/or when most of the cells are close to their full capacity. As an alternative solution, one-

by-one cell balancing method is investigated. In this case the Phase Shift Calculator keeps all but

one primary side modules in phase, as shown in Fig. 3.11. The single module that is out of the

phase is used for cell balancing of the correspondent battery cell. The phase shift of this cell can

be arbitrary set, depending on the SOC of the cell and desired speed of balancing. This control

scheme allows fast balancing of individual cells even when they are close to their full state of

charge and/or have similar state as the other cells. In this case, again, the module connected to the

cell 1 has the lowest angle with respect to the secondary side and, therefore, delivers the smallest

amount of energy. It should be noted that during motoring, plug-in, standstill, and regenerative

braking modes the same approach explained with simultaneous balancing is utilized and shown in

the right hand side of Figs. 3.11, 3.15, and 3.16 respectively. Figures 3.18, 3.19, and 3.20 shows

the simulation results for one-by-one balancing technique during motoring, energy transfer

between cell, and regenerative braking modes of operation respectively.


motoring mode.

52

Also figure 3.21 shows simulation results of light-to-heavy and heavy-to-light transient response

of the assisting DAB converter.

Figure.3.19. Simulation results for converter operation during transferring energy between

battery cells.


regenerative braking mode.

53

3.4.1.3 Secondary Side Phase-Shift Modulator (PSM) with Merged Dead-Time Generator

The secondary side phase shift modulator (PSM) generates phase-shift modulated pulses for all

four transistors on the secondary side. It also incorporates a programmable non-overlapping time

generator (dead time generator), which prevents simultaneous conduction of two transistors

sharing the same branch. The operation of this block can be described by looking at the diagram

of Fig.3.22 and its key waveforms, shown in Fig.3.23.

The architecture of the secondary side PSM is a similar to that of a counter-based pulse width

modulator (PWM) [35]. However, the main difference is that, in this case, 4 pulses having

constant on time of:

dsw

on tT

t 2

, Eq.3.5

are produced and that the starting time, i.e. the rising edge, of the pulses is varied. In Eq.3.5, Tsw

is the switching period and td is the non-overlapping time.

Figure.3.21. Simulation results of transient response of the assisting DAB converter.

54

Also, one important difference in comparison with the conventional architecture is that the circuit

of Fig.3.22 incorporates dead-time block, which in other architectures is usually designed as a

separate block.

The same digital block is used throughout all modes of operation. That is done by using a single

2-to-1 multiplexer, shown in Fig.3.22, that allows selection between a predetermined phase shift,

φreg[n], chosen such that the maximum cell current is not exceeded, used in regenerative braking

and the output of the PID compensator, φv[n], used for the rest of modes of operation.

The phase shift modulated pulses are formed through comparisons of a staircase signal, created

by the Nc-bit counter, with four values, generated by the multiple-adder. The values at the output

of the adder are created based on two digital inputs, [n] and td[n], which are digital equivalents

of the desired phase shift in the time domain and the dead time, respectively. As shown in

Figs.3.22 and 3.23 the rising edges of the control signals for Q1 and Q4 (Fig.3.6) are formed when

the counter reaches the value [n] + td[n] and SR latch 2 is triggered. This pulse ends when the

counter counts to [n] + 2Nc-1 and the latch 2 is reset, where 2Nc-1 corresponds to the duration of

a Tsw/2 interval. Similarly, the rising and falling edges of the control signals for Q2 and Q3 are

formed when the counter [n] + td[n] + 2Nc-1.

55

Figure 3.22. Secondary side phase shift modulator with integrated non-overlapping time

generator.

Ɵ

2π 4π

c[n]

0

φ[n]+td[n]

φ[n]+2Nc-1

π -Ɵtdφ+Ɵtd

Q1 & Q4

Ɵ π -Ɵtd

tTswTsw/2-td Tsw/2-td 2Tsw

t

2Nc-1 2Nc-1

Ɵ

φ[n]

Ɵ 4π 2π 0 φ

Q2 & Q3

t2TswTsw

t

c[n]

φ[n]+td[n]+2Nc-1

2Nc-1 2Nc-1

Figure 3.23. Key waveforms of the secondary side phase shift modulator with integrated

dead time block.

56

3.4.1.4 Primary Side Phase-Shift Regulators (PSM) with Merged Dead-

Time Generator

The system of Fig.3.24 produces 4 phase-modulated signals with different r angles for a k-

module assisting converter. The four PSMs have the same architecture as the previously

described system of Fig.3.22 and operate in the same manner. To synchronize operation with the

primary side modulator, all 4 PSMs use the output signal of the Nc-bit counter of Fig.3.22, c[n],

as a reference ramp.

The eight pulse-shift modulated signals at the outputs of the PSMs, i.e. 1 to 4 and their

complements, are connected to the primary side gate drivers of the k modules (Fig.3.6), through

a set of 4-to-1 multiplexers, where 2 multiplexers are assigned to each of the k modules. The

phase shifts of individual modules are determined through m1 to mk control signals, that select

Figure 3.24. K-output primary side phase shift modulator with merged dead-time generator.

57

which of the four pulse-shift modulated signals will be passed to the individual modules based

on SOC of the cells. It can be seen that in this implementation all modules share the same set of

PSM blocks, and only two additional 4-to-1 multiplexers are required per module, i.e. per battery

cell. The mode select along with four 2-to-1 multiplexers are used, as described earlier, to select

between regenerative braking mode and the rest of modes of operation. In regenerative braking

mode, 4 different primary side phase shift values, as shown in Fig.3.24, are selected such that

secondary side pulses lead primary side pulses and hence braking energy is recycled to battery

cells while implementing cell balancing.

3.4.2 Snubber Capacitors and EMI Minimization

In dead time periods, the inductor current flows through drain-to-source capacitances of the

mosfets. This current discharges the capacitances of the mosfets that turn on after the dead time

period and charge the capacitances of the mosfets that turn off after the dead time period. For

illustration, one dead time period is shown for the primary side switches where the inductor

current, iLP, charges Cpar11 and Cpar22 and discharge Cpar13 and Cpar14 as shown in Fig.3.25.

Figure 3.25. Resonant charging/discharging of the parasitic capacitances.

58

During this time period, the inductor current flows through the parasitic capacitances, total path

inductance (sum of the power transfer inductor, leakage inductances of the transformers, and

parasitic inductance of the traces), and path resistances (the resistances of the transformer, traces

and the power transfer inductor). That results in resonant charging/discharging of parasitic

capacitances and high ringing frequency and voltage overshoot across mosfets take place, as

shown in Fig. 3.26. That happens due to small values of the parasitic capacitances, inductance,

and resistance [36]. The high ringing frequency has a negative impact on electromagnetic

compatibility (EMC) and the voltage overshoot can damage mosfets. Adding a small mica or

ceramic capacitor (Cs), shown in Fig.3.28, in parallel with each switch decreases the ringing

frequency and the voltage overshoot, and also aids in decreasing mosfets turning-off power losses

[26]. It should be noted that the same situation happens for the secondary side switches as well.

Figure 3.26. Ringing and overvoltage across mosfets.

59

3.4.3 Input Filter for Reducing Battery Current Ripple

The waveform of Fig. 3.27 shows that the current provided by battery cells has large variations.

These variations could reduce battery life time due to the heating effect because of the battery

internal resistance [37]. It’s recommended to keep the input current ripples when designing dc-

dc converters for electromobility applications below a 10% of the average input current [38].

To eliminate this effect, a decoupling capacitor Cdec is placed across the primary side bridge, as

shown in Fig. 3.28. Together with the parasitic inductance of the connecting wires; this capacitor

forms a second order filter that drastically reduces the input current ripple.

Figure 3.27. Input current waveform of the DAB converter.

Figure 3.28. DAB with added snubber and filtering capacitors.

60

3.4.4 Efficiency Improvement at Light-to-Medium Loads

The DAB converter runs at low efficiency at relatively light loads because of the large inductor

current under this load condition and also due to the loss of the ZVS [24], [39]. In order to

improve efficiency of the converter at light-to-medium loads; phase-shedding [40]-[41] is

utilized. In this technique, one or modules are disconnected such that the remaining modules are

able to provide the load requirements at better overall efficiency.

It should be noted that phase shedding does not help only in extending high efficiency range, but

it also aids cells balancing. The shedded phases are chosen to be the phases with the lowest SOC

cell so that the lowest SOC cells supply further less power.

3.4.5 Gate Driving

In the introduced architecture, primary and secondary side mosfets can be driven using simple

low/high side gate drives, i.e. half bridge gate drives, and level shifters supplied from the battery

cells and one auxiliary power supply, which can be a small battery placed on top of the battery

pack in its simplest form, such that there is no need for expensive isolation techniques such as

opto-couplers or pulse transformers. This driving scheme is suitable for switching frequencies

up to hundreds of kilo hertz. The signal mosfets along with the two zener diodes are used to level-

shift the gating signals produced by the controller, which shares the same ground with the power

stage, to a suitable level where it can turn on/off the low/high side gate drives. Also, a schottky

diode is connected in parallel with the zener diode at the input of the gate drives to reduce the

negative voltage across the input of gate drives. Figures 3.29(a)-3.29(e) show the driving scheme

for the primary and secondary sides’ mosfets. It should be noted that this driving scheme can be

used with a larger number of cells by using decentralized controllers for each group of cells.

61

Controller

Power stage ground

Cell 1 groundCell 2 ground

Cell 3 ground

12V from LDO with respect to the

cell ground

HS/LSGD

To the Power mosfets of

cell 4 primary side modules

Power stage ground

(a)

Controller

Power stage ground

Level shifter mosfet

R1

R2

10V

12-18V

Cell 3 ground

Cell 3 ground


cell 3 ground

HS/LSGD


cell 3 primary side module

Power stage ground

Vz=3.3V

From Vcell 2

3KΩ

1KΩ

1KΩ

Vz=10V

Schottky

(b)

Controller

Power stage ground


R1

R2

16V

16-24V

Cell 2 ground

Cell 2 ground


cell 3 ground

HS/LSGD

To the power mosfets of


Power stage ground

Vz=3.3V

From Vcell 1

3KΩ

2.5KΩ

1KΩ

Vz=16V

Schottky

(c)

62

Controller

Power stage ground


R1

R2

22V

28-36V

Cell 1 ground

Cell 1 ground


cell 3 ground

HS/LSGD



Power stage ground

Vz=3.3V

From aux. supply on top of the battery pack

2.2KΩ

Vz=22V

1.67KΩ

2KΩ

Schottky

(d)

Controller

Power stage ground


R1

R2

28VOutput ground

Output ground


cell 3 ground

HS/LSGD

To the power mosfet of the

secondary side module

Power stage ground

Divider ratio=1.75

Vz=3.3V

28-36VFrom aux. supply on

top of the battery pack

Schottky

Vz=28V

2.2KΩ

3KΩ

3.6KΩ

(e)

Figure 3.29. Gate driving scheme, (a) gate driving scheme for cell 4 primary side module, (b)

gate driving schemer for cell 3 primary side module, (c) gate driving scheme for cell 2 primary

side module, (d) gate driving scheme for cell 1 primary side module, (e) gate driving scheme for

secondary side module.

3.4.6 Volume of Passive Components and Silicon Area Comparison

3.4.6.1 Passive Components Volume Comparison

Since the conventional DAB converter is known to have higher power density than the

conventional hard switched PWM dc-dc converters, due to a small power transfer inductor that

can be integrated with the transformer and ZVS which enables operation at higher switching

frequency without affecting converter’s efficiency significantly [24], [42], the comparison is

63

held between the assisting DAB and the conventional DAB converters. Regarding the magnetic

components volume comparison, the assisting DAB converter has a 13% smaller power transfer

inductors volume and 26% smaller total transformers volume than those of a conventional DAB

converter that runs at the same switching frequency, has lower power processing efficiency, and

processes full output power, knowing that inductors volume is proportional to their energy

storage requirements ½ LI2 and forward transformers area product is proportional to their power

handling capability [43]. Transformer core volume is related to its area product, AP, by the

following equation [44]

)(75.0

APVolume Eq.3.6

The input filter capacitors are designed, for the targeted applications, such that the assisting and

conventional DAB converters have input current ripples less than a 10% of the average cells

currents [38]. The assisting DAB converter requires 80 µF input filter capacitors rated at cell

voltage, formed using a combination of low-equivalent series resistance (ESR) capacitors such

as 8*10µF, 6.3 V capacitors with ESR as low as 2.5mΩ [45], while the conventional DAB

converter requires a 55µF input capacitor rated at the full input voltage to achieve the design

criterion. Compared to the conventional DAB converter, the assisting DAB converter has around

64% smaller input capacitor volume, knowing that capacitors volume is proportional to their

energy storage requirements ½ CV2. It should be noted that both converters have approximately

the same input voltage variation when using the previously mentioned capacitance values.

The output capacitor of the assisting DAB converter is designed such that the voltage ripples

across the flying capacitor, Cf, in addition to voltage ripples across the input due to battery

internal resistance is equal to the voltage ripples across the output of the conventional DAB

converter and less than 1% of the rated output voltage [38]. The input filter, formed by the

64

parasitic inductance and the input filter capacitors, results in less than 10% current ripples drawn

from each cell and the internal resistance of the used cells is below 8mΩ [46]. Compared to the

conventional DAB, the assisting DAB converter has around 59% less output capacitor volume

when both converters have a 1% output voltage ripples [38].

3.4.6.2 Semiconductors Area Comparison

The minimum silicon area required to implement the assisting DAB and conventional DAB

converters can be compared by looking at the summation of the semiconductor stresses for each

converter. The assisting DAB converter has around 28% less silicon area than that of the

conventional DAB converter. The savings in silicon area comes as a result for using switches

with lower blocking voltages and less current stresses compared to the conventional DAB

converter. Figure 3.30 summarizes the normalized comparison between the assisting and

conventional DAB converters.

0 20 40 60 80 100

Inductors

Transformers

Output Cap.

Input Cap.

Switches area

Assisting DAB

Conventional DAB

Figure 3.30. Normalized comparison between conventional and assisting DAB converters.

65

3.5 Experimental Result To verify the previously described concepts, a 4-cell, 200W experimental setup has been built and

tested. At the input, four 6V, 12 AH Lead-Acid cells are used. The DAB stages operate at

switching frequency of 100 kHz and provide a 48V regulated output which makes the converter

nominally processes a maximum of a 66% of the output power. In the targeted automotive

applications 48V and 200W is used in small scooters [47]-[48]. It should be noted that the same

concept can be extended to larger number of cells and higher power applications such as electric

and hybrid electric vehicles as shown in appendix B. The power stage of the prototype is

implemented with discrete components and the controller is designed using an FPGA-based

development board. In this implementation SOC information is assumed to be available from an

external circuit operates based on cells terminal voltages. Table 3.1 shows experimental setup

components and parameters.

Figures 3.31 and 3.32 shows the ZVS of the primary side and secondary side switches of the

assisting DAB converter. Figures 3.33, 3.34, and 3.35 demonstrate operation of the assisting DAB

converter as an integrated step-up converter and a balancing circuit. Figure 3.33 shows the primary

side currents, which represents the power delivered by each phase, iP1 to iP4, (Fig.3.6) and the

output voltage for the case when the four cells are balanced. It can be seen that the converter

simultaneously provides equal current sharing and tight output voltage regulation. Figure 3.34

demonstrates the simultaneous cells balancing operation when Vcell4>Vcell3>Vcell2>Vcell1. It can be

seen that iP4 >iP3 >iP2 >iP1 while tight output voltage regulation is still obtained. Figure 3.35

demonstrates the one-by-one cell balancing when the top cell is undercharged. It can be seen that

the top cell provides less current than the rest of the cells while tight output voltage regulation is

still obtained. Figure 3.36 demonstrates the mode of operation where the energy can be transferred

66

between the cells which can be used at standstill and during plug-in modes. It can be seen from

Fig. 3.36 that the top three cells transfer power to the fourth cell. Also it should be noted that the

cells currents have small ripples thanks to the input filter as described in section 3.4.3.

Figure 3.31. ZVS in primary side modules.

Figure 3.32. ZVS in secondary side module.

67

Figure 3.33. Assisting DAB converter operation when the four cells are balanced, iP1, iP2, iP3, and iP4 are the primary side currents, all current channels are 5A/Div., Vo is

the output voltage (20V/Div.).

Figure 3.34. Assisting DAB converter operation for simultaneous balancing when

Vcell4>Vcell3>Vcell2>Vcell1, iP1, iP2, iP3, and iP4 are the primary side currents, all current

channels are 5A/Div., Vo is the output voltage (20V/Div.).

68

Figure 3.35. Assisting DAB converter operation for one-by-one cell balancing when

top cell is overcharged, iP1, iP2, iP3, and iP4 are the primary side currents, all current

channels are 5A/Div., Vo is the output voltage (20V/Div.).

Figure 3.36. Assisting DAB converter operation for energy transfer between the cells, Icell1,

Icell2, Icell3, and Icell4 are the cells currents, all channels are 2A/Div.

69

Regenerative braking mode is emulated by connecting a dc voltage source, which acts as the

output voltage of the motor drive during regenerative braking, across the battery and the flying

capacitor. Figure 3.37 shows cells currents during this mode of operation. It can be seen that cells

currents are negative which means energy is recycled back to battery cells. Also cells currents

are equally shared, since battery cells are balanced, and have low current ripples due to the

presence of the input filters.

Figure 3.37. Assisting DAB converter operation during regenerative braking emulation when

the four cells are balanced, Icell1, Icell2, Icell3, and Icell4 are the cells currents, all channels are

5A/Div.

70

Figure 3.38 shows the experimental efficiency curve of the assisting DAB converter and an

estimation of the efficiency of the conventional DAB converter. It can be seen that the assisting

DAB converter runs at higher power processing efficiency for the entire operating range

compared to the conventional DAB converter after utilizing phase shedding as explained in

section 3.4.4. Also, It can be seen that the converter runs at a peak efficiency of 95.4% with a

wide range for high efficiencies (the converter operates at a 90% and above for around a 70% of

the operating range and at an 80% and above for around an 85% of the operating range), which

is comparable or even better than the best state of the art solution [25] and better than many

commercially available boost converters [49]-[50]. It should be noted that the converter

efficiency can be further improved, especially at medium and heavy loads, by using switches

rated at the required blocking voltages which is not the case for the current implementation. In

case of using switches with the required blocking voltages, a peak efficiency of around 97% can

be achieved as shown in Fig.3.38.

Figure 3.38. Efficiency curves.

71

Table 3.1. Experimental setup components and parameters

Component Parameters

Primary side switches IRLR7833

Secondary side switches IRFS4228

Low-side gate drives FAN3122

High side gate drives FAN7371

ADC (for Vo measurements) AD9221

Op-Amp (for Vo sensing) LMH6628

Inductor 10 μH

Transformers 1:4 turns ratio, 23μH magnetizing

inductance, 0.5 μH primary side leakage

inductance

Output capacitor 240μF

Primary side snubber capacitors 47nF

Secondary side snubber capacitors 2.2nF

3.6 Conclusions A new system for providing battery balancing and step-up voltage functions for electromobility

applications is introduced. The architecture is based on the assisting DAB converter concept where

72

the converter is used only to provide the difference between the battery pack voltage and the

desired output. In comparison with conventional systems this architecture drastically reduces

power processing requirements of the step-up power stage and relaxes requirements regarding

converter power processing efficiency. An implementation of this concept based on four- phase

bi-directional DAB converter prototype is demonstrated. In comparison with a single step-up

stage, the assisting DAB converter provides more functions, utilizes smaller power stage, and runs

at a higher power processing efficiency. This architecture potentially offers solution for a cost-

effective implementation of balancing feature in automotive applications, where the use of

balancing system is relatively rare.

3.7 References

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for Power Electron., Intell. Motion, Renewable Energy, and Energy Manage., 2015, pp.863-

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characterization of a high-power dual active bridge dc-to-dc converter," IEEE Trans. Ind.

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[25] F. Krismer and J. W. Kolar, "Accurate power loss model derivation of a high-current dual active

bridge converter for an automotive application," IEEE Trans. Ind. Electron., vol.57, no.3,

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[26] R. T. Naayagi, A. J. Forsyth, and R. Shuttleworth, "Performance analysis of DAB DC-DC

converter under zero voltage switching," in Proc.1st Annu. Int. Conf. on Elect. Energy Syst.,

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[27] P. S. Shenoy, K. A. Kim, B. B. Johnson, and P. T. Krein, "Differential power processing for

increased energy production and reliability of photovoltaic systems," IEEE Trans. Power

Electron., vol.28, no.6, pp.2968-2979, June 2013.

[28] H. J. Bergveld, D. Buthker, C. Castello, T. Doorn, A. de Jong, R. van Otten, and K. de Waal,

"Module-level DC/DC conversion for photovoltaic systems: the delta-conversion concept,"

IEEE Trans. Power Electron., vol.28, no.4, pp.2005-2013, April 2013.

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cascaded and bypass grid connection topologies — Design and Optimisation," in Proc. IEEE

Power Electron. Spec. Conf., 2006, pp.1-7.

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[30] P. S. Shenoy, A. Kim, B. B. Johnson, and P. T. Krein, "Differential power processing for

increased energy production and reliability of photovoltaic systems," , IEEE Trans. Power

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[31] C. Olalla, C. Deline, D. Clement, Y. Levron, M. Rodriguez, and D. Maksimovic, "Performance

of power-limited differential power processing architectures in mismatched PV systems,"

IEEE Trans. Power Electron., vol.30, no.2, pp.618-631, Feb. 2015.

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architecture with isolated differential power conversion for data centers," IEEE Trans. Power

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Portilla, Y. Torroja, M. Vasic, S. C. Huerta, M. Trocki, P. Zumel, and J. A. Cobos, "Power

balance of a hybrid power Source in a power plant for a small propulsion aircraft," IEEE Trans.

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[34] "Installation guide for electric vehicle supply equipment (EVSE): an introduction to EVSE,"

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[35] A.V. Peterchev, X. Jinwen, and S.R. Sanders, "Architecture and IC implementation of a digital

VRM controller, "IEEE Trans. Power Electron., vol. 18, pp. 356–364, Jan 2003.

[36] "AN2170", Application note, STMicroelectronics Inc., available www.st.com, June 2005.

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77

[38] Monzer Al Sakka, Joeri Van Mierlo, and Hamid Gualous, "DC/DC converters for eletric

vehicles," Book chapter, available http://www.intechopen.com.

[39] F. Krismer, "Modeling and optimization of bidirectional dual active bridge dc-dc converter

topologies," Ph.D. Dissertation, ETH Zurich, Zurich, 2010.

[40] P. Zumel, C. Fernandez, A. de Castro, and O. Garcia, "Efficiency improvement in multiphase

converter by changing dynamically the number of phases," in Proc. IEEE Power

Electron. Spec. Conf., 2006, pp. 1-6.

[41] J. Wei and F. C. Lee, "Two-stage voltage regulator for laptop computer CPUs and the

corresponding advanced control schemes to improve light-load performance," in Proc.

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[42] M. H. Kheraluwala and R. L. Steigerwald, "Efficient, high power density, high power factor

converter for very low dc voltage applications," US Patent US5481449 A, Jan 1996.

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[45] "Surface mount multilayer ceramic capacitors, general purpose & high capacitance, class 2,

X5R, 4 V to 50 V," Datasheet, Yageo, 2014, available online www.yageo.com.

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78

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79

Chapter 4

High Power Density Assisting Step-Up Converter with

Integrated Battery Balancing Feature

This chapter introduces a new step-up power converter architecture for portable applications with

multi-cell battery packs that integrates battery cell balancing function. Compared to conventionally

used boost converter, which is not providing cell balancing, the new architecture has smaller

overall volume and approximately the same power processing efficiency. The step-up function is

80

obtained using the assisting concept, where the flyback output is placed at the top of the battery

pack and, therefore, is only processing a portion of the output power. As a result, high power

processing efficiency and small converter volume are achieved. The operation of the system is

regulated by a digital controller that provides the two functions at the same time.

4.1 Introduction A typical battery power management system of a portable application with multiple battery cells,

such as a tablet computer or eBook, is shown in Fig.4.1 [1]. It usually consists of two battery cells,

a front-end dc-dc converter creating a stable intermediate bus voltage, and multiple downstream

point-of-load (PoL) power supplies providing regulated voltages for functional blocks. The power

management systems also include relatively large step-up-converters, for providing voltages for

back panel lighting [2] and flash memory programming [3]. The boost converters provide up to

60% of the total power [4]-[5] and their reactive components take a significant portion of the

overall device volume. In the targeted applications, battery-balancing circuits [6]-[16] are rarely

Figure 4.1. Conventional battery power management system of a portable device.

81

used, even though they can extend the operating time up to three times [17].This is mostly due to

large additional cost and volume conventional balancing systems introduce in those low volume-

driven and cost-sensitive applications.

The main goal of this chapter is to introduce the new high power density step-up converter

architecture of Fig.4.2 that integrates the balancing feature and, at the same time, has a smaller

overall volume of the reactive components than the commonly used boost. As such the topology

potentially allows for a further minimization of the portable devices as well as an extension of the

battery operating time.

The advantages are achieved using a single two-input interleaved flyback [18] and the concept

named assisting conversion [19], where most of the energy for the load is provided directly from

the battery and the converter only assists the battery, by providing a portion of the output power.

The digital controller of Fig.4.2 simultaneously regulates the output converter voltage vout(t) and

the input currents of the flyback, iin1(t) and iin2(t). The input currents of the two phases are

Dc-dcbus

converter

+PoL

PoL

LDO

Loads

Cbus

Vout

+

_

+Vbatt

Vcell1

Vcell2

Vbus V1

V2

Vn

+

_

_

+

_

_Vfb

Cfb

S1

S2

S3

iin1

iin2

ifb_out

Assisting flyback

Digital Controller

H

Rsifbout

SOC

S1 to S3 cnt.

Figure 4.2. Assisting flyback based architecture.

82

determined based on the information about the states of charge (SOC) of the battery cells. The

input of the flyback connected to the cell with higher SOC takes more current, gradually

minimizing the SOC difference and, therefore, extending the battery pack operating time.

4.2 Principle of Operation In the architecture of Fig.4.2, named the assisting flyback architecture, the flyback inputs are

connected to the battery cells and the output of the converter, i.e. the output capacitor, is placed on

the top of the battery pack. Therefore, the converter only processes a portion of the power,

proportional to the difference between the output and the battery pack voltages. This reduction in

the power processing requirements allows a smaller converter to be used and results in power

processing efficiency improvements.

Various forms of partial power processing, to reduce volume and improve efficiency, have been

presented in [20]-[25]. In [20]-[24] converters processing difference in power photovoltaic (PV)

applications to maximize the power delivered by each cell. Also a converter for fully electric

airplane that process a portion of the output power has been presented in [25]. The main advantage

of the presented solution over the previously described work is that it integrates balancing and

voltage control in multi-input single-output converter designed for portable applications.

4.2.1 Assisting Conversion As explained earlier in chapter 3, the advantages of the assisting flyback can be explained by

comparing it to the conventional solution and noticing that the assisting flyback only processes

only a portion of the output power and hence a smaller volume and power processing efficiency

improvement can be achieved.

83

4.2.2 Cells Balancing In order to integrate the balancing feature and the voltage step-up inside a single converter the two-

input assisting flyback converter shown in Fig.4.2 is considered. The assisting converter is a bi-

directional converter and the balancing is done by controlling the amount of current out of each

cell or by transferring the energy between cells.

4.3 Practical Implementation Figure 4.3 shows a practical implementation of the assisting flyback from Fig.4.2. In this case,

each of the ideal switches of the inputs is replaced with a pair of transistors to prevent cross

conduction of the modules. The switch on the output side of the transformer is realized as a

transistor to minimize conduction losses. Figure 4.4 shows the gating sequence of the five

switches. Also, as it can be seen from the figure small input decoupling capacitors, Cin1and Cin2,

are added at the input of each flyback phase which is a requirement for the targeted application

[26]. These decoupling capacitors are added to avoid taking pulsating currents from the battery

cells as explained in the passive components comparison section. In addition, R-C-D snubber

circuit is used with each phase to dissipate the stored energy in the leakage inductance of each

transformer winding. Figures 4.5-4.8 show simulation results for operation of the converter under

balanced and unbalanced condition during heavy and light loads respectively. Also, Fig.4.9 shows

the transient response of the converter during light-to-heavy and heavy-to-light conditions.

84

Figure 4.3. Practical implementation of the assisting flyback and its digital controller.

Figure 4.4. Gating sequences for different scenarios.

85

Figure 4.5. Simulation results for operation of the converter under balanced condition and heavy

load.

Figure 4.6. Simulation results for operation of the converter under unbalanced condition and

heavy load.

86

Figure 4.7. Simulation results for operation of the converter under balanced condition and light

load.

Figure 4.8. Simulation results for operation of the converter under unbalanced condition and

light load.

87

Figure 4.9. Simulation results for transient response of the assisting flyback converter during

heavy-to-light and light-to-heavy loads.

4.3.1 Controller For medium and heavy loads, the flyback operates in continuous conduction mode and its

operation is regulated through voltage mode pulse width regulation. The attenuated output voltage

of the assisting flyback is converted into its digital equivalent HVout[n] with the analog-to-digital

converter ADC1 and compared to the desired reference value Vref[n]. The resulting error signal e[n]

is then passed to the PID compensator that creates control signal d[n], which is proportional to the

desired duty ratio value. This value is then sent to the power sharing logic block that, based on

the state of charge (SOC) information calculates duty ratio values for both flyback inputs, labeled

as d1[n] and d2[n], and passes them to a dual-output digital pulse width modulator developed in

[27]. The information about the state of the charge of the cells is obtained from an external

monitoring circuit. For the case when the cells are balanced the power sharing logic issues the

same duty ratios to both phases and the current is shared equally. When the batteries have different

SOCs the logic issues a larger duty ratio to the phase with higher SOC resulting in a larger current

88

taken from the corresponding cell. In both cases, the two phases of the flyback are operating in

interleaved mode minimizing output capacitor ripple. In the continuous conduction mode of the

flyback information about the current through the output side of the flyback transformer is only

used for protection.

At light loads, when the flyback operates in discontinuous conduction mode the previously

described voltage mode control method cannot be directly used. This is because, in this mode, the

currents of the phases are not always proportional to their duty ratios. In a realistic converter, a

significant difference between magnetizing inductances seen by the two phases could occur and,

consequently a large mismatch in their currents. To solve this problem, a control solution based

on a measurement of the output transformer current during Q3 conduction time can be utilized. In

this case, the obtained information about the current is combined with that about the SOC, to

determine the proper duty ratios for both flyback phases.

4.3.2 Comparison of passive components volume The passive components volume comparison is done by comparing the assisting flyback converter

to a conventional boost that has almost the same efficiency curve and has the same input and output

voltage variations. In order to make both converters run at almost the same efficiency curve, the

assisting flyback converter is switched at 500 KHz while the conventional boost is switched at 400

KHz as shown in the experimental results section. In the targeted application, the main inductor is

designed such that the percentage inductor current ripples at the lowest input voltage and the

maximum output current is less than 35% of the average inductor current [26]. Also, the output

voltage ripples are required to be less than 1% of the output voltage [26]. In addition, a 10µF low-

equivalent series resistance (ESR) ceramic capacitor that acts as a decoupling capacitor is placed

at the input of the power stage [26].

89

The boost inductor and flyback transformer volumes can be compared by looking at their energy

storage requirements, i.e. ½ LI2,

knowing that most of the

transformer volume comes from

the core volume [28] especially in

low-power applications [18]. In

comparison to conventional boost

the assisting flyback converter has

a 20% less magnetic components

volume when both converters are

designed to have a 20% inductor

current ripples. The decoupling

input capacitors are designed such

that the assisting flyback

converter has the same voltage

drop across the input compared to the conventional boost. The voltage drop across the input of the

assisting flyback converter is equal to the peak current through the input capacitor multiplied by

its equivalent ESR in addition to the voltage variations across Cin placed across each cell since the

Figure 4.10. Assisting flyback converter waveforms after

adding decoupling capacitors.

90

assisting flyback

converter operates in

interleaved fashion as

shown in Fig.4.10 and Fig.

4.11.a. The voltage drop

across the input of the

boost is equal to the

inductor current

multiplied by the total sum

of the input resistance of

each cell as shown in

Fig.4.11.b. When using

two 10 µF capacitors

across the input of each

cell the voltage drop

across the input of the

assisting flyback

converter is

approximately equal to

that of the conventional boost, knowing that the ESR of decoupling ceramic capacitors of 10 µF,

6.3V gets as low as 2.5mΩ at the frequency of the flyback current [29] and the input resistance of

the cells used in the targeted application is equal to approximately 10mΩ [30]. Using two 10 µF

capacitors across each flyback input results in having the same input capacitor volume compared

(a)

(b)

Figure 4.11. (a) Input voltage drop calculation for assisting

flyback converter, (b) input voltage drop calculation for

conventional boost converter.

91

to the conventional boost, knowing that capacitors volume is proportional to their energy storage

requirements ½ CV2.

The output capacitor of the assisting flyback converter is designed such that the voltage ripples

across the flying capacitor, Cfb, in addition to voltage variations across the input is equal to the

voltage ripples across the output of the conventional boost. Compared to the conventional boost,

the assisting flyback converter has a 56% less output capacitor volume. In total, the assisting

flyback converter has a 42% less total capacitive volume compared to the conventional boost.

Taking into account that capacitors are approximately 1000 times more energy dense than

inductors in low power applications [31]-[32], the assisting flyback converter has a 23% less

reactive components volume compared to the conventional boost.

4.3.3 Silicon Area Comparison The silicon area required to implement the assisting flyback and boost converters can be compared

by looking at the summation of the semiconductor stresses for each converter. The assisting

flyback converter has around 30% more silicon area than that of the conventional boost. The

increased silicon area comes as a result for integrating cells balancing with voltage control features

in a single converter topology. However it should be noted that most of the converter volume

comes from the passive components, especially inductors, not the semiconductor devices which

means it is favorable to trade off silicon area for passive components volume. Figure 4.12

summarizes passive components volume and silicon area comparisons. Figure 4.13 shows the

contribution of the circuits’ components to the overall volume of both converters. The assisting

92

flyback converter has around a 23% smaller volume compared to conventional boost as shown in

Fig.4.13.

It worth mentioning that the assisting flyback converter uses semiconductor devices with lower

blocking voltage than that of conventional boost which can add a cost benefit in case of on-chip

implementation. In addition, using multiple switches on the primary side gives an advantage of

better heat distribution without degrading the efficiency curve compared to conventional boost as

shown in the experimental results section.

Assisting flyback

Boost

Switches area

Output Cap

Input Cap

Magnetics

Figure 4.12. Normalized values for passive components and silicon area comparisons.

Figure 4.13. Circuits’ components contribution to the overall volume of both converters.

93

4.4 Experimental Results To verify the concept presented in previous sections, an 8-to-12 V, 20W experimental prototype

has been built and tested. In the targeted portable applications 12 V supply voltage is commonly

used for LCD displays [2] and erasable memories [3]. The assisting converter processes only a

45% of the total output power .

The power stage of the prototype is implemented with discrete components and the controller is

designed using an FPGA-based development board. The input side switches operate at switching

frequency of 250 KHz and the synchronous rectifier switch on the output transformer side operates

at 500 kHz. In this implemenation, the SOC information is assumed to be available from an

external circuit that operates based on cells terminal voltage. Table 4.1 shows circuit parameters

and values.

Figures 4.14 and 4.15 demonstrate the operation of the assisting flyback converter at heavy loads.

Figure 4.14 shows the flyback currents, ifb1 and ifb2, (Fig.4.3) and the output voltage for the case

when the two cells are balanced. It can be seen that the converter simultaneously provides equal

current sharing and tight output voltage regulation. Figure 4.15 demonstrates the operation when

the top cell is overcharged. It can be seen that ifb2 is larger than ifb1 and the output voltage is still

regulated at 12V.

Figures 4.16 and 4.17 demonstrate the operation of the assisting flyback converter at light loads.

Figure 4.16 shows the flyback currents and the output voltage when the two cells are balanced.

Figure 4.17 shows the flyback currents and the output voltage when the top cell is overcharged.

94

Figure 4.14. Assisting converter operation when the two cells are balanced at heavy load,

flyback currents, Vout is the output voltage measurement (10V/Div.). ifb1 is the bottom

module flyback current (2A/Div.). ifb2 is the top module flyback current (2A/Div.).

Figure 4.15. Assisting converter operation when the top cell is overcharged at heavy load,

flyback currents. Vout is the output voltage measurement (10V/Div.). ifb1 is the bottom


95

Figure 4.16. Assisting converter operation when the two cells are balanced at light load,

flyback currents. Vout is the output voltage measurement (10V/Div.). ifb1 is the bottom


Figure 4.17. Assisting converter operation when the top cell is overcharged at light

load, flyback currents. Vout is the output voltage measurement (10V/Div.). ifb1 is the

bottom module flyback current (2A/Div.). ifb2 is the top module flyback current

(2A/Div.).

96

The results of the processing efficiency measurements for the assisting flyback and the

conventional boost are shown in Fig.4.18. The conventional boost is controlled at a lower

switching frequency of 400 KHz to achieve almost the same efficiency curve of the assisting

flyback running at 500KHz at the secondary side. It should be noted that at 500 KHz the assisting

flyback converter achieves peak efficiency of a 93.4% and runs at a 90% efficiency and above for

a 90% of the operating range (from 2W to 20W).

Table.4.1. Experimental setup parameters.

Components Parameters

Mosfets IRF8788 Transformer 1:1:1 turns ratios and 4.7 μH magnetizing

inductance Output Cap 22μF Gate drives FAN3122

Op-Amp (for output voltage sensing) LMH6628 ADC (for output voltage measurements) AD9221

Figure 4.18. Power processing efficiency curves comparison.

83

85

87

89

91

93

95

0 5 10 15 20

EF

FIC

IEN

CY

%

Po (W)

Flyback

Boost

Flyback 500KHz

Boost 400KHz

97

4.5 Conclusions A new step-up power converter architecture for portable applications with multi-cell battery packs

that integrates battery cell balancing function is introduced. The architecture is based on the

assisting flyback converter concept where a low-power converter is used only to provide the

difference between the battery pack voltage and the desired output. The passive components

volume, silicon area, and power processing efficiency of the assisting flyback and conventional

boost designed for the targeted application are compared in this chapter. In comparison with the

conventional boost, the assisting flyback converter integrates the balancing feature, has a 22%

smaller reactive components volume while having approximately the same power processing

efficiency curve.

4.6 References

[1] "TI Tablet Solutions," Datasheet, Texas Instrument, 2013, [online]. Available:

http://www.ti.com.

[2] "TFT LCD Specification: Model Name: TD080WGCA1," Datasheet, Toppoly, [online].

Available: www.distrib-informatique.com.

[3] "MAX662A," Datasheet, Maxim, [online]. Available: http://www.maximintegrated.com.

[4] R. M. Soneira, "Tablet Display Technology Shoot-Out," [online]. Available:

http://www.displaymate.com/Tablet_ShootOut_2.htm.

[5] D. Schmidt, "Samsung Galaxy Tab S 10.5 Tablet Review," [online]. Available

http://www.notebookcheck.net/Samsung-Galaxy-Tab-S-10-5-Tablet

Review.124253.0.html.

98

[6] J. W. Kimball, B. T. Kuhn, and P. T. Krein, "Increased Performance of Battery Packs by

Active Equalization," in Proc. IEEE Vehicle Power and Propulsion Conf., 2007, pp.323-

327.

[7] J.-W. Shin, G.-S. Seo, C.-Y. Chun, and B.-H. Cho, "Selective flyback

balancing circuit with improved balancing speed for series connected Lithium-ion

batteries," in Proc. Int. Power Electron. Conf., 2010, pp.1180-1184.

[8] C. Bonfiglio and W. Roessler, "A cost optimized battery management system with active

cell balancing for lithium ion battery stacks," in Proc. IEEE Vehicle Power and Propulsion

Conf., 2009, pp.304-309.

[9] W. C. Lee, D. Drury, and P. Mellor, "Comparison of passive cell balancing and active cell

balancing for automotive batteries," in Proc. IEEE Vehicle Power and Propulsion Conf.,

2011, pp.1-7.

[10] X. Lu, W. Qian, and F. Z. Peng, "Modularized buck-boost + Cuk converter for high voltage

series connected battery cells,” in Proc. 27th Annu. IEEE Appl. Power Electron. Conf.

Expo., 2012, pp.2272-2278.

[11] C.-H. Kim, M.-Y. Kim, and G.-W. Moon, "A Modularized Charge Equalizer Using a

Battery Monitoring IC for Series-Connected Li-Ion Battery Strings in Electric Vehicles," ,

IEEE Trans. Power Electron., vol.28, no.8, pp.3779-3787, Aug. 2013.

[12] M.-Y. Kim, J.-H. Kim, and G.-W. Moon, "Center-Cell Concentration Structure of a Cell-

to-Cell Balancing Circuit with a Reduced Number of Switches," IEEE Trans. Power

Electron., vol.29, no.10, pp.5285-5297, Oct. 2014.

99

[13] F. Mestrallet., L. Kerachev, J-C. Crebier, and A. Collet, "Multiphase interleaved converter

for lithium battery active balancing,” in Proc. 27th Annu. IEEE Appl. Power Electron.

Conf. Expo., 2012, pp.369-376.




[15] T. Kim, W. Qiao, and L. Qu, "A series-connected self-reconfigurable multicell battery

capable of safe and effective charging/discharging and balancing operations," in Proc.

27th Annu. IEEE Appl. Power Electron. Conf. Expo., 2012, pp.2259-2264.

[16] J. Yun, T. Yeo, and J. Park, "High efficiency active cell balancing circuit with soft-

switching technique for series-connected battery string," in Proc. 28th Annu. IEEE Appl.

Power Electron. Conf. Expo., 2013, pp.3301-3304.

[17] P. T. Krein and R. S. Balog, "Life extension through charge equalization of lead-acid

batteries," in Proc. Telecommun. Energy Conf., 2002, pp.516-523.

[18] R. W. Erickson and D. Maksimović, "Fundamentals of Power Electronics", Second

Edition, New York, Springer Science+Business Media, 2001.


US8779700, July 2014.

[20] L. Linares, R.W. Erickson, S. MacAlpine, M. Brandemuehl, "Improved Energy Capture in

Series String Photovoltaics via Smart Distributed Power Electronics," in Proc. 24th Annu.

IEEE Appl. Power Electron. Conf. and Expo., pp.904-910, 2009.

[21] H. J. Bergveld, D. Buthker, C. Castello, T. Doorn, A. de Jong, R. van Otten, and K. de

Waal, "Module-Level DC/DC Conversion for Photovoltaic Systems: The Delta-

100

Conversion Concept," IEEE Trans. Power Electron., vol.28, no.4, pp.2005-2013, April

2013.

[22] G. R. Walker and J. C. Pierce, "PhotoVoltaic DC-DC Module Integrated Converter for

Novel Cascaded and Bypass Grid Connection Topologies — Design and Optimisation," in

Proc. IEEE Power Electron. Spec. Conf., 2006, pp.1-7.

[23] P. S. Shenoy, A. Kim, B. B. Johnson, and P. T. Krein, "Differential Power Processing for

Increased Energy Production and Reliability of Photovoltaic Systems," , IEEE Trans.

Power Electron., vol.28, no.6, pp.2968-2979, June 2013.

[24] C. Olalla, C. Deline, D. Clement, Y. Levron, M. Rodriguez, and D. Maksimovic,

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Mismatched PV Systems," IEEE Trans. Power Electron., vol.30, no.2, pp.618-631, Feb.

2015.

[25] E. Bataller-Planes, N. Lapea-Rey, J. Mosquera, F. Orti, J. A. Oliver, O. Garcia, F. Moreno,

J. Portilla, Y. Torroja, M. Vasic, S. C. Huerta, M. Trocki, P. Zumel, and J. A. Cobos,

"Power balance of a hybrid power Source in a power plant for a small propulsion

aircraft," IEEE Trans. Power Electron., vol.24, no.12, pp.2856,2866, Dec. 2009.

[26] "TPS61085 650-kHz, 1.2-MHz, 18.5-V Step-Up DC-DC Converter," Datasheet, Texas

Instrument, 2014, [online]. Available: www.ti.com.

[27] Z. Lukic, C. Blake, S. C. Huerta, and A. Prodic, "Universal and Fault-Tolerant Multiphase

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IEEE Appl. Power Electron. Conf. Expo., 2007, pp.42-47.

[28] Brandon Grainger, Oreste Scioscia, and Gregory Reed, "Bringing down the Volume,"

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101

[29] "Surface Mount Multilayer Ceramic Capacitors, General Purpose & High Capacitance,

Class 2, X5R, 4 V to 50 V," Datasheet, Yageo, 2014, [online]. Available: www.yageo.com.

[30] "Polymer Lithium-ion Battery Product Specification, Model：PL-7035130-10C,"

Datasheet, [online]. Available: www.batteryspace.com.

[31] "MetacapacitorsTM Next-generation power electronics for LED lighting and other

applications," [online]. Available:

http://energy.cuny.edu/energy/downloads/metacapacitors/MetacapacitorsExecutiveSumm

ary.pdf.

[32] M. D. Seeman, "A Design Methodology for Switched-Capacitor DC-DC Converters,"

Ph.D. Dissertation, Dept. Elect. Eng. and Comput. Science, Univ. of California, Berkley,

2009.

102

Chapter 5

Other Possible Applications for Assisting Converters

This chapter investigates using the assisting converter architecture in different applications other

than automotive and portable electronics systems. Three possible applications are investigated in

this chapter which are; uninterruptible power supplies (UPS) and data centers, low-to-medium

scale grid storage systems and smart homes, and finally datacom and wireless applications.

103

Investigating these systems leads to the conclusion that the assisting converter architecture is a

potent candidate that can be utilized in these systems.

5.1 Introduction The advantages of small volume and high power processing efficiency in addition to combining

cells balancing and voltage control functions can be beneficial to different battery powered

applications besides the two applications covered in this dissertation. The following sections

suggest utilizing the assisting converter architecture in data centers, low-to-medium scale grid

storage systems and smart homes, and datacom and wireless communication systems.

5.2 Uninterruptible Power Supplies and Data Centers Recently, the scale of data centers increases vastly especially after the internet traffic has rapidly

increased. The most two important requirements for data centers, and UPS systems as well, are

small volume and high power processing efficiency [1]-[11] which opens a potent possibility for

the assisting converters to be used in these systems.

These systems have three possible architectures which are ac, low-voltage dc, and high voltage dc

architectures [12]. Ac architecture based data centers consist of an ac-dc rectifier to convert the

utility voltage to a common dc bus voltage, dc-ac inverter to convert the voltage of the common

dc bus back to ac voltage that feeds different information and communication technology (ICT)

devices, a backup battery pack connected to the common dc bus using a step-up converter, and a

battery charger as shown in Fig.5.1. The main disadvantages of this system are the multiple

successive conversions stages which limit system efficiency and reduce reliability [12] in addition

to the need for having an ac-dc converter inside each ICT device or a front end ac-dc converter for

104

multiple ICT devices. Low voltage dc architecture based data centers consist of an ac-dc rectifier,

a dc-dc converter inside each ICT device, a battery pack connected to the dc bus using a step-up

dc-dc converter, and a battery charger as shown in Fig.5.2. This system avoids the problems

associated with the ac system but still have some demerits which are the relatively high currents

associated with the used low voltage, large copper area, and higher voltage drop [12]. In order to

solve the problems of the previously mentioned systems, dc based data centers that operate at

relatively higher voltage are introduced. It should be noted that low voltage dc based data centers

have the same architecture of the higher voltage ones but they operate at different voltage levels

as the latter ones operate at 300-400V and the low voltage data centres operate at 48V [12].

Ac-dcRectifier

Utility

Step-up dc-dc

converter

Dc-acInverter

Ac-dcRectifier

Dc-dcconverter

load

ICT device1

ICT device2

ICT devicen

Charger

Battery pack

Figure 5.1. Ac system based data center.

105

Since these systems have a battery pack that needs cells balancing to prolong battery lifetime and

a step-up dc-dc converter; the assisting converter architecture can be used to provide these two

functionalities in a single power stage that is smaller in volume and higher in power processing

efficiency than those of the conventional solution.

5.3 Low-to-Medium Scale Grid Storage Systems and Smart Homes Grid storage systems and smart homes are introduced to modernize and optimize the operation of

the electric utility systems by providing more communication and power exchange between

different power sources such as batteries, flywheel systems, and supercapacitors. In recent years,

the electricity generated by storage systems has significantly increased in US to reach around 24.6

GW which forms a 2.3% of the total electricity production capacity while Japan and Europe have

Ac-dcRectifier

Utility

Step-up dc-dc

converter

Dc-dcconverter

load

ICT device1

ICT device2

ICT devicen

Charger

Battery pack

48VDC or 300-400VDC

Figure 5.2. Dc system based data center.

106

significantly higher percentage of grid storage [13]. The US department of energy has

recommended reducing the cost of the grid storage system to compete with other technologies

providing similar services [13]. Batteries are considered one of the most important energy storage

components and their power management systems still have an overhead to improve. Using the

assisting converter architecture leads to prolonging battery lifetime and reducing battery power

management system cost significantly.

Low-to-medium scale grid storage systems and smart homes have different power sources such as

batteries, super capacitor, flywheel, wind farms, PV panels, in addition to the grid power [13]. All

of these power sources are connected using power electronics converters such as ac-dc, dc-dc, and

dc-ac converters as shown in Fig.5.3. It can be seen that batteries are connected to the common

300-400Vdc bus using a bidirectional dc-dc converter to provide the voltage control and power

exchange between the battery and different sources. The assisting converter can be used to replace

Ac-dcRectifier

Utility

Bi-directional dc-dc converter

Battery pack

300-400V common dc bus Step-updc-dc

Converter

Ac-dcrectifier

PV PanelsStep-up dc-dc

converter

Supercapacitors

Ac-dcRectifier

FlywheelAc-dc

RectifierDc-ac

Inverter

Wind Farm

Figure 5.3. Low-to-medium scale grid storage system and smart home architecture.

107

this converter and provide an additional important feature which is cells balancing with a smaller

and more efficient power stage. Also, the step-up dc-dc converter interfacing the PV panels to the

dc-ac inverter in addition to the step-up dc-dc converter interfacing the supercapacitors to the

common dc bus can be replaced by a multi-phase or single phase assisting converters that has

smaller volume and better efficiency than the conventional solution and may provide maximum

power point tracking feature as well [14]-[16].

5.4 Datacom and Wireless Communication systems Most of the GSM and 3G communication devices are designed to operate in dc systems with a

nominal voltage of -48V [17]. These systems are powered up from batteries with a voltage

variation from 18-36V [17]. A step-up dc-dc converter is needed to provide a regulated dc voltage

for the communication devices and these converters could have a power range changing from

200W to 6KW [18]-[19]. The assisting converter can be used in these systems to provide voltage

control and cells balancing functionalities for these systems.

Step-updc-dc

Converter18-36V

Charger

Load

Figure 5.4. Communication systems power architecture.

108

5.5 Conclusions Three possible applications that can benefit from the assisting converter architecture are

investigated in this chapter. These applications are uninterruptible power supplies (UPS) and data

centers, low-to-medium scale grid storage systems and smart homes, and finally datacom and

wireless applications. Studying these systems leads to the conclusion that the assisting converter

architecture is a strong candidate that can be used in these systems.

5.6 References

[1] R. Simanjorang, H. Yamaguchi, H. Ohashi, T. Takeda, M. Yamazaki, and H. Murai, "A

High Output Power Density 400/400V Isolated DC/DC Converter with Hybrid Pair of SJ-

MOSFET and SiC-SBD for Power Supply of Data Center," in Proc. 25th Annu. IEEE Appl.


[2] R. Simanjorang, H. Yamaguchi, H. Ohashi, T. Takeda, H. Murai, and M. Yamasaki,

"Estimating performance of high output power density 400/400V isolated DC/DC

converter with hybrid pair SJ-MOSFET and SiC-SBD for power supply of data center," in

Proc. 31st IEEE Int. Telecommun. Energy Conf. 2009, pp.1-5.

[3] M. Noritake, T. Ushirokawa, K. Hirose, and M. Mino, "Verification of 380 Vdc

distribution system availability based on demonstration tests," in Proc. 33rd Annu.

IEEE Int. Telecommun. Energy Conf., 2011, pp.1-6.

[4] S. Abe, Y. Ishizuka, T. Ninomiya, Y. Sihun, M.Shoyama, and M. Kaga, "Prototype

Evaluation of Over 10W/cm3 Hgh Power Density Converter for 400V-DC Power

Distribution System in Data Center," in Proc. 10th Annu. IEEE Power Electron. and Drive

Syst., 2013, pp.1280-1284.

109

[5] K. Takao, H. Irokawa, Y. Hayashi, and H. Ohashi, "Novel Exact Power Loss Design

Method for High Output Power Density Converter," in Proc. 37th Annu. IEEE Power

Electron. Spec. Conf., 2006, pp.1-5.

[6] B. Subramaniam and W-C. Feng, "Enabling Efficient Power Provisioning for Enterprise

Applications," in Proc. 14th Annu. IEEE/ACM Cluster, Cloud and Grid Computing, 2014,

pp.71-80.

[7] R. R. Schmidt, E. E. Cruz, and M. Iyengar, "Challenges of Data Center Thermal

Management," IBM Journal of Research and Development, vol.49, no.4.5, pp.709-723,

July 2005.

[8] Y. Hayashi, K. Takao, T. Shimizu, and H. Ohashi, "Power Loss Design Platform for High

Output Power Density Converters," in Proc. European Power Electron. and Applicat.

Conf., 2007, pp.1-10.

[9] Y. Hayashi, K. Takao, T. Shimizu, and H. Ohashi, "High Power Density Design

Methodology," in Proc. Power Conversion Conf., 2007, pp.569-574.

[10] Y. Hayashi, "High-Power-Density Versatile DC-DC Converter for Environmentally

Friendly Data Centre," in Proc. 15th Annu. Power Electron. and Motion Control Conf.,

2012, pp.DS3b.16-1-DS3b.16-7.

[11] J. Biela, U. Badstuebner, and J. W. Kolar, "Design of a 5-kW, 1-U, 10-kW/dm3 Resonant

DC–DC Converter for Telecom Applications," IEEE Trans. Power Electron., vol.24, no.7,

pp.1701-1710, July 2009.

[12] K. Hirose, "DC Powered Data Centers in the World,” NTT Facilities Presentation, 2011,

[online]. Available: http://ze.bot.free.fr/NTT_DC_Datacenter.pdf.

110

[13] , "Grid Energy Storage," US Department of Energy, Dec.2013, [online].

Available:http://energy.gov/sites/prod/files/2013/12/f5/Grid%20Energy%20Storage%20

December%202013.pdf.

[14] N. Chaintreuil, F. Barruel, X. Le Pivert, H. Buttin, and J. Merten, "Effects of Shadow on a

Grid-Connected PV System," in Proc. 23rd European Photovoltaic Solar Energy Conf.,

2008, pp. 3417-3421.

[15] W. Herrmann, W. Wiesner, and W. VaaBen, "Hot Spot Investigations on PV Modules -

New Concepts for a Test Standard and Consequences for Module Design with Respect to

Bypass Diodes," in Proc. 26th Annu. IEEE Photovoltaic Spec. Conf., 1997, pp. 1129-1132.

[16] H. J. Bergveld, D. Buthker, C. Castello, T. Doorn, A. de Jong, R. van Otten, and K. de

Waal, "Module-Level DC/DC Conversion for Photovoltaic Systems: The Delta-

Conversion Concept," IEEE Trans. Power Electron., vol.28, no.4, pp.2005-2013, April

2013.

[17] D. Cooper, "Wide Input Voltage Range in DC Systems," August 2006, [online]. Available:

http://www.psma.com/sites/default/files.

[18] , "12-V and 24-V Input DC-DC Converters,” [online]. Available:

http://www.wilmoreelectronics.com/Prodlit/1675a.pdf

[19] , "24VDC 48VDC Power Systems for Datacom,

Wireless & Communication Power, DC to DC

Converters, Datacom Inverters from Newmar Telecom," [online]. Available:

http://sbcdatapower.com/Newmar-Telecom-DC-Power-48VDC-24VDC-Datacom-

Communication-Wireless-DC-Converters-Inverters-Battery-Chargers.html.

111

Chapter 6

Conclusions and Future Work

This Thesis presents a new and cost-effective battery management architecture for two major

applications, namely automotive and portable electronics applications. This battery management

architecture integrates balancing circuit and the step-up converter in a single stage topology.

Moreover, the converter processes only a portion of the output power and hence smaller volume

112

and better power processing efficiency can be achieved. In the following sections, the thesis main

contributions and possible extension of this work are discussed.

6.1 Conclusions The assisting dual active bridge converter [1]-[2] introduced in chapter 3 integrates cells balancing

and voltage control functionalities in a single power stage offering a cost-effective implementation

of the balancing feature for automotive applications. Also, the introduced topology has a smaller

volume and better power processing efficiency, i.e. higher power density, than conventional

solution not incorporating the balancing feature. Improving power density is achieved by making

the converter processes only a portion of the output power, unlike the existing solutions. Cells

balancing can be done during motoring, charging, and standstill modes of operation. During

motoring mode, the balancing is achieved by controlling the amount of power, and hence the

current, out of each cell while the balancing during charging and standstill modes of operation is

done by transferring energy between battery cells. Two balancing algorithms are introduced,

namely one-by-one cells balancing and simultaneous cells balancing. Phase shedding is utilized to

improve light-to-medium load efficiencies and to aid cells balancing as well. The converter runs

at a peak efficiency of a 95.4% and operates at an efficiency of a 90% and above for a 70% of the

operating range which is comparable or even better than the best of the state-of-art solutions [3]

and many commercially available boost converters. The operation of the converter is regulated by

an efficient digital controller that provides the cells balancing and voltage control functionalities.

The effectiveness of the assisting DAB converter to provide these functions and the high efficiency

range operation are verified experimentally in chapter 3.

113

The assisting flyback converter [4] introduced in chapter 4 utilizes the same concept and integrates

voltage control and balancing functions in a single converter designed for portable electronics

applications such as tablets and smartphones. While providing these functionalities, the assisting

flyback converter has about 23% smaller overall volume compared to the conventional boost. The

assisting flyback converter runs at a peak efficiency of a 93.4% with an efficiency of a 90% and

above for a 90% of the operating range which is comparable to the state of the art solutions.

The effectiveness of the assisting flyback converter to provide these functionalities and the high

efficiency range operation are demonstrated experimentally in chapter 4.

6.2 Future Work The state of charge (SOC) information is assumed to be available from an external circuit operates

based on cells terminal voltage in the practical implementation of the assisting DAB and flyback

converters. SOC estimation techniques and the challenges facing those techniques [5] can be an

interesting topic for future research.

The assisting DAB converter can be controlled to estimate (SOC) and state of health (SOH) of

battery cells using electrochemical impedance spectroscopy (EIS) [6]. That can be done by

injecting variable frequency small amplitude sinusoidal currents to each cell and measure the

voltage across cells terminals. These variable frequency small amplitude sinusoidal currents can

be obtained by introducing a variable frequency perturbation signal to the digital reference number

shown in Fig.3.5.

The IC implementation that integrates the power stage switches and controller of the assisting

DAB and flyback converters is another potential subject of future research. The integrated version

of the two topologies can benefit from the reduced die area for the switches and the reduced effect

114

of parasitic elements which has a negative impact on electromagnetic interference (EMI) and

associated parasitic losses. Also, further efficiency improvement can be achieved in the integrated

version by utilizing integrated switches with needed blocking voltage which is hard to achieve

with discrete implementation.

6.3 References


US8779700, July 2014.

[2] M. Shousha, Z. Gong, A. Prodić, V. Martin, and J. Milios "Assisting converter based

integrated battery management system for automotive applications," in Proc. Int.

exhibition and Conf. for Power Electron., Intell. Motion, Renewable Energy, and Energy

Manage., 2015, pp.863-870.

[3] F. Krismer and J. W. Kolar, "Accurate power loss model derivation of a high-current dual

active bridge converter for an automotive application," IEEE Trans. Ind. Electron., vol.57,

no.3, pp.881-891, March 2010.

[4] M. Shousha, T. McRae, A. Prodic, and V. Marten, "Assisting converter based integrated

battery management system for low power applications," in Proc. 29th Annu. IEEE Appl.


[5] V. Pop, H.J. Bergveld, P.H.L. Notten, and P.P.L. Regtien, "State-of-charge indication in

portable applications," in Proc. Int. Symp. on Ind. Electron., 2005, pp.1007-1012.

[6] A. Cuadras and O. Kanoun, "SoC Li-ion battery monitoring with impedance spectroscopy,"

in Proc. Int. multi-Conf. on Syst., Signals and Devices, 2009, pp.1-5.

115

Appendix A

Calculation of Power Transfer Equations for Square Wave Ac

Voltages

The average power delivered by the input port of Fig.3.6 (i.e. battery cell) can be calculated

through integration of the DAB voltage and current waveforms over a Tsw/2, where the waveforms

are shown in Fig.1A:

116

Figure A.1. DAB main waveforms.

dttitvndttPT

P Lsw

k

TT swsw

)(*)(∫)(1

10

10

dttiT

Vndttitvn

TL

sw

cellL

sw

TT swsw

)(2

)(*)(2 2

01

2

0 , Eq.A.1

The current waveform over the two distinctive time intervals of Fig.A.1 can be described with

the following equations:

117

L

tVnViti cfcellLL 0)( for 0 < t < tφ , Eq.A.2

L

ttVnViti cfcellLL

)( for tφ < t < Tsw/2, Eq.A.3

The initial conditions of Eq.A.2 and Eq.A.3, i.e. expressions for iL0 and iL, are obtained by

substituting t with tφ in Eq.A.2 and with Tsw/2 in Eq.A.3, and using the half cycle symmetry (i.e.

iL0= -iL (Tsw/2)). As a result the following expressions are obtained:

)2

-2(2

-)2

-(2

0Tt

L

VT

L

Vni

swcfswcellL , Eq.A.4

)2

(2

)22

-(2

T

L

VtT

L

Vni

swcfswcellL , Eq.A.5

By substituting Eq.A.2 to Eq.A.5 into Eq.A.1 and taking into account that f

tsw

k

2 , the

following power transfer equation is obtained:

L

VVn

Lf

VVnP

kkcfcell

sw

kkcfcellk

|)|-(

2

|)|-(2

. Eq.A.6

118

Appendix B

Cost Analysis and Suitability for a Large Number-of-Cells

System

In this appendix, design of the introduced architecture for a 93 Li-Ion cells, 400V for 35 KW

automotive applications [1] is given and cost analysis is provided as well. In this case, 93 primary

side modules are used and each four primary side modules share one secondary side module to

119

form what so called a block except for the last group of primary side modules. Each module is

rated at 100W such that the system is able to supply the rated output power at the desired output

voltage when all battery cells are close to their cut-off voltage. The following figures show the

power losses for each component, power processing efficiency of the converter, and system’s

cost. The cost is calculated for mass quantities (>5000 pieces) using standard suppliers websites

[2]-[3]. It should be noted that system’s cost is comparable to that of the conventional boost only

[4]-[5]. The reasons are: first, the converter processes only a portion of the output power and

hence can be built with less expensive components. Second, the used primary side switches are

low-voltage inexpensive switches used frequently in voltage regulator modules (VRMs) unlike

the conventional solution.

Figure B.1. Losses per primary side switch.

Figure B.2. Losses per secondary side switch.

120

Figure B.3. Losses per inductor.

Figure B.4. Losses per transformer.

Figure B.5. Losses per output capacitor.

Figure B.6. System losses.

121

Figure B.7. System’s efficiency.

Figure B.8. System’s cost.

97.5%

122

B.1 References

[1] "Innovative electric drive," [online]. Available: http://www.thesmart.ca/ca/en/index/smart-

fortwo-electric-drive/drive.html.

[2] www.digikey.ca.

[3] www.mouser.com.

[4] "Request for Proposal: The 2013 International Future Energy Challenge (IFEC’13),"

[online].Available, http://www.ieee-pels.org/about-pels/governing

documents/doc_download/332-2013-international-future-energy-challenge-request-for-

proposals-v-2.

[5] Discussion with industry people.

Documents

HIGH-E B M S · 2016-07-19 · would like to thank my friends, Hazem Soliman, Ahmed Mohamed, Mohamed Ramadan, Essam El Sahwi, and Sherif Helmy. My deepest gratitude goes to my parents,