Double

2504 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL 42, NO 12, DECEMBER 1994

A Monolithic 250 GHz Schottky-Diode Receiver Steven S. Gearhart and Gabriel M. Rebeiz, Senior Member, IEEE

Abstract-A 250 GHz monolithic Schottky-diode receiver based on a double-slot antenna is presented. The double-slot antenna is placed on an extended hemispherical high-resistivity silicon substrate lens. The measured DSB conversion loss and noise temperature at 258 GHz are 7.8 & 0.3 dB and 1600 f l00K for the antenna-mixer, respectively. A nonoptimal polyethylene Xd/4 matching-cap layer for the silicon lens improves the conversion loss and noise temperature by 1 dB, and another 0.7 dB improvement could be obtained with the use of a more optimal matching cap layer. The uniplanar double-slot antenna receiver is less than 0.3 x 1 mm in size including the IF filter and represents the first fully monolithic 250 GHz receiver to date. The measured performance is within 2-3 dB of the best 200' GHz waveguide receivers using planar Schottky diodes.'

I. INTRODUCTLON

"GRATED-CIRCUIT receivers consisting of a planar I antenna integrated with a matching network and a planar Schottky-diode or a three-terminal device offer many ad- vantages over waveguide-based receivers at millimeter-wave frequencies. They are smaller, lighter, and less expensive to build than waveguide systems and can be easily produced in. large numbers for millimeter-wave applications. A potential candidate for excellent millimeter-wave performance is the double-slot antenna [2]-[4]. In this work, a planar Schottky diode and a CPW-line matching network are integrated with a double-slot antenna on GaAs to form a fully monolithic Schottky-diode receiver. The monolithic integration should result in minimum parasitic capacitance and series resistance. After fabrication, the double-slot receiver is placed on an extended hemispherical high-resistivity silicon substrate lens to result in high-gain patterns with high Gaussian coupling efficiency [5] . This design requires no via holes or a backing ground-plane. The GaAs substrate is therefore not thinned down to 100 pm (or less) thereby increasing the yield of the fabrication process. The application areas for this receiver are in millimeter-wave imaging arrays for remote-sensing and radio-astronomical systems.

Manuscript received April I , 1994; revised June 20, 1994. This work was supported by the NASA Center for Space Terahertz Technology at the University of Michigan.

S. S. Gearhart was with the Electrical Engineering and Computer Science Department, University of Michigan, Ann Arbor, MI 48109-2122 USA. He is now with the Department of Electrical and Computer Engineering, University of Wisconsin, Madison, WI 53706-1691 USA.

G. M. Rebeiz is with the Electrical Engineering and Computer Science Department, University of Michigan, Ann Arbor, MI 48109-2122 USA.

IEEE Log Number 9405400.

'Duroid is a trademark of the Rogers Corporation, Chandler, AZ 85226 USA.

Antennas I

190 p Polveth

Monolithic Double-slot

Receiver

630 CaAs pm-thick wafer

570p"hick high- p Si wafer

Ma

GEARHAUT AND REBEIZ A MONOLITHIC 250 GHz SCHOTTKY-DIODE RECEIVER 2505

L/R

40 0.15 0.28 (1.29 0.36 0.44

100

35

30

25

eo .-- ,- __ 15 1 , , , , , , , , ~ H ~ e r l / ~ l @ ; t ~ ; c ~ , P ~ l t l o 4 4

6 0 I_ILL 10

Extension Length (pm)

Fig. 2. Antenna pattern Gaussicily and directivity versus extension length. The extension length is defined as the distance from the planar surface of the antenna to the hemispherical plane of the silicon lens.

inon 1400 1800 2200 2600 3000

n

-5

h

m a - -10 C m * -15 .-

0

4 .I

2 -20 D:

-25

-30 -60 -40 -20 0 20 40 60

Angle (degrees) Measured E-, H-, and 45"-plane patterns at 258 GHz on a 13.7-mm Fig. 3.

extended hemispherical lens.

close enough that no substrate modes and associated power loss occur when a GaAs wafer is placed on a silicon lens. The power radiated to the backside is minimal, only 9% (0.4 dB), and therefore no backing cavity is used to recover this power loss. In this work, the double-slot receiver is centered on a 13.7 mm diameter extended hemispherical silicon lens. An extension length L, defined as the distance from the planar surface of the antenna to the hemispherical plane of the silicon lens, of 2300 pm ( L / R = 0.34) is chosen to yield an antenna pattern with high Gaussicity and high directivity [5] (Fig. 2). The resulting radiation pattern has a directivity of 29 dB and a 90% Gaussian-coupling efficiency at 250 GHz. The Gaussian- coupling efficiency does not include power loss to the backside (0.4 dB) or the reflection loss at the silicon-air interface. The reflection loss is calculated to be 1.7 dB for no matching-cap layer on the silicon lens and 0.2 dB for a X,/4 matching-cap layer with a dielectric constant equal to JElens and uniform thickness [5].

The integrated Schottky diode forms the center conductor of the CPW line and is placed in series between the slot antennas, resulting in a sum-mode antenna pattern (Fig. 3). The 0.30&-long slot antennas (at 250 GHz) are 15 pm wide

Fig. 4. of 2.5 p m is due to the gold (1.0 /Am) and n+-GaAs (1.5 /Am).

Coplanar Waveguide (CPW) transmission line. The metal thickness

(b) (C)

Fig. 5. Receiver matching network. (a) The two antennas are represented by a Thevenin equivalent circuit Vant, Zant. (b) The antenna impedance is transferred across the CPW-transmission line to Z1. (c) The RF embedding impedance 221.

and are near the second resonance region of a slot antenna on a silicon or GaAs half-space [9]. The input impedance of the double-slot antenna is the sum of the slot self-impedance (211) and the double-slot mutual impedance (212) since the two slots are fed in phase. The self and mutual impedances of a slot antenna on a semiinfinite dielectric have been calculated recently in [3], [9]. Using the program of Eleftheriades at the design frequency of 250 GHz, Zll = 25.5 + j0 .4 R, [9]. The RF matching network consists of two short sections (90 pm) of 35 R CPW-line. The 35 R CPW-line dimensions are s = 18 pm and w = 6 pm at the antenna feed point with a metal thickness (gold and n+-layer) of 2.5 pm (Fig. 4). The CPW line widens to s = 30 pm and w = 10 pm near the diode, and the impedance remains 35 R on the CPW line. For the 90 pm-long CPW line, X,E is found using EEsoff Linecalc [lo] to be 490 pm at 250 GHz, and therefore, the line has an electrical length of 66". The maximum total width of the CPW-line (s + 2w = 50 pm) may allow some loss to radiation [ l l ] , but this width is necessary to accomodate the ohmic contact of the planar Schottky diode. The antenna input impedance Zant is transformed across the 90 pm-long CPW line to 21. The diode embedding impedance, defined as the impedance seen at the diode terminals, is 221, since the diode is in series with the CPW-line (Fig. 5). At 250 GHz, 21 = 29.3 + j10.6 f? and 221 = 59 + j 2 l R. Since the slots have a wideband input impedance in this region and the matching networks are short, the diode embedding impedance 221 is also relatively wideband. The antenna input impedance Z;, and diode embedding impedance 221 are displayed over a 20% bandwidth from 225 to 275 GHz in Fig. 6.

2 1 2 = 2.4 - j9 .7 R, and Zant = 211 + 2 1 2 = 27.9 - j9 .3 0

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2506 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES. VOL. 42, NO. 12, DECEMBER I994

Fig. 6. Theoretical impedance design over a f 10% bandwidth: (a) antenna input impedance ZZi,,,, (b) the RF embedding impedance 2 2 , as seen by the diode.

The theoretical receiver performance was analyzed using the reflection algorithm of [ 121. The Schottky-diode parameters are assumed to be as follows: a reverse saturation current of I , = 1 x A, a diode ideality factor of 7 = 1.15, a junction capacitance of C,, = 3 E, a diode barrier height of Vb, = 0.76 V and a series resistance of R, = 15 R. A parasitic capacitance of C, = 2 fF is included in parallel with the embedding impedance 221 to model the parasitic capacitance at the diode fingertip. These diode parameters are typical of a good monolithic 1.2 pm-anode Schottky diode with an etched surface channel. By varying the RF embedding impedance, it is determined that minimum conversion loss occurs with an RF embedding impedance of approximately 60 + j50 R, and a minimum noise temperature occurs for an RF embedding impedance of 40 + j40 R. For both cases the available LO power is 2 .g2.5 mW. The effects of the embedding impedances at the 2nd, 3rd, and 4th harmonics of the RF and LO were tested by making them open circuits, short circuits, and reactive loads in the analysis. While varying the higher order harmonic embedding impedances caused the conversion loss and noise temperature to change by 60.5 dB, the regions of minimum conversion loss and noise temperature remained constant. However, due to the wide CPW lines which would tend to radiate at the higher harmonics (500 GHz, 750 GHz), the embedding impedances at these harmonics are set to 50 R for the remainder of the analysis. The diode IF impedance is typically 100 to 120 R. For the designed embedding impedance of 221 = 59 + 321 R at 250 GHz with an available LO power of 2.5 mW, the diode RF impedance is 71 - j 4 5 R, the diode LO impedance is 57 - j 5 3 0, and the IF impedance is 105 52. The corresponding diode bias is Vb = 500 mV and Ib = 3.5 mA. This results in a theoretical LO reflection loss of 0.8 dB, an SSB conversion loss of 6.2 dB, and an SSB noise temperature of 830 K. The receiver design is wideband: over the 20% bandwidth from 225 GHz to 275 GHz, the theoretical SSB conversion loss is less than 7.2 dB and the SSB noise temperature is less than 1500 K.

The CPW line is short-circuited to the ground-plane at the left slot antenna, providing the dc return for biasing the diode. On the right slot-antenna, the CPW line is connected to a low- pass IF filter. The IF network cons i s t s of a 4-sect ion low-pass

/ 2500 .~ SIN,

Schottky metal TiiPtlAu

Diode finger ,,

. . . . . . . . . . . , . . .

(e)

Fig. 7. Integrated receiver fabrication process. (a) PECVD Si,N, growth on the epi-doped GaAs wafer. (b) Schottky anode well definition. (c) Ohmic contact etch, evaporation, and anneal. (d) Diode finger definition and Schottky metal evaporation. (e) Isolation etch.

CPW filter with a 3-dB corner frequency of 150 GHz and a short-circuit rejection of -13 dB from 220 to 280 GHz. A six-section filter with a rejection of -20 dB could have been included, but the 4-section filter was chosen to conserve space on the GaAs wafer. The IF filter is followed by a X/4 CPW matching network at 1.4 GHz on a low-loss Duroid 6006 (6,. = 6.15) substrate *"(Rogers Corp., Chandler, AZ 85226) with an impedance of 75 R to match the 110 R IF impedance of the LO pumped diode. to the 50 R impedance of the IF amplifier.

111. FABRICATION

The receiver was fabricated on a 525 iim-thick GaAs at the University of Michigan. An n- doping density of 3 x lo1' cmP3 and an anode diameter of 1.2 pm is chosen for minimum R,C,,-product at 250 GHz [13]. The n- thickness is chosen to be the zero-bias depletion width of 700 A to minimize the series resistance R,. The n+ layer is 1.5 pm- thick with a doping greater than 5 x 10" cm-". The 1.5 pm is less than the 250 GHz skin depth of 8 pm and was chosen for the possibility of an oxygen ion bombardment isolation. The receiver fabrication process is as follows (Fig. 7):

1) Deposit a 2400 8, stress-balanced PECVD Si,N, layer. 2) Pattem the Schottky anode while the surface is com-

pletely planar. 3 ) RIE etch the "anode well" in the Si,N, until only

200-250 8, Si,N, remains on the wafer. The dry etch


GEARHAKT AND REBEIZ A MONOLITHIC 250 GHz SCHOTTKY-DIODE RECEIVER 2507

insures minimal undercut, but must not go completely through the Si,N, layer since the RIE etch will damage the GaAs surface.

4) Pattern the ohmic contact (and receiver), RIE etch through the Si,N, layer, wet etch 1500 8, of the GaAs (through the n- layer) in HC1:Hz02:H20 at a volume ratio of 1:4:40.

5) Evaporate the ohmic layer: Ni/Ge/Au/Ti/Au at 250/325/ 650/200/1500 A, respectively.

6) Anneal the ohmic contact at 230/405/230 C for 20/40/20 sec, respectively.

7) Pattern the finger layer. This layer forms the finger for the diode, increases the thickness of the gold in other portions of the receiver, and forms the Schottky anode metal (the finger pattern covers the anode well).

8) Etch the remaining 200-250 8, of Si,N, in BHF acid. 9) Etch the GaAs oxide in HCl:H20 at 1:l. 10) Evaporate the finger/Schottky anode metal: Ti/Pt/Au at

200/200/3000 A. 11) Isolate the diode using a wet chemical etch (NH40H:

H202) that undercuts the diode finger [14]. This etch also removes the nS-layer in the antenna and in between the conductors of the CPW-line. Otherwise, the n+-layer would short circuit the slot antennas and CPW transmission lines.

Unfortunately, the diode isolation etch renders the receiver surface nonplanar and prohibits the integration of air-bridges to equalize the CPW ground planes. The slot-antenna structure is small compared to a guided wavelength and perfectly symmetrical, and therefore no even modes are triggered in the structure. As will be seen later, the exclusion of air-bridges do not seem to have a detrimental effect on the measured receiver performance at 250 GHz.

During the fabrication process, the anode diameter increased to 1.5-1.6 pm. The diode junction capacitance is therefore approximately C,, = 4 fF. A major advantage of the monolithic design is the elimination of the pad-to-pad parasitic capacitance. This capacitance becomes part of the RF transmission line which is connected to the diode terminals. The small remaining parasitic capacitance is primarily due to the region under the diode finger and around the anode and is estimated to be C,, = 2 fF (Fig. 9). The parasitic capacitance C,z from the finger to ohmic contact should be much less

Fig. 9. Top view. (b) Side view.

Parasitic capacitance due to the finger region around the anode. (a)

than C,, since the separation from the finger to the anode of 5 pm is much greater than the distance from the finger to the underlying GaAs 0.25 pm.

The dc parameters are calculated by measuring the current- voltage characteristic and curve fitting to the following stan- dard diode I-V equation:

I ( V ) = I , ( exp 4(Vv-F3) - 1) where 1, is the reverse saturation current, q is the electronic charge (1.6 x C), R, is the series resistance, v is the diode ideality factor, IC is the Boltzmann constant (1.38 x J/K), and T is the physical temperature in degrees Kelvin [15]. If the area A of the diode is known, the diode barrier height ab can also be found as

KT A**T2A a b = -1n (T)

4

where A** is the effective Richardson constant [15]. The measured dc parameters of the fabricated diode are R, =

This yields a figure-of-merit cutoff frequency given by f~ = 13 0, 7) = 1.18, a b = 0.70 v and I , = 1.0 X

1 / 2 ~ R ~ ( C j o + C,) of 1700 to 2000 GHz. A.

IV. MEASUREMENTS A polyethylene ( E , = 2.3) matching cap was developed to

reduce the 1.7 dB RF reflection loss at the silicon (elens = 11.7) lens-air interface. Ideally, the matching cap should have a dielectric constant of match = 6 = 3.42 with a uniform thickness of Xmatch/4 = 162 pm resulting in a reflection loss of only 0.2 dB [ 5 ] . This residual reflection loss of 0.2 dB is present because the substrate lens is of an extended hemispherical design instead of a pure hemispherical design. The effect of the nonoptimum matching cap can be estimated by considering the matching cap as a section of transmission line with a characteristic impedance of Zcap = Z,/+ = 249 f2 and the silicon lens as a load termination with impedance Zcap = Z,/& = 110 R where Z, is the free-space impedance of 377 0. For a polyethylene


2.508 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 42, NO. 12. DECEMBER 1994

5 : .3 n - d 300 1 . - w 5: 200

a 0 W

> 100 Angle (degrees )

(a)

-

-

-

Angle (degrees )

(b)

Fig. 10. Measured E-, H-, and 45"-plane patterns at (a) 237 GHz and (b) 280 GHz. The high E-plane sidelobes at 280 GHz are due to interference from the IF filter (see text for more detail).

( E , = 2.3) matching cap, a minimum reflection loss from the air to the silicon of 0.2 dB occurs for a thickness of X,,,/4 = 198 pm. Therefore, the overall reflection loss for the silicon lens with a polyethylene matching cap is approximately 0.2 + 0.2 dB = 0.4 dB, a 1.3 dB improvement over the reflection loss with the silicon lens alone. The matching cap is fabricated by melting a 220 pm-thick polyethylene layer over the silicon lens on a hotplate at 225" C . Several matching caps were fabricated and then removed to measure their thickness. Typically, the matching caps were 190 +C 15 pm-thick in the center and 175 f 15 pm-thick on the edge.

The double-slot antenna is centered on a 13.7 mm diameter extended hemispherical silicon lens with the polyethylene matching cap. The pattems are measured by dc biasing the diode and using it as a video detector. Fig. 3 shows the measured E- and H-plane patterns at 258 GHz, which agree well with theory [5]. A crosspolarization peak of -20 dB at broadside was measured in the E-, H- and 45O-planes due to the widening of the CPW line between the slot- antennas. A forward pattern directivity of 28.5 dB at 258 GHz is calculated by geometrically averaging the measured direc- tivities of the E-, H-, and 45"-plane patterns. The 28.5 dB directivity agrees well with the predicted value of 29 dB [5]. For the 13.7-mm diameter silicon lens, this results in a measured aperture efficiency (coupling to a plane wave) of 55%. The radiation patterns were also measured at 237 GHz and 280 GHz to test the bandwidth of the double- slot antenna. The patterns are symmetric at 237 GHz, but the 280 GHz E-plane shows unsymmetric sidelobes (Fig. 10).

b a a h a w i t h matching cap * . r * * t n o matching cap

0 L- - - - -_ j 230 240 250 260 270 280 290

Frequency ( G H z )

Fig. 1 I . matching cap for the silicon lens.

The receiver video responsivity with and without a polyethylene

This slight antisymmetry in the E-plane is due to the IF filter performance at 280 GHz. We believe that the filter does not present a perfect short circuit at the antenna terminals, and the two slot antennas are fed with slightly different magnitude and phase at 280 GHz. This results in our non- symmetric E-plane radiation patterns with high sidelobes. A similar double-slot antenna has been tested and resulted in symmetric patterns from 220-280 GHz with a polyimide capacitor replacing the IF filter [5]. In the future, a 7-section IF filter can be integrated with a -20 dB rejection over the 230-280 GHz range.

The video responsivity, defined as the detected diode voltage divided by the total RF power incident on the 13.7 mm lens, aperture, of the receiver was tested with and without the polyethyene matching cap. The incident RF power was measured using a large area bismuth bolometer on a dielectric membrane [16], and the detected diode voltage is measured across a 100 KR load. The matching cap was found to increase the video responsivity by 1.0 f 0.2 dB from 246 to 258 GHz while having no measured effect on the antenna radiation patterns. The measured video responsivity with the matching cap on the silicon lens is 410 V/W at 246 GHz and is 330 V/W at 258 GHz (Fig. 11). Outside of the 246 to 258 GHz frequency range, the receiver video responsivity falls off due to the drop in antenna aperture efficiency from the lower gain antenna radiation patterns (Fig. 10). The video responsivity referenced to the diode junction, defined as the detected diode voltage divided by the RF power available at the diode terminals, may be calculated with the inclusion of the following losses: the antenna aperture efficiency (coupling to a plane wave, estimated to be 55%), the silicon-lens reflection loss with a matching cap-layer (0.7 dB), the absorption loss in the high-resistivity silicon lens (0.2 dB), and the power lost to the backside of the antenna (0.4 dB). The resulting video responsivity referred to the diode terminals is lo00 V/W at 246 GHz and is competitive with the performance of whisker- contacted diodes at 250 GHz.

The quasioptical measurement setup (Fig. 12) is designed for 258 GHz because our LO source exhibits a peak power at this frequency. The Mach-Zender interferometer (designed for


GEARHART AND REHEIZ: A MONOLITHIC 250 GHz SCHOTTKY-DIODE RECEIVER 2509

(b)

Fig. 12. The 258 GHz quasioptical measurement setup (a). The Gaussian beam for double-slot on the extended hemispherical lens. The minimum beamwaist of 3. I mm occurs 38 mm behind the surface of the lens (b).

a center frequency of 250 GHz) allows the RF and LO signals to be combined at the receiver, and the two teflon objective lenses focus all of the LO power supplied by the corrugated LO feedhorn to the phase center of the double-slot antenna on the extended silicon lens. A tunable Gunn diode with a tripler is the 258 GHz LO source. The corrugated LO feedhom has a beamwaist of 2.0 mm. A 65 mm-diameter f/0.85 objective lens is located 60 mm from the horn aperture. This lens focused the Gaussian beam through the interferometer to the seond teflon objective lens, a 65 mm-diameter fll.4 lens. Both teflon lenses are smooth and have no matching grooves to reduce reflections. Finally, the Gaussian beam is focused 107 mm behind the fA.4 lens to the 3.1 mm minimum beamwaist of the double-slot antenna on the extended hemispherical silicon lens. This minimum beamwaist is 38 mm behind the surface of the substrate lens as per the work of Filipovic, ef al. [SI (Fig. 12).

The Mach-Zender interferometer is tuned to an IF of 1.4 GHz. The 1.4 GHz IF-chain has a noise temperature of 105 K and a gain of 97 dB with a bandwidth of 100 MHz. The measured DSB conversion loss and noise temperature versus bias and available LO power are presented in Fig. 13. The best performance was achieved at a dc bias of 0.73 V and a dc current of 1.2 mA. The available maximum LO power at the silicon-lens aperture is 1.65 mW. The available LO power at the diode terminals is calculated by multiplying the above power by the antenna backside power loss (0.4 dB) and Gaussian-coupling efficiency (0.5 dB), the lens-air reflection loss with the matching cap layer (0.7 dB), and the silicon lens absortion loss (0.2 dB). This results in a maximum available LO power at the diode terminals of 1.1 mW, which is just enough to optimally pump the diode (Fig. 13). The reflection and absorption loss (estimated at 0.3 dB and 0.3 dB,

0.0 0.2 0.4 0.8 0.8 1.0 1.2 1.4 1 8 Bias Current (ma)

Available LO Power at diode terminal-. (mW) 0.13 0.35 0.57 0.78 1.01

16 I , , , - , ~ , ,

0 2 0.4 O B 0 8 1.0 1.2 1.4 1.6 L.B Averlable LO Power at lens aperture, (mw)

(b)

Fig. 13. Measured antenna-mixer conversion loss and noise temperature (including the IF-chain contribution) at 258 GHz versus: (a) bias current at an available LO power of 1.65 mW at the lens aperture, (b) available LO power at a bias of 1.2 mA.

respectively) of the 65 mm-diameter fA.4 teflon objective lens and a 0.2 dB insertion loss in the interferometer have been normalized out of the measurements [18]. The measured DSB conversion loss and receiver noise temperature is 9.0 f 0.3 dB and 2850 f lOOK at 258 GHz without a matching cap layer and 8.0 f 0.3 dB and 2250 f lOOK with the polyethylene matching cap layer. The measured IF reflection coefficient of 0.2 dB at a bias current of I b = 1.0-1.2 mA and the noise contribution of the IF chain can be normalized out of the receiver measurements. The minimum DSB antenna-mixer conversion loss and noise temperature is therefore 7 .8f0.3 dB and 1600 f 100 K. These results are within 2-3 dB of the best tuned waveguide mixers using planar diodes [ 11 and represent the first monolithic 250 GHz receiver to date.

The theoretical receiver performance is analyzed using the reflection algorithm of Held and Kerr [12] with the actual diode parameters of I , = 1 x A, rl = 1.18, C,, = 4 F, Cp = 2 fF, = 0.70 V, and R, = 13 R. The RF embedding impedance is 2Z1 = 72 + j 2 8 R at 258 GHz and the 1.4 GHz IF embedding impedance is 110 R due to the Duroid matching network. For an LO pump power of 3.5 mW and a bias current of 1.4 mA, the theoretical single sideband conversion loss is 6.9 dB and the single sideband noise temperature is 1930 K. Concerning the theoretical receiver performance, we are more interested in the theoretical conversion loss because we have noticed that the program is somewhat inaccurate in predicting noise temepratures. Note that these figures do not include the


2510 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 42, NO. 12. DECEMBER 1994

residual silicon lens-air reflection loss of 0.7 dB, the silicon lens absorption loss of 0.2 dB, the back-side power loss of 0.4 dB, or the Gaussian coupling efficiency loss of 0.5 dB. Including these losses and employing the equivalent noise temperature of an attenuator [ 171, the overall theoretical single sideband conversion loss is 8.7 dB and the noise temperature is 3075 K. Assuming that the receiver responds to both Eidebands identically, these results correspond to a double- sideband conversion loss of 5.7 dB and a noise temperature of 1540 K. This discrepancy between the predicted and measured receiver performance is probably due to the increased series resistance from the skin effect at 258 GHz. Additionally, i t has been shown that the parasitic shunt capacitance for a planar Schottky diode increases with frequency at millimeter- wave frequencies [ 181. Regardless, the agreement between the experimental results (L,,, = 7.8 dB, T,, = 1600 K) and theory ( L , = 5.7 dB, T,,, = 1540 K) is quite good for a 250 GHz receiver.

The imaging properties of the receiver have also been tested by measuring the radiation patterns of adjacent antenna elements in the imaging array at 258 GHz. The central antenna element is aligned to the center of the extended lens, and the other two elements are separated by a spacing of 800 pm each (Fig. 14). The 800 pm separation was chosen to facilitate electron beam alignment if necessary for the Schottky anodes but is too large to sufficiently sample the data in an imaging application. As can be seen from Fig. 14, the off- axis H-plane patterns do not greatly vary for elements spaced 1600 pm (5X,!) from the central element. This corresponds to a 2 8 = 80 field of view indicating that double-slot antennas on extended hemispherical lenses have a good field of view with minimal off-axis aberrations. The measurements in Fig. 14 are in good agreement with theoretical patterns developed using a ray optics diffraction model [ 191.

V. CONCLUSIONS

The monolithic double-slot receiver on an extended silicon substrate lens is one of the best planar Schottky-diode receivers for millimeter-wave and submillimeter-wave frequencies. The double-slot antenna on an extended hemispherical lens exhibits high-gain high-Gaussian coupling efficiency radiation patterns which are not seriously degraded as the double-slot is moved off the axis of the hemispherical lens, making the receiver suitable for 2-D imaging arrays. At 258 GHz, the measured DSB antenna-mixer conversion loss and noise temperature is 7.8 -+ 0.3 dB and 1600 k 100 K. These results are within 2-3 dB of the best tuned waveguide mixers using planar diodes [ I ] and represent the first monolithic 250 GHz receiver to date.

ACKNO w LEDGMENT The authors would like to thank D. Filipovic of the Uni-

versity of Michigan and G. Eleftheriades, formerly at the University of Michigan and currently at EFPL in Switzerland for their excellent theoretical support. The authors would also like to thank the talented William Bishop of the University of

-60 -40 -20 0 20 40 6 0 0 -m-mr- P;;; Center 7

I Angle (degrees)

Fig. 14. on a 13.7 mm diameter silicon substrate lens.

Measured H-plane patterns of three antenna elements at 258 GHz

Virginia for his suggestions about Schottky-diode fabrication procedures. Lastly, the authors thank the Rogers Corporation for the donation of the substrate.

REFERENCES

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GEARHART AND REBEIZ: A MONOLITHIC 250 GHz SCHOITKY-DIODE RECEIVER 2511

II-experiment, IEEE Tram Microwave Theory Tech., vol. MlT-26, pp. 49-61, Feb. 1978.

[13] T. W. Crowe and R. J. Mattauch, Analysis and optimization of millimeter-and submillimeter-wave mixer diodes, IEEE Trans. Mi- crowave Theory Tech., vol. MTT-35, pp. 159-168, Feb. 1987.

1141 W. L. Bishop, K. McKinney. R. J. Mattauch, T. W. Crowe, and G. Green, A novel whiskerless Schottky diode for millimeter and submillimeter wave applications, Proc. IEEE MTT-S Int. Symp., pp. 607-610, June 1987.

[15] S. M. Sze, Physics of Semiconductor Devices. New York: Wiley, 1981. /16] C. C. Ling and G. M. Rebeiz, A wide-band monolithic quasi-optical

power meter for millimeter and submillimeter-wave applications, IEEE Trans. Microwave Theory Tech., vol. 39, no. 8, pp. 1257-1261, Aug. 1991.

[ 171 F. T. Ulaby, R. K. Moore, and A. K. Fung, Microwave Remote Sensing- Acme and Passive, Vol. I, Reading, MA, 1981.

[18] J. A. Wells and N. J. Cronin, Frequency-dependent simulation of planar millimeter-wave mixer diodes, presented at 17th Int. Conf. Infrared Millimeter Waves, Dec. 1992, pp. 214-215.

[I91 D. F. Filipovic and G. M. Rebeiz, Off-axis imaging properties of substrate lens antennas, in Pmc. 5th Int. Symp. Space THz Technology, May 1994.

P Steven S. Gearhart received the B.S.E.E. degree from the Virginia Polytechnic Institute and State University, the M.S.E.E. and Ph.D. degrees from the University of Michigan, Ann Arbor, in 1988, 1990, and 1994, respectively.

In September 1994, he joined the faculty of the Department of Electrical and Computer Engi- neering at the University of Wisconsin-Madison as an Assistant Professor. His research interests include millimeter-wave planar antennas, integrated receivers, imaging arrays, ind silicon and GaAs

fabrication. He was the recipient of a best student paper award at the 1994 IEEE-MTT Intemational Symposium.

Gabriel M. Rebeiz (S86-M88SM93), for a biography, see the April issue of this TRANSACTIONS, page 545.


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