Design of Compact and Auto-Compensated Single-Layer Chipless RFID Tag

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 60, NO. 9, SEPTEMBER 2012 2913

Design of Compact and Auto-CompensatedSingle-Layer Chipless RFID Tag

Arnaud Vena, Etienne Perret, Member, IEEE, and Smail Tedjini, Senior Member, IEEE

Abstract—This paper presents a new radio frequency identifica-tion (RFID) chipless tag that is highly compact and potentially low-cost. This tag has a lot of advantages, such as being fully printableon products since no ground plane is needed for fabrication. Theactual issue of the chipless tag family having a single layer, that is,their detuning effect, is compensated for the first time by a correc-tion technique based on the use of a sensing resonator. The designis based on multiple coplanar strip-line resonators where res-onant frequencies can be shifted by setting an additional short cir-cuit at particular locations. An accuratemodel is proposed to easilylink the footprint of the structure to the resonant frequency. Con-sidering a frequency resolution of 50 MHz for the reading systemand a tag dimension of 15 20 mm , 9 b can be encoded in the fre-quency band 2.0–5.5 GHz. Several experimental results validatethe proposed design as well as its implementation in a realistic ap-plication and environment.

Index Terms—Auto-compensation, chipless radio frequencyidentification (RFID), compact size, coplanar strip resonator.

I. INTRODUCTION

R ADIO frequency identification (RFID) is an enablingtechnology that allows remote identification and authen-

tication by using RF waves. RFID systems are widely used inthousands of applications such as fare collection, road tolls,and item tracking. To conquer the huge market of consumergoods tracking that is addressed today by optical barcodesand representing every year a market of ten trillion items [1],chipless RFID tags, also called RF barcodes, have severaladvantages. Indeed, RF signals allow nonline of sight (NLOS)and larger read range as well as less manual positioning. Astudy from [1] predicts that, in 2019, chipless tags will be soldat 75 billion units annually. These features explain the growinginterest of businesses for such an emerging and promisingtechnology. It is generally claimed that a target unit cost ofless than 0.1 cent has to be achieved before this technologycan be implemented in practice. One way for cost reduction isto develop a chipless tag solution that is directly printable onproducts. To satisfy these constraints, the chipless tags have to

Manuscript received January 25, 2012; revised May 25, 2012; accepted June04, 2012. Date of publication July 10, 2012; date of current version August28, 2012. This was supported in part by the French National Research Agencyunder the ANR-09-VERS-013 Program and the Grenoble Institute of Tech-nology through the BQR Program.The authors are with the Laboratoire de Conception et d’Intégration des Sys-

témes (LCIS)/Grenoble Institute of Technology, 26000 Valence, France (e-mail:[email protected]; [email protected];[email protected]).Color versions of one or more of the figures in this paper are available online

at http://ieeexplore.ieee.org.Digital Object Identifier 10.1109/TMTT.2012.2203927

be realized using only one conductive layer, and their size hasto be fully compatible with item tracking applications.Chipless RFID is fundamentally an alternative solution

to conventional high-frequency (HF) or ultrahigh-frequency(UHF) RFID [2]. In terms of application, a chipless approachis quite comparable to optical barcode technology in termsof tag fabrication and cost. However, the detection principle,based on RF waves, has many improvements; among them arelarger read range and the possibility to detect the tag regardlessof its orientation in a large detection area. Consequently, theuse of an automatic reading system becomes possible. In thecase of conventional RFID tags based on an IC chip, a specificrequest is sent to the tag by a reader using an ASK modulationscheme. The IC chip decodes the signal, processes the request,and, finally, in turn provides the response using the impedancemodulation technique. On the contrary, a chipless tag doesnot require any transmission protocol. Essentially, it can beseen as a simple target having a specific temporal/frequencyelectromagnetic signature (EMS). There is no rewritable ca-pability to encode data in this case. Indeed, such a capabilityrequires a modification of the EMS, which can be obtained byphysical modification of the footprint of the tag only. Thus, forchipless tags, the information is directly coded and embeddedinto the antenna of the tag. In the literature, two approaches forencoding data into a chipless tag can be found: time-domainapproach (TDA) and frequency-domain approach (FDA).The TDA is based on controlling the presence and position

of multiple reflections of a tag in response to an incident shortpulse. This technique is used by the SAW chipless tag tech-nology [3] that operates in the ISM band at 2.45 GHz. However,the SAW tag is quite expensive because of the use of a piezo-electric substrate. Other solutions have been proposed using astructure with delay line and coupled distributed [4] or lumpedcapacitor [5] to generate reflections. The former approach hasa capacity of 8 b and a length of 400 mm. For comparison, aSAW tag can encode 256 b in a ceramic package of size 12.56.25 mm , but an antenna has to be connected to this package

as for conventional RFID tags.The FDA is based on controlling the amplitude and phase of

the multiple resonances of the tag. Binary information can besimply coded by the presence or the absence of the resonances.The structure that codes the largest amount of data was intro-duced by Preradovic et al. [6], with 35 resonators correspondingto 35 b of data. However, the structure is cumbersome since itssize is quite larger than the standard credit card format becausethe presence of two wideband antennas: one for receiving andone for transmitting. On the other hand, the most compact sizedesign was proposed by Jalaly et al. [7]. It uses multiple

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2914 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 60, NO. 9, SEPTEMBER 2012

short-circuited dipoles of different lengths. However, the pre-sented structure has a low coding capacity of only 5 b and stillhas a ground plane.Regarding all previous solutions, it can be highlighted that all

proposed tags are much larger in size compared with an opticalbarcode. Moreover, most of the tags are designed with two con-ductive layers, making the technique of direct printing on theproduct impossible. However, a configuration based on the useof a space-filling curve to create a planar resonator has shownthe possibility to use only one conductive layer to generate res-onances [8]. The proposed design in [8] is highly miniaturizedand has the possibility to encode 5 b between 0.7 and 0.9 GHzin a surface of 200 30 mm .The novel chipless tag structure presented in this paper is

based on the FDA. It has several advantages especially in termsof compactness. Moreover, unlike the structure reported in [6],each element of the proposed tag acts, at the same time, asthe receiving antenna, resonator, and transmitting antenna. Res-onators used are based on a coplanar strip line (CPS) with oneside as a short circuit and the other side as an open circuit. Anaccurate model has been extracted to take into account these dis-continuities and to fit the well-known formula for CPS line. Thisstep is necessary in order to make a direct link between a spe-cific electromagnetic footprint having multiple resonances andthe physical size of each resonator. As a consequence, the fabri-cation process could be fully automated because, for a specificcode, a set of values for the length is computed using the mod-ified CPS formula. Unlike most of the previous designs, thereis no ground plane, and only one conductive layer is needed.This improvement is essential for decreasing the cost of fabri-cation and opens the way to the techniques of direct printing onproduct. However, the ground plane has a key role because itallows very good isolation between the container and the tag.This prevents any detuning effect. Thus, it is a great challengeto design a single-layer tag robust to the environment. Similarto UHF RFID [2], the tag can be designed to be placed on a spe-cific container with a given material and thickness. In this case,its use becomes very complex because, for every new item toidentify, a new design should be prepared. Therefore, in orderto overcome this problem, a compensation (or calibration) tech-nique based on the use of an additional resonator to probe theobject is implemented for the first time.Finally, it is possible to configure the tag after fabrication

just by setting small short circuits. Additionally, to improve thecoding capacity for each resonator, a technique based on fre-quency-shift encoding has been exploited.This paper is organized as follows. Section II presents the

physical description of the tag and the performed encodingmethod. Section III introduces the used measurement setup andthe results obtained for design validation. Section IV providesa discussion about the way to implement such a technology fora real application before concluding on the performance of thetag in comparison with the conventional designs.

II. DESIGN AND CODING TECHNIQUE

In the FDA, data are encoded for each tag by controlling thepresence of resonances in the bandwidth of the reading system.The coding technique widely used in literature [6], [7] is based

Fig. 1. Frequency coding principle with a 2-b chipless tag, using two differentcoding approaches: (a) absence/presence of resonances and (b) frequency shiftand absence/presence of resonances.

on presence/absence of resonance for specific frequencies asshown in Fig. 1(a). In this figure, two coplanar strip resonatorsare used for encoding 2 b. This technique is reliable but poor interms of density of coding per surface, because one resonatorprovides 1 b. Another technique presented by Balbin [9] usesthe phase encoding. Indeed, for a given frequency, the phasecan exhibit several values. This technique allows higher codingcapacity, but it needs a phase reference to be used in practice.The technique presented here [see Fig. 1(b)] already introducedin a previous work [10] [11] combines multiple bits in just onecoplanar strip resonator. The resonant frequency can be shiftedto take several values in a defined frequency band. In the ex-ample in Fig. 1(b), the density of coding is enhanced and allowsminiaturization of the tag by a factor of 2, corresponding to 2 bof capacity.In order to discriminate a large number of frequency values

in a defined bandwidth, the frequency shift has to be as small aspossible. This condition is attainable only if the quality factorof each resonance is sufficiently high, but an improved qualityfactor also means a poor immunity to the surrounding environ-ment. In addition, increasing the number of resonators is an evi-dent way to increase the capacity of coding, but the couplingeffect has to be taken into consideration. Hence, for a giventag, the number of coding resonators is limited by their size andtheir mutual coupling. The tag introduced in this work depictedin Fig. 2 is based on a multiple-band coplanar-strips resonatorwith open-ended extremities. The basic principles of this struc-ture have already been introduced in a previous work [10] [11],and here we develop a model in order to provide a direct linkbetween a specific code and the physical parameters of the tag.Furthermore, a technique to recover the initial resonance fre-quencies if the tag is placed on objects having various dielectricproperties is developed.

A. Tag Description

With respect to the previous considerations, the structure al-lows a good selectivity for each resonator and a good decou-

VENA et al.: DESIGN OF COMPACT AND AUTO-COMPENSATED SINGLE-LAYER CHIPLESS RFID TAG 2915

Fig. 2. Photography of designed (a) tag 1, (b) tag 3, and (c) tag 4. (d) Geometryof the “C” shape chipless RFID tag. The substrate is FR-4, with andthickness of 0.8 mm. The initial slot lengths with no short-circuit are denoted

, , and . Short circuits denoted , , and allow changing thefrequency of each resonator. Dimensions of the tags are given in Table II.

Fig. 3. . Simulation result of tag presented in Fig. 2 for various valuesof slot length L1, keeping other slot lengths constant.

pling between them. When the structure is excited with a verti-cally polarized wave [vertical arrow denoted as E in Fig. 2(d)],it gives a footprint having four resonances, which are producedby resonators denoted as 1, 2, 3, and 4 in Fig. 2(d). The fre-quency of resonators 1, 2, and 4 can be controlled independentlyas shown with simulation results plotted in Figs. 3–5. It was ob-served from simulation using CST Microwave Studio that res-onator 3 is largely influenced by the changes in surface currentof the other modes. As a result, it cannot be used for the en-coding. However, its presence is mandatory, as it ensures a gooddecoupling between resonators 1 and 2 on one side and res-onator 4 on the other side. On the other hand, it will be shown, inSection IV, that this resonator can be used to probe the relativepermittivity of the environment. It is to be noted that openingthe coplanar strip line on one extremity and setting a short cir-cuit on the other side produces a quarter-wavelength resonance.The combination of the quarter-wavelength resonator and thepresence of the right angle in their slot as for resonators 1 and 2allow an appreciable size reduction. However, the RCS of sucha resonator is weaker than for the case for which no short circuitis used (see Fig. 6). Indeed, a value of 28 dBsm is observed forthe resonant peaks when the half-wavelength resonator is trans-formed into two quarter-wavelength resonators, instead of 25dBsm reached for the open-circuit case.

B. Model and Encoding

As previously indicated, to encode information, the resonantfrequency of each resonator is varied around a reference. The

Fig. 4. . Simulation result of tag presented in Fig. 2 for various valuesof , keeping other slot lengths constant.

Fig. 5. . Simulation result of tag presented in Fig. 2 for various valuesof , keeping other slot lengths constant.

Fig. 6. . Simulation result for various bended coplanar strips resonators.

slot lengths can be modified by adding some short circuits, de-noted as , , and in Fig. 2(d). Several short-circuit con-figurations can be observed in Fig. 2(a)–(c) showing, respec-tively, the realized tags 1, 3, and 4. The resonant frequency ofone resonator can be split into two resonant modes if the shortcircuit is not symmetrical with respect to the horizontal axis. Inorder to obtain the physical length of the resonator as a func-tion of resonant frequencies, closed-form expressions [12], [13]can be used to calculate the effective wavelength of a coplanarstrip line (CPS) with a cross section shown in Fig. 7(a). Havingthe effective permittivity, to obtain the physical length of thecoplanar strip resonator, one can take as a first approximationthe quarter wavelength as the slot length value. However, forencoding with a frequency resolution as low as 50 MHz, length


Fig. 7. (a) Cross section of CPS. (b) Resonator in configuration 1. (c) Resonatorin configuration 2 having a bended section.

variations close to 0.1 mm have to be realized. Consequently, amore realistic model has to be defined.As shown in Fig. 7(b) and (c), the slot line contains several

discontinuities, first due to the open and short terminations andsecond due to the right-angle section. Therefore, it is necessaryto add or subtract some portion of length denoted , , and, as a function of the shape of the slot. Indeed, as explained in

[14] and [15], a discontinuity in a transmission line is equivalentto an extension or a reduction of its electrical length. In order toestimate these parasitic lengths, the method used in this paperconsists of fitting simulation results obtained for various valuesof and to the analytical formulas [12], [13], giving a verygood accuracy. Equation (1) gives a relation between resonancefrequency and current path length for all configurations

(1)

(2)

(3)

where is the slot length and is the length extension dueto discontinuities.It has been found in simulation that is a function of

configuration and lengths and . A plane-wave simulationof the structure using CST has been done for various valuesof , , and . From the obtained results and using a linearregression model, simple relations to compute in millimetersfor configurations 1 (2) and 2 (3) have been extracted. Theserelations are valid within the entire frequency bandwidth of thetag, that is, between 2 and 6 GHz. The slot length variationas a function of the resonant frequency is plotted in Fig. 8. Weobserve that both configurations [see Fig. 7(b) and (c)] evolvein the same way. One can notice that a length variation willproduce a larger frequency shift at higher frequency.To define an encoding for such a specific structure, we use the

frequency span between 2 and 5.5 GHz. The lower frequencyis determined by the maximal slot length available for such acompact structure 15 20 mm , while the higher frequencyis chosen to be lower than the first higher order modeof resonator 1, that is, 6 GHz. For mode 1, the frequency valuecan vary between 2 and 2.35 GHz, leading to a bandwidth of350 MHz. The frequency span of mode 2 is between 2.4 and2.65 GHz, leading to a 250-MHz bandwidth, and, for mode 4,the frequency value is between 4 and 5.5 GHz, leading to a1.5-GHz bandwidth. One can notice that the frequency span be-tween 2.7 and 4 GHz is not used. The first reason is due to the

Fig. 8. Calculated resonance frequency as a function of slot length ofcoplanar strip-line resonator according to (1)–(3). Permittivity is 4.6 andsubstrate height is 0.8 mm.

presence of unwanted mode 3 resonating at 2.73 GHz. Second,to keep a compact structure, the inner slot length (denoted as 4)has to be lower than 13 mm, producing a minimum resonancefrequency of 3.5 GHz.For one resonator, the frequency shift can take a lot of values.

By defining a bandwidth allowed for resonator , a fre-quency resolution chosen as a function of measurement ac-curacy, and the number of resonators , the following equationgives the number of achievable combinations:

(4)

Having , one can find the number of bits that can be en-coded for each resonator. To determine the resolution , oneof the limiting factors is the accuracy of realization (chemicaletching) certified to be some 20 m, so that a minimum short-circuit variation length of 50 m was used. For resonator 4, atthe maximal frequency 5.5 GHz, a step length of 50 m gives a30-MHz variation in the worst case, so that a frequency resolu-tion of 50 MHz is possible. However, the other limiting factoris due to the full-width at half-maximum (FWHM) of the peak.For modes 1 and 2, it is equal to 20 and 40 MHz so that a fre-quency resolution of 50 MHz is still possible, but, for mode 4, itis equal to 100 MHz. As a result, the of resonator 4 is equalto 100 MHz. Using (4), it can be found that eight different fre-quency values can be encoded for the first mode, six frequencyvalues for the second mode, and 16 for mode 4, giving a globalcapacity of 8 6 16 768 combinations or 9.58 b.Table I shows the various short-circuit lengths [calculated

with (1)–(3)] to provide a specific resonance frequency and,thus, a specific code for each resonator. The initial slot lengthvalues are set to 24.7, 20.3, and 11.1 mm for resonators 1, 2, and4, corresponding to the initial resonance frequencies 2, 2.4, and4 GHz, respectively. The dark cells represent the configurationsselected for the realized tags.

III. DESIGN VALIDATION

The tag shown in Fig. 2 was designed to have three modes,with resonant frequencies starting at 2, 2.4, and 4 GHz corre-sponding to slots 1, 2, and 4, respectively. The thickness of thesubstrate is 0.8 mm. The gap widths are0.5 mm and 1 mm. The strip width is 2 mm and


TABLE ISHORT-CIRCUIT LENGTHS CALCULATION ACCORDING TO (1)–(3)

TABLE IIREALIZED TAG DIMENSIONS IN mm

the metal is copper with a thickness of 35 m. Five tags withvarious short-circuit length configurations have been realized,and their dimensions are summarized in Table II. For each con-figuration, a code can be associated depending on the value of, , and , as presented in Table I.

A. Measurement Setup

The measurements presented in this paper were done in ananechoic chamber and were confirmed in a real environmentusing a bistatic configuration depicted in Fig. 9(a) and (b). Weused an Agilent 8720D vector network analyzer (VNA) con-nected to two identical horn antennas that have a gain rangingfrom 10 to 12 dBi in the frequency band from 1.5 to 6 GHz. Thetag is placed 65 cm from the antennas separated by a distanceof 60 cm. The power delivered by the VNA is approximately0 dBm in the entire frequency band. To eliminate the spuriouscoupling effect between the two antennas, an isolation measure-ment is done without any tag as shown in Fig. 10. Then a ref-erence measurement with a target having a known RCS (e.g.,conducting plate) is done to compensate for the filtering effectof the antennas. To get the calibrated complex RCS value of themeasured tags, the following equation, derived from [16] in thecase of vertical or horizontal polarization only, is used:

(5)

In this equation, the measurements results are , ,and , respectively, for empty chamber (Isolation in Fig. 10),reference scatterer (Ref. in Fig. 10), and tag under test (Tag 1in Fig. 10). To obtain the complex RCS value of the tag ,the complex RCS value of the reference scatterer must beknown. For this purpose, a flat copper plate, of size 50 50 mmwas chosen. To determine , we find that the value extractedfrom the simulation provides the most accurate results. One can

Fig. 9. (a) Measurement setup using bistatic configuration. (b) Photographshowing the tag positioning inside the anechoic chamber.

Fig. 10. Raw measurements of the empty chamber, the reference scatterer, andthe Tag 1.

notice that we cannot extract any coding information from theraw measurement of the Tag 1 (Tag 1 in Fig. 10) before cali-bration is done. On the other hand, the flat copper plate givesa strong response. It reveals some distortions in the spectrumthat will be compensated when extracting the tag RCS responseusing (5).

B. Measurement Results

The measured RCS response of tags 1, 3, and 4 are presented,respectively, in Fig. 11(a)–(c) compared with simulation results.One can notice that the simulation and the measurement re-sults show a good agreement. The RCS levels and the frequencyvalues of the resonance peaks are in close agreement except formode 4, which has an RCS level lower than the one found withsimulation. Moreover, the notches observed between two peaksare less sharp. Indeed, even if a subtraction by isolation mea-surement is done, a noise floor level of 80 dBm is still presentlimiting the dynamic range of the measurement setup.Table III shows both measured and calculated resonant

frequencies for each of the realized tags depicted in Table II. Theaverage error between the calculated and measured resonancefrequency is lower than 30 MHz. That confirms the accuracyof the proposed closed-form model and makes it possible toaddress all of the tag configurations using an analytical model.To confirm the level of decoupling between resonators, the


Fig. 11. Simulation (dashed curve) and measurement (solid curve) results for (a) tag 1, (b) tag 3, and (c) tag 5. (d) Zoom of measurement results for tags1, 3, and 5 around the resonant frequency of mode 2. For measurement, the frequency resolution is 2.8 MHz.

TABLE IIIRESONANCE FREQUENCY MEASURED IN GIGAHERTZ

RCS measurement responses of tags 1, 3, and 5, having thesame value and different values between 0 and 1 mm,are plotted in Fig. 11(d). It can be clearly seen that only theresonant frequency of mode 2 is shifted in this case. However,a few unwanted shifts can be noticed and the most criticalone is observed for mode 2 of tag 2 (see Table III). Insteadof having a value equal to 2.4 GHz, we get 2.44 GHz, sothat the value is closer than 2.45 GHz, thereby producinganother code. This unwanted shift is due to coupling withthe first mode resonating at 2.25 GHz for this configuration.To limit this effect, the frequency span of the first mode canbe limited from 2 to 2.2 GHz, or the frequency span of thesecond mode can start at 2.45 GHz.In the other configurations, a maximal unwanted frequency

shift of 20MHz for mode 1 between tag 1, tag 2, and tag 5 canalso be observed, but this value is kept lower than the chosenfrequency resolution equal to 50 MHz for modes 1 and 2 and100 MHz for mode 4, so that this coding technique is reliable.

IV. APPLICATION CONSIDERATIONS

As described in the previous section, measuring the electro-magnetic response of such a chipless tag is a difficult task.Whendealing with real applications and practical implementations,several issues are to be considered by the designer. The mainaspects are discussed here.

A. Calibration Technique

The trick used is to remove the effect of the static environ-ment by subtracting a reference measurement without any tag.Implementing this technique in a real environment is possibleprovided certain precautions are taken. Indeed, the environmenthas to be well defined and surrounding objects generating spu-rious effects (e.g., unwanted echoes) have to be static. This canbe the case if the tag is read throughout a detection tunnel withconfined walls as illustrated in Fig. 12.In a real environment, we have several narrowband wire-

less communications sharing the free space channel, mainly at860–960 MHz and 2.45 and 5.8 GHz. If a chipless tag oper-ates in the UWB frequency span (between 3.1 and 10.6 GHz),we can simply avoid using the ISM bands for coding. More-over, the noise floor is relevant outside an anechoic chamber,and the backscattered signal to detect is generally very closeto this noise level. To increase the signal-to-noise ratio, we canapply an averaging on several identical measurements. For ex-ample, we found that a value superior or equal to ten allows areliable detection. Finally, in a real environment, there are mul-tiple paths due to reflections of the electromagnetic signature


Fig. 12. Reading system using a detection tunnel with confined walls makingpossible some calibration techniques as for reference subtraction.

by the surrounding objects. This provides some distortions onthe frequency-domain response. However, we can apply a timegating on the time-domain response to damp this effect.

B. FCC and ECC Rules Considerations

To validate our design, a VNA has been used, making a fre-quency sweep from 2.0 to 5.5 GHz with a constant transmittingpower of 0 dBm. A prototype reader based on this approachhas been designed by Preradovic et al. [17] for a conveyor beltsystem. Even if some specific applications, where transmittingpower is not an issue (e.g., confined room), can be addressed,most of the standard RFID applications need to be FCC- andECC-compliant regarding the rules for UWB systems. How-ever, a possible solution is to design an impulse radio-basedreading system. Indeed, using a very short pulse as an inter-rogation signal, it is possible to cover a very large frequencyband and to decrease the power spectral density (PSD) underthe limit of 41.3 dBm/MHz [18] in the UWB band from 3.1to 10.6 GHz. The pulse repetition rate can be as low as 1 MHz,giving a very low duty cycle signal. In a previous work [10],a time-domain reading system based on a pulse generator and adigital sampling oscilloscope (Agilent DSO91204A) was exper-imented in this way and demonstrated the possibility to detectthe tag identifier both inside and outside an anechoic chamber.

C. Anticollision Considerations

One tag can be detected at a time in a given space becausethere is no collision managing as for conventional RFID. How-ever, if the tags are distant in space, the transient response ofeach tag can be extracted individually using time windowing. Adetection system such as the one described in [10] can be usedfor this purpose.For spatial separation, another technique can be considered.

It consists of using high-focusing beam antennas. Generatinga spot size lower than 12 cm at 2.2 m is possible [19]. In thefrequency domain, if designing a tag with a low coding capacity,it can be defined by several frequency windows fully orthogonalbetween them so that, even if several tags are in the field, theycan all be detected.

D. Handling of Detuning Effect Due to Container Permittivity

One of the strengths of this design is that there is no groundplane so the tag can be potentially directly printed on product.

Fig. 13. (a) View of tag 1 applied on a container of 10 mm thickness. (b) Pho-tograph of tag 1 applied on a box full of paper.

Fig. 14. Simulated frequency response, when tag is put on a container of var-ious permittivity with a loss . Simulation was done using plane-wave excitation under CST Microwave Studio.

However, it is also one of its weaknesses if considering the de-tuning of resonators when the tag is put on a container as shownin Fig. 13(a). In Fig. 14, the simulation results of the tag puton a container of various permittivities and for a thickness of1 mm are given. One can see a global frequency shift for all ofthe peaks and troughs if the loss of the container is significantas for cardboard (i.e., at 2.45 GHz).Now, if by varying the thickness of the object, having a per-

mittivity of 3.6, as shown in Fig. 15, the overall RCS level isincreased and for 10 mm, it is very hard to distinguish each res-onant peak in the frequency spectrum, because the tag responseis overlapped by the container response. However, it can be ad-vantageous to consider the response in the time domain, in orderto separate the early time response due to the wideband responseof the container and the late time response that contains the mostresonant modes.However, solutions can be implemented to handle this un-

wanted effect. First of all, a chipless tag may be designed takinginto account the permittivity of the tagged container. In this way,a previous work in the field of classical UHF RFID [20] showedthe possibility of designing robust tags for a given family ofsubstrate (high k, low k, or conductor). However, if the mate-rial’s properties of the carrier are unknown, designing a specifictag does not provide a solution. Instead, a compensation tech-nique to take into account the variability of the container couldbe more efficient.


Fig. 15. Simulated frequency response, when the tag is applied on a containerhaving a permittivity of 3.6, a loss , and various thicknesses.Simulationwas done using plane wave excitation under CSTMicrowave Studio.

Fig. 16. Relative frequency shift according to (8) for each resonant mode as afunction of the permittivity of the tagged container [see Fig. 15(a)]. The valueshave been extracted from simulation results.

To increase the detection reliability and handle some vari-ability in the container permittivity, one or several resonatorsof the tag can be used to sense it. As a result, the real effec-tive permittivity of the tag can be found, and the reading systemcan compensate these eventual shifts to get the right code. In aprevious work [21], a single-layer chipless tag having 20 res-onators and base on an on–off keying approached has been de-signed. The two extreme resonators have been used as effectivepermittivity sensors to compensate the detuning effect thanksto a linear approximation of the frequency shifts between thelowest and highest frequencies. This paper introduces a com-pensating technique that uses only one resonator to sense thepermittivity. Moreover in this design, the tag ID is coded witha frequency-shifting technique. In the tag design introduced inFig. 2, it was said that the inner slot 3 is not used for encodingand serves to limit the coupling between others resonators. But itcan also be used as a “resonance sensor.” Its frequency is equalto 2.73 GHz when there is no container back to the tag. If a devi-ation is detected on this mode, deviations on all other modes canbe estimated so that their initial resonance frequencies can berecovered. Indeed, because the frequency value of this mode isconstant and known whatever the tag configuration, it can serveas a deviation sensor for the other modes.

To fully exploit this “correction” method, it is necessary tofind a relation between the frequency deviation of the knownresonance frequency (mode 3) and the other ones. We startwriting the ratio of the measured resonance frequency with thecontainer by the one without the container as shown in

(6)

This gives the relative frequency deviation due to the environ-ment. In this equation, and are, respectively, the mea-sured frequency of the encoding resonator (modes 1, 2, and 4)and its effective permittivity in the presence of the environmentwhile and are the initial free space values to recover.In the real case, the effective permittivities and

are not known, but one can assume that they are linked to theeffective permittivities and of the sensing res-onance (mode 3) by a constant as shown in

(7)

Finally, using (6) and (7) and substituting effective permittivityby their expression, one can obtain

(8)

showing that the relative frequency shift of a given mode isequal to those of the sensing mode. In this equation, isthe measured frequency value and is the known initial fre-quency value (2.73 GHz) of the sensing mode. The followingequation allows recovery of the initial frequency value of modes1, 2, and 4 depending on the relative shift of mode 3:

(9)

To check the validity of this assumption, the variation of therelative frequency shift, for eachmode and for several simulatedenvironments is plotted in Fig. 16. The permittivity of the con-tainer varies from 1 to 10. One can see that for permittivity until

, the value of the relative shift is quasi-equal for all of themodes and a maximal error of is observed between modes1 and 3. This error will define theminimum frequency resolutionusable for a specific application. At 2 GHz, means a fre-quency shift equal to 40 MHz, while at 2.5 GHz it is equal to50 MHz and at 5 GHz, to 100 MHz. Consequently, if the tag

is put on a container of unknown permittivity, a frequency res-olution of 100 MHz has to be adopted for modes 1 and 2, whilea frequency resolution of 200 MHz must be used for mode 4.This leads to a decrease of the coding capacity compared withthe ideal case when there is no strong dielectric loading fromthe container. With (4), using these new parameters, a codingcapacity equal to combinations or 6.58 b can beestimated.To validate this useful equation, measurements have been

done varying the size, thickness, and permittivity of the con-tainer, and, at each time, the corrected resonance frequency iscalculated for all of the modes. We used some rectangular plates


Fig. 17. Measured response of the tag 1 put on a PTFE ( ,) or a Plexiglas ( , ) plate with a size equal

to 5 5 cm and thickness 1.5 mm.

Fig. 18. Measured response of tag 1 put on a Plexiglass ( ,) plate of various thickness with a size equal to 5 5 cm .

made of PTFE, Plexiglass, and Carp. Using the cavity method[22], we measured relative permittivities of 2.1, 3, and 4.1 andloss tangents of 0.002, 0.005, and 0.1 at 2.5 GHz, respectively.All of the rectangular plates were 1.5 mm thick and their sizecan be 5 5 cm or 10 10 cm . Keeping the same measure-ment setup that was used to validate the design, the RCS re-sponses for all of the tags put on the various rectangular plateswere measured.Fig. 17 shows the RCS magnitude response of tag 1 put on

a plate of size 5 5 cm . One can notice that the greater thepermittivity of the container the greater is the frequency shift.The level of response is also affected by the container, and thetroughs are not as sharp as for the “no container” case, particu-larly for the highest frequencies.Fig. 18 shows the influence of the container thickness on the

RCS response of tag 1. The frequency shift is linked to the con-tainer thickness until a certain value. After that, only the levelof the RCS response seems to be affected.In applying (9) to recover the initial resonance frequencies

of the tag put on a container made of PTFE with various thick-nesses, Figs. 19 and 20 show the frequency deviation due tothe environment, respectively for the first mode and for mode4, before and after correction. In Fig. 19, the correction allowsrecovering the correct frequency value with an error of 20 MHzfor both tags 1 and 4. Fig. 20 shows that, for mode 4, an errorof 50 MHz is still present after correction for tag 3, but large

Fig. 19. Measured and corrected resonance frequency for mode 1, when thetag is put on a PTFE plate having thickness between 0 and 4.5 mm.

Fig. 20. Measured and corrected resonance frequencies for mode 4, when thetag is put on a PTFE plate having thickness between 0 and 4.5 mm.

TABLE IVFREQUENCY SHIFT MEASURED AND CORRECTED IN MEGAHERTZ FOR TAG 1

shifts up to 280 MHz have been compensated using this ap-proach. In Tables IV and Table V, the measured shifts are sum-marized, respectively, for tags 1 and 3. In most of the cases, thedeviation error after compensation is approximately 30 MHz.However, one can notice a maximal error equal to 49.3 MHzafter compensation for mode 4 of tag 3.To prove that this correction technique is reliable in practice,

a measurement was made in the case of a tag put on a box full ofpaper, as shown in Fig. 13(b). The measured response is shownin Fig. 21 for tag 1. Before correction, one can see a large fre-quency shift for all of the resonant peaks as well as a slight atten-uation. After having applied the correction technique accordingto (9), the resonant peaks of the curve obtained with the box are


TABLE VFREQUENCY SHIFT MEASURED AND CORRECTED IN MEGAHERTZ FOR TAG 3

Fig. 21. Measured response for tag 1 put on a box full of paper ac-cording to Fig. 15(b) before and after correction using (9).

close to those obtained in the free space case. The residual fre-quency shifts for resonators 1 to 4 are, respectively, 6, 11, 0, and34 MHz after compensation, so that no decoding error is made.As a conclusion, this correction technique is a way of operatingthis tag even if it is put on a container of unknown permittivity.Indeed, it was proved that the influence of materials with lowpermittivity value can be compensated by this technique. How-ever, the frequency resolution, which is linked to the coding ca-pacity of the tag, is degraded because of the residual errors thatare still present after the compensation. Therefore, dependingon the environment parameters, the performance of the designwill change, and as a general rule, more reliability means lowercoding capacity.

V. PERFORMANCES OF THE DESIGN

To compare with previous designs, a figure of merit is neededto represent each density of coding per surface (DPS) in b/cmas a function of density of coding per frequency (DPF) in b/GHz,as shown in Fig. 22. The design proposed by Preradovic et al.[6] embeds 35 b for a size of 88 65 cm making a density ofcoding of 0.61 b/cm . Jalaly et al. [7] have designed a 5-b tag ina surface of 18 35 mm giving 0.81 b/cm . While Balbin etal. [9] proposed a tag encoding 3 b for a size of 120 50 mm ,giving a DPS of 0.05 b/cm . To finish, the design proposed byMcVay et al. [8], encodes 5 b within 200 30 mm giving aDPS of 0.083 b/cm , and a DPF of 25 b/GHz. In this new de-sign, it can be seen that DPS is largely improved with a value of3 b/cm , giving a high degree of miniaturization. This is a max-imum value reachable in a practical case when the residual error

Fig. 22. DPS as a function of DPF of several chipless tag designs found inliterature.

Fig. 23. Calculated capacity of coding in bits as a function of the number ofresonators according to (4). The resolution frequency is chosen to be 50 MHz.

after compensation is lower than 25 MHz for resonators 1 and 2and 50MHz for resonator 4. On the other hand, the DPF was de-graded to 4 b/GHz compared with the other designs. A tradeoffhas to be found between coding efficiency for a given size andbandwidth. In future work, in order to increase the capacity ofcoding, a tag will be designed with a larger bandwidth and moreresonators. For example, if it is defined for a maximum fre-quency of 9 GHz, the minimum frequency will be 9/3 3 GHzin order to avoid overlap between the fundamental mode andthe higher order mode. That gives a total bandwidth of 6 GHz.Keeping the same parameters as for the actual tag, with a size of55 32.7 mm , which is smaller than a credit card format, thetag can have resonators and for each with a bandwidthof 545 MHz. With (4), one can find a capacity of coding of 36b for a frequency resolution 50 MHz.Now, independent of the presented design, if plotting (see

Fig. 23) (4) for a frequency resolution of 50 MHz and a totalbandwidth corresponding to the UWB frequency span, the max-imal reachable coding capacity can be deduced. Some negativeslopes are presented in Fig. 23, when the bandwidth is not amultiple of the resolution frequency. With this technique it canbe highlighted that 60 b are reachable with only 20 resonatorswithin a frequency span between 3.5–10.5 GHz. To go further, apossible way to increase the capacity could be the hybrid codingtechnique as described in [23] where the frequency shift and thephase deviation are used independently to make a code.

VI. CONCLUSION

In this paper, we introduced a new design for a chipless tag.It has a reduced size and can embed a large data capacity that is


currently close to 10 b. To our knowledge, this coding density isthe largest published until now. In terms of capacity, the actualcoding technique can reach 36 b on a smaller surface than thatof a standard credit-card format. In addition to its reduced size,the cost of fabrication for this tag is very low; mainly becauseof the absence of a ground plane, it is potentially fully print-able with conductive inks. Indeed, printing techniques such asflexography on paper or inkjet printing on plastic substrate canbe used for this purpose. Based on our current experiments ona similar design, we found that, with flexography, a unit costless than one cent is already achievable, and the measured per-formances were quite similar. The resonant peaks are slightlymitigated and enlarged.Configurability of the tag using short-circuit parts has been

demonstrated and allows significant advantage when imple-mented in practice. Indeed, it is possible to produce initially alarge number of “virgin” tags and then to configure them justby adding some conductive bridges as shorts.Many experimental measurements have been conducted in an

anechoic chamber and an open space. The measurements havevalidated the design in the ideal case with no container. To makea proof of concept in a realistic case, a study has been con-ducted to evaluate the performance of the design in a real envi-ronment having various parameters. Indeed, a technique basedon a sensing resonator was established to compensate the devi-ation effect due to the container proximity. Its implementationallowed successful decoding of the tag’s identifier for variouscontainers. So, this confirms that this technology is reliable.A detection range of 65 cm has been demonstrated under

0-dBm excitation without any amplification. However, prac-tically, this performance makes sense only if addressing spe-cific applications that are not necessarily FCC- and ECC-com-pliant. In order to address the huge majority of the RFID ap-plications, the emitting power has to be very low and must fitthe requirements for UWB communications within the range3.1–10.6 GHz. To do so, a possible solution consists in de-signing a reader based on impulse radio sending short pulse oftens of picoseconds wide, with a pulse repetition frequency aslow as 1 MHz. We are currently working on the design of sucha reading system, and some results will be presented in a futurecommunication.

ACKNOWLEDGMENT

The authors would like to thank Prof. L. Duvillaret andDr. F. Garet for guidance and fruitful discussions on a part ofthis work.

REFERENCES

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[6] S. Preradovic, I. Balbin, N. C. Karmakar, and G. F. Swiegers, “Mul-tiresonator-based chipless RFID system for low-cost item tracking,”IEEE Trans. Microw. Theory Tech., vol. 57, no. 5, pp. 1411–1419, May2009.

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Arnaud Vena received the Eng. Dipl. degreein electrical engineering from the Institut Na-tional Polytechnique de Grenoble (Grenoble-INP),Grenoble, France, in 2005, where he is currentlyworking toward the Ph.D. degree.From 2005 to 2009, he was an R&D Engineer with

ACS Solution France SAS. He was in charge of de-velopment of RFID contactless card readers and con-tributed to evolution of the ISO/IEC 14443 regula-tion. In October 2009, he began his research with theLaboratoire de Conception et d’Intégration des Sys-

témes (LCIS)/Grenoble Institute of Technology, mainly focused on design ofchipless RFID systems.


Etienne Perret (M’05) was born in Albertville,Savoie, France, on October 30, 1979. He receivedthe Eng. Dipl. degree from the Ecole NationaleSupérieure d’Electronique, d’Electrotechnique,d’Informatique, d’Hydraulique et des Télécommu-nications, Toulouse, France, in 2002, and the M.Sc.and Ph.D. degrees from the Toulouse Institute ofTechnology, France, in 2002 and 2005, respectively,all in electrical engineering.From 2005 to 2006, he held a postdoctoral position

with the Institute of Fundamental Electronics (IEF),Orsay, France. His research activities cover the electromagnetic modeling ofpassive devices for millimeter and submillimeter-wave applications. His currentresearch interests are in the field of wireless communications, especially radiofrequency identification (RFID) with the design and development of antennasfor RFID tags. Since September 2006, he has been an Assistant Professor ofelectronics with the Laboratoire de Conception et d’Intégration des Systémes(LCIS)/Grenoble Institute of Technology. He has authored and coauthored morethan 50 technical conferences, letters, and journal papers and one book chapter.Dr. Perret is a Technical Program Committee member of IEEE-RFID. He

was keynote speaker and the Chairman of the 11th Mediterranean MicrowaveSymposiumMMS’2011. He also served as the Co-Chair of the 29th PIERS 2011in Marrakesh, Morocco.

Smail Tedjini (SM’92) received the Ph.D. degree inphysics from Grenoble University, Grenoble, France,in 1985.He was an Assistant Professor from 1981 to 1986

and a Senior Researcher for the CNRS from 1986to 1993. He became a University Professor in 1993,and, since 1996, he is a Professor with the ESISARDepartment, Institut National Polytechnique deGrenoble (Grenoble-INP), Grenoble, France. Hisspecialization topics concern electromagnetism,RF, wireless and optoelectronics. He serves as a

Coordinator and Staff Member in numerous academic programs both foreducation and research. In 2006 to 2007, he served as the Director of ESISAR,Embedded Systems Department, Grenoble-INP. He is involved in academicresearch supervision since 1982. He is the Founder and past Director of theLCIS Lab. Now, he is the ORSYS Group Leader. His main topics in researchare applied electromagnetism, modeling of devices and circuits at both RFand optoelectronic domains. Current research concerns wireless systems withspecific attention to RFID and nanoRF. He supervised 27 Ph.D. candidates andhe has authored or coauthored more than 250 papers.Dr. Tedjini is a member of several TPC and serves as expert/reviewer for na-

tional and international scientific committees and conferences including ISO,Piers, IEEE, URSI, ISO, ANR, OSEO, and FNQRT. He organized several con-ferences/workshops. He is President and founder of the IEEE-CPMT FrenchChapter, Vice-President of the IEEE Section France, and elected as the Vice-Chair of URSI Commission D “Electronics & Photonics” in 2008. He was re-elected as Vice-Chair of the IEEE France section and will serve as the Chair ofURSI Commission “D3” for the triennium 2011–2014.

Documents

Design of Compact and Auto-Compensated Single-Layer Chipless RFID Tag