78
© Vishal Saxena -1- Delta-Sigma Analog-to-Digital Converters Oversampling Data Converters Basics Vishal Saxena

Delta-Sigma Analog-to-Digital Converters Notes/Delta...© Vishal Saxena-3- Oversampling y v f s 2 f s OSR L PF OSR Dec i m at i o n D o u t @ f s @ f s /OSR 0 50 100 150 200 250 300-8-6-4-2

  • Upload
    others

  • View
    1

  • Download
    0

Embed Size (px)

Citation preview

  • © Vishal Saxena -1-

    Delta-Sigma Analog-to-Digital

    Converters

    Oversampling Data Converters Basics

    Vishal Saxena

  • © Vishal Saxena -2-

    Oversampling and Noise-Shaping

  • © Vishal Saxena -3-

    Oversampling

    yv

    fs

    2

    sf

    OSR

    LPF

    OSR

    Decimation

    Dout@fs @fs/OSR

    0 50 100 150 200 250 300-8

    -6

    -4

    -2

    0

    2

    4

    6

    8

    0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45 0.5-80

    -70

    -60

    -50

    -40

    -30

    -20

    -10

    0

    10

    e2=0.29136

    V(f)

    Oversampling ratio (OSR)

    Conversion bandwidth: fB = fs/2·OSR

    SQNR = 6.02·N + 1.76 + 10·log10(OSR)

    0.5 bits increase in resolution per doubling in OSR

  • © Vishal Saxena -4-© Vishal Saxena

    Oversampling

    0 50 100 150 200 250 300 350 400 450 500-20

    -10

    0

    10

    20Oversampling: Time-domain waveforms

    u[n]

    v[n]

    vf[n]

    0 50 100 150 200 250 300 350 400 450 500-1

    -0.5

    0

    0.5

    1

    e[n]

    ef[n]

    File: oversampling1.m

  • © Vishal Saxena -5-© Vishal Saxena

    Oversampling

    0 0.1 0.2 0.3 0.4-80

    -70

    -60

    -50

    -40

    -30

    -20

    -10

    0

    10Quantization Noise Spectrum

    e2=0.30483

    V(f)

    0 0.1 0.2 0.3 0.4-80

    -70

    -60

    -50

    -40

    -30

    -20

    -10

    0

    10Filtered Quantization Noise Spectrum

    q2=0.0082827

    Vf(f)

    File: oversampling1.m

  • © Vishal Saxena -6-

    Oversampling with Feedback

    Can we use feedback with high loop-gain (A·kq) to reduce the error

    e=|u-v| ?

    Since quantizer output can not be equal to the input

    The loop will be unstable as the error gets unbounded

    Ay v

    u

    fs

    High-gain

    error

    qA k e u v

  • © Vishal Saxena -7-

    Oversampling with Feedback

    Use large loop-gain in the signal band and small loop-gain at higher frequencies

    At low frequencies

    At high frequencies, low loop-gain stabilizes the loop

    L(z) is the loop-filter

    This feedback arrangement is called a (noise) modulator

    0e u v

    vu

    fs

    error2

    sf

    OSR

    L(z)

    y

  • © Vishal Saxena -8-

    First-order Modulator

    Differencing (Δ) followed by an accumulator (Σ) ΔΣ modulator

    At low frequencies 0e u v

    y vu

    fs1

    1-z-1

    z-1

    Δ

    Σ

  • © Vishal Saxena -9-

    First-order Noise Shaping

    Linearized model for the modulator

    Noise transfer function (NTF) (1-z-1) : first-order differentiator

    High-pass shaping of quantization noise

    Signal transfer function (STF) Unit delay

    y vu

    fs1

    1-z-1

    z-1

    e[n]

    1 1( ) 1 ( )V z z U z z E z STF NTF

  • © Vishal Saxena -10-

    First-order ΔΣ Modulator

    y vu

    fs1

    1-z-1

    z-1

    e[n]

    Sv(f)v[n]

    0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45 0.5-140

    -120

    -100

    -80

    -60

    -40

    -20

    0DSM Output Spectrum

    0 100 200 300 400 500 600 700 800 900 1000-8

    -6

    -4

    -2

    0

    2

    4

    6

    8DSM time-domain Simulation

    File: First_Order_DSM.m

  • © Vishal Saxena -11-

    First-order ΔΣ Modulator SQNR

    In-band quantization noise (IBN)

    SQNR = 6.02·N –3.4 + 30·log10(OSR)

    1.5 bits increase in resolution per doubling in OSR

    Out-of-band noise is filtered out using a digital filter

    0

    2

    0

    22

    0

    22

    0

    2 3

    0

    2 2

    3

    ( )

    ( )12

    4sin / 212

    12

    12 3

    36

    OSRj

    v

    OSRj

    OSR

    OSR

    OSR

    IBN S e d

    NTF e d

    d

    d

    OSR

    2 23

    12 3OSR

    ππ

    OSR

    |NTF(ejω

    )|

    0

    2

  • © Vishal Saxena -12-

    Delta-Sigma (ΔΣ) ADC

    Use oversampling (fs=2·OSR·BW) to shape the quantization noise out of the signal band

    Use low-resolution ADC and DAC to higher much higher resolution

    Digitally filter out the out-of band shaped (modulated) noise

    Trades-off SQNR with oversampling ratio (OSR)

    +

    vinL(z)

    DAC

    ADCvDSM Decimation

    Filter

    vout

    ΔΣ Modulator

    E(z)

    f

    STF

    NTF•E

    E

    fs/2•OSR fs/2

    |VDSM(f)|vin

    ffs/2

    |Vout(f)|vin

    fs/2•OSR

  • © Vishal Saxena -13-

    Second-Order Noise Shaping

    2

    1 1( ) 1 ( )V z z U z z E z

    STF NTF

    Linearized model for the modulator

    Second-order noise-shaping

    In-band quantization noise (IBN):

    2.5 bits increase in resolution per doubling in OSR

    Can be extended to higher orders

    2 45

    12 5OSR

    v1

    1-z-1

    z-1

    u 1

    1-z-1

    y

    e[n]

  • © Vishal Saxena -14-

    Second-order ΔΣ Modulator

    v1

    1-z-1

    z-1

    u 1

    1-z-1

    y

    e[n]

    Sv(f)v[n]

    0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45 0.5-180

    -160

    -140

    -120

    -100

    -80

    -60

    -40

    -20

    0

    20

    0 100 200 300 400 500 600 700 800 900 1000-8

    -6

    -4

    -2

    0

    2

    4

    6

    8

    File: Second_Order_DSM.m

  • © Vishal Saxena -15-

    -1 -0.5 0 0.5 1-1

    -0.8

    -0.6

    -0.4

    -0.2

    0

    0.2

    0.4

    0.6

    0.8

    1

    X2

    NTF(z)

    NTF

    zeroes

    Second-order ΔΣ Modulator

    10-4

    10-3

    10-2

    10-1

    -180

    -160

    -140

    -120

    -100

    -80

    -60

    -40

    -20

    0

    SNDR = 73.8 dB

    ENOB = 11.97 bits

    @OSR = 32

    Frequency

    dB

    Second-Order KD1S Output Spectrum

    40 dB/dec

    Sv(f)

    NTF(z) = (1-z-1)2

    Two zeroes at DC

    Out-of-Band Gain (OBG) (i.e gain at ω≈π) = 4

  • © Vishal Saxena -16-© Vishal Saxena

    2nd order DSM

    -1 -0.5 0 0.5 1-1

    -0.8

    -0.6

    -0.4

    -0.2

    0

    0.2

    0.4

    0.6

    0.8

    1

    0 50 100 150 200 250 300 350 400 450 500-8

    -6

    -4

    -2

    0

    2

    4

    6

    8

    v

    u

    File: Second_Order_DSM_Zero_Opt.m

    Set variable opt=0.

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9-0.12

    0.304

    0.728

    1.152

    1.576

    2

    2.424

    2.848

    3.272

    3.696

    4.12

    Normalized Frequency ( rad/sample)

    Magnitu

    de

    Magnitude Response

  • © Vishal Saxena -17-© Vishal Saxena

    2nd order DSM: contd.

    10-4

    10-3

    10-2

    10-1

    -180

    -160

    -140

    -120

    -100

    -80

    -60

    -40

    -20

    0

    SNDR = 73.8 dB

    ENOB = 11.97 bits

    @OSR = 32

    Frequency

    dB

    Second-Order KD1S Output Spectrum

  • © Vishal Saxena -18-© Vishal Saxena

    Comparison: 1st and 2nd order modulator waveforms

    60 80 100 120 140 160 180 200 220 240

    -6

    -4

    -2

    0

    2

    4

    6

    DSM time-domain Simulation

    860 880 900 920 940 960 980

    -6

    -4

    -2

    0

    2

    4

    6

    NTF(z) = (1-z-1)

    OBG = 2

    Max LSB jump = 1

    NTF(z) = (1-z-1)2

    OBG = 4

    Max LSB jump = 3

  • © Vishal Saxena -19-

    Higher-Order ΔΣ Modulators

  • © Vishal Saxena -20-

    Higher-Order NTFs

    Higher order noise shaping Reduced in-band noise, higher SQNR

    For NTF=(1-z-1)-N, in-band noise (IBN):

    Ideally (N+1/2) bits increase in resolution per doubling in OSR

    y vu L(z)

    ( ) 1

    ( ) ( )1 ( ) 1 ( )

    L zV z U z E z

    L z L z

    STF NTF

    2 2(2 1)

    12 2 1

    NNOSR

    N

  • © Vishal Saxena -21-

    Higher-Order NTFs

    NTF gain increases at high frequencies (around ω≈π)

    Can we go on increasing the order?

    0.5 1 1.5 2 2.5 3-200

    -150

    -100

    -50

    0

    50

    |NT

    F(e

    j )|

    Bode Diagram

    Frequency (rad/s)

  • © Vishal Saxena -22-

    Third-order ΔΣ Modulator Example

    NTF(z) = (1-z-1)3

    OBG = 8, Full-scale input.

    Unstable after few samples (look at quantize input (y) blowing up!). Signature for ΔΣ instability

    Worst case for a single-bit quantizer.

    0 100 200 300 400 500 600 700 800 900 1000-10

    -5

    0

    5

    10DSM time-domain Simulation

    u

    v

    0 100 200 300 400 500 600 700 800 900 1000-4

    -2

    0

    2

    4x 10

    7

    y

    File: Third_Order_DSM.m

  • © Vishal Saxena -23-

    Third-order ΔΣ Modulator Example

    Stable for 50% of full-Scale amplitude

    Signal dependent stability Need to develop intuition for modulator stability

    Reference: Stability theory from the Yellow Bible of delta-sigma

    0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45

    -180

    -160

    -140

    -120

    -100

    -80

    -60

    -40

    -20

    0

    20

    DSM Output Spectrum

    0 100 200 300 400 500 600 700 800 900 1000-10

    -5

    0

    5

    10DSM time-domain Simulation

    u

    v

    0 100 200 300 400 500 600 700 800 900 1000-10

    -5

    0

    5

    10

    y

    File: Third_Order_DSM.m

  • © Vishal Saxena -24-

    Systematic NTF Design

    NTFs of the form (1-z-1)N have stability issues The OBG (2N) are too high

    A larger OBG causes more wiggling at the quantizer input This saturates the quantizer for even smaller inputs

    Irrecoverable quantizer saturation causes loop instability

    For high-OBG the maximum stable (input) amplitude (MSA) is small

    The stability is worse for low quantizer resolutions

    Thus we need to reduce OBG while maintaining high in-band noise shaping

  • © Vishal Saxena -25-

    Systematic NTF Design Procedure

    Introduce poles into the NTF

    NTF realizability criterion No delay-free loops in the modulator

    • First sample of the NTF impulse response (i.e. h[0])=1

    NTF()=1

    D(z=)=1

    Commonly used pole positions: Butterworth, Inverse Chebyshev and maximally flat poles (maxflat)

    1(1 )( )

    ( )

    NzNTF z

    D z

    -1 -0.8 -0.6 -0.4 -0.2 0 0.2 0.4 0.6 0.8 1-1

    -0.8

    -0.6

    -0.4

    -0.2

    0

    0.2

    0.4

    0.6

    0.8

    1Pole-Zero Map

    Real Axis

    Imagin

    ary

    Axis

    X3

  • © Vishal Saxena -26-

    NTF Response with Poles

    Select appropriate OBG for the NTF to assure stability

    Trade-off between stability and increased in-band noise MSA vs SQNR for a given order and quantizer resolution

    0 50 100 150 200 250 300 350 400 450 500-10

    -5

    0

    5

    10OBG=8

    u

    v

    0 50 100 150 200 250 300 350 400 450 500-10

    -5

    0

    5

    10OBG=3

    u

    v

    0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45 0.50

    1

    2

    3

    4

    5

    6

    7

    8

    9

    /

    |NT

    F(e

    j )|

    NTF response

  • © Vishal Saxena -27-

    Systematic NTF Design Example

    Specifications SQNR > 120 dB

    A signal bandwidth which results in an OSR = 64

    • Study optimal clock rate for the given process and quantizer design.

    Designer’s Choice Order = 3

    Quantizer levels (nLev) = 16

    Butterworth high-pass response for the NTF

    Use MATLAB for finding coefficients of the HPF response. [b,a] = butter(order, ω3dB, ‘high’) The cutoff frequency ω3dB specifies the transfer function.

  • © Vishal Saxena -28-

    Systematic NTF Design contd.

    Start with cutoff frequency ω3dB=π/8, for the butterworth HPF H(z).

    Derive a realizable NTF using NTF(z)=H(z)/b0

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 10

    0.5

    1

    1.5

    /

    Magnitude

    H(z)

    NTF(z)

  • © Vishal Saxena -29-

    Systematic NTF Design contd.

    Map the NTF response to a loop-filter architecture (details later)

    Simulate the modulator for all possible amplitudes and input tone frequencies.

    Compute the peak SQNR and MSA. simulateDSM function in the toolbox. Can use Risbo’s method shown later

  • © Vishal Saxena -30-

    Systematic NTF Design contd.

    Peak SNR = 107 dB

    MSA = 0.9

    -100 -80 -60 -40 -20 00

    10

    20

    30

    40

    50

    60

    70

    80

    90

    100

    110

    120

    130

    140

    Input Level, dB

    SN

    R d

    B

    peak SNR = 107.2dB

  • © Vishal Saxena -31-

    Systematic NTF Design contd.

    If SNR is not enough, repeat the entire procedure with a higher cutoff frequency for the Butterworth HPF IBN ↓, SQNR ↑

    OBG ↑ and MSA ↓

    If SNR is too high, repeat the entire procedure with a lower cutoff frequency for the Butterworth HPF IBN ↑, SQNR ↓

    OBG ↓ and MSA ↑

  • © Vishal Saxena -32-

    Systematic NTF Design contd.

    ω3dB=π/4.

    Peak SNR = 119 dB, OBG = 2.25, MSA = 0.8

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 10

    0.5

    1

    1.5

    2

    2.5

    /

    Magnitude

    H(z)

    NTF(z)

    -100 -80 -60 -40 -20 00

    10

    20

    30

    40

    50

    60

    70

    80

    90

    100

    110

    120

    130

    140

    Input Level, dB

    SN

    R d

    B

    peak SNR = 119.7dB

  • © Vishal Saxena -33-

    Systematic NTF Design contd.

    ω3dB=2π/7.

    Peak SNR = 121 dB, OBG = 2.54, MSA = 0.8. Design closed !

    -100 -80 -60 -40 -20 00

    10

    20

    30

    40

    50

    60

    70

    80

    90

    100

    110

    120

    130

    140

    Input Level, dB

    SN

    R d

    B

    peak SNR = 121.1dB

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 10

    0.5

    1

    1.5

    2

    2.5

    3

    /

    Magnitude

    H(z)

    NTF(z)

  • © Vishal Saxena -34-

    Systematic NTF Design contd.

    An advanced version of this iterative process is implemented as the function synthesizeNTF in the delta-sigma Toolbox. Several ‘opt’ params for NTF zero (and pole) optimization

    Use synthesizeChebyshevNTF for low OSR and low OBG designs.

  • © Vishal Saxena -35-

    NTF-Zero Optimization

    Spread zeros in the signal band to minimize in-band noise Complex zeros on the unit circle

    8dB increase in SQNR for 3rd order modulator

    Bandwidth normalized NTF-zero locations obtained by toolbox function ds_optzeros(order, 1)

    Already implemented in synthesizeNTF function for opt=1

    10-3

    10-2

    10-1

    -120

    -100

    -80

    -60

    -40

    -20

    0

    20

    /

    |NT

    F(e

    j )|

    NTF response

    -1 -0.8 -0.6 -0.4 -0.2 0 0.2 0.4 0.6 0.8 1-1

    -0.8

    -0.6

    -0.4

    -0.2

    0

    0.2

    0.4

    0.6

    0.8

    1Pole-Zero Map

    Real Axis

    Imagin

    ary

    Axis

  • © Vishal Saxena -36-© Vishal Saxena

    2nd order DSM: NTF Zero Optimization

    File: Second_Order_DSM_Zero_Opt.m

    Set variable opt=1.

    -1 -0.5 0 0.5 1-1

    -0.8

    -0.6

    -0.4

    -0.2

    0

    0.2

    0.4

    0.6

    0.8

    1

    0 50 100 150 200 250 300 350 400 450 500-8

    -6

    -4

    -2

    0

    2

    4

    6

    8

    v

    u

  • © Vishal Saxena -37-© Vishal Saxena

    2nd order DSM: NTF Zero Optimization contd.

    10-4

    10-3

    10-2

    10-1

    -160

    -140

    -120

    -100

    -80

    -60

    -40

    -20

    0

    SNDR = 79.3 dB

    ENOB = 12.89 bits

    @OSR = 32

    Frequency

    dB

    Second-Order KD1S Output Spectrum

    •5.5 dB increase in SQNR.

    NTF pole (if any) optimization to be discussed later.

  • © Vishal Saxena -38-

    Estimating MSA (Maximum Stable Amplitude)

    MSA is found through extensive simulation

    Simulate for input sinusoids of varying amplitudes for all possible signal frequencies in the signal band. For every input amplitude compute in-band SNR.

    Beyond the MSA, the NTF poles move out of the unit circle.

    Noise shaping is disrupted and the in-band SNR drops.

    At this point the quantizer input (y[n]) blows up.

    simulateSNR function in the toolbox does exactly the same

    Time consuming and often impractical for iterative design

  • © Vishal Saxena -39-

    Estimating MSA using Risbo’s Method

    Use a slow ramp input from 0 to FS value. Plot log10|y[n]|. Observe where this plot blows up.

    Take 90% of the input amplitude where log10|y[n]| blows up as a conservative estimate for MSA.

    Estimated MSA is close to that predicted by the sinewave input method.

    Much quicker than the sinewave technique (simulateSNR function)

    y v

    u

    L

    t0

    FS

    Slow ramp input

  • © Vishal Saxena -40-

    Estimating MSA using Risbo’s Method

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-6

    -4

    -2

    0

    2

    4

    6

    8

    10

    12

    14

    log |y|

    Input, u

    MSA = 0.843

    MSA Estimation by Simulation

    File: MSA_Risbo_Method.m

  • © Vishal Saxena -41-© Vishal Saxena

    Simulation with input with MSA

    0 100 200 300 400 500 600 700 800-20

    -10

    0

    10

    20DSM time-domain Simulation

    u

    v

    0 100 200 300 400 500 600 700 800-20

    -10

    0

    10

    20Loop-filter States

    x1

    x2

    y=x3

    File: MSA_Risbo_Method.m

  • © Vishal Saxena -42-© Vishal Saxena

    Simulated SNR with input with MSA

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-250

    -200

    -150

    -100

    -50

    0

    SNDR = 119.3 dB

    ENOB = 19.52 bits

    @OSR = 64

    /

    dB

    FS

    Modulator Output Spectrum

    File: MSA_Risbo_Method.m

  • © Vishal Saxena -43-© Vishal Saxena

    Simulation with input with 1.2*MSA

    0 100 200 300 400 500 600 700 800-20

    -10

    0

    10

    20DSM time-domain Simulation

    u

    v

    0 100 200 300 400 500 600 700 800-50

    0

    50Loop-filter States

    x1

    x2

    y=x3

    File: MSA_Risbo_Method.m

  • © Vishal Saxena -44-© Vishal Saxena

    Simulation with input with 1.2*MSA

    0 100 200 300 400 500 600 700 800-20

    -10

    0

    10

    20DSM time-domain Simulation

    u

    v

    0 100 200 300 400 500 600 700 800-2

    -1

    0

    1

    2x 10

    6 Loop-filter States

    x1

    x2

    y=x3

    File: MSA_Risbo_Method.m

  • © Vishal Saxena -45-© Vishal Saxena

    Bode Sensitivity Integral

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-0.4

    -0.3

    -0.2

    -0.1

    0

    0.1

    0.2

    0.3

    /

    log|H

    |

    C1=0.083472, C

    2=0.083569

    File: BodeSensitivity1.m

    Single pole/zero transfer function with pole/zero inside the unit circle.

    Area above and below the 0-dB axis are equal.

  • © Vishal Saxena -46-© Vishal Saxena

    Bode Sensitivity Integral

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-160

    -140

    -120

    -100

    -80

    -60

    -40

    -20

    0

    20

    /

    10 log|N

    TF

    |

    C1=3.27, C

    2=3.367

    File: BodeSensitivity2.m

    Butterworth NTF.

    Area above and below the 0-dB axis are equal.

  • © Vishal Saxena -47-© Vishal Saxena

    Bode Sensitivity Integral

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-70

    -60

    -50

    -40

    -30

    -20

    -10

    0

    10

    20

    /

    10 log|H

    |

    C1=3.949, C

    2=4.0776

    File: BodeSensitivity3.m

    Inverse Chebyshev NTF.

    Area above and below the 0-dB axis are equal.

  • © Vishal Saxena -48-© Vishal Saxena

    Bode Sensitivity Integral

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-70

    -60

    -50

    -40

    -30

    -20

    -10

    0

    10

    20

    /

    10*l

    og|H

    |

    File: BodeSensitivity4.m

    Better in-band performance results in worse out-of-band performance.

  • © Vishal Saxena -49-© Vishal Saxena

    Bode Sensitivity Integral

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-70

    -60

    -50

    -40

    -30

    -20

    -10

    0

    10

    20

    /

    10*l

    og|H

    |

    File: BodeSensitivity5.m

    Complex NTF zeros result in better in-band performance for the same OBG.

  • © Vishal Saxena -50-© Vishal Saxena

    Bode Sensitivity Integral

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-70

    -60

    -50

    -40

    -30

    -20

    -10

    0

    10

    20

    /

    10*l

    og|H

    |

    File: BodeSensitivity6.m

    Higher-order NTF results in better in-band performance for the same OBG.

  • © Vishal Saxena -51-

    Loop Filter Architectures

  • © Vishal Saxena -52-

    Loop-Filter Architectures

    Several loop-filter discrete-time architectures possible

    Toolbox function realizeNTF maps the synthesized NTF to loop-filter co-efficients[a,g,b,c] = realizeNTF(H, form);

    Dynamic Range Scaling (DRS) performed to scale loop-filter states to a bounded value Scaling performed using ABCD matrix representation of the loop-

    filter

    See any introductory text on Linear SystemsABCD = stuffABCD(a,g,b,c,form);

    [ABCDs umax] = scaleABCD(ABCD, nLev, f0, xLim);

    [a,g,b,c] = mapABCD(ABCDs, form);

  • © Vishal Saxena -53-

    CIFB (Cascade of Integrators with Distributed Feedback)

    Cascade of delaying integrators: Feedback coefficients a’s realize the zeros of L1 and thus the

    NTF and STF poles.

    Feed-in coefficients b’s determine zeros of L0 and thus the STF zeros.

    State scaling coefficients c’s are used for dynamic range scaling.

    -g1

    c1

    -a1 -a2

    x1(n) x2(n) y(n)c2

    z-1

    1-z-1

    z-1

    1-z-1

    b2b1

    DAC

    v(n)

    -a3

    x3(n)c2

    z-1

    1-z-1

    b3

    -a4

    b4

    u(n)

  • © Vishal Saxena -54-

    CRFB (Cascade of Resonators with Distributed Feedback)

    Combine a non-delaying and a delaying integrator with local feedback around them, to form a stable resonator Local feedback coefficients g’s realize the complex zeros in the

    NTF.

    Implements NTF with complex zeros.

    For odd-order, use an integrator in the front to avoid noise coupling due to g

    -g1

    c1

    -a1 -a2

    x1(n) x2(n+1) y(n)c2

    z-1

    1-z-1

    1

    1-z-1

    b2b1

    DAC

    v(n)

    -a3

    x3(n)c2

    z-1

    1-z-1

    b3

    -a4

    b4

    u(n)

  • © Vishal Saxena -55-

    CIFF (Cascade of Integrators with Feed-Forward Summation)

    Feedforward summation of states

    For b1=bN+1=1 and b2 to bN=0, STF=1 Loop-filter only processes quantization noise, low power and

    distortion

    Feedforward loop-filters typically result in lower-power implementation

    DAC

    -g1

    c2

    -c1

    x1(n) x2(n) y(n)c3

    v(n)

    x3(n)a3

    a2

    a1

    z-1

    1-z-1

    z-1

    1-z-1

    z-1

    1-z-1

    b2b1 b3 b4

    u(n)

  • © Vishal Saxena -56-

    CRFF (Cascade of Resonators with Feed-Forward Summation)

    Use resonators with feedforward summation Implements NTF with complex zeros

    For odd-order, use an integrator in the front to avoid noise coupling due to g

    -g1

    c1

    -a1 -a2

    x1(n) x2(n) y(n)c2

    z-1

    1-z-1

    1

    1-z-1

    -b2-b1

    DAC

    v(n)

    -a3

    x3(n)c2

    z-1

    1-z-1

    -b3

    -a4

    -b4

    u(n)

  • © Vishal Saxena -57-© Vishal Saxena

    CIFB Example 1

    CIFB, order = 4

    All NTF zeros at z=1, i.e. opt =0.

    OBG = 3, OSR = 16, nLev = 15.

    Only single input coupling is used

    b(2:end) = 0

    Maxflat poles in STF

    a = [0.16 0.86 1.9 2.1]

    b = [0.16 0 0 0]

    c = [1 1 1 1]

    g = [0 0]

    File: CIFB_4th_Order_1.m

    -1 -0.5 0 0.5 1-1

    -0.5

    0

    0.5

    1NTF(z)

    -1 -0.5 0 0.5 1-1

    -0.5

    0

    0.5

    1STF(z)

    -1 -0.5 0 0.5 1-1

    -0.5

    0

    0.5

    1

    L0(z)

    -1 -0.5 0 0.5 1-1

    -0.5

    0

    0.5

    1

    L1(z)

  • © Vishal Saxena -58-© Vishal Saxena

    CIFB Example 1 contd. : NTF and STF

    File: CIFB_4th_Order_1.m

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 10

    0.5

    1

    1.5

    2

    2.5

    3

    /

    Mag

    NTF

    STF

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-200

    -150

    -100

    -50

    0

    50

    /

    Mag,

    dB

    NTF

    STF

  • © Vishal Saxena -59-© Vishal Saxena

    CIFB Example 1 contd. : Loop-Filter

    States

    File: CIFB_4th_Order_1.m

    0 100 200 300 400 500 600 700 800 900 1000-10

    -8

    -6

    -4

    -2

    0

    2

    4

    6

    8

    10Transient Simulation

    samples, n

    u

    v

    0 100 200 300 400 500 600 700 800 900 1000-20

    -15

    -10

    -5

    0

    5

    10

    15

    20Integrator States

    samples, n

    x1

    x2

    x3

    x4

  • © Vishal Saxena -60-© Vishal Saxena

    CIFB Example 1 contd. : Simulated

    Spectrum

    File: CIFB_4th_Order_1.m

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-200

    -180

    -160

    -140

    -120

    -100

    -80

    -60

    -40

    -20

    0

    SNDR = 80.7 dB

    ENOB = 13.11 bits

    @OSR = 16

    /

    dB

    FS

    Modulator Output Spectrum

  • © Vishal Saxena -61-© Vishal Saxena

    Other Examples of Feedback

    Topologies

    File: CIFB_4th_Order_2.m

    CRFB with single feed-in CRFB_4th_Order_1.m

    Low-distortion CRFB topology CRFB_4th_Order_2.m

    CIFB with single feed-in and optimized NTF zeros CIFB_Opt_4th_Order_1.m

    Low-distortion CIFB topology with optimized NTF zeros CIFB_Opt_4th_Order_2.m

  • © Vishal Saxena -62-© Vishal Saxena

    CIFF Example 1

    CIFF, order = 4

    All NTF zeros at z=1, i.e. opt =0.

    OBG = 3, OSR = 16, nLev = 15.

    Low-distortion topology

    b(1) =b(5)= 1

    b(2:4)=0

    a = [2.1 1.9 0.86 0.16]

    b = [1 0 0 0 1]

    c = [1 1 1 1]

    g = [0 0]

    File: CIFB_4th_Order_1.m

    -1 -0.5 0 0.5 1-1

    -0.5

    0

    0.5

    1NTF(z)

    -1 -0.5 0 0.5 1-1

    -0.5

    0

    0.5

    1STF(z)

    -1 -0.5 0 0.5 1-1

    -0.5

    0

    0.5

    1

    L0(z)

    -1 -0.5 0 0.5 1-1

    -0.5

    0

    0.5

    1

    L1(z)

  • © Vishal Saxena -63-© Vishal Saxena

    CIFF Example 1 contd. : NTF and STF

    File: CIFF_4th_Order_1.m

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 10

    0.5

    1

    1.5

    2

    2.5

    3

    /

    Mag

    NTF

    STF

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-200

    -150

    -100

    -50

    0

    50

    /

    Mag,

    dB

    NTF

    STF

  • © Vishal Saxena -64-© Vishal Saxena

    CIFF Example 1 contd. : Loop-Filter

    States

    File: CIFF_4th_Order_1.m

    0 100 200 300 400 500 600 700 800 900 1000-10

    -8

    -6

    -4

    -2

    0

    2

    4

    6

    8

    10Transient Simulation

    samples, n

    u

    v

    0 100 200 300 400 500 600 700 800 900 1000-6

    -4

    -2

    0

    2

    4

    6Integrator States

    samples, n

    x1

    x2

    x3

    x4

  • © Vishal Saxena -65-© Vishal Saxena

    CIFF Example 1 contd. : Simulated

    Spectrum

    File: CIFF_4th_Order_1.m

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-200

    -180

    -160

    -140

    -120

    -100

    -80

    -60

    -40

    -20

    0

    SNDR = 80.4 dB

    ENOB = 13.06 bits

    @OSR = 16

    /

    dB

    FS

    Modulator Output Spectrum

  • © Vishal Saxena -66-© Vishal Saxena

    CIFF Example 2

    CIFF, order = 4

    All NTF zeros at z=1, i.e. opt =0.

    OBG = 3, OSR = 16, nLev = 15.

    Only single input feed-in used

    b(2:end)=0

    a = [2.1 1.9 0.86 0.16]

    b = [1 0 0 0 0]

    c = [1 1 1 1]

    g = [0 0]

    File: CIFB_4th_Order_2.m

    -1 -0.5 0 0.5 1-1

    -0.5

    0

    0.5

    1NTF(z)

    -1 -0.5 0 0.5 1-1

    -0.5

    0

    0.5

    1STF(z)

    -1 -0.5 0 0.5 1-1

    -0.5

    0

    0.5

    1

    L0(z)

    -1 -0.5 0 0.5 1-1

    -0.5

    0

    0.5

    1

    L1(z)

  • © Vishal Saxena -67-© Vishal Saxena

    CIFF Example 2 contd. : NTF and STF

    File: CIFF_4th_Order_2.m

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 10

    0.5

    1

    1.5

    2

    2.5

    3

    3.5

    /

    Mag

    NTF

    STF

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-200

    -150

    -100

    -50

    0

    50

    /

    Mag,

    dB

    NTF

    STF

    Notice the significant STF peaking !

  • © Vishal Saxena -68-© Vishal Saxena

    CIFF Example 2 contd. : Loop-Filter

    States

    File: CIFF_4th_Order_2.m

    0 100 200 300 400 500 600 700 800 900 1000-10

    -8

    -6

    -4

    -2

    0

    2

    4

    6

    8

    10Transient Simulation

    samples, n

    u

    v

    0 100 200 300 400 500 600 700 800 900 1000-50

    -40

    -30

    -20

    -10

    0

    10

    20

    30

    40

    50Integrator States

    samples, n

    x1

    x2

    x3

    x4

    Last integrator output has significant signal content Use dynamic range scaling.

    Last integrator will burn more power in this case.

  • © Vishal Saxena -69-© Vishal Saxena

    CIFF Example 2 contd. : Simulated

    Spectrum

    File: CIFF_4th_Order_2.m

    0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-200

    -180

    -160

    -140

    -120

    -100

    -80

    -60

    -40

    -20

    0

    SNDR = 82.1 dB

    ENOB = 13.34 bits

    @OSR = 16

    /

    dB

    FS

    Modulator Output Spectrum

  • © Vishal Saxena -70-© Vishal Saxena

    Other Examples of Feed-forward

    Topologies

    Low-distortion CRFF topology CRFF_4th_Order_1.m

    CRFF with single feed-in CRFF_4th_Order_2.m

    Low-distortion CIFF topology with optimized NTF zeros CIFF_Opt_4th_Order_1.m

    CIFF with single feed-in and optimized NTF zeros CIFF_Opt_4th_Order_2.m

    STF peaking in FF topologies with single feed-in is an issue CT FF DSM will have STF peaking as full-feedforward branch can’t be

    used.

    The feed-in coefficients b’s can be strategically used to realize CIFF/CRFB topology with better out-of-band STF attenuation.

  • © Vishal Saxena -71-

    ΔΣ Modulator Architectures

    Cascade/ MASH architecture:

    •Eg. Two first order modulators are used to implement second order

    modulator.

    •Stability concerns are relaxed but mismatch in the two forward paths

    should be properly monitored.

  • © Vishal Saxena -72-

    ΔΣ Modulator Architectures

    Feedforward modulators

    •Most popular architecture.

    •Input signal is summed at Nth stage integrator output.

    •Summation block may be required at higher order modulators.

    •Multibit quantizer is necessary.

  • © Vishal Saxena -73-

    Key Terminologies :

    SQNR – Signal to quantization noise ratio Thermal/electrical noise are not included.

    SNR - Signal to noise ratio Distortion is not included.

    SDNR - Signal to noise and distortion ratio All noise sources are included.

    ENOB – Effective number of bits (resolution) This is very important than actual number of output bits

    Dynamic Range (DR) Measured with input of the modulator shorted.

    Harmonic Distortion THD is usually total harmonic distortion. Or Third??

    Spur Free Dynamic Range (SFDR) Very key parameter in communication systems

  • © Vishal Saxena -74-

    Frequency Domain Measurements

    Dynamic range

    Peak SNR

    Peak SNDR

  • © Vishal Saxena -75-

    Spurious (tone) Free Dynamic Range (SFDR)

    SFDR

  • © Vishal Saxena -76-

    References

    Data Conversion Fundamentals

    A.1 M. Gustavsson, J. Wikner, N. Tan, CMOS Data Converters for Communications, Kluwer Academic Publishers, 2000.

    A.2 B. Razavi, Principles of Data Conversion System Design, Wiley-IEEE Press, 1994.

    A.3 ADC and DAC Glossary by Maxim.

    A.4 B. Murmann, "ADC Performance Survey 1997-2009," [Online].

    A.5 S. Pavan, N. Krishnapura, EE658: Data Conversion Circuits Course at IIT Madras [Online].

    A.6 The Fundamentals of FFT-Based Signal Analysis and Measurement, T.I. App Note here.

    http://pdfserv.maxim-ic.com/en/an/AN641.pdfhttp://www.stanford.edu/~murmann/adcsurvey.htmlhttp://www.ee.iitm.ac.in/vlsi/courses/ee658_2009/starthttp://www.lumerink.com/courses/ECE697A/docs/The Fundamentals of FFT-Based Signal Analysis and Measurements.pdf

  • © Vishal Saxena -77-

    References

    Delta-Sigma Data ConvertersB.1 R. Schreier, G. C. Temes, Understanding Delta-Sigma Data Converters,

    Wiley-IEEE Press, 2005 (the Green Bible of Delta-Sigma Converters).B.2 S. R. Norsworthy, R. Schreier, G. C. Temes, Delta-Sigma Data Converters:

    Theory, Design, and Simulation, Wiley-IEEE Press, 1996 (the Yellow Bible of Delta-Sigma Converters).

    B.3 S. Pavan, N. Krishnapura, “Oversampling Analog to Digital Converters Tutorial,” 21st International Conference on VLSI Design, Hyderabad, Jan, 2008.

    B.4 S. H. Ardalan, J. J. Paulos, “An Analysis of Nonlinear behavior in Delta-Sigma Modulators,” IEEE TCAS, vol. 34, no. 6, June 1987.

    B.5 R. Schreier, “An Empirical Study of Higher-Order Single-Bit Delta-Sigma Modulators,” IEEE TCAS-II, vol. 40, no. 8, pp. 461-466, Aug. 1993.

    B.6 J. G. Kenney and L. R. Carley, “Design of multibit noise-shaping data converters,” Analog Integrated Circuits Signal Processing Journal, vol. 3, pp. 259-272, 1993.

    B.7 L. Risbo, “Delta-Sigma Modulators: Stability Analysis and Optimization,” Doctoral Dissertation, Technical University of Denmark, 1994 [Online].

    B.8 R. Schreier, J. Silva, J. Steensgaard, G. C. Temes,“Design-Oriented Estimation of Thermal Noise in Switched-Capacitor Circuits,” IEEE TCAS-I, vol. 52, no. 11, pp. 2358-2368, Nov. 2005.

    http://eivind.imm.dtu.dk/publications/PhD_thesis/LarsRisbo_PhD_Part1.ps.gz

  • © Vishal Saxena -78-

    References

    CAD for Mixed-Signal Design

    D.1 K. Kundert, “Principles of Top-Down Mixed-Signal Design,” Designer’s Guide Community [Online].

    D.2 R. Schreier, Matlab Delta-Sigma Toolbox, 2009 [Online], [Manual], [One page summary].

    D.3 Jose de la Rosa, “SIMSIDES Toolbox: An Interactive Tool for the Behavioral Simulation of Discrete-and Continuous-time SD Modulators in the MATLAB,” University of Sevilla, Spain, [Contact the authors for the software].

    D.4 P. Malcovati, Simulink Delta-Sigma Toolbox 2, 2009. Available [Online].

    Example Datasheets

    E.1 A 16-bit, 2.5MHz/5 MHz/10 MHz, 30 MSPS to 160 MSPS Dual Continuous Time Sigma-Delta ADC – AD9262, Analog Devices, 2008.

    E.2 A 24-bit, 192 kHz Multi-bit Audio ADC – CS5340, Cirrus Logic, 2008.

    http://www.designers-guide.org/Design/tdd-principles.pdfhttp://www.mathworks.com/matlabcentral/fileexchange/19http://www.lumerink.com/courses/ECE697A/docs/DSToolbox.pdfhttp://www.lumerink.com/courses/ECE697A/docs/OnePageStory.pdfhttp://www.mathworks.de/matlabcentral/fileexchange/25811-sdtoolbox-2http://www.analog.com/en/analog-to-digital-converters/ad-converters/ad9262/products/product.htmlhttp://www.cirrus.com/en/pubs/proDatasheet/CS5340_F2.pdf