DEERE AFFIDAVIT Attachment QQQ - AT&T® Official M. Anderson Ephraim Arnon James Aslanis Keith Atwell Hiromitsu Awai Jein Baek Scott J. Baer Rupert Baines H. Charles Baker LeRoy Baker

DEERE AFFIDAVITAttachment QQQ

ANSI®T1E1.4/99-002R4 (DRAFT T1.XXX-1999)

American National Standardfor Telecommunications

Spectrum Managementfor Loop Transmission Systems

Secretariat

Alliance for Telecommunications Industry Solutions

Approved <Date to be determined>

American National Standards Institute, Inc.

Abstract

This standard provides spectrum management requirements and recommendations for theadministration of services and technologies that use metallic subscriber loop cables. Spectrummanagement is the administration of the loop plant in a way that provides spectral compatibility forservices and technologies that use pairs in the same cable. In order to achieve spectral compatibility,energy that transfers into a loop pair, from services and transmission system technologies on otherpairs in the same cable, must not cause an unacceptable degradation of performance. In addition,energy in a particular loop pair must not transfer into other pairs in a manner that causes anunacceptable degradation in the performance of services and technologies on those pairs. Thisstandard includes signal power limits, technology deployment restrictions, and loop assignmentguidelines for certain digital subscriber line spectrum management classes. It also provides a genericanalytical method to determine spectral compatibility.

AmericanNationalStandard

Approval of an American National Standard requires verification by ANSI that therequirements for due process, consensus, and other criteria for approval have beenmet by the standards developer.

Consensus is established when, in the judgment of the ANSI Board of StandardsReview, substantial agreement has been reached by directly and materially affectedinterests. Substantial agreement means much more than a simple majority, but notnecessarily unanimity. Consensus requires that all views and objections beconsidered, and that a concerted effort be made toward their resolution.

The use of American National Standards is completely voluntary; their existence doesnot in any respect preclude anyone, whether he has approved the standards or not,from manufacturing, marketing, purchasing, or using products, processes, orprocedures not conforming to the standards.

The American National Standards Institute does not develop standards and will in nocircumstances give an interpretation of any American National Standard. Moreover,no person shall have the right or authority to issue an interpretation of an AmericanNational Standard in the name of the American National Standards Institute.Requests for interpretations should be addressed to the secretariat or sponsor whosename appears on the title page of this standard.

CAUTION NOTICE: This American National Standard may be revised or withdrawnat any time. The procedures of the American National Standards Institute require thataction be taken periodically to reaffirm, revise, or withdraw this standard. Purchasersof American National Standards may receive current information on all standards bycalling or writing the American National Standards Institute.

Published by

American National Standards Institute

11 West 42nd Street, New York, New York 10036

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ContentsPage

Foreword ................................................................................................................ v

1. Scope, purpose, and application........................................................................ 1

2. Normative references......................................................................................... 3

3. Definitions, abbreviations, acronyms, and symbols ........................................... 3

4. General Information ........................................................................................... 7

5. Signal power limits and other deployment restrictions ..................................... 13

6. Conformance testing methodology .................................................................. 18

Tables

1 Spectrum management class 1 PSD template definition............................ 23

2 Minimum transveres balance template for the xTU-C ............................... 23



5 PSD mask definition for downstream transmission from spectrummanagement class 4 TU-C ......................................................................... 25

6 PSD mask definition for upstream transmission from a spectrummanagement class 4 TU-R ......................................................................... 25


8 Termination impedances............................................................................. 25

9 Resolution bandwidth for measuring a DUT PSD for conformance withspectrum management classes 1, 2, 3, and 4. ........................................... 26

10 Resolution bandwidth for measuring a DUT PSD for conformance withspectrum management class 5................................................................... 26

11 Resolution bandwidth for measuring a DUT PSD for conformance withspectrum management class 6................................................................... 26

12 Summary of transverse balance testing criteria.......................................... 26

A.1 Code for DFE PAM SNR computation ........................................................ 45

A.2 Code for DFE QAM/CAP computation........................................................ 46

A.3 Code for DMT margin computation............................................................. 47

A.4 Data points for Unger NEXT model ............................................................ 48

A.5 Coefficients for T1 repeater input filtering gain equation ............................ 48

B.1 American wire gauge (AWG) and metric wire............................................. 73

B.2 Cable model parameters for TP1(0.4 mm or 26-gauge twisted pair) ......... 74

B.3 Primary constants for TP1 (0.4 mm or 26-gauge twisted pair) ................... 74

B.4 Cable parameters for 26-AWG PIC air core ............................................... 75

B.5 Cable parameters for 26-AWG filled PIC.................................................... 77

B.6 Cable model parameters for TP2 (0.5 mm or 24-gauge twisted pair) ........ 79

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B.7 Primary constants for TP2 (0.5 mm or 24-gauge twisted pair) ................... 79



B.10 Cable model parameters for TP3 (DW-10 reinforced .5 mm copperPVC-insulated steel strength member, polyethelene sheath)..................... 84

B.11 Primary constants for TP3 (DW-10 reinforced .5 mm copper PVC-insulated steel strength member, polyethelene sheath) ............................. 84

B.12 Cable model parameters for FP (1.14 mm flat cable)................................. 85

B.13 Primary constants for FP(1.14 mm flat cable) ............................................ 85

B.14 Cable model parameters for category 5 twisted pair .................................. 86

B.15 Primary constants for category 5 twisted pair ............................................. 86

B.16 Cable parameters, two-pair twisted drop .................................................... 87

B.17 Cable parameters, two-pair quad drop ....................................................... 89

B.18 Cable parameters, flat-pair drop ................................................................. 91

Figures

1 Configuration for evaluation of effect of NEXT and FEXT intodownstream................................................................................................. 27

2 Configuration for evaluation of effect of NEXT and FEXT into upstream ... 27

3 Spectrum management class 1 PSD template ........................................... 28



6 Spectrum management class 4 PSD mask for downstreamtransmission from spectrum management class 4 TU-C ........................... 29

7 Spectrum management class 4 PSD mask for upstream transmissionfrom spectrum management class 4 TU-R................................................. 30

8 Spectrum management class 7 template ................................................... 30

9 PSD and total average power measurement setup .................................... 31

10 Example PSD and total average power measurement setup ..................... 31

11 Illustrative test configuration for transverse balance conformancetesting.......................................................................................................... 32

A.1 Unger NEXT model and simplified NEXT model of 1% NEXT for 18 kftof 22GA PIC................................................................................................ 49

B.1 Loop ABCD parameters, impedance and voltages..................................... 93

B.2 Two-port network model ............................................................................. 94

B.3 Incremental section of twisted-pair transmission line.................................. 94

B.4 Simple load circuit for power analysis ......................................................... 94

B.5 Examples of two-port cascades for twisted-pair transmission lineconfigurations.............................................................................................. 95

B.6 Near end crosstalk (NEXT)......................................................................... 96

B.7 NEXT power sum losses for 25 pairs of PIC cable binder group................ 96

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B.8 Comparison of ANSI NEXT with measured NEXT ..................................... 97

B.9 Far end crosstalk (FEXT)............................................................................ 97

B.10 Comparison of ANSI FEXT with measured FEXT ...................................... 98

E.1 Examples of T1 power spectral density variations.................................... 104

E.2 Examples of DDS power spectral density variations................................. 104

E.3 Data dependent power changes in a wide band due to T1 data pattern... 105

E.4 Data dependent power changes in a narrow band due to T1 datapattern ....................................................................................................... 105

F.1 Standard ringing potential with best case start/end .................................. 108

F.2 Standard ringing pattern with worst case start/end................................... 108

F.3 Ringing waveforms (worst case generalization)........................................ 109

F.4 Triple ringing interval................................................................................. 109

F.5 Simple battery feed arrangement.............................................................. 110

Annexes

A. Evaluation of interference from new technologies into existingtechnologies ................................................................................................ 33

B. Loop information ......................................................................................... 50

C. Probability of error estimation ..................................................................... 99

D. Additional spectrum management topics currently under study by theformulating committee of this standard..................................................... 102

E. Time varying, user data-dependent crosstalk from T1 and DDSservices ..................................................................................................... 103

F. Non-continuous CO signaling events........................................................ 106

G. Informative references .............................................................................. 110

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Foreword (This foreword is not part of American National Standard T1.XXX-1999)

Accredited Standards Committee T1, Telecommunications serves the publicthrough improved understanding between carriers, customers, andmanufacturers. Technical Subcommittee T1E1 of Committee T1 developstelecommunications standards and technical reports related to various digitalsubscriber line technologies. This standard is intended to be a living document,subject to revision and updating as warranted by advances in network andequipment technology.

This standard provides spectrum management requirements andrecommendations for the administration of services and technologies that usemetallic subscriber loop cables. Spectrum management is the administration ofthe loop plant in a way that provides spectral compatibility for services andtechnologies that use pairs in the same cable. In order to achieve spectralcompatibility, energy that transfers into a loop pair, from services andtransmission system technologies on other pairs in the same cable, must notcause an unacceptable degradation of performance. In addition, energy in aparticular loop pair must not transfer into other pairs in a manner that causes anunacceptable degradation in the performance of services and technologies onthose pairs. This standard includes signal power limits, technology deploymentrestrictions, and loop assignment guidelines for certain digital subscriber linespectrum management classes. It also provides a generic analytical method todetermine spectral compatibility.

Because of the wide range of network switching systems, network transportsystems, subscriber loop plant, and customer installations in North America,conformance with this standard does not guarantee spectral compatibility oracceptable performance under all possible operating conditions. In some cases,additional spectrum management measures will be needed to ensure spectralcompatibility. This need for additional measures is necessitated by the significantdifferences between various loop technologies.

ANSI guidelines specify two categories of requirements: mandatory andrecommendation. The mandatory requirements are designated by the word shalland recommendations by the word should. Where both a mandatory requirementand a recommendation are specified for the same criterion, the recommendationrepresents a goal currently identifiable as having distinct compatibility orperformance advantages.

There are 7 annexes in this standard. Annex A is normative and considered tobe part of this standard; Annexes B-G are informative and are not consideredpart of this standard, that is, they do not include requirements but provideinformation that may be helpful to users of this standard.

Suggestions for improvement of this standard are welcome. They should be sentto the Alliance for Telecommunications Industry Solutions, T1 Secretariat, 1200 GStreet NW, Suite 500, Washington, DC 20005.

This standard was processed and approved for submittal to ANSI by AccreditedStandards Committee on Telecommunications, T1. Committee approval of thestandard does not necessarily imply that all members voted for its approval. Atthe time it approved this standard, the T1 Committee had the following members:

G. H. Peterson, ChairE. R. Hapeman, Vice-ChairS. D. Barclay, SecretaryOrganization Represented Name of Representative

EXCHANGE CARRIERS

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Exchange Carrier Member..............................................................................Name of RepresentativeName of Alternate (Alt)

Organization Represented Name of Representative

INTEREXCHANGE CARRIERS

Interexchange Carrier Member .......................................................................Name of RepresentativeName of Alternate (Alt.)

MANUFACTURERS

Manufacturer Member ....................................................................................Name of RepresentativeName of Alternate (Alt.)

GENERAL INTEREST

General Interest Member................................................................................Name of RepresentativeName of Alternate (Alt.)

Technical Subcommittee T1E1 on Interfaces, Power and Protection of Networks,which is responsible for the development of this standard, had the followingmembers:

Ed Eckert, ChairDick Brandt, Vice-ChairJohn Roquet, Secretary

Organization Represented Name of Representative

Member Name Name of RepresentativeOrganization Represented Name of Representative

Member Name .......................................................................................Name of RepresentativeName of Alternate (Alt.)

Working Group T1E1.4 on DSL Access, which had the technical responsibilityduring the development of this standard, had the following members:

Thomas J. J. Starr, ChairmanMassimo Sorbara, Vice-ChairmanRon McConnell, SecretaryDick McDonald, Editor

Syed A. AbbasRobyn AberOscar AgazziCajetan M. AkujuobiRon AllenSubra AmbatiTariq AmjedCandare M. AndersonEphraim ArnonJames AslanisKeith AtwellHiromitsu AwaiJein BaekScott J. BaerRupert BainesH. Charles BakerLeRoy BakerJohn T. BalinskiChuck BaloghArt BarabellUri BarorJohn Barselloti

Roy BatruniDon BellengerDaniel BengtssenRafi Ben-MichaelBen BennettBill BergmanDev BhattacharyaNigel BillingtonBora BirayLarry BishopRichard BishopRay BlackhamR. T. BobilinGary BoltonJan BostromMark F. BowenBruce BowiePeter BrackettRichard BrandtDave BrierLes BrownRandy Brown

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Curtis BrownmillerWilliam BuckWilliam BuckleyBill BurtonJohn BushRichard CamJohn CamagnaJohn CamagnaPatrick CameronJim CarloPaulus CarpelanC. A. CarpenterKen CavanaughGuy CerulliPaul ChangYen T. ChangTrang Chan-virakJoe CharboneauAdam ChellaliS. John ChenW. I. H. ChenWen S. ChenRaymond ChenDaniel ChenHoover ChenJacky ChowPeter ChowJohn CioffiAlan CohenNigel ColeTerry ColeMarty ColombattoKevin ConeGreg CopelandGraham G. CopleyLawrence H. CorbettMauro CostaRay CountermannBill CranePhil CrawbyDavid CummingsKim CurrieAaron DagenTom DalyTamar DanonJim DellMichael DemjanenkoShuang DengAndre' P. des RosiersPhilip DesJardinsFranz DielacherCurtis DoddJean-Louis DolmetaGuojie dongBernard DugerdilCraig EdwardsGeorge EislerTsur EitanEarl EmersonEdward S. ErlichDan Etz-HadarDave EvansVedat Eyuboglu

Charles FadelGuy FedorkowMichael FirthRocky FlaminioKay FleskesSteve FollettAl ForcucciKlaus FosmarkKevin FosterVladimir FriedmanHans-Joerg FrizlenRobin GangopadhyaClete GardenhourJuan GarzaAmit GattaniLajos GazsiTom GearyNabil GebraelAl GharakhanianEmil GhelbergMike GilbertJim GirardeauHugh GoldbergYuri GoldsteinDavid GoodmanRichard GoodsonSteven GordonLinda GosselinPeter T. GriffithsGlen GrochowskiJohn GruberSanjay GuptaL. B. GwinnCliff HallRabah HamdiRodney HannemanChris HansenGopal HarikumarRoy HarveyRoy HarveyJosef HausnerTom HaycockShahin HedayatChris HeegardPeter Niels HellerDia HelmyBrian HenrichsMalcolm HerringHanan HerzbergCurt HicksAmir HindieMinnie HoDavid HoerlDavid HolienMahbub HoqueJames C. HorngGary R. HoyneGang HuangLes HumphreyMarlis HumphreyCannon HwuIshai IlaniGreg Ioffe

Mikael IsakssonTomokazu ItoKrista S. JacobsenKen JacobsonCharlie JenkinsRalph JensenScott JezwinskiAlbin JohanssonDavid C. JonesEdward JonesRagnar JonssonJohn JoyceVern JunkmannWen-Juh KangSatoru KawanagoKen KerpezBabak KhalajSayfe kiaeiPaul KishAvi KligerRon KnipperRobert KniskernKen KoYosef KofmanJouni KoljonenHajime KotoTetsu KoyamaJames KrollPhilip J. KyeesRobert LaGrandT. K. LalaChi-Ying LanJohn LangevinMartin LaRoseSteven C. LarsenMike LassandrelloGeorge J. LawrenceDong Chul LeeHoward LevinGabriel LiHaixiang LiangZe'ev LichtensteinSimon LinJari LindholmStan LingJames LiouDave LittleFuling Frank LiuQing Li LiuValentino LivaG. W. (Wayne) LloydBob LocklearAnatoli LoewenGuozhu LongPini LozowickPerry LuAhmed MadaniRabih MakaremMarcus MaranhaoDan MarchokDoug W. MarshallAl MartinKazuya Matsumoto

Bo MatthysJack MaynardGary McAninchKent McCammonJohn McCarterShawn McCaslinRonald C. McConnellKieth McDonaldRichard A. McDonaldPeter MelsaDenis J. G. MestdaghHarry MildonianDave MillironKhashayar MirfakhraeiSteve MlikanCory ModlinMichael MoldoveanuSteven MontiDavid R. MoonKevin MullaneyJoe MullerBabak NabiliDonovan NakRandy NashFrank NavaviGil NavehGunter F. NeumeierMai-Huang NguyenRamin NobakhtAndy NorrellRao NuthalapatiStephen OhFranz OhenHans ÖhmanYusaku OkamuraKazu OkazakiVladimir OksmanAl OmranMike O'NeillAidan O'RourkTom O'SheaEric PanethPanos PapamichalisYatendra K. PathakShimon PelegMichael PellegriniMatt PendletonLarry PerronWillie PickenAshley PickeringThierry PolletMichael PolleyBob PoniatowskiBoaz PoratRon PoratCarl PosthumaPhilip PotterAmit PreussAleksandar PurkovicGordon PurtellDan QueenJim QuiliciJack Quinnell

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Ariel RadskySelem RaduSreen RaghavanAli RahjouJeffrey M. RakosAvi RapaportAvi RapaportJanice RathmannDennis J. RauschmajerGord ReesorJohn ReisterBehrooz RezvaniRon riegertTerry RileyBoaz RippinJorge RiveraRichard RobertsSilvana RodriguesJohn E. RoquetJohn RosenlofEric J. RossinMike RudeMark RussellChristopher J. RustKimmo K. SaarelaKen SakanashiDebbi SalleeHenry SamueliHal SandersWayne SandersonJamal SarmaSabit Say

Denny SchartKevin SchneiderGary SchultzBob ScottLinda SealeReuven SegevRadu SeleaAhmed ShalashMark ShannonDonald P. ShaverGreg SherrillTzvi ShukhmanEli ShustermanRex SiefertKevin SievertRichard SilvaDoug SilveiraPeter SilvermanMark SimkinsKamran SistanizadehDon SkinfillJoe SmithP. Norman SmithR. K. SmithStephen SmithEdwin J. SoltysiakMassimo SorbaraAndrew SorowkaWalt SotoJ. Scott SpradleyPaul SpruytTom Starr

William StewartJames StisciaJeff StraitCaleb StrittmatterRichard StuartRay SubbankarHenri SuyderhoudJames SzeligaHiroshi TakatoriDaryl C. TannisLarry TaylorMatthew TaylorSteve TaylorGary TennysonRainer ThoenesVernon TiceEd TirakianChi-Lin TomAntti TommiskaJ. Alberto TorresRichard L. TownsendBob TracyDwen-Ren TsaiMarcos TzannesMasami UedaJohn UlanskasJuan Ramon UribePeter VaclavikCraig ValentiNick van BavelHarry van der MeerFrank Van der Putten

Dick van GelderJeff Van HorneM. VautierRobert L. VealDale VeenemanRami VerbinPieter VersavelRaman ViswanathanJeff WaldhuterJosef WaldingerBrian WaringDewight WarrenCurtis WatersAlan WeissbergerJ. J. WernerRick WeselGreg WhelanAlbert WhiteSong WongBernard E. WorneHan YehSoobin YimKyung-Hyun YooGavin YoungIrvin YoungbergXiaolong YuShaike ZalitzkyXuming ZhangGeorge Zimmerman

T1E1.4/99-002R4 DRAFT ANSI T1.XXX-1999

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American National Standardfor Telecommunications

Spectrum Managementfor Loop Transmission Systems

1. Scope, purpose, and application

1.1 Scope

This standard provides spectrum management requirements and recommendations for the administrationof services and technologies that use metallic subscriber loop cables. Spectrum management is theadministration of the loop plant in a way that provides spectral compatibility for services and technologiesthat use pairs in the same cable. In order to achieve spectral compatibility, energy that transfers into aloop pair, from services and transmission system technologies on other pairs in the same cable, must notcause an unacceptable degradation of performance. In addition, energy in a particular loop pair must nottransfer into other pairs in a manner that causes an unacceptable degradation in the performance ofservices and technologies on those pairs.

This standard includes the following types of requirements and recommendations for certain defineddigital subscriber line spectrum management classes:

- power spectral density (PSD)

- total average power

- transverse balance

- deployment restrictions

- loop assignment guidelines

The standard also specifies a generic analytical method to determine the spectral compatibility of looptechnologies that do not qualify for one of the spectrum management classes defined in this standard.

Requirements in this standard are specified for insulated solid copper conductor twisted-pair cables usedin the subscriber loop environment.

A transmission system conforms to this standard only if it meets all of the applicable requirements for oneof the DSL spectrum management classes defined in this standard. Guarded systems do not necessarilyconform with this standard.

DSL transmission systems that meet all of the specifications associated with one of the DSL spectrummanagement classes are assumed to be spectrally compatible in the same binder group, unless otherwiseindicated, with all of the guarded systems defined in this standard. Meeting the specifications associatedwith one of the spectrum management classes in this standard does not assure spectral compatibility withunguarded loop transmission systems.

The requirements in this issue of this standard assume that the DSL system is deployed between a wirecenter (WC) and a customer installation (CI). Applications that locate the TU-C at an intermediate pointbetween the WC and the CI or applications that use intermediate repeaters are not specified, and arebeyond the scope of the requirements in this issue of this standard. These applications can, in somecases, cause crosstalk that is greater than that from a WC deployment. Guidelines for such applicationswere not available at the time this standard was written.


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Electromagnetic Compatibility (EMC) is outside of the scope of this standard. In addition, the spectrummanagement of privately owned twisted-pair cables or customer premises twisted-pair cabling are beyondthe scope of this standard although the information in this standard may be useful in such applications.

The signals that the network and customer installation (CI) apply to the loop are basically of two types:normal telecommunications transmission system voltages and currents, and voltages and currents due tomaintenance activities. The normal network and CI signals are addressed in this standard. Voltages andcurrents due to network maintenance activities and abnormal voltages and currents that are the result ofthe environment (e.g., induced voltages and currents or lightning) are not covered in this standard.

1.2 Purpose

The purpose of this standard is to minimize the potential for crosstalk interference in twisted-pairsubscriber loop cables that are shared by multiple carriers. When a single carrier deploys technologies inloop plant, it alone has the responsibility for spectral compatibility and may select any combination ofcompatible loop technologies. In an unbundled loop environment however, multiple carriers utilize pairs inthe same loop cables. In such instances, if services and technologies are deployed in an uncoordinatedmanner, they may interfere with each other. This standard assumes that loop cables are shared bymultiple carriers and that all carriers share the responsibility for spectral compatibility.

This standard provides information that will help to ensure that twisted-pair transmission systems can co-exist – usually in the same binder group - without impaired operation due to crosstalk interference. Thestandard is intended to be used by carriers to manage the loop plant and by manufacturers in the designof loop transmission systems.

This standard was also developed to assist carriers, manufacturers, and users of products to beconnected to local loops, to understand the characteristics of twisted-pair loop cables. In addition, thisstandard can be used to determine if new services and loop transmission system technologies arespectrally compatible with certain guarded services and technologies that are defined in this standard.

This standard is intended to be consistent with Part 68, Subpart D, of the FCC Rules and Regulations thatcontains requirements for the registration of customer installation terminal equipment to protect thenetwork from harm. Some of the digital subscriber line spectrum management classes defined in thisstandard are not covered by Part 68. If Part 68 rules are subsequently established for technologies thatfall into those categories, the requirements in this standard can be referenced. Tariffs, contracts, orregulatory acts in various jurisdictions may contain requirements different from those in this standard.

The provisions of this standard are also intended to be consistent with applicable requirements concerningsafety and environmental conditions.

1.3 Application

This standard is applicable to twisted-pair cables that are used by multiple carriers in the local loopenvironment.

Availability of the loops described and defined in this standard depends upon the network facilities servinga particular CI. All of the loops described in this standard may not be universally available. For example,a loop that supports Basic Rate ISDN can only be provided if the facilities serving the CI are qualified tosupport such technology.

Because of the wide range of network switching systems, network transport systems, subscriber loopplant, and CIs in North America, conformance with this standard does not guarantee spectral compatibilityor acceptable performance under all possible operating conditions. In some cases, additional spectrummanagement measures will be needed to ensure spectral compatibility. This need for additionalmeasures is necessitated by the significant differences between various loop transmission systemtechnologies.


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2. Normative references

The following standards contain provisions that, through reference in this text, constitute provisions of thisAmerican National Standard. At the time of publication, the editions indicated were valid. All standardsare subject to revision, and parties to agreements based on this standard are encouraged to investigatethe possibility of applying the most recent editions of the standards indicated below.

ANSI T1.413-1998, American National Standard for Telecommunications – Network and CustomerInstallation Interfaces – Asymmetrical Digital Subscriber Line (ADSL) Metallic Interface.

ANSI T1.601-1998, American National Standard for Telecommunications – Integrated Services DigitalNetwork (ISDN) – Basic Access Interface for Use on Metallic Loops for Application on the Network Side ofthe NT (Layer 1 Specification).

ANSI T1.401-992, Carrier-to-Customer Metallic Interface - Digital Data at 64kb/s and Subrates.

3. Definitions, abbreviations, acronyms, and symbols

3.1 Definitions

3.1.1 american wire gauge: A unit used to measure the diameter of round wire.

3.1.2 balance, balanced: See longitudinal balance and transverse balance.

3.1.3 binder group: A collection of twisted pairs consisting of 12, 13, or 25 pair units, or a multiunitconsisting of 50 or 100 twisted pairs, that have been assembled together during the manufacturingprocess and bound together with colored tape for identification.

3.1.4 bit error ratio: A performance measure consisting of the ratio of bits in error to the total numberof bits transmitted.

3.1.5 carrier: An organization that provides telecommunications services to customer installations.

3.1.6 central office: Refers to a switching entity for an NNX code. The term "wire center" is used torefer to the building when referring to the origin of the outside loop plant. Sometimes "central office" isused to mean "wire center" when the intent is clear. A wire center often has more than one central officeNNX code.

3.1.7 conductor: A continuous solid copper or aluminum wire that has a circular cross-section.

3.1.8 crosstalk: Electromagnetic energy that couples into a metallic cable pair from signals on otherpairs in the same cable.

3.1.9 customer installation: All cabling and equipment on the customer side of the network interface.

3.1.10 customer premises equipment: Telecommunications equipment located at the customerinstallation on the customer side of the network interface.

3.1.11 demarcation point: See network interface.

3.1.12 disturbed pair: A cable pair that has a service or technology that is experiencing anunacceptable degradation in performance (e.g., decreased bit error ratio, decreased data rate, ordecreased loop reach) because of crosstalk interference from one or more other pairs in the same cable.

3.1.13 disturbing pair: A pair with a signal that is contributing to crosstalk interference that is degradingthe performance (e.g., decreased bit error ratio, decreased data rate, or decreased loop reach) of aservice or technology on another pair in the same cable.

3.1.14 downstream: The direction of transmission from the carrier Central Office to the CI.


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3.1.15 drop wire: A type of loop cable, consisting or one or more pairs, that is used between the loopcable terminal and the network interface device.

3.1.16 far-end crosstalk: Crosstalk that occurs when the receiver on a disturbed pair is located at theother (far) end of the cable as the transmitter of a disturbing pair.

3.1.17 guarded: A term used in this standard to describe a loop transmission system with which DSLsystems and other new loop transmission systems are required to demonstrate spectral compatibility.

3.1.18 insulated conductor: A conductor that has been surrounded with insulation that is often color-coded.

3.1.19 insulation: The dielectric material that surrounds a conductor and prevents it from contactingother conductive material.

3.1.20 longitudinal balance: Describes the degree of symmetry with respect to ground of a two-conductor transmission line. Longitudinal balance may be expressed as 20 times the log10 of themagnitude of the ratio of an applied longitudinal voltage (referenced to ground) to the resultant metallicvoltage.

3.1.21 loop: A communication path between the distributing frame in a carrier Central Office and thenetwork interface at a customer location.

3.1.22 multiunit: Two or more subunits that are assembled together into 50 or 100 pair groups andbound with colored binder tape for identification.

3.1.23 near-end crosstalk: Crosstalk that occurs when the receiver on a disturbed pair is located at thesame (near) end of the cable as the transmitter of a disturbing pair.

3.1.24 network: All equipment and facilities, including loop plant, located on the carrier side of thenetwork interface.

3.1.25 network interface: The physical demarcation point between carrier network loop facilities and theCI.

3.1.26 pair: Two insulated conductors.

3.1.27 pair unit: Twelve, thirteen, or twenty-five twisted pairs that have been assembled together andbound with colored binder tape for identification.

3.1.28 power spectral density (PSD): The power level and frequency content of a transmitted signal.

3.1.29 short-term stationary: A term used in this standard to describe a loop transmission system inwhich an “ON” condition (in which the transmitter generates a signal) alternates with an “OFF” condition(in which the transmitter is silent or generates only a pilot tone).

3.1.30 spectral compatibility: The capability of two loop transmission system technologies to coexist inthe same cable and operate satisfactorily in the presence of crosstalk noise from each other.

3.1.31 spectrum management: In this standard, the term refers to processes that are intended tominimize the potential for interference and maximize the utility of the frequency spectrum of metallic loopcables.

3.1.32 spectrum management class: In this standard, the term refers to the classes defined in 5.2,classifying the technologies in terms of their PSD. Abbreviated SM class.

3.1.33 subunit: A term used to identify the 12 or 13 pair units that are used in a 50 pair multiunit or the25 pair units that are used in a 100 pair multiunit.

3.1.34 transverse balance: A comparison of the voltage of a transmitted metallic signal to the voltage ofany resulting longitudinal or transverse signal. See 6.5.

3.1.35 type I PSDS: A legacy loop transmission system based on 56 kbps digital data service that usesAMI operating at 56 kbps on two loop pairs to provide a 4-wire full-duplex digital channel. Network


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signaling is accomplished using bipolar patterns that include bipolar violations. For more information, seeTIA/EIA-596.

3.1.36 type II PSDS: A legacy loop transmission system that functions in two modes: analog and digital.Analog signaling is used to perform network supervisory and address signaling. The system is switchedto the digital mode after a connection is established. Type II PSDS uses Time Compression Multiplexingand AMI operating at 144 kbps to provide a full-duplex 56 kbps service on a 2-wire loop. For moreinformation, see TIA/EIA-596.

3.1.37 type III PSDS: A legacy loop transmission system that uses Time Compression Multiplexing andAMI operating at 160 kbps to provide two full-duplex digital channels on a 2-wire loop. One digital channelis an 8 kbps signaling channel for supervisory and address signaling and the other is a 64 kbps datachannel. For more information, see TIA/EIA-596.

3.1.38 twisted pair: A balanced transmission line consisting of two insulated conductors that have beentwisted together during the manufacturing process to reduce coupling to and from external circuits. Seebalanced.

3.1.39 unguarded: A term used in this standard to describe a loop transmission system with which DSLsystems and other new loop transmission systems are not required to demonstrate spectral compatibility.

3.1.40 upstream: The direction of transmission from the CI to the carrier Central Office.

3.1.41 voicegrade: A term used to qualify a channel, facility, or service that is suitable for thetransmission of speech, data, or facsimile signals; generally with a frequency range of about 300 to 3000Hz.

3.1.42 wire center: Telephone building that is the origin of the outside loop plant. Usually has one ormore central offices. See central office.

3.2 Abbreviations, acronyms, and symbols

The following acronyms are used throughout this document.

ADSL Asymmetric Digital Subscriber Line

ANS American National Standard

ANSI American National Standards Institute

AWG American Wire Gauge; see definition

BER bit error ratio; see definition

bps bits per second

CI customer installation; see definition

CO central office; see definition

CPE customer premises equipment; see definition

CSA carrier serving area

dB decibel

dBm decibel referenced to 1 milliwatt

dBrn decibel referenced to noise


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dBrnC decibel referenced to noise with C-message weighting

DDS digital data service

DSL digital subscriber line

DUT device under test

FCC Federal Communications Commission

FEXT far-end crosstalk; see definition

HDSL high-bit-rate digital subscriber line

HDSL2 high-bit-rate digital subscriber line over a single pair

Hz hertz

ISDN Integrated Services Digital Network

ITU-T International Telecommunication Union – Telecom Sector

kHz kilohertz

mH millihenry

ms millisecond

NEXT near-end crosstalk; see definition

NI network interface; see definition

PSD power spectral density

PSDS public switched digital service. See definitions of type 1, type II, andtype III.

RADSL rate adaptive digital subscriber line

RLCG resistance, inductance, capacitance, and conductance

RRD revised resistance design

SM Spectrum management, e.g., SM class 1. See definition.

T1 type of 4-wire metallic 1.544 Mbps transmission system

TU-C Transceiver Unit – Central office end. Sometimes combined withanother letter; e.g., ATU-C for a central office ADSL transceiver

TU-R Transceiver Unit – Remote terminal end. Sometimes combined withanother letter; e.g., ATU-R for a remote ADSL transceiver

VDSL very-high-bit-rate digital subscriber line

WC wire center; see definition


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4. General Information

Most of the subscriber loop plant in North America consists of metallic cables that were designed primarilyfor voicegrade services. Several other types of services and technologies use these loop cables howeverincluding, but not limited to, digital data services, T1-carrier systems, and Digital Subscriber Line (DSL)transmission systems.

Metallic loop cables generally contain several solid copper conductors that are circular in cross-section.Each conductor is surrounded by insulation that is usually color-coded. During manufacturing, pairs ofinsulated conductors are twisted together. Several twisted pairs are then assembled together into unitscalled binder groups that are bound with colored tape for identification.

The signals that are transmitted on a loop cable pair create an electromagnetic field that surrounds nearbypairs and induces voltages into those pairs. The twisting of the insulated conductors into pairs minimizesthis coupling as does the bundling of pairs into binder groups. Despite these measures however, acapacitive coupling still exists between the pairs of a multipair loop cable.

This clause provides general information about crosstalk interference in metallic loop cables, the spectralcompatibility of loop transmission systems, and various aspects of spectrum management. Clause 5provides signal power limitations and deployment restrictions for several DSL spectrum managementclasses. Conformance testing methodology is provided in clause 6.

4.1 Crosstalk

The electromagnetic energy that couples into a metallic cable pair from services and transmission systemtechnologies on other pairs in the same cable is unwanted energy and is called crosstalk noise or justplain “crosstalk”. Crosstalk may, or may not, be disturbing. When crosstalk causes an unacceptabledegradation in the performance of victim services or technologies in the same cable, it is called crosstalkinterference. Preventing crosstalk interference requires the careful manufacturing, installation,maintenance, and administration of loop cables.

Crosstalk is sensitive to frequency, signal strength, and exposure. High frequency energy couples intoother pairs more than low frequency energy because as the signal frequency increases, the crosstalkcoupling loss between the pairs of a cable decreases. Hence, for two signals of equal strength, the higherthe frequency, the greater the crosstalk noise.

A strong signal will transfer more power into other pairs than will a weaker signal. The amount ofcrosstalk noise is directly proportional to the power of the disturbing signal. The stronger the disturbingsignal, the greater the crosstalk noise. Thus, one of the most effective means of controlling crosstalknoise is to limit the signal energy that is applied to cable pairs. Signal power limitations for several DSLclasses are provided in clause 5.

Exposure is a measure of the proximity of metallic pairs at various points along a cable and the lengthover which pairs are in close proximity. The greater the exposure, the greater the total crosstalk noise.Since it is impossible to predict the exact amount of exposure between any two pairs in a cable, statisticalexposure models are used for the crosstalk margin evaluations described in Annex A.

Crosstalk noise that occurs when a receiver on a disturbed pair is located at the same end of the cable asthe transmitter of a disturbing pair is called Near-End-Crosstalk (NEXT). Crosstalk noise that occurs whena receiver on a disturbed pair is located at the other end of the cable as the transmitter of the disturbingpair is called Far-End-Crosstalk. NEXT is more troublesome than FEXT when transmission takes place inboth directions in a binder group. In this situation, which is typical of the loop environment, a strongtransmitted signal can interfere with a nearby weak receive signal.

4.2 Spectral compatibility

Spectral compatibility is the capability of two loop transmission system technologies to coexist in the samecable and operate satisfactorily in the presence of crosstalk noise from each other. Spectral compatibilityis defined pair-wise for specific well-defined transmission systems.


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Technology A is considered to be a spectrally compatible disturber into technology B when signalsgenerated by A, on any number of pairs in a binder group, do not cause a service degradation fortechnology B when B is using a pair in the same binder group.

Likewise, technology B is considered to be a spectrally compatible disturber into technology A whensignals generated by B, on any number of pairs in a binder group, do not cause a service degradation fortechnology A when A is using a pair in the same binder group.

Service degradation is defined as not meeting the Bit Error Ratio (BER) and Signal-to-Noise Ratio (SNR)margin requirements defined for the specific transmission system for all loop lengths, model loops, or lossvalues within the requirements for the specific transmission system.

This standard also defines signal power limitations and deployment restrictions associated with severalDSL spectrum management classifications. Any loop transmission system that meets all of the signalpower limits and deployment restrictions of a particular DSL spectrum management class is assumed tobe spectrally compatible with all of the guarded systems defined in this standard.

4.3 Spectrum management

In this standard, the term spectrum management refers to processes that are intended to minimize thepotential for crosstalk interference and maximize the utility of the frequency spectrum in multipair metallicloop cables.The spectrum management requirements and recommendations in this standard include signal powerlimitations, technology deployment restrictions, and a generic analytical method that can be used to definenew DSL spectrum management classifications or determine the spectral compatibility of differenttechnologies. The requirements and recommendations in this standard are intended to provide spectralcompatibility with certain defined guarded loop transmission systems and thereby maximize the use of thebandwidth provided by metallic loop cables.

This standard recognizes that the United States telecommunications industry is operating in an unbundledloop environment where multiple carriers utilize pairs in the same loop cables. In this situation, theresponsibility for spectral compatibility has to be shared by all carriers. When new loop technologies areintroduced by different carriers, each independently proving spectral compatibility with guarded systems,the possibility remains that, without proper coordination, the new technologies may interfere with eachother.

4.3.1 Guarded loop services and technologiesThis standard defines certain guarded loop services and technologies. Guarded systems are defined asloop transmission systems with which the DSL spectrum management classes defined in this standard,and other new loop transmission systems, are required to demonstrate spectral compatibility. Theguarded systems defined in this standard are systems that have been deployed in high numbers as wellas standards-based DSL systems that are expected to be deployed in high numbers in the near future.

In order to assure spectral compatibility with the anticipated mix of current and future technologies on loopbinder groups, this standard has defined a set of loop transmission systems1 as follows to which spectralcompatibility shall be demonstrated:

- Voicegrade services2

- Digital Data Service (DDS) based on T1.410

- Basic Rate Integrated Services Digital Network (ISDN) based on T1.601. Note that this includes 2-channel digital systems (UDC-2) based on ISDN technology.

–––––––1 Very-High-Bit Rate Digital Subscriber Line (VDSL) technology is currently in the standards development process.VDSL is expected to be added to the guarded technology list. In anticipation of this, some accomodation for VDSLhas already been made in the signal power limitations for the digital subscriber line spectrum management classesdefined in clause 5.2 Voicegrade services include speech, data, and tone signals that use the frequency spectrum from 0 to 4 kHz. Formore information, see Annex A.


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- High-Bit-Rate Digital Subscriber Line (HDSL) based on G.991.1, Annex A.

- Asymmetrical Digital Subscriber Line (ADSL) based on T1.413-1998 with non-overlappedupstream/downstream mode

- RADSL based on Committee T1 TR-59.

- Splitterless ADSL based on G.992.2 with non-overlapped upstream/downstream mode

- Repeatered T1 (1.544 Mbps) technology based on T1.403

- HDSL2 (DS1 payload on single pair) based on the draft standard for HDSL2

This set is defined to take into account: 1) voluntary DSL standards based on industry consensus andopen specifications and 2) several legacy loop transmission systems. Spectral compatibility – generally inthe same binder group - with the guarded services and technologies listed above shall be demonstratedby meeting all of the signal power limitations and other criteria for one of the DSL spectrum managementclasses defined in clause 5 (Method A).

4.3.1.1 Voicegrade services

Voicegrade services include speech, network signaling, data, and tone signals that use the frequencyspectrum from 0 to 4 kHz. (See Annex A.)

4.3.1.2 Digital Data Services (DDS)

Digital Data Services, based upon T1.410, operate at 64 kbps and subrates of 2.4, 4.8, 9.6, 19.2 38.4, 56kbps. Secondary channel services are also available for all subrates. While all DDS subrates andsubrates with secondary channels are guarded, the DDS analytical evaluation procedure in Annex Afocuses on 56 kbps and 64 kbps DDS in order to reduce the number of DDS evaluations that a newsystem must undergo. Since Type 1 Public Switched Digital Service (PSDS) uses the same physical layeras 56 kbps DDS, any new technology that demonstrates compatibility with 56 kbps DDS will also becompatible with Type 1 PSDS.

4.3.1.3 Basic Rate ISDN (BRI)

In the context of this standard, BRI represents a family of guarded loop transmission systems that use thetransceiver technology described in T1.601. The family includes traditional BRI that uses the ISDN datalink layer protocols described in T1.602 as well as other systems that have adapted the T1.601 layer 1transceiver technology for use as:

- a packet network access system (IDSL)

- a point to point transport system sometimes referred to as a Universal Digital Channel (UDC).

BRI, IDSL, and UDC are defined in this standard as systems that use the 2B1Q line code, operate at 80kbaud for transmission at 160 kbps, and may be transported via DLC by using BRI termination extension(BRITE) devices. The entire BRI family is guarded. The analytical method for demonstrating compatibilitywith BRI in Annex A does not differentiate between the members of the BRI family and adequately guardsall members of the family.

4.3.1.4 High-Bit-Rate Digital Subscriber Line (HDSL)

HDSL systems are designed to transport 784 kbps over Carrier Serving Area (CSA) distances on a singlenon-loaded twisted pair. The most common application transports a 1.544 Mbps payload on two non-loaded twisted pairs but some applications may use a single pair. Some HDSL applications extend thereach by the use of intermediate repeaters. Guarded HDSL systems are echo canceller hybrid systemsthat use the 2B1Q line code and operate at 392 kbaud. The analytical method for demonstratingcompatibility with HDSL in Annex A does not differentiate between one pair and two pair applications.


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4.3.1.5 HDSL2

HDSL2 is a second generation HDSL loop transmission system that is currently in the standardsdevelopment process. The system is designed to transport a 1.544 Mbps payload on a single non-loadedtwisted pair at Carrier Serving Area (CSA) distances.

4.3.1.6 ADSL, RADSL, and Splitterless ADSL

The guarded asymmetrical DSL systems operate using different frequency bands (non-overlapped) forupstream and downstream operation. The analytical method for demonstrating compatibility with thesesystems in Annex A is described in terms of the relevant line code (i.e., DMT, CAP, or QAM).

4.3.1.7 Repeatered T1 systems

T1 is compatible in the same binder group with many of the other guarded systems; however, separatebinder groups are recommended wherever possible to separate T1 from spectrum management class 5systems (i.e., ADSL, RADSL, and Splitterless ADSL). The analytical method for demonstratingcompatibility with T1 in Annex A, permits a new technology to assume that T1 is in the same, or anadjacent, binder group.Unguarded loop services and technologies.

4.3.2 Unguarded loop services and technologies

Meeting the signal power limitations and other criteria for one of the spectrum management classes inclause 5 does not assure spectral compatibility with unguarded loop transmission systems including, butnot limited to, the following services and technologies:

- 15 kHz program audio

- Type II PSDS

- Type III PSDS

- Local Area Data Channel CPE

- Data-over-voice multiplexers associated with CO-LAN services

- Analog Carrier

Spectral compatibility with the services and technologies listed above may be determined by using theanalytical method described in Annex A however the user will have to provide the unguarded systeminformation that is necessary for margin calculations since this is not provided in Annex A.

4.3.3 Signal power limitations (method A)

Since strong signals transfer more power into other pairs than weaker signals, the most widely used andmost successful method of controlling crosstalk interference and achieving spectral compatibility isthrough the use of signal power limitations. Signal power limitations specify the amplitude, frequencydistribution, and total power of electrical signals at the point where the signal enters the subscriber loopcable.

Clause 5 of this standard defines signal power limits for several DSL spectrum managementclassifications. The requirements apply to signals transmitted by DSL transceivers units whether locatedin a Central Office (TU-C) or a remote terminal location (TU-R). The remote terminal location is usually onor near the customer premises.

TU-C and TU-R equipment that meets the signal power limitations and other criteria for one of the DSLspectrum management classes defined in clause 5 is expected to achieve spectral compatibility – in thesame binder group unless otherwise specified - with the guarded transmission systems defined in thisstandard.

The characterization of a transmitted signal by power level and frequency content is called the powerspectral density (PSD) of the signal. The primary signal power requirements in this standard are specifiedthrough the use of PSD masks and templates. The PSD mask shows the maximum power boundary orlimit, in dBm per Hz, for the transmitted signal. The use of the PSD masks and templates is describedmore fully in 6.1, 6.2, and 6.3.


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4.3.3.1 Transceiver unit – remote terminal end (TU-R)

Part 68 of the FCC Rules and Regulations contain mandatory signal power limits for several types ofcustomer premises equipment (CPE) including voice, voiceband data, DDS subrates, public switcheddigital services (PSDS), ISDN, local area data channel (LADC), and DS1. Clause 5 of this standarddefines signal power limits for several DSL spectrum management classifications that are not currentlycovered by Part 68.

The TU-R equipment used with DSL systems is usually CPE however in some cases it may be networkequipment. The TU-R signal power limits in clause 5 shall be applicable regardless of whether or not theTU-R is network equipment or CPE. Any TU-R that transmits a signal into a metallic loop cable shall meetthe relevant upstream signal power limitations and other criteria associated with one of the DSL spectrummanagement classifications defined in section 5.

4.3.3.2 Transceiver unit – central office end (TU-C)

Historically, carriers have controlled the transmitted signal power of network elements through thedevelopment and use of voluntary industry standards related to particular technologies. Clause 5 of thisstandard defines signal power limits for several DSL spectrum management classifications. The DSLclassifications defined in clause 5 are based on the industry’s current view of requirements for spectrummanagement.

The TU-C is network equipment. Any TU-C that transmits a signal into a metallic loop cable shall meetthe relevant downstream signal power limitations and other criteria associated with one of the DSLspectrum management classifications defined in section 5.

4.3.4 Technology deployment restrictions

Some loop transmission system technologies can be deployed in a manner that substantially increasesthe likelihood of crosstalk interference. To prevent interference in such instances, it is necessary toadhere to certain technology deployment restrictions in addition to signal power limitations. Suchadditional requirements may include deployment restrictions and loop assignment guidelines.

Technology deployment restrictions, if applicable, have been provided along with the signal powerlimitations for each of the DSL spectrum management classes defined in clause 5. Any service or looptransmission system that meets the signal power limitations for one of the DSL spectrum managementclasses defined in this standard shall be deployed according the relevant deployment restrictions that arespecified for that DSL spectrum management class in clause 5.

4.3.4.1 Deployment restrictions

Deployment restrictions constrain the way loop transmission systems are operated so that theassumptions on which spectral compatibility was determined will remain valid. Deployment restrictionsinclude such things as prohibitions against spectrum management class 5 systems using reverseupstream/downstream operation or overlapping upstream/downstream frequency spectrums. If a DSLspectrum management class has applicable deployment restrictions, they shall be specified in clause 5.

4.3.4.2 Loop assignment guidelines

In order to achieve and maintain spectral compatibility with some of the guarded systems, some DSLspectrum management classes may require loop assignment guidelines. Such guidelines are intended toreduce crosstalk by decreasing exposure.

Although it is impossible to predict the exact amount of exposure between any two pairs in a cable, wherebinder group integrity has been maintained, some general assumptions can be made concerningexposure. For example, one can assume that pairs in the same binder group have more exposure thanpairs in adjacent binder groups.

An example of a loop assignment guideline is the requirement that services in a specific DSL spectrummanagement class be assigned to pairs that are not in the same binder group as one of the guardedsystems (e.g., T1). If a DSL spectrum management class has applicable loop assignment guidelines, they


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will be specified in clause 5.Loop assignment guidelines assume that a reasonable degree of binder groupintegrity exists in the loop environment. This is often not the case. Therefore, technologies thatdemonstrate spectral compatibility by using the analytical method in Annex A (Method B) shall not relyupon binder group separation in order to achieve full compatibility with any guarded transmission systemother than T1. However, this standard does not preclude the use of binder group separation.

4.3.4.3 Expected performance levels

In some instances, a particular DSL spectrum management class may need to reduce its expectedperformance level (i.e., data rate, BER, etc.) in order to achieve and maintain spectral compatibility withone or more of the guarded systems. If this standard permits such a reduction for a particular DSLspectrum management class, it shall be explicitly stated for that class in clause 5. This standard does not,and shall not, permit a reduction in the performance of a guarded system.

4.3.4.4 Loop reach values

In some instances, a particular DSL spectrum management class may need to reduce its expected loopreach in order to achieve and maintain spectral compatibility with one or more of the guarded systems. Ifthis standard permits such a reduction for a particular DSL spectrum management class, it shall beexplicitly stated for that class in 5. This standard does not, and shall not, permit a reduction in the loopreach of a guarded system.

4.3.5 Analytical method of determining spectral compatibility (method B)

It is recognized that future technologies may transmit signals that do not conform to the signal powerspecifications for one of the spectrum management classes defined in Clause 5, but which might still bespectrally compatible with the guarded loop services and technologies listed in 4.3.1. In order to nurtureinnovation in the development of new technologies which further maximize the utility of the copper loopplant, an analytical method for evaluating new technologies is provided in Annex A.

This method (referred to as Method B) involves the computation of signal to noise margins for guardedsystems, and provides an industry-approved method of determining the spectral compatibility of any looptransmission system with the guarded loop transmission systems defined in this standard. For each ofthese guarded systems, Annex A provides the specific NEXT margin formulas, evaluation loops, anddefined crosstalk environments required by Method B.

The analytical method in Annex A should be used to develop new signal power limits and deploymentrestrictions for new DSL spectrum management classes. It is expected that this analytical method will alsobe used to provide guidance during new system development. However, as noted in 4.3, such use couldlead to the introduction of several new technologies that would be compatible with guarded systems butnot necessarily compatible with each other. Therefore, system developers are encouraged to bring newDSL technologies that do not fit into existing spectrum management classes into the formulating group forthis standard, so that the creation of a new class and any associated deployment restrictions can beconsidered. Other processes, such as the disclosure of verifiable methods to assess spectral compatibilitywith the new technology, may also help avoid the uncoordinated introduction of new technologies thatcould result in crosstalk interference.

4.3.5.1 Margin computations

Margin computations determine the crosstalk margin in decibels (dB). Unless otherwise specified inAnnex A, each guarded system shall have a margin of at least 6 dB with BER ≤ 10 –7. Margin is a functionof many variables including:

a) Crosstalk coupling loss,

b) Loss characteristics of loop cables,

c) Characteristics of the disturbed signal,

d) Receiver technology of the disturbed system, and

e) Characteristics of the disturber signal.


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Annex A provides the margin formula and the information associated with items a) thorough d). The userwill have to supply the information for item e).

The configuration in figure 1 shall be used when the effect of system B NEXT and FEXT interference intoa system A downstream receiver is evaluated and both systems do not have the same loop reach.

This simulation set-up assumes that all of the head-end transmitters (ATU-C, HTU-C, etc) of both systemsare co-located at a central location and that the distance-limited system B does not use range-extendingrepeaters. It is also assumed that all of the system B upstream transmitters are co-located at the longestsupported loop length. This gives a worst case view of the effect of system B interference upon theoperation of system A.

The first cable section is adjusted to cover the maximum reach distance of system B, and the secondcable section is adjusted to cover the remaining length of any test loop under consideration. The systemB FEXT noise generator shall generate FEXT noise equivalent to a system B output signal passedthrough the FEXT coupling loss and through the whole cable from the system B transmitter location to thesystem A receiver location (sum of first and second sections).

The configuration in figure 2 shall be used to simulate the effect of system B interference onto theupstream operation of system A.

This simulation set-up assumes that all of the head-end receivers (ATU-C, HTU-C, etc) of both systemsare co-located at a central location. In this case, the system B FEXT noise generator shall generate FEXTnoise equivalent to a system B output signal passed through the FEXT coupling loss and through thecable section from the system B transmitter location to the system A receiver location (first cable sectiononly).

Evaluation loopsFor each guarded system, Annex A provides a set of loops that shall be used foranalytical evaluations.

4.3.5.2 Crosstalk environment

System performance computations shall be based on a crosstalk environment that reflects the expectedsix-year 99% worst-case statistical probability of binder fills.

5. Signal power limits and other criteria

Crosstalk noise is controlled primarily through the use of signal power limits that consist of Power SpectralDensity (PSD) limitations and total average power limitations. Additional criteria, such as transversebalance requirements and deployment restrictions, are also important. This clause provides all of thesespecifications for several DSL spectrum management classes. The conformance testing methodology inclause 6 shall be used to determine compliance with the requirements in this clause.

DSL transmission systems that meet the PSD limitations, total average power limitations and transversebalance requirements for one of the DSL spectrum management classes defined in this clause shall bedeployed according to the applicable deployment restrictions and loop assignment guidelines that arespecified in this clause for each specific DSL spectrum management class. The deployment restrictionsfor some of the DSL classes limit the distance that a system can operate at in order to ensure thatcrosstalk from systems in that class will not impair guarded systems.

A multimode DSL system shall be deployed according to the applicable deployment restrictions and loopassignment guidelines associated with the class for which it is configured.

5.1 Short-term stationary systems

Some types of DSL transmitters operate in transmission modes in which an “ON” condition (in which thetransmitter generates a signal) alternates with an “OFF” condition (in which the transmitter is silent orgenerates only a pilot tone). Examples of such transmitters include burst transmission systems andsystems which use quiescent modes to reduce power consumption during idle data periods. Such


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transmitters are referred to as “short-term stationary,” since during the ON condition the transmitted signalhas the same effect as a stationary (or cyclo-stationary) signal when observed over an appropriately shorttime interval. Due to the relative frequency of ON/OFF and OFF/ON transitions in short-term stationarytransmitters, additional conformance criteria are applied to these transmitters.

Clause 6.4 defines a test to determine whether short-term stationary conformance criteria shall be appliedto a DUT and defines the short-term stationary conformance criteria.

5.2 Spectrum management classes

5.2.1 Spectrum management class 1 (formerly VLB)

Spectrum management class 1 is intended for DSL transmission systems that operate in the frequencyspectrum up to about 115 kHz, including most, but not all, T1.601 compliant systems.

5.2.1.1 Spectrum management class 1 PSD and total average power limitation

Spectrum management class 1 TU-C and TU-R equipment shall meet the PSD conformance criteria in6.1 using the PSD template described in table 1 and figure 3. The total average power into 135 Ohms andbelow 115 kHz that is transmitted by the spectrum management class 1 TU-C and TU-R equipment shallbe 14.0 dBm or less.

5.2.1.2 Spectrum management class 1 transverse balance requirement

The transverse balance of spectrum management class 1 TU-C and TU-R equipment shall be measuredover the applicable frequency range using the procedures and 135-ohm measurement configurationspecified in clause 6. The transverse balance of spectrum management class 1 TU-C and TU-Requipment exceed the values in table 2 over the entire range of frequencies between the upper and lower–20 dB points of the signal pass-band.

5.2.1.3 Spectrum management class 1 deployment restrictions and loop assignment guidelines

Non-repeatered loop transmission systems that meet the signal power and transverse balancerequirements associated with Spectrum Management Class 1 may use any loop facility and may beassigned to pairs that are in the same binder group as any of the guarded systems defined in thisstandard.

5.2.2 Spectrum management class 2 (formerly LB)

Spectrum Management Class 2 is intended for DSL transmission systems that operate in the frequencyspectrum from 0 to about 238 kHz.


Spectrum management class 2 TU-C and TU-R equipment shall meet the PSD conformance criteria insection 6 using the PSD template described in table 3 and figure 4.

The total average power below 238 kHz that is transmitted by Spectrum Management Class 2 TU-C andTU-R equipment shall be 14.0 dBm or less.

5.2.2.2 Spectrum management class 2 transverse balance requirementThe transverse balance of spectrum management class 2 TU-C and TU-R equipment shall be measuredover the applicable frequency range using the procedures and 135-ohm measurement configurationspecified in clause 6. The transverse balance of spectrum management class 2 TU-C and TU-Requipment exceed the values in table 2 over the entire range of frequencies between the upper and lower–20 dB points of the signal pass-band.

5.2.2.3 Spoectrum management class 2 restrictions and loop assignment guidelines

Spectrum management class 2 DSL transmission systems shall use non-loaded loop facilities. In order tobe compatible in the same binder group as guarded systems, non-repeatered SM class 2 DSLtransmission systems shall use loop facilities that have a working length TBD or less. To assureacceptable performance of guarded systems, loop length restrictions may be needed for this spectrum


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management class. The specification of loop length restrictions for this class is expected to be provided ina future version of this standard.

5.2.3 Spectrum management class 3 (formerly MB)Spectrum management class 3 is intended for DSL transmission systems that operate in the frequencyspectrum up to about 370 kHz.


Spectrum management class 3 TU-C and TU-R equipment shall meet the PSD conformance criteria insection 6 using the PSD template described in table 4. At frequencies at or below 1.05 MHz, linearinterpolation of the frequency and PSD entries of table 4 is used to define the template. At frequenciesabove 1.05 MHz, the template is -143-10log10(f1.5/1.134x1013). The template is shown graphically infigure 5.

The total average power below 370 kHz that is transmitted by spectrum management class 3 TU-C andTU-R equipment shall be 14.0 dBm or less.




Spectrum management class 3 DSL transmission systems shall use non-loaded loop facilities. Basedupon a conservative model using the 1% Unger near-end crosstalk coupling model (see figure A.1), with24 disturbers, 6 dB margin, co-located CPE, and the assumed acceptable performance objectives for theguarded systems, the loop length restriction limit for spectrum management class 3 is CSA reach.3 This isa provisional value and may be modified in a future version of this standard.

5.2.4 Spectrum management class 4 (formerly HBS)

Spectrum management class 4 class is intended to include standard compliant HDSL2 equipment andother DSL transmission systems that have TU-C equipment that operates in the frequency spectrum up toabout 440 kHz and TU-R equipment that operates in the frequency spectrum up to about 300 kHz.


Spectrum management class 4 TU-C equipment shall meet the PSD conformance criteria in section 6.2.4using the downstream PSD mask described in table 5 and figure 6. Spectrum management class 4 TU-Requipment shall meet the PSD conformance criteria in section 6.2.4 using the upstream PSD maskdescribed in table 6 and figure 7.

–––––––3 CSA-reach is defined as a loop distance that meets Carrier Serving Area (CSA) length guidelines but not the CSArestrictions on bridged tap and the number of different gauges. Thus, the working length of a CSA-reach loop iswithin CSA range (9 kft of 26 AWG or 12 kft of 24, 22, or 19 AWG) but the length of the bridged tap and the totalcable length including bridged tap may exceed CSA guidelines. The working length of a CSA-reach multigauge cablethat contains 26 AWG cable may not exceed 12 kft minus the length of the 26 AWG cable in kft divided by three [12kft – (L26 ÷ 3) ]. Deployment is limited here on the basis of crosstalk impact. Bridged tap has very little effect on thepower of disturbing crosstalk. This is not the same as limiting the transmission range of a system based onperformance, which can be noticeably affected by bridged tap. Similarly, multiple gauge changes have very littleeffect on crosstalk power.


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The total average downstream power (into 135 Ohms) below 450 kHz that is transmitted by the spectrummanagement class 4 TU-C shall not exceed 17.3 dBm. The total average upstream power (into 135Ohms) below 350 kHz that is transmitted by the spectrum management class 4 TU-R shall not exceed17.0 dBm.




Spectrum management class 4 DSL transmission systems shall use non-loaded loop facilities. Basedupon a conservative model using the 1% Unger near-end crosstalk coupling model (see figure A.1), with24 disturbers, 6 dB margin, co-located CPE, and the assumed acceptable performance objectives for theguarded systems, the loop length restriction limit for the spectrum management class 3 is CSA reach.3

This is a provisional value and may be modified in a future version of this standard.

5.2.5 Spectrum management class 5 (formerly HBA)

Spectrum management class 5 s intended for DSL transmission systems that have TU-C equipment thatoperates in the frequency spectrum from about 138 kHz to about 1104 kHz and TU-R equipment thatoperates in the frequency spectrum from about 25 kHz to about 138 kHz.


Spectrum management class 5 TU-C equipment shall meet the PSD conformance criteria in section 6using the reduced-NEXT downstream PSD mask defined in Annex F of T1.413-1998 and TR-59.Spectrum management class 5 TU-R equipment shall meet the PSD conformance criteria in section 6using the upstream PSD mask defined in T1.413-1998 and TR-59.

The total average downstream power between 138 kHz and 1104 kHz that is transmitted by the spectrummanagement class 5 TU-C shall not exceed 19.9 dBm.

The total average upstream power below 138 kHz that is transmitted by the spectrum management class5 TU-R shall not exceed 12.5 dBm.


The transverse balance of spectrum management class 5 TU-C and TU-R equipment shall be measuredover the applicable frequency range using the procedures and 100 ohm measurement configurationspecified in clause 6. The transverse balance of spectrum management class 5 TU-C and TU-Requipment shall exceed the values in table 2 over the entire range of frequencies between the upper andlower –20 dB points of the signal pass-band


Spectrum management class 5 DSL transmission systems shall use non-loaded loop facilities. Non-repeatered spectrum management class 5 systems may be assigned to pairs that are in the same bindergroup as any of the guarded systems except for T1. In order to assure compatibility with T1, Spectrummanagement class 5 DSL transmission systems and T1 systems should be assigned to pairs that are indifferent binder groups whenever possible.

Spectrum management class 5 systems shall not be deployed in the following modes:

- Overlapping downstream PSD mode defined in T1.413 that allows the TU-C to transmit significantdownstream power in the 25 kHz to 138 kHz frequency band.

- Power boost mode described in the first version of the ADSL standard (ANSI T1.413-1995).


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- Transceivers located at the customer end of the loop transmitting in the downstream frequencyband (188-1104 kHz). This does not preclude adjacent colocation configurations, but suchconfigurations should use a dedicated binder.

5.2.6 Spectrum management class 6 (formerly VHB)

Spectrum management class 6 is intended for DSL transmission systems that operate in the frequencyspectrum up to about 10 - 20 MHz.


Spectrum management class 6 TU-C and TU-R equipment shall meet the PSD conformance criteria insection 6 using a PSD template (or templates) that are TBD. The spectrum management class 6 PSDtemplate should be based on emerging VDSL standards, which were not completed in time for this issueof this standard. Spectrum management class 6 should be frequency-division duplex (FDD), with distinctPSD templates for upstream and downstream transmission. There may also be distinct PSD templates forsymmetric spectrum management class 6 systems and for asymmetric spectrum management class 6systems.

The total average power that is transmitted by spectrum management class 6 TU-C and TU-R equipmentshall be 11.5 dBm or less.


The transverse balance of spectrum management class 6 TU-C and TU-R equipment shall be measuredover the applicable frequency range using the procedures and 100 ohm measurement configurationspecified in clause 6. The transverse balance of spectrum management class 6 TU-C and TU-Requipment shall exceed the values in table 2 over the entire range of frequencies between the upper andlower –20 dB points of the signal pass-band. Above 3 MHz, the transverse balance requirement is TBD.


Spectrum management class 6 DSL transmission systems shall use non-loaded loop facilities. Unlikeother DSLs, spectrum management class 6 systems were created to offer high bit rates over short rangeswhen deployed from remote fiber-fed terminals, pedestals, or cases. Deployment restrictions for spectrummanagement class 6 systems shall address both CO and remote deployments.

5.2.7 Spectrum management class 7 (formerly SRVHBS)

Spectrum management class 7 is intended for DSL transmission systems that operate in the frequencyspectrum from 0 to about 776kHz.

5.2.7.1 The seventh class PSD and total average power limitation

The seventh class TU-C and TU-R equipment shall meet the PSD conformance criteria in section 6 usingthe PSD template described in table 7 and figure 8.

The total average power below 776kHz that is transmitted by the seventh class TU-C and TU-Requipment shall be 14.0dBm or less.

5.2.7.2 Class 7 transverse balance requirement

The transverse balance of spectrum management class 7 TU-C and TU-R equipment shall be measuredover the applicable frequency range using the procedures and 100 ohm measurement configurationspecified in clause 6. The transverse balance of spectrum management class 6 TU-C and TU-Requipment shall exceed the values in table 2 over the entire range of frequencies between the upper andlower –20 dB points of the signal pass-band.

5.2.7.3 Class 7 deployment restrictions and loop assignment guidelines

The seventh class symmetric DSL transmission systems shall use non-loaded loop facilities. In order to becompatible with in the same binder group as guarded systems, non-repeatered seventh class symmetric


18

DSL transmission systems shall use loop facilities that have working length equal to or less than 7kft(provisional) of 26 AWG ,or equivalent.

Note: The ITU-T currently has a project (G.shdsl) that addresses data rates similar to those intended for thisclass. It is expected that this class will be superceded by a new one that reflects the outcome of that effort.When the new class is defined in a future version of this standard, it is expected that any new deployments usingthis class (Class 7) will not be compliant with the new version of this standard.

6. Conformance testing methodology

The conformance testing methodology in this clause shall be used to determine compliance with thesignal power limitations and transverse balance requirements in clause 5.

6.1 General conformance criteria

The conformance testing methodology is designed for the purpose of lab evaluation of the compliance ofequipment to the SM classes defined in section 5. A SM class is defined by its SM template, and theassociated SM mask is the SM template plus 3.5 dB.

The first step in the testing process is measurement of the transmitted PSD of the equipment under test.The appropriate termination impedance and resolution bandwidth will a function of the SM class, andtherefore defined in sections 6.2 – 6.7. The result of the PSD measurement, in units of dBm/Hz and at acenter frequency of BWn × , is denoted by ( )nPa , where rBW is the resolution bandwidth in kHz for the

SM class in question. The range of n is rBWn 300001 ≤≤ .

Conditions for compliance:

Compliance is achieved by meeting the following conditions:

a) For rBWn 300001 ≤≤ , ( ) ( )nPnP Ma ≤ , where ( )nP is the PSD of the SM mask at

frequency rBWn × , and ( ) ( ) dB5.3+= nPnP TM .

b) ( )( ) 1

1log10

1

10 ≤

×× ∑

−+

=

lm

mn T

a

nP

nP

ldB for all 1300001 +−≤≤ lBWm r and rBWl 100= . In other words,

the average over the measured PSD normalized by the template for the number of points equivalentto 100 kHz must be less than the 1 dB.

c) The total power of the transmitted PSD shall be no greater than the total power limit for that SMclass, as defined in sections 6.2-6.7.

d) The transverse balance of the associated TU-C and TU-R shall be greater than or equal to therequirement for that SM class, as defined in section 5.

NOTE: numerical values 100 kHz and 3.5 dB are TBD.

6.2 PSD conformance criteria unique to specific spectrum management classes

6.2.1 Specific conformance criteria for spectrum management class 1

There are no specific PSD conformance criteria for the Spectrum Management Class 1.



6.2.3 Spectrum management class 3



19


The general PSD conformance criteria in section 6.1 does not apply the spectrum management class 4. APSD mask is specified for the spectrum management class 4 instead of a PSD template. A member ofspectrum management class 4 shall have a measured PSD that shall not exceed the PSD mask that isspecified for spectrum management class 4 in tables 5 and 6 and figures 6 and 7 at any frequency. Amember spectrum management class 4 shall also meet the total average power limitations, transversebalance requirement, and deployment restrictions defined in 6.5 of this standard as well meeting as allother applicable requirements in this standard.

6.2.5 Specific conformance criteria for the spectrum management class 5

There are no specific PSD conformance criteria for spectrum management class 5.

6.2.6 Specific conformance criteria forspectrum management class 6




6.3 PSD and total average power measurement procedure

The test methodology for measuring the PSD and the total average power of a device under test (DUT)are defined in this subsection. For each spectrum management class, there are two different transmitPSD test cases:

a) Downstream (CO to Remote) transmission: the measured output of a central office transmissionunit (TU-C).

b) Upstream (Remote to CO) transmission: the measured output of a remote transmission unit (TU-R).

A DUT shall have total average power and PSD measured as described in this subsection in both theupstream case and the downstream case in order to determine compliance with the total average power,PSD conformance test, and other applicable conditions of a spectrum management class as defined inthis standard. Unless otherwise stated, all specifications apply to both the upstream case and thedownstream case. All measurements are performed directly at the transmitter output of the DUT with noadditional attenuation.

6.3.1 Test circuit for PSD and total average power measurement

A test setup as pictorially shown in figure 9 shall be used for measuring total average power and PSD. Anexample of a specific embodiment of this test setup is the circuit in figure 10. VOUT is connected to a high-impedance wideband rms voltmeter or spectrum analyzer. The PSD may be tested while line powered orlocally powered as required by the intended application of the DUT.

If the DUT is line powered then the test circuit shall contain provisions for DC power feed. If the DUT is notline powered then the DC power-feed circuitry may be omitted from the test circuit. For line poweredapplications, if the DUT is a TU-C the test shall be performed with the line power supply activated and anappropriate DC current sink (with high AC impedance) attached to the test circuit. If the DUT is a TU-R thetest shall be performed with power (DC voltage) applied at the line interface (TIP/RING) by an externalvoltage source feeding through an AC blocking impedance. Note that the DC current source/sink mustpresent a high impedance (at signal frequencies) to common ground. The test circuit contains provisionsfor transformer isolation for the measurement instrumentation. Transformer isolation of theinstrumentation input prevents measurement errors from unintentional circuit paths through the commonground of the instrumentation and the DUT power feed circuitry. When the termination impedance of thetest circuit seen by the DUT output meets the calibration requirements defined in 6.3.2 the test circuit willnot introduce more than ± 0.25 dB error with respect to a perfect test load of exactly the specifiedresistance.


20

The DUT shall be measured by equipment that is not synchronous with the transmitted symbols of theDUT, and there shall be no synchronization between the measurement equipment and the DUT. This is toavoid making an inaccurate measurement because of the effects of cyclostationarity.

6.3.2 Calibration of the test circuit and termination impedance

The nominal termination impedance of the test circuit as seen by the DUT output shall be resistive with aresistance of R Ohms as specified in table 8 for the appropriate spectrum management class. Theminimum return loss with respect to the termination impedance R over the frequency band of 1 kHz to 5MHz shall be 35 dB from 10 kHz to 2 MHz with a slope of 20 dB/decade below and above these cornerfrequencies for measuring a DUT for conformance with Spectrum Management Classes 1, 2, 3, 4, and 5.

The minimum return loss with respect to the termination impedance R over the frequency band of 1 kHz to30 MHz shall be 35 dB from 10 kHz to 20 MHz with a slope of 20 dB/decade below and above thesecorner frequencies for measuring a DUT for conformance with spectrum management class 6.

Note: 35 dB return loss will allow ±0.20 dB measurement error with respect to the nominal terminationimpedance value, R.

6.3.3 Operation of the DUT

The DUT shall be tested while it transmits the maximum power, and the maximum PSD levels at allfrequencies, at which it can transmit data when deployed. The DUT shall not have any power cutbackenabled. The DUT shall be tested under steady state conditions, after all start-up and initializationprocedures have been completed and while the DUT is transmitting data. To ensure that the DUT is in asteady-state condition, while undergoing test the DUT shall not have measured total average powers inany distinct 1.25 millisecond time intervals that differ by more than 8 dB. Although specific measurementsof average power and PSD during start-up and other non-data transmission phases are not provided, aDUT that transmits inordinately high power or PSD levels during these phases may be considered to be innon-compliance with this standard.The DUT input shall consist of a pseudo-random uniformly distributeddata sequence, and the DUT output shall be a fully modulated transmit signal with all overhead, framing,coding, scrambling, modulation, filtering and all other operations performed on the data stream that themodem would normally perform while transmitting data.

6.3.4 Total average power measurement procedure

The average power of a DUT shall meet the total average power requirements as specified in Section 5 ofthis standard over the bandwidth specified in Section 5 of this standard for conformance with a spectrummanagement class. The total average power may be tested while line powered or locally powered asrequired by the intended application of the DUT. The total average power shall be measured andaveraged over a time span of at least 10 seconds.

6.3.5 Power spectral density (PSD) measurement procedure

6.3.5.1 PSD resolution bandwidth

PSDs are recorded by averaging the observed output power of the DUT on each of a number ofcontiguous, regularly-spaced, small frequency bands; with each frequency band having a definedresolution bandwidth. The PSD of a DUT that is measured for conformance with Spectrum ManagementClasses 1, 2, 3, or 4 shall be recorded with frequency spacing equal to the resolution bandwidths specifiedin table 9 at all frequencies from 1 kHz to 30 MHz.

The PSD of a DUT that that is measured for conformance with spectrum management class 5 shall berecorded with frequency spacing equal to the resolution bandwidths specified in table 10 at all frequenciesfrom 1 kHz to 30 MHz.

The PSD of a DUT that that is measured for conformance with spectrum management class 6 shall berecorded with frequency spacing equal to the resolution bandwidths specified in table 11 at all frequenciesfrom 1 kHz to 30 MHz.


21

6.3.5.2 PSD Measurement time duration

Each frequency point (corresponding to a measurement in single resolution bandwidth) of a PSD shall bemeasured by averaging the power in the resolution bandwidth of that frequency point for a time period ofat least 2.0 seconds. This requirement is equivalent to setting the sweep time for a single sweep of aspectrum analyzer for a duration equal to at least 2.0 seconds per frequency point.

Note: this requirement is based on a statistical derivation that showed that to measure the average powerin a given resolution bandwidth within 0.1 dB accuracy with 99% confidence required observation of about9,000 transmitted symbols, and the slowest common signal is an ADSL tone which is at a 4 kHz rate.Measuring an entire PSD for 2.0 seconds in all of each of the resolution bandwidths in tables 9, 10, and 11requires a minimum observation time of 44 minutes.

6.4 Short-term stationary conformance criteria

6.4.1 Determination of whether to apply short-term stationary conformance criteria

The short-term stationary conformance criteria in clause 6.4.2 through 6.4.4 shall be applied to a DUT ifthe total average power transmitted by the DUT in any two non-overlapping 1.25 millisecond timeintervalsseparated by less than 60 seconds can differ by more than 8 dB. This includes variation due tothe presence or absence of input data for transmission or the presence of specific input data sequences,but does not include variations due to external stimuli such as the application of externally controlledpower management, externally initiated retrain, or a change in crosstalk levels or loop conditions whichcauses automatic retrain.

Equipment to which short-term stationary criteria are applied shall transmit at TBD dB below the SMmask. In addition, the short-term stationary transmitter shall continuously transmit in the ON condition fora minimum of 500 µsec.

6.4.2 Continuous mode for conformance testing

Equipment to which short-term stationary conformance criteria are applied shall provide a testconfiguration in which the transmitter remains in the ON condition continuously. In the ON condition, theDUT shall transmit the maximum power, and the maximum PSD levels at all frequencies, at which it cantransmit data when deployed. The DUT shall not have any power cutback enabled. The DUT shall nothave measured total average powers in any distinct 1.25 millisecond time intervals that differ by more than8 dB, including variation due to the presence or absence of input data for transmission or the presence ofspecific input data sequences.

6.4.3 Frequency domain requirements

6.4.3.1 Continuous mode testing

Equipment to which short-term stationary conformance criteria are applied shall be tested in thecontinuous ON condition specified in clause 6.4.2 using the conformance testing methodology defined in6.1, 6.2, and 6.3.

6.4.3.2 Short-term stationary mode testing

Equipment to which short-term stationary conformance criteria are applied shall be tested with inputconditions which generates the most frequent mode transitions permitted by the equipment. Theconformance testing methodology shall be as defined in section 6.1, 6.2, and 6.3 with the followingexceptions:

- The requirement in clause 6.3.3 that the DUT shall not have measured total average powers in anydistinct 1.25 millisecond time intervals that differ by more than 8 dB shall be waived.

- Each frequency point (corresponding to a measurement in single resolution bandwidth) of a PSDshall be measured by averaging the power in the resolution bandwidth of that frequency point for atime period of at least 4.0 seconds. This requirement is equivalent to setting the sweep time for a


22

single sweep of a spectrum analyzer for a duration equal to at least 4.0 seconds per frequencypoint. This requirement is used in place of the requirement in section 6.3.5.2.

- The equipment vendor shall identify the input conditions necessary to generate the modetransitions for this test.

6.4.4 Time domain requirements

Equipment to which short-term stationary conformance criteria are applied shall transmit in the ONcondition for a cumulative total of 10 milliseconds minimum in any 4 second period. This requirement isintended to facilitate detection of crosstalk from short-term stationary equipment by other receivers withina defined time interval.

6.5 Transverse balance testing methodology

Transverse balance is a comparison of the voltage of a transmitted metallic signal to the voltage of anyresulting longitudinal signal. It is the ratio of the metallic voltage VM at any frequency (f) to the transversevoltage VL at frequency (f). The result in dB is expressed as:

( ) ( )[ ]fVfV LMLM 10Log20BalanceTransverse =−

where VM (f) = the metallic voltage applied across the tip and ring conductors of the port under test at anyfrequency (f) between F1 and F2 is from a balanced source with a metallic impedance ZM, and VL (f) = theresultant longitudinal voltage appearing across a longitudinal impedance ZL .

The greater the VM to VL ratio, the better the transverse balance of the transceiver unit and the lesslikelihood that it will contribute to a crosstalk interference problem.

When calibrating the testing arrangement, the source metallic voltage should equal VM volts for each DSLclass when a metallic termination of ZM is substituted for the equipment under test. The metallicimpedance (ZM) shall be either 100 or 135 ohms as specified in clause 5.

The applicable ZL , ZM , F1, F2, and VM values for each DSL class are summarized in table 12.

The minimum transverse balance requirements for the TU-C and TU-R equipment under test shall beequaled or exceeded for the range of applicable frequencies (from F1 to F2) at all 2-wire loop ports with allvalues of loop current that the port under test is capable of drawing when attached to the appropriate loopsimulator circuit.

An illustrative test configuration for transverse balance conformance testing is shown in figure 11.

The equipment under test, at the CO end, must meet the transverse balance requirements in table 12.The testing methods or equivalent are given in TIA/EIA TSB-31-B.

Table 2 provides a template to be used for the transverse balance requirements; the actual frequencyrange over which the requirements apply and to be included in testing is dependent on the system undertest.

Transverse balance testing shall only be performed over the range of frequencies included in the powerspectral density (PSD) applicable to the equipment under test and actually used in data transmission. Forthat purpose, all of the signal pass-band shall be included, between the upper and lower –20 dB points.

Transverse balance may be measured while the DUT is line powered or locally powered. If the DUT isline powered then the test circuit shall contain a dc voltage source. In such applications, if the DUT is aTU-C the test shall be performed with TU-C line power activated and an appropriate dc current sink (withhigh ac impedance) attached to the test circuit. If the DUT is a TU-R, the test shall be performed with theappropriate dc voltage source applied between the tip and ring conductors through an ac blockingimpedance. The dc current source or sink must present a high impedance (at signal frequencies) tocommon ground. In line powered applications, the test circuit shall contain provisions for isolation of themeasurement instrumentation from unintentional circuit paths through the common ground of theinstrumentation and the DUT power feed circuitry.


23

Table 1 - Spectrum management class 1 PSD template definition

Frequency Range, f (Hz) PSD Template (dBm/Hz)

0< f ≤ 25000 Hz -32.5

25000< f ≤ 76000 Hz

×−−

25000log35.105.32 10

f

76000< f ≤ 79000 Hz

−×−−

3000

760005.05.37

f

79000 < f ≤ 85000 Hz

−×−−

10000

69000log6.1938 10

f

85000< f ≤ 100000 Hz15000

85000442

−×−− f

100000 < f ≤ 115000 Hz15000

100000746

−×−− f

115000 Hz < f ≤ 120000 Hz -53

120000 Hz < f ≤ 225000 Hz

×−

000120log5553 10 ,

f-

225000 Hz < f ≤ 635000 Hz

×−

000225log7068 10 ,

f-

635000 Hz < f

× 13

23

10101341

log10143.

f - -

Table 2 – Minimum transverse balance template for the xTU-C

Frequency band Minimum transverse balance

200 Hz -12 kHz 40 dB

12 kHz - 1544 kHz 35 dB

1544 kHz - 3000 kHz 30 dB


24

Table 3 - Spectrum management class 2 PSD template definition

Frequency Range, f (Hz) PSD Template (dBm/Hz)

0 < f ≤ 238000 Hz ( )34-

000,1321

1

000,250

000,250sin

log1042

2

10

+×

×/ffð

fð

238000 Hz < f ≤ 335000 Hz -71

335000 Hz < f ≤ 625000 Hz

×−

000335log10571 10 ,

f-

625000 Hz < f

× 13

23

10101341

log10143.

f - -

Table 4. Spectrum management class 3 PSD template definition

Frequency (khz) PSD template(dBm/Hz)

0 -37

50 -37

125 -38

210 -41

310 -57

370 -73

550 -75

670 -85

750 -97

980 -98

1050 -102.75


25

Table 5 - PSD mask definition for downstream transmission from a spectrum management class 4TU-C

Frequency(kHz)

MaximumPower(dBm/Hz)

Frequency(kHz)


Frequency(kHz)


≤1 -54.2 280 -35.7 1000 -89.22 -42.2 375 -35.7 2000 -99.712 -39.2 400 -40.2 ≥3000 -108190 -39.2 440 -68.2236 -46.2 600 -76.2

Table 6 - PSD mask definition for upstream transmission from a spectrum management class 4TU-C

Frequency(kHz)


Frequency(kHz)


Frequency(kHz)


≤1 -54.2 220 -34.4 555 -102.62 -42.1 255 -34.4 800 -105.610 -37.8 276 -41.1 1400 -108175 -37.8 300 -77.6 ≥2000 -108

Table 7 – Spectrum management class 7 PSD template definition

Frequency(kHz)

Power(dBm/Hz)

Frequency(kHz)

Power(dBm/Hz)

Frequency(kHz)

Power(dBm/Hz)

0 -39 390 -42 1000 -77

100 -39 420 -43 1100 -80

150 -40 500 -51 2000 -107

300 -42 775 -77

Table 8 – Termination impedances

Spectrum management class Termination impedance R (Ohms)Class 1 135Class 2 135Class 3 135Class 4 135Class 5 100Class 6 100Class 7 TBD


26

Table 9 – Resolution bandwidth for measuring a DUT PSD for conformance with spectrummanagement classes 1, 2, 3, and 4.

Frequency Region resolution bandwidthf <= 10 kHz 1 kHz

10 kHz <= f <= 3.1 MHz 3 kHz3.1 MHz <= f <= 30 MHz 100 kHz

Note: Values above 10 kHz are TBD

Table 10 – Resolution bandwidth for measuring a DUT PSD for conformance with spectrummanagement class 5.


10 kHz <= f <= 3.1 MHz 10 kHz3.1 MHz <= f <= 30 MHz 100 kHz

Note: values above 10 kHz are TBD

Table 11 – Resolution bandwidth for measuring a DUT PSD for conformance with spectrummanagement class 6.


10 kHz <= f <= 20 MHz 10 kHz20 MHz <= f <= 30 MHz 100 kHz

Note: values above 10 kHz are TBD

Table 12 – Summary of transverse balance testing criteria

SM Class1

SM Class2

SM Class3

SM Class4

SM Class5

SM Class6

SM Class7

ZL 500/90 (1) 500/90 (1) 500/90 (1) 500/90 (1) 90 TBD TBDZM 135 135 135 135 100 TBD TBDF1 200 200 200 200 200 TBD TBDF2 109 238 370 450 3000 TBD TBDVM 0.367 V 0.367 V 0.367 V 0.367 V 0.316 V TBD TBD

NOTES:

Numbers in this table are under study.

(1): The longitudinal impedance (ZL) shall be 500 ohms for frequencies from 200 Hz to 12 kHz and 90ohms for frequencies above 12 kHz.


27

System ATransmitter

System AReceiver

First cablesection

Second cablesection+ +

System B NEXTnoise generator

System B FEXTand AWGNnoise generator

xTU-C xTU-R

Figure 1 – Configuration for evaluation of effect of NEXT and FEXT into downstream

System ATransmitter

System AReceiver

Test Loop++

System B NEXTnoise generator

System B FEXTand AWGNnoise generator

xTU-C xTU-R

Figure 2 – Configuration for evaluation of effect of NEXT and FEXT into upstream


28

Figure 3 - Spectrum management class 1 PSD template

-110

-100

-90

-80

-70

-60

-50

-40

-30

0 100 200 300 400 500 600 700 800

Frequency (kHz)

PS

D (

dB

m/H

z)

Figure 4 - Spectrum management class 2 class PSD Template

-110

-100

-90

-80

-70

-60

-50

-40

-30

0 100 200 300 400 500 600 700 800

Frequency (kHz)

PS

D T

em

pla

te (

dB

m/H

z)


29

0 200 400 600 800 1000 1200-110

-100

-90

-80

-70

-60

-50

-40

-30

Frequency (kHz)

PS

D (

dBm

/Hz)

Figure 5. Spectrum management class 3 PSD template

-100

-90

-80

-70

-60

-50

-40

-30

0 100 200 300 400 500 600 700 800 900 1000

Frequency (kHz)

PS

D (

dB

m/H

z)

Figure 6 - PSD mask for downstream transmission from a spectrum management class 4 TU-C


30

Figure 7 - PSD mask for upstream transmission from a spectrum management class 4 TU-R

Figure 8 – Spectrum management class 7 PSD template

-100

-90

-80

-70

-60

-50

-40

-30

0 100 20 0 300 40 0 5 00 600

Freq uen cy (kHz)

PS

D (

dB

m/H

z)


31

Deviceunder test (DUT)

Return loss

as per calibration

of the test circuit

Tip

Ring

1 uF (min)

1 uF (min)

Vout

DCcurrent

sink

ResistiveTermination, R Ohms

(ground isolatedinput)

Figure 9 – PSD and total average power measurement setup

Deviceunder test (DUT)

Return lossas per calibration

of the the test circuit

Tip

Ring

1 uF (min)

1 uF (min)

R

1:1(+/- 1%)

(To high-impedance

load)

Vout20 mH (min)

DCcurrentsink

Figure 10 – Example PSD and total average power measurement setup


32

TrackingGenerator

Note 3 R1

SpectrumAnalyzer

EL

1:1

3 pF

Note 1

EL = longitudinal voltage

R1 = longitudinal impedance (90 or 500h )

20 pF

Note 2

SpectrumAnalyzer

EM

EM = metallic voltage

Note 1: The 3 pF capacitor may be placed on either side of the test set, as required, to obtain proper balancing of the testb idNote 2: Use a 12.4 to 24.5 differentialt iNote 3: Effective output impedance of tracking generator should match the metallic impedance ZM.

R2

R2 = ZM - 50 Ω (R2 is used for calibration. ZM is 100 or 135 ohms.)

T1

S3

S3

S1S2

S2

EquipmentUnder Test

T1 = 100 Ω : 100 Ω wideband transformer with center taps.

Figure 11 – Illustrative test configuration for transverse balance conformance testing


33

Annex A(normative)

Evaluation of interference from new technologies into existing technologies

A.1 Goals and framework for evaluation

The goal of spectral compatibility analysis described in this section is twofold:

a) to provide tests to validate that new services technologies will not interfere with existingtechnologies, and

b) to allow sufficient flexibility to nurture innovation in new subscriber line transmission technologieswhich further maximize the utility of the copper loop plant.

To achieve both goals simultaneously, this section describes computations which may be performed onnew signals to demonstrate spectral compatibility with existing technologies.

Fitting within the spectrum management class PSD masks of the main body of this standard provide asimplified test for spectral compatibility.However, this test alone would preclude large classes of newtransmission schemes which are spectrally compatible, and would stifle creativity for providing copperaccess solutions. In order to nurture spectrally compatible innovation, this section describes a second,more complicated evaluation(Test #2) Which may be used to demonstrate compatibility technology bytechnology with guarded transmission technologies in the local loop. Test # 2 follows established industrypractices for demonstrating compatibility of new technologies during the definition of a standard. Thesepractices have been used successfully in the T1E1.4 working group for technical evaluation of services forHDSL, HDSL2, ADSL, and VDSL, and would be sufficient for demonstrating compatibility of newtechnologies. These analyses should be used to add to the spectrum management classes in thisstandard at later dates. When followed rigorously, such analyses may be used as the basis for agreementon spectral compatibility between parties sharing loop facilities, in the interim between updates of thisstandard. Such agreements would be entirely between the parties and are outside the scope of thisstandard.

A.2 Analytical Method:Detailed crosstalk margin evaluations

Detailed margin calculations are required to demonstrate spectral compatibility of new technologiesoutside of the established spectrum management categories.. These calculations are described in thissection and must be calculated for each technology which may be interfered with. Because sometechnologies are spectrally asymmetric, that is, use a different transmit spectrum in each direction,evaluations must be performed in both the upstream and downstream directions.

The use of this section establishes non-interference with existing technologies by comparing theperformance of existing technologies in the presence of the new technology with industry-standardreference performance levels in the presence of existing crosstalk. In this method, the establishedreference disturbers are replaced in equal number by the new technology under trial, and the performancemargin of the technology being disturbed by the new technology is compared to the established referencecase.

This section is organized as follows. The subsections of A.2 describe the margin calculations andmethodology for a variety of technologies. Subsequent sections give the transmission and performanceparameters, and reference performance levels associated with each existing technology. As newtechnologies become established, a subsection can be inserted into future versions of this standarddetailing the established performance benchmarks and method for calculating compatibility with the newtechnology.

There are four types of margin computations described in this section: DFE-based PAM signals (e.g.,2B1Q ISDN and HDSL), DFE-based QAM/CAP signals (e.g. CAP signals), DMT-based signals (e.g.,T1.413-1995 ADSL), and linear-equalization based signals (e.g., T1). Which computation is useddepends on the existing technology being effected, not the nature of the proposed technology.


34

A.2.1 DFE-based PAM signals (e.g., 2B1Q ISDN and HDSL)

Margin for DFE-based PAM technologies is computed using an Optimal DFE calculation for PAM:

dB))(_1(10log101

0SNR_reqdffSNRf

fbaudMargin

fbaud

−+∗⌡⌠=

where f_SNR(f) is the folded received signal-to-noise ratio, defined as:

∑−= ×+

×+×+=1

2

2

)(

|)(|)()(_

nnfbaudfN

nfbaudfHnfbaudfSfSNRf

and S(f) is the desired signal’s (e.g., ISDN or HDSL) transmit power spectral density, |H(f)|2 is themagnitude squared of the wireline loop transfer function, and N(f) is the total noise power spectral density(crosstalk plus background noise) computed as described above. SNR folding calculated out to 4 timesthe Nyquist rate (twice the baud rate) is sufficient for all current xDSL signals. If future signals use morebandwidth, they may require expansion of the range of n in the summation.

The C code in table A.1 computes the optimal DFE SNR for PAM signals, from given two arrayscontaining received signal and received noise power spectral densities. By using the following code andsubtracting the required SNR from the result, one can compute PAM DFE margins as described above.

A.2.2 DFE-based QAM/CAP signals

Margin for DFE-base CAP/QAM technologies is computed using an Optimal DFE calculation for QAM:

dB_))(_1(10log101

2/

2/⌡⌠ −+∗=

+

−

fbaudf

fbaudf

c

c

reqSNRdffSNRffbaud

Margin

where f_SNR(f) is the folded received signal-to-noise ratio, defined as:

∑= ×+

×+×+=3

0

2

)(

|)(|)()(_

nnfbaudfN

nfbaudfHnfbaudfSfSNRf

and S(f) is the desired signal’s transmit power spectral density, |H(f)|2 is the magnitude squared of thewireline loop transfer function, and N(f) is the total noise power spectral density computed as describedabove.

One important difference from the PAM calculation is that for QAM/CAP, S(f) = 0 for f < 0.

As in the PAM case, SNR folding is calculated out to 4 times the Nyquist rate, yet for QAM this is 4 timesthe baud rate. As for PAM, future signals which use more bandwidth may require expansion of the rangeof n in the summation.

Unlike PAM signals, it is important that the region of folding be sufficient to include any offset for thecarrier frequency of the QAM/CAP signal. This may be included either by changing the limits of integrationor by changing the limits on n in the SNR folding summation to adequately span the frequencies used bythe signal.

The C code in table A.2 computes the optimal DFE SNR for QAM/CAP signals, given two arrayscontaining received signal and received noise power spectral densities. By using the code in table A.2and subtracting the required SNR from the result, one can compute QAM/CAP DFE margins as describedabove.

A.2.3 DMT margin computations

DMT systems allocate bits to individual carriers based on the Shannon capacity of the tones. Margin forthese systems is determined by the determining the Shannon capacity (minus appropriate SNR gap, andplus coding gain), and then degrading the SNR at all frequencies until the capacity is equal to the desireddata rate. Capacity at an individual frequency is given by:


35

Γ+=

)(

|)(|)(1log)(

22

fN

fHfSfC

where S(f) is the received signal power spectral density at frequency f, |H(f)|2 is the magnitude squared ofthe wireline loop transfer function, N(f) is the noise power spectral density at the receiver, as before, and Γis the effective SNR gap, as above. For coded systems, SNR gap is defined as (9.75 - (effective codinggain)) dB . For the purposes of margin calculations, the effective SNR gap is increased by the desiredmargin, and is defined as Γ = 9.75 - (effective coding gain) + Margin (dB).

Total capacity for the DMT system is then computed by integrating C(f) across the frequency band usedby the DMT system. Some DMT systems have a minimum number of bits per tone, (such as T1.413-1995, T1,413-1998, and ITU-T G.992.2, all of which support a minimum of 2 bits/tone (MINBITS=2)). Incalculations for these systems, C(f) must be further limited not to exceed the prescribed maximum.

When computing DMT capacity, the resulting integration is conditional at each frequency:

∫=bandwidthDMT

dffCC )(' ,

where C’(f) = min(C(f), MAXBITS) if C(F)>MINBITS , and C’(f) = 0 if C(f)<MINBITS, and DMTbandwidth isthe frequency range used by the data carrying tones of the desired DMT signal. It is worth noting thatimplemented DMT systems go through a process of bit loading and adjustment of powers to each of thetones. However, studies have shown that margins achieved by such algorithms closely match thoseachieved by the less implementation dependent capacity calculation shown here.

The C-code in table A.3 computes DMT margins.

A.2.4 Margin computations for linear equalization systems (e.g., T1)

To be added later.

A.3 Compatibility with voicegrade services and technologies

A.3.1 Description of voicegrade services and technologies

Voicegrade services and technologies use the frequency spectrum from 0 to 4 kHz and often employvarious types of dc and ac signaling. There are several types of voicegrade signals and the impact ofcrosstalk interference varies depending upon the type of disturbed signal. For example, voice systemsare concerned about the subjective effects of background noise during silent intervals when no speech ispresent, whereas analog voiceband data systems are concerned about the signal-to-noise ratio duringdata transmission.

Voicegrade services and technologies transmit signals that can be placed into one of five generalcategories:

- speech signals

- single and dual tone signals

- low frequency signals

- digital data

- analog data.

A.3.1.1 Speech signals

Speech signals include live voice as well as recorded announcements. Most of the speech energy is inthe frequency range from 200 to 3400 Hz. The most sensitive speech receiver is the human ear. It hasbeen found that background noise during silent intervals when no speech is present is the most disturbingnoise to the average listener. Background noise is measured with a C-message weighting filter that


36

simulates the effects of the average human ear with a 500-type telephone set. A background noise levelof 20 dBrnC or less is considered to be acceptable.

A.3.1.2 Single and dual tone signals

Single and dual frequency tones are used as network control and addressing signals, call progresssignals, and alerting signals. Network control and addressing signals include dual-tone Multi-frequency(DTMF) signaling, multi-frequency (MF) signaling, single frequency (SF) signaling, and coin depositsignals. Call progress signals include dial tone, busy tone, reorder tone, audible ring, special informationtones, and receiver off-hook tone. Call waiting tone is an example of a single frequency alerting tone thatis used with a supplemental feature on analog access lines.

Single and dual-tone signals range in frequency from 440 Hz to 2600 Hz and require signal-to-noise ratioson the order of 16 to 28 dB for reliable detection.

A.3.1.3 Low frequency (< 100 Hz) signals

Ringing, maintenance signals, and subvoice data systems are examples of signals that use low (< 100Hz) frequencies. The actual frequency range of the various signals is from about 17 to 83 Hz. Thesesignals have a relatively high tolerance for noise compared to other voicegrade signals.

A.3.1.4 Digital data

Digital data subrates use voiceband frequencies. The lowest digital data rates are entirely within thevoiceband. Digital data at 2.4 kbps has nulls at 0 and 2.4 kHz with maximum power at 1.2 kHz. Digitaldata at 3.2 kbps has nulls at 0 and 3.2 kHz with maximum power at 1.6 kHz. Digital data rates at 4.8 kbpsand above use bandwidths that are wider than the 4 kHz voiceband. For example, the 4.8 kbps digitaldata signal has nulls at 0 and 4.8 kHz with energy concentrated at 2.4 kHz.

The maximum loop loss for digital data services is 31 dB between 135-ohm terminations at the frequencythat represents one-half of the data rate. The minimum signal-to-noise ratio that provides acceptableperformance is 20 dB.

A.3.1.5 Analog data

Several types of analog data are used in the loop environment. The most common types are:

- Low-speed frequency shift keying (FSK) associated with supplemental network features such asCalling Number Delivery, Calling Name Delivery, and Visual Message Waiting Indicator.

- Customer data using one of the ITU-T standards such as V.34 or V.90.

The network-originated FSK data messages associated with network supplemental features on analogaccess lines generally require a signal-to-noise ratio of at least 25 dB. V.34 modems require a signal tonoise ratio of 39 dB. V.90 modems are the most sensitive voiceband data modems requiring a 50 dBsignal-to-noise ratio to operate at the maximum speed. The high signal-to-noise ratio makes the V.90modem the most sensitive of all of the voicegrade technologies.

A.3.2 Voicegrade evaluation

Because of the subjective effects of speech crosstalk, particularly intelligible crosstalk, specialconsideration must be given to crosstalk between loops that carry speech signals. In addition, voicebandsignals that have narrow spectral characteristics also require complicated evaluations to determine thesubjective effects of single frequency crosstalk interference on a human listener. This standard does notprovide guidance for evaluating the subjective effects of speech crosstalk or single frequency interference.This standard assumes that the transmission system under evaluation is a DSL system that has spectralenergy that is dispersed across a portion of the voiceband and that the crosstalk noise from such asystem will have a Gaussian noise distribution.

The voicegrade spectral compatibility evaluation assumes that the V.90 modem is the victim technology. Ifthe DSL system under evaluation passes this evaluation, then it is unlikely that crosstalk interferenceproblems will result with the other, more robust, types of voicegrade systems.


37

It is convenient to evaluate V.90 performance in terms of the total crosstalk noise power that occurs in thefrequency band from 0 to 4 kHz.

A.3.2.1 Evaluation loop

The loop used for voicegrade evaluations shall be 15 kft of 26-gauge cable. This loop has a resistance of1250 ohms and a 1 kHz loss of 7 dB when terminated at each end with 900 ohms.

A.3.2.2 Reference crosstalk environment

Spectral compatibility evaluations that use the V.90 modem as the victim technology shall assume forty-nine disturbers in a 50-pair binder group.

A piece-wise linear crosstalk model is used for evaluations (see figure A.1 and table A.4). A simplified 49-disturber model which has 67 dB of loss at 20 kHz and a linear (log-log) slope of –4 dB per decade can beexpressed as:

NEXT49 = 10 log10[(f)2/5 ÷ 2.11 x 108]

where, (f ) is in Hz from 200 to 20,000.

A.3.2.3 Crosstalk noise and peak power levels computation

Evaluations shall be performed in both the upstream and downstream directions. The DSL system underevaluation shall be considered spectrally compatible with the V.90 modem, and voicegrade services andtechnologies in general, if the NEXT caused by 49-disturbers in the same binder group meets thevoiceband NEXT PSD and total voiceband NEXT noise objective. The DSL system under evaluation shallbe considered spectrally compatible with voicegrade services and technologies in general, if the NEXTcaused by 49-disturbers in the same binder group meets the voiceband NEXT PSD requirement and totalvoiceband NEXT noise requirement.

A.3.2.3.1 Voiceband NEXT PSD

The NEXT PSD at any frequency from 200 to 4,000 Hz caused by 49-disturbers on a victim pair in thesame binder group shall not exceed –97.5 dBm. To determine compliance, the 200 to 4,000 Hz PSD ofthe system under evaluation is passed through the 49-disturber crosstalk model. The resultant NEXTpower level for each frequency is compared to the requirement.

PSDD(f) + 10 log10[(f )2/5 ÷ 2.11 x 108] ≤ – 97.5 dBm per Hz

The voiceband NEXT PSD requirement is met by any DSL system that has a transmit PSD is less than-29 dBm/Hz across the frequency band from 200 to 4000 Hz.

If the voiceband NEXT PSD requirement is not met, the system under evaluation has failed todemonstrate spectral compatibility with the V.90 modem, and voicegrade systems in general.

A.3.2.3.2 Total voiceband NEXT noise limit

The total NEXT noise on a victim pair caused by 49-disturbers in the same binder group should notexceed –75 dBm (15 dBrn). To determine compliance, the 200 to 4,000 Hz PSD of the system underevaluation is passed through the 49-disturber crosstalk model at each frequency and the NEXT noise ateach frequency is then summed on a power basis. The resulting total voiceband NEXT noise is thencompared to the requirement.

[ ] dBm751011.2)()(log10 85/24000

200

10 −≤×÷×∫ dfffPSDD ;

where the PSD is expressed in linear units (e.g., mW/Hz).

This objective is met by any DSL system that has a transmit PSD that is less than –41 dBm/Hz across thefrequency band from 200 to 4000 Hz.


38

If the NEXT power level objective of ≤ -75 dBm is not met, the system under evaluation has failed todemonstrate spectral compatibility with the V.90 modem. In order to demonstrate spectral compatibilitywith voicegrade systems in general, the total NEXT noise in the frequency band from 1 to 4000 Hz on avictim pair caused by 49-disturbers in the same binder group shall not exceed –66 dBm (24 dBrn). Thisrequirement is met by any DSL system that has a transmit PSD less than –32 dBm/Hz across thefrequency band from 200 to 4000 Hz.

A.3.3 Spectral compatibility of voicegrade systems with guarded systems

The FCC has adopted rules and regulations in Part 68 for CPE in order to protect the network from harm.One of the harms recognized by the FCC is crosstalk interference. The FCC has adopted signal powerlimitations and longitudinal balance limitations to prevent crosstalk interference from being caused byvoicegrade CPE.

CPE that meets the voice or voiceband data signal power limitations in Part 68 will have spectralcompatibility with all of the guarded loop transmission systems listed in 4.3.

Likewise, network equipment that meets the encoded analog content specifications in Part 68 will havespectral compatibility with all of the guarded loop transmission systems listed in 4.3.

A.4 Compatibility with T1.410

ANSI T1.410-1992 (alternatively known as the Digital Data System, or DDS), operates at rates from 2.4kbps to 64 kbps, symmetrically, using simplex transmission over two non-loaded wire pairs. It is theprimary means for low rate connections for Frame Relay service, and is still quite popular, with over200,000 new installations each year. While 56 or 64 kbps service is primarily used for Frame Relay, therestill is a significant deployment of subrate (2.4, 4.8 or 9.6 kbps) service for automated teller machines andlottery networks.

T1.410 uses 50% duty-cycle AMI transmission, similar to that of T1. The main lobe of the transmitspectrum lies in the frequencies between 0 and the signaling rate, with the peak at ½ the bit rate. Asspecified in the standard, the transmit filter is 1st order, with a 3 dB point at 1.3 times the signaling rate (atrates below 19.2 kbps, some additional filtering is present.) Maximum transmit power is 6 dBm into 135Ohms, except at the 9.6 kbps rate, where the transmit power is limited to 0 dBm (both number computedfor equi-probable 0s and 1s, since T1.410 does not employ data-randomizing scramblers). For singlechannel service up to 56 kbps, the signaling rate is the same as the service rate. For a service rate of 64kbps, the signaling rate is 72 kbps. Optionally, at rates of 56 kbps and below, a secondary channel ispresent, which increases the signaling rate by approximately 30%.

T1.410 specifies that transceivers operate on loops where the insertion loss at the 1/2 the signalingfrequency is 34 dB. Additional loop deployment practices limit the length of bridged taps that can bepresent on the line. At rates below 19.2 kbps, single and total bridged tap lengths are limited to 6 kft. Atrates of 19.2 kbps and above, the total bridged tap length is limited to 2.5 kft with no single bridged tapexceeding 2.0 kft.

A.4.1 Computation of DDS Performance – Margin Computation for AMI Transceivers

DDS uses AMI transmission with a 50% duty cycle. Historically, the receivers have used a rather simplestructure which incorporates a linear equalizer with only a single zero and a 3rd order lowpassfilter (See[1]).

The optimal (from a minimum mean squared error perspective) linear receiver for a 50% duty cycle pulsecan be obtained through the procedure described in [2]. For DDS, the resulting equalized channelresembles a 60% raised cosine channel, which rolls off much faster than the third order lowpass filtersuggested in [1].

Since bipolar violations are used as control codes, the DDS receiver is not able to fully exploit thecorrelation in the AMI signal for maximum performance. To derive the optimum receiver margin, weassume a 2 level signal, and then increase the required SNR to compensate for the power differencebetween the AMI and 2 level signals. (In fact, the result is nearly the same as we get if the correlation istaken into account.) Starting from the work in [1], we can obtain


39

,1)()(

)(1 2/

2/df

fLfM

fM

fbaudMSE

fbaud

fbaud∫− +=

where L(f) is defined as:

∑∞

−∞=

×−×−

×=

n

nfbaudfHnfbaudfG

nfifbaudNfL

)()(

2()(

and M(f) is spectrum of the message sequence (sin 2 shaped for AMI.) G(f) represents the transmit pulseshape; for AMI that includes the 50% duty cycle and any other filtering (1 st order for DDS). H(f) is thechannel response, and N(f) is the noise spectrum. When M(f) is a constant (M0), this can be reduced tothe familiar margin equation for linear equalization:

dB_1)(_

1110log10arg

2/

2/reqSNRdf

fSNTffbaudinM

fbaud

fbauddB −

+

−= ∫−∗

and f_SNR, the folded SNR is given by

20

)(

)()(_ ∑

∞

−∞= ×−×−×−

=n

nfbaudfN

nfbaudfHnfbaudfGMSNRf

2

)(

)()(∑∞

−∞= ×−×−×−

=n

nfbaudfN

nfbaudfHnfbaudfS.

To account for the transmit power increase caused by the AMI correlation, we increase the required SNRby 3 dB (the power difference for a ternary signal compared to a binary signalwith the same levelseparation.) Then for a 10-7 error rate, the SNR_Required for a pseudo-optimum AMI receiver isapproximately 17.3 dB.

Since actual receivers have additional impairments (mis-equalization, timing jitter, etc.), the actualrequired SNR is often higher than the 17.3 dB listed here. In addition, since the actual receive filters don’troll off as fast at the optimal receiver, additional noise power may reach the decision device, reducing theactual SNR from that theoretically calculated. These conditions noted, we present the optimal calculationas the basis for the relative performance measures to be used in this section.

A.4.2 Evaluation loops

The maximum metallic loop loss for T1.410 is 34 dB at ½ the signaling frequency (Nyquist frequency.)Loop loss shall be calculated assuming 135 ohm terminations. Because DDS transceivers use linearequalization, both upstream and downstream scenarios use the worst case loops listed below:

For the 56 kHz signaling rate, the Nyquist frequency is 28 kHz. ANSI T1.601 test loop 6 is representativeof a worst case loop, and is used for 56 kbps evaluation.

For the 72 kHz signaling rate, the Nyquist frequency is 36 kHz. ANSI T1.601 test loop 10 is representativeof a worst case loop, and is used for 64 kbps evaluation.

For the 9.6 kHz signaling rate, the Nyquist frequency is 4.8 kHz. 27 kft of 26 AWG is representative of a34 dB loop, and is used for 9.6 kbps evaluation.

A.4.3 Reference crosstalk environment

T1.410 is deployed today in the same loop plant with T1.601 ISDN. ISDN is the worst existing disturber forDDS. To assess the effect of crosstalk from new technologies on DDS, a relative comparison will be madewith ISDN crosstalk. If a new technology produces the same or higher margins than that obtained with


40

ISDN crosstalk, then it is deemed compatible with DDS. ISDN crosstalk will be represented by the SMclass 1 template.

Since DDS only transmits on 1 of 2 pairs in use, spectral compatibility studies that use DDS as thedisturber technology should assume 24 disturbing DDS systems in a 50-pair binder group.

The reference crosstalk environment against which new technologies will be compared is 49 SpectrumManagement Class 1 disturbers. The two-piece Unger model for NEXT described in figure A.1 and tableA.4 is to be used for crosstalk into DDS due to the low frequency nature of the signal.

A.4.4 Margin computation

DDS relative margin is computed as described above (A.X.1) for AMI signals. For the new technology tobe considered spectrally compatible with DDS, the following scenarios must produce margins no lowerthan that computed using the same loop and noise-coupling models with 49 spectrum management class1 disturbers:

a) 49 new technology NEXT/FEXT

b) 24 SM class 1 NEXT/FEXT + 24 new technology NEXT/FEXT.

DDS evaluations at 9.6 kbps and 64 kbps should be sufficient to ensure spectral compatibility with all DDSrates.

Required SNR ( SNR_req) for DDS is 17.3 dB. The transmit signal spectrum used in the calculation is thatof a 50% duty cycle bipolar signal, balanced about DC (50% positive pulses, 50% negative pulses) andpassed through a 1 pole filter with 3 dB point at 1.3 times the signaling rate. Transmit power is 6 dBm for56/64 kbps, and 0 dBm for 9.6 kbps. A frequency resolution of approximately 100 Hz (FDELTA=100 Hz)should be used for 56/64 kbps DDS margin calculations and 20 Hz for 9.6 kbps DDS margin calculationsdue to the narrow bandwidth of the signal.

A.5 Compatibility with ISDN DSL

Using the transmit spectrum for ISDN described in Annex B of T1.413-1995, spectral compatibility withISDN is verified by performing an Optimal DFE margin calculation for DFE-based PAM signals todetermine ISDN margin in the presence of the proposed signal. The remainder of this section defines thetest parameters.


Upstream Direction: Since ISDN DSL uses spectrally symmetric echo-canceled transmission, in theupstream a worst-case near-end crosstalk event would occur when the ISDN loop is longest and the newtechnology is crosstalking into the ISDN signal. ANSI T1.601 Loop 1 (18 kft comprised of 16.5 kft 26AWGand 1.5 kft 24AWG) should be used for this test.

Downstream Direction: Evaluation should be performed on the shorter of either (a) the longest singlelength of 26 AWG copper that the proposed technology will run on, or (b) ANSI T1.601 Loop 1 (as for theupstream).

A.5.2 Reference Crosstalk environment

The reference crosstalk environment against which new technologies will be compared is:

49 Spectrum Management Class 1 template (self-NEXT) disturbers.

The two-piece Unger model for NEXT described in figure A.1 and table A.4 is to be used for crosstalk intoISDN due to the low frequency nature of the ISDN signal.

A.5.3 Margin Computation

ISDN DSL margin is computed as described for DFE-based PAM signals. The computed margin for ISDNagainst the proposed technology as a disturber should be compared against a calculation using the sameloop and noise coupling models for:

a) 49 new technology NEXT/FEXT, or


41

b) 24 SM class 1 Template NEXT/FEXT + 24 new technology NEXT/FEXT.

Spectral compatibility requires that computations for the maximum allowable numbers of the proposedtechnology (based on self-crosstalk limitations) disturbing ISDN produce no lower ISDN margins than 49ISDN DSL NEXT. Required SNR (SNR_req) for ISDN is 21.5 dB. The baud rate for the 2B1Q ISDNsignal is 80 kHz. Model resolution of approximately 100 Hz (FDELTA=100 Hz) should be used for ISDNmargin calculations due to the narrow bandwidth of the signal.

A.6 Compatibility with HDSL

Using the transmit spectrum for HDSL described in Annex B of T1.413-1995, spectral compatibility withHDSL is verified by performing an Optimal DFE margin calculation for DFE-based PAM signals todetermine HDSL margin in the presence of the proposed signal. The remainder of this section defines thetest parameters.


Upstream Direction: In practice, CSA4 has been shown to be a greater impediment to HDSL transmissionthan the longest loops (CSA6 and CSA8). CSA4 should be used for margin evaluation.

Downstream Direction: Evaluation should be performed on the shorter of either (a) the longest singlelength of 26 AWG copper that the proposed technology will run on, or (b) CSA4.

A.6.2 Reference crosstalk environment

The reference crosstalk environment against which new technologies will be compared is:

49 SM class 3 template disturbers.

Either the simplified T1E1 NEXT model or the two-piece Unger model may be used for crosstalkevaluation. See figure A.1 and table A.4.


HDSL margin is computed as described for DFE-based PAM signals. The computed margin for HDSLagainst the proposed technology as a disturber should be compared against a calculation using thesameloop and noise-coupling models for


b) 24 SM class 3 template NEXT/FEXT + 24 new technology NEXT/FEXT.

Spectral compatibility requires that computations for the maximum allowable numbers of the proposedtechnology (based on self-crosstalk limitations) disturbing HDSL produce no lower HDSL margins than 49SM class 3 template NEXT. The baud rate for the HDSL signal is 392 kHz. Required SNR (SNR_req) forHDSL is 21.5 dB. Model resolution of at least 500 Hz (FDELTA <= 500) should be used for the HDSLmargin calculations.

A.7 Compatibility with ADSL and RADSL technologies

ADSL compatibility is inherently more complicated than for single fixed-rate technologies. Compatibilitywith ADSL must consider different performance levels at differing loop reaches, as appropriate to thedeployment reach of the technology being evaluated as a disturber to ADSL.. This section addressesT1.413-1998, CAP/QAM RADSL and ITU Recommendations G.992.1 and G.992.2.

A.7.1 Evaluation loops and performance levels

4 performance classes of ADSL are determined:

a) 5440 kbits downstream, 640 kbits upstream at reaches up to 9 kft 26 AWG (Loop CSA 6).

b) 1720 kbits downstream, 176 kbits upstream at reaches up to 13.5 kft 26 AWG (ANSI T1.601Loop 7)

c) 1720 kbits downstream, TBD kbits upstream at reaches up to 12 kft 26 AWG.


42

d) 256 kbits downstream, 96 kbits upstream on T1.601 Loop 1 and on T1.601 Loop 2.

Downstream evaluation loops: For the downstream direction, evaluation will be on the shorter of either thelongest reach of the proposed system, or the reach of the desired ADSL performance level. In the casewhere the evaluation is limited by the reach of the proposed system, the performance level required ofADSL will be the next longer reach level (e.g., if the proposed system reaches 10 kft on 26 AWG, then theperformance level should be 1720 kbits downstream, 400 kbits upstream). In these cases, performanceat the limited reach is compared with margins given by the reference crosstalk environment at the targetedreach of the desired performance level (in the example, at 12 kft 26 AWG).

Upstream evaluation loops: For the upstream direction, evaluation need consider all four performancelevels, regardless of the reach of the technology being evaluated. In practice, however, meeting level 1,the more stringent of the three should be sufficient.

A.7.2 Reference crosstalk environments

Downstream:

Performance Class 1: (5440/640 kbits, CSA reach): 20 SM class 3 template NEXT/FEXT disturbers.

Performance Classes 2&3: (1720/176 kbps, 13.5 kft reach): 24 SM class 3 template NEXT/FEXTdisturbers.Performance class 4: (256/96 kbps, T1.601 loops 1 & 2): 10 Spectrum Management Class 1NEXT/FEXT disturbers.

Upstream:

Performance classes 1, 2 &3: 20 SM class 3 template NEXT/FEXT disturbers: 10

Performance class 4: Spectrum Management Class 1 template NEXT/FEXT disturbers

Either the simplified T1E1 NEXT model or the two-piece Unger model may be used for crosstalkevaluation. See figure A.1 and table A.4.


T1.413-1998, G.992.1, and G.992.2 ADSL Margins are computed as described for DMT signals.CAP/QAM RADSL margins are computed as described for CAP/QAM DFE signal.

For the purposes of these evaluations, the ADSL or RADSL transmit PSDs (and baud rates for RADSL)defined in the relevant standards or recommendations, should be used for the ADSL or RADSL signals.

Evaluations will be performed for each type of ADSL (T1.413-1998, (non-overlappedupstream/downstream spectra with the reduced NEXT transmit spectra of annex F), G.992.1 (also withnon-overlapped upstream/downstream spectra), G.992.2, and CAP/QAM, according to the parameterswithin the relevant standards).

The data rates and noise models from A.7.2 are used in the formulas in the formulas from A.2. Noadditional overhead is added to these rates; the results are relative, not absolute.

In order to reduce the sensitivity of this procedure to model accuracy, the computed margin for ADSL orRADSL with the proposed new technology as a crosstalker should be compared against a calculationusing the same models for the reference crosstalkers for each performance/reach class, rather thanagainst a particular fixed minimum specified performance margin; e.g., 6 dB.

Downstream

Performance class 1:



Performance classes 2 & 3:


b) 12 Spectrum Management Class 1 template NEXT/FEXT + 12 new technology NEXT/FEXT.


43




Upstream

Performance classes 1, 2, &3:






Spectral compatibility requires that computations for the same number of disturbers as in the referencecase (up to the maximum allowable number of the proposed technology based on self-crosstalklimitations) disturbing ADSL produce no lower ADSL margins than the reference cases. Model resolutionof at least 4 times the tone spacing of the DMT signal should be used for ADSL margin calculations.

A.8 Compatibility with AMI T1

The test for compatibility with repeatered AMI T1 assumes the following:

- The repeater section is operating with 3 dB of margin,

- The margin, after taking the proposed DSL system into account, must be at least 2.0 dB

- The loss of the first repeater section out of an office is assumed to be 22.5 dB at 772 kHz.

- The loss of subsequent sections is assumed to be 32 dB at 772 kHz.

Power-summing of margins, to obtain the 2.0 dB of margin, yields a required minimum margin, due to theDSL system(s) alone, of 9.0 dB.

It has been found empirically that, on a repeater span having a loss of 22.5 dB at 772 kHz, the maximumnoise that can be tolerated at the repeater input, while maintaining a BER of 10-7 , is -27.5 dBm. Themaximum noise due to DSL system(s) on the first repeater section out of an office, then, shall be equal toor less than –36.5 dBm (-27.5 – 9.0). Similarly, the maximum noise for a 32 dB span is -40.5 dBm. Themaximum noise due to DSL system(s) for subsequent repeater sections, then, shall be equal to or lessthan –49.5 dBm (-40.5 – 9.0).

When evaluating the noise coupled into the repeater, the following equation, developed via curve-fitting,shall be used to model the repeater input filtering.

012

23

34

45

5)( afafafafafadBGain +++++=

where f is in MHz and the coefficients for both the 22.5 and 32 dB sections are shown in table A.5

Defining C(f) as the 1% Unger two-piece model (see figure A.1 and table A.4) and Gain(f) as given above,the following conditions for compatibility with T1 carrier must be met:

dBmdffGainfCPSDdBSectionEndMHz

Disturber 5.36)()(:)5.22(544.1

0

−≤∗∗∫

dBmdffGainfCPSDdBSectionSpanMidMHz

Disturber 5.49)()(:)32(544.1

0

−≤∗∗− ∫


44

A.8.1 Evaluation Loops

To be added

A.8.2 Reference crosstalk environments

To be added


To be added

A.9 Compatibility with HDSL2

To be added

A.10 Combination of crosstalk sources: composite crosstalk model

See B.4.3.


45

Table A.1 - Code for DFE PAM SNR computation

/* OPTIMAL DFE PAM SNR computation */float pamsnr (

float *signal, /* array of received signal psd samples (resolution =FDELTA Hz)*/

float *noise, /* array of received noise psd samples (resolution =FDELTA Hz) */

int baud, /* PAM baud rate expressed in units of FDELTA (frequencyresolution) */

int end, /* Maximum number of frequency samples */int in_dB) /* FLAG: 1 = PSDs given in deciBels, 0 = PSDs given in

linear units */ int i,cnt; float snr,temp; temp = 0; i = 0; cnt = 0; while(i<end && i < baud)

if( in_dB == 1 ) if (2*baud-i) < end) temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+ pow(10.0,0.1*(signal[baud-i]-noise[baud-i]))+ pow(10.0,0.1*(signal[i+baud]-noise[i+baud))+ pow(10.0,0.1*(signal[2*baud-i]-noise[2*baud-i]))+ +1.0); ) else if (i+baud < end)

temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+ pow(10.0,0.1*(signal[baud-i]-noise[baud-i]))+ pow(10.0,0.1*(signal[i+baud]-noise[i+baud]))+ +1.0);

else if (baud-i < end) temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+ pow(10.0,0.1*(signal[baud-i]-noise[baud-i]))+ +1.0);

else temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+1.0);

else if (2*baud-i < end temp += log(signal[i]/noise[i]+signal[baud-i] / noise[baud-i] +signal[i+baud]/noise[i+baud] + signal[2*baud-i]/noise[2*baud-i] +1.0);

else if (i+baud < end) temp += log(signal[i]/noise[i]+signal[baud-i] /

noise[baud-i] + signal[i+baud]/noise[i+baud] +1.0); else if (baud-i < end) temp += log(signal[i]/noise[i]+signal[baud-i] /

noise[baud-i]+1.0); else temp += log(signal[i]/noise[i] +1.0); cnt ++;i++;

temp /= (float) cnt; snr=10.0*temp*log10(exp(1.)); return(snr); /* dB */


46

Table A.2 - Code for DFE QAM/CAP computation

/* OPTIMAL DFE QAM/CAP SNR computation */float qamsnr (

float *signal, /* array of received signal psd samples (resolution =FDELTA Hz)*/

float *noise, /* array of received noise psd samples (resolution =FDELTA Hz) */

int baud, /* PAM baud rate expressed in units of FDELTA (frequencyresolution) */

int end, /* Maximum number of frequency samples */int in_dB) /* FLAG: 1 = PSDs given in deciBels, 0 = PSDs given in

linear units */ int i,cnt; float snr,temp;

temp = 0; i = 0; cnt = 0; while(i<end && i < baud)

if( in_dB == 1 ) if (i+3*baud < end) temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+ pow(10.0,0.1*(signal[i+baud]-noise[i+baud]))+ pow(10.0,0.1*(signal[i+2*baud]-noise[i+3*baud]))+ pow(10.0,0.1*(signal[i+3*baud]-noise[i+3*baud]))+ +1.0);

else if (i+2*baud < end)temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+ pow(10.0,0.1*(signal[i+baud]-noise[i+baud]))+ pow(10.0,0.1*(signal[i+2*baud]-noise[i+2*baud]))+ +1.0);

else if (i+baud < end) temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+ pow(10.0,0.1*(signal[i+baud]-noise[i+baud]))+ +1.0);

else temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+1.0);

else if (i+2*baud < end) temp += log(signal[i]/noise[i] +

signal[i+baud]/noise[i+baud]+signal[i+2*baud]/noise[i+2*baud] +1.0);

else if (i+baud < end) temp +=log(signal[i]/noise[i] +

signal[i+baud]/noise[i+baud]+1.0); else temp += log(signal[i]/noise[i] +1.0); cnt ++;i++;

temp /= (float) cnt; snr=10.0*temp*log10(exp(1.));

return(snr); /* dB */


47

Table A.3 - Code for DMT margin computation

float dmtmrgn(float *signal, /* array of received signal psd samples (resolution =

FDELTA Hz)*/float *noise, /* array of received noise psd samples (resolution =

FDELTA Hz) */int rate, /* desired bit rate, expressed in units of bits per second

per FDELTA ) */int start, /* start of DMT bandwidth (sample number) */int end, /* end of DMT bandwidth (sample number) */int in_dB) /* FLAG: 1 = PSDs given in deciBels, 0 = PSDs given in

linear units */

int j, firstpass;float snr;float snr_margin ;float delcap, totcap ;

snr_margin = MAXIMUM_VALUE;firstpass = 1;snr_margin += MARGIN_STEP;do

snr_margin -= MARGIN_STEP;

/* Compute capacity */totcap = 0.;for (j = start; j < end; j++)

if (in_dB) snr = sig[j]-noise[j];else snr = 10.*log10(sig[j]/noise[j]);delcap = log(1. + pow(10., .1*(snr -snr_margin-SNRGAP))) / log(2);if (delcap > MAXBITS) delcap = MAXBITS;

if (delcap < MINBITS) delcap = 0;totcap += delcap;

if (totcap > rate && firstpass)

snr_margin +=10.; totcap=0.; else firstpass = 0;

while (totcap < rate);return (snr_margin);

SNRGAP, MAXBITS, MINBITS, are all adjusted based on the DMT system being evaluated.MAXIMUM_VALUE and MARGIN_STEP are control how fast and how accurately the routinecomputes margin. MAXIMUM_VALUE is the maximum margin of interest, the integrationbegins there. MARGIN_STEP defines the accuracy of the result.


48

Table A.4 – Data Points for Unger NEXT Model (see figure A.1)

Frequency in kHzNo. of

Disturbers 0.2 2 20 200 2000

1 88 82 76 61 46

10 80 75 70 56 42

49 74 70 66 52 38

Table A.5 – Coefficients for T1 repeater input filtering gain equation

Coefficient Value for 22.5 dB section Value for 32 dB section

A0 -12.91476008173899 -21.84038057235726

A1 15.74168401196194 40.22938541210919

A2 20.75952294972729 -2.99965401635352

A3 -36.60781681972960 -31.38386179570797

A4 13.09484055899603 18.63736172126514

A5 -0.91231176505002 -3.26384215013252


49

20000 Hz

30

40

50

60

70

80

90

100 1000 10000 100000 1000000

Frequency - Hz

1%

NE

XT

Lo

ss -

dB

Simplified Model for 49 Disturbers(57 dB at 80 kHz; -15 dB/Decade)

1

10

49

Number of Disturbers

-15

-15

-14

-15

-4

-5

-6

slope

slope

Notes:

1. Terminated with cable characteristic impedance Z0 at each frequency

2. NEXT disturbers in the same cable binder unit of 50 pairs

3. See table A.4 for data points of Unger NEXT model

Figure A.1 – Unger NEXT model and simplified NEXT model of 1% NEXT for 18kft of 22GA PIC


50

Annex B(normative)

Loop Information

B.1 General

B.1.1 The loop environment

There are 700 million metallic twisted pair cables delivering communications services to customersaround the world. It is predicted that most of this embedded base of copper wire will eventually bereplaced with wider bandwidth transmission media such as optical fiber and coaxial cable. However,twisted pair copper wire will be the main method of delivery for several years to come4) .

Recent years have seen the increase in demand for customer information bandwidth escalate dramaticallyfrom 3 kHz analog voice services to digital services requiring several megabits per second. Advances inintegrated circuit density, digital signal processing techniques and information compression algorithms areresulting in the introduction of ever higher bandwidth twisted pair transmission systems that can transportthese new services.

These new systems must fit into a outside loop transmission environment with several existingtransmission systems and other systems that may be introduced later. For the voice frequency servicesthe twisting of the wire pairs and construction of the cables such that no two pairs traveled together forvery long, helped to control crosstalk coupling. Interference between pairs was held to acceptable levels.As signal bandwidth increases, the crosstalk coupling between pairs increases at the same time as thetransmission loss increases making the circuits more susceptible to interference. Interference can comefrom other transmission systems of the same kind or from different type systems that overlap the signalspectrum.

The 1.5Mb/s T1 line system was originally developed for application in the intra-office cable plant whoseconstruction is very carefully controlled. To control crosstalk interference between T1 systems, T1 signalsin the two directions were placed on separate pairs located in different binder groups that had shieldsbetween them. T1 repeaters were spaced and placed to minimize differences in signal levels. When T1systems began to be deployed in the customer outside loop plant, the situation became much morechallenging. As will be described later, the loop plant is designed and constructed to deliver voice servicesto customers at acceptable quality and minimum cost. In recent years, many T1 lines have been deployedin the outside plant to deliver 1.5 Mb/s services to business customers. The engineering design andconstruction of these lines is a challenge in minimizing interference and cost.

Over the years several high bandwidth analog carrier systems were also deployed in the outside plant withmixed results for compatibility and interference.

The use of the loop plant to transport high rate digital signals was not envisioned at the beginning. Indeed,for over 100 years the loop plant has been optimized for the reliable delivery of voice frequency services atlowest cost and acceptable quality. In the last several years, the design of new loop plant has beenmodified slightly to ease the introduction of digital transmission.

As newer digital transmission systems have been developed for the loop plant, each one has beensubjected to hard scrutiny for potential interference with like systems. ISDN Basic Access digitalsubscriber line (DSL) systems had to account for other DSLs in the cable and existing systems like T1lines and Digital Data Service (DDS). In turn, high-bit-rate DSL (HDSL) had to show compatibility withDSL, T1 and DDS. Asymmetric DSL (ADSL) had to account for all of the above.

Development and deployment of these new transmission systems is very costly in time and money. Itwould be very desirable to predict how a system will perform before it is actually built. Testing a systemagainst all reasonable cases of interferers on all reasonable loop configurations is not feasible. To test the

–––––––4) As will be discussed later, not all telephony wire is made up of twisted pairs. Some single-pair aerial/overheaddrop wire uses parallel/flat/non-twisted conductors for lengths up to 700 feet. Also, not all of the wire is copper.Copperclad steel and copper-cadmium mixes are used where strength is needed in drop cables.


51

performance of a system and the mutual interference with other systems, a combination of analysis,simulation, laboratory and field testing is done. Analysis and computer simulation are the first steps indeveloping new systems. Needed are accurate models for: the transmission systems (the new proposedsystem as well as the other possible conflicting systems), the primary transmission constants of thecables, and the crosstalk coupling for the frequency spectrum, representative set(s) of test looptopologies, a reasonable set of interferer systems (types and numbers and combinations), broadbandbackground (thermal) noise models, impulse noise models, etc.

B.1.1.1 Background noise

According to a recent Bellcore study, the residence background noise level in the band of interest could beat a level of around -140 dBm/Hz. This background noise level is higher than that achievable by a receiverfront end electronic circuit. On the other hand, attention still has to be paid to make the receiver front endelectronic circuit noise level below the assumed -140 dBm/Hz level.

B.1.1.2 Impulse noise

Impulse noise is of major concern for higher speed twisted wire pair type systems, especially due to thehigher subscriber loop loss. Compared with the very weak received signal, a majority of impulses collectedby the same Bellcore study would cause receiver detection error. It has been shown that forward errorcorrection coding is effective at minimizing the impact of impulse noise. The effect of impulse noise needsto be included in transmission performance simulation. Forward error correction codes are typically usedto handle impulse noise. Section tbd and Appendix tbd describe the results of field measurements ofimpulse noise.

B.1.1.3 Radio frequency interference (RFI)

In addition to coupling within a cable, radio frequency interference (RFI) also becomes a concern as thesignal frequency increases, the wavelength shortens and approaches the dimensions of the cablestructure components, and overlaps radio services. Radio frequency energy may radiate from a wire pairand interfere with radio services (egress). Radio frequency energy may enter a wire pair and interfere withthe wire pair transmission system (ingress).

Modern wire pair systems operate with signals in a "metallic" mode where currents in the two conductorsare equal and opposite in direction thus tending to reduce radiation either entering or exiting. Currents thattravel on both conductors in the same direction are said to be in a "longitudinal" mode. These longitudinalcurrents are much more likely to radiate. External radio frequency fields tend to couple to the pair in thelongitudinal mode. The balance of the individual wire pair conductors and the connecting circuitry relativeto the environment determines the conversion of the normal metallic signal conduction to longitudinalcurrents and the conversion of longitudinal currents to metallic signals.

B.1.1.4 Structural cable faults

Structural cable faults (degraded splices, shorts, opens, grounds, crossed pairs, conductor pairreversals, … do occur and will prevent a transmission system from working. Such mechanical cable faultsare beyond the scope of this effort.

B.1.1.5 The loop environment

Early digital twisted pair transmission systems needed to have cables with very simple make-ups. T1carrier system was originally intended for the interoffice cable plant to replace the loaded cable voice-frequency pairs. Interoffice voice-frequency cable used only one gauge of wire. No bridged taps wereallowed. The T1 repeater spacing matched that of the loading coils starting at 3000 feet and at intervals of6000 feet afterwards to take advantage of the loading coil mounting locations between central offices.

When T1 carrier began to be deployed in the subscriber loop plant for connections to Digital Loop Carriersystems and for high-capacity digital services to business customers, it encountered a much tougherenvironment in terms of cable makeup. Repeater spacing had to be reduced to 4000 or 3000 feet. Bridgedtaps had to be removed.


52

Modern digital twisted pair transmission systems are intended to deliver digital information to averagehouseholds and businesses through the copper loop plant as it exists without modifying the makeup.Depending on the desired information rate and noise environment, the serving distance from a centraloffice (CO) or a remote terminal (RT) could be different. To have a low overall cost, the deploymentprocedure for a new digital transmission system should be as simple as possible. In other words, it wouldbe ideal if the system terminals could simply be installed on the selected loop and turned on. Additionalengineering work such as field trips and loop qualification should be avoided. From the telephonecompany point of view, a service using the transmission system as a delivery vehicle should be pre-qualified for a known type of loop plant, such as resistance design range, CSA, etc. Any loop qualificationshould be on a bulk area basis, not for each individual loop.

B.1.1.6 Telephone cable and subscriber loop structures

This section describes the nature of the structure of the outside loop plant in the US.

A subscriber loop consists of sections (typically 500 feet long) of copper twisted5) pairs of different gauges.A section of a subscriber loop could be aerial (hung on poles), buried (directly in the ground), orunderground (pulled inside protective conduit). Electrical joints, called splices, for cable sections could bemade on a telephone pole for aerial cables or in a manhole for underground cables. These splices are notsoldered as in most electronic circuits, but are made with some form of compression technique. For manyyears the most common splice was made by stripping the insulation from the wire ends, hand twisting thebare wire ends together and covering the splice with tape. Modern splices use connectors which use ahand compression tool to generate the force to penetrate the insulation and make a solid connection.Properly performed, the compression splice results in a metal to metal connection that is impervious toliquid or gas.

Twisted pair cables have large cross sections near the central office. There could be 10, 25 or up to 50pairs in a cable binder group and up to 50 binder groups per cable. Binder groups are combined to formcables of from 50 pairs to several thousand pairs. Cables share a common electrical and physicalstructure, with metallic electrical sheathing and plastic covering. Cables intended for application near thecustomer premises may have fewer pairs.

Functionally, a subscriber loop can be divided into portions that belong to feeder cable, distribution cable,and drop wire6). Wiring inside the customer premises that connects to the drop wire at the networkcustomer interface does not count as part of the network loop7). The interface between the network loopand the customer premises wiring is usually made as close as practical to the point of entry to thepremises.

For large multi-tenant buildings and campuses, the network may provide cabling past the minimum pointof entry if permitted by state regulations. Feeder cables provide links from a central office to aconcentrated customer area. Distribution cables then carry on from feeder cables to potential customersites. Since the loop plant construction is completed before customer service requests, distribution cablesare usually made available to all existing and potential customer sites. Hence, it is a common practice toconnect a twisted pair from a feeder cable with more than one distribution cable to maximize theprobability of reaching a potential customer. These multiple connections from a feeder or a distributioncable to more than one customer location are called "bridged taps." At any one time only customer isconnected and the other taps are left open8). As customers connect and disconnect service, these bridgedtap appearances allow the operating company flexibility in the use of the wire.

–––––––5) As noted elsewhere, an exception to twisted pair cable is single-pair aerial drop wire.6) The term "drop" refers to the drop downward from a pole to a house. Today, most "drop" wires in new constructionare buried.7) Of course, any significant length of customer premises wiring were included before a transmission system terminalwould obviously contribute to transmission effects. Customer premises wiring can vary from a few feet to thousandsof feet in length. The analyses here assumes that the terminal is at the interface.8) Of course, old fashioned party lines had all the customers on the line tied to the same loop back to the wire center.


53

At voice frequency the transmission effects of bridged taps are relatively small and can be controlledwithin acceptable limits by design. The loop plant design rules, such as Resistance Design and CSA, limitthe total bridged tap length to minimize adverse effects, mainly loss and spectrum distortion, on POTStransmission. From a transmission line point of view, these bridged taps are open ended shunts. Abovethe voice band the transmission effects become more significant as the frequency increases and thesignal wavelengths approach the tap lengths.

Connection points between feeder cables and distribution cables are commonly located in cabinets, calledFeeder Distribution Interfaces (FDI). Connection points in distribution cables are commonly in pedestalsfor underground cables or terminals for aerial cables.

Single aerial drop wires often consist of parallel copperclad steel wires, sometimes called "flat pairs." Fornew construction in recent years, multiple (2, 4 or 6) twisted copper pairs are being used, and are buried ifpossible. The drop wire is usually short and has a proportionately small effect on the loop transmissioncharacteristics except for potential radiation effects. A typical rule of thumb was to allow the drop wiring tobe less than 700 feet or 25 ohms in resistance.

The loop and drop wire potentially could pick up other high frequency radiation noises. It could also couldradiate signals to other high frequency electronic devices.

B.1.2 Loop plant design rules: resistance design

Most of the embedded outside loop plant in the US has been constructed using the guidelines calledResistance Design or one of its variations.

POTS loop plant design must accomplish three goals: ensure that there is sufficient direct current flowfrom the network battery plant to operate station sets, allow dc/low-frequency call process signaling(dialing, ringing), and limit transmission loss and frequency roll-off to acceptable levels. As mentioned,telephone cables are designed with different gauges of wire from 26 AWG (thin, with higher resistance) to19 AWG (thicker with lower resistance). These different gauges are designed to have close to the samecapacitance between conductors per unit length (nominal 0.083 (µf/mile). It happens that limiting themaximum dc resistance also controls the maximum voice frequency loss and roll off with frequency.

For modern switching systems a maximum loop resistance (DC resistance) of 1500 ohms9) meetspowering, signaling and transmission objectives. The maximum transmission loss at 1004 Hz is about 9dB with a rolloff of 6 dB at 2804 Hz. From survey data, the average loop has a dc resistance of 600 ohmswith 4 dB of loss at 1 kHz.

Since distances from a central office to each customer are different, distribution cables of different gaugesare utilized to keep the amount of copper (and dollars) used to a minimum while meeting designguidelines. To reduce overall loop resistance the end sections of a long subscriber loop tend to havecoarser twisted pairs. Whereas finer gauge twisted pairs are used closer to the central office in order toreduce the diameter of cables in crowded ducts and minimize cost.

However, some customers are so far away from the central office that a direct implementation of twistedpair cables would result in a dc resistance much higher than the specified 1500 ohms and hence a poorvoice channel service quality. A procedure of installing loading coils and coarser gauge cables has beenused to extend the central office serving distances for the voice channel. Inductive loading results in a loopwith reduced loss within the voice band for a given gauge of cable and acts as a low pass filter above

–––––––9) Different types of switching equipment have different dc loop resistance limits, depending on the battery feedvoltage, the feed resistance, the typical set resistance and the desired minimum current. Step-by-step (SXS)switches with nominal 48 volt dc batteries (and a 41.5 volt emergency minimum) typically have a 1300 ohm designlimit to achieve a minimum of 23 mAdc through a rotary dial 500-type station set with about 150 ohm dc resistance.Thus, references are common to "1300 ohm Resistance Design" even though SXS switches have been retired in allmajor operating companies. Newer electronic switches typically have a 1500 ohm loop design limit from a nominal52 volt battery, for a minimum of 20 mAdc through 400 ohm-dc Touchtone station sets.


54

3000 Hz.10).The original rule for loading cable was 18 kft working length excluding any bridged taps.11)

Under the current rules, loading coils are installed for cables with a total length exceeding 15 kft includingbridged taps. For the most common loading plan, called "H88" with 88mH inductors, the first loading coil isinstalled at 3 kft from the central office. Loading coils are installed every 6 kft thereafter. There may be nobridged taps between loading coils. Bridged taps on the end sections at the central office and customerends may be left connected, up to a total tap length of 6 kft.

B.1.3 Loop plant design rules: carrier serving area (CSA)

DLC systems were originally developed to serve POTS customers beyond the Resistance Design range.Early DLC systems are based on copper twisted pairs using the 1.5 Mb/s T-carrier, T1-Line technology.Twenty-four voice channels are carried on one T1-line by use of time division multiplex. With the use ofoutside plant digital repeaters/regenerators it is possible to reach out 100 miles. Fiber based DLC systemsare now more popular. Depending on the cost of DLC electronics, it becomes more economical to servecustomers with DLC systems beyond a certain distance. This "prove in" distance has been decreasing asDLC electronics costs have come down.

The concept of Carrier Serving Area (CSA) engineering guidelines was originally developed in the early1980's to support 56 kb/s Digital Data Service (DDS) delivery to customers served by DLC systems. Theconcept was then revised very slightly and has been used as the guide for voice grade special servicesand POTS deployment from the DLC remote terminal. A CSA is roughly defined as a serving distance of 9kft for 26 gauge loops and 12 kft for 24 gauge loops from a DLC remote terminal the term is also appliedto loops that originate from a central office as well if they meet CSA guidelines. Short loops around acentral office may be consistent with CSA rules even though constructed using Resistance Design rules. Arecent (1991) survey shows that over 60% of DLC loops meet the CSA guidelines. (References in thisdocument to "CSA" loops or "CSA-type" loops mean wire pairs that meet CSA design guidelines whetherthey originate from a central office or from a network remote terminal site.)

As the operating company have deployed DLC systems CSA rules have proven a useful rule of thumb forHDSL system deployment. They were also chosen as the loop reach target for 6Mb/s ADSL-3 systems.Carrier serving area wire pairs from the remote terminal of a DLC system to the network interface on thecustomer's premises are expected to meet the following design guidelines.

a) Non-loaded cable only

b) Multi-gauge cable is restricted to two gauges (excluding short cable sections used for stubbing orfusing).

c) Total bridged tap length may not exceed 2.5 kft. No single bridged tap may exceed 2.0 kft.

d) The amount of 26 gauge cable (used alone or in combination with another gauge cable) may notexceed a total length of 9 kft including bridged tap.

e) For single gauge or multi-gauge cables containing only 19, 22 or 24 gauge cable, the total cablelength including bridged tap may not exceed 12 kft.

f) The total cable length including bridged tap of a multi-gauge cable that contains 26 gauge cablemay not exceed

kft 9

)26(312

LBTAP

L

−−

–––––––10) Transmission analysis shows that loss is minimum for certain ratios of resistance, conductance, capacitance andinductance. Normal cable has a small inductance relative to resistance and capacitance. Lumped inductive loadingachieves close to the ratios for minimum cable loss within the voice band.)11) The length of the cable that connects directly from the network to the customer, excluding any bridged taps, iscalled the "working length." The working length of the cable corresponds to the dc resistance path from the networkbattery to the customer interface. Bridged taps are open-circuited to dc flow.)


55

where L26 is the total length of 26 gauge cable in the cable (excluding any 26 gauge bridged tap)and LBTAP is the total length of bridged tap in the cable. All lengths are in kilofeet (kft).

The limits defined above are the maximum permissible outer bounds for a CSA. Nothing in the CSAconcept prohibits the restriction of CSA cables to shorter lengths.

CSA guidelines do not include central office wiring on the switch side of the protector frame, drop wire orcustomer building wiring. Detailed statistics for central office wiring or customer premises are notavailable. Central office cabling is typically 24 or 26 gauge and may be up to 1 kft long. Customerpremises wiring is typically 26 gauge and may be up to 1kft or more. In some installations, some drop andwiring inside the customer premises may be part of the network. Although not a transmission requirement,it is suggested that no more than two gauges of cable be used. Note: all wire gauge references in thisdocument are American Wire Gauge (AWG).

B.1.4 Distribution area (DA)

A CSA is often further divided into 1 to 6 Distribution Areas (DA). A DA is characterized by a single FeederDistribution Interface (FDI) where cross-connects are located. A DA typically serves about 500 customers.The cable pair group from a RT to all DAs could have different service capacity than that of all DAscombined. Distribution cables emanating from an FDI usually have a 1.5 to 2 pairs for all potentialcustomer living units. On the other hand, cable pairs from RT to FDI are installed based on the number ofreal customer lines with a smaller spare ratio. This strategy is aimed at an overall minimized installationcost. The average serving distance of each DA is usually significantly shorter than that of a CSA.12) Arecent (1991) survey shows that most DA distribution loops are less than 6 to 8 kft in length (26 and 24gauge respectively) or about 2/3 of the maximum CSA lengths.

B.1.5 Loop statistics

The Resistance Design and Carrier Serving Area design do not define how much of each type of cablingis actually used. Major surveys of loop topology in the old Bell System were conducted in 1976 and 1983.

B.2 AWG and metric cable: diameters and DC resistance and capacitance

Test loop sets have been developed for AWG and metric cables by T1E1.4 and ETSI for ISDN DSLs,HDSL, ADSL and VDSL. It is sometimes useful for interested parties who are familiar with one set ofcables, but not the other, to make a rough judgment on which cable in one set compares to which cable inthe other set, if any. One can get into the right ballpark or at least out of the wrong one, by comparingconductor diameters and diameters, DC (0 Hz) resistance and DC capacitance and insulation materials.Table B.1 summarizes this data for the most common types of metric and AWG telephony cables. Non-telephony 18 and 20 AWG gauges are also included for comparison because their conductor diametersare close to 0.8mm and 1.00 mm metric cables.

Attenuation versus frequency data (say at 1 kHz, 10 kHz, 100 kHz, 1 MHz, 10 MHz and 30 MHz) wouldallow further contrasts and comparisons. Polyethylene is the most common insulation for feeder anddistribution cables. Polyethylene is a very good dielectric whose properties change very little withfrequency. PVC is the most common insulation for single-pair, overhead/aerial drop wires exposed to theexternal environment. PVC dielectric properties vary much more with frequency than those ofpolyethylene.

Only AWG 26 PIC and metric 0.40 PE are really close in transmission characteristics.

B3. Cable primary constants (RLGC) characterization

It is not feasible to perform laboratory or field tests to represent all likely environments that a transmissionsystem will encounter. Computer simulation provides a means to test schemes against anything that canbe quantified numerically. Fundamental to the simulation of wire systems are accurate models of thetransmission characteristics of the wire itself versus frequency and temperature.

–––––––12) However, the maximum serving distance of a DA might still be very close to that of a CSA.


56

The primary constants of resistance (R, ohms/km), inductance (L µH/km), capacitance (C, nF/km), andconductance (G, mho/km) are used to model most transmission lines. Secondary parameters such asimpedance, attenuation and phase or the chain parameters ABCD may be calculated from the primaryconstants. These "constants" actually vary in value with frequency, temperature and humidity. To a firstorder, signal attenuation increases as the square root of the frequency. Variation of the "constants" andinductive reactance becoming larger with frequency relative to resistance and capacitance result in theactual attenuation versus frequency curve being more complex.

Chain parameters ABCD allow cascading of models of two port electronic devices such as wire pairs.Complex loop topologies with changes of gauge and bridged taps can be constructed with ABCDmatrices. See Figure B.1.

The existing primary constant RLCG models of the common AWG PIC cables were based on carefulmeasurements and curve-fitting in the early 1970s. They were believed to be valid to 10 MHz and torepresent nominal values for expected manufacturing variations. VDSL and newer proposed schemesmay well have spectral components to 30 MHz. It is vital to have models that reflect the transmissionbehavior of the cables in the real world to the frequency and temperature ranges needed.

The primary constant data can be presented in either as R, G, C and G values versus frequency or asparameters to equations that have been curve fitted to measured data. (See T1E1.4/96-015.)

B.3.1 Transmission-Line Characterization

This section directly addresses the transmission characteristics of twisted-pair phone lines.

Most twisted-pair phone lines can be well-modeled for transmission at frequencies up to at least f<30 MHzby using what is known as two-port modeling or “ABCD” theory. Such ABCD theory is well covered inbasic electromagnetic texts, but is often not in a form convenient for use in DSLs. Werner presentedessential results of such translation to DSLs in a 1991 JSAC paper “The HDSL Environment” (August1991) and this section essentially repeats that effect, but provides more detail along with updates basedon various studies in standards bodies that have led to DSL characterization to at least 30 MHz.

Section A.1.1 first describes ABCD modeling in general before Section A.1.2 specializes to the case oftwisted-pair transmission lines. Sections A.1.3 considers the special case of bridge taps before SectionA.1.4 shows how to compute the transfer characteristics of a subscriber loop consisting of many sections.Section A.1.5 shows how to measure RLCG parameters for loop characterization as well as lists modelsfor several popular twisted-pair types.

B.3.1.1 “ABCD” modeling

Figure B.2 shows a general two-port linear circuit. There is a voltage at each port and a current on theupper path on each port. The voltages and currents will depend on the source (port 1) and load (port 2)impedances and voltage source(s), but nevertheless always relate to each other by the matrix relationship:

221

221

2

2

2

2

1

1 or DICVI

BIAVV

I

V

I

V

DC

BA

I

V

+=+=

⋅Φ=

⋅

=

where Φ is a 22× matrix (nonsingular in all but trivial situations not of interest) of 4 possibly frequency-dependent parameters, A, B, C, and D, which all depend only on the network and not on externalconnections. The quantities have circuit definitions as in the table below:

A open-load voltage ratio

B shorted-load impedance

C open-load admittance

D shorted-load current ratio


57

The transformation is reversed by 1−Φ so that

⋅Φ=

⋅

−

−−

=

−

1

11

1

1

2

2 1

I

V

I

V

AC

BD

BCADI

V.

When I=Φ , that is an identity, the network is a trivial connection of the upper path and lower path acrossthe network, essentially meaning there is no network. A relationship of interest is the ratio

( )1

2

V

VfT =

where the frequency dependence is shown explicitly for T(f), but not for the other voltages to simplify

notation. This ratio depends on the load impedance attached at port 2, or the ratio 2

22 I

VZZL ==

( )BZA

Z

ZBA

fTL

L

L+⋅

=+

= 1

can be related to a transfer function )(fH between an input voltage supply SV (with finite internal

impedance SZ ) to the output voltage 2VVL = (across a load 2ZZL = ).

( )( )

( )( )

( )( ) )()(

1

12

2fT

ZZ

Z

fV

fV

fV

fVfH

fV

fV

SS

L

S

L ⋅+

=⋅== ,

where 1

11 I

VZ = is the input impedance of the terminated two-port. 1Z must be computed as in the

second equation below and is the ratio of input voltage to current when load LZ is attached at the output.

A cascade of two-ports has a two-port matrix that is the product, in order, of the two ports

⋅Φ=

⋅Φ⋅Φ⋅Φ=

−

N

N

N

NN I

V

I

V

I

V121

1

1 ... ,

allowing for the calculation of transfer functions, and insertion losses of more complicated networks aslong as a two-port model can be found for each subsection in the cascade. The inverse is found byreversing the order and taking the product of the inverse matrices. The input impedance of the two-port is

DCZ

BAZ

Z

DC

Z

BA

I

VZ

L

L

L

L

++=

+

+==

1

11 .

Two-port networks are very useful in the analysis of twisted-pair transmission lines as in the next severalsections. In these sections, the transmission line is modeled as a cascade of two ports that arecharacterized by resistance, inductance, capacitance, and conductance per unit length, and by the lengthof the transmission-line segment.

B.3.1.2 Transmission-line RLCG characterization

The two-port characterization of a transmission line derives from the per-unit length two-port model inFigure B.3. The R, L, C, and G parameters represent resistance, inductance, capacitance, andconductance per unit length of the transmission line.

A segment of transmission line can be viewed as a cascade of such sections that are infinitesimally smallin length. At any point x, the two-port voltages and currents relate through the differential equations


58

( )

( ) VCjGdx

dI

ILjRdx

dV

⋅ω+=−

⋅ω+=−

at any given frequency fπ=ω 2 . V and I are phasor quantities representing peak amplitudes of

sinusoids at frequency f (or amplitudes of the complex exponential ftje π2 ). The R, L, C, and Gparameters themselves can vary with frequency, but are presumed constant with respect to length at anygiven frequency in the analysis to follow. This set of differential equations is equivalent to the pair ofsecond-order differential equations

Idx

Id

Vdx

Vd

⋅γ=

⋅γ=

22

2

22

2

,

where

( ) ( ) YZCjGLjRj ⋅=ω+⋅ω+=β+α=γ

is the frequency-dependent propagation constant for the twisted pair, and characterizes the segment oftransmission line. The impedance per unit length, Z, and the admittance per unit length, Y, are alsodefined in Figure B.3. The attenuation constant is and the phase constant is . The attenuation constantis very important for twisted-pair. As can be inferred from equations to come, the attenuation of a twisted-pair is approximated by 8.668 α dB per unit length at the frequency of interest. The phase constant isrelated to speed of propagation on the twisted pair: At each frequency fπ=ω 2 , a sinusoid propagates onthe twisted pair with phase given by

( ) xtx β−ω=ωθ ,

and has envelope amplitude attenuated as xe α− . The wavelength is the length (at fixed frequency andtime) over which the sinusoid undergoes a full cycle and is thus given by

βπ=λ 2

.

Remembering that β is tacitly a function of frequency, different frequencies thus have different

wavelengths. The sinusoidal wave at frequency appears to propagate along the twisted pair at phasevelocity

βω=pv ,

and the phase delay per unit length at this same frequency is ωβ==τ

pvp1 . When β is a linear function

of frequency, the channel is said to have linear phase and the phase velocity and delay are constant over

all frequencies. An example is the case where R=G=0, and then LCω=β - and (when L and C are

constant with respect to frequency) means that all frequencies move at the same phase velocity

LCv p

1= . Such a transmission line is said to be dispersionless. Note that it is possible to subtract out

the linear (proportional) part of β without introducing error to the time domain response of a cable pair onlywhen there is no reflected wave being propagated along the pair. Such a condition (no reflection) occursonly when the impedance of the load is matched to the characteristic impedance of the cable pair. Thismay be particularly important when modeling bridged taps. In practical DSLs, dispersionless transmission


59

never occurs and different frequencies travel with different velocities, leading to dispersion of signalenergy (and to the intersymbol interference). For a dispersive transmission line, it is of interest toinvestigate the speed at which a group of frequencies centered around propagates. To understand thisconcept of “group” or “envelope” velocity, suppose one investigates the differing speeds of the twofrequencies ω∆±ω where the offset or difference is small and the corresponding values of β∆±β , but

both have the same amplitude. The resultant sum waveform is

( ) ( )[ ] ( ) ( )[ ] [ ] [ ]xtxtwAxtwAxtwA β−ω⋅⋅β∆−⋅∆=β∆−β−∆−ω+β∆+β−∆+ω coscos2coscos ,

the right side this equation is an “envelope-modulated” sinusoid, a product of two sinusoids. When thephase velocity is constant and there is no dispersion, the phase velocity of the first term on the is thesame as that of the second term, and the phase velocity equals the group velocity. However, when phasevelocity is not constant, the first term moves at a different (often much slower) speed given by β∆ω∆ / .

This slower speed is the group velocity and in general computed by the inverse of the group delay

ωβ=τ

d

dg or .

Group delay in essence measures the spread in delay between the fastest and slowest movingfrequencies in the immediate vicinity of ω. The greater the group delay, the greater the dispersion in thetransmission line.

The solution to the set of differential equations is easily modeled as the sum of two opposite-directionvoltage/current waves:

( )( ) xx

xx

eIeIxI

eVeVxV

γ−γ−+

γ−γ−+

⋅+⋅=

⋅+⋅=

00

00 .

By insertion of either of these solutions into the appropriate first-order voltage/current differentialequations, the ratio of the positive-going voltage to the positive-going current, as well as the (negative ofthe) ratio of the negative-going voltage to the negative-going current is equal to a constant characteristicimpedance of the transmission line

Y

Z

CjG

LjR

I

V

I

VZ =

ω+ω+=−==

−

−

+

+

0

0

0

00 .

One easily verifies that the R, L, C, and G parameters are equal to

γℜ=

γℑ

ω=

⋅γℑω

=

⋅γℜ=

0

0

0

0

1

1

ZG

ZC

ZL

ZR

.

For twisted-pair transmission and DSLs, it is rare that any of these 4 parameters are zero and sosimplifications in textbooks or other developments that lead to so-called “lossless transmission lines” or“dispersion-less” transmission are not of interest for DSLs. Furthermore, these parameters are frequency-dependent for transmission lines and are best determined by measurement as in Subsection 4.3.7.

A segment of transmission line of length d has solution dL VV = and dL II = and thus


60

( )( ) dd

L

ddL

eIeIdII

eVeVdVV

γ−γ−+

γ−γ−+

⋅+⋅==

⋅+⋅==

00

00 .

Since the two voltage waves in each direction are related to the same-direction current waves by the

common ratio 0Z , one can solve the above two equations for +0V and −

0V to get:

( )( ) d

LL

dLL

eZIVV

eZIVV

γ−−

γ+

⋅⋅−=

⋅⋅+=

021

0

021

0.

By substituting these constants back into the solution in general and evaluating for the voltage andcurrents at x=0 in terms of those at x=d , one obtains the following two-port representation

( )( )

( ) ( )( ) ( )

( )( )

⋅

⋅

⋅=

dI

dVdd

Z

dZd

I

Vγγ

γγ

coshsinh1

sinhcosh

0

0

0

0.

The ABCD entries can be read from the matrix, or equivalently, can be computed from the R, L, C, Gvalues through in relations for γ and for 0Z . Then, for a given length of transmission line d, the engineer

may model that transmission line as a single “lumped” two-port, replacing the distributed model in FigureB.3.

Knowing the load impedance so that LZdIdV =)(/)( , the insertion loss then becomes

( ) ( )( )

( )d

d

dZ

Zd

T

LZZ

L

γ+

γ=γ⋅

+γ

=tanh1

sech

sinhcosh

100

.

The input impedance of the two-port is V(0)/I(0) or

( )( )dZZ

dZZZZ

L

L

γ⋅+γ⋅+

⋅=tanh

tanh

0

001 .

The input impedance of a very long line reduces to 01 ZZ = , since ( ) 1tanh →γd for large d.

The transfer function in any case becomes

( )( ) ( )

γ⋅+⋅+

γ+⋅

γ⋅=

+=

dZdZ

dZT

ZZ

ZH

LL ZZ

ZZ

SS tanh1tanh

sech

000

0

1

1 .

Thus, this type of model applies to the upper example in Figure B.5. Note also there the two-port modelsthat characterize the source and load. Thus, general principle of multiplying matrices when cascadingtwo-ports can be directly applied. If several transmission line segments with different R, L, C, and G werecascaded, then each would have its own two-port model. This situation corresponds to connection oftwisted pairs (splicing) with different gauges.

B.3.1.3 Power for transmission lines

A sinusoid at any frequency on a transmission line represented by the phasor voltage V and phasorcurrent I has average (rms) power

( ) [ ]*21 VIfP ℜ= .

Figure B.4 shows a simple phase circuit having input current I and voltage V across a load withimpedance LLL jXRZ += . From basic circuit theory, a sinusoidal current with peak amplitude I delivers

power


61

( ) [ ]*21

2

212

21 VIR

Z

VRIfP L

LL ℜ=== ,

thus providing interpretation for the relation in the previous equation.

Maximum power is transferred from the power supply to the load when the source impedance is the

conjugate of the load impedance in Figure B.4, LLLoptS jXRZZ −== *, . This corresponds to one-half the

total power of the source being dissipated in the load. An example of the use of this maximum-power-transfer result is when one investigates the termination of a twisted-pair transmission line. To transfermaximum power from the line to the load, the load impedance should be designed to be the conjugate ofthe line impedance viewed going back into the line. When the line is long, this impedance will be thecharacteristic impedance of the line itself, meaning the best loading is

*0, ZZ optL ≅ ,

meaning half the power in the line is transferred to the load (with the other half dissipated within the lineitself). Similarly, the optimum driving impedance is the conjugate of the line impedance, which again forlong lines is the characteristic impedance, so

*0,, ZZZ optLoptS ≅= .

Again, half the source power will be delivered to the line. For a lossless transmission line, the half of thesource power delivered to the line is the same half of power delivered to the load. At higher frequencies,all transmission lines become lossless and so the best load and source impedances become resistive andequal to the (real) characteristic impedance of the line.

The condition for maximum power transfer is not the same condition for elimination of reflections (seenext subsection) unless the line is lossless.

B.3.1.4 Reflection coefficients

When the load impedance is equal to the characteristic impedance (and not the conjugate of the

characteristic impedance), the negative-going wave constant 00 =−V in the above equations. There is

then no reflected wave and all the above relationships simplify somewhat. In practice, such matching isnot likely to occur, and the solution for the differential equation at x=d has general ratio of positive-goingwave to negative-going wave as

0

0

0

0

ZZ

ZZ

eV

eV

L

Ld

d

+−

=⋅

⋅=ρ

γ+

γ−−.

This reflection coefficient is clearly zero when the transmission line is “matched” or terminated in its ownimpedance, 0ZZL = . The return loss is defined as the inverse of the reflection coefficient for any

interface to a two-port, and usually expressed as a positive quantity in decibels. This situation prevents“bouncing” of signals on a transmission line and thus reduces the dispersion (relative delay) of signals onthe line. In this case of 0ZZL = , the input impedance is then also 01 ZZ = . When the transmission line

impedance is approximately real, then the situation of no bouncing corresponds also to maximum energytransfer in Subsection 4.3.2.1 from the line into the matched load. However, when (as usual for twistedpairs), the line characteristic impedance is complex, then maximum energy transfer occurs when the loadis the conjugate of the characteristic impedance, and thus elimination of bouncing does not guaranteemaximum energy transfer for lossy lines. On many lines as the frequency increases, the R and G termsbecome negligible and so for these frequencies, maximum energy transfer and elimination of bouncing

occur when the load impedance is matched to CLZZL ≈= 0 .


62

A similar source reflection coefficient can be written as

0

0

ZZ

ZZ

S

SS +

−=ρ .

This source reflection coefficient measures the reflected positive-going wave amplitude with respect to anegative-going wave that flows into the source impedance. The return loss at the interface to between thesource and line is therefore the inverse (in dB) of the source reflection coefficient. Note that the source

impedance that leads to maximum power transfer into the line *1ZZS = again is not necessarily the same

as that leading to no reflection at the source end. A wave launched from a source will traverse the loopwith phase and group velocities, will be reflected at one end, reflected again at the source end, and so on.This series of reflections leads to a transient on the loop, unless the loop is terminated in a loadimpedance equal to the characteristic impedance of the line. Again when the line can be approximatedover the used frequency range as lossless, and thus having real characteristic impedance, then themaximum energy transfer and reduction of bouncing objectives coincide.

Formally the return loss of a transmission line is the inverse ratio of reflected power to incident power onthe load (or next section of circuitry). This return loss is simply the square of the reflection coefficient, thus

dB. 1

log10 loss return2

10 ρ=

B.3.1.5 Characterization of a bridge-tap section – a three-port

For modeling of loops, a bridge-tap is a three-port section, but one of the ports appears as a loadimpedance to the line, between the two sections on each side of the bridge tap. Such a situation can bemodeled by the two-port with ABCD matrix shown in the last example of Figure B.5.

The impedance of the tap section tZ is computed according to the formula above for the input impedance

of a section of transmission line terminated with an open circuit ( ∞=LZ ), which simplifies to

( )( )d

dZZ tt γ

γsinh

cosh0 ⋅= .

If the tap were not terminated in an open circuit, then the general formula for the input impedance Z1

(above)of the section should be used.

Circuits with bridge-taps on bridge-taps have an impedance that is calculated by working backwards fromall open taps to points of the taps, modeled as the two-tap section’s impedances in parallel. The resultantimpedance then becomes a termination (load) impedance for the next section working backwards towardsthe main transmission pair of interest. While perhaps tedious, the calculation process is straightforwardand recursive.

B.3.1.6 Computation of transfer function

The computation of the transfer functions for twisted-pair transmission lines with multiple sections thensimply becomes a process of multiplying in cascade the corresponding two-port ABCD matrices for eachsection. Some examples are provided in Figure B.5, with the corresponding two-port matrices below eachexample. The matrices are multiplied left to right in the natural order of appearance in the figure. That isthe overall two port is just

NΦ⋅⋅Φ⋅Φ=Φ !10

where the source voltage divider is modeled by the two-port

=Φ

10

10

SZ.


63

The final output voltage and current are related by the usual LLL ZIV ⋅= , which allows the transfer

function to be computed from the ratio

S

L

V

VH = .

In the upper example of Figure B.5, a simple section of twisted pair with characteristic impedance 0Z and

propagation constant γ is modeled by the cascade of a two-port matrix description 1Φ for a length d and

the source two-port matrix 0Φ . This upper example is straightforward application of the two-port theory.

The lower example additionally has a bridge-tap section with 02Z and 2γ of length 2d and a second

section of the transmission line with yet a third characteristic impedance and propagation constant. Thetwo sections of transmission line are modeled as usual, where the impedance and propagation constantcan be computed for each frequency from the known R, L, C, G parameters for each section. The bridge-tap section is modeled as a parallel (shunt) impedance that is computed according to the formula for anopen-ended transmission line of length 2d (if the tap were terminated, the impedance shown need only be

replaced by the more general expression for the inverse of the input impedance of that section). Theoverall two-port matrix is simply the product of the 4 two-port matrices shown.

A variety of simplifications are sometimes studied assuming each section is very long and so appears tobe terminated in its own characteristic impedance, leading to expressions for the transfer function andinput impedance in various situations. While sometimes useful for interpretation, with modern day signalprocessing analysis tools (for instance, matlab, etc.), it is often easier to compute the transfer functionwithout simplifying assumptions and then analyze the corresponding results.

B.3.1.7 Relationship of transfer function and “insertion loss”

Transmission engineers sometimes also directly measure the transfer characteristics of a transmissionline at several frequencies. It is hard to measure the transfer function directly because of loading effects,but it is possible to measure easily the insertion loss, from which the transfer function can be computed ifload and source impedances for the measurement are known.

The insertion loss is computed using a configuration in Figure B.4 by first measuring the voltage noV , and

then inserting the transmission line at the point where noV was measured initially and again measuring

LV , the voltage across the load with the line inserted. Thus the insertion loss is

( ) ( )fV

fVfT

no

LIL

)(= .

The desired transfer function is instead SL VVH /= so

( ) ( )fTZZ

Z

V

V

V

VfH IL

LS

L

no

L

S

no ⋅+

=⋅= .

Note that when LZZ =1 , meaning the line is terminated in its own impedance as often in practice, then the

equation can be rewritten in terms of the T(f) as

( ) ( )fTZZ

Z

V

V

V

VfH

S

L

S⋅

+=⋅=

1

1

1

1 ,

which also then shows that in the matched-termination case, ( ) ( )fTfT IL= . In most cases of interest in

DSL, the line is long and so the source impedance is matched to the characteristic impedance (whichequals the input impedance of the line when the line is long) and all impedances are real over the higherfrequencies used for DSL transmission. In this case, the transfer function is simply 6 dB lower than the


64

insertion loss. A program that computes transfer functions of twisted pairs using two-port theory is on thesecond author’s worldwide web page at http://www-isl.stanford.edu/people/cioffi.

A crucial point of note: When the transfer function is computed for a circuit using RLCG parameters, thenthe insertion loss may be computed from the transfer function and is roughly 6 dB higher under theapproximations above. The insertion point is exactly the point at which a transmit power constraintapplies. Thus for instance, input voltage levels computed from a power constraint for a DSL (for instance,in performance calculation or SNR computation) undergo a channel that is the insertion loss, and not thetransfer function. A common mistake is to compute data rates and performance as if the transmit powerwere 6 dB lower by incorrectly using the transfer function instead of the insertion loss.

Transmission lines are characterized in this Appendix by 4 parameters, the Resistance R in Ohms/km, theInductance L in Henrys/km, the Capacitance C in Farads/km, and the Conductance G in Mhos/km.

The RLCG parameters in this appendix were provided by the following measurement and curve-fittingprocedures:

B.3.1.7.1 Measurement Procedure

The open-circuit impedance, OCZ , and short-circuit impedance, SCZ , for a length, l , of twisted-pair

transmission line are measured versus frequency. An l=10 m length is used for measurements below 2MHz and an l=1 m length is used for measurements between 2 MHz and 30 MHz. The characteristicimpedance and propagation constant are computed from the measured impedance according to:

characteristic impedance:

SCOC ZZZ ⋅=0

propagation constant:

=γ −

OC

SC

Z

Z

l1tanh

1

From the characteristic impedance and propagation constant, RLCG can be computed as:

( )0ZR γℜ=

( )01

ZL γℑω

=

ℑ=

0

1

ZC

γω

γℜ=0Z

G .

B.3.1.7.2 Curve-fitting

Because of error in practical measurements of the impedance, the RLCG values may not follow smoothcurves with frequency so parameterized (smooth) models of RLCG are then fit to the measured values.The models are:

4 244 24

111

)(

farfar

fR

SOSCOC ⋅++

⋅+

=

where r0C is the copper DC resistance and r0s is (any) steel DC resistance, while ac and as are constantscharacterizing the rise of resistance with frequency in the “skin effect.”


65

b

m

b

m

ff

ffll

fL

+

+=

∞

1

)(0

where l0 and ∞I are the low-frequency and high-frequency inductance, respectively and b is a parameter

chosen to characterize the transition between low and high frequencies in the measured inductancevalues.

ecfccfC −∞ ⋅+= 0)(

where ∞c is the “contact” capacitance and c0 and ce are constants chosen to fit the measurements.

egfgfG +⋅= 0)(

where g0 and ge are constants chosen to fit the measurements.

Further information on smoothing of test measurements is found in ASTM D 4566.

B.3.2 TP1

TP1 is representative of .4 mm or 26-gauge phone-line twisted pair. The specific cable measured wasprovided by Bell South to BT and measurements were validated by GTE to produce an acceptable fitbetween measured responses and projected insertion loss as computed from the parameters in table B.2using methods in B.3.1.7. The primary constants produced using the parameters are given in Table B.3.Measurements by Bellcore, whose results are listed in tables B.4 and B.5, have indicated that their resultsfor 26-AWG PIC lines have found strong agreement with the values in the model of this document.

B.3.3 TP2

TP2 is representative of .5 mm or 24-gauge phone-line twisted pair. Parameters are found in table B.6computed using methods in B.3.1.7. Primary constants are found in table B.7. Measurements by Bellcore,whose results are listed in table B.8, have indicated that their results for 24-AWG PIC lines are in strongagreement with the values in the model of this document.

B.3.4 22-Gauge Phone-Line Twisted Pair

Measurements by Bellcore for 22-gauge twisted pair are found in table B.9.

B.3.5 TP3

TP3 is representative of DW10 Reinforced cable with .5 mm copper PVC-insulated conductors, PVC-insulated steel strength member, and Polyethylene sheath. Parameters computed using methods inB.3.1.7 are found in table B.10. Primary constants are found in table B.11.

B.3.6 FP

FP is representative of ETSI 1.14 mm flat (no twists) phone-line twisted pair. Parameters computed usingmethods in B.3.1.7 are found in table B.12. Primary constants are found in table B.13.

B.3.7 Category-5 Twisted Pair

Table B.14 gives parameters and table B.15 gives primary constants computed using methods in 3.1.7for cables thatmeet or exceed EIA/TIA Category 5 twisted-pair specifications.

B.3.8 Two-Pair Twisted Drop

Bellcore measured a two-pair twisted service drop cable where the tip and ring are twisted for each pair,and the two pairs are then twisted together. The conductor gauge is 22 AWG, and the tested cable was of


66

length 228.6 m = 750 ft. The values for R, L, C and αααα are averaged over the two pairs since theyexhibited a high degree of symmetry.

The measurements were made with an HP-3577A Network Analyzer connected to an HP-356711A S-Parameter test set. Measurements at equally spaced log frequencies between 772 kHz and 40 MHz wereobtained.

C and αααα were directly measured with the network analyzer. The short and open complex impedances,

scZ and ocZ of the drop cable were also measured with the network analyzer, and the characteristic

impedance, 0Z , calculated in the usual fashion.

It was not feasible to obtain accurate measurements of the conductance G due to a lack of a precisemeasurement of impedance angle. This is a result of measurement equipment limitations and thetransformer baluns used to perform the impedance conversion. Additionally, the drop wire jackets directlycontact the pair insulation, hence altering the effective dielectric constant and tan delta. Moreover, thecapacitance is not flat over the entire frequency range. Fortunately, at high frequencies, G is of littleimportance for transmission.

Using the relationship

C

LZ =0 ,

which holds when G<<ω C and R<<ω L, the inductance values are calculated.

Using the relationship

0/34.4 ZR∗=α

the resistance values over the range 0.772 - 40 MHz are evaluated. Results are given in table B.16.

B.3.9 Two-Pair Quaded Drop

Bellcore measured a two-pair quaded service drop cable where the four conductors comprising the twopairs are twisted together as a unit. The conductor gauge is 22-AWG, and the tested cable was of length228.6 m = 750 ft.

Test equipment, measurement setup and the equations used to perform the calculations are identical tothose used in B.3.8 on Two-Pair twisted drop. Results are found in table B.17.

B.3.10 Flat-Pair Drop

Bellcore measured a flat-pair service drop cable where the tip and ring conductors of a single pair areparallel. The conductor gauge is 18-1/2 AWG, and the tested cable was of length 291 m = 954 ft.

Test equipment, measurement setup and the equations used to perform the calculations are identical tothose used in B.3.8 on Two-Pair twisted drop. Results are found in table B.18.

B.3.11 Additional Models

For additional models for European and other types of cable, see ETSI STC TMC6 Permanent Document# TM6(97)02.

B.4 Cable crosstalk models

Accurate models of crosstalk coupling between pairs in typical cable structures are as vital as the primaryRLGC constant models to system simulation. For the DSL family of transmission systems the limitingfactor on loop range has been crosstalk coupling of signal energy from like or unlike transmission systemson other pairs in the cable and not from the end-to-end attenuation of the signal.

The current crosstalk models were developed in the 1980s based on computer simulations of the physicalstructure of the cables and later compared with measurements. Quantitative crosstalk models for lessthan full binder groups or small cables are not available.


67

B.4.1 Near end crosstalk, NEXT

Telephone twisted pairs are organized in binder groups of 10, 25, or 50 pairs. Many binder groups share acommon physical and electrical shield in a cable. Due to capacitive and inductive coupling,

there is crosstalk between each twisted pair even though pairs are well insulated at DC. The crosstalk invoice frequency band is minimal, i.e. one can hardly hear the voice energy from an adjacent pair becausethe crosstalk loss is usually more than 80 dB, compared with a voice channel loss of less than 20 dB.

In general the effect of cable crosstalk is minimized not only by the use of good insulation materialbetween pairs but also by adapting different twist distances among different pairs in a binder group. Thebinder groups are also twisted such that no two groups are adjacent for long runs. For digitalcommunication via digital subscriber line technology, where the signal bandwidth reaches into the MHzrange, the crosstalk is a limiting factor to the achievable throughput.

Near-end-crosstalk (NEXT) is defined as the crosstalk effect between transmit and receive pairs at thesame end of a cable section. In other words, NEXT is a measure of the crosstalk effect between atransmitter and a receiver at the same end of a twisted pair cable. See Figure B.6. NEXT is usuallyconsidered for full duplex digital subscriber line systems such as DSL and HDSL where the transmit andreceive spectra at each end are the same (or overlap).

NEXT is strongest on the cable at the point where the transmitter of the crosstalking signal puts the signalon the pair. Any receivers near to this transmitter will receive NEXT as well as the intended signal. TheNEXT path attenuates the unintended signal greatly, but the relevant issue is the signal to noise ratiobetween the intended signal and the NEXT. Therefore, NEXT becomes a problem if the intended signal isattenuated enough. Symmetrical systems such as the ISDN DSL have transmitters at both ends of everypair on which it is installed. The worst case NEXT is then usually the NEXT produced by a binder groupfull of similar collocated transceivers. The NEXT received from similar systems, i.e. DSL to DSL, HDSL toHDSL, or T1 to T1, in this way is called “self-NEXT.”

For DSL and HDSL, full duplex communication on a single pair is achieved by the use of the echocancellation technique. This requires transmit and receive signal paths be as fully separated as practicalwith signal processing techniques13) even though the transmit and receive signals share the samefrequency spectra. However, transmit signals in other adjacent pairs are not available to the particularreceiver. Thus, any energy coupled into a pair used by a transmission system can not be effectivelyremoved from the received signal.

For the T1 line system, bipolar (AMI) encoding of the 1.544 Mb/s signal results in a transmitted in atransmitted spectrum centered around 772 kHz. This 772 kHz signal is much higher in frequency thanvoice signals and crosstalk coupling is much higher. For T-carrier or T1 system, the full duplexcommunication is based on two separate twisted pairs. In the interoffice cable plant, special cables areused with a shield between the binder groups. T1 signals going in one direction from all T1 systems areplaced in one binder groups. All signals going the other direction are in the other binder group with theshield between them. This binder group separation of transmit and receive pairs and shielding greatlyreduce, but does not eliminate, the NEXT effects.

In the outside customer loop plant, the special cables are not readily available. Binder group separation ispracticed as much as possible for T1 in the loop plant. Shorter repeater spacing and very careful attentionto placing of repeaters relative to other T1 systems helps compensate somewhat for the much moresevere crosstalk environment in the loop plant.

For ADSL systems using FDM to separate the upstream and downstream transmissions, there is no self-NEXT to limit transmission range as is the case for DSL and HDSL. For EC-based ADSL systems usingecho cancellation with overlapping downstream and upstream spectra, there will be self-NEXT in the

–––––––13) Two-to-four-wire hybrid circuits which act as balanced bridge networks perform the first level of separationbetween transmit and receive signals. Ten to twenty dB of isolation can be achieved with active and passive analogcompromise balance impedance networks. Digital echo cancellers can provide 30 to 40 dB of additional isolation.)


68

overlap region. Analyses indicate that self-NEXT will not be a limiting factor for EC-based ADSL looprange, but compared to non-overlapped systems (e.g., FDM), overlapped systems would causesubstantially more crosstalk into ADSL upstream and downstream transmissions on other pairs..

There could be 1225 different NEXT values at a particular frequency for a 50 pair binder group, assumingpair-to-pair NEXT is symmetrical. The measured NEXT can be approximated with a gamma or atruncated Normal distribution on log scale. The truncated Normal distribution has a better physicalmeaning since the number of NEXT pairs is limited. In practice, we might be concerned about NEXT frommore than one disturber. We need to calculate a power sum for multiple disturbers. We have 50, 3.16 ×1015 and 4.1 × 1011 different power sum NEXT values for 49 disturbers, 24 disturbers, and 10 disturbersrespectively in a 50 pair binder group. The manipulation of large numbers of power sums for 24 and 10disturbers is not easy. Hence, a direct computer simulation approach has been used in the past.

NEXT is dependent on frequency as well as on the relative location of the pairs in the binder group.However, location is not relevant for a full binder group. Cables differ from one another with respect toNEXT due to the cable design and manufacturing variations. The NEXT loss is usually stated in terms ofthe power sum of crosstalk from signals in all other pairs in the cable binder group at the given frequency.The NEXT model used for studies such as the one reported here is stated as expected 1% worst casepower sum crosstalk loss as a function of frequency. This means that on the average, 1% of the cablestested have power sum crosstalk loss worse (less) than the model at the given frequency. Such a model isa smooth curve Vs frequency, in which the loss decreases at about 15 dB per decade of frequency.Individual pair-to-pair loss on a single sample of cable is not a simple curve, and individual pairs generallyexhibit different loss Vs frequency curves. The power sum loss for less than a full binder group dependson the distribution of the pairs on which the crosstalking signal appears. Measurements for a 25-pairbinder group of a 24-AWG PIC cable are given in Figure B.7:

The study of transmission issues related to T1 systems established a first step in dealing with NEXTmodeling for simulations. The study not only tried to model NEXT loss with mean and standard deviationbut also initiated the use of 1% worst case NEXT value for overall system requirements. The reason isthat people were expecting better than 95% satisfactory T1 service at an error ratio of less than 10-6. Theuse of the 1% worst case for transmission engineering would allow multiple spans of T1 systems in anend-to-end service connection and also provide room for some unforeseen impairments.

The same better than 95% satisfactory service objective also applies to other digital subscriber linesystems such as DSL, HDSL, and ADSL. The 1% worst case NEXT model has also been used for DSLand HDSL simulation studies and test procedures. The piece-wise linear (log-log scale) NEXT modelsused for DSL and HDSL have loss values of 57 dB, 61 dB, and 67 dB for 49 disturbers, 10 disturbers, and1 disturber, respectively, at a frequency of 80 kHz. A simplified 49 disturber NEXT model that has 57 dB ofloss at 80 kHz and a linear (log-log scale) slope of -15 dB/decade has been most frequently used by ANSIT1E1.4 and can be expressed by

2/349 fxNEXT n ×=

where ( ) 6.014 4910818.8 nxn ××= − and n is the number of disturbers. Experimental results for a 25-pair

binder group of a 24-AWG PIC cable, support this model. In Figure B.8, those results are shown fitted tothe model. The difference between the fitted results and the ANSI model can be explained as thedifference between a 50-pair binder group of 22-AWG PIC cable (ANSI model) and a 25-pair binder groupof 24-AWG PIC cable.

B.4.2 Far end crosstalk, FEXT

Far-end crosstalk (FEXT) is defined as the effect of crosstalk due to adjacent transmitters. In other words,FEXT is due to crosstalk from adjacent transmitters at the transmitter end that couples to the receiver ofanother system. See Figure B.9. FEXT loss is similar but not equal to the combination of NEXT and thesubscriber loop channel losses over the coupling length. FEXT was also considered during T1transmission engineering efforts but was classified as a minor factor compared with NEXT. The effect ofFEXT for DSL and HDSL is very small and, hence, has been omitted in test procedures.


69

The effect of ADSL system self-FEXT can not be simply ignored. At high frequencies and for upstreamtransmitter disturbers on short loops, ADSL self-FEXT noise power can exceed that of HDSL NEXT andwhite background noise combined. A simplified FEXT model has been most frequently used by ANSIT1E1.4 and is expressed by

2249 )( klffHFEXT channel ×=

Where 2

)(fHchannel is the channel transfer function, k = 8 × 10-20 ( ) 6.049n× , n = number of disturbers, l

= the loop length in feet, and f = frequency in Hz. Experimental results for a 25-pair binder group of a 24-AWG PIC cable, support this model. In Figure B.10, those results are shown fitted to the model. Thedifference between the fitted results and the ANSI model can be explained as the difference between a50-pair binder group of 22-AWG PIC cable (ANSI model) and a 25-pair binder group of 24-AWG PICcable.

The simplified FEXT model assumes the channel transfer function and length of the coupling path matchthose of the disturbed system or more simply that the disturber system FEXT sources (transmitters) areco-located with the transmitter of the disturbed system. In the upstream direction, this underestimates theFEXT where the disturbers are closer to the central office than the victim signal transmitter.

B.4.3 Method for combining crosstalk contributions from unlike types of disturber

B.4.3.1 Base models for NEXT and FEXT

The modelling of interference contributions to an access DSL system due to crosstalk from other DSLsystems in the same cable is a fundamental part of spectral compatibility studies. The widely acceptedbase models due to work by Werner and others for near end crosstalk (NEXT) and far end crosstalk(FEXT) which are commonly used (see B.4.2) for this modelling are of the form:

[ ] 6.023

],[ nfXfSnfNext N=

[ ] 6.022][],,[ nlfXfHfSlnfFext F=

These expressions are for that interference power likely to be exceeded in 1% or less of cases where f isfrequency, n is the number of disturbing systems, l is the length of the cable, XN and XF are scalarconstants, S[f ] is the PSD of the interfering systems and H[f ] is the pair signal transfer function. There isan implicit assumption in these models that all the pairs involved are in the same binder group of the samecable and have a common length and also that all the interferers are of the same type.

There is a counter-intuitive aspect of these models relating to the n0.6 term. Intuitively it would beexpected for the interference power to be proportional to the number of disturbers (since the disturbersare independent) but instead there is the n0.6 factor. This is due to the fact that the quantity being dealtwith is not an average value or an expectation of any sort, but a 1% worst case.

If the proximity of pairs in a cable segment is maintained along its length, certain pairs (usually theproximate ones) contribute much more to the interference in a given pair than others do. When there arefew interferers (n small) if a single member is one of the proximate pairs the contribution to interference isdisproportionately increased. For this reason the model has to be biased for small numbers of interferersand this is the reason for the exponent of n being less than unity.

A difficulty arises when modelling complex access network scenarios though, where there may be manytypes of interferer. Suppose for example that the NEXT from n1 systems of spectrum S1[f ] and n2

systems of spectrum S2[f ] is considered. The obvious way of extending the model to cope with this is toadd the crosstalk power contributions according to the base model for each:

[ ] [ ] [ ] 6.022

6.011

23

23

nfXfSnfXfSfNext NN +=


70

The difficulty here is that each term in this expression is pessimistic enough for the 1% worst case, buttheir joint probability is much lower, so the combined model is excessively pessimistic. This can be seenby taking this expression and allowing S2[f ] = S1[f ] (the interferers have become of the same type). In thiscase the expression can be simplified to:

[ ] [ ] ( )6.02

6.011

23

nnfXfSfNext N +=

whereas the base model would in this case predict the lesser interference of:

[ ] [ ] ( ) 6.0211

23

nnfXfSfNext N +=

This appendix describes the recommended method for calculation of NEXT and FEXT contributions fromgroups of unlike disturbers. The method avoids making an over pessimistic calculation of total crosstalkcontribution which arises when assuming that all sub-groups of n interfering systems are using the worst npairs in a multi-pair cable. It does so without treating any sub-group differently so that there is only oneway of making the computation. The computation is such that in various limiting or trivial cases itconverges asymptotically to the base model for the reduced state. Also it never predicts a lower crosstalklevel when more disturbers are added.

The method is equally applicable to the calculation of NEXT and FEXT models.

B.4.3.2 Combining crosstalk from mixed disturber types

Instead of directly adding the crosstalk power terms, each term is first arbitrarily raised to the power 1/0.6before carrying out the summation. Then, after the summation, the resultant expression is raised to thepower 0.6. There is no simple physical justification for this process but it has been shown both analyticallybelow and elsewhere by means of Monte Carlo simulations that the method has many sound and realisticproperties.

B.4.3.3 Application to two NEXT terms

Take the example from B.4.3.1. The combined NEXT power would take the form:

[ ] [ ] [ ]6.0

6.01

6.022

6.01

6.011

23

23

+

= nfXfSnfXfSfNext NN

The first sound property is that if either inner term vanishes the model returns to the base model.Suppose for example that S2≡0 or n2=0. In this case the second term would vanish. This would leave thetwo arbitrarily introduced exponents acting on a single expression, so that they cancel out, returning theexpression to the base model.

The second sound property arises when S2≡S1. In this case the common factors S1[f ] XN f 3/2 can betaken out of the two inner terms, and further brought outside the enclosing brackets, leaving:

[ ] [ ] ( ) ( )6.0

6.01

6.02

6.01

6.011

23

+= nnfXfSfNext N

This in turn quickly collapses to:

[ ] [ ] ( ) 6.0211

23

nnfXfSfNext N +=

which is identical to the base model for the case of n1+n2 identical disturbers.

The same process can be applied to collections of more than two interference contributions.

B.4.3.4 Application to FEXT terms

The same process can also be applied to collections of FEXT interferers.


71

Take the case of three sources of FEXT at a given receiver. In this case there are n1 systems ofspectrum S1[f ] at range l1, a further n2 systems of spectrum S2[f ] at range l2 and yet another n3 systemsof spectrum S3[f ] at range l3.

The expected crosstalk is built in exactly the same way as before, taking the base model for each source,raising it to power 1/0.6, adding these expressions, and raising the sum to power 0.6:

[ ]( ) [ ]( )[ ]( )

6.0

6.01

6.033

2233

6.01

6.022

2222

6.01

6.011

2211

][

][][][

+

+=

nlfXfHfS

nlfXfHfSnlfXfHfSfFext

F

FF

In this case it is assumed that H1[f ] is the transfer function of the length l1 etc.

Even in this more complex case the same sound properties appear.

The first sound property is that if any of the inner terms vanishes the model returns to the simpler caseuntil when there is only one inner term left it returns to the base model. For FEXT though there are manymore ways in which a term can disappear. Instead of just S2≡0 or n2=0 there are also the possibilities l2=0and l2→∞. The latter arises because the product l2 H2

2[f ]→0 as l2→∞. In any of these cases the secondterm would vanish, and the equation is exactly as it would appear if the second crosstalk subgroup hadnot been considered in the first place. If in addition the third term disappears, for example because n3=0,the resulting equation is easily reduced to the base model for just the first subgroup of interferers.

The second sound property arises when for example S2≡S1 and l2=l1. This means that the first two termsactually relate to identical system types causing FEXT at the same location. As l2=l1 it can be assumedthat H2

2[f ]≡H12[f ]. In this case the common factors (S1[f ] H1

2[f ] XF f 2 l1)1/0.6 can be taken out of the first

two inner terms, leaving the expression:

[ ]( ) ( ) ( ) [ ]( )6.0

6.01

6.033

2233

6.01

6.02

6.01

6.01

6.01

122

11 ][][][

+

+= nlfXfHfSnnlfXfHfSfFext FF

The exponents around n1 and n2 now collapse to yield the sum n1+n2 which can then be taken back insidethe common factor to yield:

[ ] ( )( ) [ ]( )6.0

6.01

6.033

2233

6.01

6.0211

2211 ][][][

++= nlfXfHfSnnlfXfHfSfFext FF

This is exactly the form that would be obtained if the new method were applied to the simplified modellingsituation (of an increased number of identical disturbers at the same location) in the first place.

In addition if the terms subscripted with 3 were to vanish, for example because n3=0, then the expressionwould further simplify to the base model for the remaining interferers.

B.4.3.5 Crosstalk is non-decreasing

It will be apparent that the exponentiation operations, which are applied in this process, are applied toquantities of dimension power. This means of course that they are applied to real positive functions.After exponentiation the functions are still real and positive. As adding more disturbers is modelled byadding together these real positive functions and then applying a monotonic mapping to the sum (thesubsequent exponentiation with exponent 0.6) it follows that adding more disturbers always increases thecrosstalk.

B.4.3.6 All disturbers are treated equally

It should be apparent from the absolute symmetry of the method that all disturbers are treated equally. Itdoes not matter what order the disturbers are taken in the resulting expression is the same.


72

B.4.3.7 Adding NEXT and FEXT

The method should be separately applied to the NEXT terms and the FEXT terms to arrive at separateNEXT and FEXT disturbance power spectra. These power spectra should then be added.

The method should not itself be used for adding NEXT to FEXT. This is because it is perfectly feasiblefor the same proximate disturbing pair to contribute both NEXT and FEXT powers from different disturbingtransceivers, whereas it cannot contribute two lots of NEXT or two lots of FEXT from different disturbingtransceivers.


73

Table B.1 – American wire gauge (AWG) and metric wire

Metric & AWG Wire Gauges: R & C (0Hz/DC, 20deg C or 70deg F)

mm AWG mils nF/km nF/mi Ω/km Ω/mi Ω/kft NOTES Loss

(0.253) 30 10.03 mil 40 64.4 677 1090 2060.30 PE 40 64.4 494 792 1500.32 PE 28 12.64 mil 40 64.4 409 658 125

0.40 PE 50 80.5 280 451 84.4(0.405) 26 PIC 15.94 mil 51.6 83.0 273 439 83.1(0.455) 25 MAT 40.0 64.3 213 343 64.9 Metro. Area Trans., PIC

0.50 DW10 50.9 82.0 181 291 55.10.50 DUG 55 88.5 179 288 55.60.50 PE 50 64.4 179 288 54.5(0.511) 24 PIC 20.10 mil 51.6 83.0 172 276 52.3

0.60 PE 40 64.4 123 198 37.50.63 PE 45 72.4 113 182 34.5(0.644) 22 PIC 25.35 mil 51.6 83.0 108 174 32.9

0.70 PE 40 64.4 90 145 27.5

0.80 PE 40 64.4 69 111 21.0(0.812) 20 31.96 mil 66.6 107 20.3 for comparison, not telephony

0.90 DW12 51.1 82.3 55.5 89.3 16.90.90 PE 40 64.4 55 88.5 16.8(0.912) 19 PIC 51.6 83.0 53.8 86.6 16.4 35.89(0.965) 18 ½ 1) 72-118 116-19 141 227 43 PVC, copperclad steel, parallel

1.0 DW8 22.7 36.5 41.2 66.3 12.6 PVC, copper &cadmium(1.024) 18 40.3 mil 41.9 67.4 12.8 for comparison, not telephony

DW1 28.0 45.1 63.5 102 19.4 PVC, copper & cadmiumDW3 24.4 39.3 266 428 81.0 PVC, copperclad steelDW5 29.3 47.2 258 415 78.6 PVC, copperclad steelDW6 27.9 44.9 200 322 61.0 PVC, copperclad steelNOTES: ( ) = AWG conductor diameter → (mm) = not a normal metric size

PE = metric Polyethylene insulated cablePIC = AWG Polyethylene insulated cable, sometimes called "plastic insulated cable"as contrasted to older pulp or paper insulated cable.PVC = Polyvinyl chloride insulated cableDW = European drop wire, overhead/aerialDUG = European underground drop cable

1) F Drop Wire, AT-8668, aerial, parallel (flat, not twisted) 18 ½ AWG copperclad steelconductors, solid black PVC insulation, oval cross section, conductor diameter = 0.038 inch,43 ohm/kft = 227 ohm/mi, C = 0.116 µf/mi dry, C = 0.190 µf/mi wet (US Drop wire limits: 700feet or 25 ohms)

Attenuation/Loss at 1 kHz, 10 kHz, 100 kHz, 1 MHz, 10 MHz, 30 MHz: FUTURE?


74

Table B.2 - Cable model parameters for TP1 (0.4 mm or 26-gauge twisted pair)

Resistance cr0 sr0 ca xa

(value) 286.17578 Ω/km ∞ Ω/km 0.14769620 0.0

Inductance 0l ∞l b mf

(value) 675.36888µH/km

488.95186µH/km

0.92930728 806.33863 kHz

Capacitance ∞c 0c ec

(value) 49 nF/km 0.0 nF/km 0.0

Conductance 0g eg

(value) 43 nS/km .70

Table B.3 - Primary constants for TP1 (0.4 mm or 26-gauge twisted pair)

Frequency Resistance Inductance Capacitance Conductance

(Hz) (ΩΩΩΩ/km) (H/km) (F/km) (S/km)

5000 286.21516 673.7277e-6 49.e-9 16.701192e-6

10000 286.3332 672.26817e-6 49.e-9 27.131166e-6

20000 286.8039 669.55152e-6 49.e-9 44.074709e-6

50000 290.03566 662.28605e-6 49.e-9 83.70424e-6

100000 300.77488 651.94136e-6 49.e-9 135.97794e-6

1.e6 626.85069 572.86886e-6 49.e-9 681.50407e-6

10.e6 1.9606119e3 505.33352e-6 49.e-9 3.4156114e-3

10.5e6 2.0090081e3 504.66857e-6 49.e-9 3.5342801e-3

30.e6 3.3955368e3 495.20494e-6 49.e-9 7.3697598e-3


75

Table B.4: Cable parameters for 26-AWG PIC air core

MHz R

(ΩΩΩΩ/Km)

G

(µµµµS/Km)

L

(mH/Km)

C

(nF/Km)

αααα

(dB/Km)

ββββ

(rad/Km)

0Z (ΩΩΩΩ)

0.304 354.02 95 0.579 49.61 14.26 10.23 108.05

0.327 358.78 103 0.579 49.87 14.50 11.05 107.71

0.357 364.39 112 0.577 50.09 14.78 12.05 107.37

0.388 372.51 123 0.576 50.29 15.16 13.13 107.07

0.418 380.64 133 0.576 50.46 15.52 14.16 106.84

0.456 392.35 145 0.575 50.60 16.04 15.44 106.59

0.496 405.63 158 0.574 50.71 16.63 16.80 106.35

0.534 420.44 170 0.573 50.82 17.27 18.10 106.16

0.582 440.01 186 0.571 50.89 18.11 19.71 105.94

0.633 462.03 203 0.569 50.92 19.06 21.41 105.72

0.682 485.14 218 0.568 50.96 20.05 23.04 105.53

0.743 512.91 238 0.565 50.93 21.25 25.04 105.32

0.809 542.09 258 0.562 50.88 22.50 27.16 105.09

0.871 568.12 278 0.559 50.83 23.63 29.18 104.90

0.949 596.53 303 0.556 50.74 24.87 31.67 104.67

1.033 622.32 329 0.552 50.64 26.01 34.32 104.43

1.112 643.03 354 0.550 50.60 26.94 36.86 104.22

1.212 665.05 385 0.547 50.55 27.93 40.03 103.98

1.319 687.17 419 0.544 50.53 28.94 43.44 103.73

1.421 709.85 451 0.542 50.56 29.97 46.71 103.51

1.548 741.08 492 0.539 50.56 31.37 50.78 103.25

1.684 776.22 535 0.536 50.56 32.95 55.11 102.98

1.814 809.18 576 0.534 50.57 34.43 59.24 102.76

1.977 844.21 628 0.531 50.54 36.03 64.34 102.49

2.151 873.33 683 0.528 50.53 37.38 69.81 102.22

2.317 898.41 736 0.526 50.56 38.56 75.08 101.99

2.525 936.27 803 0.523 50.59 40.30 81.63 101.71

2.748 978.03 874 0.521 50.60 42.23 88.62 101.44

2.959 1014.30 942 0.519 50.63 43.91 95.29 101.21

3.225 1051.68 1026 0.516 50.65 45.67 103.60 100.94

3.509 1094.11 1118 0.514 50.68 47.66 112.51 100.67

3.78 1135.57 1205 0.512 50.73 49.59 121.02 100.44

(continued)


76

Table B.4 (concluded)

MHz R

(ΩΩΩΩ/Km)

G

(µµµµS/Km)

L

(mH/Km)

C

(nF/Km)

αααα

(dB/Km)

ββββ

(rad/Km)

0Z (ΩΩΩΩ)

4.119 1181.04 1314 0.509 50.76 51.74 131.59 100.18

4.482 1229.07 1430 0.507 50.80 54.01 142.92 99.92

4.827 1274.64 1542 0.505 50.85 56.15 153.76 99.70

5.26 1326.70 1682 0.503 50.89 58.63 167.24 99.44

5.724 1383.81 1831 0.501 50.92 61.34 181.65 99.19

6.165 1433.71 1975 0.499 50.98 63.71 195.47 98.98

6.718 1496.64 2153 0.497 51.01 66.71 212.61 98.74

7.31 1555.32 2345 0.495 51.05 69.53 230.98 98.50

7.874 1610.79 2529 0.494 51.11 72.20 248.56 98.30

8.58 1679.19 2758 0.492 51.15 75.49 270.44 98.07

9.337 1747.18 3003 0.490 51.19 78.77 293.84 97.84

10.06 1809.56 3238 0.489 51.25 81.79 316.25 97.66

10.96 1883.19 3532 0.487 51.29 85.37 344.13 97.44

11.92 1959.05 3846 0.485 51.34 89.07 373.99 97.23

12.84 2030.18 4148 0.484 51.40 92.53 402.58 97.06

14 2113.47 4524 0.483 51.44 96.60 438.13 96.86

15.23 2204.55 4926 0.481 51.48 101.05 476.19 96.66

16.4 2285.22 5313 0.480 51.55 105.00 512.65 96.50

17.87 2381.34 5793 0.479 51.58 109.73 557.99 96.32

19.45 2484.60 6309 0.477 51.62 114.80 606.52 96.14

20.95 2577.25 6803 0.476 51.69 119.36 653.04 95.99

22.83 2691.58 7419 0.475 51.72 125.00 710.86 95.82

24.84 2811.07 8079 0.474 51.76 130.90 772.74 95.65

26.76 2915.22 8711 0.473 51.82 136.07 832.08 95.52

29.16 3058.27 9499 0.472 51.85 143.12 905.83 95.36

31.73 3189.29 10343 0.470 51.89 149.65 984.79 95.21

34.17 3320.49 11153 0.470 51.94 156.16 1060.47 95.09

37.24 3469.12 12160 0.469 51.98 163.58 1154.59 94.95

40 3606.84 13077 0.468 52.03 170.45 1240.11 94.83


77

Table B.5 – Cable parameters for 26-AWG filled PIC

MHz R

(ohm/Km)

G

(µµµµS/Km)

L

(mH/Km)

C

(nF/Km)

αααα

(dB/Km)

ββββ

(rad/Km)

0Z(ohms)

0.304 397.8 48.3 0.685 46.44 14.267 10.767 121.409

0.327 398.7 52 0.682 46.77 14.376 11.618 120.79

0.357 399.5 56 0.68 47.09 14.488 12.677 120.134

0.388 400.3 60.2 0.677 47.38 14.592 13.813 119.54

0.418 401.6 64.9 0.676 47.64 14.701 14.902 119.076

0.456 403.9 69.7 0.674 47.9 14.854 16.259 118.579

0.496 407.3 75.1 0.672 48.15 15.042 17.715 118.127

0.534 413.9 80.8 0.671 48.37 15.336 19.112 117.77

0.582 423.1 86.9 0.67 48.59 15.734 20.851 117.386

0.633 437.7 93.6 0.668 48.78 16.328 22.712 117.032

0.682 454.6 101 0.667 48.95 17.005 24.487 116.751

0.743 478.8 108 0.665 49.07 17.959 26.678 116.446

0.809 506.4 117 0.663 49.14 19.045 29 116.164

0.871 533.3 125 0.661 49.18 20.1 31.197 115.937

0.949 565.9 135 0.658 49.15 21.376 33.902 115.689

1.033 595.1 145 0.654 49.08 22.528 36.772 115.457

1.112 616.4 156 0.652 49.05 23.378 39.512 115.27

1.212 635.4 168 0.649 48.98 24.153 42.923 115.064

1.319 649.9 181 0.646 48.95 24.758 46.595 114.871

1.421 665 195 0.644 48.97 25.376 50.143 114.713

1.548 688.1 209 0.643 48.99 26.309 54.578 114.539

1.684 721.9 226 0.641 49 27.65 59.311 114.374

1.814 758 243 0.639 48.99 29.074 63.801 114.239

1.977 796 261 0.637 48.93 30.583 69.341 114.089

2.151 821.3 281 0.634 48.86 31.607 75.261 113.947

2.317 840 302 0.633 48.86 32.377 80.973 113.83

2.525 871.4 326 0.632 48.85 33.646 88.12 113.699

2.748 913.4 351 0.63 48.82 35.321 95.73 113.575

2.959 948.7 377 0.628 48.81 36.73 102.987 113.472

3.225 979.9 406 0.627 48.77 38.004 112.019 113.358

3.509 1018 436 0.625 48.75 39.56 121.726 113.248

3.78 1057 471 0.624 48.75 41.111 131.005 113.158

(continued)


78


MHz R

(ohm/Km)

G

(µµµµS/Km)

L

(mH/Km)

C

(nF/Km)

αααα

(dB/Km)

ββββ

(rad/Km)

0Z(ohms)

4.119 1100 506 0.623 48.72 42.837 142.545 113.056

4.482 1153 545 0.621 48.69 44.956 154.887 112.959

4.827 1196 586 0.62 48.68 46.712 166.675 112.879

5.26 1243 630 0.619 48.66 48.622 181.378 112.789

5.724 1300 679 0.618 48.63 50.907 197.094 112.702

6.165 1347 730 0.617 48.62 52.81 212.146 112.631

6.718 1403 786 0.616 48.6 55.087 230.88 112.551

7.31 1467 846 0.614 48.57 57.697 250.927 112.473

7.874 1519 909 0.614 48.57 59.805 270.094 112.41

8.58 1581 980 0.613 48.54 62.34 293.981 112.338

9.337 1650 1054 0.612 48.52 65.163 319.541 112.269

10.06 1712 1134 0.611 48.52 67.691 344.03 112.212

10.96 1785 1222 0.61 48.5 70.701 374.474 112.148

11.92 1872 1312 0.609 48.48 74.235 407.112 112.086

12.84 1943 1415 0.608 48.48 77.176 438.268 112.035

14 2024 1522 0.608 48.46 80.519 477.129 111.978

15.23 2129 1637 0.607 48.43 84.794 518.684 111.923

16.4 2206 1763 0.606 48.44 87.994 558.535 111.877

17.87 2307 1894 0.605 48.42 92.156 608.048 111.826

19.45 2431 2042 0.605 48.4 97.247 661.129 111.776

20.95 2513 2196 0.604 48.4 100.693 711.891 111.736

22.83 2636 2363 0.603 48.38 105.808 774.997 111.69

24.84 2759 2545 0.603 48.36 110.912 842.743 111.646

26.76 2886 2734 0.603 48.37 116.164 907.614 111.609

29.16 2996 2947 0.602 48.35 120.82 988.173 111.568

31.73 3149 3170 0.601 48.33 127.189 1074.618 111.529

34.17 3301 3411 0.601 48.35 133.523 1157.406 111.496

37.24 3470 3673 0.6 48.33 140.573 1260.329 111.46

40 3671 3946 0.6 48.34 148.837 1353.774 111.429


79

Table B.6 – Cable model parameters for TP2 (0.5 mm or 24-gauge twisted pair)

Resistance cr0 sr0 ca ca

(value) 174.55888Ohms/km

∞ Ohms/km 0.053073481 0.0


(value) 617.29539 µH/km 478.97099µH/km

1.1529766 553.760 kHz


(value) 50 nF/km 0.0 nF/km 0.0

Conductance 0g eg

(value) 234.87476fMho/km

1.38

Table B.7 – Primary constants for TP2 (0.5 mm or 24-gauge twisted pair)



5000 174.62121 616.69018e-6 50.e-9 29.882364e-9

10000 174.8078 615.95674e-6 50.e-9 77.774343e-9

20000 175.54826 614.35345e-6 50.e-9 202.42201e-9

50000 180.48643 609.15855e-6 50.e-9 716.82799e-9

100000 195.44702 600.41634e-6 50.e-9 1.8656765e-6

1.e6 482.06141 525.43983e-6 50.e-9 44.754463e-6

10.e6 1.5178833e3 483.72215e-6 50.e-9 1.0735848e-3

30.e6 2.6289488e3 480.34357e-6 50.e-9 4.8894913e-3


80

Table B.8 – Cable parameters for 24-AWG PIC air core

MHz R

(ΩΩΩΩ/Km)

G

(µµµµS/Km)

L

(mH/Km)

C

(nF/Km)

αααα

(dB/Km)

ββββ

(rad/Km)

0Z

(ΩΩΩΩ)

0.304 269.87 98 0.581 51.3 11.05 10.42 106.39

0.327 280.59 105 0.578 51.22 11.51 11.19 106.24

0.357 291.94 115 0.574 51.13 12.01 12.14 105.99

0.388 302.05 125 0.57 51.07 12.46 13.16 105.67

0.418 310.67 134 0.567 51.09 12.86 14.14 105.35

0.456 321.11 146 0.563 51.12 13.34 15.36 104.96

0.496 333.53 159 0.559 51.19 13.92 16.67 104.53

0.534 346.45 172 0.556 51.29 14.51 17.92 104.16

0.582 362.84 188 0.552 51.36 15.27 19.47 103.71

0.633 378.68 204 0.548 51.41 16.01 21.12 103.27

0.682 391.83 220 0.545 51.45 16.63 22.68 102.89

0.743 405.72 240 0.54 51.48 17.29 24.63 102.46

0.809 420.66 262 0.537 51.53 18.01 26.71 102.03

0.871 436.52 282 0.534 51.62 18.76 28.72 101.69

0.949 455.25 308 0.53 51.66 19.64 31.2 101.29

1.033 472.06 335 0.526 51.69 20.45 33.85 100.92

1.112 487.37 362 0.524 51.74 21.18 36.38 100.62

1.212 507.85 394 0.521 51.78 22.15 39.54 100.28

1.319 527.94 429 0.518 51.8 23.11 42.91 99.96

1.421 545.55 463 0.515 51.84 23.95 46.14 99.7

1.548 568.5 504 0.513 51.86 25.03 50.15 99.42

1.684 590.64 549 0.51 51.87 26.09 54.44 99.16

1.814 611.79 592 0.508 51.9 27.09 58.54 98.95

1.977 636.98 645 0.506 51.9 28.28 63.65 98.72

2.151 662.83 702 0.504 51.9 29.5 69.11 98.51

2.317 686.62 756 0.502 51.92 30.62 74.34 98.35

2.525 715.83 823 0.5 51.9 32 80.84 98.17

2.748 744.89 896 0.498 51.88 33.37 87.78 98

2.959 772.18 965 0.497 51.9 34.65 94.44 97.87

3.225 804.85 1051 0.495 51.87 36.19 102.72 97.73

3.509 838.52 1143 0.494 51.84 37.77 111.57 97.61

3.78 869.26 1231 0.493 51.84 39.21 120.05 97.51

(continued)


81


MHz R

(ΩΩΩΩ/Km)

G

(µµµµS/Km)

L

(mH/Km)

C

(nF/Km)

αααα

(dB/Km)

ββββ

(rad/Km)

0Z

(ΩΩΩΩ)

4.119 905.99 1341 0.492 51.81 40.93 130.59 97.41

4.482 944.23 1458 0.49 51.77 42.73 141.86 97.31

4.827 978.86 1570 0.489 51.77 44.35 152.67 97.24

5.26 1021.06 1709 0.488 51.72 46.33 166.1 97.16

5.724 1064.19 1859 0.487 51.68 48.35 180.48 97.1

6.165 1104.2 2002 0.487 51.67 50.22 194.25 97.05

6.718 1152.18 2179 0.486 51.63 52.47 211.37 96.99

7.31 1201.2 2369 0.485 51.58 54.77 229.7 96.95

7.874 1246.16 2551 0.484 51.57 56.88 247.25 96.91

8.58 1300.4 2778 0.484 51.52 59.43 269.08 96.88

9.337 1356.48 3020 0.483 51.47 62.06 292.44 96.85

10.06 1407.6 3252 0.482 51.46 64.46 314.84 96.82

10.96 1469.55 3540 0.482 51.41 67.37 342.67 96.8

11.92 1533.1 3849 0.481 51.37 70.36 372.47 96.78

12.84 1591.69 4144 0.481 51.35 73.12 401.04 96.77

14 1662.07 4512 0.48 51.3 76.44 436.54 96.76

15.23 1735.64 4905 0.48 51.26 79.91 474.55 96.75

16.4 1802.69 5282 0.48 51.25 83.08 510.99 96.75

17.87 1886.11 5750 0.479 51.2 87.03 556.28 96.74

19.45 1973.16 6251 0.479 51.15 91.14 604.76 96.74

20.95 2049.06 6732 0.479 51.14 94.75 651.24 96.74

22.83 2145.43 7329 0.478 51.09 99.32 709.04 96.75

24.84 2251.42 7968 0.478 51.05 104.34 770.89 96.75

26.76 2338.22 8581 0.478 51.04 108.48 830.23 96.76

29.16 2453.81 9342 0.477 51 113.98 903.98 96.76

31.73 2573.37 10157 0.477 50.95 119.68 982.89 96.77

34.17 2678.96 10938 0.477 50.94 124.73 1058.57 96.78

37.24 2816.8 11909 0.477 50.9 131.31 1152.64 96.79

40 2935 12792 0.477 50.9 136.97 1238.16 96.79


82

Table B.9 – Cable parameters for 22-AWG PIC air core

MHz R

(ΩΩΩΩ/Km)

G

(µµµµS/Km)

L

(mH/Km)

C

(nF/Km)

αααα

(dB/Km)

ββββ

(rad/Km)

0Z

(ΩΩΩΩ)

0.304 197.3 96 0.537 50.27 8.33 9.92 103.38

0.327 202.3 104 0.537 50.33 8.55 10.69 103.28

0.357 207.61 113 0.536 50.41 8.79 11.65 103.08

0.388 214 123 0.534 50.51 9.09 12.67 102.83

0.418 220.28 133 0.533 50.64 9.38 13.64 102.57

0.456 227.97 145 0.531 50.76 9.74 14.86 102.25

0.496 236.78 159 0.528 50.89 10.16 16.15 101.89

0.534 246.08 171 0.527 51.03 10.59 17.39 101.58

0.582 258.3 187 0.524 51.15 11.16 18.93 101.21

0.633 271.25 204 0.521 51.27 11.76 20.56 100.83

0.682 285.09 220 0.519 51.38 12.41 22.13 100.52

0.743 301.4 240 0.516 51.45 13.17 24.06 100.15

0.809 317.76 262 0.513 51.51 13.93 26.11 99.79

0.871 333.2 282 0.51 51.56 14.66 28.07 99.5

0.949 349.26 308 0.507 51.58 15.42 30.5 99.16

1.033 363.23 335 0.504 51.59 16.09 33.09 98.84

1.112 374.44 361 0.502 51.64 16.64 35.58 98.58

1.212 386.35 394 0.499 51.67 17.23 38.68 98.29

1.319 399.1 429 0.497 51.72 17.85 42.01 98.02

1.421 412.51 462 0.495 51.79 18.5 45.21 97.8

1.548 431.41 504 0.493 51.83 19.4 49.19 97.56

1.684 452.73 549 0.491 51.86 20.42 53.43 97.34

1.814 471.26 592 0.49 51.89 21.3 57.47 97.15

1.977 492.08 645 0.488 51.89 22.3 62.49 96.96

2.151 508.82 701 0.486 51.89 23.11 67.88 96.78

2.317 525.41 756 0.485 51.93 23.91 73.06 96.63

2.525 552.67 824 0.483 51.94 25.21 79.49 96.48

2.748 580.94 896 0.482 51.92 26.55 86.35 96.33

2.959 602.38 965 0.481 51.92 27.57 92.89 96.22

3.225 621.3 1052 0.479 51.9 28.5 101.06 96.1

3.509 650.94 1144 0.478 51.9 29.91 109.84 95.99

3.78 679.27 1233 0.477 51.9 31.25 118.21 95.9

(continued)


83


MHz R

(ΩΩΩΩ/Km)

G

(µµµµS/Km)

L

(mH/Km)

C

(nF/Km)

αααα

(dB/Km)

ββββ

(rad/Km)

0Z

(ΩΩΩΩ)

4.119 704.13 1342 0.476 51.87 32.45 128.6 95.81

4.482 733.72 1460 0.475 51.85 33.87 139.78 95.73

4.827 765.09 1573 0.475 51.85 35.36 150.44 95.67

5.26 795.07 1713 0.474 51.82 36.8 163.73 95.6

5.724 835.86 1862 0.473 51.78 38.74 177.94 95.54

6.165 861.63 2006 0.472 51.78 39.99 191.54 95.5

6.718 900.91 2184 0.471 51.75 41.87 208.49 95.45

7.31 937.53 2375 0.471 51.71 43.63 226.63 95.41

7.874 974.62 2558 0.47 51.71 45.41 243.97 95.37

8.58 1016.13 2786 0.47 51.67 47.41 265.58 95.34

9.337 1057.87 3029 0.469 51.63 49.42 288.69 95.31

10.06 1097.92 3262 0.469 51.63 51.35 310.85 95.29

10.96 1145.72 3552 0.468 51.59 53.66 338.4 95.27

11.92 1191.83 3862 0.468 51.55 55.9 367.9 95.26

12.84 1236.98 4159 0.468 51.54 58.08 396.17 95.24

14 1290.75 4529 0.467 51.51 60.69 431.33 95.23

15.23 1349.41 4925 0.467 51.47 63.54 468.97 95.23

16.4 1403.08 5304 0.467 51.46 66.14 505.05 95.22

17.87 1464.42 5775 0.466 51.42 69.14 549.89 95.22

19.45 1534.46 6280 0.466 51.38 72.54 597.91 95.21

20.95 1591.96 6763 0.466 51.38 75.36 643.91 95.21

22.83 1668.08 7364 0.465 51.34 79.08 701.15 95.21

24.84 1737.61 8008 0.465 51.3 82.51 762.44 95.22

26.76 1808.44 8624 0.465 51.3 85.99 821.18 95.22

29.16 1889.94 9391 0.465 51.26 90.02 894.26 95.22

31.73 1994.92 10212 0.465 51.23 95.14 972.48 95.23

34.17 2070.38 10999 0.465 51.23 98.9 1047.44 95.23

37.24 2175.93 11977 0.464 51.19 104.11 1140.68 95.24

40 2278.32 12866 0.464 51.19 109.13 1225.38 95.25


84

Table B.10 – Cable model parameters for TP3 (DW10 reinforced .5 mm copper PVC-insulated steelstrength member, polyethelene sheath)


(value) 180.93 Ohms/km ∞ Ohms/km .0497223 0


(value) 728.87 µH/km 543.43 µH/km .75577086 718888 Hz.


(value) 51 nF/km 63.8 nF/km .11584622

Conductance 0g eg

(value) 89 nMho/km .856

Table B.11 – Primary constants for TP3 (DW10 reinforced .5 mm copper PVC-insulated steelstrength member, polyethelene sheath)



5000 180.98245 724.62777e-6 74.722723e-9 130.65969e-6

10000 181.13951 721.81902e-6 72.886768e-9 236.50605e-6

20000 181.76372 717.26896e-6 71.192474e-9 428.0977e-6

50000 185.96294 707.04768e-6 69.151688e-9 938.00385e-6

100000 199.01927 694.78496e-6 67.745589e-9 1.6978732e-3

1.5e6 579.72026 611.02577e-6 63.21713e-9 17.246997e-3

10.e6 1.493348e3 565.7413e-6 60.79255e-9 87.504681e-3

30.e6 2.5864318e3 553.86667e-6 59.613734e-9 224.11821e-3


85

Table B.12 – Cable model parameters for FP (1.14 mm flat cable)


(value) 41.16 Ohms/km ∞ Ohms/km .001218 0


(value) 1000 µH/km 911 µH/km 1.195 174.2 kHz


(value) 22.68 nF/km 31.78 nF/km .1109

Conductance 0g eg

(value) 53 nMho/km .88

Table B.13 – Primary constants for FP (1.14 mm flat cable)



5000 41.268736 998.73982e-6 35.041871e-9 95.360709e-6

10000 41.589888 997.16583e-6 34.127572e-9 175.49949e-6

20000 42.805363 993.76481e-6 33.280903e-9 322.98493e-6

50000 49.316246 983.62766e-6 32.257008e-9 723.38496e-6

100000 62.284991 969.66713e-6 31.548702e-9 1.3312998e-3

1.e6 186.92411 920.40732e-6 29.550852e-9 10.098942e-3

10.e6 590.76171 911.20963e-6 28.003118e-9 76.608308e-3

30.e6 1.023223e3 910.69563e-6 27.392833e-9 201.43854e-3


86

Table B.14 – Cable model parameters for category 5 twisted pair


(value) 176.6 Ohms/km ∞ Ohms/km .0500079494 0.0


(value) 1090.8 µH/km 504.5 µH/km 0.705 32570 kHz


(value) 48.55 nF/km 0.0 nF/km 0.0

Conductance 0g eg

(value) 1.47653 nS/km .91

Table B.15 – Primary constants for category 5 twisted pair



5000 176.656720 967.308142e-6 48.55e-9 3.430086e-6

10000 176.826554 913.078780e-6 48.55e-9 6.445287e-6

20000 177.501041 847.551900e-6 48.55e-9 12.110988e-6

50000 182.020084 753.691218e-6 48.55e-9 27.880784e-6

100000 195.898798 687.417012e-6 48.55e-9 52.389261e-6

1.e6 475.172462 552.634084e-6 48.55e-9 425.835904e-6

10.e6 1.4954809e3 514.663928e-6 48.55e-9 3.461324e-3

30.e6 2.5901370e3 509.228994e-6 48.55e-9 9.406382e-3


87

Table B.16 – Cable parameters, two-pair twisted drop

MHz R

(ΩΩΩΩ//Kft)

L

(mH/Kft)

C

(nF/Kft)

αααα

(dB/Kft)

0Z(ohm)

0.772 112.6412 0.143576 14.15747 4.85443771 100.7043

0.819094 116.4074 0.141203 13.94644 5.02088326 100.6214

0.869062 120.112 0.140489 13.89817 5.18481854 100.5408

0.922077 124.0752 0.140184 13.88958 5.36006449 100.4626

0.978327 128.3016 0.139989 13.89123 5.54683894 100.3867

1.038008 132.5838 0.139838 13.89671 5.73618274 100.313

1.101329 137.0843 0.139706 13.90334 5.93512422 100.2415

1.168514 142.5063 0.139581 13.91021 6.17415048 100.172

1.239797 147.1853 0.139459 13.91676 6.38116738 100.1046

1.315428 151.9907 0.139336 13.92271 6.59381465 100.0391

1.395673 157.6941 0.13921 13.92782 6.84559689 99.97558

1.480813 163.022 0.137695 13.79328 7.0812544 99.91388

1.571148 168.7727 0.138949 13.93559 7.3354461 99.85398

1.666992 174.1886 0.138813 13.93821 7.57525379 99.79583

1.768684 180.6494 0.138676 13.94017 7.86066953 99.73937

1.876579 187.1776 0.138536 13.94142 8.14921318 99.68456

1.991056 194.3419 0.138396 13.94219 8.46564642 99.63135

2.112517 201.1778 0.138254 13.94235 8.7679671 99.5797

2.241387 209.5537 0.138115 13.94233 9.13761936 99.52955

2.378118 216.4363 0.137974 13.94175 9.44235495 99.48086

2.523191 223.9826 0.137835 13.94097 9.77621976 99.43359

2.677113 232.2693 0.137699 13.94009 10.142589 99.3877

2.840425 240.6053 0.137566 13.93908 10.5113113 99.34315

3.0137 250.1631 0.137436 13.9381 10.9336262 99.2999

3.197545 259.1278 0.137309 13.93699 11.3302283 99.25792

3.392605 268.8572 0.137186 13.93593 11.7604684 99.21715

3.599564 279.5037 0.137066 13.93491 12.2310533 99.17758

3.819149 290.5045 0.13695 13.9339 12.7173706 99.13916

4.052129 300.6599 0.136837 13.93283 13.1668983 99.10186

4.299321 311.7282 0.136727 13.93179 13.6566036 99.06565

4.561593 323.8352 0.13662 13.93077 14.1920396 99.0305

4.839864 336.0791 0.136515 13.92969 14.7337063 98.99637

5.13511 349.7153 0.136413 13.92858 15.3366472 98.96324

5.448368 363.3367 0.136312 13.92734 15.939193 98.93107

5.780735 377.6032 0.136212 13.92597 16.5702788 98.89984

6.133378 393.3168 0.136114 13.92445 17.2651257 98.86953

6.507533 407.97 0.135994 13.92044 17.9136803 98.84009

6.904512 426.5291 0.135918 13.92074 18.7340146 98.81152

7.325709 443.104 0.135701 13.90626 19.4674802 98.78378

(continued)


88


MHz R

(ΩΩΩΩ//Kft)

L

(mH/Kft)

C

(nF/Kft)

αααα

(dB/Kft)

0Z(ohm)

7.772599 461.1752 0.135721 13.91594 20.266952 98.75685

8.246752 479.3825 0.135621 13.91307 21.072676 98.7307

8.749829 499.3066 0.13552 13.90983 21.9541433 98.70532

9.283595 518.4216 0.135417 13.90623 22.8003059 98.68068

9.849923 539.9883 0.135312 13.9022 23.7545736 98.65676

10.4508 562.2894 0.135205 13.89775 24.7414418 98.63353

11.08833 586.0891 0.135096 13.89286 25.7945582 98.61098

11.76475 610.2553 0.134984 13.88752 26.8641089 98.58909

12.48244 635.0644 0.13487 13.88174 27.962258 98.56784

13.2439 662.0816 0.134752 13.87545 29.1579437 98.54721

14.05182 691.5593 0.134631 13.86865 30.462329 98.52718

14.90903 720.1421 0.134508 13.86144 31.7276262 98.50774

15.81852 749.6236 0.134382 13.85377 33.0328382 98.48886

16.7835 783.4812 0.134252 13.84553 34.5312289 98.47053

17.80735 817.8867 0.13412 13.83685 36.0541353 98.45274

18.89365 851.7503 0.133985 13.82779 37.5534985 98.43547

20.04622 890.3597 0.133846 13.81817 39.2624684 98.4187

21.2691 925.5264 0.133706 13.80831 40.819977 98.40242

22.56658 969.5871 0.133561 13.79778 42.7701295 98.38661

23.94321 1010.435 0.133416 13.78702 44.5789511 98.37127

25.40382 1060.989 0.133265 13.77562 46.816403 98.35637

26.95353 1104.402 0.133115 13.76416 48.7392081 98.34191

28.59778 1152.924 0.132962 13.75225 50.8878273 98.32787

30.34234 1191.673 0.132811 13.74044 52.6053964 98.31424

32.19331 1244.775 0.132654 13.72787 54.9569524 98.30101

34.1572 1319.738 0.132489 13.71443 58.2741802 98.28816

36.2409 1367.755 0.132332 13.70164 60.4020976 98.27569

38.4517 1441.865 0.132167 13.68794 63.682754 98.26358

40 1488.43 0.132058 13.67882 65.7446239 98.25571


89

Table B.17 – Cable parameters, two-pair quad drop

MHz R

(ΩΩΩΩ//Kft)

L

(mH/Kft)

C

(nF/Kft)

αααα

(dB/Kft)

0Z

(ohms)

Pair 1 Pair 2 Pair 1 Pair 2 Pair 1 Pair 2 Pair 1 Pair 2 Pair 1 Pair 2

0.772 125.4 129.2 0.156 0.178 12.53 13.69 4.879 4.923 111.5 113.9

0.82 130.1 134.7 0.14 0.141 11.28 10.83 5.065 5.132 111.5 113.9

0.87 135 139.9 0.145 0.148 11.67 11.38 5.259 5.331 111.4 113.9

0.923 139.5 145.4 0.146 0.149 11.75 11.51 5.439 5.54 111.3 113.9

0.979 143.4 149 0.146 0.15 11.79 11.57 5.592 5.676 111.3 113.9

1.038 148.3 154.1 0.146 0.151 11.81 11.6 5.786 5.87 111.2 113.9

1.103 153 158.3 0.146 0.151 11.82 11.62 5.971 6.031 111.2 113.9

1.169 158.1 164.2 0.142 0.151 11.5 11.63 6.174 6.256 111.1 113.9

1.241 163.8 170.3 0.146 0.148 11.84 11.43 6.402 6.487 111.1 113.9

1.317 170.3 176.6 0.146 0.153 11.84 11.75 6.658 6.729 111 113.9

1.396 176.3 182.6 0.146 0.151 11.85 11.64 6.896 6.957 111 113.9

1.483 180.7 188.3 0.146 0.151 11.85 11.64 7.07 7.175 110.9 113.9

1.572 187.2 194.3 0.146 0.151 11.85 11.64 7.33 7.404 110.9 113.9

1.668 193.5 202.3 0.146 0.151 11.85 11.64 7.579 7.71 110.8 113.9

1.771 200.4 209.2 0.145 0.151 11.85 11.64 7.851 7.969 110.8 113.9

1.877 207.1 215.6 0.145 0.151 11.85 11.64 8.118 8.214 110.7 113.9

1.993 214 223 0.145 0.151 11.85 11.64 8.39 8.497 110.7 113.9

2.114 221.6 232.1 0.145 0.151 11.85 11.64 8.691 8.843 110.6 113.9

2.243 229.6 239.2 0.145 0.151 11.85 11.64 9.01 9.116 110.6 113.9

2.381 237.2 248.3 0.145 0.151 11.85 11.64 9.312 9.46 110.6 113.9

2.523 245.6 257.7 0.145 0.151 11.85 11.64 9.642 9.819 110.5 113.9

2.68 254.9 267.4 0.145 0.151 11.85 11.64 10.01 10.19 110.5 113.9

2.843 263.5 276.3 0.145 0.151 11.85 11.64 10.35 10.53 110.5 113.9

3.016 273.1 287.4 0.144 0.151 11.85 11.64 10.74 10.95 110.4 113.9

3.201 283.3 297.2 0.144 0.151 11.84 11.64 11.14 11.32 110.4 113.9

3.393 293.1 307.7 0.144 0.151 11.84 11.65 11.53 11.72 110.4 113.9

3.604 303.9 318.5 0.144 0.15 11.84 11.59 11.96 12.13 110.3 113.9

3.822 314.7 331.8 0.144 0.149 11.84 11.46 12.39 12.64 110.3 113.9

4.055 326.9 343.3 0.144 0.151 11.84 11.63 12.87 13.08 110.3 113.9

4.304 338.9 355.5 0.144 0.151 11.84 11.63 13.34 13.54 110.2 113.9

4.562 351.8 370.7 0.144 0.151 11.84 11.63 13.85 14.12 110.2 113.9

4.845 364.6 384.7 0.144 0.151 11.84 11.63 14.36 14.66 110.2 113.9

5.139 377.6 400.6 0.144 0.151 11.84 11.63 14.88 15.26 110.2 113.9

5.453 391.6 413.7 0.144 0.151 11.84 11.63 15.43 15.76 110.1 113.9

5.788 407 429.1 0.144 0.151 11.84 11.63 16.04 16.35 110.1 113.9

(continued)


90


MHz R

(ΩΩΩΩ//Kft)

L

(mH/Kft)

C

(nF/Kft)

αααα(

dB/Kft)

0Z

(ohms)

Pair 1 Pair 2 Pair 1 Pair 2 Pair 1 Pair 2 Pair 1 Pair 2 Pair 1 Pair 2

6.133 424.1 445.9 0.143 0.151 11.84 11.63 16.72 16.99 110.1 113.9

6.515 439.6 463.9 0.143 0.151 11.84 11.63 17.34 17.68 110.1 113.9

6.91 456.8 482.3 0.143 0.151 11.84 11.63 18.02 18.38 110 113.9

7.331 474.7 501.6 0.142 0.151 11.77 11.63 18.73 19.11 110 113.9

7.782 494.7 520.4 0.143 0.151 11.79 11.63 19.52 19.83 110 113.9

8.247 513.7 542.6 0.143 0.151 11.83 11.63 20.27 20.67 110 113.9

8.76 533.8 565.7 0.143 0.151 11.83 11.63 21.07 21.55 109.9 113.9

9.291 553.8 594.5 0.143 0.151 11.83 11.63 21.87 22.65 109.9 113.9

9.857 578.8 610.9 0.143 0.151 11.83 11.63 22.85 23.28 109.9 113.9

10.46 596.9 635.1 0.143 0.151 11.83 11.63 23.57 24.2 109.9 113.9

11.09 620.5 661.2 0.143 0.151 11.83 11.63 24.51 25.19 109.9 113.9

11.78 646.7 685.5 0.143 0.151 11.83 11.63 25.55 26.12 109.9 113.9

12.49 672.1 712.6 0.143 0.151 11.83 11.63 26.55 27.15 109.8 113.9

13.25 701.4 740.6 0.143 0.151 11.83 11.63 27.72 28.22 109.8 113.9

14.07 728.4 772.8 0.143 0.151 11.82 11.63 28.79 29.44 109.8 113.9

14.91 757.8 804.1 0.143 0.151 11.82 11.63 29.95 30.64 109.8 113.9

15.84 789.1 837.8 0.142 0.151 11.82 11.63 31.2 31.92 109.8 113.9

16.8 821.2 872.1 0.142 0.151 11.82 11.63 32.47 33.23 109.8 113.9

17.82 854.9 906.3 0.142 0.151 11.82 11.63 33.81 34.53 109.7 113.9

18.92 892.9 945.9 0.142 0.151 11.82 11.63 35.32 36.04 109.7 113.9

20.05 925.2 991.2 0.142 0.151 11.82 11.63 36.6 37.77 109.7 113.9

21.29 964.6 1028 0.142 0.151 11.82 11.63 38.16 39.18 109.7 113.9

22.58 1012 1078 0.142 0.151 11.81 11.63 40.03 41.08 109.7 113.9

23.96 1053 1120 0.142 0.151 11.81 11.63 41.66 42.67 109.7 113.9

25.43 1104 1169 0.142 0.151 11.81 11.63 43.69 44.56 109.7 113.9

26.95 1142 1226 0.142 0.151 11.81 11.63 45.19 46.71 109.7 113.9

28.63 1194 1283 0.142 0.151 11.81 11.63 47.26 48.9 109.6 113.9

30.37 1229 1319 0.142 0.151 11.81 11.63 48.65 50.26 109.6 113.9

32.22 1306 1378 0.142 0.151 11.81 11.63 51.71 52.49 109.6 113.9

34.2 1352 1439 0.142 0.151 11.8 11.63 53.53 54.83 109.6 113.9

36.24 1395 1507 0.142 0.151 11.8 11.63 55.26 57.4 109.6 113.9

38.5 1482 1555 0.142 0.151 11.8 11.63 58.69 59.24 109.6 113.9

40 1530 1622 0.142 0.151 11.8 11.63 60.6 61.81 109.6 113.9


91

Table B.18 – Cable parameters, flat-pair Drop

MHz R

(ΩΩΩΩ//Kft)

L

(mH/Kft)

C

(nF/Kft)

αααα

(dB/Kft)

0Z(ohms)

0.772 170.5 0.129 10.5 6.681 110.8

0.82 173.7 0.13 10.43 6.76 111.5

0.87 177.6 0.131 10.39 6.87 112.2

0.923 182.4 0.132 10.36 7.012 112.9

0.979 188 0.133 10.34 7.185 113.6

1.038 194.5 0.135 10.33 7.389 114.2

1.103 201.8 0.136 10.33 7.625 114.8

1.169 209.9 0.138 10.32 7.892 115.5

1.241 219 0.139 10.33 8.19 116.1

1.317 228.9 0.141 10.34 8.519 116.6

1.396 239.8 0.142 10.35 8.88 117.2

1.483 251.5 0.144 10.36 9.273 117.7

1.572 264.2 0.145 10.38 9.696 118.3

1.668 277.8 0.147 10.39 10.15 118.8

1.771 292.3 0.148 10.41 10.64 119.3

1.877 307.8 0.149 10.42 11.16 119.7

1.993 324.2 0.151 10.43 11.7 120.2

2.114 341.6 0.152 10.45 12.28 120.7

2.243 359.9 0.153 10.46 12.9 121.1

2.381 379.1 0.155 10.47 13.54 121.5

2.523 399.4 0.156 10.49 14.21 121.9

2.68 420.6 0.157 10.49 14.92 122.3

2.843 442.8 0.158 10.5 15.66 122.7

3.016 466 0.159 10.51 16.43 123.1

3.201 490.1 0.16 10.52 17.23 123.5

3.393 515.3 0.161 10.52 18.06 123.8

3.604 541.4 0.162 10.53 18.92 124.2

3.822 568.5 0.163 10.54 19.81 124.5

4.055 596.6 0.164 10.54 20.74 124.9

4.304 625.7 0.165 10.54 21.69 125.2

4.562 655.8 0.166 10.55 22.68 125.5

4.845 686.9 0.167 10.55 23.7 125.8

5.139 719.1 0.168 10.55 24.75 126.1

5.453 752.2 0.169 10.56 25.83 126.4

(continued)


92


MHz R

(ΩΩΩΩ//Kft)

L

(mH/Kft)

C

(nF/Kft)

αααα

(dB/Kft)

0Z(ohms)

5.788 786.3 0.169 10.56 26.95 126.6

6.133 821.4 0.17 10.56 28.09 126.9

6.515 857.5 0.171 10.56 29.27 127.2

6.91 894.7 0.171 10.56 30.47 127.4

7.331 932.8 0.172 10.56 31.71 127.7

7.782 972 0.173 10.56 32.98 127.9

8.247 1012 0.173 10.56 34.28 128.1

8.76 1053 0.174 10.56 35.62 128.3

9.291 1095 0.175 10.56 36.98 128.6

9.857 1139 0.175 10.56 38.38 128.8

10.46 1183 0.176 10.56 39.8 129

11.09 1228 0.176 10.56 41.26 129.2

11.78 1274 0.177 10.56 42.75 129.4

12.49 1322 0.177 10.55 44.27 129.6

13.25 1370 0.178 10.55 45.82 129.7

14.07 1419 0.178 10.55 47.41 129.9

14.91 1469 0.178 10.54 49.02 130.1

15.84 1521 0.179 10.55 50.67 130.2

16.8 1573 0.179 10.55 52.34 130.4

17.82 1626 0.18 10.54 54.05 130.6

18.92 1680 0.18 10.55 55.79 130.7

20.05 1736 0.181 10.54 57.57 130.9

21.29 1792 0.181 10.54 59.37 131

22.58 1849 0.181 10.54 61.2 131.1

23.96 1908 0.182 10.53 63.07 131.3

25.43 1967 0.182 10.53 64.96 131.4

26.95 2027 0.182 10.53 66.89 131.5

28.63 2089 0.182 10.53 68.85 131.7

30.37 2151 0.183 10.53 70.84 131.8

32.22 2214 0.183 10.52 72.86 131.9

34.2 2279 0.183 10.51 74.92 132

36.24 2344 0.184 10.52 77 132.1

38.5 2410 0.184 10.51 79.12 132.2

40 2455 0.184 10.5 80.55 132.3


93

DCZ

BAZZ

L

Lin +

+=

ACZ

BDZZ

S

Sout +

+=

inS

inSin ZZ

ZVV

+=

SLSL

LSLout DZZCZBAZ

ZVVV

+++==

=

ininin

ZVP

1Re||

2

1 2

=

Loutout

ZVP

1Re||

2

1 2

BAZ

Z

V

V

L

L

in

out

+=

Loop Insertion Loss = SLSL

SL

DZZCZBAZ

ZZ

++++

Mean Squared Loss (MSL) = ∑=

N

i iin

iout

fP

fP

N 1 )(

)(1

Figure B.1 – Loop ABCD parameters, impedance and voltages


94

+–

DC

BAVs

+V1

–

I1 I2

+V2=VL

–ZL

ZS

SourceTwo-port Network

Load

Figure B.2 – Two-port network model.

+V(x)

–

I(x) I(x+dx)

Rdx LdxCdx Gdx

+V(x+dx)

–

Two-port Network

CjGY

LjRZ

ω+=ω+=

Figure B.3 – Incremental section of twisted-pair transmission line.

VS

+–

SSS jXRZ +=

LLL jXRZ +=+

V

–

I

Figure B.4 – Simple load circuit for power analysis.


95

+–

ZS

ZLVS

Line Length dZ0 , γ

+

VL

–

Source

Load

+–

ZS

VS

Line Length d1

Z01 , γ1

Source

ZL

+

VL

–

Load

Line Length d3

Z03 , γ3

Line Length d2

Z02 , γ2

=Φ

101

0SZ

=Φ

101

0SZ

γγ⋅

γ⋅γ=Φ )cosh()sinh(1

)sinh()cosh(

111101

1101111 dd

Z

dZd

γγ⋅


)sinh()cosh(

333303

3301333 dd

Z

dZd

γ⋅=Φ 1)tanh(101

2202

2 dZ

γγ⋅


)sinh()cosh(

0

01 dd

Z

dZd

Figure B.5 – Examples of two-port cascades for twisted-pair transmission line configurations


96

NEXT

Same Binder Group

Transmit

Receive

Figure B.6 – Near end crosstalk (NEXT)

NEXT PO W ER SUM LO SS(dB)1000 FT, 24 AW G PIC

0

10

20

30

40

50

60

70

0.1 1 10 100

FREQUENCY(M Hz)

Pair 1Pair 2Pair 3Pair 4Pair 5Pair 6Pair 7Pair 8Pair 9Pair 10Pair 11Pair 12Pair 13Pair 14Pair 15Pair 16Pair 17Pair 18Pair 19Pair 20Pair 21Pair 22Pair 23Pair 24Pair 251% Case

Figure B.7 – NEXT power sum losses for 25 pairs of PIC cable binder group


97

1% N EXT PO W ER SU M LO SS1000 FT, 24 AW G PIC

0

10

20

30

40

50

60

70

0.1 1 10 100

FR EQ U EN CY(M H z)

.3 - 40M H z fit

A N S I

1.5 - 30M H z fit

Figure B.8 – Comparison of ANSI NEXT with Measured NEXT

FEXT

Same Binder Group

Transmit

Receive

Figure B.9 – Far end crosstalk (FEXT)


98

1% FEXT PO W ER SU M LO SS1000 FT, 24 AW G PIC

0

10

20

30

40

50

60

70

0.1 1 10 100

FR EQ U EN CY(M H z)

.3 - 40M H z fit

AN SI

1.5 - 30M H z fit

Figure B.10 – Comparison of ANSI FEXT with Measured FEXT


99

Annex C(Informative)

Probability of error estimation

As in all digital transmission, the most common measure of DSL performance is the probability of error.Usually, probability of bit error is desired, but sometimes probability of symbol error is also of interest. Ineither case, the engineer attempts to measure the probability of error by observation of a system’sperformance – this is usually achieved with transmission hardware through the use of a BERT. Typically,the bert allows a test engineer to select one of a number of different bit streams (typically various lengthsof pseudorandom patterns). The bert essentially synchronizes to the receiver output bit pattern andcompares it to the input pattern, while counting the number of error positions. The number of errorsaccumulated is periodically divided by the total number of bits measured to estimate the probability of biterror. As time increases, this average bit error rate should converge to the actual system value (when thesystem is not time-varying). A reset button allows bit error counts to be restarted at zero when necessary.

In general, bit error rate measurements become more reliable with time. The designer then needs toknow how long the bit error rate needs to be observed before any derived bit error rate is sufficientlyaccurate. This is a basic statistical problem that involves measurement of a distribution. Let us supposethat bit (or symbol) errors are made with some unknown, but fixed, probability p. One measures p bycounting errors in successive observations of the channel output. Let the kth experiment be denoted by

kp where

−=

)1( measured error no0

)( measured error1

p

ppk .

Then, an estimate of the probability of error, based on N independent measurements, is

∑−

=⋅=

1

0

1)(ˆ

N

kkp

NNp .

This estimate has an average value

[ ] pNpE =)(ˆ

and a variance about this average of

( )[ ] ( )N

pp

N

pNpp ≈−==σ 1ˆvarˆ 2 .

Clearly, this estimate converges to the true probability of error as N gets large. However, N can be muchlarger than sometimes expected. For instance, the standard deviation is the square root of the variance.Thus, for a system where p=10-7 , for the probability of error estimate deviation from to have a standard

deviation of 10% of the value of p , then 97

8 10 or 10

10 ==−

− NN

.

In fact, a single standard deviation may not be sufficient to guarantee good accuracy of measuredprobability of error.

The distribution of the random variable )(ˆ Np has a binomial distribution given by

( ) ( ) ( ) kkNp pp

k

NkNpNkf −−

==⋅= 1ˆPrˆ .


100

The test engineer desires to ensure that the probability of error estimate deviates less than an amount

L

p=ε from the true value with a high degree of confidence. Let us say that we desire ( )δ−1 (equal say

90%) confidence that the measurement deviates less than ε from the true value. Corresponding to thisvalue of ε is a range of values for the index ( )εk such that the estimate is close enough, mathematically

stated precisely as

( ) ( )

( ) kkN

kpp

k

NpNp −∑

ε−

=δ−>ε<− 11ˆPr .

Clearly, just to have a non-trivial set for ( )εk , then pLN /≥ . Evaluation of the sum can be excessively

intensive and so a rough use of the central limit theorem is applied to the distribution to say that for largeN , the distribution is approximately Gaussian and so The probability is then approximated by

( ) δ−=

−=⌡

⌠π

≈ε<−−

−121

2

1ˆPr2

2

L

NpQdxepNp L

px

L

p

L

p,

or then

2

δ=

L

NpQ .

For 90% confidence, 1.=δ that the error is less than p/L%, then the above equation produces

p

LN

28.14 ⋅≥ ,

so, for instance, 10% accuracy at 7e1 −=p , requires that nearly 3 billion bits be tested. Thus, at a speed

of 10 Mbps, this takes about 300 seconds, or approximately 5 minutes. For 1 Mbps transmission, the testwould require 3.5 hours. The measurement time can be reduced most easily by reducing L to 2, whichcorresponds to only about a .2 dB SNR difference. Even then, 1 Mbps DSL transmission at 1e-7 errorrate may take 2 minutes for a measurement, while a lower speed of 100 kbps would take 20 minutes.Such measurement intervals are typical in, for instance, performance comparision tests sponsored bystandards groups like ANSI.

C.1 Effect of input bit sequence

Clearly, the input bit sequence will need to be periodic for any practical implementation of a bert. Theperiod of this sequence should be such that it exceeds the memory of the transmission systemsignificantly. Such sequence length is necessary to ensure that all possible channel output conditions areexcited. Given that DSL transmission systems may have long memory, a 24-bit pseudorandom pattern ismost likely used (with a period of 224-1 bits and running through all 24-bit sequences once and only onceper period. Some sequences with lengths greater than 24 will not have equal likelihood of occurrence andcan bias probability of error measurements, but this effect is usually presumed small by DSL engineers.

C.2 Period of injected “Gaussian” noise

ANSI T1E1.4 studies note that most commercial line simulators make use of pseudorandom noise ingenerating Gaussian noise measuring DSL performance. An unfortunate consequence is that the peaknoise samples generated do not accurately follow the Gaussian distribution tails, thus biasing probability oferror measurements in an optimistic direction. Typically, line simulators generate noise by using someinternal analog noise source and adding digitally generated noise to it. If the period of the latter digital“Gaussian” noise is M, then the peak value of the noise in a set of M samples is also Gaussian with mean


101

σ−=µ

MQ

11

and variance

( )3

22

2

2

21

M

eMpeak

σµ

⋅πσ⋅−=σ .

To eliminate an optimistic bias, the tester would need M>107, which complicates line simulator design.For the more typical value of M=8192, the bias is optimistic by 2.4 dB (see also [17]), meaning that labmeasurements for 8192=M are then optimistic by 2.4 dB and should be reduced by such for fieldperformance.

C.3 6 dB margin and importance sampling

To avoid long measurement times, importance sampling is a method used by test engineers to test onlythe worst-case situations by increasing the occurrence of peak noise samples with respect to Gaussiannoise. Such importance sampling must be very carefully applied for informative results. However, DSLengineers use a form of importance sampling in the concept of margin. Recalling that DSLs are specifiedto have a probability of bit error of 10-7 with a 6 dB margin. This means that the actual probability of errorwould be below 10-24, requiring centuries of measurement time. Instead, testing is executed with noiseincreased by 6 dB so that reasonable measurement times can be used. The margin concept is onemechanism for importance sampling. DSL engineers, however, prefer the supposed practicalinterpretation that unforeseen noise disturbances of a temporarily nature will not cause an error with sucha large margin, although justification for such unforeseen noises at a level of 6 dB is difficult (either thenoise change is much smaller for crosstalk changes or much larger for impulse or temporary RFdisturbances).


102

Annex D(informative)

Additional spectrum management topics currently under studyby the formulating committee of this standard

The formulating committee of this standard has considered several additional topics for which specificrequirements or recommendations could not be finalized in the short time available for the development ofthe first edition of this standard. These topics include, but are not limited to, the following items which theformulating committee feels are important and should be addressed in future editions of this standard:

- Spectrum management guidelines associated with remotely deployed TU-C equipment, such asADSL ATU-C implementations that are collocated with a digital loop carrier remote terminal somedistance from, the Central Office.

- Spectrum management guidelines associated with repeatered DSL applications such as mid-spanISDN or HDSL repeater implementations.

- Revision of non-DSL out-of-band metallic and longitudinal signal power limits to provide anadequate level of protection for DSL systems.

- Addition of VDSL to the guarded list. When VDSL is standardized, it is expected to be added to theguarded list along with the information for the analytical method. When it is standardized, it is to bespectrally compatible.

- Possible extension of the Spectrum Management Class 5 (formerly High Band Asymmetric)upstream band to lower frequencies.

- Methods for optimizing PSDs, maximizing throughput and binder group capacity.

- Trade-offs between loop length restrictions and spectral characteristics.


103

Annex E(informative)

Time varying, user data-dependent crosstalk from T1 and DDS services

Both T1 AMI and 56 kbps DDS are well established and growing services in North America. Thus, it isimportant that the effects of these services on other services are properly considered. Specifications forT1 AMI can be found in ANSI T1.403-1992. Specifications for DDS can be found in ANSI T1.410-1992and AT&T Technical Reference PUB 54075.

Neither T1 nor DDS include scrambling of user data. As a consequence, both the spectral behavior andthe time domain behavior of T1 and DDS line signals are highly dependent on the user data beingtransmitted at any moment. This behavior thus manifests itself in the crosstalk interference of T1 and DDSinto other services. Such behavior strongly contrasts with traditional stationary crosstalk models used toanalyze and test subscriber loop technologies.

T1 and DDS services host many types of user data and protocols. A consequence is that user datacontent and data patterns cannot be predicted nor controlled and it should be assumed that any patterncan be transmitted, that the duration of a pattern is indeterminable and that changes from one pattern toanother can occur at any moment. Examples include bursts of “random” user data followed by idle periodsconsisting of HDLC flags or ONEs.

One consequence of data dependency is that the transmit power spectral density (PSD) and the signalenergy in a given frequency band can vary greatly as user data patterns change. The time duration ofeach PSD variant is caused solely by the time changes of user data content, and thus the time duration ofeach PSD variant may vary from less than a millisecond to many hours. Changes from one PSD to anyother may occur at any moment.

An option for scrambling is defined in T1.410. Currently, however, it is not widely used.

Figure E.1 and Figure E.2 show examples of stationary PSD variants for T1 and DDS. Figure E.3 andFigure E.4 show examples of how the power in frequency bands can vary with time. (It is cautioned thatthese are but examples and are not inclusive of all possible PSD variants. Note also that other DDS datarates exist.)

Several conclusions may be drawn regarding crosstalk from DDS and T1:

1) Crosstalk can exceed that commonly modeled based on a random data assumption for T1 andISDN in certain frequency bands by as much as 20 dB.

2) Crosstalk should be considered to be time varying. The time duration and time change of eachPSD variant is not predictable nor controllable. It is caused by user data content.

3) Crosstalk in a wide band (for example, tens of kilohertz) can change at least 25 dB .

4) Crosstalk in a narrow band (for example 3 kHz) can change at least 45 dB.

5) The above affects all frequency bands from near DC to the highest range of T1.


104

20kH z

120kH z

220kH z

320kH z

420kH z

520kH z

620kH z

720kH z

820kH z

920kH z

1020kH z

All ZER O s

All O NEs

Q RR S “R andom ” D ata1/8 O NEs

T1 Pow er Spectral Density Variations

-25

-45

-35dB

-55

-95 dB m/H z

Figure E.1 – Examples of T1 power spectral density variations

-25 dB m /Hz

-35 dB m /Hz

-45 dB m /Hz

-55 dB m /Hz

-65 dB m /Hz

-75 dB m /Hz

20kH z

220kH z

120kH z

320kH z

420kH z

520kH z

620kH z

720kH z

820kH z

D D S Power Spectral D ensity V ariations

920kH z

2047 Random,all O NE s,

H DL C flag patterns

Figure E.2 – Examples of DDS power spectral density variations


105

Data Dependent Power Change in the 20 - 420 kHz Band

-30

-25

-20

-15

-10

-5

0

5

Time

dBAny

Duration

AnyDuration

Random Data Level

All ONEs Level

Figure E.3 – Data dependent power changes in a wide band due to T1 data patterns

Data Dependent Power Change in a Narrow Band at 193 kHz

-90

-80

-70

-60

-50

-40

-30

Time

dBm/HzRandom Data Level

ZEROs Data Level

ONEs Data Level

Figure E.4 – Data dependent power changes in a narrow band due to T1 data patterns


106

Annex F(informative)

Non-continuous CO signaling events

F.1 Ringing

Ringing in North America is an AC voltage superimposed on a DC bias. Many installations in the US usenon-sinusoidal 20 Hz ringing with a nominal rms. 90 volts at the ringing source. Other frequencies in userange from 16 2/3 to 66 2/3 with voltages from 85 to 13514). One ANSI standard sets the maximumvoltage limit to 150V rms15) and notes cases where it can attain 175V rms.

Ringing is a non-continuous disturber. At the beginning of each ringing burst there is a transition from -48-Volt battery feed to -48-Volt with superimposed AC ringing. Nominal interrupts are 2 seconds on and 4seconds off. Custom ringing cadences with multiple ringing, such as triple cadences, are common. Theringing waveform is ideally a sine wave with its axis of symmetry shifted -48-Volts from zero. The ringingburst can be characterized in terms of 100's of milliseconds as shown in figure F.1. In this depiction, thesine wave starts and stops in unity with the DC bias and represents the best case relative to instantaneouspower changes as a result of ring application and trip.

Elements of synchronization are related to the application of ringing in many applications, such as the useof a common ringing bus serving hundreds of lines. Central office implementations, in many cases,simultaneously ring multiple lines with concurrent cadence. As such, the application and withdraw ofringing is generally without regard to the phase angle of AC energy. The peak voltage when ringing istripped can be the sum of the DC and greatest AC or approximately 170 volts as shown in figure F.2.

In its worst case, a generated ringing waveform is a trapezoidal shape, which means it has higherfrequency components occurring at 25 mS intervals. Transient energies often result from gap switching inthe ringing generator as shown in figure F.3.

Various forms of ringing cadence exist as noted above such as "triple," "double," "long/short," "coded,"and "teen ringing.16) For example, triple ringing bursts three times within 1800 mS as shown in figure F.4.These have the effect of increasing random, ring application and removal impulse effects as shownabove.

Telephone Switching systems typically have the capability of ringing as many as one-fourth of theconnected lines. Accordingly, in the worst case, an average of 6 of the 25 pairs in a binder group could bein some phase of ringing application or removal.

F.2 Supervision (hook flash)

As shown in figure F.5, the DC potential is applied to the customer loop through a battery-feed deviceconsisting of two inductive coils in series with tip and ring. An idle circuit is nominally 48 Volts with nocurrent flowing.

During service initiation, the customer closes the loop and a transient voltage migration occurs within thecable pair of greater than 40 volts, that is, it drops to 6 volts across the telephone set.

A sudden voltage change in the presence of distributed capacitance can couple as not all of it getscancelled out. A wave front of the sudden change in loop voltage is unbounded and currently unrestricted.POTS filters for DSL are only on the pair connected to and adjacent pairs are susceptible to the type ofinductive kick as described above. This exists throughout the network today.

–––––––14) GTE Customer Handbook - 500, Issue 1, 197215) T1.401-1993, "Interface Between Carriers and Customer Installations--Analog Voicegrade Switched Access LinesUsing Loop-Start and Ground-Start Signaling."16) ANSI T1.401.02-1995, "Interface between Carriers and Customer Installations--Analog Voicegrade SwitchedAccess Lines with Distinctive Alerting Features."


107

F.3 Dial Pulse

These are periodic transitions from on-hook to off-hook in order to convey numeric values typically at 10pulses per second in North America. Usually, 40 ms make (close) versus 60 ms break (open) as there isless time required to build the magnetic flux versus lose it. As soon as the dial on the phone is turned, allof the resistance in the circuit (all the handset circuitry) is shunted. There is a solid short in the circuit inorder to get ready to go to maximum current.

The shorter the loop the higher the current but the less the cross talk potential. This is just the opposite oflonger loops. These phenomena exist on short and longer loops.


108

4200 mS

1800 mS

-138 V nominal peak

90 Vac rms.

Figure F.1 – Standard ringing potential with best case start/end

~170V worst case

20Hz or 50 mS Peak to Peak

0

+ 48Vdc

- 48Vdc

- 90Vrms ac 2

Figure F.2 – Standard ringing potential worst case start/end


109

The phase at the transition edge of ringing can be > 500Hz

< .5 mS

25 mS

Time i

Time i+1

Time i+2

1 mS

infinity

Figure F.3 – Ringing waveforms (worst case generalization)

1800 mS

Figure F.4 – Triple ringing interval

48Vdc

Figure F.5 – Simple battery feed arrangement


110

Annex G(informative)

Informative references

[1] B. J. Dunbar, et. al., “Dataport – Channel Units for Digital Data System 56-kb/s Rate”, BSTJ, vol 61 no.9, November 1982.

[2] T. Berger & D. W. Tufts, “Optimum Pulse Amplitude Modulation Part I: Transmitter – Receiver Designand Bounds from Information Theory,” IEEE Transactions on Information Theory, vol. IT-13, no. 2, April1967.

[3] T1.410

[4] GTE Customer Handbook - 500, Issue 1, 1972

[5] Transmission Systems for Communications, Bell Telephone Laboratories, Fifth Edition, 1982.

[6] ASTM D 4566, Standard Test Methods for Electrical Performance of Insulations and Jackets forTelecommunications Wire and Cable, 1994.

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