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ICTI INFORMATION AND COMMUNICATION

TECHNOLOGY INSTITUTE

COM 101 TELECOMMUNICATION SYSTEMS

COURSE NOTES

Prepared by: Angel Diez

KABUL, August 15, 2005

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TABLE OF CONTENTS

1.- INTRODUCTORY CONCEPTS

1.1.- MEASURING UNITS 1.2.- NOISE CONCEPTS 1.3.- TIME AND FREQUENCY DOMAINS 1.4.- RLC RESONANT CIRCUITS

2.- ANALOG MODULATION 2.1.- AMPLITUDE MODULATION 2.2.- FREQUENCY MODULATION

3.- DIGITAL MODULATION 3.1.- ASK, FSK, PSK 3.2.- BPSK, QPSK, QAM

4.- DIGITAL COMMUNICATIONS 4.1.- THE TELEPHONE NETWORK 4.2.- PCM, ADPCM, ISDN 4.3.- WIDE AREA NETWORKS

4.3.1.- PHYSICAL INTERFACES 4.3.2.- LINK PROTOCOLS 4.3.3.- PDH, SDH

4.4.- INTERNET AND HIGHER LEVEL PROTOCOLS 4.5.- VOICE OVER IP

5.- WIRELESS NETWORKS 5.1.- CELLULAR NETWORKS 5.2.- WI-FI, WI-MAX, 3G

6.- TRANSMISSION LINES 6.1.- SMITH CHART 6.2.- IMPEDANCE MATCHING

7.- ANTENNAS AND WAVE PROPAGATION 8.- FIBRE OPTICS 9.- COMMUNICATION SATELLITES

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1.0.-INTRODUCTION A typical telecommunication system is described below.

The frequency spectrum is a valuable natural resource of any country. It is highly important to control its use and allocation.

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1.1.- MEASURING UNITS DECIBEL:

• Describes ration between two values • The basic definition applies to power gain ratio • Addition property

POWER GAIN: For an amplifier of gain G (absolute value), its value in dB is :

12

log*10)( 10 PP

dBG =

Where P2 is output power and P1 is input power Examples:

1.- Gain of a power amplifier = 2 (1 W input = 2 W output) Gain (dB) = 10 * log 2/1 = 3.01 dB 2.- Attenuation in a network = 1000 (1000 W input = 1 W output) Attenuation (dB) = 10 log 1000/1 = 30 dB Also: Gain (dB) = 10 log 1/1000 = - 30 dB

POWER REFERENCE POINTS IN dB UNITS It is common in telecommunication networking practice to define absolute power in dB format as per the following units of measure: dBm is absolute power in reference to 1 mW.

)(1)(

log*10)( 110 mW

mWPdBmP =

P1 in this case is measured in mW. dBW is absolute power in reference to 1W.

)(1)(

log*10)( 110 W

WPdBWP =

P1 in this case is measured in W.

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1.2.- NOISE CONCEPTS 1.2.1.- THERMAL NOISE.- Generated by the movement of electrons within matter. Is also called WHITE NOISE and JOHNSON NOISE. Its power is proportional to the ambient temperature and to the bandwidth of the signal under measure.

kTBPn = watts

nP = Noise power available on output of a resistor.

k= Boltzmann's constant = 2310*38,1 −

J/K (Joules per Kelvin degree) B= Bandwidth in Hz. T = Temperature in Kelvin

Example: For B= 4 MHz and ambient temperature of 290 K (17 °C) we obtain 1410*6,1 −=nP W. = -108 dBm.

It appears in all telecommunication systems. The quality of an amplifier depends much on how to minimize noise.

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1.2.2.- NOISE ACROSS A RESISTOR OF R OHMS. Often it is important to know the noise voltage generated across a resistor. Since the maximum power is delivered when the internal resistance equals the load resistance:

( )

kTBRV

kTBRVR

VkTB

RV

RV

RV

P

n

n

n

n

L

n

L

nL

4

44

442/

2

2

222

=

=

=

===

1.2.3.- SIGNAL TO NOISE RATIO It is the ratio of Signal power over Noise power in one particular point of the system.

kTBRV

VV

RVRV

NS s

n

s

n

s

4//

/2

2

2

2

2

===

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1.2.4.- NOISE FACTOR AND NOISE FIGURE Noise Factor F is the ratio of S/N at the input of the system over S/N at the output. If the system (e.g amplifier) does not introduce any noise F = 1.

( )( ) ( ) ( )sononisi

o

i PPPPNSNS

F /*///

==

Noise Figure is the Noise Factor in dB. FFdB 10log*10=

For an amplifier of gain G: si

so

PP

G = or

ni

no

GPP

F =

If the noise at the input is thermal noise : kTBPni =

Therefore: FGkTBPno = . This is equivalent to having an ideal amplifier (no noise) of Gain = G, with an input noise of :

FkTBP totalni =,

If the amplifier introduces thermal noise, its contribution is:

kTBFkTBFGkTBPna )1( −=−=

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1.2.5.- AMPLIFIER CHAIN.- FRIISS FORMULA Given a chain of amplifiers:

It is useful to consider them as ideal (no noise) and calculate the total noise of the chain at the input. For two amplifiers in series of gains G1 and G2 with noise factors F1 and F2:

Noise at input: kTBPni =

Total noise at input due to first amplifier: = =−+ kTBFkTB )1( 1 kTBF1

Noise at input of second amplifier:= kTBFkTBFG )1( 211 −+

Noise at the output of second amplifier: ))1(( 2112 kTBFkTBFGGPno −+= The noise factor of the chain is:

1

21

21

2112

21

1))1((G

FF

kTBGGkTBFkTBFGG

PGGP

Fni

no −+=

−+==

For a chain of several amplifiers, we obtain the Friiss Formula:

...11

21

3

1

21 +

−+

−+=

GGF

GF

FF

NOTE: G and F are absolute values (no unit of measure). When these values are stated in dB, it is necessary to convert them to its absolute value, in order to apply Friiss formula.

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1.2.6.- NOISE EQUIVALENT TEMPERATURE A popular way of specifying noise by manufacturers is by the Noise Temperature.

The noise associated at the input of an amplifier is: kTBF )1( − . This noise may be represented by a hypothetical temperature Te such that:

kTBFBkTe )1( 1 −=

And the Equivalent Temperature is: TFTe )1( 1 −= The Friiss formula becomes:

...21

3

1

21 +++=

GGT

GT

TT eeee

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1.3.- TIME AND FREQUENCY DOMAINS Electrical signals may be represented as they travel in time (time domain) or as a series of frequency components (frequency domain).

Any periodic signal may be generated as the addition of sine waves of higher frequency, harmonics of the original periodic signal. This was demonstrated by Fourier, using the Fourier series.

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We use oscilloscopes to display signals in the Time domain and Spectrum Analyzers to visualize them in the frequency domain. 1.4.- RLC RESONANT CIRCUITS A series resonant circuit consists of a resistor, an inductor and a capacitor in series. The impedance of the circuit on output at resonance is minimum at the resonant frequency and is given by:

22 )( CL XXRZ −+=

From the expression of the impedance we obtain the resonance frequency:

LCfr π2

1=

The Bandwidth of the resonant circuit is defined as the frequency range delimited by the

impedance being min*2 Z (with Z min = R2). We obtain:

LR

Bπ2

= Also:

r

r

Qf

B =

12

A parallel resonant circuit consists of a resistor, an inductance and a capacitor in parallel.

In this case the bandwidth is obtained when the impedance of the circuit is

max2

1Z = max*707.0 Z .

For this circuit: r

r

Qf

B = Also: R

XQ L

r =

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2.- ANALOG MODULATION 2.1.- AMPLITUDE MODULATION The need for modulation arises because the range of frequencies contained in a baseband signal is not, in general, the same as the range of frequencies which can be transmitted by the communications channel

§ An RF oscillator delivers an output which can be expressed as:

)2cos()( θπ += tfAtf cc

suffix c – carrier wave § Ac – carrier amplitude § fc – carrier frequency (Hz) § θ – phase angle of carrier at t = 0

§ if latter 3 remain constant - no information

§ will consider amplitude or frequency changing A modulating signal m(t) will modify the amplitude of the carrier so that the modulated signal becomes:

)2cos())(1()( θπ ++= tftmAtf ccam

If m(t) is sinusoidal: )2cos()( tfmtm mπ= so that:

)2cos())2cos(1()( θππ ++= tftfmAtf cmcam Where m= Modulation Index The resultant signal, with θ = 0, after filtering with a low-pass filter, becomes:

tffmA

tffmA

tfAtf mcc

mcc

ccam )(2cos2

)(2cos2

)2cos()( ++−+= πππ

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The total power of the modulated signal is:

)2

1(2

22 mAP pc +=

The noise power at the input is:

mifin fNBNN 00 2==

If we call 2

2pA

C = then the S/N ratio at input is:

min fN

mCNS

0

2

2)2/1(

]/[+

=

If the demodulation is done with an envelope detector the S/N ratio on output is:

inout NSmm

NS ]/[22

]/[ 2

2

+=

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Where 2

2

22mm+

=γ is the merit factor. With a maximum value of 2/3 when m=1.

AM GENERATION: A simple AM modulator can be manufactured by placing the sources of carrier and modulating signal in series. The amplified signal is passed on to a resonant circuit tuned at the frequency of the carrier.

AM RECEPTION.- ENVELOPE DETECTOR

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HETERODINING

SUPERHETERODINING

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2.2.- FREQUENCY MODULATION 2.2.1.- FREQUENCY MODULATION: The instant frequency of the carrier is proportional to the value of the baseband signal m(t).

)(tmKff fci +=

fK is the proportionality constant and cf is the

carrier frequency. The frequency deviation

|)(|max tmKDeltaff f==∆

The spectral analysis of FM and PM is difficult.

If we use )2cos()( tftm mπ= it becomes easier.

We define: mff∆

=β as the modulation index.

The bandwidth of the modulated signal (according to the Carson rule) is:

)1(2 += βmif fB

)(2 ffB mif ∆+=

If β is low (<0.5) we obtain Short Band FM and then : mif fB 2≈

If β is high (>> 1) we obtain Large Band FM and then: )2 fBif ∆≈

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On demodulation the S/N ratio at input, calculated for the ifB band is:

ifin BN

CNS

0

]/[ =

)1(2]/[

0 β+=

min fN

CNS

Supposing a carrier power much larger than the noise power, the S/N ratio on output is obtained to be:

inm

out NSfNC

NS ]/)[1(323

]/[ 2

0

2

βββ+==

As measured in the baseband ],0[ mf The S/N at the output increases with the modulation index β and with the signal's bandwidth. The merit factor is in this case:

in

out

NSNS

]/[]/[

)1(3 2 =+= ββγ

Therefore we have advantages if we operate with a bandwidth as large as possible. In commercial FM we have the following band allocation:

FM has better performance against noise than AM since the spurious noise related to amplitude changes become eliminated by a limiter circuit.

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THRESHOLD EFFECT One problem is the fading problem that occurs when the carrier power is not mauch larger than the noise. There is a threshold under which the signal degrades rapidly. This occurs around 10 dB S/N ratio on input.

One way of improving the quality of FM in high frequencies is by introducing a high-pass filter before modulation also called Preemphasis and a low-pass filter after demodulation also called Deemphasis. Where high frequency signals are amplified on transmission and attenuated on reception.

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The filter circuits that create preemphasis and deemphasis are as follows:

FM TRANSMITTER.- A typical FM transmitter can be seen in the following figure.

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FM RECEIVER.- A simple FM receiver is displayed in the following figure.

A standard FM transmitter is shown in blocks in the following figure.

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FM DISCRIMINATOR.-

PHASE DEMODULATOR .- A typical phase demodulator is shown in the following figure.

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PHASE LOCK LOOP.- The phase lock loop is a feed-back circuit that locks the frequency at a fixed value.

SYNTHESIZER.- Using a PLL very precise frequency generators can be constructed. The device is called a synthesizer since the frequencies are generated in a digital fashion.

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2.2.2.- PHASE MODULATION Phase modulation is often employed with digital modulation systems, an example of which is the BPSK. BPSK MODULATION

BPSK IMPLEMENTATION

BPSK DEMODULATOR

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3.- DIGITAL MODULATION.- ASK turns one bit into a different amplitude of the carrier. FSK turns one bit into a different frequency of the carrier. PSK turns one bit into a different phase of the carrier.

• Amplitude Shift Keying

• Frequency Shift Keying

• Phase Shift Keying

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FSK TRANSMITTER.- A typical asynchronous modem of the type Bell 202 employs FSK as modulation scheme. Speeds of 1,200 bps could be obtained (from the 1970s).

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QPSK MODULATION

In QPSK each symbol represents two bits. The symbol is encoded as a fixed amplitude with a particular phase. It is possible to say: PHASE: ∆φ = + 45° = 00 ∆φ = +135 ° = 10 ∆φ = +225° = 11 ∆φ = - 45° = 01

QPSK MODULATOR (I & Q SIGNALS).- Differential QPSK is also called I&Q modulation.

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QAM (QADRATURE AMPLITUDE MODULATION).- If instead of four states (2 bits per state) we manage to use modems that can recognize phase one out of sixteen symbols (or states) at a time, using variations of phase and amplitude, we obtain QAM. Today it is possible to build QAM modems that can recognize up to 256 symbols at the same time (eight bits per symbol)

QPSK ENCODER.- A typical QPSK encoder is shown below.

OTHER TYPES OF DIGITAL MODULATION

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In all cases sampling of the analog signal is required. In the case of the following PAM signal we are sampling at the rate of 11 samples every 3 Hz. If the analog signal is 3000 Hz, the sampling rate is 11,000 samples/sec.

In the following case we use four levels with two bits per sample.

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ANALOG TO DIGITAL CONVERSION.- The process of sampling is carried out with a DAC (Analog to Digital converter)

NYQUIST SAMPLING THEOREM.- Nyquist elaborated a theorem whereby it is possible to recover fully an analog signal as long as the speed of sampling is at least twice the bandwidth of the original analog signal . In the case of an analog voice channel (4,000 Hz bandwidth) the minimum sampling rate required is:

000,82 == Bfs samples/s PCM.- PULSE CODE MODULATION In PCM the voice channel is sampled at the rate of 8,000 samples per second, 8 bits per sample. Therefore a total of 256 levels are available. 128 are positive and 128 are negative. The output data rate of a PCM codec is 64 Kbps. COMPANDING Low voltage signals are often more common in telephony than high voltage signals. In order to improve the quality of the system the CODEC contains a Compander (Compressor and Decompressor) that provides more samples to the low-voltage signals. Companding is actually implemented with a non-linear filter of logarithmic shape. In Europe the scheme is referred as A-law, while in North America is called µ-law

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QUANTIZATION ERROR Digital modulation of analog signals introduce a quantization error as shown in the following figure.

If the maximum amplitude of the signal is +20 volts and there are 128 levels, each level amounts to 125 mV. We shall call the "sample voltage" = s. The average quantization error is:

∫+

−

=−=2/

2/

222

12)(

1 sm

smi

i

i

sdmmm

se

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S/N RATIO OF A PCM SIGNAL

If )(2 tm is the average power of the signal, it is possible to evaluate the ratio

)()(

2

2

tetm

The maximum value is Ms/2, with M= number of quantization levels.

22

2

312/)2/(

MsMs

noisesignal

==

It is often expressed in dB: [S/N]dB = 4,8 + 20 logM = 4,8 + 6m We see that if we increase the number of bits per sample from m to m+1 the S/N ratio improves by 6 dB.

DELTA MODULATION

DELTA MODULATION.- DECODER CIRCUIT

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SPREAD SPECTRUM MODULATION FREQUENCY HOPPING SYSTEM.- The frequency hopping sequence is generated with a pseudo-random generators. The signal looks as if it were noise to other receivers.

DIRECT SEQUENCE .- DSS.- Instead of frequencies, the DSS spread spectrum scheme generates codes of higher data rate that are added to the information stream of data. The scrambling codes are also generated by a pseudo-random generator, with a sequence only known to receiver and transmitter. Only one carrier frequency exists but the bandwidth of the resulting modulated signal is far greater than the band of the information data rate. On reception, the reverse procedure takes place.

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DIGITAL LINE CODES To study only: NRZ-L (Non- Return to Zero L), RZ-Bipolar, RZ-AMI (Alternate Mark Inversion), Differential Manchester

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ERROR RATE PERFORMANCE OF DIGITAL MODULATION SCHEMES. By statistical analysis it is possible to calculate the error probability of the different digital

modulation schemes, which is a function of the ratio 0/ NEb where:

bE = Energy per bit in Watts/bit; 0N = Normalized noise power in Watts/Hz

For instance the Bit Error Probability of coherent FSK is : )(0N

EQP b

b =

where Q(x) is: )2

(21 xerfc ; erfc is the Error Function.

A practical way of describing the relationship between Pb and 0/ NEb is with the following graph:

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The key relationship is as follows:

DNC

NEb

00

= (a1)

where C is the received signal power and D is the data rate in bits per second. EXAMPLE:

A satellite link working at the frequency of 7.3 GHz has 0N

C= 72 dB and uses FSK with

coherent demodulation. The desired bit error rate is 510−=bP . According to the graph, we

should maintain 0/ NEb >12,6 dB. Applying the expression (a1) above, we obtain:

D= 870 Kbps. NOTE: The expression (a1) above is calculated in absolute values. Any dB values have to be converted to absolute values in order to reach the right answer.

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4.- WANS AND DIGITAL NETWORKS

See Power Point Presentation for this chapter.

4.5.- VOICE OVER IP.

See Power Point Presentation for this chapter.

5.- WIRELESS NETWORKS.

See Power Point Presentation for this chapter.

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6.- TRANSMISSION LINES 6.1.- CHARACTERISTIC IMPEDANCE of a line Zo.- Characteristic impedance of a transmission line is the input impedance when the line is infinite. There are no reflections. It is a function of the physical properties of the particular line. The line can be mathematically simulated with a series of distributed inductors L/2 and capacitors C :

We obtain: CL

Z =0

For a coaxial cable of type RG-8 we have: C= 100 nF/m; L = 250 µH/m

Ω== 500 CL

Z

REFLEXION FACTOR Γ .- Is the ratio of the maximum amplitudes of the reflected wave and the incident wave.

0

0

ZZZZ

V

V

L

L

inc

ref

+−

==Γ

Where LZ = Load impedance and 0Z = Characteristic Impedance.

If LZ = 0Z then Γ = 0 (no reflexion) VOLTAGE STATIONARY WAVE RATIO VSWR. Is defined as:

LZZ

VSWR 0

11=

Γ−+Γ

=

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REFLECTED POWER. Power of the reflected wave in watts.

L

reflincrefl R

VPP

22 * =Γ=

THE SMITH CHART.- This is a practical chart that helps to avoid complex calculations when dealing with transmission line impedances. The impedance at each point is calculated as follows:

0

0

)()(

)(ZzZZzZ

zin

in

+−

=Γ

If we convert: θρ je*=Γ we obtain:

θρθρθρθρ

sincos1sincos1

jj

Z in −−++

=

inZ is obtained by the intersection of the circles inX = const. with the circles inR = const. which forms the basis of the Smith chart.

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EXERCICES. 1.- IMPEDANCE VS. ADMITTANCE.

2.- MATCHING TRANSFORMER λ/4 When the load is purely resistive and not equal to the characteristic impedance

0ZRZ LL ≠= , a popular way of matching impedances is by adding, just before the

load, a portion of a line of length λ/4.

In this case LRZZ *001 =

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For the example of the figure: If ZL = 200 ohms and Z0 = 75 ohms

Ω== 5.122200*7501Z

The length of the new segment is λ/4. 3.- USE OF VSWR AND STUBS. Exercice : If VSWR = 2 and Vmin = 0.125 λ, we wish to adjust the load with a "short-circuit stub". This is a short piece of line whose length can be changed and is applied in parallel to the transmission line. In this case no cutting of the original line occurs. All measurements are done with admittances. (not impedances) PROCEDURE:

1.- Trace a circle with centre at 1=Z and radius = VSWR = 2.

2.- Start at the point 2=G (Vminimum).

3.- Find the point intersecting the circle: 1=G

The distance obtained is: λλ 098.0)250.0348.0( =−=d

4.- The admittance at this point 2 is: 7.01 jYin −= Therefore we have to add an

admittance of +j0.7 in order to reach the matching point. This we do with the short-circuit stub. 5.- We calculate the length of the stub by displacement :

The stub is short-circuited. O ohm resistance = ∞ admittance. So we start from ∞ (right side) of the chart. The admittance we want is 0 + j0.7. Therefore we move 0.25λ (half a circle) then we add + j0.7 . From this point we trace a straight line passing by the centre of the chart and we measure the distance.

λλ 347.0)097.025.0(1 =+=d Note: The point of minimum voltage is always situated left of the central point Z=1. A point of maximum voltage will always be found right of the central point. For admittances is the reverse.

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EXERCICE 3.- USE OF STUBS

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7.- AIR WAVE PROPAGATION AND ANTENNAS 7.1.- PROPAGATION EQUATION

For a signal of average power tP radiated isotropically (equally in all directions),

the power density tP of the wave at the distance d is :

24 dP

P ti π=

If we use an antenna of maximum gain tG , the radiated power is:

24 dGP

GPP ttti π== W/ 2m

For a receiving antenna, with an effective area effA . The power supplied by the

antenna to the load is:

efftt

eff AdGP

PAP 24π==

For all antennas we have: πλ4

2

=G

Aeff

If the maximum gain of the receiving antenna is rG we have:

2)4(

dGG

PP

rtt

r

πλ

=

Which is the fundamental propagation equation in vacuum. If the frequency f is expressed in MHz and the distance in Km, we obtain:

44

2

3

)(10*57.0

dfGG

PP

rtt

r−

=

Converting in dB we obtain:

)log20log205.32()()()( fdGGPP

dBrdBtdBt

r ++−+=

Always with f in MHz and d in Km. This last term, named L, is called the propagation loss.

fdL log20log205.32 ++= We then have:

dBdBrdBtdBt

r LGGPP

)()()()( −+=

7.2.- CHARACTERISTICS OF RADIO PROPAGATION REFRACTION.- When an electromagnetic wave passes from one medium to another the wave changes direction according to Snell's law:

2211 sinsin θθ nn = where n1 and n2 are the refraction indexes of the two media.

1θ

2θ

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DIFFRACTION.- When an electromagnetic wave passes near a sharp obstacle, the wave does not continue in straight line, but rather it curbs. This effect allows for reception beyond the straight line of vision. The effect is stronger for HF spectrum than for higher frequencies. Reason why land mobile communications in mountainous terrain is preferable in the HF band. INTERFERENCE.- Two waves leaving the same antenna may reach a point where both are in reverse phase. This may occur when one of the waves has been reflected by an obstacle (the ground for instance). In that case the received signal may be completely lost (opposite phase) . MAXIMUM DISTANCE BETWEEN ANTENNAS.- Considering diffraction of the earth curvature, a standard extended earth radius is 4/3 of the real radius. The maximum line of

"diffracted vision" of an antenna of height th is: tt hd 4= where td is the

maximum distance in Km that the antenna can reach and th is the height of the antenna in meters. Therefore the maximum distance between two antennas is:

rt hhd 44 +=

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7.3.- MICROWAVE and OTHER RADIO LINKS TERRAIN TOPOGRAPHY.- When selecting an antenna site for a base station or microwave tower the following aspects have to be considered:

• Altitude over sea level • Vegetation • Reflective zones • Obstructions

Refraction of the earth surface is taken into consideration. We use the concept of effective earth radius which is 4/3 time the real radius. Reflections: The receiving antenna may receive multiple reflections of the main beam, causing lobes of reception, where at certain antenna heights the signal fades substantially. It is important to adjust the height of the microwave antenna in order to avoid the nulls of the reflective lobes. Attenuation due to rain, fog, snow.- Attenuation due to rain depends on the shape of the raindrops. For frequencies under 8 GHz it is insignificant. Above 8 GHz it should be considered. One way of fighting attenuation is to increase the fading marging. Link Reliability.- It is measured in %. It is defined as the number of outage hours per year. Often we use tables applicable to Rayleigh fading as follows:

REQUIRED RELIABILITY % FADING MARGIN dB 90 % 8 dB 99 % 18 dB

99.9 % 28 dB 99.99 % 38 dB

99.999 % 48 dB

For intermediary values interpolation can safely be used. Diversity.- In order to increase the reliability of the link, two types of diversity are used:

• Space Diversity.- One transmitter is used but two receivers (and two antennas) are used. Both signals are combined after reception to produce the one with best quality. Useful in mobile radio systems when the terrain is mountainous and prone to cause reflections.

• Frequency Diversity.- The same signal is transmitted with two separate radios and antennas. The signal is received with two separate receivers and antennas. It is more effective than Space Diversity but far more expensive. Used mainly in the military.

47

Interference.- Every antenna that functions as a relay station must regenerate the signal with a new set of frequencies in order to avoid interference with the adjacent link. A multitude of interference signals exist outside the ones used by the operator. It is imperative to employ a strict control of the radio spectrum in the country in order to avoid interference. Polarisation.- An electromagnetic signal may be polarised depending of the position of the electric field E. If it is horizontal polarisation the E field is horizontal to the earth surface. If vertically polarized, the E filed is vertical to the earth surface. This is used in microwave signals to transmit two beams of the same carrier which are independent from each other (orthogonal)

48

7.4.- ANTENNAS

Radiation Intensity: where P is the power of the transmitter fed to the antenna. A typical parallel-plate capacitor is shown in figure (a), where it is connected to a D.C. voltage supply (represented by an E inside of a circle). A voltage applied to two plates produces an electric field between the plates The electric field lines are shown by arrows pointing from the positively charged plate (+) to the negatively charged plate (–). Figures (b) and (c) show the change in electric field lines due to a change in the orientation of the capacitor’s plates.

π40

PU =

An Isotropic Radiator i is “a radiator which radiates the same amount of power in all directions.” IItt iiss aa ppuurreellyy hhyyppootthheettiiccaall rraaddiiaattoorr!!!!!!!!

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As an extension of the capacitor's behaviour we can see how a dipole antenna operates. The λ/2 (half wave) dipole antenna is very similar to the open capacitor. Shown in Figure a, the λ/2 dipole has two conductive plates. Changing the polarity (alternating current) we note the change in the direction of the electric field lines. As this sequence of alternating voltages continues, an alternating electric field loop “detaches” from the antenna, and radiates outward.

half wave

one wave length

1.5 wave length

very short dipole

Radiation Pattern is defined as “ the variation of the magnitude of the electric or magnetic field as a function of direction (at a distance far from the antenna).”

50

Beamwidth is the angle defined by the 3 dB points of power loss of the strongest lobe. For the dipole antenna of the figure BW= 60º

Since the Radiation Intensity U is independent of the distance of observations but only depends upon antenna inherent parameters, it can be taken to describe the Radiation Pattern of an Antenna

The Directive Gain g is “ the ratio of the radiation intensity U of the antenna to that of an isotropic radiator U0 radiating the same amount of power.” It is a function of direction!!!

PU

UU

g π40

==

Hertzian Dipole

( ) θθ 2sin5.1 ⋅=g

0.1 0.2 0.3 0.5 1

30

210

60

240

90270

120

300

150

330

180

0

3 dB

51

Where e is radiation efficiency and D is the maximum directional gain.

For all antennas: πλ4

2

=M

e

GA

where Gm is the maximum gain of the antenna and λ is the wavelength.

The Effective Area Ae characterises the antenna’s ability to absorb the incident power density w and to deliver it to the load. (Receiving Antenna!)

inputPU

UU

G π4input ,0

==inputP

UG max

max 4π=

DG

gG

e max==

The Power Gain G is “ the ratio of the radiation intensity U of the antenna to that of an isotropic radiator U0 radiating an amount of power equal to the power accepted by the antenna.”

Lossinput PPP

PP

e+

==

The Radiation Efficiency e is the ratio of the radiated power P to the total power Pinput accepted by the antenna. Pinput = P + PLoss

wP

A loade =

52

ANTENNA COUPLING.- The Antenna's impedance has to be matched to the output impedance of the transmitter. (or input impedance of the receiver) in order to avoid reflections and maximize radiated power. Stubs and matching transformers are used to that purpose. YAGI-UDDA ANTENNA.- Antenna of low cost and medium gain. It s composed of the following elements:

• Active element.- Power is fed here. Usually a 2λ

dipole.

• Reflectors.- Longer than the active element, separated form the active element by

a distance of 4λ

. It acts as a mirror concentrating the electromagnetic waves in

the opposite direction. Its length is approximately 0.55 λ.

• Director Shorter than the active element, separated form the active element by a

distance of 4λ

. It acts as a magnifying lens concentrating the power in the desired

direction. Its length is approximately 0.45 λ.

53

PARABOLIC ANTENNA.- Used in UHF and higher bands. The physical area of the antenna with diameter D is:

4

2Dπ

From the expression of Effective area, we have:

2

)*

(*)(λ

πθ

DIG = where I is the illumination factor or efficiency factor.

If we use an efficiency factor of 0.6 (typical) , we obtain:

2

)(6λD

G =

The Beamwidth of a parabolic antenna, given in degrees, is:

DBW dB

λ703 =−

The width of the beam between two consecutive zeroes of the pattern for the main beam is exactly double. (see Figure of Antenna Pattern)

DBWzeroes

λ140=

8.- FIBRE OPTICS SYSTEMS

(SEE POWER POINT PRESENTATION)

54

9.- SATELLITE COMMUNICATIONS.- 9.1 ATMOSPHERIC ABSORPTION The figure below indicates the average atmospheric absorption as a function of frequency at different altitudes above sea-level and the effects of rain and fog. Note that the figures cover different frequency ranges. AVERAGE ATMOSPHERIC ABSORPTION

55

ATMOSPHERIC ABSORPTION OF ATMOSPHERE DUE TO FOG AND RAIN The figure to the right indicates that at frequencies below 15 GHz attenuation due to rain is still far lower than at frequencies in the optical range, GEOSTATIONARY SATELLITES.-

• occupy fixed position with respect to earth above the equator - no tracking required • 3 satellites provide coverage for most of earth's surface (not polar regions)

LINK BUDGET

56

We use the following Propagation Equation:

dBdBRdBTdBWTdBWR LR

GGPP −−++= )4

log(20λπ

Where L, due to rain is typically 5 dB. EXERCICE.- Calculate the power that must be transmitted from a geostationary satellite to

give a power of -116dBW (2110*5.2 −

W) at a receiver on the earth. Assume f=10 GHz, Gr = 40 dB = Gt = 30 dB and additional losses of 5 dB. R = altitude = 36,0000 Km.

SOLUTION.- We obtain: 52034030116 −−++=− TP dBW

Therefore: TP = 22 dBW = 159 W. EIRP = 22 dBW + 30 dB = 52 dBW