Application Note 32

AN32-1

an32f

March 1989

High Effi ciency Linear RegulatorsJim Williams

Figure 1. Typical Switching Supply Arrangement Showing Linear Post-Regulators

Introduction

Linear voltage regulators continue to enjoy widespread use despite the increasing popularity of switching approaches. Linear regulators are easily implemented, and have much better noise and drift characteristics than switchers. Ad-ditionally, they do not radiate RF, function with standard magnetics, are easily frequency compensated, and have fast response. Their largest disadvantage is ineffi ciency. Excess energy is dissipated as heat. This elegantly sim-plistic regulation mechanism pays dearly in terms of lost power. Because of this, linear regulators are associated with excessive dissipation, ineffi ciency, high operating temperatures and large heat sinks. While linears cannot compete with switchers in these areas they can achieve signifi cantly better results than generally supposed. New components and some design techniques permit reten-tion of linear regulator’s advantages while improving effi ciency.

One way towards improved effi ciency is to minimize the input-to-output voltage across the regulator. The smaller this term is, the lower the power loss. The minimum input/output voltage required to support regulation is referred to as the “dropout voltage.” Various design techniques and technologies offer different performance capabilities. Appendix A, “Achieving Low Dropout,” compares some approaches. Conventional three terminal linear regulators have a 3V dropout, while newer devices feature 1.5V drop-out (see Appendix B, “A Low Dropout Regulator Family”) at 7.5A, decreasing to 0.05V at 100μA.

Regulation from Stable Inputs

Lower dropout voltage results in signifi cant power sav-ings where input voltage is relatively constant. This is normally the case where a linear regulator post-regulates a

switching supply output. Figure 1 shows such an arrange-ment. The main output (“A”) is stabilized by feedback to the switching regulator. Usually, this output supplies most of the power taken from the circuit. Because of this, the amount of energy in the transformer is relatively unaf-fected by power demands at the “B” and “C” outputs. This results in relatively constant “B” and “C” regulator input voltages. Judicious design allows the regulators to run at or near their dropout voltage, regardless of loading or switcher input voltage. Low dropout regulators thus save considerable power and dissipation.

VREG “B”

V

VREG

+VIN

“C”

V

SWITCHINGREGULATOR

“A”

AN32 F01

L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners.

Application Note 32

AN32-2

an32f

Regulation from Unstable Input—AC Line Derived Case

Unfortunately, not all applications furnish a stable input voltage. One of the most common and important situations is also one of the most diffi cult. Figure 2 diagrams a classic situation where the linear regulator is driven from the AC line via a step-down transformer. A 90VAC (brownout) to 140VAC (high line) line swing causes the regulator to see a proportionate input voltage change. Figure 3 details ef-fi ciency under these conditions for standard (LM317) and low dropout (LT®1086) type devices. The LT1086’s lower dropout improves effi ciency. This is particularly evident at 5V output, where dropout is a signifi cant percentage of

the output voltage. The 15V output comparison still favors the low dropout regulator, although effi ciency benefi t is somewhat reduced. Figure 4 derives resultant regulator power dissipation from Figure 3’s data. These plots show that the LT1086 requires less heat sink area to maintain the same die temperature as the LM317.

Both curves show the deleterious effects of poorly con-trolled input voltages. The low dropout device clearly cuts losses, but input voltage variation degrades obtainable effi ciency.

+

LINEARVOLTAGE

REGULATOR

VIN—MOVES WITHAC LINE VOLTAGE

90 TO 140VAC LINE

VOUT

AN32 F02

AC LINE VOLTAGE80

POW

ER D

ISSI

PATI

ON IN

WAT

TS

9

12

151413

1011

78

45

12

110 130 140

AN32 F04

6

3

090 100 120

LM317 VO = 15V, 1A

LT1086 VO = 15V, 1A

LT1086 VO = 5V, 1A

LM317 VO = 5V, 1A

AC LINE VOLTAGE80

EFFI

CIEN

CY (%

)

40

100

110

120

100 120 130

AN32 F03

20

10

80

60

30

90

0

70

50

90 110 140 150

VIN = 16.5V

LT1086

LM317

VIN = 25.6VVIN = 28V

VIN = 12.4VVIN = 10.1V

VIN = 6.5VVIN = 18V

VIN = 8V

DATA DOES NOT INCLUDE RECTIFIERDIODE LOSSES. SCR PRE-REGULATORINCLUDES SCR LOSSES

VOUT = 15V, 1A

LT1086VOUT = 5V, 1A

LT1086 WITHSCR PRE-REGULATORVOUT = 15V, 1A

LM317 DROPOUT = 3VLT1086 DROPOUT = 1.5V

LM317

LT1086 WITHSCR PRE-REGULATOR

VOUT = 5V, 1A

Figure 4. Power Dissipation for Different Regulators vs AC Line Voltage. Rectifi er Diode Losses are not Included

Figure 2. Typical AC Line Driven Linear Regulator

Figure 3. Effi ciency vs AC Line Voltage for LT1086 and LM317 Regulators

Application Note 32

AN32-3

an32f

SCR Pre-Regulator

Figure 5 shows a way to eliminate regulator input variations, even with wide AC line swings. This circuit, combined with a low dropout regulator, provides high effi ciency while retaining all the linear regulators desirable characteristics. This design servo controls the fi ring point of the SCRs to stabilize the LT1086 input voltage. A1 compares a portion of the LT1086’s input voltage to the LT1004 reference. The amplifi ed difference voltage is applied to C1B’s negative input. C1B compares this to a line synchronous ramp (Trace B, Figure 6) derived by C1A from the transformers rectifi ed secondary (Trace A is the “sync” point in the fi gure). C1B’s pulse output (Trace C) fi res the appropriate

SCR and a path from the main transformer to L1 (Trace D) occurs. The resultant current fl ow (Trace E), limited by L1, charges the 4700μF capacitor. When the transformer out-put drops low enough the SCR commutates and charging ceases. On the next half-cycle the process repeats, except that the alternate SCR does the work (Traces F and G are the individual SCR currents). The loop phase modulates the SCR’s fi ring point to maintain a constant LT1086 input voltage. A1’s 1μF capacitor compensates the loop and its output 10kΩ-diode network ensures start-up. The three terminal regulator’s current limit protects the circuit from overloads.

+

–

+

–

+

20VACAT 115VIN

L2

20VACAT 115VIN

1N4148

L1500μH

1N4148

1N4148

1k

1N40021N4002

1N4002 10k

158k*(TRIM)

121Ω*

1.37k*

12k*

200k

2.4k

10k

0.1μF

+V

+V

+V

0.001

1μF

10k 10k+V

AN32 F05

LT10041.2V

750Ω

L1 = PULSE ENGINEERING, INC. #PE-50503L2 = TRIAD F-271U (TYPICAL)* = 1% FILM RESISTOR

22μF

+4700μF

≈16.5V TO 17V

+20μF

15VOUT1A

“SYNC”

90VAC TO140VAC

≈30V TO ALL“+V” POINTS

C1A1/2 LT1018

C1B1/2 LT1018

–

+A1

LT1006

LT1086

= G.E. C-106B

Figure 5. SCR Pre-Regulator

Application Note 32

AN32-4

an32f

This circuit has a dramatic impact on LT1086 effi ciency versus AC line swing*. Referring back to Figure 3, the data shows good effi ciency with no change for 90VAC to 140VAC input variations. This circuit’s slow switching preserves the linear regulators low noise. Figure 7 shows slight 120Hz residue with no wideband components.*The transformer used in a pre-regulator can signifi cantly infl uence overall effi ciency. One way to evaluate power consumption is to measure the actual power taken from the 115VAC line. See Appendix C, “Measuring Power Consumption.”

DC Input Pre-Regulator

Figure 8a’s circuit is useful where the input is DC, such as an unregulated (or regulated) supply or a battery. This circuit is designed for low losses at high currents. The LT1083 functions in conventional fashion, supplying a regulated output at 7.5A capacity. The remaining com-ponents form a switched-mode dissipation regulator. This regulator maintains the LT1083 input just above the dropout voltage under all conditions. When the LT1083 input (Trace A, Figure 9) decays far enough, C1A goes high,

Figure 8a. Pre-Regulated Low Dropout Regulator

+

–

+

C1B1/2 LT1018

–

+

C1A1/2 LT1018

12V

1N4148

1N4148 ≈30V

1N4148

L1820μH

12VIN(7V TO 20V)

10μF

+22μF

+47μF

+1000μF

+10μF

5VOUT7.5A

≈6.75VD S

MBR1060

1N96616V

L2335μH

Q1P50N05E

330Ω

0.01

10k

100Ω

100k D

S

Q2VN2222

270k

1M

20k 12k12V

12k*

AN32 F08a

56k*(TRIM)

LT10041.2V47pF

5.1k

10k

2kL1 = PULSE ENGINEERING, INC. #PE-52630L2 = PULSE ENGINEERING, INC. #PE-51518P50N05E = MOTOROLA

12V

LT1083-5

Figure 6. Waveforms for the SCR Pre-Regulator Figure 7. Output Noise for the SCR Pre-Regulated Circuit

A = 50V/DIV

B = 10V/DIV

C = 50V/DIV

D = 20V/DIV

E = 10A/DIV

F = 10A/DIV

G = 10A/DIV

HORIZ = 5ms/DIV AN32 F06

A = 5mV/DIV(AC-COUPLED)

HORIZ = 5ms/DIV AN32 F07

Application Note 32

AN32-5

an32f

allowing Q1’s gate (Trace B) to rise. This turns on Q1, and its source (Trace C) drives current (Trace D) into L2 and the 1000μF capacitor, raising regulator input voltage. When the regulator input rises far enough C1A returns low, Q1 cuts off and capacitor charging ceases. The MBR1060 damps L2’s fl yback spike and the 1M-47pF combination sets loop hysteresis at about 100mV.

Q1, an N-channel MOSFET, has only 0.028Ω of satura-tion loss but requires 10V of gate-source turn-on bias. C1B, set up as a simple fl yback voltage booster, provides about 30V DC boost to Q2. Q2, serving as a high voltage pull-up for C1A, provides voltage overdrive to Q1’s gate. This ensures Q1 saturation, despite its source follower connection. The Zener diode clamps excessive gate-source

overdrives. These measures are required because alterna-tives are unattractive. Low loss P-channel devices are not currently available, and bipolar approaches require large drive currents or have poor saturation. As before, the linear regulator’s current limit protects against overloads. Figure 10 plots effi ciency for the pre-regulated LT1083 over a range of currents. Results are favorable, and the linear regulator’s noise and response advantages are retained.

Figure 8b shows an alternate feedback connection which maintains a fi xed small voltage across the LT1083 in ap-plications where variable output is desired. This scheme maintains effi ciency as the LT1083’s output voltage is varied.

470Ω

1N4148

120

1k

+10μF

VOUTLT1083

1N4148

AN32 F08b

4N28

10k

20k

1M

TO REMAININGCIRCUITRY

47pF

12k12V

12V

FROML2

VREF ≈ 1.8V

–

+

1/2 LT1018

Figure 8b. Differential Sensing for the Pre-Regulator Allows Variable Outputs

Figure 9. Pre-Regulator Waveforms Figure 10.Effi ciency vs Output for Figure 8a

OUTPUT CURRENT (A)0

0

EFFI

CIEN

CY (%

)

10

30

40

50

100

• • • •70

2 4 5

AN32 F10

20

80

90

60

1 3 6 7 8

VIN = 12V

A = 100mV/DIV(AC-COUPLED)

B = 20V/DIV

C = 20V/DIV

D = 4A/DIV

HORIZ = 100μs/DIV AN32 F09

Application Note 32

AN32-6

an32f

10A Regulator with 400mV Dropout

In some circumstances an extremely low dropout regulator may be required. Figure 11 is substantially more complex than a three terminal regulator, but offers 400mV dropout at 10A output. This design borrows Figure 8A’s overdriven source follower technique to obtain extremely low satura-tion resistance. The gate boost voltage is generated by the LT1072 switching regulator, set up as a fl yback converter.* This confi guration’s 30V output powers A1, a dual op amp. A1A compares the regulators output to the LT1004 reference and servo controls Q1’s gate to close the loop. The gate voltage overdrive allows Q1 to attain its 0.028Ω saturation, permitting the extremely low dropout noted. The Zener diode clamps excessive gate-source voltage and the 0.001μF capacitor stabilizes the loop. A1B, sensing current

across the 0.01Ω shunt, provides current limiting by forcing A1A to swing negatively. The low resistance shunt limits loss to only 100mV at 10A output. Figure 12 plots current limit performance for the regulator. Roll-off is smooth, with no oscillation or undesirable characteristics.

+

–

+A1B

1/2 LT1013

LT1072

VIN

VIN L1150μH

VCGND

VSW

FB

100k*

D S

Q1P50N05E

1N96616V

1N4148

0.001

LT10041.2V

L1 = PULSE ENGINEERING, INC. #PE-52645P50N05E = MOTOROLA* = 1% FILM RESISTOR** = DALE RH-10

0.01Ω**

10k*

180Ω

10k*

VIN

100k*

28k

1N4148

1.2k

1k

–

+

A1A1/2 LT1013 10k

VIN

12k*

AN32 F11

38k*

22μF

+4.7μF

+1μF

+47μF

≈30V

5V10A

Figure 11. 10A Regulator with 400mV Dropout

OUTPUT CURRENT (A)8

OUTP

UT V

OLTA

GE (V

)

4

5

6

• • • • •

•

•11 13

AN32 F12

3

2

9 10 12 14 15

1

0

MAXIMUM RATEDOUTPUT CURRENTFOR 5VOUT

Figure 12. Current Limit Characteristics for the Discrete Regulator

*If boost voltage is already present in the system, signifi cant circuit simplifi cation is possible. See LTC Design Note 32, “A Simple Ultra-Low Dropout Regulator.”

Application Note 32

AN32-7

an32f

+

– +1/2

LT10

18

12V

10k

270k

L1 820μ

H5.

1k

10μF

+22

μF

+47

μF+

22μF

+10

00μF

≈5.4

5VP5

0N05

E 1N96

7A18

V1N

967A

18V

L233

5μH

MBR

1060

330Ω

100k

VN22

22

1N41

48

12V I

N(5

.9V

TO 2

0V)

1N41

48

1N41

48≈3

0V

10k

2k

L1 =

PUL

SE E

NGIN

EERI

NG, I

NC. #

PE-5

2630

L2 =

PUL

SE E

NGIN

EERI

NG, I

NC. #

PE-5

1518

* =

1% F

ILM

RES

ISTO

R0.

01Ω

= D

ALE

RH-1

0

12V

0.01

P50N

05E

5VOU

T10

A

– +

1/2

LT10

1820

k12

k

0.01

Ω

1M47

pF

0.00

1

12V

LT10

041.

2V

12k*

100k

*

42.5

k*(T

RIM

)10

0Ω

10k*

10k*

180Ω

100k

*

1N41

48

– +1/2

LT10

13

– +

1/2

LT10

13

AN32

F13

38k*

12k*

Figu

re 1

3. U

ltral

ow D

ropo

ut L

inea

r Reg

ulat

or w

ith P

re-R

egul

ator

Application Note 32

AN32-8

an32f

Ultrahigh Effi ciency Linear Regulator

Figure 13 combines the preceding discrete circuits to achieve highly effi cient linear regulation at high power. This circuit combines Figure 8a’s pre-regulator with Figure 11’s discrete low dropout design. Modifi cations include deletion of the linear regulators boost supply and slight adjustment of the gate-source Zener diode values. Similarly, a single 1.2V reference serves both pre-regula-tor and linear output regulator. The upward adjustment in the Zener clamp values ensures adequate boost voltage under low voltage input conditions. The pre-regulator’s feedback resistors set the linear regulators input voltage just above its 400mV dropout.

This circuit is complex, but performance is impressive. Figure 14 shows effi ciency of 86% at 1A output, decreasing to 76% at full load. The losses are approximately evenly distributed between the MOSFETs and the MBR1060 catch diode. Replacing the catch diode with a synchronously switched FET (see Linear Technology AN29, Figure 32) and trimming the linear regulator input to the lowest possible value could improve effi ciency by 3% to 5%.

Micropower Pre-Regulated Linear Regulator

Power linear regulators are not the only types which can benefi t from the above techniques. Figure 15’s pre-regulated micropower linear regulator provides excellent effi ciency and low noise. The pre-regulator is similar to Figure 8a. A drop at the pre-regulator’s output (Pin 3 of the LT1020 regulator, Trace A, Figure 16) causes the LT1020’s comparator to go high. The 74C04 inverter chain switches,

OUTPUT CURRENT (A)0

EFFI

CIEN

CY (%

)

60

80

100

• • • • • •

8AN32 F14

40

20

50

70

90

30

10

021 43 6 7 95 10

VIN = 12V

Figure 14. Effi ciency vs Output Current for Figure 13

+VIN VOUT

2.5VREF GND

LT1020

FB

4 9 11

2

1M*1M*

909k*1M

200kOUTPUT TRIM

825k*

270pF

220k 

* = 1% METAL FILM RESISTORGROUND UNUSED 74C04 INPUTS

8

7

674C04

200kPRE-REGTRIM

0.001μF680pF 10μF+

220μF1N5817

IRFD9120

HP5082-2810

AN32 F15

100mHDALE TE-5Q4-TA

DS

+V6V TO 10V

5VOUT35.2V

–

+

LT1020COMP

Figure 15. Micropower Pre-Regulated Linear Regulator

Application Note 32

AN32-9

an32f

biasing the P-channel MOSFET switch’s grid (Trace B). The MOSFET comes on (Trace C), delivering current to the inductor (Trace D). When the voltage at the inductor-220μF junction goes high enough (Trace A), the compara-tor switches high, turning off MOSFET current fl ow. This loop regulates the LT1020’s input pin at a value set by the resistor divider in the comparator’s negative input and the LT1020’s 2.5V reference. The 680pF capacitor stabilizes the loop and the 1N5817 is the catch diode. The 270pF capacitor aids comparator switching and the 2810 diode prevents negative overdrives.

The low dropout LT1020 linear regulator smooths the switched output. Output voltage is set with the feedback pin associated divider. A potential problem with this circuit is start-up. The pre-regulator supplies the LT1020’s input but relies on the LT1020’s internal comparator to function. Because of this, the circuit needs a start-up mechanism. The 74C04 inverters serve this function. When power is applied, the LT1020 sees no input, but the inverters do. The 200k path lifts the fi rst inverter high, causing the chain to switch, biasing the MOSFET and starting the circuit. The inverter’s rail-to-rail swing also provides good MOSFET grid drive.

The circuit’s low 40μA quiescent current is due to the low LT1020 drain and the MOS elements. Figure 17 plots ef-fi ciency versus output current for two LT1020 input-output differential voltages. Effi ciency exceeding 80% is possible, with outputs to 50mA available.

References

1. Lambda Electronics, Model LK-343A-FM Manual

2. Grafham, D.R., “Using Low Current SCRs,” General Electric AN200.19. Jan. 1967

3. Williams, J., “Performance Enhancement Techniques for Three-Terminal Regulators,” Linear Technology Corporation. AN2

4. Williams, J., “Micropower Circuits for Signal Condition-ing,” Linear Technology Corporation. AN23

5. Williams, J. and Huffman, B., “Some Thoughts on DC-DC Converters,” Linear Technology Corporation. AN29

6. Analog Devices, Inc, “Multiplier Application Guide”

OUTPUT CURRENT (mA)0

EFFI

CIEN

CY (%

)

60

80

100

40

AN32 F17

40

20

50

70

90

30

10

0105 2015 30 35 4525 50

VDIFF LT1020 = 0.2VOUT OF

REGULATION

VDIFF LT1020 = 0.5V

Figure 17. Effi ciency vs Output Current for Figure 15Figure 16. Figure 15’s Waveforms

Note: This application note was derived from a manuscript originally prepared for publication in EDN magazine.

A = 50mV/DIV(AC-COUPLED)

B = 10V/DIV

C = 10V/DIV

D = 100mA/DIV

HORIZ = 500μs/DIV AN32 F16

Application Note 32

AN32-10

an32f

APPENDIX A

Achieving Low Dropout

Linear regulators almost always use Figure A1a’s basic regulating loop. Dropout limitations are set by the pass elements on-impedance limits. The ideal pass element has zero impedance capability between input and output and consumes no drive energy.

PASSELEMENT REGULATING

LOOP

–

+

VREF

AN32 FA1a

DRIVE

INPUT OUTPUT

Figure A1a. Basic Regulating Loop

VIN VOUT

VIN

VIN

VOUT

VOUT

Followers CompoundCommon

Emitter/Source

–V

VOUT

AN32 F1b

VIN VOUT

VIN VOUT VIN

VIN

VOUT

Figure A1b. Linear Regulator with Some Pass Element Candidates

A number of design and technology options offer various trade-offs and advantages. Figure A1b lists some pass element candidates. Followers offer current gain, ease of loop compensation (voltage gain is below unity) and the drive current ends up going to the load. Unfortunately, saturating a follower requires voltage overdriving the input (e.g., base, gate). Since drive is usually derived directly from VIN this is diffi cult. Practical circuits must either generate the overdrive or obtain it elsewhere. This is not easily done in IC power regulators, but is realiz-able in discrete circuits (e.g., Figure 11). Without voltage overdrive the saturation loss is set by Vbe in the bipolar case and channel on-resistance for MOS. MOS channel

on-resistance varies considerably under these conditions, although bipolar losses are more predictable. Note that voltage losses in driver stages (Darlington, etc.) add di-rectly to the dropout voltage. The follower output used in conventional three terminal IC regulators combines with drive stage losses to set dropout at 3V.

Common emitter/source is another pass element option. This confi guration removes the Vbe loss in the bipolar case. The PNP version is easily fully saturated, even in IC form. The trade-off is that the base current never arrives at the load, wasting substantial power. At higher cur-rents, base drive losses can negate a common emitter’s saturation advantage. This is particularly the case in IC designs, where high beta, high current PNP transistors are not practical. As in the follower example, Darlington connections exacerbate the problem. At moderate currents PNP common emitters are practical for IC construction. The LT1020/LT1120 uses this approach.

Common source connected P-channel MOSFETs are also candidates. They do not suffer the drive losses of bipolars, but typically require 10V of gate-channel bias to fully saturate. In low voltage applications this usually requires generation of negative potentials. Additionally, P-channel devices have poorer saturation than equivalent size N-channel devices.

The voltage gain of common emitter and source confi gura-tions is a loop stability concern, but is manageable.

Compound connections using a PNP driven NPN are a reasonable compromise, particularly for high power (be-yond 250mA) IC construction. The trade-off between the PNP Vce saturation term and reduced drive losses over a straight PNP is favorable. Also, the major current fl ow is through a power NPN, easily realized in monolithic form. This connection has voltage gain, necessitating attention to loop frequency compensation. The LT1083-6 regulators use this pass scheme with an output capacitor providing compensation.

Readers are invited to submit results obtained with our emeritus thermionic friends, shown out of respectful courtesy.

Application Note 32

AN32-11

an32f

Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

APPENDIX B

A Low Dropout Regulator Family

The LT1083-6 series regulators detailed in Figure B1 feature maximum dropout below 1.5V. Output currents range from 1.5A to 7.5A. The curves show dropout is signifi cantly lower at junction temperatures above 25°C. The NPN pass transistor based devices require only 10mA load current for operation, eliminating the large base drive loss characteristic of PNP approaches (see Appendix A for discussion).

In contrast, the LT1020/LT1120 series is optimized for lower power applications. Dropout voltage is about 0.05V at 100μA, rising to only 400mV at 100mA output. Quiescent current is 40μA.

LT1083 Dropout Voltage vs Output Current

OUTPUT CURRENT (A)0

MIN

IMUM

INPU

T/OU

TPUT

DIF

FERE

NTIA

L (V

) 2

••

•

••••

•

8

AN32 FB1a

1

021 43 6 7 95 10

INDICATES GUARANTEED TEST POINT

–55°C ≤ TJ ≤ 150°C0°C ≤ TJ ≤ 125°C

TJ = 150°C

TJ = –55°C

TJ = 25°C

LT1084 Dropout Voltage vs Output Current LT1085 Dropout Voltage vs Output Current

OUTPUT CURRENT (A)0

MIN

IMUM

INPU

T/OU

TPUT

DIF

FERE

NTIA

L (V

) 2

••

•

• ••••

AN32 FB1b

1

021 43 65

INDICATES GUARANTEED TEST POINT

–55°C ≤ TJ ≤ 150°C0°C ≤ TJ ≤ 125°C

TJ = 150°C

TJ = 25°C

TJ = –55°C

OUTPUT CURRENT (A)0

MIN

IMUM

INPU

T/OU

TPUT

DIF

FERE

NTIA

L (V

) 2

••

•

•••

•••

AN32 FB1c

1

021 43

INDICATES GUARANTEED TEST POINT

–55°C ≤ TJ ≤ 150°C

TJ = 150°C

TJ = 25°C

0°C ≤ TJ ≤ 125°C

TJ = –55°C

LT1086 Dropout Voltage vs Output Current LT1020/LT1120 Dropout Voltage and Supply Current

OUTPUT CURRENT (A)0

MIN

IMUM

INPU

T/OU

TPUT

DIF

FERE

NTIA

L (V

) 2

••

•

•

•••

AN32 FB1d

1

01.00.5 1.5

INDICATES GUARANTEED TEST POINT

–55°C ≤ TJ ≤ 150°C

TJ = 150°C

TJ = 25°C

0°C ≤ TJ ≤ 125°C

TJ = –55°C

OUTPUT CURRENT (mA)

DROP

OUT

VOLT

AGE

(V) SUPPLY CURRENT (m

A)

0.10

1.00

1 10 1000100

AN32 FB1e

0.01

1

10

0.10.1

Figure B1. Characteristics of Low Dropout IC Regulators

Application Note 32

AN32-12

an32f

Linear Technology Corporation1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 1989

IM/GP 389 20K • PRINTED IN USA

APPENDIX C

Measuring Power Consumption

Accurately determining power consumption often neces-sitates measurement. This is particularly so in AC line driven circuits, where transformer uncertainties or lack of manufacturer’s data precludes meaningful estimates. One way to measure AC line originated input power (Watts) is a true, real time computation of E-I product. Figure C1’s circuit does this and provides a safe, usable output.

BEFORE PROCEEDING ANY FURTHER, THE READER IS WARNED THAT CAUTION MUST BE USED IN THE CONSTRUCTION, TESTING AND USE OF THIS CIRCUIT. HIGH VOLTAGE, AC LINE-CONNECTED POTENTIALS ARE PRESENT IN THIS CIRCUIT. EXTREME CAUTION MUST BE USED IN WORKING WITH AND MAKING CON-NECTIONS TO THIS CIRCUIT. REPEAT: THIS CIRCUIT CONTAINS DANGEROUS, AC LINE-CONNECTED HIGH VOLTAGE POTENTIALS. USE CAUTION.

The AC load to be measured is plugged into the test socket. Current is measured across the 0.01Ω shunt by A1A with additional gain and scaling provided by A1B. The diodes and fuse protect the shunt and amplifi er against severe

overloads. Load voltage is derived from the 100k-4k divider. The shunts low value minimizes voltage burden error.

The voltage and current signals are multiplied by a 4-quad-rant analog multiplier (AD534) to produce the power prod-uct. All of this circuitry fl oats at AC line potential, making direct monitoring of the multipliers output potentially lethal. Providing a safe, usable output requires a galvanically isolated way to measure the multiplier output. The 286J isolation amplifi er does this, and may be considered as a unity-gain amplifi er with inputs fully isolated from its output. The 286J also supplies the fl oating ±15V power required for A1 and the AD534. The 286J’s output is re-ferred to circuit common ( ). The 281 oscillator/driver is necessary to operate the 286J (see Analog Devices data sheet for details). The LT1012 and associated components provide a fi ltered and scaled output. A1B’s gain switching provides decade ranging from 20W to 2000W full scale. The signal path’s bandwidth permits accurate results, even for nonlinear or discontinuous loads (e.g. SCR choppers). To calibrate this circuit install a known full-scale load, set A1B to the appropriate range, and adjust the trimpot for a correct reading. Typical accuracy is 1%.

–

+286J

A = 1

–15VISO

15VISO

FLOATINGCOMMON

Z = X • Y10

AD534 ZY

DANGER! LETHAL POTENTIALSPRESENT IN SCREENED AREA!DO NOT CONNECT GROUNDEDTEST EQUIPMENT—SEE TEXT

X

281 OSCILLATOR/DRIVER

56.2k

150k0.47

1M

10k

100k

1M10k

100k20A

4k

115VAC LINE

0.01Ω*

1N9162

1N11832

TEST LOADSOCKET

15.8k

20W

200W

2kW665k

100k

15V

–

+

–

+

–

+

LT1012

15V

–15V

WATTS OUT0V TO 2V

ALL RESISTORS = 1% FILM RESISTORS* = DALE RH-25

= FLOATING COMMON—AC LINE COMMON

= CIRCUIT COMMONA1B

1/2 LT1057

AN32 FC1

A1A1/2 LT1057

USE A LINE ISOLATED ±15V SUPPLY TO POWER COMPONENTSOUTSIDE SCREENED AREA. DO NOT TIE AND POINTS TOGETHER

Figure C1. AC Wattmeter DANGER! Lethal Potentials Present—See Text