An Impedance-Matching Technique for Increasing the Bandwidth of Microstrip Antennas

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 37, NO. 11, NOVEMBER 1989 1345

An Impedance-Matching Technique for Increasing the Bandwidth of Microstrip Antennas

Abshnct-The nature of the inherent narrow bandwidth of conven- tional microstrip patch antennas is considered. It is observed that, except for single-feed circularly polarized elements, their bandwidth is limited only by the resonant behavior of the input impedance and not by radiation pattern or gain variations, which usually are negligible over a moderate 10 to 20 percent bandwidth. Therefore, broad-band impedance- matching is proposed as a natural solution to increase the bandwidth. The maximum obtainable bandwidth is calculated using Fanos broad- band matching theory. It is found that by using an optimally designed impedance-matching network, the bandwidth can be increased by a factor of at least 3.9, the exact value depending OD the degree of matching required. In view of practical realizations, a transmission-line prototype for a proper matching network is developed. Attention is paid to the translation of this prototype network into a practical structure (e.g. a microstrip or stripline circuit). Practical design examples along with experimental results are given which clearly show the validity of the technique.

I. INTRODUCTION

ICROSTRIP ANTENNAS have many interesting prop- M erties (e.g., low profile, light weight, cheapness), but their application in many systems is impeded by their inherent narrow bandwidth [l]. Many elements with enhanced bandwidth have already been investigated; e.g., electrically thick elements [2], stacked multipatch, multilayer elements [3], multiple-resonator elements [4], [5]. All these wider band elements, however, are characterized by increased complexity and/or enlarged size of the radiating structure. Mostly, their increased impedance bandwidth is also paid for by poorer radiation characteristics. For example, multiple-resonator elements [4], [5] exhibit frequency-dependent array effects that disturb, more or less, the radiation pattern. Increasing the substrate thickness 121, [3], causes increased excitation of substrate waves [6]. Apart from lowering the radiation efficiency, these substrate waves diffract at the substrate edges and deteriorate the quality of the radiation pattern. Although the excitation of substrate waves can be largely avoided by using substrate materials with very low dielectric constant (i.e., er x l), the application of electrically thick antennas only be- comes feasible if proper feeding techniques can be developed

In this paper, broad-band impedance-matching [8] is proposed as a method for bandwidth enhancement of microstrip

[11,[31, ~71.

Manuscript received October 9, 1987; revised March 24, 1988. H. F. Pues is with Emerson & Cuming Europe N.V., Nijverheidsstraat 7,

A. R. Van de Capelle is with the Department of Elektrotechniek, Afd.

IEEE Log Number 8929284.

2431 Westerlo, Belgium.

Microgolven and Lasers, B-3030 Leuven-Heverlee, Belgium.

antennas [9], [lo]. The method is unique in that it does not alter the radiating element itself. Instead, a reactive matching network is added to compensate for the rapid frequency variations of the input impedance. The validity of the technique is based upon the relative frequency insensitivity of the radiation pattern and gain characteristics as compared to the resonant behavior of the input impedance. This is explained in Section 11. In Section 111, both the normally obtained bandwidth and the bandwidth that can be obtained using broad-band matching, are calculated. Dividing the latter quantity by the former one, a bandwidth-enlargement factor is found which depends only on the bandwidth criterion expressed as a maximum al- lowable voltage standing-wave ratio (VSWR). In Section IV, a transmission-line matching-network prototype is derived that can serve as a basis for practical designs. A complete design procedure for an impedance-matched microstrip antenna is outlined in Section V. It is indicated that because of approximations in both the derivation of the prototype and the translation of this prototype to a practical structure, good final designs can be obtained only if proper use is made of computer simulation and optimization. Finally, in Section VI, two practical design examples are described. Both concern S-band microstrip antenna elements: a single substrate rectangular element with a coplanar microstrip matching network, and a square multilayer element with a stripline matching network.

II. FREQUENCY DEPENDENCE OF ANTENNA PARAMETERS An experimental investigation of the frequency depen-

dence of the operational characteristics of common microstrip patch antennas clearly shows that the impedance variations are the dominant bandwidth-limiting factor, whereas the gain (=directivity x radiation efficiency) and radiation pattern variations are almost negligible over a moderate 10 to 20 percent bandwidth. This behavior can be explained easily using the theory of modal expansion in cavities [ 111 as applied in microstrip antenna cavity analysis models [ 121. According to these models, the total input impedance can be written as a sum of modal impedances where each modal impedance behaves as a parallel-resonant circuit. In the same way, the total radiation field can be written as a vector sum of modal radiation fields where each modal field is given as the product of a nearly frequency independent normalized pattern and a frequency dependent scalar excitation coefficient. Thus, it follows that in all cases where only one dominant mode is ex- cited, the input impedance will behave as a parallel-resonant circuit, whereas the (relative) radiation pattern will show almost no frequency variation. Because the operation of single-

0018-926X/89/1100-1345$01.00 O 1989 IEEE

1346 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 31, NO. 11, NOVEMBER 1989

feed circularly polarized (SFCP) microstrip antennas [ 121, [13] is based upon the simultaneous excitation of two orthog- onal modes, the above does not apply for SFCP elements. But in nearly all other cases, there will exist a band of some 10 to 20 percent, where the excitation level of higher order modes is negligible, and the impedance is the only bandwidth-limiting factor. This even applies to microwave scanning arrays [14].

III. BANDWIDTH-ENLARGEMENT FACTOR

In the vicinity of its fundamental resonant frequency, the input impedance of a microstrip antenna can be modeled by either a series-resonant or a parallel-resonant RLC circuit. Indeed, it suffices to choose a proper reference plane on the feed line (preferably as close as possible to the element) or to devise some very simple impedance-transforming circuit, for such a behavior to occur in a more or less approximate fashion. So, assuming an exp (jut) time dependence, the input impedance can be written as either

in the series-resonant case, or as

RO zi, = ~ 1 + jQu

in the parallel-resonant case. In these equations Ro is the resonant resistance, Q is the quality factor and

(3) J r J

where f is the frequency variable and f r the resonant frequency. If the feed line has a characteristic impedance Zo, the input VSWR is given by

(4)

If the bandwidth criterion is taken to be VSWR 5 S, and f, and f2 are the lower and upper band edge frequencies, respectively, so that VSWRCfl) = VSWR(j.2) = S , the bandwidth is given by

It follows from (1)-(5) that

where T = ZO/RO in the series-resonant case, and T = Ro/Zo in the parallel-resonant case. Because, normdy, an antenna is designed to be perfectly matched at its resonant frequency (e.g., by properly locating the position of a coxial feed probe or by using a quarter-wavelength transformer), T normally equals unity. Equation (6) then reduces to the well-known ex- pression [ 121

(7)

feed l i n e reac t i ve matching r a d i a t i n g element

network

Fig. 1. Principle of broad-band matching.

Note, however, that, in order to maximize B, it would be best to take T = Topt # 1 where

Topt = I 2 (S + i) . The application of (8) turns out to be the most elementary form of broad-band impedance-matching (case n = 1 as explained below).

It is evident that the above-calculated bandwidth (7) can be increased, at least in principle, by using an impedance- matching network, as shown in Fig. 1. Ideally, this network would transform the frequency -dependent complex antenna impedance Zi, to a pure real resistance ZO over as large a bandwidth as required. However, there appear to exist some theoretical limitations on such a transformation which are im- posed by nature itself [8]. Indeed, it is impossible to realize a perfect match over a continuous band of frequencies by means of a purely reactive (i.e., linear, passive and lossless) network. The best one can do is to realize a constant (but not perfect) match within the band of operation and a total mismatch out- side this band. In that way, one can either optimize the degree of matching if the bandwidth is given a priori, or maximize the bandwidth if the degree of matching (e.g., VSWR 5 S) is given. The maximum VSWR = S bandwidth obtainable for a series- or parallel-resonant circuit, can be calculated in a straightforward manner using Fano's theory 181, 1151. The result is given by

(9)

This equation expresses that the maximum realizable bandwidth is inversely proportional to both the element quality factor and the specified return loss (expressed in dB).

Because (9) represents the optimum that is theoreti- cally achievable using broad-band matching and (7) gives the normally obtained bandwidth, the maximum bandwidth- enlargement factor is found by dividing both quantities:

Fig. 2 shows this factor which only depends on S and has a minimum value of 3.90 for S = 2.64.

IV. ~NSMISSION-LINE MATCHING-NETWORK PROTOTYPE For increasing the bandwidth by impedance matching, a

proper matching network has to be designed. In this section, a transmission-line matching-network consisting of half-

PUES AND VAN DE CAPELLE: INCREASING BANDWIDTH OF MICROSI'RIP ANTENNAS 1347

I I I I 3

J 7 9 1 1

s - Fig. 2. Bandwidth-enlargement factor versus specified VSWR

1 ,%, fie

Z c 1 I _I

(a) (b) Fig. 3. Transmission-line models for antenna impedance. (a) Parallel-

resonant case. @) Series-resonant case.

wavelength open-circuited stubs and quarter-wavelength inter- connecting lines, is derived that is useful as a prototype for practical realizations at microwave frequencies. This prototype has enough degrees of freedom to ensure practical real- izability in microstrip or stripline, if the design bandwidth is not less than about 4 percent. It is clear that other prototypes could be devised depending on the desired practical realization form of the matching network (e. g . , quasi-lumped-element prototypes for MMIC realizations or coupled-transmission- line prorotypes for compact interdigital realizations), but such other prototypes are not considered in this paper (except for some short references to lumped-element approaches in this and the following section).

In general, the design of a broad-band matching network is a difficult network synthesis problem. Therefore, published results are used as much as possible in the present derivation. Particularly, the modified Chebyshev equal-ripple characteristic as proposed by Fano [8], is adopted. In [16], normalized low-pass prototype element values for an LC-ladder circuit having this characteristic, are given for the case of either a parallel-RC or a series-RL load. These normalized design parameters (called gi -parameters) are used below to synthesize the present prototype.

The parallel-RC or series-RL loads of the low-pass prototype are transformed to the band-pass resonant models of Fig. 3 by setting

where

Fig. 4. Intermediate matching-network prototype consisting of open- circuited transmission-line stubs and admittance inverters (series-resonant case).

and fLp is the low-pass frequency variable. By this frequency transformation, parallel-C elements are transformed into parallel open-circuited half-wavelength stubs and series-L elements into series short-circuited half-wavelength stubs. Be- cause the latter are not physically realizable, they are removed from the matching network by using admittance inverters J characterized by their Y-matrix

In this way, the intermediate matching-network of Fig. 4 is obtained in the series-resonant case, and a similar one (con- taining an additional inverter J12) in the parallel-resonant case.

The transmission-line resonant models of Fig. 3 are almost equivalent (at least over a moderate bandwidth) to the lumped- element RLC-circuits used in Section I11 (using f L p = V / B instead of (1 1) would have yielded these). Their quality factor is given by

in the parallel-resonant case (Fig. 3(a)), and

in the series-resonant case (Fig. 3(b)). With respect to Figs. 3 and 4, it can be observed that all line

sections are a half-wavelength long at the resonant frequency f r , Ro is the resonant antenna resistance, Yci(Zci ) is the characteristic admittance (impedance) of the ith transmission- line resonator, Jij+l is the admittance inverter between resonators i and i + 1, Jn,,,+1 is a final impedance-scaling admittance inverter, and 2 0 is the (real) system impedance (usually 50 0). It can be seen that the first resonator (i = 1) be- longs to the antenna model itself, whereas the following ones (i = 2 , 3, . . . ,n) belong to the matching network. If one re- stricts the antenna model to the patch element proper so it does not include a possible feed probe inductance, the latter can be included in the i = 2 resonator [7], [17], [18], as discussed in Section V.

The different network parameters Yci and Jij+l must satisfy the following: I

t tan ( ; B ) (parallel-resonant case) (16)

1348 IEEE TRANSACT1 ONS ON ANTENNAS AND PROPAGATION, VOL. 37, NO. 1 1 , NOVEMBER 1989

Fig. 5 . Final transmission-line prototype for broad-band matching network (series-resonant case).

yC2 = (series-resonant case) (17) M O

The g;-parameters are found from [16], and are a function of the order of the network n (to be chosen by the designer) and the decrement

(20) A = - 7r

2AQ

Observe that, by definition, go = 1 and g l = 1/6. To obtain a prototype that is practically realizable, the ad-

mittance inverters are replaced by quarter-wavelength lines. Furthermore, to increase the number of degrees of freedom, the half-wavelength stubs are splitted up in two quarter- wavelength sections with different characteristic impedances. In this way, the final prototype is obtained which is depicted in Fig. 5 for the series-resonant case. For the networks of Fig. 4 and Fig. 5 (series-resonant case) and their corresponding ones (parallel-resonant case) to be approximately equivalent, the following equations have to be satisfied for i = 2, 3, . . . , n [15]:

yY+l= J;j+l cos (:B)

where

r =tan ( :B) and the ai-parameters can be freely chosen. In the parallel- resonant case, (21) also applies for i = 1, and in the series- resonant case, the first term between the inner parentheses in (22) vanishes for i = 2.

V. DESIGN PROCEDURE FOR AN IMPEDANCE-MATCHED ANTENNA

This section summarizes the complete procedure for de- signing a broad-band impedance-matching network for a given

microstrip antenna element. First, the antenna impedance is made to be resonant at the center frequency of the band, as explained in Section 111. Then, the antenna model parameters f ?, Ro, and Q are determined. This has to be done very carefully, by preference trough accurate measurements, because most analysis models are not accurate enough for this purpose [ 151.

Once the antenna parameters are known, the order n and the bandwidth B (if not given a priori) are to be determined. Us- ing (20) and [ 161, a deliberate choice can be made. The choice of n typically reflects a trade-off between increased bandwidth and/or degree of matching (the larger n, the nearer the optimum (9) will be approached) on the one hand and increased complexity (the network will become larger and lossier) on the other. Typical values for n are 2, 3, or 4. The case n = 1 is trivial and has been dealt with in Section I11 (8). The approaches of [7] and [17] could be described as n = 1.5 (feed probe inductance resonated by series capacitor at center frequency without first optimizing the inductance value) whereas [ 181 gives a real n = 2 lumped-element approach.

Knowing n and 6, the g;-parameters (i = 2, 3, . . . , n) are found from [16]. The parameters of the intermediate prototype (Fig. 4) then follow from (16) or (17), (18) and (19). Subsequently, the parameters of the final prototype are derived from (21)-(23). In this process, there are 2n - 3 degrees of freedom in the series-resonant case and 2n - 2 in the parallel-resonant case. One could, for example, choose freely the Yc;-parameters (except Yc2 in the series-resonant case) and the a;-parameters. By making these choices in a deliberate fashion, it is normally possible to obtain a practically realizable prototype, i.e., a prototype that, when translated to a physical lay-out, yields line widths that are neither too wide nor too narrow.

The final step of translating the prototype to a practical circuit is a very critical one. Indeed, for getting good results, it is absolutely essential that the effects of discontinuities (such as open ends, steps and T-junctions) are compensated. Therefore, to avoid lengthy trial-and-error tuning procedures, the application of computer simulation and optimization techniques is highly desirable. This also allows to compensate for the different approximations in the design of the prototype itself, i.e., the use of approximate transmission-line models for the antenna impedance (Fig. 3) and the approximation of the intermediate prototype (Fig. 4) by the final prototype (Fig. 5 ) .

VI. APPLICATIONS

A . Single-Substrate Impedance-Matched Rectangular Antenna

The first design example concerns an integrated impedance- matched antenna consisting of a rectangular microstrip antenna and a coplanar microstrip impedance-matching network. The whole structure is laid out on top of a 20 cm x 15 cm x 1.6 mm RT/duroid 5880 substrate (er = 2.20), as shown in Fig. 6. A similar antenna with a shielded-microstrip matching network (where the shield height was tuned to optimize the network response), has been described elswehere [lo], [ 191.

PUES AND VAN DE CAPELLE: INCREASING BANDWIDTH OF MICROSTRIP ANTENNAS 1349

Fig. 6 . Layout of rectangular impedance-matched antenna (antenna #l).

The following antenna parameters, calculated from an im- proved transmission-line model [20], were used in the present design: fr = 3.027 GHz, Ro = 48.88 R and Q = 22.64 (parallel-resonant case). The design of the circuit was based on the following choices: n = 3, B = 10 percent, Z:3 = 130 R, Yc2 = Yc3, and a2 = a3 = 1 . With ZO = 50 51, this yielded: 2: = 65.72 R, Z:4 = 72.28 R, ZL2 = 2: = 25.78 R and ZL3 = Z f 3 = 25.33 R. When translating these values to the microstrip circuit shown in Fig. 6, both the i = 2 and i = 3 resonators were realized as two parallel identical stubs in order to reduce their line width.

To be able to judge the performance of this impedance- matched antenna properly, a reference antenna (Fig. 7) has been built in the same process (a piece of substrate cut from the same sheet was used). This reference antenna is completely identical to the impedance-matched antenna except that the matching network is replaced by a simple 50 R microstrip line. Note that the calculated edge-fed impedance of the antenna element (i.e., 48.88 51) is very nearly equal to 50 R. Hence, the reference antenna should be well matched at f = fr. Fig. 8 shows the return loss of both antennas. The reference antenna has its best match at 3.025 GHz (-21.5 dB) and has a higher order mode dip at 3.424 GHz. This higher order mode dip is very much suppressed by the matching network as shown by the other curve. Within the band of operation, the impedance-matched antenna has its worst match at 3.035 GHz (-8.8 dB). It can be seen, that the bandwidth at this level (S = 2.14) has been increased by a factor of 3.2 to a value of 275 MHz or 9.1 percent, whereas the theoretical maximum bandwidth-enlargement factor for this degree of matching equals 4.0 (Fig. 2).

Fig. 7. Layout of reference antenna (antenna #2).

Si1 L M l o g MAG REF 0 . 0 dB

2 . 5 dB/

V

START 2.600000000 GHz STCP 3.600000800 R l z

Fig. 8. Return loss versus frequency of antennas #1 and #2.

fr, the mismatch loss of antenna #1 (impedance-matched antenna) within its band of operation is less than that of antenna #2 (reference antenna). However, because the matching network will inevitably be somewhat lossy, one could ask if the decrease of the mismatch loss is not annihilated by the increase of the dissipation loss. That this is not the case, is demon- strated by Fig. 9 which shows the transmission performance of both antennas. Particularly, a radiation link was established between a standard gain horn on the one side and antenna #1 or &2 on the other. The figure shows the transmission co- It is clear from Fig. 8 that, except in a small band around _ _ ,, - _ .~ . ~ ~ . ~~~~ --U

1350 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 31, NO. 1 1 , NOVEMBER 1989

S a l 8 M l o g MAG REF -28.0 dB

2 . 0 dB/

START 2.600000000 GHz STOP 3.400000000 GHz

Fig. 9. Transmission characteristic versus frequency of antennas #1 and #2.

+

+

t +

+ +

-t Cu -Clad

Fig. 10 Multdayer impedance-matched antenna (antenna #3).

efficient measured in these two cases. This characteristic is almost proportional to the realized gain. It follows that antenna #1 is a more efficient radiator over the 2.832 - 2.988 GHz band and the 3.055 - 3.174 GHz band, whereas antenna #2 is more efficient in between. The maximum difference in this center band equals 0.61 dB and occurs at 3.026 GHz (i.e., the frequency of best match of antenna #2).

Concerning radiation patterns, E- and H-plane cuts for both antennas have been measured at 2.9, 3.0, and 3.1 GHz [15].

They do not show any appreciable difference, which proves that the matching network, although it is coplanar with the patch, does not affect the radiation characteristics. It is to be observed, however, that only copolar patterns were measured.

B . Multilayer Impedance-Matched Square Antenna The second design example concerns a multilayer square

microstrip antenna with a stripline matching network situ- ated underneath the antenna ground plane. A similar antenna


521 B M l o g MAG REF -25.0 dB

2 . 5 dB/ v -29.48 dB

1 START 2.800000000 GHz STOP 3.800000000 GHz

Fig. 12. Transmission characteristic of antenna #3 and standard gain horn (antenna #4).

with an underneath microstrip matching network has been described elsewhere [2 l].

The present antenna is shown in Fig. 10. It is a sandwich structure consisting of (from top to bottom) a 0.5 mm Cu-Clad 217 substrate bearing the antenna patch, a 6.4 mm Eccofoam PP-2 layer, a first metal ground plate (the antenna ground plane), two 1.6 mm Cu-Clad 217 substrate layers bearing the stripline matching netwbrk, and a second bottom ground plate onto which an OSM 203-1 stripline connector is attached. The overall dimensions (apart from the connector and four connecting screws) are 70 mm x 70 mm x 14 mm.

The antenna model parameters were fr = 3.28 GHz, RO = 33.3 R and Z,1 = 151.5 R (series-resonant case). Choosing n = 2, b = 12 percent, a2 = 0.3 and Z O = 50 R, the design was carried out straightforwardly. Using computer-

aided simulation and optimization, adjustments were made to compensate for the different approximations. The measured return loss diagram is shown in Fig. 11. Considering the -16.44 dB (or S = 1.35) level, which is the maximum level in the band of operation, a bandwidth of 324 MHz or 9.9 percent is obtained. Using (7) and (15), the unmatched antenna is found to have a bandwidth of only 4.2 percent at this level. Observe also that a better match than -14 dE3 is obtained within the design bandwidth of 12 percent.

The transmission performance is illustrated in Fig. 12. This figure shows the transmission coefficient between a log-periodic dipole array antenna on the one side and the impedance-matched antenna or a standard gain horn (Narda Model 644) on the other side. It follows that the realized gain is about 8 dB over a bandwidth of 12 percent. This high gain


I , dB

to-polar -5

-10

I

tI

dB -25

Fig. 13. (a) Measured radiation patterns at 3.100 GHz of antenna #3. (b) Measured radiation patterns at 3.300 GHz of antenna #3. (c) Measured radiation patterns at 3.500 GHz of antenna #3.


(C) Fig. 13. (Continued.)

value for a single square element is partly due to the deliberate choice of the horizontal dimensions (70 mm x 70 mm). Mounted on a large ground plane, the gain would be somewhat less.

Finally, Fig. 13 shows the E- and H-plane CO- and cross- polar patterns at 3.1, 3.3, and 3.5 GHz. These patterns do not show any significant change within the band of operation.

VII. CONCLUSION In this paper, broad-band impedance matching has been

proposed as a powerful technique to increase the bandwidth of microstrip antennas. The theoretical limitations have been described and a practical design method for the required matching networks has been outlined. The validity of this design procedure has been illustrated by two representative design examples. However, it should be stressed that impedance-matching is a very general technique and that many other design procedures and realization forms could be devised, which possibly could yield better results.

REFERENCES

[l] D. M. Pozar, An update on microstrip antenna theory and design including some novel feeding techniques, ZEEE Antennas Propagat. Soc. Newsletter, vol. 28, no. 5, pp. 5-9, Oct. 1986. E. Chang, S. A. Long, and W. F. Richards, An experimental investigation of electrically thick rectangular microstrip antennas, ZEEE Trans. Antennas Propagat., vol. AP-34, pp. 767-772, June 1986. C. H. Chen, A. Tulintseff, and R. M. Sorbello, Broadband two-layer microstrip antenna, in ZEEE Antennas Propagat. Soc. Znt. Symp. Dig., 1984, pp. 251-254. G. Kumar and K. C. Gupta, Directly coupled multiple resonator wideband microstrip antennas, ZEEE Trans. Antennas Propagat., vol. AP-33, pp. 588-593, June 1985.

[ 5 ] H. Pues, J. Bogaers, R. Pieck, and A. Van de Capelle, Wideband quasi-log-periodic microstrip antenna, Inst. Elec. Eng. P m . , vol.

128, pt. H, pp. 159-163, June 1981. [6] A. K. Bhattacharyya and R. Garg, Effect of substrate on the efficiency

[2]

[3]

141

of an arbitrarily shaped microstrip patch antenna, ZEEE Trans. An- tennas Propagat., vol. AP-34, pp. 1181-1188, Oct. 1986. K. S. Fong, H. F. Pues, and M. J. Withers, Wideband multilayer coaxial-fed microstrip antenna element, Elactron. Lett., vol. 21, pp. 497-499, May 23, 1985. R. M. Fano, Theoretical limitations on the broadband matching of ar- bitrary impedances, J. Franklin Inst., vol. 249, nos. 1-2, pp. 57-83 and 139-154, Jan.-Feb. 1950. H. F. Pues and A. R. Van de Capelle, Impedance-matching of microstrip resonator antennas, in URSZ North Amer. Radio Sci. Meet. Dig., Quebec, June 1980, p. 189. Broad-band microstrip antenna, U.S. Patent 4445 122, Apr. 24, 1984. R. F. Harrington, Time-Harmonic Electromagnetic Fields. New York: McGraw-Hill, 1961, pp. 431-440. K. R. Carver and J. W. Mink, Microstrip antenna technology, ZEEE Trans. Antennas Propagat., vol. AP-29, pp. 2-24, Jan. 1981. P. C. Sharma and K. C. Gupta, Analysis and optimized design of single feed circularly polarized microstrip antennas, ZEEE Trans. Antennas Propagat., vol. AP-31, pp. 949-955, Nov. 1983. J. S. Lee and W. J . Furlong, An extremely lightweight fuselage- integrated phased array for airborne applications, ZEEE Trans. An- tennas Propagat., vol. AP-29, pp. 178-182, Jan. 1981. H. F. Pues, Study of the bandwidth of microwave integrated antennas: Development of design models for wideband microstrip antennas (in Dutch), Ph.D. dissertation, Microwaves and Lasers Div., Catholic Univ. Louvain, Louvain, Belgium, 1983. G. L. Matthaei, L. Young, and E. M. T. Jones, Microwave Filters, Impedance-Matching Networks, and Coupling Structures. New York: McGraw-Hill, 1964, sec. 4.09-4.10. J. M. Griffin and J. R. Forrest, Broadband circular disc microstrip antenna, Electron. Lett., vol. 18, pp. 266-269, Mar. 18, 1982. D. A. Paschen, Practical examples of integral broadband matching of microstrip antenna elements, in Proc. 1986 Antenna Appl. Symp., Monticello, IL, Sept. 17-19, 1986. H. F. Pues and A. R. Van de Capelle, Wideband impedance-matched microstrip resonator antennas, in Inst. Elec. Eng. Conf. Pub. 195 (Antennas and Propagation), pt. 1, pp. 402-405, Apr. 1981. - , Accurate transmission-line model for the rectangular microstrip antenna, Inst. Elec. Eng. P m . , vol. 131, pt. H, pp. 334-340, Dec. 1984. H. Pues, A. Van Kauteren, J. Vercruysse, and A. Van de Capelle, Broadband microstrip radar antenna element, in Proc. Znt. Conf. Radar, Paris, May 1984, pp. 298-303.


Hugo F. Pues (S76-M82) was born in Leuven, Belgium, on May 2, 1954. He received the degrees of Electromechanical Engineer in 1977 and Doctor in Applied Sciences in 1983, both from the Katho- lieke Universiteit Leuven.

His doctoral research focused on bandwidth- enhancement techniques for microstrip antennas. In 1983-1984, he worked for ERA Technology Ltd., Leatherhead, UK, in the field of antenna design and numerical analysis of electromagnetic problems. He then returned to the Katholieke Universiteit Leuven

where he was involved in research work on microstrip antennas and circuits, nucrowave power applications and numerical analysis. Since January 1987, he has worked for Grace N.V., (formerly Emerson & Cuming Europe N.V ), Westerlo, Belgium, where he is now the R & D manager of the Microwave Product Line with interests mainly directed towards computer-aided measure- ment and design of advanced absorbing materials.

Antoine R. Van de Capelle (S70-M84) was born in Nazareth, Belgium, in 1946. He received the M.Sc., Ph.D., and Special Doctors degrees from the Katholieke Universiteit Leuven in 1970, 1973, and 1979, respectively

In 1970 he joined the Department of Electrical Engineering of the K. U. Leuven, where he is now a Professor. In 1974 he established a research group on antennas, which for the past 15 years has been concentrating on microstrip antennas. The groups current research programs involve radio communi-

cation systems with projects on maritime satellite terminals, antenna mea- surement techniques, propagation on high-frequency communication links, S.S.R.-radar systems and mobile telephone systems. As a Professor at the K. U. Leuven, he teaches courses on telecommunication systems and antennas and propagation.

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An Impedance-Matching Technique for Increasing the Bandwidth of Microstrip Antennas