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A 32 GHz Low-Power Low-Phase-Noise VCO
Implemented in SiGe BiCMOS Technology
Yuhua Qi, Rulong He, and Zhao Shen Electronic Engineering College, Naval University of Engineering, PLA, Wuhan 430033, China
Email: [email protected]; [email protected]; [email protected]
Abstract—A low-phase-noise, low-power Ka-band Voltage-
Controlled Oscillator (VCO) using cross-coupled pair
configuration is presented. The Ka-band VCO circuit uses 0.18
μm SiGe BiCMOS technology. The VCO has low phase noise
of -114.6 dBc/Hz at 1 MHz offset from 32.12 GHz carrier
frequency and can be tuned from 30.17 to 33.48 GHz. The
figure of merit is -202.1 dBc/Hz. The power consumption of the
VCO with 0.27 mm2 chip area is 1.8 mW from a 1.5 V power
supply.
Index Terms—VCO, low power, low phase noise, SiGe
BiCMOS
I. INTRODUCTION
The recent advances in wireless communications
indicate a trend toward integrating multiple
communication standards into a single device. Such
devices could be implemented with multiple dedicated
front-ends integrated in parallel. However, that solution
requires larger silicon area and possibly more power
consumption. Reconfigurable radio transceivers, on the
other hand, optimize the functionality versus power and
area tradeoffs by programming a wideband front-end to
the desired standard. In such multistandard reconfigurable
radio transceivers, one of the most challenging building
blocks is the frequency synthesizer, which needs to
generate the local-oscillator signal with both a wide
tuning range and sufficiently high spectrum purity [1],
Phase-locked loops are widely used in wireless
communication systems. For broadband or multiband
systems, such as TV broadcasting systems and Global
Navigation Satellite System [1]. Voltage-controlled
oscillators (VCOs) are crucial building blocks in radio-
frequency phase-locked loops (RF PLLs), which are
usually used as frequency synthesizers to provide stable
and reliable frequency sources in wireless communication
systems [2], [3].
VCOs are important blocks in any communication or
test and measurement system. They are widely used in
wireless and wired receiving, data communication and
clock recovery systems [1]. Hence, extensive efforts have
Manuscript received December 19, 2016; revised February 28, 2017.
This work was supported by the Advance Research project of China under Grant No. 51307xxxx09, and the Advance Research project of
PLA in China under Grant No. KJ2013××××. Corresponding author email: [email protected].
doi: 10.12720/jcm.12.2.98-104
been made to improve the performance of VCOs. Several
parameters to evaluate the performance of a VCO are
available, including oscillation frequency, oscillation
amplitude, frequency tuning range, phase noise and
power consumption. Frequency tuning range and phase
noise are the most critical two parameters among all these
specifications [2], [3]. In this paper, we focus on the
improvement of power and the reduction of phase noise.
Traditionally, such VCO circuits have been the
exclusive domain of III-V compound semiconductors
such as GaAs or InP [4], [5]. However, over the past
several years, silicon and silicon germanium (SiGe) have
emerged as low cost alternatives that deliver the
performance of III-V compound semiconductors. A
number of RF VCOs have been reported using SiGe
BiCMOS [6]-[8], and CMOS technology [9]-[11]. With
rapid developments in silicon technologies, these systems
become feasible to be implemented using BiCMOS or
CMOS to achieve low cost, low power, and high
integration. In summary, the VCOs which are highly
integrated, having low phase noise, low power and wide
band are the trend for the future. So this paper presents
the design of an integrated millimeter wave VCO which
has a low power consumption.
The phase noise of VCO is determined by the variable
device used in the resonating tank, for instance, varactor
diodes commonly control the phase noise in a typical LC
VCO. Therefore, in an LC VCO varactor size should be
carefully chosen to decrease the phase noise. At
millimeter-wave frequencies, two topologies (cross-
[9]) are widely
used. However, it is well known that the maximum
oscillation frequency of a Colpitts VCO is higher than
that of a cross-coupled VCO, meaning that a Colpitts
VCO can achieve relatively lower phase noise. One way
to further extend the output frequency range is to use
push-push oscillators. However, as the frequency
increases, the output power from a push-push oscillator is
usually low, as reported in. The output power can be
improved through a tuning network at the output, but
only for a narrow bandwidth [11]. It can be concluded
that both topologies are capable of very good phase noise.
However, it is well known that the maximum oscillation
frequency of a Colpitts VCO is higher than that of a
cross-coupled VCO.
Much attention has been devoted to the power
consumption performance of oscillators in the past, while
Journal of Communications Vol. 12, No. 2, February 2017
98©2017 Journal of Communications
[2].
[6], coupled VCO [7] and Colpitts VCO
phase noise was not concerned with much. Since power
consumption is precious and more stages lead to more
parasitic problems and risks in circuit design at high
frequency, the low-phase-noise oscillator core make it
predominant in applications, especially where cost, power
consumption and reliability are important. In recent years,
some researches on oscillator phase-noise were carried
out. By introducing transform method into oscillator
design, low phase noise was achieved but power rose at
the same time. In this paper, a Ka-band cross-coupled
VCO was made with low phase noise and low power for
direct use without any additional stage. For this purpose,
the analysis for low power consumption has been done in
detail and the inductors of the resonating tank were
constructed by microstrip lines to obtain high quality
factor (Q). Design optimizations of the resonating tank
were made carefully, and an excellent result was achieved.
The cross-coupled VCO achieves an oscillation
frequency range from 30.17 to 33.48 GHz. The phase
noise is -114.6 dBc/Hz at 1 MHz offset from the
oscillation frequency 32.12 GHz, which is very excellent
result for low-power Ka-band oscillators.
II. CIRCUIT DESIGN
A. Design for Low Power
A general LC-tuned VCO can be symbolized as in Fig.
1. The oscillator consists of an inductor L and a capacitor
C, building a parallel resonance tank, and an active
element -R, compensating the losses of the inductor
represented by resistor RL in Fig. 1, and the losses of the
capacitor represented by the resistor RC. This circuit
results in an oscillator with angular center frequency:
C
1
LC (1)
The general LC-tuned VCO of Fig. 1 can be redrawn
in Fig. 2, neglecting the capacitor losses, as the series
resistance of the integrated inductors largely dominates
the tank losses. Using the energy conservation theorem,
the maximal energy in the inductor must equal the
maximal energy stored in the capacitor:
2 2
peak peak
2 2
CV LI (2)
where Vpeak is the peak amplitude voltage of the sine
wave voltage across the capacitor and Ipeak is the peak
amplitude current of the sine wave current through the
inductor. This current is flowing to the resistor RS, so
the effective loss in the tank can be calculated as:
2
peak2S Sloss peak
2 2
CVR RP I
L (3)
or with (4):
2 2 2S Sloss peak peak2 22 2
C
C
R RP C V V
L
(4)
L
C RC
RL
-R
Fig. 1. Basic LC-VCO.
L
C
RS
Fig. 2. Basic LC-resonator tank.
This loss must be compensated by the active part of the
VCO to sustain the oscillation. Ploss in the above equation
is the fundamental minimum for the power consumption
of a LC-VCO. The equations lead to some significant
conclusions for the design of low power consumption of
any LC-VCO:
1) Power consumption decreases linearly with the
series resistances in the resonance;
2) Power consumption decreases quadratically when
the tank inductance is increased so that the oscillator
frequency is assumed to be a constant;
3) Power consumption decreases quadratically when
the tank capacitance is decreased so that the oscillator
frequency is assumed fixed;
4) Power consumption decreases quadratically with the
oscillator frequency when the tank inductance is a
constant. Power consumption increases quadratically with
the oscillation frequency when the tank capacitance is a
constant.
B. Design for Low Phase N oise
The phase noise (PN) for a cross-coupled VCO is
given by [2]:
2
o
sig t ank
2PN 10log
2
kT
P Q
(5)
where K is Boltzmann constant, T is the absolute
temperature, ωo is the oscillation frequency, Qtank is the
quality factor of the LC resonating tank, Δω is the
frequency offset from the carrier, and Psig is the average
Journal of Communications Vol. 12, No. 2, February 2017
99©2017 Journal of Communications
signal power consumption in the resistive part of the tank,
which is given by
tank
2
sig
P2
VP
R (6)
where Vtank is the voltage swing of the LC tank and RP is
the parallel equivalent resistance.
From (5) and (6), the phase noise can be deduced as
2
oP
2
t ank t ank
4PN 10log
2
kTR
V Q
(7)
Equation (7) indicates that the phase noise at a given
frequency offset depends on the voltage swing of the tank
Vtank and the quality factor of the LC resonating tank Qtank.
To improve the phase noise, one can increase either the
Vtank or the Qtank.
However, the Vtank is limited by the bias current of the
oscillator and the breakdown voltage of the active devices.
In order to obtain high Vtank, the size of the transistors
should be selected large enough to apply high bias current,
but elevating the bias current leads to high power
consumption. Therefore, there is always a compromise
between the high Vtank of an LC VCO and its power
dissipation.
The Qtank is another one of the most important
parameters in LC VCOs. It affects most VCO properties,
including the overall tank loss, and the phase noise. Qtank
is also a key factor for power calculation and estimation.
The better Qtank is, the lower power achieves. In the low-
frequency bands (<10 GHz), the tank is dominated by the
Q of the inductor (QL). However, as the frequency
increases to higher than 10 GHz, the Q of the varactors
(Qvar) decreases significantly [12]. Thus, the Qvar
determines the overall tank at the millimeter-wave
frequencies because Qvar is far less than QL.
tan k C L var
1 1 1 1
Q Q Q Q (8)
At the same time, the tuning range is also determined
by the varactor size. Thus, it is critical to obtain high-Q
and large-ratio (Cmax/Cmin) varactors to achieve better
VCO performance. However, there is a tradeoff between
the Qvar and the capacitance ratio. If the large capacitance
but the Qvar is minimized, in turn to degrade the phase
noise. In addition, the varactor should be carefully laid
out to reduce parasitic.
Although an oscillator is a strongly nonlinear circuit
which operates under large-signal conditions, an efficient
method to gain insight into the limitations of the available
transistor technology is to start off with a small signal (S-
parameter) analysis. Compared with transient or
harmonic balance simulations, small-signal simulations
are fast, do not suffer from nonconvergence problems,
and do not require initial conditions. This is particularly
important when simulating bipolar VCOs at millimeter-
wave frequencies, the transistor model must capture self-
heating and avalanche multiplication, which are two
positive feedback phenomena that cause significant
nonconvergence problems at large voltage swings.
During the transient simulations, the impedance of the S-
parameter signal source is set to a small value (i.e.,10–20),
to clearly reflect the expected effective resistance of the
tank inductance at the oscillation frequency, and thus to
allow for the onset of oscillations to occur.
As illustrated in Fig. 3, by plotting the real and
imaginary parts of the impedance looking into the base of
the HBT, Rin and Xin, respectively, one can quickly assess
whether the circuit produces an adequate negative
resistance at the desired oscillation frequency, fosc,
(approximately the frequency at which Xin=0 and Rin<0 ),
and one can estimate the tuning range of the VCO. From
Fig. 3, it can be concluded that fosc is about 32 GHz and
36.5 GHz when Vtune is equal to be 0 V and 1.5 V,
respectively. Fig. 4 shows the simulated oscillation
frequency of the VCO by the means of harmonic balance
simulation, it can be seen that the oscillation frequency
range is from 31.2 GHz to 35.89 GHz, which is in
accordance with that concluded with S-parameter analysis.
25 30 35 40-1.5x10
-2
-1.0x10-2
-5.0x10-3
0.0
5.0x10-3
1.0x10-2
1.5x10-2
2.0x10-2
Imp
eda
nce
/
Frequency /GHz
Rin @V
tune=0 V
Xin @V
tune=0 V
Rin @V
tune=1.5 V
Xin @V
tune=1.5 V
Fig. 3. Simulated input impedance as the tune voltage Vtune is changed
from 0 V to 5 V.
0.0 0.5 1.0 1.529
30
31
32
33
34
35
36
37
Fre
qu
ency
/G
Hz
Vtune
/V
Fig. 4. Simulated oscillation frequencies of the VCO.
III. IMPLEMENTATION AND MEASUREMENT RESULTS
The schematic of cross-coupled VCO designed in this
paper is shown in Fig. 5, in which the inductors (Ltank)
and the MOS varactors (Ctank) form the LC resonating
tank. A cross-coupled differential transistor pair (Q1 and
Q2) with positive capacitive feedback (C) provide the
necessary negative resistance to compensate the tank
Journal of Communications Vol. 12, No. 2, February 2017
100©2017 Journal of Communications
(6).
ratio is chosen, the Tuning Range (TR) can be maximized,
losses for sustaining oscillation. A simple current source
(Io) provides a constant current for the VCO.
Ltank
Ctank
Vtune
VDD
Q2
Io
Ltank
Ctank
Q1
Vout- Vout+
Vbias
Rb Rb
C C
Fig. 5. Schematic of cross-coupled VCO designed in this paper.
The VCO has been designed and fabricated in a 0.18
μm SiGe BiCMOS technology. The die microphotograph
of the chip is shown in Fig. 6 and the whole chip area
including all testing pads is 0.47 × 0.58 mm2. The voltage
and current source HP4142B was used to supply the dc
voltages, meanwhile the output was connected through a
Ground-Signal-Ground (GSG) probe to the spectrum
analyzer Agilent N9030A with a phase-noise
measurement utility.
of the VCO
Fig. 7 shows the measured tuning characteristic of the
VCO versus Vtune. At the supply voltage of 1.5 V, the tail
current of the VCO core is set to be 1.2 mA current. The
tuning range is 3.31 GHz or 10.4% from 30.17 to 33.48
GHz when the Vtune changes from 0 to 1.5 V. The linear
range is 2 GHz from 31.02 to 33.02 GHz while the Vtune
varies from 0.45 to 1.05 V, thereby the linear gain of
VCO is calculated to be 3.33 GHz/V. The measured
phase noise at 32.12 GHz is shown in Fig. 8. At 100 kHz
and 1 MHz offset frequency from the carrier, the phase
noise is -78.8 dBc/Hz and -114.6 dBc/Hz, respectively.
In order to prove the feasibility of the design in the
paper, the comparison of simulated results and measured
results is shown in Table I.
TABLE I: COMPARISON OF SIMULATED RESULTS AND MEASURED
RESULTS
fosc (GHz) PN (dBc/Hz) △f
(MHz) PVCO (mW)
simulated 31.20~35.89 -116.8 1 2.0
measured 30.17~33.48 -114.6 1 1.8
0.0 0.5 1.0 1.529
30
31
32
33
34
35
Fre
qu
ency
/G
Hz
Vtune
/V
of the VCO
102
103
104
105
106
-140
-120
-100
-80
-60
-40
-20
0
Frequency offset (Hz)
Ph
ase
no
ise
(dB
c/H
z)
of the VCO
We compare the performance of VCO circuit at
different frequencies with previous reports, and the
measured characteristics are normalised to the figure of
merit (FOM), defined as:
osc VCOFOM PN 20log 10log1mW
f P
f
(9)
where PN is the measured phase noise at the offset
frequency Δf from the carrier frequency fosc. PVCO is the
VCO power consumption in mW. A larger |FOM| denotes
a better oscillator. The FOM of this 32 GHz VCO circuit
is -202.1 dBc/Hz at 1 MHz offset. Table II summarises
the measured performance of this VCO and includes
other reported performances for comparison. The VCO
Journal of Communications Vol. 12, No. 2, February 2017
101©2017 Journal of Communications
Fig. 7. Measured oscillator frequency .
Fig. 6. Die microphotograph .
Fig. 8. Measured phase noise .
reported in this paper achieves a low power and low
phase noise, which compares well with other published
results [2], [6], [10]-[20].
The important parameters of a VCO are the center
oscillation frequency fosc, the phase noise PN measured at
an offset of Δf from fosc, the output power Pout, the power
dissipation PVCO, the tuning range TR achieved by
varying the tuning voltage Vtune, and the variation of
tuning voltage ΔVtune. To fairly compare the VCOs
reported in the available literatures, several figure-of-
merits (FOMs) have been presented.
in is the most widely used, but it
excludes the TR which is one of the most important
properties of a VCO. Even though some systems do not
require a wide range or need only operate at a single
frequency, a wide TR is nevertheless required because of
variations caused by the parasitics at millimeter-wave
frequencies. There is a direct tradeoff between the TR
and other properties. Therefore, the TR should be
included, as described in FOMT depicted in 10).
oscT
VCO
FOM PN 20log10%
10log1mW
f TR
f
P
(10)
However, FOMT does not take Pout into account, and as
a result, output power will not affect the FOMT even if a
VCO generates a very little output power. It may be
supposed that the output power is already considered in
the phase noise, but the phase noise is determined by the
tank swing, which should be distinguished from the
output power. As the tank swing is large, the output
power can be low if the output buffer is improperly
designed. In addition, if the output power is low, then
more buffers and more power dissipation are required to
satisfy the system specifications because mixers, for
example, require a large amount of LO power. Therefore,
the output power should be included, as described in
FOMTP depicted in 11).
oscTP
VCOout
FOM PN 20log10%
10log1mW
f TR
f
PP
(11)
Lastly, the ΔVtune is another important parameter of a
VCO, because there is a direct tradeoff between the
ΔVtune and the TR. The bigger the ΔVtune is, the wider the
TR achieves. The importance of ΔVtune is described in
FOMTV depicted in 12) [11]. However, FOMTP and
FOMTV do not simultaneously account for ΔVtune and Pout
to be beneficial to VCO comparison. A new ΔVtune-added
and Pout-added FOM factor, called FOMTVP, is hence
proposed as 13).
oscTV
tune
VCO
1FOM PN 20log
10%
10log1mW
f TR
f V
P
(12)
oscTVP
tune
VCOout
1FOM PN 20log
10%
10log1mW
f TR
f V
PP
(13)
Table II shows the comparison of the performance of
the designed VCO in this paper with those of previously
reported VCOs in the Ku and X band. As can be seen
from Table I, The VCO reported in this work has an
excellent FOMTVP comparing with the other oscillators
processed in SiGe BiCMOS, GaAs PHEMT or CMOS
technology.
TABLE II: COMPARISON OF KU AND X VCOS
Ref. fosc
(GHz)
PN
(dBc/Hz)
△f
(MHz)
PVCO
(mW)
FOMTVP
(dB/Hz) Technology
[2] 22.7 -114 1 18 -150.9 SiGe HBT
[6] 32.55 -97 1 20 -140.6 SiGe
BiCMOS
[10] 24 -119.4 1 3.86 -165.3 CMOS
[12] 19 -112 1 200 -153.9 SiGe
BiCMOS
[13] 13.6 -116.2 1 3.78 -153.8 SiGe
BiCMOS
[14] 23.1 -94 1 2.5 -157.3 SiGe
BiCMOS
[15] 20.89 -97.2 1 40 -167.6 SiGe
BiCMOS
[16] 24.27 -100.3 1 7.8 -152.2 CMOS
[17] 19 -112 1 200 -140.5 SiGe
BiCMOS
[18] 31.6 -94 1 410 -130.3 GaAs
PHEMT
[19] 24 -119.4 1 3.86 -160.1 CMOS
[20] 24.3 -106.05 1 6.2 -158.3 CMOS
This
work 31.8 -114.6 1 1.8 -162.1
SiGe
BiCMOS
IV. CONCLUSION
A fully integrated 0.18 μm SiGe BiCMOS Ka-band
VCO, with cross-coupled pair configuration shows good
circuit performance, in terms of phase noise and low
power dissipation. This Ka-band VCO exhibits a -114.6
dBc/Hz phase noise at a 1 MHz offset and 3.31 GHz
tuning range. The FOM achieves -202.1 dBc/Hz.
ACKNOWLEDGMENT
This work was supported by the Advance Research
project of China (Grant No. 51307xxxx09), and the
Advance Research project of PLA in China
(KJ2013××××).
Journal of Communications Vol. 12, No. 2, February 2017
102©2017 Journal of Communications
(
(9) FOM depicted
(
(
(
BAND
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Yuhua Qi was born in Shishou, Hubei
province, China, in
1977. He received
the bachelor degree in Naval University
of Engineering, Wuhan, China, in
2000.
He received the master degree in Naval
University of Engineering, Wuhan,
China, in
2006. His research is focused
on information network technology,
modeling of HBTs and radio-frequency circuit design.
Rulong He was born in Hubei Province,
China, in 1977. He received the B.S.
degree from Naval University of
Engineering, PLA(NUEPLA), Wuhan,
in 2000 and the M.S. degree in 2006,
both in
communication engineering. His
research interests include submarine
cable communication, signal processing,
and information network theory.
Journal of Communications Vol. 12, No. 2, February 2017
103©2017 Journal of Communications
Zhao Shen was born in Hubei Province,
China, in 1980. He received the B.S.
degree from Naval University of
Engineering, PLA(NUEPLA), Wuhan,
in 2002 and the M.S. degree in 2005,
both in communication engineering. He
is currently pursuing the Ph.D. degree
with the School of Electronic
Information and Communications, Huazhong University of
Science and Technology(HUST), China. His research interests
include wireless communication, signal processing, and
information theory.
Journal of Communications Vol. 12, No. 2, February 2017
104©2017 Journal of Communications