Mitigating EMI of Powerline Communications
Using
Carrier-less UWB Pulses
Der Fakultät für Ingenieurwissenschaften der
Universität Duisburg-Essen
zur Erlangung des akademischen Grades eines
Doktors der Ingenieurwissenschaften
( Dr.-Ing. )
genehmigte Dissertation
von
M.Sc. Getahun Mekuria Kuma
aus Äthiopien
Referent: Prof. Dr.-Ing. Holger Hirsch
Korreferent: Prof. Dr. rer. nat. Achim Enders
Tag der mündlichen Prüfung: 2. September 2008
Acknowledgements
I thank the Lord my God with “all my heart and with all my soul and with all my mind” for his
boundless provisions, physical and spiritual, “…wisdom and power are His” Daniel 2:21.
“Therefore, I will praise you, O LORD,, ……I will sing praises to your ame.”2. Samuel 22:50
I am very fortunate to have Prof. Dr.-Ing. Holger Hirsch as my PhD supervisor. Since the time
I applied to do my research under your supervision and during each and every single day of my
stay at the Institute of Energietransport und -speicherung (ETS), dear Prof. Hirsch, your
willingness to help me, your wide ranges of theoretical and practical knowledge and the way
you bring light to any of my difficulties and problems have helped me hugely in bringing my
studies to where it stands today. You are not only my advisor; the friendly atmosphere you
have created with all of us is what makes the working environment at ETS exceptional. I am
greatly indebted to you, I simply say: thank you so much.
I would like to greatly thank Prof. Dr. rer. nat. Achim Enders from Institute of EMC, TU
Braunschweig, for willing to be my external examiner despite his extremely tight schedules .
Thank you, Prof. Enders, for giving me this chance to be my examiner and for your important
feedbacks.
I owe a great deal of thanks to Prof. Dr.-Ing. Peter Jung, Prof. Dr.-Ing. Axel Hunger and
Prof. Dr. rer. nat. Franz J. Tegude for giving me their precious time, for the discussions I have
had with each of them and for their willingness to be member of my PhD examination
committee.
Had it not been for the full financial support I have been receiving from Deutscher
Akademischer Austausch Dienst (DAAD), it would have been unthinkable to persuade my
PhD here in Germany. I am very indebted to DAAD and personally to Dr. Ronald Weiß, Mrs.
Dagmae Eckert and Mrs. Jennifer Schenk from Referat 413 (Afrika/Sub Sahara) for extending
the generosity of DAAD to me. Thank you so much.
My many thanks are also to each staff member of ETS. Each one of them has helped me
throughout my stay at ETS.
Dr.-Ing. Fekadu Shewarega and his family, thank you for encouraging me during my studies,
thank you and I owe you too much.
My many relatives and friends back home, my father and my mother, my brothers and my
sisters, all my friends whose wish and prayers are to see and to hear that I remain healthy, and
that I be successful in my studies, I say: thank you so much.
Last but by no means the least, I say thank you to my wife Dr. Solomie Jebessa. The days were
tough, but thank God they are gone.
Getahun Mekuria Kuma
September 2008,
Duisburg, Germany
- iv -
Table of Contents
Acknowledgements........................................................................................................ ii
Table of Contents ..........................................................................................................iv
Figures.........................................................................................................................vi
Tables ........................................................................................................................vii
Glossary and Acronyms .............................................................................................. viii
1. Introduction ....................................................................................................11
1.1. Introduction: Powerline Communication ............................................................................................ 11
1.2. Why Interference is an Important Subject in PLC .............................................................................. 13
1.3. The Ultra-Wideband............................................................................................................................ 14
1.4. Governing EMC Standards for PLC ................................................................................................... 16
1.5. Current Status of PLC Technology ..................................................................................................... 17
2. Characterization of a Powerline Channel .........................................................19
2.1. Modelling ............................................................................................................................................ 19
2.1.1. Non-branched Channel ............................................................................................................. 20
2.1.2. One-branched Channel ............................................................................................................. 21
2.1.3. General n-branched Channel .................................................................................................... 22
2.2. Simulation Results .............................................................................................................................. 23
2.2.1. Non-branched Channel ............................................................................................................. 24
2.2.2. One-branched Channel ............................................................................................................. 24
2.2.3. Two-Branched Channel............................................................................................................ 26
2.2.4. Three-Branched Channel.......................................................................................................... 28
2.2.5. Effect of Position and length of Branches ................................................................................ 29
2.3. Impulse Echo Characterization ........................................................................................................... 32
2.3.1. Modelling Reflection Types ..................................................................................................... 32
2.3.2. Localization of Strong Reflection Points.................................................................................. 35
2.3.3. Line Attenuations on Symmetrical and Asymmetrical Signals ................................................ 37
2.3.4. Effect of Distribution Board on Received Signal Amplitudes.................................................. 38
2.3.5. TCTL and LCTL ...................................................................................................................... 40
3. Transmission of UWB Pulses over Powerline Channel ......................................46
3.1. Formulation of the UWB Signal Pulses .............................................................................................. 46
3.1.1. Gaussian and its Derivative Pulses ........................................................................................... 46
3.1.2. Power Spectral Density (PSD) ................................................................................................. 49
3.2. Transmission of UWB Pulse Signals .................................................................................................. 50
3.2.1. Pulse Parameters....................................................................................................................... 50
3.2.2. Improving Reception ................................................................................................................ 51
- v -
3.2.3. Simulations............................................................................................................................... 53
3.2.4. Transmission Setup .................................................................................................................. 55
3.2.4.1. Reference Transmission.................................................................................. 55
3.2.4.2. Transmission on Test Bench ............................................................................ 56
4. Theoretical Analysis of Interferences from UWB Signals ..................................59
4.1. Radiated Power Loss from 2PWN ...................................................................................................... 59
4.2. Power Spectral Density of UWB Signals............................................................................................ 61
4.3. Effect of Modulation in Minimizing Spectral Lines ........................................................................... 63
4.3.1. Un-modulated pulses ................................................................................................................ 64
4.3.2. Modulated pulses...................................................................................................................... 65
4.3.2.1. Binary Phase Shift Keying (BPSK) ................................................................... 65
4.3.2.2. Amplitude Shift Keying (ASK) ........................................................................ 65
4.3.2.3. Pulse Position Modulation (PPM)..................................................................... 68
4.3.2.4. ON-OFF Keying (OOK) ................................................................................. 69
4.3.2.5. Other Alternatives ......................................................................................... 69
4.3.3. Carrier-based Transmissions .................................................................................................... 70
4.4. UWB Signals on a Narrow-band Receiver.......................................................................................... 71
4.5. Pulse width and Measurement frequency............................................................................................ 72
4.6. Low/High PRF Region, PDCF and Effective Duty Cycle .................................................................. 74
5. EMI Measurement Setups and Results .............................................................77
5.1. Measured Signal Spectrum.................................................................................................................. 77
5.2. EMI Measurement Setups ................................................................................................................... 83
5.2.1. Disturbance Voltage Measurement Setup ................................................................................ 83
5.2.2. Radiation Measurement Setup.................................................................................................. 85
5.3. Measurement Results .......................................................................................................................... 86
5.3.1. Disturbance Voltage Measurement Result ............................................................................... 86
5.3.2. Radiated Field Measurement Results ....................................................................................... 87
5.3.2.1. Measurement Points....................................................................................... 87
5.3.2.2. The Test-Bench ............................................................................................ 87
5.3.2.3. The Measurement Results ............................................................................... 87
6. Discussions and Conclusions ...........................................................................91
6.1. Discussions and Conclusions based on Results................................................................................... 91
6.2. Topic Proposals for related Future Researches ................................................................................... 93
References and Bibliography .........................................................................................95
- vi -
Figures
Figure 1 ABCD representation of a 2PWN ................................................................................................. 20
Figure 2 Non-branched Powerline channel ................................................................................................. 20
Figure 3 One-branched Powerline channel.................................................................................................. 21
Figure 4 General n -branched Powerline channel ....................................................................................... 23
Figure 5 Transfer Function and Impulse Response of a non-branched channel .......................................... 24
Figure 6 Transfer Function and Impulse Response of a channel with one 2 m branch ............................... 25
Figure 7 Transfer Function and Impulse Response of a channel with one 5 m opened branch................... 25
Figure 8 Transfer Function and Impulse Response of a channel with two 2 m branches............................ 27
Figure 9 Transfer Function and Impulse Response of a channel with two branches, 1 m and 3 m ............. 28
Figure 10 Transfer Function and Impulse Response of a channel with three branches ................................. 29
Figure 11 Impulse Response of three-branched channel for different branch length parameters.................. 30
Figure 12 Impulse Response of three-branched channel for different branch position parameters ............... 31
Figure 13 Measurement setup for (a) Group-1 and (b) Group-2 injections ................................................... 33
Figure 14 Modelling of echoes from Group-1 and Group-2 injections ......................................................... 33
Figure 15 Impulsive Input signals (a) Symmetrical and (b) Asymmetrical................................................... 34
Figure 16 Measurement Results from (a) Group-1 and (b) Group-2 injection types..................................... 35
Figure 17 Effect of Distribution Board on (a) Group-1 and (b) Group-2 injections...................................... 39
Figure 18 Measurement Setup for (a) TCTL and (b) LCTL as defined in ITU-T G.117 .............................. 41
Figure 19 Impulsive echo measurement results of (a) TCTL and (b) LCTL................................................. 42
Figure 20 Gaussian Pulse and its first four derivative pulses ........................................................................ 47
Figure 21 Fourier Transform of the pulses in Fig.20..................................................................................... 48
Figure 22 Plots of Autocorrelation functions of 1DGP and 2DGP................................................................ 49
Figure 23 Parameters for Pulse Design ......................................................................................................... 50
Figure 24 Representation of signals a PLC channel ...................................................................................... 51
Figure 25 Convolution of received and template pulses................................................................................ 51
Figure 26 Multiplication of received and template pulses............................................................................. 52
Figure 27 Possible circuitry to minimize effects of branches from received pulses. ..................................... 52
Figure 28 Received Pulses with and without the correction circuitry on the four channels .......................... 54
Figure 29 Measurement Configuration of UWB transmission over 2PWN .................................................. 55
Figure 30 Received 2DGP over a 1 m coax................................................................................................... 56
Figure 31 Received 2DGP over the different channels and effect of the proposed correction circuitry........ 57
Figure 32 Wave propagation to and from a 2PWN ....................................................................................... 60
Figure 33 Representation of the discrete signal pulse ................................................................................... 62
Figure 34 Format of M-Array PAM UWB modulation................................................................................. 64
Figure 35 Amplitude distribution example for 4-ASK .................................................................................. 66
Figure 36 Format of M-Array PPM UWB modulation ................................................................................. 68
Figure 37 Representation of OOK Modulation ............................................................................................. 69
Figure 38 Simulated plots of Continuous and Discrete PSDs of UWB pulses .............................................. 70
Figure 39 UWB pulses as seen over NB Receiver ........................................................................................ 71
- vii -
Figure 40 Representation of "Effective Time" .............................................................................................. 75
Figure 41 Measured spectrum comparisons at 4 MHz .................................................................................. 79
Figure 42 Measured spectrum comparisons at 8 MHz .................................................................................. 80
Figure 43 Measured spectrum comparisons at 12 MHz ................................................................................ 81
Figure 44 Re-plot of Fig.41(c), Fig.42(c), and Fig.43(c) with mathematically averaged curve .................... 82
Figure 45 Setup for Disturbance Voltage Measurement................................................................................ 84
Figure 46 Setup for Radiation Measurement ................................................................................................. 85
Figure 47 Measured Results of Disturbance Voltages from the different signals ......................................... 86
Figure 48 Electric Field from un-modulated sinusoidal and un-modulated UWB ........................................ 88
Figure 49 Electric Field from un-modulated sinusoidal and BPSK modulated UWB................................... 88
Figure 50 Electric Field from un-modulated UWB and BPSK modulated UWB ......................................... 89
Figure 51 Electric Field from BPSK modulated sinusoidal and BPSK modulated UWB............................. 89
Figure 52 Electric Fields from all the four transmissions.............................................................................. 90
Tables
Table 1 Parameter values for curves in Fig.16 .......................................................................................... 36
Table 2 Average Line Attenuations of Symmetrical and Asymmetrical signals....................................... 37
Table 3 Comparing the effect of DistBrd on received signals for the two Groups of injections............... 39
Table 4 Amplitude and Timing parameters for Symmetrical inputs and Asymmetrical outputs .............. 43
Table 5 Amplitude and Timing parameters for Asymmetrical inputs and Symmetrical outputs .............. 43
Table 6 TCTL Values in Time and Frequency domains ........................................................................... 44
Table 7 LCTL Values in Time and Frequency domains ........................................................................... 44
Table 8 Summary of the maximum spectral components of the different signals..................................... 83
- viii -
Glossary and Acronyms
1DGP 1st Derivative Gaussian Pulse
2DGP 2nd Derivative Gaussian Pulse
2PW- Two-port Wired Network
AFG Arbitrary Function Generator
ASK Amplitude Shift Keying
AV Average value of the electrical quantity measured (Voltage, Power)
AWG- Additive White Gaussian Noise
BALU- BALanced to Unbalanced signal convertor
BER Bit Error Rate
BPL Broadband Power Line
BPSK Binary Phase Shift Keying
BW Band Width
Cat-5 Category-5 UTP cable
CE-ELEC European Committee for Electrotechnical Standardization, abbreviated from its French name:
Comité Europée de Normalisation Electrotechnique
CISPR The Special International Committee on Radio Interferences, abbreviated from its French
name: Comité International Spécial des Perturbations Radioélectriques
CW Continuous Wave
DC Direct Current
DistBrd Distribution Board
DPO Digital Phosphor Oscilloscope
DS-UWB Direct Sequence UWB
EM Electromagnetic
EMC Electromagnetic Compatibility
EMI Electromagnetic Interference
ESD Energy Spectral Density
FCC Federal Communications Commissions of the USA
GHz Giga Hertz
ICT Information and Communication Technology
IF Intermediate Frequency
ITE Information Technology Equipment
kHz Kilo Hertz
LCL Longitudinal Conversion Loss
LCTL Longitudinal Conversion Transmission Loss
- ix -
MAC Media Access Layer Protocol
MB-OFDM Multi-band OFDM
MHz Mega Hertz
-B Narrow Band
-RZ Non-return-to-Zero
OFDM Orthogonal Frequency Division Multiplexing
OOK ON-OFF Keying
OPM Orthogonal Pulse Modulation
PAM Pulse Amplitude Modulation
PA- Personal Area Network
PDCF Pulse Desensitization Correction Factor
PHY Physical Layer Protocol
PK Peak value of the electrical quantity measured (Voltage, Power)
PLC Powerline Communication
PPM Pulse Position Modulation
PRF Pulse Repetition Frequency
PSD Power Spectral Density
PSK Phase Shift Keying
PVC Polyvinyl Chloride
QAM Quadrature Amplitude Modulation
QoS Quality of Service
QP Quai Peak value of the electrical quantity measured (Voltage, Power)
RBW Resolution Bandwidth
SA Spectrum Analyzer
S-R Signal-to-Noise Ratio
TCL Transverse Conversion Loss
TCTL Transverse Conversion Transmission Loss
T-IS- Telecommunications-port Impedance Stabilization Network
USB Universal Serial Bus
UTP Unshielded Twisted Pair
UWB Ultra Wide Band
WUSB Wireless USB
- x -
- 11 -
1. Introduction
Contents
• Introduction: Powerline Communication
• Why Interference is an Important Subject in PLC
• The Ultra-Wideband
• Governing EMC Standards for PLC
• Current Status of PLC Technology
1.1. Introduction: Powerline Communication
The Powerline Communication (PLC) is a technology which utilizes the existing electric
networks, both the distribution network outside and the installations inside buildings, for
broadband data communication over the band of frequencies up to 30 MHz without requiring
to install additional data cables. For many decades now electrical power utilities have been
using their power transmission and distribution infrastructure for data communications aimed
exclusively at monitoring and controlling transmission lines, optimizing power generations
distributed over an interconnected power grid, remote load connecting and disconnecting,
online kilo-watt-hour-meter (kWHM) reading, online tariff setting, and other related services
aimed at improving efficiency of the power grid and customer services. Utilization of the
electric network infrastructure as a medium for commercial broadband data communications,
however, is one of the recently evolving broadband technologies and is aimed at transforming
the electrical network to serve additionally as a communication network. The main idea behind
PLC is the reduction of cost and expenditure in the realization of new telecommunication
networks [HHAH04].
This double-faceted use of the electric network is, however, not without challenges and these
challenges are related to the different features of the power line network itself:
• It is a shared channel. The number of outlets on a single socket line inside a building,
for example, explains how much a given PLC channel is shared among the consumer
loads. These consumer loads can be switched ON or OFF any time while data
communications is taking place between nodes.
• The loading condition of the channel is highly fluctuating. The different consumer
devices connected or disconnected have wide range of load sizes and hence the
Chapter.1- Introduction
- 12 -
impedance of the network fluctuates continuously as these loads are switched ON or
OFF.
• The network is characterized by unsymmetries of wide ranges: mismatches between
characteristic impedance of the line and the connected loads, mismatches caused due to
branches, wires or cables of different sizes on the same channel, unsymmetries related
to un-patterned proximity to grounding points, presence of switch gears (Distribution
Boards), presence of noises that are impulsive in nature, etc.
These challenges, however, have never prevented utilizing electric network for broadband data
transmission within a building as a Local Area Network (LAN) channel and outside a building
as an access network.
Broad-band PLC technologies are generally implemented in the following different categories:
Access PLC
The electric supply system consists of High-Voltage (HV) networks, Medium-Voltage (MV)
networks and Low-Voltage (LV) networks. In most countries the voltage levels of the HV, the
MV, and the LV networks are 110-380 kV, 10-30 kV, and 230/240 V, respectively. These
networks are within the domain of responsibility of the utility companies owning and/or
managing the respective networks. It is additional utilization of these transmission and
distribution electric network as a communication infrastructure that is commonly categorized
as Access PLC.
Access PLC is to be used to bridge long distances to avoid building extra communication links
and is considered as one of the so called “last-mile” technologies comprising of Fiber optical
cable, Digital Subscriber’s Line (DSL), broadband cable and Fixed/Mobile broadband wireless
access networks. The telecom backbone is to be coupled at the MV front end (at the HV/MV
substation) or at the LV front end (at the distribution transformer) using either Optical fibers or
Satellite communications depending on the accessibility, cost, and Quality of Service (QoS)
[HHAH04].
In-House PLC
In-House PLC is the utilization of the electric network inside a building as a Local Area
Network (LAN) channel. This enables the configuration of LAN in a building without
requiring or installing any additional unshielded Twisted Pair (UTP) cables thereby the cost of
installation and all sorts of inconveniencies related to installing new cabling inside already
Chapter.1- Introduction
- 13 -
existing buildings are hugely reduced. With its currently emerging 200 Mbps modems,
simplicity of establishing communications between nodes, and its option for a wireless
extension of the PLC service through its PLC wireless Access Point (AP), In-House PLC is
undergoing promising transformations making it optimum solution to some of the limitations
of its competing technologies.
E-energy
This is an initiative for an Information and Communication Technology (ICT)-based energy
system of the future. This is expected and aimed at providing an efficient electric power
generations, transmissions, distributions and consumptions using the ICT. The current PLC
technology is, therefore, the basis for providing an optimized service in the energy sector.
1.2. Why Interference is an Important Subject in PLC
What makes the subject of Electromagnetic Interference (EMI) an important issue in the PLC
technology can be explained very easily through discussing what is not part of the PLC channel
but part of the other communication channels for minimizing EMI and making them highly
immune to disturbances from other external sources.
Twisted-Pair Channel:
The two wires of a pair inside a standard telecommunication cable are twisted across the whole
cable length. The radiated field at a given radial distance due to an electric current +I inside
one wire of a pair is fully compensated by an equal and opposite field due to a current −I in the
other wire of that same pair. Additionally, the channel is not shared among multiple users and
hence the channel experiences relatively uniform impedance characterization during the period
of broadband data transmission. Disturbance voltage from external sources induced on one of
the wires is also fully compensated by an equal but opposite voltage induced on the other wire
of the same pair.
Broadband Cable:
The coaxial cable of Cable-TV channel used for broadband Cable is primarily intended to carry
TV signals in the range of hundreds of MHz frequencies, which is much higher than the current
broadband PLC frequency of operation. Therefore, these cables had been primarily intended to
carry signals of much higher frequencies without major degradation in providing minimum
Chapter.1- Introduction
- 14 -
interference to the environment. In addition to carrying the returning current, the major purpose
of the braided copper sheath between the inner dielectric insulator and the outer plastic sheath
is also to provide electromagnetic “blanket” to protect the radiation of the EM field from the
inner conductor to the environment and also to protect disturbance voltages from external
sources to the signal carrying inner conductor. The characteristic impedance of the cable is also
designed to provide a reflection-free transmission as much as possible.
PLC Channel:
The parallel-wire transmission channel of the electric network has only a slight twist (for
mechanical reasons) and typically has no protection sheath one finds in other channels that are
primarily intended for minimizing interferences and maximizing immunity at broad-band
frequency of operations. This is, therefore, one of the challenges facing the PLC technology: to
minimize EMI and to maximize its immunity to external disturbances. Additionally, the
existing governing standards for the EM emissions and susceptibility limit lines have been set
for those technologies and channels equipped with the features discussed earlier for minimized
EMI and maximized immunity and as a “new-comer” the PLC is also faced with the
requirement of meeting those limit lines. Even though currently there are discussions and
deliberations taking place to straighten these EMC related unfair treatment facing the PLC the
technology the PLC has also proved itself to be robust enough to win the challenges and has
since long already become marketable in countries with strict EMC regulations.
1.3. The Ultra-Wideband
Definition
The FCC 15.503 (d) defines Ultra Wideband (UWB) signal as a signal that satisfies either of
the following two conditions:
• Signal band width (BW) of 500 MHz, or
• Fractional BW of 20%
Fractional BW, here represented as bwf , is in turn defined in FCC 15.503 (c) as:
c
lh
bwf
fff
−= (1.1)
Where:
Chapter.1- Introduction
- 15 -
cf is the center frequency at which the signal has the maximum power
emission
hf and
lf are the higher and lower frequencies, respectively, at which the power
emission of the signal is 10 dB below the maximum emission.
These two requirements are not related to each other and signals can fulfill either of the two
conditions based on areas of applications of the signals.
Since the introduction of the UWB technology, however, it has been widely implemented for
wireless applications with the 500 MHz requirement due to the assignment of the 3.1 GHz to
10.6 GHz band for wireless applications without requiring any license.
The basics behind the UWB technology, which is thought to have evolved from classical high-
power pulse transmissions for radar applications is based on exploiting the advantage of a
wider BW that comes from transmission of a narrow pulse and both approaches of UWB
realizations have now evolved in to two competing technologies in the wireless applications
and each has its own advantages and disadvantages. The two major industry alliances
promoting these different approaches of wireless UWB implementations are the WiMedia
Industry Alliance promoting the Multi-band OFDM (MB-OFDM) [WiMedia] and the UWB-
Forum promoting the carrier-less and pulse-based UWB implementation based on Direct
Sequence technology (DS-UWB) [UWBFor]. Each technology has established its own PHY
and MAC layer protocols and it is yet early to be sure which technology dominates future
wireless UWB implementations. Despite the differences, however, the central point of UWB
transmission is a minimized interference between co-existing transmissions at very high data
rate.
Improved Channel Capacity
Shannon’s equation shows that the Channel Capacity (C ) increases as a function of BW faster
than as a function of Signal-to-Noise ratio ( SR )
)1(log. 2 SRBWC += (1.2)
Where the SR can again be expressed in terms of the transmitted signal power ( P ) and the
noise power spectral density (o
) as:
oBW
PSR
.= (1.3)
Chapter.1- Introduction
- 16 -
Therefore, (1.2) and (1.3) show that a given amount of increase in the channel capacity requires
either a near-linear increase in the bandwidth or an exponential increase in the transmitted
signal power.
It is also possible to re-write (1.3) in terms of the signal Power Spectral Density, PSD , as
follows:
o
PSDSR = (1.4)
This shows that, for a constant transmitted total power an increase in the BW of a signal
linearly decreases the PSD. As will be shown in Chapter 4.1, the electromagnetic radiation
from a channel in the case of radio protection (which is addressed in standards) is directly
proportional to the signal PSD rather than the transmitted total signal power.
Therefore, here is the advantage of UWB: increasing the BW linearly increases the channel
capacity and linearly decreases the radiation interference without changing the transmitted total
power.
1.4. Governing EMC Standards for PLC
The major EMC related standards for the PLC implementations are the following two:
• CISPR-22: Information Technology Equipment (ITE), Radio Disturbance
Characteristics, limits and methods of measurement. Like any other ITE, a PLC device falls
in the scope of this standard. CISPR 22 distinguishes the frequency range below 30 MHz,
where emissions are measured as conducted signals and the frequency range above 30 MHz
where measurements of the disturbance field strength are performed with antennas. Since
current PLC technologies are limited to frequencies below 30 MHz conducted
measurements and related limits should be performed according to the CISPR 22
philosophy. However, the methods described in CISPR 22 for the conducted emissions are
not yet suitable for PLC, because they either focus on mains terminals (use of measurement
networks that blocks high frequencies) or on telecommunication terminals (use of
measurement networks that are not designed for higher voltages).
• FCC-Part 15: The PLC is commonly known as Broadband Powerline (BPL) in the US
and the FCC 15 conduction and radiation limits have been amended to incorporate the BPL
interference issue. The Public Notice by FCC on August 3, 2006 states the possibility of
Chapter.1- Introduction
- 17 -
co-existence between BPL and other communication systems by imposing a reduction of
emissions by 20 dB below the normal Part 15 emission limits to provide adequate
interference protection for radio systems and also authorizing deployment of BPL
[FCCA06].
1.5. Current Status of PLC Technology
Currently there are different international and national Working Groups (WG) mandated to
undertake designing and specifying the different Layers involved in the PLC communication
system and the EMC related matters of the PLC. Some of the major working groups are:
• IEEE: The following WGs under the IEEE take care of their particular areas of
activities:
o IEEE-P1675: is responsible to the specifications and standards related to the
PLC Hardware installations and Safety issues.
o IEEE-P1775: is the IEEE WG responsible to the EMC requirements-Testing
and Measurement methods of the PLC.
o IEEE-P1901: is the IEEE WG working on the Media Access Control (MAC)
and the PHYsical (PHY) layer protocols of the PLC.
• OPERA: the “Open PLC European Research Alliance”, is a research alliance
funded from the European Union (EU) dedicated to the improving the existing, and
developing new, PLC specifications and standards. The different sub-groups of OPERA
deal with the whole spectrum of the PLC technologies: MAC and PHY Layers, EMC
related matters, Routing Protocols, Hardware, Safety, and a wide range of subjects.
OPERA works on developing new business models that integrate the PLC technology to
other existing technologies [Opera].
• HomePlug: is an Alliance of trade groups consisting of companies involved
in the development, specifications and manufacturing of PLC products and services. The
different HomePlug product specifications include the 14 Mbps HomePlug 1.0 PLC
modem and the HomePlug AV PLC modems intended for HDTV streaming over the PLC
channel.
Chapter.1- Introduction
- 18 -
• ETSI PLT is the PLC working Group under the European
Telecommunications Standards Institute (ETSI). ETSI is an independent European based
institute widely known fore its successful standardization of the Global System for Mobile
communication (GSM). ETSI-PLT is working on harmonized standards taking care of
conformity of the PLC technology with other EU Directives. This is done by studying the
technical requirements to avoid interferences with users of the radio spectrum [Etsi].
• CISPR: CISPR/I/PLT CISPR-22-PLT, Limits and methods of measurement of
broadband telecommunication equipment over power lines, revises the available CISPR 22
measurement methods intended for ITE to accommodate the special condition of the PLC
scenario.
• CE-ELEC: The sub-committee SC205A, Mains Communicating Systems,
covers the 3 kHz-148.5 kHz for home and building control applications and for utility
remote metering [SC205A], and the technical committee TC205A, Home and Building
Electronic System (HBES), covers the PLC frequency band of up to 30 MHz for various
electronic devices that are used in homes, buildings and similar environments [TC205A].
The current PLC PHYsical Layer specifications that are widely used for PLC modem design
are the following [HPAV05]:
• Frequency Band: 2 - 28 MHz
• PHY Layer data rate 200 Mbps
• Modulation OFDM
• Number of usable carriers 917
• Sub-carrier modulations varies from BPSK to 1024 QAM
- 19 -
2. Characterization of a Powerline Channel
Contents
• Modelling
o Non-branched Channel o One-branched Channel o General n-branched Channel
• Simulation Results
o Non-branched Channel o One-branched Channel o Two-branched Channel o Three-branched Channel
• Impulse Echo Characterization
o Modelling Reflection Types o Localization of Strong Reflection Points o Line Attenuation of Symmetrical and Asymmetrical Signals o TCTL and LCTL
2.1. Modelling
Modelling and characterizing the different properties of a given communication channel is one
of the primary and important properties to be done before transmitting information over the
channel. Such investigations help to understand the possible domains of the transfer function,
the impulse response, symmetry, etc, of the channel under different conceivable scenarios to
which the channel may be subjected during the actual data transmission. The PLC channel can
be modeled as a multi-path channel and can be expressed in terms of the weighting
coefficients, attenuation coefficients and delay parameters as shown in [HHAH04],
[MZKD99]. The PLC channel can also be modeled as any two-port channel through ABCD
parameters taking care of branching parameters in the modelling, and it is this approach that is
applied here.
Let us consider a 2-Port-Wired-Network (2PWN) channel represented by its ABCD matrix
parameters as shown in Fig.1. The network parameters can be expressed as:
221 BIAVV += (2.1a)
221 DICVI += (2.1b)
=
2
2
1
1
I
V
DC
BA
I
V (2.1c)
Chapter.2- Characterization of a Powerline Channel
- 20 -
Figure 1 ABCD representation of a 2PWN
For a given general network, the ABCD parameters have the following special relations:
DA = (2.2a)
1=− BCAD (2.2b)
Consider the case where Port-1 is connected to a source of internal resistance of s
R and Port-2
is connected to a load of L
R . Then, the matrix relation of (2.1c) can be re-written as :
=
−
L
ss
RV
V
DC
BA
I
RIV
/2
2
1
1 (2.2c)
Rearranging and solving for the voltage Transfer Function, one gets:
SSLL
L
RfDRRfCfBRfA
RfH
)()()()()(
+++= (2.3)
And then the impulse response of the line can be solved from:
∫= dftfjfHth )2exp()()( π (2.4)
Equation (2.3) and (2.4) show that the transfer function and the impulse response can be easily
computed if the ABCD matrix parameters of the network are known. The advantage of the
ABCD chain matrix lies in its convenience in handling computations of multiple and cascaded
sections of a given channel. The Powerline channel is one of such channels in which each
branch and each segment along the channel length is to be treated as a separate channel unit
and then cascaded together to compute the total channel matrix coefficients.
2.1.1. -on-branched Channel
Fig.2 shows the representation of a non-branched channel of length l with the source, source
and load impedance as shown.
Figure 2 Non-branched Powerline channel
Chapter.2- Characterization of a Powerline Channel
- 21 -
Its ABCD chain matrix can be expressed as [TEFK03], [GMHH07]:
( ) ( )( ) ( )
=
ll
ll
γγ
γγ
coshsinh1
sinhcosh
C
C
Z
Z
DC
BA (2.5)
Where: c
Z is the Characteristic Impedance of the line, and
γ is the Propagation constant of the line:
The Characteristic Impedance and the Propagation Constant are both frequency dependent
parameters and can be solved from the following relations:
CfjG
LfjRfZ
C ππ
2
2)(
+
+= (2.6)
( ) ( )CfjGLfjRf ππγ 22)( +⋅+= (2.7)
Where: LGR ,, and C are the per unit resistance, conductance, inductance and capacitance of
the line, respectively.
Then, the transfer function of (2.3) for a non-branched channel can be written as:
( )( ) ( )( ) ( ) ( )( )SSL
C
CL
L
RlRRlZ
lZRl
RfH
γγγγ coshsinh1
sinhcosh
)(
+
++
= 2.8)
And the impulse response can then be computed from (2.4):
Therefore, (2.8) shows that the transfer function (and also the impulse response) of a non-
branched Powerline channel is influenced by the Characteristic Impedance, the Propagation
Constant, the length of the line; the source and the load impedances.
2.1.2. One-branched Channel
Next, consider the same channel of length l but has a branch of length 1brl at a distance x from
the source. The branch is connected to a load 1brR as shown in Fig.3 below:
Figure 3 One-branched Powerline channel
Chapter.2- Characterization of a Powerline Channel
- 22 -
For the sake of computation of the ABCD matrix, the channel is segmented into the following
three Segments:
Segment-1: The segment of length x between the source and the branch
Segment-2: The branch of length 1brl
Segment-3: The segment of length xl − between the branch and the load
The ABCD matrices of the three Segments are:
( ) ( )( ) ( )
=
xx
Z
xZx
DC
BA
C
C
chch
chch
γγ
γγ
coshsinh1
sinhcosh
1_1_
1_1_ , for Segment-1 (2.9a)
=
1
101
1_1_1_
1_1_
eqbrbr
brbr
ZDC
BA,for Segment-2 (2.9b)
( )( ) ( )( )( )( ) ( )( )
−−
−−=
xlxl
Z
xlZxl
DC
BA
C
C
chch
chch
γγ
γγ
coshsinh1
sinhcosh
2_2_
2_2_ , for Segment-3 (2.9c)
Where the term 1_eq
Z can be computed from the following relation [TEFK03]:
)tanh(
)tanh(
11
111_
brbrC
brCbr
CeqlRZ
lZRZZ
γγ
++
⋅= (2.10)
And the ABCD matrix of the whole channel is then the cascaded matrix multiplications of the
three matrices:
=
2_2_
2_2_
1_1_
1_1_
1_1_
1_1_
chch
chch
brbr
brbr
chch
chch
DC
BA
DC
BA
DC
BA
DC
BA (2.11)
Then, the transfer function and the impulse response of the channel are computed from (2.3)
and (2.4).
2.1.3. General n-branched Channel
Consider a general n branched channel as shown in Fig.4. For a general n -branched channel:
i. the ABCD matrix of segment i of the channel, ]1,1[ +∈ ni , between the thi )1( − branch
and the thi branch is given by:
( ) ( )
( ) ( )
−−
−−
=
−−
−−
)(cosh)(sinh1
)(sinh)(cosh
11
11
__
__
iiii
c
iicii
ichich
ichich
xxxxZ
xxZxx
DC
BA
γγ
γγ
(2.12)
Where:
Chapter.2- Characterization of a Powerline Channel
- 23 -
ix is the position of branch i from the source
00=x and lx
n=
+1
ii. The ABCD matrix of branch j , ],1[ nj ∈ , can also be computed from:
=
01
01
_
__
__
jeq
jbrjbr
jbrjbr
Z
DC
BA
(2.13)
Where
)tanh(
)tanh(
__
__
_
jbrjbrC
jbrCjbr
CjeqlRZ
lZRZZ
γ
γ
+
+⋅=
jbr
R _ is the load connected to branch j
jbr
l _ is the length of branch j
lbr_1
lbr_2
lbr_n
Figure 4 General n -branched Powerline channel
Hence, the ABCD matrix of the complete channel can be computed from:
=
++
++
=∏
1_1_
1_1_
1
__
__
__
__
nchnch
nchnchn
i
ibribr
ibribr
ichich
ichich
DC
BA
DC
BA
DC
BA
DC
BA
(2.14)
2.2. Simulation Results
The different mathematical formulations in the previous Section have been analyzed using
Matlab to simulate the Transfer Function and the Impulse Response of the different channel
configurations. The conductor used in the simulation is the standard 2.5 sq.mm PVC insulated
copper conductor.
Chapter.2- Characterization of a Powerline Channel
- 24 -
2.2.1. -on-branched Channel
The simulated non-branched channel is a 20 m cable connected to a load matched to the
characteristic impedance of the line. Fig.5 shows the Transfer Function and magnitude of the
impulse response.
0 20 40 60 80 100-5
-4.8
-4.6
-4.4
-4.2
-4
-3.8
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.1
0.2
0.3
0.4
0.5
0.6
0.7
Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
Figure 5 Transfer Function and Impulse Response of a non-branched channel
As expected, the Impulse Response does not have any echo depicting the ideal condition where
there are no branches and that the load connected to the channel is a perfectly matched.
2.2.2. One-branched Channel
This is the same channel as that of Section 2.2.1 but with a 2 m branch located at the middle of
the 20 m channel. The physical and electrical characteristics of the branch remain the same and
the results for the simulation are shown in Fig.6. The simulation was repeated for a 5 m branch
and the result is as shown in Fig.7 except that only the simulation result for the opened branch
is given in Fig.7 to avoid repetition.
0 20 40 60 80 100-50
-40
-30
-20
-10
0
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.1
0.2
0.3
0.4
0.5
Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
Chapter.2- Characterization of a Powerline Channel
- 25 -
0 20 40 60 80 100-50
-40
-30
-20
-10
0
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.1
0.2
0.3
0.4
Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
0 20 40 60 80 100-50
-40
-30
-20
-10
0
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.1
0.2
0.3
0.4
0.5
Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
Figure 6 Transfer Function and Impulse Response of a channel with one 2 m branch (a) opened (b) shorted and (c) matched
0 20 40 60 80 100-50
-40
-30
-20
-10
0
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.1
0.2
0.3
0.4
Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
Figure 7 Transfer Function and Impulse Response of a channel with one 5 m opened branch
The following properties can be seen from Fig.6 and Fig.7:
i. The time spacing between the direct signal and the echo of the non-matched channels
is directly related to the length of the branch despite whether the branch is open
circuited or short-circuited.
Chapter.2- Characterization of a Powerline Channel
- 26 -
ii. In all the three cases the magnitude of the direct signals are equal, implying that it does
not depend on the loading condition of the branch.
iii. Comparing the magnitude of the impulse response of Fig.6(a) and Fig.7, it can be seen
that the magnitude of the impulse response is independent of the length of the branch.
iv. As expected, frequency of oscillation in the transfer function is related to twice the
branch length and the dielectric property of the cable.
2.2.3. Two-Branched Channel
The same 20 m channel is investigated again for a two-branched case. The first branch is 2 m
long and is located at 25% of channel length from the source. The second branch is also 2 m
long but located at 75% of channel length from the source. Fig.8 shows the result of the
simulation. The simulation was repeated for branches of different lengths (1 m and 3 m) and
results are shown in Fig.9.
0 20 40 60 80 100-100
-80
-60
-40
-20
0
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.1
0.2
0.3
0.4
Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
(a) Open
0 20 40 60 80 100-100
-80
-60
-40
-20
0
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.1
0.2
0.3
0.4
Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
Chapter.2- Characterization of a Powerline Channel
- 27 -
0 20 40 60 80 100-50
-40
-30
-20
-10
0
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.05
0.1
0.15
0.2
0.25
0.3
0.35
Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
Figure 8 Transfer Function and Impulse Response of a channel with two 2 m branches (a) opened (b) shorted and (c) matched
0 20 40 60 80 100-100
-80
-60
-40
-20
0
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.05
0.1
0.15
0.2
0.25
0.3
0.35Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
0 20 40 60 80 100-100
-80
-60
-40
-20
0
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.05
0.1
0.15
0.2
0.25
0.3
0.35
Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
Chapter.2- Characterization of a Powerline Channel
- 28 -
0 20 40 60 80 100-50
-40
-30
-20
-10
0
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.05
0.1
0.15
0.2
0.25
0.3
0.35
Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
Figure 9 Transfer Function and Impulse Response of a channel with two branches, 1 m and 3 m, (a) opened (b) shorted and (c) matched
The following conclusions can be made based on Fig.8 and Fig.9
i. The echoes from equal-length branches are superimposed and hence produce an echo
stronger than the direct signal. This may lead to a wrong conclusion in making a
distinction between the two.
ii. Similar to what was said previously for a one-branched channel, varying the branch
lengths does not affect the magnitude of the direct signal response.
2.2.4. Three-Branched Channel
The number of branches is further increased to three. This time simulation of equal-length
branches are not shown here to avoid the problem related to equal-length branches as shown in
Fig.8 (a, b). The first branch is 1 m long and is located at 25% of channel length from source,
the second is 4 m long and located at 50%, and the third is 2 m long and is located at 75%.
0 20 40 60 80 100-100
-80
-60
-40
-20
0
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.05
0.1
0.15
0.2
Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
Chapter.2- Characterization of a Powerline Channel
- 29 -
0 20 40 60 80 100-100
-80
-60
-40
-20
0
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.05
0.1
0.15
0.2
Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
0 20 40 60 80 100-50
-40
-30
-20
-10
0
20.log(|H(f)|)
Frequency [ MHz ]
Transfer Function of the Channel
0 50 100 150 200 250 3000
0.05
0.1
0.15
0.18
Voltage [ V ]
Time [ ns ]
Magnitude of the Impulse Response
Figure 10 Transfer Function and Impulse Response of a channel with three branches, 1 m, 4 m and 2 m, (a) opened (b) shorted and (c) Matched
2.2.5. Effect of Position and length of Branches
Effects of the position and length of branches on the impulse response of the channel are
further analyzed by varying the branch position and the branch length parameters for the three-
branched channel. Fig.11 shows simulated results of the impulse response of the channel under
different combinations of branch lengths. Fig.12 also shows results when the position of the
branches are varied across the channel length. The indicated branch positions in Fig.12 are the
respective positions of the three branches from the input port expressed as a percentage of the
channel length.
Based on results of Fig.10 above and Fig.11 and Fig.12, the following conclusions can be made
for the three-branched PLC channels:
i. As discussed in Section 2.2.2 and Section 2.2.3 the amplitude of the direct pulse is
independent of both branch length and branch position.
ii. Amplitude of the direct pulse is independent of the loading conditions of the branches.
iii. Branches of equal lengths can possibly produce echoes much stronger than the
magnitude of the direct pulse.
Chapter.2- Characterization of a Powerline Channel
- 30 -
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0 5e-008 1e-007 1.5e-007 2e-007 2.5e-007 3e-007
Voltage [ V ]
(a) Shorted branches
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0 5e-008 1e-007 1.5e-007 2e-007 2.5e-007 3e-007
Voltage [ V ]
2m, 6m, 3m
5m,16m, 3m
2m, 2m, 2m
3m,46m, 2m
Time [ sec ]
(b) Opened Branches
(b) Opened branches
0
0.05
0.1
0.15
0.2
0.25
0.3
0.35
0.4
0 5e-008 1e-007 1.5e-007 2e-007 2.5e-007 3e-007
Voltage [ V ]
(c) Matched branches
Figure 11 Impulse Response of three-branched channel for different branch length parameters
Chapter.2- Characterization of a Powerline Channel
- 31 -
0
0.05
0.1
0.15
0.2
0.25
0.3
0.35
0 5e-008 1e-007 1.5e-007 2e-007 2.5e-007 3e-007
Voltage [ V ]
(a) Shorted branches
0
0.05
0.1
0.15
0.2
0.25
0.3
0.35
0.4
0 5e-008 1e-007 1.5e-007 2e-007 2.5e-007 3e-007
Voltage [ V ]
(b) Opened branches
0
0.05
0.1
0.15
0.2
0.25
0.3
0.35
0.4
0 5e-008 1e-007 1.5e-007 2e-007 2.5e-007 3e-007
(c) Matched branches
Figure 12 Impulse Response of three-branched channel for different branch position parameters
Chapter.2- Characterization of a Powerline Channel
- 32 -
2.3. Impulse Echo Characterization
In a real Powerline channel, there are other properties that need to be known in order to
characterize the unsymmetries with in the channel. There are always properties that can not be
measured or analyzed in the commonly done frequency-domain characterization of the
Powerline channel and therefore they need to be characterized in the time domain. Among
these properties are location of strong reflections, magnitude of these reflected waves, locations
of strong symmetrical-to-asymmetrical (and vice versa) signal conversions (or from
Differential Mode signal to Common Mode signal and vice versa). In addition to the transfer
function and impulse response discussed earlier, these additional characterizations of the
Powerline channel provide a better view into the channel to identify locations of strong
unsymmetries inside the network.
2.3.1. Modelling Reflection Types
To make this investigation, impulsive signals were injected to the Powerline channels at
different injection points across the channel. The magnitude of the direct pulse, the magnitude
of the echoes, the position in time of the echoes and the way these echoes from multiple
injections are aligned with respect to each other give important information in localizing the
source of these reflections with the network. For the purpose of analysis, the signal injection
points are classified into two groups, Group-1 and Group-2 as shown in Fig.13 (a, b).
• Group-1: Three signal injection points (P-1, P-2, P-3) are identified as injection
points on the same circuit line with the measurement point inside a
complex electrical network. As modelled in Fig.13 (a) such circuits are
assumed to have strong reflections coming from locations outside the
transmission path. Such reflection points are modeled as Type-A in this
Thesis.
• Group-2: Similarly, three points (P-4, P-5, P-6) are identified as injection points
on a different circuit line from the circuit line of measurement point.
These injections are assumed to be characterized by strong reflections of
Type-B or Type-C as shown in Fig.13 (b). Type-B reflection is that
which comes from a strong un-symmetry between the injection and
measurement points, whereas Type-C reflection is that which comes
from the other side of the measurement point as shown in Fig.13(b).
Chapter.2- Characterization of a Powerline Channel
- 33 -
Measurements were performed in an office building according to the measurement setup of
Fig.13. The expected reflection Types for the two injection Groups are also shown [GMHH08]:
Figure 13 Measurement setup for (a) Group-1 and (b) Group-2 injections
Additionally, the types of echoes these two Groups of injections produce are assumed to have
the forms shown in Fig.14, staggered echoes from Group-1 and overlapping echoes from
Group-2 injections.
Figure 14 Modelling of echoes from Group-1 and Group-2 injections
Chapter.2- Characterization of a Powerline Channel
- 34 -
It is also important to note that even though reflections from both points B and C of Fig.13 (b)
similarly produce overlapping echoes, a reverse measurement (interchanging the injection and
measurement points) helps to identify if the reflection is of Type-B or Type-C. If in the reverse
measurement the echoes are staggered similar to that of Group-1 injections, then it is of Type-
C and if the echoes still remain overlapped, then they are of Type-B and therefore the network
is assumed to have strong reflections of either Type-A or Type-B or both. The input impulsive
signals used for the measurements are shown in Fig.15 (a and b) [GMHH08].
-20
0
20
40
60
80
100
-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006
Voltage [ V ]
-15
-10
-5
0
5
10
15
20
25
30
-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006
Voltage [ V ]
Figure 15 Impulsive Input signals (a) Symmetrical and (b) Asymmetrical
Chapter.2- Characterization of a Powerline Channel
- 35 -
2.3.2. Localization of Strong Reflection Points
Measurements were performed as shown in the setup of Fig.13 and the results from the two
Groups of injections are shown in Fig.16 (a and b). The parameters in the Fig.16 are to be
understood as follows:
• x
V : is the received direct signal output at the measurement point.
• ss
V : is the measured echoes after part of the signal is reflected by a nearby strong
reflection point.
• sst : is the time delay between
xV and
ssV .
-10
-5
0
5
10
15
20
25
30
-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006
vss1
vss2
vss3
vx1
vx2
vx3
Voltage [ V ]
vx4
vx5
vx6
-3
-2
-1
0
1
2
3
4
-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006
vss5
vss4
vss6
Voltage [ V ]
Figure 16 Measurement Results from (a) Group-1 and (b) Group-2 injection types
Chapter.2- Characterization of a Powerline Channel
- 36 -
The delay time sst between the direct signal pulse and the different echoes of each curve are
further translated into a length ss
l of a polyethylene insulated conductor (rε = 6) to localize the
sources of the echoes as shown in Table 1. The measured values of the other parameters of the
curves are also tabulated [GMHH08].
Table 1 Parameter values for curves in Fig.16
Points x
V [ V ] ss
V [ V ] sst [µs]
ssl [m]
P-1 27.32 -8.60 0.326 19.96
P-2 29.64 -9.00 0.288 17.63
Gro
up-1
P-3 26.60 -9.80 0.254 15.55
P-4 2.36 -1.78 0.382 23.39
P-5 2.33 -2.22 0.408 24.98
Gro
up-2
P-6 3.53 -2.14 0.386 23.64
The different values of the length ss
l in the Table are to be interpreted as follows:
i. The short delay time from Point P-3 (0.254 µs) compared to that of P-1 and P-2 shows
that P-3 is nearer to the strong reflection point than P-1 or P-2 (Type-A reflection) and
therefore, point A is found at
o 15.55 m from P-3
o 17.63 m from P-2 and at
o 19.96 m from P-1.
ii. The soundness of this localization is further substantiated by the physical spacing
between the three points:
o Points P-1 and P-2 are physically spaced 2.32 m apart,
o Point P-2 and P-3 are physically spaced 2.09 m apart.
iii. From the Table,
o Points P-1 and P-2 19.96 – 17.63 = 2.33 m
o Points P-2 and P-3 17.63 – 15.55 = 2.08 m are in a good
harmony with the physical spacing between the three points.
iv. For Group-2 injections, strong reflections are coming from 23.39 m, 24.98 m, and
23.64 m, respectively, from the measuring point. Apart from a minor deviation from P-
5, which can be attributed to measurement uncertainties, these three points show that
the source of strong reflections is seems to be located at about 23.5 m from the
measurement point.
Chapter.2- Characterization of a Powerline Channel
- 37 -
v. By physically inspecting these locations obtained from the echoes (15.55 m from P-3
and about 23.5 m from the measurement point), it is the electrical Distribution Board
of the office building that is located there. This has given the strong impression that
even though the electrical circuit lines inside the office building were connected to
many consumer loads such as office computers, Printers, Copier, and other electrical
devices during the measurement, it is the Distribution Board that was found to have
been the source of strong unsymmetries in the network causing strong reflections.
2.3.3. Line Attenuations on Symmetrical and Asymmetrical Signals
In this experiment the computation of the line attenuations of Symmetrical and Asymmetrical
signals in the network was analyzed as follows:
i. by computing the ratio of the received pulse to the transmitted pulse in time domain,
and
ii. by converting the time domain inputs and outputs to the frequency domain and taking
statistical average of the ratios over the entire frequency spectrum.
Table 2 shows the summary of these results.
Table 2 Average Line Attenuations of Symmetrical and Asymmetrical signals
Line Attenuation [ dB ]
Symmetrical signals Asymmetrical signals
Inject.
Points
Time Domain Freq. Domain Time Domain Freq. Domain
P-1 10 5-10 12 10-15
P-2 10 5-10 14 10-15
Gro
up-1
P-3 10 5-10 14 10-15
P-4 32 25-30 42 35-40
P-5 32 25-30 44 40-45
Gro
up-2
P-6 28 25-30 40 35-40
The following conclusions can be made based on Table 2 [GMHH08]:
i. Asymmetrical signals are attenuated much stronger than Symmetrical signals, both for
Group-1 and Group-2 injections.
ii. Even though the values in Table 2 are typical to the condition at the measurement site,
Group-2 injection points generally experience much stronger attenuation of signals
Chapter.2- Characterization of a Powerline Channel
- 38 -
than Group-1 injections both for Symmetrical and the Asymmetrical signals. This fact
was re-analyzed by repeating the measurements on a different phase with different
branching topology, and the result remained the same.
iii. There is a good harmony between the transfer function values computed in the time-
domain and in the frequency domain, even though the time-domain values fall at the
lower part of the range specified by the frequency domain.
2.3.4. Effect of Distribution Board on Received Signal Amplitudes
As indicated in the previous section, the difference between the line attenuations of signals
from Group-1 and Group-2 is found to have been very considerable and hence the effect of the
Distribution Board (DistBrd), which is found out to be the source of strong reflections, is
further investigated as per the following two scenarios:
i. Scenario-1 The DB is fully operational, the injection and the measurement circuit
lines and all other circuit lines powered from the DistBrd are also operational with all
their consumer loads. The measured output under this scenario is represented as
Amplitude-I in Fig.16 for both Group-1 and Group-2 injections.
ii. Scenario-2 The feeder line to the DistBrd is disconnected, all circuits powered from
the DistBrd are disconnected and connection is established only between the injection
and measurement points. This requires no connection to the DistBrd for Group-1
injections but requires connection of the two circuit lines (the injection circuit line and
the measurement circuit line) for the case of Group-2 injections. Measurement results
of this scenario are represented as Amplitude-II in Fig.17.
Fig.17 shows only sample measurement outputs from both Groups (P-1 from Group-1 and P-4
from Group-2) since same-Group points showed almost similar results. The values of
Amplitude-I and Amplitude-II for both cases are tabulated in Table 3.
Chapter.2- Characterization of a Powerline Channel
- 39 -
-15
-10
-5
0
5
10
15
20
25
30
35
-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006
-10
-5
0
5
10
15
-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006
Voltage [ V ]
Figure 17 Effect of Distribution Board on (a) Group-1 and (b) Group-2 injections
Table 3 Comparing the effect of DistBrd on received signals for the two Groups of injections
Point
Voltage
Amplitude I
[ V ]
Voltage
Amplitude II
[ V ]
Difference
[ dB ]
P-1 27.3 29.4 0.64
P-4 2.36 13.32 15.0
Chapter.2- Characterization of a Powerline Channel
- 40 -
Based on Fig.17 and Table 3 the following conclusions can be made:
i. The effect caused by the DistBrd on transmission types characterized by Group-1
injections is in the range of only 0.64 dB or 7% difference.
ii. The effect caused by the DistBrd on transmission types characterized by Group-2
injections is in the range of 15 dB or 462% difference.
iii. Therefore, it is obvious that the DistBrd causes a very strong attenuation with in the
Powerline network and its influence on the different transmission-reception pairs is
also different. Due to this reason the location of the transmission-reception pairs
should be taken in to account when analyzing the Powerline network to get a better
impression of the level of attenuations the signals experience as they propagate along
the channel.
2.3.5. TCTL and LCTL
One common way of quantifying the level of unsymmetries inside a complex network is
through estimating by what level in the network are Symmetrical signals converted to
Asymmetrical signals, or vice versa, during signal transmissions. ITU-T G.117 defines
terminologies and measurement setups related to these conversion parameters meant to
quantify levels of unsymmetries inside complex networks [GMHH08].
The following Terminologies are defined in ITU-T G.117:
i. Transverse Conversion Loss (TCL) for One-port Networks, or Transverse Conversion
Transmission Loss (TCTL) for Two-port networks, defined as the level by which
Symmetrical signals are converted to Asymmetrical signals with in the network.
ii. Longitudinal Conversion Loss (LCL) for One-port networks, or Longitudinal
Conversion Transmission Loss (LCTL) for Two-port networks, defined as the level by
which Asymmetrical signals are converted to Symmetrical signals with in the network.
Fig.18 (a) and (b) show the diagrammatical representation for the computation of TCTL and
LCTL as defined in ITU-T G.117
Chapter.2- Characterization of a Powerline Channel
- 41 -
(a)
(b)
Figure 18 Measurement Setup for (a) TCTL and (b) LCTL as defined in ITU-T G.117
And the expressions for these parameters are also given in ITU-T G.117 as:
2
110log20
L
T
V
VTCTL = (2.15a)
2
110log20
T
L
V
ELCTL = (2.15b)
For the purpose of characterizing the PLC channel in the time domain, measurements were
performed in accordance with the setup of Fig.18 to further analyze the network. Fig.19 (a and
b) show the impulsive echo measurement on a PLC channel for characterizing the level of
unsymmetries inside the network. These figures and their different voltage and timing
information parameters are to be understood as follows:
i. Fig.19 (a):- Symmetrical signal pulse is injected and Asymmetrical signal is
measured at the output port of the network.
• y
V is what is thought to be the original signal pulse attenuated by the internal
Symmetrical/Asymmetrical conversion factor of the Macfarlane Probe used in
the measurement.
Chapter.2- Characterization of a Powerline Channel
- 42 -
• sa
V : is the Asymmetrical signal output at the measurement point due to the injected
Symmetrical signal
• sat : is the time delay between
yV and
saV indicating the location where Symmetrical
signals are strongly converted to Asymmetrical signals.
Vsa1
-1.2
-1
-0.8
-0.6
-0.4
-0.2
0
0.2
0.4
0.6
0.8
-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006
Vy1
Vas1
Vz1
-1.5
-1
-0.5
0
0.5
1
1.5
-1e-006 -8e-007 -6e-007 -4e-007 -2e-007 0 2e-007 4e-007 6e-007 8e-007 1e-006
Figure 19 Impulsive echo measurement results of (a) TCTL and (b) LCTL
ii. Fig.19 (b):- Asymmetrical signal pulse is injected and Symmetrical signal is
measured at the output port of the network.
• z
V : is what is assumed to be the original signal received after being attenuated by
the internal Asymmetrical/Symmetrical conversion factor of the Macfarlane
probe used in the measurement.
Chapter.2- Characterization of a Powerline Channel
- 43 -
• as
V : is the symmetrical signal converted from asymmetrical injection in the network.
• ast : is the time delay between
zV and
asV indicating the location where Asymmetrical
signals are strongly converted to Symmetrical signals.
These voltage amplitude and timing parameters for the different points are summarized in the
following Tables.
Table 4 Amplitude and Timing parameters for Symmetrical inputs and Asymmetrical outputs
Point sa
V [ V ] sat [µs]
P-1 -1.164 0.307
P-2 -1.227 0.293
Gro
up-1
P-3 -1.164 0.264
P-4 -0.244 --
P-5 -0.227 --
Gro
up-2
P-6 -0.333 --
Table 5 Amplitude and Timing parameters for Asymmetrical inputs and Symmetrical outputs
Point as
V [ V ] ast [µs]
P-1 -1.2 0.302
P-2 -1.2 0.281
Gro
up1
P-3 -1.55 0.240
P-4 0.216 --
P-5 0.289 --
Gro
up2
P-6 0.295 --
Both Fig.19 (a) and (b) are measurement results for injections made at point P-1 of Group-1.
Results from measurements done from injections at P-2 and P-3 also show comparatively
similar to that of P-1. Measurement results of Group-2 injections, however, exhibit very strong
and un-patterned oscillations and therefore retrieving timing parameters is found to have been
very difficult. It is for this reason that timing parameters for Group-2 injections are missing in
Table 4 and Table 5.
Comparing the time delay parameters sat of Table 4 and
ast of Table 5 with the timing
parameter sst of Table 1, it is strongly convincing that locations of strong reflections are also
locations where Symmetrical signals are converted to Asymmetrical signals and Asymmetrical
Chapter.2- Characterization of a Powerline Channel
- 44 -
signals are also converted to Symmetrical signals. Or said differently, locations of strong
reflections are also locations where Modal conversions take place. This is as per the common
practice of referring the Symmetrical signal as Differential Mode (DM) signal and the
Asymmetrical signal as Common-Mode (CM) among the EMC community.
Similar to what was discussed in Section 2.3.3 for Line Attenuations, both the TCTL and
LCTL parameters are computed in Time domain as well as in Frequency domain and results
are summarized as shown in Table 6 and Table 7.
Table 6 TCTL Values in Time and Frequency domains
Points
Time Domain
(Signal ratios)
TCTL [ dB ]
Freq. Domain
transformed
TCTL [ dB ]
P-1 38 35-40
P-2 37 35-40
Gro
up-1
P-3 38 35-40
P-4 51 45-55
P-5 52 45-55
Gro
up-2
P-6 49 45-55
Table 7 LCTL Values in Time and Frequency domains
Points
Time Domain
(Signal ratios)
LCTL [ dB ]
Freq. Domain
transformed
LCTL [ dB ]
P-1 27 15-20
P-2 27 15-20
Gro
up-1
P-3 25 15-20
P-4 42 40-45
P-5 39 40-45
Gro
up-2
P-6 39 40-45
Comparing the differences between the TCTL values of Group-1 injections and Group-2
injections in Table 6, and also comparing the differences between the LCTL values of Group-1
and Group-2 injections in Table 7 it is obvious to see that these differences are related to the
source of the unsymmetries which contributed for the same range of difference in line
Chapter.2- Characterization of a Powerline Channel
- 45 -
attenuations between Group-1 and Group-2 injections. This can be understood from what is
already discussed in Section 2.3.3 and Section 2.3.4.
In all of the impulsive echoes measurements being discussed in Section 2.3 the Macfarlane
probe is used for injections of both symmetrical and asymmetrical signals. This probe,
however, is calibrated for measurement resolution BW for frequencies below 30 MHz (i.e.
9 kHz) and its characteristics for measurement BW for frequencies above 30 MHz (i.e.
120 kHz) is not calibrated or standardized. Despite that, the probe has its own characteristic
parameters beyond 30 MHz range and that remains to be the property of the probe even if it is
not standardized. Additionally, any change in the resolution BW has the same effect on both
the symmetrical and the asymmetrical signals.
- 46 -
3. Transmission of UWB Pulses over Powerline
Channel
Contents
• Formulation of the UWB Signal Pulses
o Gaussian and its Derivative Pulses o Power Spectral Density (PSD)
• Transmission of UWB Pulse Signals
o Pulse Parameters o Improving Reception o Simulations o Transmission Setup
3.1. Formulation of the UWB Signal Pulses
In the wireless applications, different forms of pulses including different forms of Gaussian
pulses and Hermitian pulses are employed as UWB pulses [MGLM05], [XSMG06],
[Miller03], . Among these different families of pulses the second derivative Gaussian pulse is
much commonly used for many of the carrier-less UWB wireless transmissions. Therefore, this
same waveform of 2nd order Gaussian pulse is used in all the analysis made in this Dissertation.
3.1.1. Gaussian and its Derivative Pulses
The mathematical expressions of the Gaussian Pulse and its first four derivative pulses are
shown below [ASPO03], [BHNB04], [LZAH01], [RepJH01].
• Zero-order Gaussian pulse given by
−=
2
2
02
exp)(τt
tg (3.1)
• First Derivative Gaussian,
−==
2
2
210
1 2exp
)(
)()(
ττtt
Atd
tdgtg (3.2)
Chapter.3- Transmission of UWB Pulses over Powerline Channel
- 47 -
• Second Derivative Gaussian,
−
−==
2
2
2
2
2
2
2
2
02 2
exp1)(
)()(
τττttA
td
tdgtg (3.3)
• Third Derivative Gaussian
−
−==
2
2
2
2
433
3
03 2
exp3)(
)()(
τττttt
Atd
tdgtg (3.4)
• Fourth Derivative Gaussian,
−
+−==
2
2
4
4
2
2
4
4
4
4
04 2
exp6
3)(
)()(
ττττtttA
td
tdgtg (3.5)
Where the factors 1A , 2A , 3A and 4A are amplitude coefficients of each pulse intended to scale
its respective amplitude to some intended value, as shown in Fig.20 for a 1 V peak pulse.
1
0
1
g1 t( )
t0.7
0.15
1
g0 t( )
t0.7
0.15
1
g2 t( )
t
1
0
1
g3 t( )
t0.7
0.15
1
g4 t( )
t
Figure 20 Gaussian Pulse and its first four derivative pulses
The Fourier Transform for the zero-order pulse is given by:
( )222
0 2exp2)( ffG τππτ −= (3.6)
From the Fourier Transform property, the Fourier Transform of a higher order derivative
pulse, say an thn derivative pulse, can be computed from:
( ) )(2)(
)( 00 fGfjA
dt
tgdfG
n
nn
n
nπ=
ℑ= (3.7)
Where
[ ].ℑ represents the Fourier Transform and
nA is the amplitude coefficient
Chapter.3- Transmission of UWB Pulses over Powerline Channel
- 48 -
The plot of the Fourier Transform of the pulses in Fig.20 are shown in Fig.21
0 7.5 .107
1.5 .108
G0 f( )
G1 f( )
G2 f( )
G3 f( )
G4 f( )
f
Figure 21 Fourier Transform of the pulses in Fig.20
In this Thesis the first two derivatives of the Gaussian pulse are referred to as 1DGP (or
interchangeably )(1 tg ), and 2DGP (or interchangeably )(2 tg ), respectively, and attention is given
to this two.
The Autocorrelation functions of 1DGP and 2DGP are computed from (3.2) and (3.3) as
follows:
• For 1DGP
∫∞
∞−
−= λλλ dtggtR )().()( 1111
∫∞
∞−
−−
−
−
= λ
τλ
τλ
τλ
τλ
dtt
tR2
2
2
2
22112
)(exp
2exp
)()(
Solving the integral gives the autocorrelation of 1DGP as:
−
−=
2
2
2
2
114
exp2
12
)(τττ
π tttR (3.8)
• Similarly, for 2DGP
∫∞
∞−
−= λλλ dtggtR )().()( 2222
∫∞
∞−
−−
−
−−
−= λ
τλ
τλ
τλ
τλ
dtt
tR2
2
2
2
2
2
2
2
222
)(exp
2exp
)(11)(
Solving the integral, here too, gives:
−
+−=
2
22
3
4
224
exp4
3
4
3
16)(
ττ
ττπ
ttttR (3.9)
Chapter.3- Transmission of UWB Pulses over Powerline Channel
- 49 -
Plots of the two autocorrelation functions have been shown in the following figures:
R11 t( )
t
R22 t( )
t
Figure 22 Plots of Autocorrelation functions of 1DGP and 2DGP
It is also interesting to see that the autocorrelation functions (3.8) of 1DGP and autocorrelation
function (3.9) of 2DGP are themselves derivative pulses and are scaled version of the second
derivative and the fourth derivative Gaussian pulses, respectively.
3.1.2. Power Spectral Density (PSD)
The PSD of a signal is defined as the average power delivered to a 1 Ω load per unit frequency.
The classical approach for computing the PSD of a signal are:
i. Correlogram method: The Autocorrelation function of the signal is to be computed and the
signal PSD is the Fourier Transform of the Autocorrelation Function. For a signal )(tx with
Autocorrelation function )(tRxx
, the power spectral density function S can be expressed as:
∫∞
∞−
−= dtftjtRfSxx
)2exp()()( π
Therefore, computing the Fourier transforms of (3.8) and (3.9) gives the PSD of the 1DGP
and that of the 2DGP, respectively.
ii. Periodogram method: The signal PSD can also be computed as a statistical average of the
square of the Fourier Transform of the signal over a time period. Mathematically:
2)(
1)( fX
TfS =
Where )( fX is the Fourier Transform of the signal )(tx and T is the period of the signal.
Chapter.3- Transmission of UWB Pulses over Powerline Channel
- 50 -
3.2. Transmission of UWB Pulse Signals
Before dealing with the issue of minimized radiation from UWB signals over conductive
media, it is necessary to verify if it is really possible to transmit such narrow-pulses over
branched conductors. To prove this, 2DGP signals were transmitted over PLC channels of
different branching configurations in Laboratory environment (Test-Benches).
3.2.1. Pulse Parameters
Consider Fig.23 for the design of the appropriate pulse parameters for the 2DGP. Both
modulated and un-modulated streams of repetitive pulses of this shape were transmitted on the
PLC channel.
Figure 23 Parameters for Pulse Design
• Where:
o p
T Pulse width
o idT Idle time (also called Guard time)
Then, the pulse period T is therefore the sum of the pulse width and the guard time, leading to
the pulse repetition frequency (or rate) PRF to be given as:
idp
PRFTTT
f+
==11 (3.10)
The Idle time idT can be considered as sliding time, which means that its value is to be varied
on real time to proportionally decrease or increase the PRF by keeping the pulse width p
T
constant. Varying the PRF in-turn proportionally affects the PSD of the signal. There should be
some sort of synchronizations between the transmission and reception nodes when real-time
adjustment of the guard time is part of the transmission protocol. This helps to reliably track
the data streams.
Chapter.3- Transmission of UWB Pulses over Powerline Channel
- 51 -
3.2.2. Improving Reception
Transmission of pulse signals across Powerline channel needs to be treated differently from
wireless pulse transmissions so that the decision making circuitry inside the receiver makes
optimum decision based on the inherent characteristics of the Powerline channel [GMHH07].
Consider Fig.24 in which the transmitted pulse signal propagates across a Powerline channel of
impulse response )(th . An Additive White Gaussian Noise (AWGN) )(tn with a constant PSD
0 is assumed to have been added into the signal in the channel.
Figure 24 Representation of signals a PLC channel
Therefore, the received signal )(ty can be written as:
)()(*)()(2
tnthtgty += (3.11)
Where * indicates convolution in the time domain.
In the standard matched filtering of pulse signals, the received signal )(ty is passed through a
filter which has an impulse response of the template of the transmitted pulse )(2 tg . Let us
consider two types of filters to explore additional possibility for the impulse response of the
matched filter, if it is possible to optimize the decision making process. One filter would have
the impulse response of the transmitted pulse )(2 tg , while the other filter has an impulse
response of )(0 tg as shown below:
Figure 25 Convolution of received and template pulses
Since filtering is equivalent to convolution in the time domain the representations of Fig.25 is
meant to simplify the explanation. This solution however has the following challenges in
delivering the intended optimized decision making.
Chapter.3- Transmission of UWB Pulses over Powerline Channel
- 52 -
• The concept of matched filtering is based on the idea of optimally receiving signals in a
channel which is corrupted by additive and white noises in nature. The PLC channel,
however, has an impulse response )(th in addition to an AWGN response )(tn .
• The impulse response )(th from branched Powerline channels has multiple echoes with
time spacing and amplitude coefficients strongly dependent on the location, the
number, the length and the loading conditions of the branches. This has been shown in
the simulation results of Section 2.2
Now, consider a different approach in which the convolution in the time domain in Fig.26 is
replaced by a multiplication to time-delayed local templates of )(0 tg and )(2 tg as shown below:
Figure 26 Multiplication of received and template pulses
Simulations show that the template for the delayed version of the zero-order Gaussian pulse
produces a better result even if the transmitted pulse is a 2DGP. This is also true when the
1DGP is transmitted. Based on this understanding the Model of Fig.27 is thought to minimize
the effect of branches of the PLC channel if included as part of the Receiver circuit so that the
Receiver is able to make a better decision about the transmitted pulses [GMHH07].
Figure 27 Possible circuitry to minimize effects of branches from received pulses.
From Fig.27 and equation (3.3), one can write:
∫ ∫ −=− dtdtTtgATtgdelycdely
)()( 20 (3.12)
Where:
cA is amplitude coefficient to be set based on preference
Chapter.3- Transmission of UWB Pulses over Powerline Channel
- 53 -
delyT is the propagation delay which depends on the length and the dielectric property.
It is to be determined from the timing information before the actual data
transmission takes place.
Therefore, the received signal can be optimally computed from:
[ ][ ] )(.)()(*)()( 02 delayTtgtnthtgty −+= (3.13)
This helps to maximize the performance of the Receiver circuitry, as shown in the simulation
and the measurement results.
3.2.3. Simulations
The simulation for the received pulses with and without the effect of the correction circuitry
was done for the following four channels differing in the number of branches:
• CH-1: 2x2.5 sq. mm PVC insulated cable, 20m length, 1 m above perfect
ground surface, no branch along the channel
• CH-2: same as CH-1 but with one 2 m long branch at 10 m from signal
injection point
• CH-3: same as CH-1 but with two branches each of which are 2 m long at 5 m
and 15 m away from signal injection point
• CH-4: same as CH-1 but three branches each of which are 2 m long at 5 m,
10 m and 15 m away from signal injection point.
The results of the simulation are shown in Fig.28 for the different Channel configurations
shown above.
Chapter.3- Transmission of UWB Pulses over Powerline Channel
- 54 -
050 100 150 200
-0.5
0
0.5
1
1.5
Time [ ns ]
(b) CH-2 one Branch
without..
with correction
0 50 100 150 200
-0.6
-0.4
-0.2
0
0.2
0.4
0.6
0.8
Time [ ns ]
(c) CH-3 two Branches
without..
with correction
0 50 100 150 200
-0.4
-0.2
0
0.2
0.4
0.6
Time [ ns ]
(d) CH-4 three Branches
without..
with correction
Figure 28 Received Pulses with and without the correction circuitry on the four channels
Chapter.3- Transmission of UWB Pulses over Powerline Channel
- 55 -
3.2.4. Transmission Setup
The measurement setup for this procedure consists of the channels described in the previous
Section (CH-1, CH-2, CH-3 and CH-4) and the following equipment:
Figure 29 Measurement Configuration of UWB transmission over 2PWN
• The transmitting end consists of Tektronix AFG3101 Arbitrary Function Generator
(AFG3101) as its main component for generating the UWB pulses. The AFG3101 has
the following basic specifications:
o Sampling rate: 2.5G Giga samples per second (Gs/s)
o Frequency: 100 MHz for sinusoidal, 50MHz for Arbitrary Functions
• The receiving end consists of Tektronix DPO3104 Oscilloscope as its main component
with the following basic specifications:
o Sampling rate: 2.5 Gs/s
o Analog BW: 350 MHz
3.2.4.1. Reference Transmission
Before transmitting the pulse over the Laboratory test bench channels of different
configurations (CH-1, CH-2, CH-3 and CH-4), a reference test setup was analyzed. This
reference setup consists of the same transmitting and receiving setups similar to that intended
for the different PLC channels except that the channel is a 1 m coaxial cable with 50 Ω
characteristic impedance.
Chapter.3- Transmission of UWB Pulses over Powerline Channel
- 56 -
Figure 30 Received 2DGP over a 1 m coax
Setting the internal impedances of both the AFG3101 and the DPO3104 to 50 Ω makes this
reference transmission a perfectly matched one. Pulses to be received over branched PLC
channels are to be compared to this reference pulse received under such an ideal channel
condition. Fig.30 shows this received reference pulse over the 1 m coaxial cable together with
a “soft”-implementation of the correction circuitry of Fig.27.
The different curves in Fig.30 and Fig.31 are to be understood as follows:
• RED: This is the received pulse after the original pulse propagates across its
corresponding channel
• GREE-: This is the correction pulse to be multiplied at the multiplier circuitry of
Fig.27.
• BLUE: This is the resulting pulse which would help the decision making
circuitry to decide whether a pulse had been transmitted or not.
3.2.4.2. Transmission on Test Bench
Then the 2DGP signal is transmitted over the different channels with configurations previously
stated (Section 3.2.3).
(a) Received 2DGP over CH-1
Chapter.3- Transmission of UWB Pulses over Powerline Channel
- 57 -
(b) Received 2DGP over CH-2
(c) Received 2DGP over CH-3
(d) Received 2DGP over CH-4
Figure 31 Received 2DGP over the different channels and effect of the proposed correction circuitry
The following reasonable conclusions can be made based on the results shown in Fig.30 and Fig.31:
i. The PLC channel is not as dispersive as commonly thought for nano-second range base
band pulse transmissions. The major difference between the received pulses over the
reference channel and over the different Test bench channels is not distortion of pulse
shape (dispersion) but rather it is mainly amplitude difference.
Chapter.3- Transmission of UWB Pulses over Powerline Channel
- 58 -
ii. The improvement achieved by the proposed correction circuitry on the received pulses
can be easily seen. This is a “soft” implementation of the circuitry, which means that
template of the zero-order Gaussian pulse is multiplied to the received streams of pulses
within the digital oscilloscope on real-time. The BLUE pulses in Fig.30 and Fig.31 are
better decoded to data bits than the RED pulses during actual data transmissions.
- 59 -
4. Theoretical Analysis of Interferences from UWB
Signals
Contents
• Radiated Power Loss from 2PW-
• Power Spectral Density of UWB Signals
• Effect of Modulation in minimizing spectral lines
o Un-modulated pulses o Modulated pulses o Carrier-based transmissions
• UWB signals on a -arrow-band Receiver
• Pulse width and Measurement Frequency
• Low/High PRF Region, PDCF and Effective Duty Cycle
4.1. Radiated Power Loss from 2PW-
For a general 2PWN the radiated power loss from the network can be expressed in terms of the
different power loss terms as [GMHH06]:
othersdialcculossradPPPPP −−−= (4.1)
Where:
radP radiated power from the network
lossP total power loss inside the network, measured as a difference of input
and output power levels.
cuP copper loss inside the network
dialcP dielectric loss inside the network
othersP Other losses not accounted for by
radP ,
cuP and
dialcP
For a worst case radiation analysis the copper losses, the dielectric losses and all sorts of losses
can be neglected and all losses inside the network can be attributed to radiation. Considering
Fig.32, the total loss, and by implication the radiation loss, can then be expressed in terms of
the incident and reflected waves as:
2
2
2
1
2
1 bbaPrad
−−= (4.2)
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 60 -
Where:
2
1a incident power at port-1
2
1b reflected power at port 1 due to mismatch at that port
2
2b power delivered to the load at port-2.
2PWNPort-1 Port-2
b1, reflected
a2=0
b2, to load
Power losses of all sorts
a1, from source
Figure 32 Wave propagation to and from a 2PWN
Re-writing (4.2):
−−=
2
1
2
2
2
1
2
12
1 1a
b
a
baP
rad (4.3a)
( )2
21
2
111 SSPPinrad
−−= (4.3b)
Where:
inP transmitted signal power
11S reflection coefficient at port-1, and
21S forward transmission coefficient
Therefore, equation (4.3b) shows that the radiated power from a wired network is characterized
by the level of the transmitted signal power and the characteristic parameters 11S and 21S of the
network itself.
The transmitted power inP is the total power injected at the input port of the network and hence
(4.3b) does not indicate the distribution of the radiated power across a given frequency band.
Therefore, it is possible to re-write (4.3b) in a way that would better represent the radiated
power as:
( )2
21
2
11 )()(1)()( fSfSfPSDfPSDsigrad
−−= (4.4)
Where:
sig
PSD PSD of the transmitted signal
rad
PSD radiated power spectral density
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 61 -
Let us see how and when the two equations (4.3b) and (4.4) are to be used:
• For a narrow-band signal the radiated power measured by a measuring receiver is the
average power over the measuring BW of the receiver and hence equation (4.3b)
applies.
• For a signal having a BW much higher than the BW of the measuring receiver equation
(4.3b) represents a fraction of the total radiated power. Therefore, it would be
reasonable to express in terms of the radiated PSD instead of the radiated power, and
hence equation (4.4) is more suitable to UWB signals than (4.3b).
Additionally, different national and international limits on radiated fields put maximum
allowable values to the radiated power over a given frequency band (which is the radiated
PSD) or to the radiated electric or magnetic field over a given frequency band. Due to this
reason it is equation (4.4) which correctly signifies the subject of this Thesis better than
equation (4.3b).
The classical approach to minimizing EMI or disturbances from signal transmissions over a
given channel characterized by 11S and 21S is either to condition the channel to optimize 11
S and
21S or to reduce the level of the transmitted power inP . Impedance matching, balancing of points
of unsymmetries, filtering, and other different techniques are some of the conditioning
techniques to optimize the values of 11S and 21S . However, as to what level can a channel be
conditioned to minimize interference depends on the nature and the complexity of the channel
itself. Some channels are relatively easier to condition than others. Powerline channels,
unfortunately, are not among those that are easy to condition as they are not primarily intended
to be communication channels. Therefore, considering a different alternative is necessary for
such cases. Equation (4.4) explains the basic reason why implementation of lower PSD signals,
as defined bysig
PSD , can be of help in minimizing EMI from a 2PWN channel.
4.2. Power Spectral Density of UWB Signals
One way to compare the radiations from two different signals across the same channel is by
comparing the distribution of the radiated power from each signal as they are transmitted
across that same channel.
Additionally, the PSD of a signal is of great importance in analyzing the interference of the
signal on a given victim circuit. This is because of the fact that the magnitude and the
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 62 -
distribution of the power spectral lines of the signal across the BW of the victim circuit depend
not only on the shape of the transmitted pulse, but also on how the pulses are modulated in
amplitude and in position at the transmitter.
Consider the following model for a general UWB transmission:
∑−
=
−=1
0
)()(f
n
nnTtatd δ ∑
−
=
−=1
0
)()(f
n
nnTtgats
Figure 33 Representation of the discrete signal pulse
Where )(tδ is the Dirac delta function. Therefore, it is possible to re-write the UWB pulse
trains as a convolution of a single pulse with a sequence of Dirac delta function, as follows:
)(*)()( tgtdts = (4.5a)
−= ∑
∞
∞−
nTtatgtsn
(*)()( δ (4.5b)
Where * indicates the convolution operation. Then the Fourier Transform of (4.5a) is:
( )∑ −== fnTjafGfDfGfSk
π2exp).()().()( (4.6)
Where:
)( fG is the Fourier Transform of a single transmitted UWB pulse )(tg
)( fD is the Fourier Transform of )(td
Note that )( fS has spikes every T/1 due to the parameters inside the summation, and its
magnitude depends on the coefficient term k
a of the modulation.
The PSD, which is the average power per Hertz, can then be expressed as:
)()(1
)(2
fSfGT
fSddss
= (4.7)
Where
2)( fG is the energy spectral density (ESD) of a single pulse
)( fS dd is the average PSD of the Dirac delta function with amplitude
coefficients k
a .
Therefore, the PSD of the UWB signal is determined by the following two factors:
i. The ESD of a single pulse, which determines the range of frequency over which the
energy of the signal is available, and
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 63 -
ii. The PSD of the Dirac delta function )( fSdd
which is determined by how the pulses are
modulated in time and in amplitude, and in turn determines the discrete spectral
components of the PSD of the UWB signal over that spectrum.
And the total energy in a single pulse is given by:
∫∞
∞−
= dttgEg
2)( in the time domain, or (4.8a)
∫∞
∞−
= dffGEg
2)( in the frequency domain (4.8b)
And, the total average power of an UWB pulse signal of period T is then given by:
TEPgg/= (4.9)
4.3. Effect of Modulation in Minimizing Spectral Lines
Consider a general M-array PAM modulated transmission format as shown in Fig.34 with a
block of data consisting of N Symbols. The symbol time s
T and the Frame duration f
T are
equal, and hence the signal can be represented as:
∑−
=
−=1
0
)()(M
k
fkkTtgats
The parameter k
a is the kth coefficient of the amplitude random variable n
a with mean a
µ and
variance 2
aσ represented as:
1210 ,......,.....,,, −=Mkn
aaaaaa
The different timing parameters of )(ts are as shown in Fig.34 and the PSD of this data
representation is given in the Literature including [Proak01], [ASPO03] [YNAM06], [Win02],
[YNAM03] as shown in equation (4.10) below.
4444 34444 2143421discrete
f
M
k ff
a
coninuous
f
a
ssT
kf
T
kG
TfG
TfS
−
+= ∑
−
=
δµσ
21
0
2
22
2
)()( (4.10)
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 64 -
1
. . .
. . .
Figure 34 Format of M-Array PAM UWB modulation
Therefore, (4.10) shows that the PSD of the transmission has both a continuous part and a
discrete part. Selection of a particular modulation technique determines whether the continuous
or the discrete part dictates the magnitude of the PSD. The performance of different
modulations commonly implemented in UWB pulse transmission on the PSD of (4.10) are
further investigated in the following sub Sections.
4.3.1. Un-modulated pulses
First, consider an un-modulated signal characterized by a constant amplitude coefficient
random variable n
a given by:
[ ]1,01 −∈∀= nan
The mean and variance of this random process is therefore:
[ ] 11 1
0
=== ∑−
=
n
nnaa
aEµ
( )[ ] 0)(1 1
0
222 =−=−= ∑−
=
n
ananaa
aE µµσ
Where
[ ].E represents the statistical expected value of a random process.
And hence from (4.10) the PSD of an un-modulated signal transmission becomes:
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 65 -
4444 34444 21discrete
f
k ff
ssT
kf
T
kG
TfS
−
= ∑
−
=
δ
21
0
2
1)( (4.11)
Therefore, the PSD of an un-modulated UWB signal consists of spectral spikes that are
separated by frequency of f
T/1 , with the envelope of these spikes following the curve of the
energy spectrum of a single pulse. The magnitudes of these spikes are proportional to the
square of the pulse repetition frequency, fPRF
Tf /1= . From the coefficients of both the discrete
part and the continuous part in (4.10) one can see that the discrete spectrum is dBfPRF
)log(10
higher than the magnitude of the continuous part. This makes un-modulated transmissions
among the maximum radiating UWB implementations [Wentz07].
4.3.2. Modulated pulses
4.3.2.1. Binary Phase Shift Keying (BPSK)
Consider again a random coefficient variable n
a representing BPSK modulation with an equal
probability of each bit and characterized by
[ ]1,01,1 −∈∀−= nan
For this random process, the mean and the variance values are:
0=a
µ
12 =a
σ
And hence the PSD of (4.10) becomes:
43421continuous
f
ssfG
TfS
2)(
1)( = (4.12)
With the spectral spikes disappearing and the PSD of the transmission equals the PSD of a
single UWB pulse. Therefore, during measurement of EMI from UWB signals, unlike carrier-
based transmissions, it is possible that interference measurements from un-modulated pulses
may be higher than that from modulated pulses.
4.3.2.2. Amplitude Shift Keying (ASK)
Similar to the BPSK, a general M-array ASK is considered in this Section. Let the UWB
transmission with M-Array ASK with its number of bits in each symbol given by M 2log=
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 66 -
has an amplitude random variable coefficient n
a . Assuming that each symbol has an equal
probability Mp /1= and distributed symmetrically on the negative and positive halfs of the
amplitude plane as shown in Fig.35 for 4-array ASK.
Figure 35 Amplitude distribution example for 4-ASK
Consider a n amplitude coefficient random variable:
[ ]1,0,,......,,,, 1210 −∈∀= − MnaaaaaaMkn
Where each discrete value k
a has a value of
12 −= kak
for all integer
−−∈
2,1
2
MMk
The mean and the variance of the random variable n
a for an M-array ASK are given by:
01 1
0
== ∑−
=
M
n
naa
Mµ (4.13)
3
)1( 22 −=
Ma
σ (4.14)
Therefore, the PSD of a UWB transmission of a general M-array ASK with each symbol
having equal probabilities and with zero mean amplitude coefficient random variable consists
of only the continuous part of (4.10) and is given by:
44 344 21continuous
f
ssfG
TM
MfS
2
2
2
)(3
)1()(
−= (4.15)
With equation (4.4) in mind, this implies that the level of EM interference from an M-array
ASK modulated UWB transmission is a product of the EM interference from a single pulse and
the variance of the amplitude coefficient random variable n
a defined by (4.14). Equations
(4.13) and (4.14) are to be understood as follows:
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 67 -
• Equation (4.13) is ensured by a symmetrical distribution of the amplitude coefficient in
the positive and negative amplitude plane, similar to the case of 4-ASK shown in
Fig.35. Additionally, each symbol should have an equal probability Mp /1= in the
signal space.
• Derivation of equation (4.14) is given below:
( )[ ] ∑−
=
−=−=1
0
222 )(1 M
n
ananaa
MaE µµσ
( )[ ] ( )∑−=
−=−=2/
2/1
222 121 M
Mk
anak
MaE µσ (4.16)
Due to the symmetry of the distribution of the amplitude over the positive and the
negative amplitude plane, (4.16) can be re-written as:
( )∑=
−=2/
1
22 122 M
k
ak
Mσ
+−= ∑∑∑
===
2/
1
2/
1
2/
1
22 1442 M
k
M
k
M
k
akk
Mσ (4.17)
The following summation properties are applied to the three sections of (4.17) as
follows:
623
23
1
2 nnnx
n
x
++=∑=
(4.18a)
2
)1(
1
+=∑
=
nnx
n
x
(4.18b)
nn
x
=∑=1
1 (4.18c)
Inserting the appropriate parameters, we get:
24
)2)(1(
12824
232/
1
2 ++=++=∑
=
MMMMMMk
M
k
(4.19a)
8
)2(2/
1
+=∑
=
MMk
M
k
(4.19b)
21
2/
1
MM
k
=∑=
(4.19c)
Combining (4.17) and the three equations of (4.19)
+
+−
++=
22
)2(
6
)2)(1(22 MMMMMM
Ma
σ and then
3
)1( 22 −=
Ma
σ (4.20)
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 68 -
4.3.2.3. Pulse Position Modulation (PPM)
Consider the case of an M-array PPM modulated UWB transmission. Fig.35 has been re-drawn
as shown in Fig.36 to accommodate parameters corresponding to PPM modulation. As one of
the commonly used modulation in the UWB applications PPM also needs to be discussed from
the EMI point of view [YSIT05].
Now, consider a PPM transmission characterized by M equally spaced positions within the
symbol time s
T . The frame time MTTsf/= occupies only one of these M possible frames and
the placement of the pulse at one of these equally spaced frame positions can be considered a
random process with mean p
µ and variance 2
pσ . Then, the signal wave form )(ts can be
expressed as:
∑∞
−∞=
+−=
n
fT
M
nktgts )( (4.21)
1
Figure 36 Format of M-Array PPM UWB modulation
In different Literatures [YNAM06], [Wentz07], [Eisen07], and for a binary PPM in
[MBGG04] the PSD of this signal for a general M-array PPM modulation with equally
probable symbols and randomly positioned in the time frame has been expressed as:
−+
−
= ∑∑∑ ∑
−
=
−
=
∞
−∞=
−
=
21
0
1
0
2
21
0
22)(
1)(
111)(
M
n
n
M
n
n
fn f
M
n f
n
f
ssfG
MfG
MTT
nf
T
nG
TMfS δ (4.22)
As can be seen from the first term of (4.22) PPM is also characterized by spectral lines spaced
at fPRF
Tf /1= . Similar to an un-modulated signal transmission discussed earlier in Section 4.3.1
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 69 -
these spectral lines follow the envelope of the spectrum of a single pulse f
TfG /|)(| 2 and are
higher than the smooth trace of the continuous spectrum by )log(10PRFf .
4.3.2.4. O--OFF Keying (OOK)
The OOK modulation is mapping of binary digits by the presence (bit 1) and absence (bit 0) of
the transmitted pulses, as shown below:
1 1 1
Figure 37 Representation of OOK Modulation
The equation for the PSD of an OOK modulated signal is correctly given in [JHMC01] as
shown in equation (4.23). Similar expressions are also shown in other sources including
[HAZC06] but these expressions lack the factor of ¼ and cannot be substantiated using (4.10)
why this parameter is missing. Computing the mean value a
µ and the variance 2
aσ for the
random variable of the amplitude distribution of an OOK modulated wave form and inserting
these values in equation (4.10) gives the same result as that of [JHMC01] and is given by.
44444 344444 2143421discrete
f
M
k ff
coninuous
f
ssT
kf
T
kG
TfG
TfS
−
+= ∑
−
=
δ
21
0
2
2
4
1)(
4
1)( (4.23)
And it has both the discrete and the continuous part. But, as discussed earlier, the amplitude of
the discrete part dominates the magnitude of the PSD.
4.3.2.5. Other Alternatives
It is also a common practice in the wireless UWB applications to modulate signals using two
different modulations to exploit the advantages both modulations provide. For example a pulse
modulated by both ASK and PPM has the advantage of a relatively smooth spectrum due to the
ASK and a better BW efficiency due to the PPM. Other techniques, such as Differential PPM
[DSJK99], delay-based BPSK [DWAC07], Pulse-based Polarity randomization [YNAM03],
are thoroughly discussed in the literature.
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 70 -
0 2.5.107
5 .107
0
0.01
0.02
0.03
0.04
UWB un-modulated Maximum at 12 MHz
Pow
er S
pect
ral
Den
sity
, PS
D [
W/ H
z ]
0 2.5 .107
5 .107
0
0.001
0.002
UWB BPSK Maximum at 12 MHz
Pow
er S
pect
ral
Den
sity
, PSD
[ W
/ H
z ]
0 2.5.107
5 .107
0
0.01
0.02
0.03
0.04
UWB un-modulated Maximum at 8 MHz
Pow
er S
pect
ral
Den
sity
, PS
D [
W/ H
z ]
0 2.5 .107
5 .107
0
0.001
0.002
UWB BPSK Maximum at 8 MHz
Pow
er S
pect
ral
Den
sity
, PSD
[ W
/ H
z ]
0 2.5.107
5 .107
0
0.01
0.02
0.03
0.04
UWB un-modulated Maximum at 4 MHz
Pow
er S
pect
ral
Den
sity
, PS
D [
W/ H
z ]
0 2.5 .107
5 .107
0
0.001
0.002
UWB BPSK Maximum at 4 MHz
Pow
er S
pect
ral
Den
sity
, PSD
[ W
/ H
z ]
Figure 38 Simulated plots of Continuous and Discrete PSDs of UWB pulses for maximum emissions at (a) 4 MHz, (b) 8 MHz, and (c) 12 MHz
4.3.3. Carrier-based Transmissions
For the analysis of interferences from carrier-based transmissions, it is a common practice in
the EMC measurements to transmit sinusoidal signals at a given frequency and then measure
the interference level (both conducted and radiated) at that particular frequency. Therefore, this
same approach is considered here and no further analytical expressions are required other than
what is commonly practiced [Perez95], [Paul06], [Schwab96], [FKPR05].
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 71 -
4.4. UWB Signals on a -arrow-band Receiver
It is also very important to analyze how the standard EMI receivers or any other narrow-band
receivers perform or measure the level of interferences that is coming from signals having
much larger bandwidth than their own [XSMG06]. The signal frequency spectrum and the
filter response of the measuring instrument can be represented as follows:
f
Figure 39 UWB pulses as seen over NB Receiver
Where in the above figure:
)( fHIF
is the receiver IF filter IF band-pass transfer function
)( fS is the spectrum of the received UWB signal
Therefore, the filter out put can be computed from:
)().()( fSfHfYIF
= (4.24)
In this case, in which the filter pass-band )( fHIF
has a much narrower bandwidth compared to
the spectrum:
• The spectrum )( fS is relatively constant across the IF pass band frequency range of the
filter centered at 0f .
• Therefore, the exact shape of the pulse is not important and the response of the filter to
a single pulse is the same as the impulse response of the IF filter itself, )( fhIF
, which is
the inverse Fourier Transform of )( fHIF
.
Now, let us further analyze the receiver’s output signal. Using (4.6) and (4.37)
)().().()( fDfGfHfYIF
= (4.25)
And considering )( fG to be constant across the IF bandwidth of the filter with magnitude of
)( 0fG :
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 72 -
<−
≥=
0);().().(
0);().().()(
0
0
ffDfGfH
ffDfGfHfY
IF
IF (4.26)
The IF filter impulse response )( fhIF
and the Gaussian Pulse )(tg are both real-valued time
functions, and hence both )( fHIF
and )( fG are conjugate symmetric. Additionally, for the sake
of interference evaluation it is possible to take )()( 00 fGfG −= even if the phase of the later
lags behind by π radians.
Therefore, without any loss of generality:
)(.|)(|).()( 0 fDfGfHfYIF
≅ (4.27)
And the PSD of the IF output is then:
)()()()(2
0
2fSfGfHfS
ddIFyy≅ (4.28)
Therefore, one can clearly see that the PSD of an UWB pulse received by a narrow-band EMI
measurement receiver (or a general narrow-band victim circuit) is a function of the ESD of a
single pulse at the center frequency 0f of the receiver IF filter and the pulse modulation
described by the PSD )( fSdd
. The detail analysis for the computation of )( fSdd
has been made
in Section 4.3.2 based on the implemented modulation technique.
4.5. Pulse width and Measurement frequency
In the standard measurement procedures for interferences (both conduction and radiation) from
a device operating at a given frequency 0f the Receiver’s IF center frequency should be shifted
to that same 0f to synchronize both the operation and the measurement frequencies. Also in
transmissions involving multiple carriers like the OFDM, the measurement center frequencies
should be synchronized to the center frequency of each subcarrier to measure the maximum
emission contributed by the transmission at that subcarrier.
This issue, however, is different when it comes to the measurement procedures involving UWB
signal transmission. In the following section a relation between the pulse width and the
measurement frequency is derived.
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 73 -
Consider the two commonly used UWB waveforms )(1 tg of (3.2) and )(2 tg of (3.3) for the
subject being considered. The corresponding frequency spectrums of these two waveforms are
obtained from equation (3.7) by setting 1=n for )(1 fG and 2=n for )(2 fG and also inserting the
corresponding amplitude coefficients 1A and 2A from the pulses.
( ) ( )( )2222
1 22)( feeffG τππττπ −= for )(1 tg and (4.29)
( ) ( ) ( )2222222
2 24)( feffG τππττπ −= for )(2 tg (4.30)
These spectrums have magnitude maxima at frequencies 1f and 2f , respectively, which can be
computed from:
0)(
1
1 == ff
df
fdG for )(1 tg , and (4.31)
0)(
2
2 == ff
df
fdG for )(2 tg (4.32)
Solving the above equations gives the following results:
τπ2
11 =f (4.33)
τπ2
12 =f (4.34)
Additionally, referring to )(2 tg of Fig.23 for the applications on conductive media, the energy of
the pulse is localized over the [ ]ττ 44.4,44.4− range of the time axis. A similar, but smaller pulse
range is indicated in [ASPO03]. Therefore, the pulse width p
T and the pulse width factor τ can
then be reasonably approximated as:
τ88.8≅p
T (4.35)
Combining (4.34) and (4.35) for )(2 tg gives the frequency of the maximum emission as:
pT
f2
2 ≅ (4.36a)
This is true whether a guard time g
T , as shown in Fig.23, is introduced or not; if introduced it
will only affect the total pulse period T , and hence affects the pulse repetition frequency, but
does not affect or shift the point of maximum emission 2f from what is defined by (4.36a).
This point of maximum emission 2f is a function of only the pulse factor τ . Similar analysis
for )(1 tg gives the maximum emission frequency point 1f as:
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 74 -
pT
f41.1
1 ≅ (4.36b)
For GHz range frequencies for wireless applications the values of 1f and 2f may slightly shift
from the values obtained from (4.36) due to the approximations used which would affect pulses
with a sub-nano range width p
T .
4.6. Low/High PRF Region, PDCF and Effective Duty Cycle
A-must-to-know fact during measurement of emissions from pulse based transmissions is what
is commonly known as Low- and High- PRF Region. For measurements to be done using
Spectrum Analyzer (SA), or EMI Test Receiver, the transmission is said to be “Low PRF
Region” or “High PRF Region” depending on the ratio of the Resolution Bandwidth (RBW) of
the measurement instrument to the PRF of the pulse as follows [Wentz07]:
• Low PRF Region: when RBW > 1.7 PRF, and
• High PRF Region: when RBW < 0.3 PRF (preferably RBW < 0.1 PRF)
The investigation in [AppN-150] shows that in the transition region where RBW are in the
same value range with the PRF ( )PRFRBW ≅ it is difficult to predict the response of the SA
since it is fully dependent on the modulation implemented for transmission of the pulses. For
the Low PRF region, the time spacing between pulses is large enough to allow the output of the
IF Filter inside the SA to return to zero between two consecutive pulses. Due to this, the Peak
Power (PK) and the Average Power (AV) measurements of the SA are independent of the
presence or absence of modulation and therefore both PK and AV Power measured by the SA
are the same for both modulated and un-modulated pulses, and the PK and AV Power
measurements are independent of the PRF as long as RBW > 1.7 PRF is maintained [AppN-
150].
For the High PRF Region, however, the RBW is sufficiently narrow such that the “line
spectrum” of impulses spaced at PRF is visible on the SA for a repetitive pulse train. Measured
PK and AV power levels are determined by the modulation implemented.
Another important and commonly considered during measurement of emissions from UWB
pulses is the effective duty cycle, effτ , which is defined as the width of a rectangular pulse
which has the same amplitude and the same area as the transmitted UWB pulse. This is defined
by (4.37) and shown in Fig.40 [Wentz06], [Wentz07].
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 75 -
∫=PRF
o pk
effV
dttp/1
)(τ (4.37)
Where:
)(tp is the transmitted UWB pulse and
pkV is the peak voltage of the pulse as shown in Fig.40 below. For the
remaining analysis 1=pk
V is assumed.
Equal area
eff
Vpk
time
Figure 40 Representation of "Effective Time"
The computation of this effective time effτ for a 2DGP is to be done in consistence with the
relation and the expression in equations (3.2) and (4.37) and knowing that the pulse centered at
0=t crosses the time axis at points of τ− and τ in terms of the pulse width constant:
∫∫∫−
−==PRFPRF
o
effdttpdttpdttp
2/1/1
)(2)()(τ
τ
τ
τ (4.38)
This takes in to account the whole area of the 2DGP, the positive main signal part (in the time
domain) and the two negative parts of the pulse. Inserting (4.35) in (4.38) and solving the
integral, the effective pulse width effτ in terms of the pulse width factor τ can be reasonably
expressed as:
ττ 43.2=eff
(4.39)
Which means that a 2DGP expressed by (3.3) with a period of 8.88τ has the same amplitude
and equal energy to a rectangular pulse of width 2.43τ . Hence, the transmission has an
effective duty cycle of 27%. In other words, the emission from a 100% duty cycle UWB pulse
is the same as the emission from a rectangular pulse of 27% duty cycle having same amplitude.
Chapter.4- Theoretical Analysis of Interferences from UWB Signals
- 76 -
Pulse Desensitization Correction Factor (PDCF) is a controversial factor which is thought to be
used to adjust SA measured power levels characterized by the effective duty cycle to the
corresponding 100% duty cycle value. The strong understanding is however, particularly for
applications less than 1GHz frequency, not to consider the PDCF. This is an argument based
on the fact that the interference level from an UWB transmission should be considered as only
that amount received by the NB victim receiver and should not be extrapolated to what the
victim would have received had it had the same BW as the transmitting system.
Therefore, even if knowing the effective duty cycle gives a comparative impression between
the transmitted UWB pulse and an equivalent rectangular pulse of equal energy the PDCF has
no direct relevance to the subject of this work.
- 77 -
5. EMI Measurement Setups and Results
Contents
• Measured Signal Spectrum
• EMI Measurement Setups
o Disturbance Voltage Measurement Setup o Radiation Measurement setup
• Measurement Results
o Disturbance Voltage Measurement Result o Radiated Field Measurement Results
5.1. Measured Signal Spectrum
One way of comparing interferences (both conducted interferences and radiated interferences)
from a given network due to two or more different transmission protocols is done by analyzing
the voltage spectra of these different transmission protocols using a Spectrum Analyzer (SA).
This is because of the fact that level of interferences and voltage spectral components are
directly proportional [JTSI05], [Loyka00].
Therefore, continuous streams of:
• Sinusoidal waves
• Un-modulated UWB pulses, and
• BPSK modulated UWB pulses
were generated and the voltage spectra of each of these signals were measured. All of the
signals waveforms generated and measured during this investigation have the following
parameters:
• Amplitude 0 dBm (or 107dBµV on a signal generator of 50 Ω internal impedance)
• For safety of the SA, 20 dB Attenuator was connected at the input port of the SA
The generation and measurement of these signals were performed as follows:
a. Sinusoidal signal: Continuous Wave (CW) is generated using AFG3101
Function Generator. For the sake of observing the signals at few selected
frequencies 4 MHz, 8 MHz and 12 MHz points were taken as the signal
Chapter.5- EMI Measurement Setups and Results
- 78 -
frequencies. The measured voltage spectra of the CW at those three frequencies are
shown in Fig.41(a), Fig. 42(a) and Fig.43(a).
b. Un-modulated UWB: Continuous stream of pulses with mathematical equation
of (3.3) and waveform of Fig.23 (with zero idle time idT ) were generated using the
AFG3101 Function generator. The relation between the pulse width and the
frequency of maximum spectrum (4 MHz, 8 MHz and 12 MHz) were adjusted
according to the relation in (4.36a). The amplitude of the streams of pulses is the
same as that used for CW. The measured discrete spectral lines of the un-modulated
pulses at those frequencies are shown in Fig.41(b), Fig. 42(b) and Fig.43(b).
c. BPSK-modulated UWB: Continuous streams of the same pulses for the un-
modulated transmission were used except that the pulse trains were alternate
positive and inverted pulses of a pre-determined pattern. This is due to the fact that
the AFG3101 used for generating the UWB pulses is not meant to and hence does
not have the necessary speed to switch between alternate pulses for a randomized
data streams. This approach leads to a periodic sequence which contributes for the
presence of spectral lines of its own related to the measurement BW of the
Spectrum Analyzer. This can be easily noticed from Fig.41(c), Fig. 42(c) and
Fig.43(c). These spectra are re-plotted as shown in Fig.44 showing the average of
the continuous spectrum which would have been the case had the UWB pulses been
modulated by random sequences of alternate +1 and -1. During the EMI
measurements to be discussed in Section 5.2, therefore, the EMI test Receiver
measured the rather higher spectra components resulted from the pre-determined
alternate patters of stream of pulses instead of the continuous spectrum represented
by the average trace shown in Fig.44. Additionally, these discrete spectra are
relatively constant despite changes in the pulse width. The actual continuous spectra
as shown in Fig.44 (a, b, c), however, linearly decreases with a decreasing pulse
width. Therefore, the results to be shown in Section 5.2 for the BPSK modulation is
the worst case scenario and the actual emission from BPSK modulated pulses are
much lower than the results presented in that Section [Loyka98], [NLAH03]. A
summary of the maximum values of these different wave forms are shown in
Table 8.
Chapter.5- EMI Measurement Setups and Results
- 79 -
20
30
40
50
60
70
80
90
0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008
20
30
40
50
60
70
80
90
0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008
20
30
40
50
60
70
80
90
0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008
Figure 41 Measured spectrum comparisons at 4 MHz (a) sinusoidal, (b) un-mod UWB (c) bpsk UWB
Chapter.5- EMI Measurement Setups and Results
- 80 -
(a) Sinusoidal Frequency [ Hz]
(b) unmod-UWB Frequency [ Hz]
(c) bpsk-UWB Frequency [ Hz]
20
30
40
50
60
70
80
90
0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008
20
30
40
50
60
70
80
90
0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008
20
30
40
50
60
70
80
90
0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008
Figure 42 Measured spectrum comparisons at 8 MHz (a) sinusoidal, (b) un-mod UWB (c) bpsk UWB
Chapter.5- EMI Measurement Setups and Results
- 81 -
20
30
40
50
60
70
80
90
0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008
20
30
40
50
60
70
80
90
0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008
20
30
40
50
60
70
80
90
0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008
Figure 43 Measured spectrum comparisons at 12 MHz (a) sinusoidal (b) un-mod UWB (c) bpsk UWB
Chapter.5- EMI Measurement Setups and Results
- 82 -
20
30
40
50
60
70
80
90
0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008
20
30
40
50
60
70
80
90
0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008
10
20
30
40
50
60
70
80
90
0 1e+007 2e+007 3e+007 4e+007 5e+007 6e+007 7e+007 8e+007 9e+007 1e+008
(b) BPSK-UWB at 8MHz Frequency [ Hz]
(a) BPSK-UWB at 4MHz Frequency [ Hz]
(c) BPSK-UWB at 12MHz Frequency [ Hz]
averaged curve
averaged curve
averaged curve
Figure 44 Re-plot of Fig.41(c), Fig.42(c), and Fig.43(c) with mathematically averaged curve
Chapter.5- EMI Measurement Setups and Results
- 83 -
The difference in the voltage Spectra of these waves at those sample frequencies are
summarized in the following Table.
Table 8 Summary of the maximum spectral components of the different signals
Measured Voltages at the three selected measurement frequencies
[ dBµV ]
4 MHz
8 MHz
12 MHz
CW 86 86 86
Un-modulated UWB 81 81 81
known +1 and -1 pattern 71 71 71 BPSK-
modulated Average of the spectrum 68 65 62
The differences in the measured data summarized in Table 8 above are indications of the
differences expected in the level of interferences when these different forms of signals are
transmitted over the same network.
5.2. EMI Measurement Setups
The amplitudes of all the signals generated and transmitted for the measurement of conducted
and radiated Interferences are the same 0 dBm or 107 dBµV (for a generator of 50 Ω output
impedance) except that the 20 dB Attenuator connected to the input of the SA during the
spectral measurements stated in Section 5.1 is no more in use during the interference
measurements.
Additionally, a signal format of alternate positive and negative sinusoidal waves were
generated and its interference also investigated together with the other three transmission
formats explained in Section 5.1. This signal has the same pattern similar to that of the BPSK-
UWB wave streams and hence it is considered as “sinusoidal BPSK” in this section.
5.2.1. Disturbance Voltage Measurement Setup
The measurement of the conducted disturbance voltage is performed according to the setup
shown in Fig.45. The Telecom-port Impedance Stabilization Network (T-ISN) is defined in
Committee Draft CISPR/I/89/CD for coupling of symmetrical and asymmetrical signals from a
Chapter.5- EMI Measurement Setups and Results
- 84 -
given network. This Committee Draft defines a common-mode impedance of 150 Ω and an
LCL value of 30 dB for a T-ISN and hence the one used in the measurement setup of Fig.45
fulfills these requirements.
Figure 45 Setup for Disturbance Voltage Measurement
The different components (equipment) shown in the setup of Fig.45 are explained as follows:
• EMI-Receiver This is an EMI Test Receiver from Rhode & Schwarz (R&S). It is
compliant to the CISPR 16 standards for measurement equipment.
Measurements were performed using all its three detectors: Peak
Detector (PK), Average Detector (AV) and Quasi-Peak Detector (QP).
• AFG3101: This is an Arbitrary Function generator used to generate waveforms
loaded in to its memory. Different standard and pre-loaded wave forms
can be generated.
• T-IS-: as defined in CISPR/I/89/CD and explained earlier.
• BALU-: Macfarlane balun was used for injecting a purely balanced signal in to
the T-ISN similar to a PLC Modem. The Asymmetrical port of the balun
was terminated with a 50 Ω load.
• Laptop: Both for controlling the frequency of the AFG3101 Function generator
and for retrieving the measured data from the R&S EMI Test Receiver.
• Software: This software (Mess4) is for the synchronization of the AFG3101
Function generator and the EMI Test Receiver, for controlling the
frequency of the Generator, for setting the measurement point by the
EMI Test Receiver and also for retrieving the measured data to the
laptop.
• Saftey-box This is a High Pass Filter for the protection of the AFG3101 Function
Generator from being damaged by the 50 Hz, 220 V line.
Chapter.5- EMI Measurement Setups and Results
- 85 -
5.2.2. Radiation Measurement Setup
The measurement setup for the radiated field measurements is as shown in Fig.46 below.
AFG3101
DPO4034
Oscilloscope for
Observing Waveform
The 2-port wired
Channel
antenna point
2PWNS11
S12
S21
S22
GPIB
Cables
AFG3101
Arbitrary Function Generator
EMI-Receiver Laptop for Measurement
and controlling signal
generation
BALUN
50Ω
termination
Sym
Figure 46 Setup for Radiation Measurement
The following equipment and devices are used for the measurement:
• EMI-Receiver: This is the same R&S EMI Test Receiver described in the
previous Section. Radiated field measurements were performed using all
its three detectors: PK, AV and QP.
• AFG3101: Same as explained in the previous Section.
• DPO4034: A Digital Phosphorous Oscilloscope for observing the transmitted
signals. This is not mandatory but only helps to observe transmissions.
• Antenna: The Magnetic Loop antenna is calibrated for measurements below
30 MHz. Even if it is not placed in a strictly far-field region for all the
frequencies of measurements, such radiated field measurements are
commonly practiced.
• BALU-: The same Macfarlane probe as previously stated. But here the Sym port
of the balun was terminated by 50 Ω and signal injections were
Asymmetrical.
• Laptop Both for controlling the frequency of the AFG3101 Function generator
and for retrieving the measured data from the R&S EMI Test Receiver.
• Software Same as explained in the previous Section.
Chapter.5- EMI Measurement Setups and Results
- 86 -
5.3. Measurement Results
The methodology followed in the measurements is the comparison of the different
interferences (Disturbance Voltage, and radiated Field) between the following four types of
wave forms:
• Un-modulated sinusoidal wave
• BPSK modulated sinusoidal wave
• Un-modulated UWB pulse
• BPSK modulated UWB pulse
Since the carrier-based transmissions employ the sinusoidal signals as carrier signal the
difference in the level of interference between the sinusoidal and the UWB signals is a proof of
an achievable reduction in interference by employing a carrier-less UWB transmission instead
of the carrier-based transmission. Therefore, much attention is not to be given to the magnitude
of interference from each waveform since that depends on the measurement scenario and
network configuration.
5.3.1. Disturbance Voltage Measurement Result
Fig.47 shows the disturbance voltage measurements of the four different waveforms using the
QP detector. The PK and AV detectors also show relatively similar results and hence are not
plotted. Discussions on results are made in Chapter 6.
40
45
50
55
60
65
70
75
0 5e+006 1e+007 1.5e+007 2e+007 2.5e+007 3e+007
Conducted Disturbance Voltage [ dB
µV
]
Figure 47 Measured Results of Disturbance Voltages from the different signals
Chapter.5- EMI Measurement Setups and Results
- 87 -
5.3.2. Radiated Field Measurement Results
In addition to the disturbance voltage measurement described in the previous Section,
measurements of radiated fields were made on a Laboratory Test Bench Powerline channel.
The radiated magnetic field measurement was done using Magnetic Loop Antenna and the
equivalent electric field is computed based on the Antenna Factor and the 51.5 dB Ω
conversion parameter between measured magnetic field and computed electric field.
5.3.2.1. Measurement Points
For the measurement of radiated field, different measurement points along the length of the
channel have been set as antenna points. The results obtained from measurements at these
different points are relatively the same and hence only results from one of these points (P1) are
presented here.
5.3.2.2. The Test-Bench
The channel selected for the measurement of the radiated filed is CH-4 of Section 3.2.3 at a
height of 1.5 m above perfect ground and with asymmetrical signal injection.
5.3.2.3. The Measurement Results
The measured results are displayed in the following figures, discussions and conclusions based
on the figures follow in Chapter 6.
The results shown in Fig.48-Fig.52 show graphs of electric field strengths converted from the
measured radiated magnetic fields using the 51.5 dB Ω as said earlier.
Chapter.5- EMI Measurement Setups and Results
- 88 -
55
60
65
70
75
80
85
90
95
100
0 5e+006 1e+007 1.5e+007 2e+007 2.5e+007 3e+007
Figure 48 Electric Field from un-modulated sinusoidal and un-modulated UWB
45
50
55
60
65
70
75
80
85
90
95
100
0 5e+006 1e+007 1.5e+007 2e+007 2.5e+007 3e+007
Figure 49 Electric Field from un-modulated sinusoidal and BPSK modulated UWB
Chapter.5- EMI Measurement Setups and Results
- 89 -
45
50
55
60
65
70
75
80
85
90
0 5e+006 1e+007 1.5e+007 2e+007 2.5e+007 3e+007
Figure 50 Electric Field from un-modulated UWB and BPSK modulated UWB
45
50
55
60
65
70
75
80
85
90
95
0 5e+006 1e+007 1.5e+007 2e+007 2.5e+007 3e+007
Figure 51 Electric Field from BPSK modulated sinusoidal and BPSK modulated UWB
Chapter.5- EMI Measurement Setups and Results
- 90 -
Electric Field Strength [ dB
µV
/m ]
50
55
60
65
70
75
80
85
90
95
100
0 5e+006 1e+007 1.5e+007 2e+007 2.5e+007 3e+007
Figure 52 Electric Fields from all the four transmissions
- 91 -
6. Discussions and Conclusions
Contents
• Discussions and Conclusions based on results
• Topic proposals for related future researches
6.1. Discussions and Conclusions based on Results
Different sorts of findings from the different mathematical and experimental investigations
have been illustrated in the different Sections and sub-Sections of the Chapters of the
Dissertation about the certainty of the case-in-point: Mitigating EM Interferences of
Powerline Communications using Carrier-less UWB pulses. Full texts of those findings have
been placed where they would maintain the smooth flow of the particular subject discussed at
those sub-sections.
In this Section, therefore, a review of those findings and a discussion of measurement results of
Chapter 5 are summarized. Other points worth mentioning to cement the different components
are also part of the summary made here.
1. It is necessary that the technology of UWB communication system is to be understood
as one based on the definition outlined in FCC 15.502 where either of the 500 MHz or
the 20% fractional BW requirements is fulfilled. The choice between the two
requirements depends on the application area and the limitations related to band of
operations.
2. Signal BW can be increased in either of the following two ways:
a. In carrier-based transmissions: by increasing the number of carriers
b. In carrier-less transmissions: by transmitting very-narrow pulses
3. The main advantage of increased BW is a reduced interference to a co-existing
transmission without compromising the total transmitted power. This is very necessary
in high data rate communications such as broadband PLC.
Chapter.6- Discussions and Conclusions
- 92 -
4. The source of strong un-symmetry inside the Powerline channel is the Distribution
Board of the network. Strong reflections, attenuations and modal conversions take place
due to the strong un-symmetry introduced by the Distribution Board. Therefore, care
should be taken to consider the position of the two communicating ends relative to the
Distribution Board when characterizing Powerline channels.
5. The radius of coverage in wireless UWB transmission is very short (approx. 3 m) due to
the sub-nano range pulse width. Transmission of the pulses over a conductive media
with pulse width compatible to the frequency band of PLC operation, however, is not
characterized by such shortcomings. The plots of received pulses discussed in Section
3.2.4 are for pulses received over 20 m PLC channels.
6. Modulation formats resulting into smooth spectral components are to be used for a
highly reduced level of interferences from UWB transmissions.
7. Results summarized in Table 8 and Fig.41 – Fig.44 and Fig.47 - Fig.52 show that the
reduction in interference is huge if a carrier-less UWB pulse transmission is
implemented with wisely selected modulation formats instead of carrier-based
transmissions.
8. Therefore, the findings of this Dissertation can be further converged in two sentences:
• It is possible to transmit nano-range UWB pulses over PLC channels.
• Reduction of Interferences by more than 15 dB can be achieved over the
PLC band of operations below 30 MHz.
Chapter.6- Discussions and Conclusions
- 93 -
6.2. Topic Proposals for related Future Researches
UWB technology is still undergoing important phase of transformations even for the wireless
applications. The two approaches of achieving that “ultra-wide” band are either increasing the
number of sub-carriers or transmitting sub-nano range pulses in baseband. These are competing
and seemingly un-reconciling approaches towards the common goal: a huge reduction in
interferences between co-existing transmissions of high data rates. To what extent the number
of sub-carriers can be increased also depends on the available band of operation. The PLC
technology is limited to a maximum of 30 MHz range and the need for a reduced interference
level, therefore, requires looking for a solution from any sort of UWB implementations.
Therefore, for the sake of exploiting the potential of UWB pulses for wired channel
applications in general, and for PLC in particular, researches in the following directions are
believed to contribute for the maturity of the approach:
1. Performance measurement of UWB signals on wired channels
2. Methods of Interference Measurement for UWB signals over PLC band of operation
3. Conducted Interference measurements of UWB signals
4. Injections of UWB signals to wired channels
5. Looking for a possibility of expanding the PLC band of operations
6. Algorithmic and Hardware realization of wired UWB
7. Integration of wireless UWB and PLC protocols both for the wireless extension of
existing PLC transmissions and for a wired extension of existing UWB protocols. The
emerging wireless USB (WUSB) in a digital-home requires researches aimed at
integrating UWB technologies and PLC technologies.
- 94 -
References and Bibliography
- 95 -
References and Bibliography
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[BH-B04] Bo Hu, Norman C. Beaulieu, “Accurate Evaluation of Multiple-Access Performance in TH-PPM and TH-BPSK UWB Systems,” IEEE Transactions on Communications, Vol.52, No.10, October 2004.
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NETEX Project Office, DARPA, USA, April 2003 [MZKD99] Manfred Zimmermann, Klaus Dostert, “A Multi-path Signal Propagation Model for the
Powerline Channel in the High Frequency Range,” International Symposium on
Powerline Communications and its Applications (ISPLC), Lancaster, UK, April 1999. [-LAH03] Nikolaus H. Lehmann, Alexander M. Haimovich, “The Power Spectral Density of a Time
Hopping UWB Signal: A Survey,” IEEE Conference on Ultra Wideband Systems and
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Sons, Inc, 2006. [Perez95] Reinaldo Perez, Handbook of Electromagnetic Compatibility, Academic Press, 1995 [Proaki01] John G. Proakis, Digital Communications, 4th ed., pp.205, McGraw-Hill Publications,
2001. [Proph06] Graham Prophet, Powerline’s Other data channel, EDN Europe, May 2006 website:
http://www.edn.com/article/CA6294227.html [Rappa02] Theodore Rappaport, Wireless Communications, 2nd edition, Printice Hall, USA, 2002. [RepJH01] Report: UWB-GPS Compatibility Analysis Project, Strategic Systems Department, The
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high-speed communication channels: channel characterization and a test channel ensemble,” International Journal of Communication Systems, 16:381-400, 2003.
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References and Bibliography
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[XSMG06] X. Shen, M. Guizani, R.C. Qiu, T. Le-Ngoc, Ultra-Wideband wireless communications
and networks, John Wiles & Sons, Ltd, July 2006. [Y-AM03] Yves-Paul Nakache, Andreas F. Molisch, Spectral Shape of UWB Signals Influence of
Modulation Format, Multiple Access Scheme and Pulse Shape, Mitsubishi Electric Research Laboratory (MERL), TR-2003-40, May 2003.
[Y-AM06] Yves-Paul Nakache, Andreas F. Molisch, “Spectral Shaping of UWB Signals for Time-Hopping Impulse Radio,” IEEE Journal on Selected Areas in Communications, Vol.24, No.4, April 2006.
[YSIT05] Yasuo Suzuki, Ichihiko Toyoda, Masahiro Umehira, “Interference Analysis from Impulse Radio UWB Systems Using Simple Signal Models,” Institute of Electronics, Information and Communication Engineers (IEICE) Transactions, Fundamentals, Vol.E88-A No.11, November 2005.
[YWXD07] Yue Wang, Xiaodai Dong, Ivan J. Fair, “Spectral Shaping and NBI Suppression in UWB Communications,” IEEE on Wireless Communications, Vol.6, No.5, May 2007.
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CURRICULUM VITAE Full Name Getahun Mekuria Kuma Date of Birth 01.12.1969 Place of Birth Illubabor, Ethiopia Marital Status Married Permanent Email: [email protected]
Education April 2005 – Sept. 2008: Dr.-Ing. (PhD) in Electrical Engineering
Energietransport und –speicherung, Universität Duisburg-Essen, Duisburg, Germany Title: “Mitigating EMI of Powerline Communication using
Carrier-less UWB pulses” Supervisor: Prof. Dr.-Ing. Holger Hirsch Sept.1997 – Jul.1999 M.Sc.in Electrical Engineering Major in Communication Engineering Addis Ababa University, Addis Ababa, Ethiopia Sept.1987 – Jul.1992 B.Sc.in Electrical Engineering Addis Ababa University, Addis Ababa, Ethiopia
Additional Certifications
CCAI: Cisco Certified Academy Instructor (No. 3266948CCNA) Port Elizabeth Technikon University, Port Elizabeth, South Africa
CCNA: Cisco Certified Network Associate (No. CSO10787298)
Employment History:
Oct.2002 -- Sept.2004 Lecturer Dept. of Electrical and Computer Eng., Faculty of Technology, Addis Ababa University, Addis Ababa, Ethiopia
Mar.2000 -- Aug.2002 Head of Engineering Department Siemens Ethiopia Ltd., Addis Ababa, Ethiopia Oct.1996 -- Nov.1997 Hardware Engineer GATE Private Limited Company, Addis Ababa, Ethiopia Jul.1992 -- Sept.1996: Electrical Distribution Network Design Engineer
Ethiopian Electric Light and Power Authority, Jimma, Ethiopia.
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Publications 1. Getahun Mekuria, Holger Hirsch, “Measurement of Radiated Field from UWB signal over
Powerline Channel, IEEE-EMC Symposium, Detroit, USA, 18-22 August 2008. 2. Getahun Mekuria, Holger Hirsch, “Charakterisierung der elektrischen Unsymmetrie von
komplexen Leitungsnetzen im Zeitbereich,” (German), Elektromagnetische Verträgleichkeit-2008 (EMV2008), Düsseldorf, Germany, February 2008.
3. Getahun Mekuria, Holger Hirsch, “Powerline Communication: Untapped ICT Infrastructure in the
Developing Countries,” World IT Forum (WITFOR 2007), Addis Ababa, Ethiopia, 22 – 24 August 2007 (Invited Paper).
4. Getahun Mekuria, Holger Hirsch, “UWB Pulse Transmission over Powerline Channel”, IEEE-
International Symposium on Powerline Communication and its Applications (ISPLC), Pisa, Italy, 26-28 March 2007.
5. Getahun Mekuria, Holger Hirsch, “EMC Analysis of Pulse Transmission on a Wired Network”,
18th International Wroclaw Symposium and Exhibition on Electromagnetic Compatibility, Wroclaw, Poland, 28-30 June 2006.
6. Getahun Mekuria, Eneyew Adugna, Deva Rajan, “Information Secrecy and Public-Key
Cryptography.” Zede-Journal of Ethiopian Engineers and Architects, Addis Ababa, Ethiopia, Dec 2001.
7. Getahun Mekuria, Eneyew Adugna, Deva Rajan, “Cryptography-an Ideal Solution to Privacy,
Data Integrity & Non-Repudiation.” Zede-Journal of Ethiopian Engineers and Architects, Addis Ababa, Ethiopia, Dec 1999