LINEAR TECHNOLOGYLINEAR TECHNOLOGYLINEAR TECHNOLOGYAUGUST 1998 VOLUME VIII NUMBER 3
, LTC and LT are registered trademarks of Linear Technology Corporation. Adaptive Power, Burst Mode, C-Load,FilterCAD, Hot Swap, Linear View, Micropower SwitcherCAD, No RSENSE, SwitcherCAD and UltraFast are trademarks ofLinear Technology Corporation. Other product names may be trademarks of the companies that manufacture the products.
The LTC1625 CurrentMode DC/DC ControllerEliminates theSense Resistor by ChristopherB. UmmingerIntroductionPower supply designers have a newtool in their quest for ever higherefficiencies. In the past, when design-ing a step-down DC/DC converter,one had to choose between the highefficiency of voltage mode control andthe many benefits of current modecontrol. Although voltage mode con-trol offers high efficiency and a simpletopology, it is difficult to compensate,has poor rejection of input-voltagetransients and does not inherentlylimit output current under fault con-ditions, such as an output shortcircuit. Current mode control over-comes these problems by adding acontrol loop to regulate the inductorcurrent in addition to the output volt-age. Unfortunately, a sense resistor isrequired to measure this current,which adds cost and complexity whilereducing converter efficiency. How-ever, with the new LTC1625 NoRSENSE™ controller, one can enjoy allof the benefits of current mode con-trol without the penalties of using asense resistor.
The LTC1625 is a step-down DC/DC switching regulator controller thatincorporates popular features fromthe LTC1435 and related parts. It iscapable of a wide range of operationwith inputs from 3.7V to 36V. Fixedoutput voltages of 5V and 3.3V can beselected or an external resistive dividercan be used to obtain output voltages
from 1.19V up to nearly the full inputvoltage. The controller provides syn-chronous drive for N-channel powerMOSFETs and retains the advantageof low dropout operation typicallyassociated with P-channel MOSFETs.Burst Mode™ operation maintainsefficiency at low load currents, butcan be overridden to assist second-ary-winding regulation by forcingcontinuous operation. In addition toeliminating the sense resistor, theLTC1625 further reduces the exter-nal parts count by incorporating theoscillator timing capacitor. The oscil-lator frequency can be set to 150kHz,225kHz, or can be injection locked toany frequency between these points.
Current Mode Controlwithout a Sense ResistorHow does one implement currentmode control without a current senseresistor? The answer is to make thestandard power MOSFET switches dodouble duty as current senseelements. By measuring the drain-to-source voltage of a MOSFET operatingin the triode region, one can use theRDS(ON) of the device as a senseresistance. Although conceptuallysimple, this is tricky to implement inpractice because inductor currentinformation can only be obtained whena MOSFET is turned on. Figure 1
continued on page 3
IN THIS ISSUE…COVER ARTICLEThe LTC®1625 Current Mode DC/DCController Eliminates theSense Resistor ............................... 1Christopher B. Umminger
Issue Highlights ............................ 2
LTC in the News… ......................... 2
DESIGN FEATURESLTC1627 Monolithic SynchronousStep-Down Regulator MaximizesSingle or Dual Li-Ion Battery Life..................................................... 5
Jaime Tseng
Clock-Tunable, High Accuracy,Quad 2nd Order, Analog FilterBuilding Blocks ............................. 8Philip Karantzalis
New DC/DC Controller Enables HighStep-Down Ratios ........................ 11Greg Dittmer
VID Voltage Programmer forIntel Mobile Processors ................ 13Peter Guan
Fixed Frequency, 500kHz, 4.5A Step-Down Converter in an SO-8Operates from a 5V Input ............ 16Karl Edwards
48 Volt Hot Swap™ Controller for Negative Voltages ................. 19Henry Yun and Robert Reay
Hot Swapping the PCI Bus Usingthe LTC1643 ............................... 22Robert Reay
Eliminate Pipeline Headaches withNew 12-Bit 3Msps SAR ADC ......... 25Dave Thomas and William C. Rempfer
DESIGN INFORMATIONComponent and MeasurementAdvances Ensure 16-Bit DACSettling Time (Part One) .............. 30Jim Williams
New 16-Bit Bipolar Output DAC inNarrow SO-16 Package ................ 33Hassan Malik
DESIGN IDEAS ....................... 34–37(complete list on page 34)
New Device Cameos ..................... 37
Design Tools ................................ 39
Sales Offices ............................... 40
Linear Technology Magazine • August 19982
EDITOR’S PAGE
LTC in the News…Issue HighlightsThe lead article this month fea-
tures the new LTC1625 No RSENSEDC/DC controller. The LTC1625 is astep-down DC/DC switching regula-tor controller that incorporatespopular features from the LTC1435and related parts. The LTC1625 pro-vides all of the benefits of currentmode control without the penalties ofusing a sense resistor. How does oneimplement current mode control with-out a current sense resistor? Theanswer is to make the standard powerMOSFET switches do double duty ascurrent sense elements.
Several other new power controlproducts are also introduced in thisissue. The LTC1627 is a new additionto a growing family of power manage-ment products optimized for Li-Ionbatteries. Li-Ion batteries, with theirhigh energy density, are becomingthe chemistry of choice for manyhandheld products. The LTC1627monolithic, current mode synchro-nous buck regulator was specificallydesigned to meet these demands. Withits operating supply range of 2.65V to8.5V, it can operate from one or twoLi-Ion batteries as well as 3- to 6-cellNiCd and NiMH battery packs.
LTC1435A is an improved versionof the LTC1435 synchronous DC/DCcontroller. It has all the outstandingfeatures of the LTC1435, with areduced minimum on-time of 300nsor less, and improved noise immunityat low output voltages. With theseimprovements, high performance atoutput voltages down to 1.3V can beachieved with operating frequenciesin excess of 250kHz from input sup-ply voltages above 22V.
To accommodate the precise, yetvaried voltage requirements of thenext generation of microprocessors,LTC introduces the LTC1706-19 VIDvoltage programmer. This device is aprecision, digitally programmableresistive divider that enables an entirefamily of LTC’s DC/DC converterswith onboard 1.19V references to pro-duce output voltages from 1.30V to2.00V in 0.05V steps.
The LT®1506 is a 500kHz mono-lithic buck mode switching regulator,functionally identical to the LT1374but optimized for lower input voltageapplications. Its high 4.5A switch rat-ing makes this device suitable for useas the primary regulator in small tomedium power systems.
In the data conversion arena, wedebut the LTC1412, a new 12-bit3Msps ADC that brings new levels ofperformance and ease of use to highspeed ADC applications. By raisingthe speed of the successive approxi-mation (SAR) method to 3Msps, iteliminates the many drawbacks ofpipelined and subranging ADCs inthat speed range. It is the first clean,simple to use alternative to pipelinedADCs for applications up to 3Msps.
Also in this issue, we premier sev-eral new members of the LTC1068family of universal switched capaci-tor filters. Each product contains fourmatched, low noise, high accuracy2nd order switched capacitor filtersections. High precision, high perfor-mance, quad 2nd order, dual 4thorder or single 8th order filters can bedesigned with an LTC1068 familyproduct.
Rounding out the Design Featuressection, we have four new Hot Swap™controllers. The LTC1640H andLTC1640L provide a simple, flexiblesolution to –48V hot swapping. Thechips allow a board to be safelyinserted into or removed from a livebackplane with a supply voltage from–10V to –80V. The “H” and “L” partsare intended for use with moduleswith active high and active low enableinputs, respectively. The LTC1643 isa Hot Swap controller for PC busapplication. The LTC1643H isdesigned for motherboard applica-tions, whereas the LTC1643L isdesigned for CompactPCI™ applica-tions where the chip resides on theplug-in board.
The Design Information sectionincludes part one of an article onachieving and measuring settlingtimes for 16-bit DACs and theLTC1650, a new 16-bit bipolar out-put DAC in the SO-16 package. TheDesign Ideas section includes a lownoise 33V varactor bias supply, a lownoise 5V to –5V/200mA DC/DC con-verter, a crystal oscillator with 50%duty cycle and complementaryoutputs, a circuit to provide a ground-referenced output voltage that isproportional to a measured current.We conclude with a quintet of NewDevice Cameos.
The July 13, 1998 issue of Busi-ness Week included the magazine’sannual “Business Week Global1000” rankings of the top U.S. andinternational corporations accord-ing to market capitalization. LinearTechnology made both the U.S.and global lists, coming in at num-ber 373 in the United States and768 worldwide. The Company alsomade Equities magazine’s annual“America’s Most Profitable Com-panies” issue at the end of June,ranking at number 100 among themost profitable public companiestraded on the Nasdaq.
On July 21, Linear Technologyannounced its financial results forfiscal year 1998. Said Robert H.Swanson, President and CEO, “Fis-cal 1998 was a good year for LinearTechnology as we achieved recordlevels of sales and earnings.”Specifically, the Company reportednet sales for the fourth quarter of$132,011,000 (a 27% increase overnet sales for the fourth quarter ofthe previous year), net income forthe quarter of $49,503,000 (up 32%from the previous fiscal year’sfourth quarter). For the year, theCompany ’s net sa les were$484,799,000 (up 28% comparedto the previous year), with net in-come of $180,902,000 (up 35%versus the prior year).
CompactPCI is a trademark of the PCI Industrial ComputerManufacturers’ Group.
Linear Technology Magazine • August 1998 3
DESIGN FEATURES
–
+
–
+
–
+Σ
BA×11
TA×11
TK
SW
PGND
ITH
M1
M2
IL
I
t t t
t
V SW
V T
V S ∝
IL
V B
VIN
Figure 1. The LTC1625 senses VDS across the power MOSFETs to infer inductor current.
V TUO)V(
V NI )V(
5 21 81 42 03
5.1 ✓ ✓
8.1 ✓ ✓ ✓
5.2 ✓ ✓ ✓ ✓
3.3 ✓ ✓ ✓ ✓ ✓
0.5 ✓ ✓ ✓ ✓
0.21 ✓ ✓ ✓
Table 1. Some common input and output voltage combinations available
with fOSC = 150kHz
LTC1625, continued from page 1
0
I O(M
AX) (
A)
10
DI_1068_02a. EPS
10050RDS(ON) (mΩ)
0
5
Si4420DY
Si4410DY
Si4412DYSi9936DY
Figure 2. Maximum output current isdetermined primarily by the on-resistanceof the power MOSFETs.
illustrates how this is accomplished inthe LTC1625. The drain-to-source volt-ages of the power MOSFETs are sensedthrough the TK, SW and PGND pins.Sense amplifiers TA and BA measureand amplify these voltages only whenthe respective MOSFET is conductingcurrent. The two resulting signals (VTand VB) are summed to create a sensevoltage ramp (VS) that is proportionalto the inductor current throughoutthe entire switching cycle. A currentcomparator (I) uses the sense voltageand a current threshold voltage (ITH) tocontrol the power MOSFETs. Becausethe LTC1625 fully recovers the induc-tor current information, it behaves
identically to traditional current modecontrollers that use a sense resistor.
An important feature of currentmode control is its ability to inher-ently limit load current simply byrestricting the range of the current-threshold voltage. The maximumallowed current is then inversely pro-portional to the sense resistance. Withthe LTC1625, the power MOSFET on-resistance determines the maximumoutput current. An approximate graphof this relationship is shown in Figure2, along with data points for somepopular Siliconix MOSFETs. The on-resistance is typically guaranteed tobe below a maximum specified by themanufacturer, but it will vary sub-stantially between lots and overtemperature. Thus, a design accom-modating the maximum possibleon-resistance will have a significantlyhigher maximum output current whenthis resistance is lower than normal.To ensure reasonable current levelsin case of a fault, such as an outputshort to ground, the LTC1625 willfold back the current if the outputvoltage falls significantly.
Easing the Layout ChallengeEnsuring that switching transientsdo not interfere with the control loopis a common difficulty with high out-put-current power converters. A greatdeal of effort is often expended inboard layout and component place-ment to subdue coupling effects, suchas pulse pairing or grouping. Elimi-nating the sense resistor and itsassociated signal lines makes this jobeasier. In addition, the LTC1625 usessome internal blanking around theswitching transitions to furtherreduce the possibility of jitter or pulsepairing in the control loop. The resultis a controller that is relativelyimmune to these coupling effects andremarkably easy to lay out success-fully. On the other hand, the additionof blanking increases the minimumtime in which the controller can turnthe top MOSFET on and off. Thisplaces a limit on the minimum dutycycle but does not unduly restrict themaximum input voltage for a givenoutput voltage. Table 1 shows themany common input and output volt-age combinations that the LTC1625can accommodate.
Additional FeaturesThe LTC1625 controller is designedfor synchronous, step-down applica-tions with two N-channel powerMOSFETs. Using an N-channel MOS-FET for the topside switch is morecost-effective than using a P-channeldevice, but it requires a floating top-side driver. This driver is poweredusing an external bootstrap capaci-tor and diode. If the input voltage
Linear Technology Magazine • August 19984
DESIGN FEATURES
+
+
1�
2�
3�
4�
5�
6�
7�
8
16�
15�
14�
13�
12�
11�
10�
9
EXTVCC
SYNC
RUN/SS
FCB
ITH
SGND
VOSENSE
VPROG
VIN
TK
SW
TG
BOOST
INTVCC
BG
PGND
LTC1625+
5V
CSS 0.1µF
RC1
10k
CC1 820pF
CC2 220pF
RF 4.7Ω
CF0.1µF
CIN10µF30V× 2
M1Si4410DY
M2Si4410DY
L1 7µH
R211k1%
R110k1%
COUT100µF10V0.065Ω ESR× 3
CBO.22µF
CVCC4.7µF
VOUT2.5V/5A
VIN5V–28V
DB = CMDSH-3
D1MBRS140T3
*DB
*
Figure 3. 2.5V/5A adjustable-output supply
continued on page 18
drops close to the output voltage, theLTC1625 will begin skipping cycles,leaving the top MOSFET on longer inorder to maintain low dropout opera-tion. Once in dropout, the boostcapacitor is recharged by turning offthe top MOSFET and turning on thebottom MOSFET briefly every tenthcycle. This maintains a duty cyclegreater than 99% in dropout.
Another external part was elimi-nated by incorporating the oscillatortiming capacitor into the LTC1625. Thecontroller runs at a nominal 150kHzfrequency that can be increased 50%by taking the SYNC pin above 1.2V.Frequency-sensitive applications cansynchronize the oscillator by applyinga clock signal between 165kHz and200kHz to this pin.
Providing gate charge to the powerMOSFET switches is one of the mainsources of efficiency loss in switchingregulators. The LTC1625 includes twofeatures to minimize this loss. Nor-mally, both MOSFETs are turned onand off once per oscillator cycle. Whenthe load current drops to near zero,the gate charge required for switchingbecomes a significant fraction of thetotal input current. The LTC1625addresses this problem with BurstMode operation. As the load currentdecreases below approximately onefifth of its maximum value, both MOS-FETs are turned off for a few cycleswhile the output capacitors supportthe load. In this way, gate charge issaved at the expense of fixed frequencyoperation. For applications that requireit, fixed frequency operation can bemaintained by tying the FCB pin low toforce continuous operation at low loadcurrents. This pin can also be used toadaptively override Burst Mode opera-tion when regulating the output of asecondary winding. Another problemwith gate charge is that it is normallydrawn from the high voltage inputsupply. However, power can be savedby obtaining the gate charge from ahigh efficiency external supply(between 5V and 7V) connected to theEXTVCC pin. This could be the outputvoltage of the regulator, a boosted ver-sion of the output or some other systemsupply, if available.
Removing Output VoltageConstraints along with RSENSEThe LTC1625 can be configured for awide variety of output voltages usingthe VPROG and VOSENSE pins. For basic3.3V and 5V applications, one canuse an internal resistive divider withthe output connected directly to theVOSENSE pin. Connect the VPROG pin toINTVCC for a 5V output or to groundfor a 3.3V output. Leaving the VPROGpin open disables the internal resistivedivider and connects the error-amplifier feedback node directly tothe VOSENSE pin. With an externalresistive divider, the LTC1625 iscapable of regulating an output volt-age anywhere between the inputvoltage and the 1.19V internal refer-ence. This represents a significantimprovement over previous currentmode controllers, which were con-strained by the common mode rangeof the current sense signal lines.
Start-up and shutdown of theLTC1625 can be controlled via theRUN/SS pin. This pin is connected toan external capacitor that is chargedfrom an internal 3µA current source.The controller will be shut down ifthis pin is held below 1.4V. After thepin is released, the part stays shutdown while the capacitor is charged,permitting a controlled delay forsequencing the power supply start-up. When the voltage rises above 1.4V,the controller will start but with aclamp on the current threshold volt-
age that limits the load current toapproximately one third of its maxi-mum value. Limiting the load currentreduces the inrush current into theconverter. As the voltage increasesabove 1.4V, the clamp is graduallyraised to its normal value.
Design ExamplesFigure 3 shows the LTC1625 in anapplication supplying a 2.5V outputusing an external feedback divider.Si4410DY MOSFETs from Siliconixallow this converter to deliver up to 5Aof load current. Ripple current is 1.8A(36% of full load) and current limitoccurs around 6A. Note also that theEXTVCC pin is connected to an exter-nal 5V supply. This increases efficiencyby drawing the roughly 7mA gatecharge current from a supply lowerthan VIN.
An efficiency plot of this circuit isshown in Figure 4. An LTC1435 withidentical components in the powerpath is also plotted for comparison. Atlower output voltages such as this,the sense resistor is responsible for anincreasing share of the total powerloss. By eliminating this source ofloss, the LTC1625 is easily able todeliver an efficiency greater than 90%at high load current. The benefit ofreduced I2R loss is readily apparent atthe highest loads. The controller makesa transition to Burst Mode operationbelow around 1.1 A which keeps theefficiency high at moderate loads.
Linear Technology Magazine • August 1998 5
DESIGN FEATURES
IntroductionThe LTC1627 is a new addition to agrowing family of power managementproducts optimized for Li-Ion batter-ies. Li-Ion batteries, with their highenergy density, are becoming thechemistry of choice for many hand-held products. As the demand forlonger battery operating time contin-ues to increase and the operatingvoltages of submicron DSPs andmicrocontrollers decreases, moredemands are placed on DC/DC con-version. The LTC1627 monolithic,current mode synchronous buck regu-lator (Figure 1) was specificallydesigned to meet these demands.
Single and DoubleLi-Ion Cell OperationThe LTC1627, with its operating sup-ply range of 2.65V to 8.5V, can operatefrom one or two Li-Ion batteries aswell as 3- to 6-cell NiCd and NiMHbattery packs. Figure 2 shows a typi-cal discharge voltage profile of a single
LTC1627 Monolithic SynchronousStep-Down Regulator MaximizesSingle or Dual Li-Ion Battery Life
by Jaime Tseng
Li-Ion battery. As shown, a fullycharged single-cell Li-Ion batterybegins the discharge cycle around 4V(it may be slightly higher or lower,depending upon the manufacturer’scharge-voltage specifications). Dur-ing the bulk of the discharge time, thecell produces between 3.5V and 4.0V.Finally, towards the end of discharge,the cell voltage drops quickly below3V. When the voltage drops further,the discharge must be terminated toprevent damage to the battery. A pre-cision undervoltage lockout circuittrips when the LTC1627’s supply volt-age dips below 2.5V, shutting the partdown to only 5µA of supply current.
MaximizingBattery Run TimeThe LTC1627 incorporates power sav-ing Burst Mode operation and 100%duty cycle for low dropout to maxi-mize the battery operating time. InBurst Mode operation, both power
MOSFETs are turned off for increas-ing intervals as the load current drops.Along with the gate-charge savings,unused circuitry is shut down betweenburst intervals, reducing the quies-cent current to 200µA. This extendsoperating efficiencies exceeding 90%to over two decades of output loadrange (see Figure 3). As the batterydischarges, the LTC1627 smoothlyshifts from a high efficiency switch-mode DC/DC regulator to a lowdropout (100% duty cycle) switch. Inthis mode, the voltage drop betweenthe battery input and the regulatoroutput is determined by the load cur-rent, the series resistance of the
–
+
350kHzOSC
2.5VUVLO
PDR
NDR
CONTROL
SYNC
RUN/SS
VFB
0.8V
ITH
SW
VDR
VOUT
VIN
LTC1627
DISCHARGE CAPACITY (mAh)0
1.5
Li-Io
n CE
LL V
OLTA
GE (V
)
2.0
3.0
3.5
4.0
5.0
400 800
2.5
4.5
200 600 1000 1200 1400
2C
1C
0.2C
SANYO 186501300mAhTA = 25°C
OUTPUT CURRENT (A)0.01
70
EFFI
CIEN
CY (%
)
75
80
85
90
100
0.10 1.00
95
L1 = 15µHVOUT = 3.3V
VIN = 5V
Figure 1. LTC1627 block diagram
Figure 2. Typical single-cell Li-Iondischarge curve
Figure 3. Efficiency vs output load current
Linear Technology Magazine • August 19986
DESIGN FEATURES
internal P-channel power MOSFETand the inductor resistance.
The internal power MOSFETswitches provide very low resistanceeven at low supply voltages. Figure 4is a graph of switch resistance vssupply voltage for both switches. TheRDS(ON) is typically 0.5Ω at 5V andonly rises to approximately 0.65Ω at3V, for both switches. This low switchRDS(ON) ensures high efficiency switch-ing as well as low dropout DCcharacteristics at low supply voltages.
ExtendingLow Supply OperationAt low supply voltages, the LTC1627is most likely to be running at highduty cycles or in dropout, where theP-channel main switch is on continu-ously. Hence, the I2R loss is due mainlyto the RDS(ON) of the P-channel MOS-FET. When VIN is below 4.5V, theRDS(ON) of the P-channel MOSFET canbe lowered further by driving its gatebelow ground. The top P-channelMOSFET driver makes use of a floatingreturn pin, VDR, to allow biasing below
GND. A simple charge pump boot-strapped to the SW pin realizes anegative voltage at the VDR pin, asshown in Figure 5. Each time the SWnode cycles from low to high and thenfrom high to low, charge is trans-ferred from C2 to C1 producing anegative voltage at VDR equal in mag-nitude to VIN – (2 • VDIODE). In dropout,when the P-channel MOSFET isturned on continuously, a dropoutdetector counts the number of oscil-lator cycles that the P-channelMOSFET remains on and periodicallyforces a brief off period to allow C1 torecharge. When 100% duty cycle isdesired, VDR can be grounded to dis-able the dropout detector.
Constant-Frequency,Current Mode ArchitectureThe LTC1627 uses a constant-frequency, current mode step-downarchitecture that provides excellentrejection of input line and output loadtransients and also provides cycle-by-cycle current limiting. Input linetransients are rejected by the feed-forward characteristics inherent incurrent mode control. The output loadtransients are rejected by the greatererror-amplifier bandwidth afforded incurrent mode control. In currentmode, the circuit behaves as if therewere a constant current feeding theparallel combination of the outputcapacitor and output load, yieldingonly a 90° rather than a 180° phaselag. This simplifies the feedback-loop design and the circuitry aroundthe error amplifier required forstabilization.
Current mode limits the peak cur-rent cycle-by-cycle, protecting theinternal main switch and synchro-nous rectifier. In extreme cases, whenthe output is shorted to ground, thefrequency of the oscillator is reducedto one-tenth of its nominal frequencyto allow the inductor current time todecay and prevent inductor-currentrunaway. The oscillator’s frequencygradually increases back to its nomi-nal frequency when VFB rises above0.3V.
The internal oscillator is set for afixed switching frequency of 350kHz,allowing the use of small surfacemount inductors. In switching-noise-sensitive applications, the LTC1627can be externally synchronized to fre-quencies of up to 525kHz. Duringsynchronization, Burst Mode opera-tion is inhibited and pulse-skippingmode is used. In this mode, when theoutput load is very low, the currentcomparator remains tripped for morethan one cycle and forces the mainswitch to stay off for the same num-ber of cycles. Increasing the outputload slightly allows constant frequencyPWM operation to resume.
Minimal ExternalComponentsSize is extremely important in mod-ern portable electronics, so theLTC1627 is designed to work with aminimum number of external compo-nents. The loop compensation,current sense resistor and the mainand synchronous switches are inter-nal. An internal catch diode is alsoprovided across the internal synchro-nous switch, eliminating parasiticcurrents or latch-up if the externalSchottky diode is omitted. Only an
Figure 4. RDS(ON) for both switchesvs input voltage
Figure 5. Using a charge pump to bias VDR
Figure 6. Dual-output 1.8V/0.3A and 3.3V/100mA application
INPUT VOLTAGE (V)1
0
R DS(
ON) (
Ω)
0.1
0.3
0.4
0.5
7
0.9
0.2
4 10
0.6
0.7
0.8
2 3 5 6 8 9
TJ = 25˚C
MAIN AND SYNCHRONOUS SWITCH
VDR SWL1
100µFCOUT
LTC1627
VOUT
VINVIN
Linear Technology Magazine • August 1998 7
DESIGN FEATURES
inductor, input and output filtercapacitors and two small resistorsand capacitors are needed to con-struct a high efficiency DC/DCswitching regulator (see Figure 7).The 47pF filter capacitor connectedto the ITH pin (error-amplifier output)filters out switching noise. If the loopcompensation needs to be adjustedfor a specific application, the ITH pincan also be used for externalcompensation.
Auxiliary-Winding ControlUsing the SYNC/FCB PinBesides higher efficiency and lowerswitching noise, synchronous switch-ing provides a means of regulating asecondary flyback winding. In non-synchronous regulators, power must
be drawn from the inductor primarywinding in order to extract powerfrom auxiliary windings. But withcontinuous synchronous operation,power can be drawn from the auxil-iary windings without regard to theprimary output load.
The LTC1627, with its synchro-nous switching and attendantcircuitry, provides the means of easilyconstructing a secondary flybackregulator, as shown in Figure 6. Thisflyback regulator is regulated by thesecondary feedback resistive dividertied to the SYNC/FCB pin. This pinforces continuous operation wheneverit drops below its ground-referencedthreshold of 0.8V. Power can then bedrawn from the secondary flybackregulator whether the main output isloaded or not.
ITH
RUN/SS
VFB
GND
SYNC/FCB
VDR
VIN
SW
LTC1627
CITH47pF
CSS0.1µF
COUT†100µF6.3V
25µH*
R1249k
1%
R280.6k
1%
1
2
3
4 5
6
7
8
VIN ≤ 8.4V
CIN††22µF16V
VOUT3.3V/0.5A
SUMIDA CD54-250AVX TPSC107M006R0150AVX TPSC226M016R0375
+
+*†
††
ITH
RUN/SS
VFB
GND
SYNC/FCB
VDR
VIN
SW
LTC1627
CITH47pF
CSS0.1µF
COUT✝ 100µF6.3V
D1
C10.1µF
15µH*
R1169k
1%
R280.6k
1%
1
2
3
4 5
6
7
8
VIN2.8V–4.5V
CIN††22µF16V
C20.1µF
VOUT2.5V/0.5A
SUMIDA CD54-150ZETEX BAT54SAVX TPSC107M006R0150AVX TPSC226M016R0375
D2
BAT54S**
++
***
†††
Typical Applications
1 or 2 Li-IonStep-Down ConverterFigure 7 is a schematic diagram show-ing the LTC1627 being powered byone or two Li-Ion batteries. All thecomponents shown in this schematicare surface mount and have beenselected to minimize the board spaceand height. The output voltage is setat 3.3V, but is easily programmed toother voltages.
Single Li-IonStep-Down ConverterThe circuit in Figure 8 is intended forinput voltages below 4.5V, making itideal for single Li-Ion battery applica-tions. Diodes D1 and D2 andcapacitors C1 and C2 comprise thebootstrapped charge pump to realizea negative supply at the VDR pin, thereturn pin for the top P-channelMOSFET driver. This allows Figure8’s circuit to maintain low switchRDS(ON) all the way down to the UVLOtrip voltage.
ConclusionThe new LTC1627 monolithicsynchronous buck regulator is a ver-satile, high efficiency, DC/DCconverter that is at home in a widerange of low input voltage applica-tions. Features such as precisionUVLO and optional bootstrapped gatedrive make it particularly well suitedto single-cell Li-Ion power.
Figure 7. Dual lithium-ion to 3.3V/0.5A regulator
Figure 8. Single lithium-ion to 2.5V/0.5A regulator
forthe latest information
on LTC products, visit
www.linear-tech.com
Linear Technology Magazine • August 19988
DESIGN FEATURES
Clock-Tunable, High Accuracy,Quad 2nd Order, Analog FilterBuilding Blocks
by Philip Karantzalis
IntroductionThe LTC1068 product family consistsof four monolithic, clock-tunable filterbuilding blocks. Each productcontains four matched, low noise,high accuracy 2nd order switchedcapacitor filter sections. An externalclock tunes the center frequency ofeach 2nd order filter section. TheLTC1068 products differ only in theirclock-to-center frequency ratio. Theclock-to-center frequency ratio is setto 200:1 (LTC1068-200), 100:1(LTC1068), 50:1 (LTC1068-50) or 25:1(LTC1068-25). External resistors canmodify the clock-to-center frequencyratio. High precision, high perfor-mance, quad 2nd order, dual 4thorder or single 8th order filters can bedesigned with an LTC1068 familyproduct. Designing filters with anLTC1068 product is fully supportedby the FilterCAD™ 2.0 design soft-
ware for Windows®. The internal sam-pling rate of all the LTC1068 devicesis twice the clock frequency. Thisallows the frequency of input signalsto approach twice the clock frequencybefore aliasing occurs. Maximumclock frequency for LTC1068-200,LTC1068 and LTC1068-25 is 6MHzwith ±5V supplies; that for theLTC1068-50 is 2MHz with a single 5Vsupply. For low power filter applica-tions, the LTC1068-50 power supplycurrent is 4.5mA with a single 5Vsupply and 2.5mA with a single 3Vsupply. The LTC1068 products areavailable in a 28-pin SSOP surfacemount package. The LTC1068 (the100:1 part) is also available in a 24-pin DIP package. The following fourcircuits are typical examples ofapplication-specific filters that theLTC1068 products can realize.
LTC1068-200 UltralowFrequency Linear-PhaseLowpass FilterFigure 1 shows an LTC1068-200linear-phase 1Hz lowpass filter sche-matic and Figure 2 shows its gain andgroup delay responses. The clock fre-quency of this filter is 400 times the–3dB frequency (f–3dB or fCUTOFF). Thelarge clock-to-fCUTOFF frequency ratioof this filter is useful in ultralow fre-quency filter applications whenminimizing aliasing errors could bean important consideration. Forexample, the 1Hz lowpass filter shownin Figure 1 requires a 400Hz clockfrequency. For this filter, the inputfrequencies that can generate alias-ing errors are in a band from 795Hz to805Hz (2 × fCLK ±5 × f–3dB). For mostvery low frequency signal-processingapplications, the signal spectrum isless than 100Hz. Therefore, Figure1’s filter will process very low fre-quency signals without significantaliasing errors, since its clock fre-quency is 400Hz and the aliasinginputs are in a small band around800Hz.Windows is a registered trademark of Microsoft Corp.
INV B
HPB/NB
BPB
LPB
SB
NC
AGND
V+
NC
SA
LPA
BPA
HPA/NA
INV A
1
2
3
4
5
6
7
8
9
10
11
12
13
14
INV C
HPC/NC
BPC
LPC
SC
V–
NC
CLK
NC
SD
LPD
BPD
HPD/ND
INV D
28�
27�
26�
25�
24�
23�
22�
21�
20�
19�
18�
17�
16�
15
LTC1068-200RL2 14.3k
R22 15.4k
R32 10k
R34 10k
R24 15.4k
R43 12.4k
R23 10k
R33 12.4k
R11 14.3k
R41 15.4k
R21 12.4k
R31 10k
400kHz
VOUT
5V
VIN
RL1 23.2k
0.1µF
RB3 23.2k
RL3 23.2k
R52 5.11k
R62 9.09k
–5V
R54 5.11k
R64 9.09k0.1µF
GAIN
(dB)
10
0�
–10�
–20�
–30�
–40�
–50�
–60�
–70�
–80�
–90
0.1 10FREQUENCY (Hz)
1
1.0
0.9�
0.8
0.7�
0.6�
0.5�
0.4�
0.3�
0.2�
0.1�
0.0
GROUP DELAY (s)
GAIN
GROUPDELAY
Figure 1. Linear-phase lowpass filter: f–3dB = 1Hz = fCLK/400Figure 2. Gain and group delay responseof Figure 1’s circuit.
Linear Technology Magazine • August 1998 9
DESIGN FEATURES
LTC1068-50 Single 3.3VLow Power Linear-PhaseLowpass FilterFigure 3 is a schematic of an LTC1068-50-based, single 3.3V, low power,lowpass filter with linear phase. Theclock-to-fCUTOFF ratio is 50 to 1 (fCUTOFFis the –3dB frequency). Figure 4 showsthe gain and group delay response.
The flat group delay response in thefilter’s passband implies a linearphase. A linear-phase filter has atransient response with very smallovershoot that settles very rapidly. Alinear-phase lowpass filter is usefulfor processing communication sig-nals with minimum intersymbolinterference in digital communica-tions transmitters or receivers. The
INV B
HPB/NB
BPB
LPB
SB
NC
AGND
V+
NC
SA
LPA
BPA
HPA/NA
INV A
1
2
3
4
5
6
7
8
9
10
11
12
13
14
INV C
HPC/NC
BPC
LPC
SC
V–
NC
CLK
NC
SD
LPD
BPD
HPD/ND
INV D
28�
27�
26�
25�
24�
23�
22�
21�
20�
19�
18�
17�
16�
15
LTC1068-50
RH2 34k
RL2 9.09k
R22 43.2k
R32 43.2k
R42 196k
R44 34.8k
R34 14.3k
R24 16.9k
R43 48.7k
R23 10.7k
R33 12.7k
R11 22.6k
R41 22.6k
R21 20.5k
R31 10k
500kHz
VOUT
3.3V
VIN
RB1 13.3k
RA1 56.2k
0.1µF
1µF
RB3 24.9k
RL3 26.7k
GAIN
(dB)
10
0�
–10�
–20�
–30�
–40�
–50�
–60�
–70�
–80
1k 100kFREQUENCY (Hz)
10k
GAIN
GROUPDELAY
150�
140�
130�
120�
110�
100�
90�
90�
70�
60
GROUP DELAY (µs)
maximum clock frequency for thisfilter is 1MHz with a single 3.3V sup-ply and 2MHz with a single 5V supply.Typical power supply current is 3mAwith a single 3.3V supply and 4.5mAwith a single 5V supply.
LTC1068-25 SelectiveBandpass Filter is ClockTunable to 80kHzFigure 5 shows a 70kHz bandpassfilter based on the LTC1068-25 oper-ating with dual 5V power supplies.The clock-to-center frequency ratio is25 to 1. Figure 6 shows the gainresponse of Figure 5’s bandpass fil-ter. The passband of this filter extendsfrom 0.95 × fCENTER to 1.05 × fCENTER.The stopband attenuation is greaterthan 40dB at 0.8 × fCENTER and 1.15 ×fCENTER. The center frequency can beclock tuned to 80kHz with dual 5Vsupplies and to 40kHz with a single5V supply. With FilterCAD, theLTC1068-25 can be used to realizebandpass filters less selective thanthat shown in Figure 6, which can beclock tuned up to 160kHz with dual5V supplies.
INV B
HPB/NB
BPB
LPB
SB
NC
AGND
V+
NC
SA
LPA
BPA
HPA/NA
INV A
1
2
3
4
5
6
7
8
9
10
11
12
13
14
INV C
HPC/NC
BPC
LPC
SC
V–
NC
CLK
NC
SD
LPD
BPD
HPD/ND
INV D
28�
27�
26�
25�
24�
23�
22�
21�
20�
19�
18�
17�
16�
15
LTC1068-25
RL2 23.2k
RH2 11.3k
R22 4.99k
R32 107k
R34 63.4k
R44 17.4k
R24 7.5k
R43 42.3k
R23 4.99k
R33 59k
R11 29.4k
R41 20.5k
R21 4.99k
R31 24.9k
1.75MHz
VOUT
5V
VIN
RH1 28k
0.1µF
RH3 15.4k
RL3 45.3k
R52 4.99k R62 56.2k
–5V0.1µF
R544.99k
R64 10k
R61 11.3k
R51 4.99k
GAIN
(dB)
10
0�
–10�
–20�
–30�
–40�
–50�
–60�
–70�
–80�
–90
20 100FREQUENCY (kHz)
6030 40 50 70 80 90
Figure 3. Low power, single 3.3V supply, 10kHz, 8th order, linear-phase lowpass filter
Figure 5. 70kHz, 8th order, bandpass filter
Figure 4. Gain and group delayresponse of Figure 3’s filter
Figure 6. Gain response of Figure 5’s filter
Linear Technology Magazine • August 199810
DESIGN FEATURES
INV B
HPB/NB
BPB
LPB
SB
AGND
V+
SA
LPA
BPA
HPA/NA
INV A
1
2
3
4
5
6
7
8
9
10
11
12
INV C
HPC/NC
BPC
LPC
SC
V–
CLK
SD
LPD
BPD
HPD
INV D
24�
23�
22�
21�
20�
19�
18�
17
16
15
14
13�
LTC1068
RL2 12.4k
RH2 55.2k
R22 10k
R32 12.4k
R24 10k
R34 10k
R11 22.4k
R21 10k
R31 12.4k
RL1 12.4k
0.1µF
RH3 38.4k
RL3 16.5k
–5V
0.1µF
5V
R33 12.4k
R23 10k
fCLK = 128 × fOUT
*COSINE WAVE OUT
*SINE WAVE OUT
3�
4�
5�
6�
11�
12�
13�
14
1�
2�
16�
10�
7�
8�
9�
15
CD4520
1Q0�
1Q1�
1Q2�
1Q3�
2Q0�
2Q1�
2Q2�
2Q3
CLK
GND
÷2
÷4
÷8
÷16
÷32
÷64
÷128
÷256
5V
0.47µF
0.1µF
fOUT =fCLK128
PIN 16'S COSINE WAVE OUTPUT IS REFERENCED TOPIN 15'S SINE WAVE OUTPUT
*
LTC1068 Square-Wave-to-Quadrature Oscillator FilterFigure 7 shows the schematic of aLTC1068 based filter that is specifi-cally designed to produce a lowharmonic distortion sine and cosineoscillator from a CMOS-level squarewave input. The reference sine waveoutput of Figure 7’s circuit is on pin15 (BPD on the 24-pin LTC1068 pack-age) and the cosine output is on pin16 (LPD on the 24-pin LTC1068 pack-age). The output frequency of thisquadrature oscillator is the filter’sclock frequency divided by 128. Theoutput of a CMOS CD4520 divide-by-128 counter is coupled with a 0.47µFcapacitor to the input to the LTC1068filter operating with dual 5V powersupplies. The filter’s clock frequencyis the input to the CD4520 counter.
The LTC1068 filter is designed topass the fundamental frequency com-ponent of a square wave and attenuateany harmonic components higherthan the fundamental. An ideal squarewave (50% duty cycle) will have onlyodd harmonics (3rd, 5th, 7th and soon), whereas a typical practical squarewave has a duty cycle less or morethan 50% and will also have evenharmonics (2nd, 4th, 6th and so on).The filter of Figure 7 has a stopbandnotch at the 2nd and 3rd harmonicsfor a square wave input with a fre-quency equal to the filter’s clockfrequency divided by 128. The filter’ssine wave output (pin 15) is 1VRMS fora ±2.5V square wave input and hasless than 0.025% THD (total har-monic distortion) for input frequenciesup to 16kHz and less than 0.1% THD
for frequencies up to 20kHz. The co-sine output (on pin 16, referenced topin 15’s sine wave output) is 1.25VRMSfor a ±2.5V square wave input andhas less than 0.07% THD for frequen-cies up to 20kHz.
The 20kHz frequency limit is dueto the CD4520; with a 74HC typedivide-by-128 counter, sine and cosinewaves up to 40kHz can be generatedwith the LTC1068-based filter ofFigure 7.
The LTC1068Product Family: VersatileFilter Building Blocks,1Hz to 200kHzThe previous four filter examples aretypical of the great variety of filtersthat can be designed with the LTC1068product family. As a general selectionguide, the following is recommended:for low noise and low frequency filtersless than 20kHz, use the LTC1068-200; for low noise and low frequencyfilters less than 40kHz, use theLTC1068; for low power filters up to20kHz operating with single 3V to 5Vsupplies, the LTC1068-50 should bethe choice; and for filters in the fre-quency range 40kHz to 200kHz, usethe LT1068-25. The FilterCAD designsoftware will recommend the appro-priate LTC1068 device for most filterapplications. For some application-specific filter designs, the choice ofthe proper LTC1068 device may notbe obvious and the assistance of aLinear Technology applications engi-neer will be helpful.
LTC1068-BasedSemicustom FiltersFor application-specific filter require-ments of single 8th order or dual 4thorder filters, a customized version ofan LTC1068-family product can beobtained in a 16-pin SO package withinternal thin-film resistors. Clock-to-center frequency ratios higher or lowerthan 200:1, 100:1, 50:1 or 25:1 canalso be obtained. Please contact LTCMarketing for details.
Figure 7. Square-wave-to-quadrature oscillator converter
Authors can be contactedat (408) 432-1900
Linear Technology Magazine • August 1998 11
DESIGN FEATURES
New DC/DC Controller EnablesHigh Step-Down Ratios
by Greg DittmerImportance ofMinimum On-TimeAs processor voltage requirements arepushed lower and lower, input supplyvoltages remain high, forcing DC/DCconverters to operate at lower andlower duty cycles. Since operatingfrequencies also remain high to mini-mize noise and the size of components,the on-time of the topside switch in aconstant-frequency converter mustcontinue to decrease to regulate thelower and lower output voltages. Therequired on-time is given by TON =VOUT/(VIN • f)
Unfortunately, there is a limit tohow small this time can be. In atypical current mode DC/DC con-verter, once the main switch is turnedon at the beginning of each switchingcycle, the speed at which it can beturned off is limited by the responsetime of the current comparator, thetime required for the turn-off com-mand to propagate through the logicand output driver and the timerequired to discharge the capacitanceof the topside gate. These delays addup to a few hundred nanosecondsand constitute the minimum time thetopside switch must stay on duringeach switching cycle. If the maximumVIN and frequency are fixed, this mini-mum on-time sets a lower limit on
output voltage. If an output voltagebelow this limit is required, the onlychoice is to lower the operating fre-quency, which is usually not desirable.Figure 1 shows the on-times requiredwith VIN = 22V as a function of outputvoltage for various frequencies.
Capabilities of the LTC1435The LTC1435 high efficiency synchro-nous DC/DC controller has beenextremely popular for notebook com-puters and other battery-poweredequipment due to its low noise, con-stant-frequency operation and its dualN-channel drive for outstanding highcurrent efficiency without sacrificinglow dropout operation. However, its400ns–500ns minimum on-timerequires lower operating frequencies(
Linear Technology Magazine • August 199812
DESIGN FEATURES
is approaching TON(MIN), there may bea minimum ripple current amplitude,and hence, a maximum inductancenecessary to prevent cycle skipping.Or, expressed differently, the lowerthe inductance, the higher the maxi-mum VIN that can be achieved beforethe minimum on-time is violated andcycle skipping occurs. For mostapplications, 40% ripple not onlyreduces the minimum on-time butalso optimizes efficiency.
22V to 1.6VConverter at 250kHzFigure 4 shows the LTC1435A config-ured in an all N-channel synchronousbuck topology as a 22V to 1.6V/3Aconverter running at 250kHz. The43pF COSC capacitor sets the internal
oscillator frequency at 250kHz andthe 33mΩ sense resistor sets themaximum load current at 3A. For a22V to 1.6V converter, the on-timerequired is:
TON = 1.6/(22 × 250kHz) = 291nsCan the LTC1435A do this? At
maximum VIN the inductor ripple is
∆IL =
VOUT • (1 – VOUT/VIN)
F • L
1.6 • (1 – 1.6/22)250kHz • 4.7µH
= = 1.3A
which is 43% of the 3A maximumload. From Figure 3, 43% ripple givesa minimum on-time of 235ns, whichis well below the 291ns required by
+
+
+
M1Si44412DY
M2 Si44412DY
CIN10µF, 30V×2
L1 4.7µHDB*
RSENSE0.033Ω 35.7k
1%
102k1%
COUT100µF, 6.3V×2
D1MBRS
-140T3
4.7µF
1000pF
0.1µF
COSC 43pF
CSS0.1µF
CC330pF
CC251pF
100pF
VIN
TG
SW
INTVCC
BOOST
BG
PGND
COSC
RUN/SS
ITH
SGND
VOSENSE
13�
16�
14�
12�
15�
11�
10
1
2
3
5
6
VOUT1.60V/3A
VIN4.5V–22V
SENSE– SENSE+
7 8
LTC1435A
*DB = CMDSH-3CENTRAL(516) 435-1110
RC10k
+
+ +
M1Si4410DY
M2 Si4410DY
CIN10µF, 30V×2
L1 3.3µHDB*
RSENSE0.015Ω
COUT820µF4V×2
D1MBRS
-140T3
4.7µF
1000pF
0.22µF
COSC 43pF
CSS0.1µF
CC1000pF
CC2220pF
51pF
VIN
TG
SW
INTVCC
BOOST
BG
PGND
COSC
RUN/SS
ITH
SGND
VOSENSE
13�
16�
14�
12�
15�
11�
10
1
2
3
5
6
VOUT1.30V–2.00V/7A
VIN4.5V–22V
SENSE– SENSE+
7 8
LTC1435A
*DB = CMDSH-3CENTRAL(516) 435-1110
RC10k
RF4.7Ω
CF0.1µF
VCC
FB
VID0
VID1
VID2
VID3 GND
SENSE
LTC1706-19
FROM µP
3
5
7 8 1 2 4
6
this application, so no cycle skippingwill occur. If a 10µH inductor is used,the ripple amplitude drops to 0.6A or20% and the minimum on-timeincreases to 280ns. This does notprovide much margin below the 291nson-time required, and thus the 4.7µHinductor is a better choice.
Intel Mobile ProcessorVID Power ConverterFigure 5 shows the LTC1435A usedwith an LTC1706-19 to implement anIntel Mobile Pentium® II ProcessorVID power converter. This DC/DCconverter provides digitally selectableoutput voltages over the range of 1.3Vto 2.0V in 50mV increments at 250kHzand a 7A load current. The selectableoutput voltage is implemented byreplacing the conventional feedbackresistor network with the LTC1706-19, which provides the appropriatefeedback resistor ratios internally (seethe accompanying article in thisissue). The proper ratio is selectedwith the 4-bit digital input pins.
ConclusionThe LTC1435A retains all the out-standing features of the LTC1435,such as constant-frequency opera-tion, dual N-channel MOSFET driveand low dropout, while addingenhancements such as reducedminimum on-time and improved per-formance at low output. With theseenhancements, the LTC1435A is aperfect fit for notebook computersand battery-powered equipmentrequiring high frequency, low dutycycle DC/DC converters.Pentium is a registered trademark of Intel Corp.
Figure 4. LTC1435A 22V to 1.6V/3A converter (f = 250kHz)
Figure 5. Intel Mobile Pentium II processor VID power converter
Linear Technology Magazine • August 1998 13
DESIGN FEATURES
VID Voltage Programmerfor Intel Mobile Processors by Peter Guan
Microprocessor manufacturers’relentless push for higher speed andlower power dissipation, especially inareas of mobile laptop computer pro-cessors, is forcing supply voltages tothese processors to a level previouslythought impossible or impractical. Infact, the supply voltage has becomeso critical that different microproces-sors demand different yet precisesupply voltage levels in order to func-tion optimally.
To accommodate this new gene-ration of microprocessors, LTCintroduces the LTC1706-19 VID(voltage identification) voltage pro-grammer. This device is a precision,digitally programmable resistivedivider designed for use with an en-tire family of LTC’s DC/DC converterswith onboard 1.19V references. Theseconverters include the LTC1433,LTC1434, LTC1435, LTC1435A,LTC1436, L TC1438, LTC1439,LTC1538-AUX, LTC1539 andLTC1624. (Consult the factory forfuture compatible DC/DC converter
+
+ +
M1Si4410DY
M2 Si4410DY
CIN10µF, 30V×2
L1 3.3µHDB*
RSENSE0.015Ω
COUT820µF4V×2
D1MBRS
-140T3
4.7µF
1000pF
0.22µF
COSC 43pF
CSS0.1µF
CC1000pF
CC2220pF
51pF
VIN
TG
SW
INTVCC
BOOST
BG
PGND
COSC
RUN/SS
ITH
SGND
VOSENSE
13�
16�
14�
12�
15�
11�
10
1
2
3
5
6
VOUT1.30V–2.00V/7A
VIN4.5V–22V
SENSE– SENSE+
7 8
LTC1435A
*DB = CMDSH-3
RC10k
RF4.7Ω
CF0.1µF
VCC
FB
VID0
VID1
VID2
VID3 GND
SENSE
LTC1706-19
FROM µP
40k
VCC
VID0
40k
VCC
VID1
40k
VCC
VID2
40k
VCC
VID3
RFB2SWITCH-CONTROL
LOGIC
RFB113k
RFB2
SENSE
FB
SGND
VCC
Figure 1. Intel Mobile Pentium II processor VID power converter
Figure 2. LTC1706-19 block diagram
Linear Technology Magazine • August 199814
DESIGN FEATURES
products.) The LTC1706-19 is fullycompliant with the Intel mobile VIDspecifications and comes in a tinySO-8 package. Four digital pins areprovided to program output voltagesfrom 1.3V to 2.0V in 50mV steps withan accuracy of ±0.25%.
Figure 1 shows a VID-programmedDC/DC converter for an Intel mobileprocessor that uses the LTC1435Aand LTC1706-19 to deliver 7A of out-put current with a programmable VOUTof 1.3V to 2.0V from a VIN of 4.5V to22V. Simply connecting the LTC1706-19’s FB and SENSE pins to theLTC1435A’s VOSENSE and SENSE–
pins, respectively, closes the loopbetween the output voltage sense andthe feedback inputs of the LTC1435Aregulator with the appropriate resis-tive divider network, which iscontrolled by the LTC1706-19’s fourVID input pins.
Figure 2 shows a simplified blockdiagram of the LTC1706-19. A 40kresistor in series with a diode fromVCC pulls up each VID input pin.Therefore, the VID pin must begrounded or driven low to produce adigital low input, whereas a digitalhigh input can be generated by eitherfloating the VID pin or connecting it toVCC. Series diodes from VCC areincluded to prevent the inputs frombeing damaged or clamped by apotential higher than VCC. This allowsthe LTC1706-19 to be fully TTL com-patible and operational over a VIDinput voltage range that is muchhigher than VCC. When all the inputsare high, the LTC1706-19 has a typi-
cal quiescent current of 0.1µA fromVCC, because all active devices areturned off. However, due to the pull-up resistors on each of the VIDprogramming inputs, each groundedinput contributes approximately68µA, (VCC – 0.6)/40k of supply cur-rent in a 3.3V system.
The top feedback resistor in theblock diagram, RFB1, connectedbetween SENSE and FB, is a 15kresistor whose value is not modifiedby the state of the VID program inputs.The bottom feedback resistor, RFB2,however, is modified by the four VID
7�
8�
1�
2
VID0�
VID1�
VID2�
VID3
6
5
SENSE
FBGND
VCC
4
3LTC1706-19
10µF
VCC 2.7V–5.5V
+
+
1�
2�
3�
4
8�
7�
6�
5
SENSE–
ITH/RUN
VFB
GND
VIN
BOOST
TG
SW
LTC1624
470pF
6.8k
100pF
1000pF
0.1µFRSENSE0.05Ω
Si4412DY
10µH
MBRS340T3
CIN22µF35V×2
VOUT1.3V–2.0V
COUT100µF10V×2
VIN4.8V–20V
inputs. The precision of the ratio be-tween RFB2 and RFB1 results in a±0.25% output accuracy.
Table 1 shows the VID inputs andtheir corresponding output voltages.VID3 is the most significant bit (MSB)and VID0 is the least significant bit(LSB). When all four inputs are low,the LTC1706-19 sets the regulatoroutput voltage to 2.00V. Eachincreasing binary count is equivalentto decreasing the output voltage by50mV. Therefore, to obtain a 1.30Voutput, the three MSBs are left float-ing while only VID0 is grounded. Incases where all four VID inputs aretied high or left floating, such as whenno processor is present in the system,a regulated 1.25V output is gener-ated at VSENSE.
Figure 3 shows a combination ofthe LTC1624 and the LTC1706-19configured as a high efficiency step-down switching regulator with aprogrammable output of 1.3V to 2.0Vfrom an input of 4.8V to 20V. Usingonly one N-channel power MOSFET,the two SO-8 packaged LTC partsoffer an extremely versatile, efficient,compact regulated power supply.
edoC 3DIV 2DIV 1DIV 0DIV tuptuO
0000 DNG DNG DNG DNG V00.2
1000 DNG DNG DNG taolF V59.1
0100 DNG DNG taolF DNG V09.1
1100 DNG DNG taolF taolF V58.1
0010 DNG taolF DNG DNG V08.1
1010 DNG taolF DNG taolF V57.1
0110 DNG taolF taolF DNG V07.1
1110 DNG taolF taolF taolF V56.1
0001 taolF DNG DNG DNG V06.1
1001 taolF DNG DNG taolF V55.1
0101 taolF DNG taolF DNG V05.1
1101 taolF DNG taolF taolF V54.1
0011 taolF taolF DNG DNG V04.1
1011 taolF taolF DNG taolF V53.1
0111 taolF taolF taolF DNG V03.1
Table 1. VID inputs and corresponding output voltages
Figure 3. High efficiency SO-8, N-channel switching regulator with programmable output
Linear Technology Magazine • August 1998 15
DESIGN FEATURES
Figure 4 shows the LTC1436A-PLLand the LTC1706-19, a combinationthat yields a high efficiency low noisesynchronous step-down switchingregulator with programmable 1.3V to2V outputs and external frequencysynchronization capability.
Besides the LTC family of 1.19V-referenced DC/DC converters, theLTC1706-19 can also be used to pro-
+
+ +
M1Si4412DY
M2 Si4412DY
CIN22µF, 35V×2
L13.3µH
DB*
RSENSE0.02Ω
COUT100µF10V×2
D1MBRS
-140T3
4.7µF
1000pF
0.22µF
COSC 39pF
CSS0.1µF
CC510pF
100pF
VIN
TGL
TGS
SW
INTVCC
BOOST
BGL
PGND
COSC
RUN/SS
ITH
SGND
VOSENSE
18�
21�
19
20�
17�
22�
16�
15
2
3
4
6
8
VOUT1.30V–2.00V/5A
VIN4.5V–22V
SENSE– SENSE+
9 10
LTC1436A-PLL
*DB = CMDSH-3
RC10k
VCC
FB
VID0
VID1
VID2
VID3 GND
SENSE
LTC1706-19
FROM µP
M3IRLML2803
PLL LPF PLLIN
10k 0.1µF
EXTERNALFREQUENCY
SYNCHRONIZATION
1 24
+
7�
8�
1�
2
VID0�
VID1�
VID2�
VID3
1�
2�
3�
4
8�
7�
6�
5
SHDN
VIN
GND
FB
IPOS
INEG
GATE
COMP
6
5
SENSE
FBKGND
VCC
4
3LTC1706-19
1µF
VCC 3.3V
VIN12V
10pF7.5k
1000pF
24µF
5.1Ω
220µF
IRFZ24
3.3V
VOUT1.27V–2.03IN 50.8mV STEPS
LT1575
gram the output voltages of regula-tors with different onboard references.Figure 5 shows the LTC1706-19programming the output of theLT1575, an UltraFast™ transientresponse, low dropout regulator thatis ideal for today’s power-hungry desk-top microprocessors. However, sincethe LT1575 has a 1.21V referenceinstead of a 1.19V reference, the out-
put will range from 1.27V to 2.03V insteps of 50.8mV.
The LTC1706-19 is the ideal com-panion chip to provide precise,programmable low-voltage outputs foran entire family of LTC DC/DC con-verters. Its compact size, compatibilityand high accuracy are just the rightfeatures for today’s portable electronicequipment.
Figure 4. High efficiency, low noise, synchronous step-down switching regulator with programmable output and external synchronization
Figure 5. UltraFast transient response, low dropout regulator with adjustable output voltage
Linear Technology Magazine • August 199816
DESIGN FEATURES
Fixed Frequency, 500kHz, 4.5AStep-Down Converter in an SO-8Operates from a 5V Input by Karl EdwardsIntroductionThe LT1506 is a 500kHz monolithicbuck mode switching regulator, func-tionally identical to the LT1374 butoptimized for lower input voltage ap-plications. Its high 4.5A switch ratingmakes this device suitable for use asthe primary regulator in small tomedium power systems. The smallSO-8 footprint and input operatingrange of 4V to 15V is ideal for localonboard regulators operating from5V or 12V system supplies. The 4.5Aswitch is included on the die, alongwith the necessary oscillator, controland logic circuitry to simplify design.The part’s high switching frequencyallows a considerable reduction inthe size of external components, pro-viding a compact overall solution.
The LT1506 is available in stan-dard 7-pin DD and fused-lead SO-8packages. It maintains high efficiencyover a wide output current range bykeeping quiescent supply current to4mA and by using a supply-boostcapacitor to saturate the power switch.The topology is current mode for fasttransient response and good loop sta-bility. Full cycle-by-cycle short-circuitprotection and thermal shutdown areprovided. Both fixed 3.3V and adjust-able output voltage parts are available.
5V to 3.3V Buck ConverterThe circuit in Figure 1 is a step-downconverter suitable for use as a localregulator to supply 3.3V logic from a5V power bus. The high efficiency,shown in Figure 2, removes the needfor bulky heat sinks or separate powerdevices, allowing the circuit to beplaced in confined locations. Sincethe boost circuit only needs 3V tooperate, the boost diode can still beconnected to the output, improvingefficiency. Figure 1’s circuit showsthe shutdown pin option. If this pin is
pulled to a logic low, the output isdisabled and the part goes into shut-down mode, reducing supply currentto 20µA. An internal pull-up ensurescorrect operation when the pin is leftopen. The SYNC pin, an option for theDD package, can be used to synchro-nize the internal oscillator to a systemclock. A logic-level clock signal appliedto the SYNC pin can synchronize theswitching frequency in the range of580kHz to 1MHz.
Current SharingMultiphase SupplyThe circuit in Figure 3 uses multipleLT1506s to produce a 5V, 12A powersupply. There are several advantagesto using a multiple switcher approachcompared to a single larger switcher.The inductor size is considerablyreduced. Inductor size is proportional
to the energy that needs to be storedin the core. Three 4A inductors storeless energy (1/2Li2) than a single 12Acoil, so they are much smaller. Inaddition, synchronizing three con-verters 120° out of phase with eachother reduces input and output ripplecurrents. This reduces the ripple rat-ing, size and cost of the filtercapacitors.
Current Sharing/Split Input SuppliesCurrent sharing is accomplished byconnecting the VC pins to a commoncompensation capacitor. The outputof the error amplifier is a gm stage, soany number of devices can be con-nected together. The effective gm ofthe composite error amplifier is theproduct of the individual devices. InFigure 3, the compensation capaci-tor, C4, has been increased by 3×.Tolerances in the reference voltagescause small offset currents to flowbetween the VC pins. The overall effectis that the loop regulates the outputat a voltage somewhere between theminimum and maximum referencesof the devices used. Switch-currentmatching between devices will betypically better than 300mA over thefull current range. The negativetemperature coefficient of the VC-to-switch-current transconductanceprevents current hogging.
BOOST
LT1506-3.3
VINOUTPUT3.3V4A
INPUT5V
1506 TA01
C20.68µF
CC1.5nF
D1MBRS330T3
C1100µF, 10VSOLIDTANTALUM
C310µF TO
50µFCERAMIC
D21N914
L15µH
VSW
SENSESHDN
OPENOR
HIGH= ON GND VC
+
+EF
FICI
ENCY
(%)
0 4LOAD CURRENT (A)
21 3
90�
85�
80�
75�
70
Figure 1. 5V to 3.3V step-down converter
Figure 2. Efficiency vs load current forFigure 1’s circuit
Linear Technology Magazine • August 1998 17
DESIGN FEATURES
A common VC voltage forces eachLT1506 to operate at the same switchcurrent, not at the same duty cycle.Each device operates at the duty cycledefined by its input voltage. This is auseful feature in a distributed powersystem. The input voltage to eachdevice could vary due to drops across
the backplane, copper losses, con-nectors and so on. The common VCsignal ensures that loading is stillshared between the devices.
SynchronizedRipple CurrentsA ring counter generates three syn-chronization signals at 600kHz, 33%duty cycle, phased 120° apart. Thesync input will operate over a widerange of duty cycles, so no furtherpulse conditioning is needed. At fullload, each device’s input ripple cur-rent is a 4A trapezoidal wave at600kHz, as shown in Figure 4. Sum-ming these waveforms gives theeffective input ripple for the completesystem. The resultant waveform,shown at the bottom of Figure 4,remains at 4A but its frequency hasincreased to 1.8MHz. The higher fre-quency eases the requirements onthe value of input filter without the 3×increase in ripple current rating thatwould normally occur. Although onlya single input capacitor is required,practical layout restrictions usuallydictate an individual capacitor at eachdevice. Figure 5 shows the outputripple current waveforms. The result-ant 1.8MHz triangular waveform hasa maximum amplitude of 350mA atan input voltage of 10V. This issignificantly lower than would beexpected for a 12A output. Interest-ingly, at inputs of 7.6V and 15V, the
++
C468nF25V
C1, C3: MARCON THCS50E1E106ZD1: ROHM RB051L-40D2: 1N914L1: DO3316P-682
+ C3C10µF25V
C2C330nF10V
D1C
D2C
1506 F15
L1C6.8µH
+ C110µF25V
5V12AR1
5.36k1%
R24.99k1%
1.8MHz
3-BIT RINGCOUNTER
+
+ C3B10µF25V
C2B330nF10V
D1B
D2BL1B6.8µH
+
+ C3A10µF25V
INPUT6V TO 15V
C2A330nF10V
D1A
D2AL1A6.8µH
VC SYNC SW GND
LT1506-SYNC
VIN BOOST FBVC SYNC SW GND
LT1506-SYNC
VIN BOOST FBVC SYNC SW GND
LT1506-SYNC
VIN BOOST FB
TIME
CURR
ENT
TIME
CURR
ENT
TIME
CURR
ENT
TIME
CURR
ENT
PHASE 1
PHASE 2
PHASE 3
TOTAL
TIME
CURR
ENT
TIME
CURR
ENT
TIME
CURR
ENT
TIME
CURR
ENT
PHASE 1
PHASE 2
PHASE 3
TOTAL
theoretical summed output ripplecurrent cancels completely. To reduceboard space and ripple voltage, C1and C3 are ceramic capacitors. Loopcompensation capacitor C4 must beadjusted when using ceramic outputcapacitors, due to the lack of effectiveseries resistance (ESR). The typical
Figure 3. Current-sharing 5V/12A supply
Figure 4. Input current Figure 5. Output current
Linear Technology Magazine • August 199818
DESIGN FEATURES
tantalum compensation value of 1.5nFis increased to 22nF (×3) for theceramic output capacitor. If synchro-nization is not used and the internaloscillators free run, the circuit willoperate correctly, but ripple cancella-tion will not occur. Input and outputcapacitors must be ripple rated forthe individual output currents.
Redundant OperationThe circuit shown in Figure 3 is faulttolerant when operating at less than8A of output current. If one powerstage fails open circuit, the outputwill remain in regulation. The feed-back loop will compensate by raisingthe voltage on the VC pin, increasingthe switch current of the two remain-ing devices.
5V to 3.3V at 2.5Aon 0.25in2 of board space,0.125in HighIn many space-sensitive applications,the component that dominates bothboard area and overall height is theinductor. One of the factors affectinginductor value choice is maximumripple current. Using the high cur-rent switch rating of the LT1506,
higher ripple current can be toler-ated, allowing the use of small, lowvalue, high current inductors. Aceramic output capacitor also reducesboard area and improves voltageripple. Using Figure 1’s circuit withthe SO-8 LT1506 and the componentchanges in Table 1, a very small, lowprofile, step-down converter can beimplemented.
Conclusion The LT1506 is a compact, easy touse, monolithic switcher. The inter-nal 4.5A switch covers a wide range ofmedium power applications. Its inputoperating range of 4V to 15V andavailability in SO-8 or DD packagesmake it ideal for very space-efficient,local onboard DC/DC converters.
traP eulaVV /rodne
#traP
3C,1C V01,Fµ22nikoT
Z622GZ0121
CC Fn22
1L Hµ2.2adimuS2R234DC
Table 1. Component changes for a low profileversion of Figure 1’s circuit
A circuit demonstrating the wideoutput range of the LTC1625 is shownin Figure 5. This application usesSi4412DY MOSFETs to deliver a 12Voutput at up to 2.2A. Note that theSYNC pin is tied high for 225kHzoperation in order to reduce theinductor size and ripple current.
1
EFFI
CIEN
CY (%
)
100�
95�
90�
85�
80
DI_1068_02a. EPS
3LOAD CURRENT (A)
2 4 50
VIN = 20VVOUT = 2.5V
LTC1625
LTC1435 +
+
1�
2�
3�
4�
5�
6�
7�
8
16�
15�
14�
13�
12�
11�
10�
9
EXTVCC
SYNC
RUN/SS
FCB
ITH
SGND
VOSENSE
VPROG
VIN
TK
SW
TG
BOOST
INTVCC
BG
PGND
LTC1625+
CSS 0.1µF
RC1
10k
CC11nF
CC2 220pF
RF 4.7Ω
CF0.1µF
CIN22µF35V× 2
M1Si4412DY
M2Si4412DY
L1* 39µH
R235.7k1%
R13.92k1%
COUT100µF16V0.030Ω ESR
CBO.1µF
CVCC4.7µF
VOUT12V/2.2A
VIN12V–28V
L1 = SUMIDA CDRH127-390MCDB = CMDSH-3
INTVCC
D1MBRS140T3
**DB
***Figure 4. Efficiency vs load current
Figure 5. 12V/2.2A adjustable-output supply
ConclusionThe LTC1625 step-down DC/DC con-troller offers true current mode controlwithout the expense and difficulty ofusing a sense resistor. Popular fea-tures from Linear Technology’s othercontrollers, such as fixed frequencyoperation, N-channel MOSFET drive,Burst Mode operation, soft-start andoutput voltage programming make
this controller useful in a variety ofapplications. By eliminating the powerloss in the sense resistor, even higherefficiencies can be achieved than werepreviously possible, making theLTC1625 an excellent choice forDC/DC converter designs requiringthe highest performance.
LTC1625, continued from page 4
Authors can be contactedat (408) 432-1900
Linear Technology Magazine • August 1998 19
DESIGN FEATURES
48 Volt Hot Swap Controllerfor Negative VoltagesIntroductionAs supply voltages for PC boards con-tinue to drop, designers face thedifficult task of minimizing the volt-age drops through distributed powersystems. At operating voltages of 3.3Vor lower, the voltage drops acrosspower busses, connector pins andinrush control circuitry can cause asupply voltage to drop out of toler-ance. A solution to this problem is todistribute power at a high voltage,commonly 48V, and then step thevoltage down to the final desired valueon each board in the system, usingpower modules.
Most 48V power modules requirean input bypass capacitor with a typi-cal value of hundreds of microfarads.When the board is hot-plugged into alive 48V power rail, the input capaci-tor can draw huge inrush currents asit charges. The inrush current cancause permanent damage to theboard’s components and createglitches on the system power supplythat can make the system functionimproperly.
The LTC1640H and LTC1640L pro-vide a simple, flexible solution to –48Vhot-swapping problems. The chipsallow a board to be safely inserted intoor removed from a live backplane witha supply voltage from –10V to –80V.They feature programmable inrushcurrent control, programmableundervoltage and overvoltage protec-
by Henry Yunand Robert Reay
tion, a programmable electroniccircuit breaker and direct power-mod-ule-enable control.
Power Supply ControlA typical LT1640 application is shownin Figure 1. The input voltage of thepower module on a circuit board iscontrolled by gradually increasing thegate voltage of the external N-channelpass MOSFET (Q1) in the power path.R1 provides current fault detectionand R2 prevents high frequencyoscillation. Resistors R4, R5 and R6provide undervoltage and overvoltagesensing. Resistor R3 and capacitorC2 act as a feedback network to accu-rately control the inrush current. Thewaveforms are shown in Figure 2. Theinrush current can be calculated withthe following equation:
IINRUSH = (45µA × CL)/ C2where CL is the total load capacitance.
Resistor R3 helps keep Q1 off whenthe power pins first make contact.When the power pins make contact,they bounce several times. While thecontacts are bouncing, the LT1640senses an undervoltage condition andthe GATE is immediately pulled low.
Once the power pins stop bounc-ing, the GATE pin starts increasinguntil when Q1 turns on and the GATEvoltage is held constant by the feed-back network of R3 and C2. When the
DRAIN voltage has finished increas-ing, the voltage on the GATE pin thenrises to its final value.
Electronic Circuit BreakerThe LT1640 features an electroniccircuit-breaker function that protectsagainst short circuits or excessivesupply currents. By placing a sense
+
VEE
VDD
LT1640L
SENSE
C133nF25V
C30.1µF100V
C4100µF100V
C5100µF16V
Q1IRF530
R210Ω5%
R310k5%
C23.3nF100V
R4562k
1%
R59.09k
1%
R610k1%
R10.02Ω
5%
4
OV
GND
3
2
–48V
OV = 71V
UV = 37V UV
5 6
8
7
1
GATE DRAIN
LUCENTJW050A1-E
PWRGD
95V
1640 TA01
8
7
6
5
1
2
4
+
VOUT+
SENSE+
TRIM
SENSE–
VOUT–
VIN+
ON/OFF
VIN–
Figure 1. Typical LT1604 application with a power module
Figure 2. Inrush-current control waveforms
Figure 3. Short-circuit protection waveforms
Linear Technology Magazine • August 199820
DESIGN FEATURES
resistor between the VEE and SENSEpins, the circuit breaker will be trippedwhenever the voltage across the senseresistor is greater than 50mV for morethan 3µs, as shown in Figure 3.
When the circuit breaker trips, theGATE pin is immediately pulled toVEE and the external N-channel MOS-FET is turned off. The GATE pin willremain low until the circuit breaker isreset by pulling UV low then high orcycling power to the part. A circuitthat automatically resets the circuitbreaker after a current fault is shownin figure 4.
Transistors Q2 and Q3, along withR7, R8, C4 and D1, form a program-mable one-shot circuit. Before a shortoccurs, the GATE pin is pulled highand Q3 is turned on, pulling node 2 toVEE. Resistor R8 turns off Q2. Whena short occurs, the GATE pin is pulledlow and Q3 turns off. Node 2 starts tocharge C4, and Q2 turns on, pullingthe UV pin low and resetting thecircuit breaker. As soon as C4 is fully
charged, R8 turns off Q2, UV goeshigh and the voltage on the GATEstarts to increase. Q3 turns back onand quickly pulls node 2 back to VEE.Diode D1 clamps node 3 one diodedrop below VEE. The duty cycle is setto 10% to pr event Q1 f romoverheating.
Undervoltage andOvervoltage DetectionThe UV (3) and OV (2) pins can beused to detect undervoltage and over-voltage conditions at the power supplyinput. The UV and OV pins are inter-nally connected to analog comparatorswith 20mV of hysteresis. When theUV pin falls below its threshold or theOV pin rises above its threshold, theGATE pin is immediately pulled low.The GATE pin will be held low untilUV is high and OV is low.
The undervoltage and overvoltagetrip voltages can be programmed usinga 3-resistor-divider, as shown inFigure 5.
VEE
VDD
LT1640L PWRGD
SENSE
C133nF25V
C41µF
100V C3100µF100V
Q1IRF530
R210Ω5%R8
510k5%
R310k5%
C23.3nF100V
R4562k
1%
R71M5%
R59.09k
1%
R610k1%
Q3ZVN3310
Q22N2222
D11N4148
R10.02Ω
5%
4
3
2OV
GND
–48V
UV
5 6
8
7
1
GATE DRAIN
1640 F10a
3
2
+
With R4 = 562k, R5 = 9.09k and R6= 10k, the undervoltage threshold isset to 37V and the overvoltage thresh-old is set to 71V.
PWRGD/PWRGD OutputThe PWRGD/PWRGD output can beused to directly enable a power mod-ule when the input voltage to themodule is within tolerance. TheLT1640H has a PWRGD output formodules with an active-high enableinput, and the LT1640L has a PWRGDoutput for modules with an active-low enable input.
When the DRAIN pin of theLT1640H is more than VPG (1.4V)above VEE (see Figure 6), internaltransistor Q3 is turned off and R7and Q2 clamp the PWRGD pin onediode drop (~0.7V) above the DRAINpin. Transistor Q2 sinks the module’spull-up current and the module turnsoff.
VEE
VDD
LT1640HLT1640L
R4
R5
R64
1640 F11
OV
GND
3
2
–48V
UV
8
VUV = 1.223R4 + R5+ R6
R5 + R6( )VOV = 1.223
R4 + R5+ R6R6( )
+
VEE
VDDLT1640H
SENSE
C1
C3
Q1
R2R3
C2
R4
R5
R6
R1
4
3
2OV
GND
–48V
UV
5 6
8
1
7
GATE
ACTIVE HIGHENABLE MODULE
VOUT+
VOUT–
VIN+
1640 F12
VIN–
PWRGD
DRAIN–
+
VEE
R75k
Q2
Q3
ON/OFF
+
–1.4V
Figure 4a. This circuit resets the circuit breaker after a current fault.Figure 4b. Waveforms of Figure 4a’s circuit
Figure 5. Undervoltage and overvoltagesensing Figure 6. Active-high enable module
Linear Technology Magazine • August 1998 21
DESIGN FEATURES
+
VEE
VDDLT1640L
SENSE
C1
C3
Q1
R2R3
C2
R4
R5
R6
R1
4
3
2OV
GND
–48V
UV
5 6
8
1
7
GATE
ACTIVE LOWENABLE MODULE
VOUT+
VOUT–
VIN+
1640 F13
VIN–
PWRGD
DRAIN–
+ VEE
Q2ON/OFF
+ –1.8V
VEE
VDD
LT1640L PWRGD
SENSE
C133nF25V
C3100µF100V
Q1IRF530
R210Ω5%
R751k5%
R310k5%
C23.3nF100V
R4562k
1%
R59.09k
1%
R610k1%
R10.02Ω
5%
4
3
2OV
GND
–48V
UV
5 6
8
7
14N25
GATE DRAIN
1640 F14
PWRGD
+
When the DRAIN pin drops belowVPG, Q3 will turn on, shorting thebottom of R7 to VEE and turning Q2off. The pull-up current in the modulethen flows through the R7, pulling thePWRGD pin high and enabling themodule.
When the DRAIN pin of the LT1640Lis more than VPG (1.4V) above VEE, theinternal pull-down transistor, Q2, isoff and the PWRGD pin is in highimpedance state (see Figure 7). ThePWRGD pin will be pulled high by themodule’s internal pull-up currentsource, turning the module off. Whenthe DRAIN pin drops below VPG , Q2will turn on, and the PWRGD pin willbe pulled low, enabling the module.
The PWRGD signal can also beused to turn on an LED or optoisola-tor to indicate that the power is good,as shown in Figure 8.
Figure 7. Active-low enable module
Figure 8. Using PWRGD to drive an optoisolator
Gate Pin Voltage RegulationWhen the supply voltage to the chip ismore than 15.5V, the GATE pin volt-age is regulated at 13.5V above VEE. Ifthe supply voltage is less than 15.5V,
the GATE voltage will be about 2Vbelow the supply voltage. At the mini-mum 10V supply voltage, the gatevoltage is guaranteed to be greaterthan 6V and no greater than 18V forsupply voltages up to 80V.
ConclusionLT1640 provides a simple and flexiblesolution for hot swap applications. Itis the first part that allows systemdesigners to connect an 80V supplydirectly to the chip without any volt-age step-down circuitry. It can beprogrammed to control the outputvoltage slew up rate and the inrushcurrent. It has programmable under-voltage and overvoltage protection, andthe PWRGD/PWRGD output can betied directly to a power-module enablepin. The LT1640 simplifies the designof high voltage hot-swap control sys-tems and combines all of these featuresin an 8-pin SO/PDIP package.
Authors can be contactedat (408) 432-1900
forthe latest information
on LTC products, visit
www.linear-tech.com
Linear Technology Magazine • August 199822
DESIGN FEATURES
Hot Swapping the PCI BusUsing the LTC1643 by Robert ReayHot Circuit InsertionWhen a circuit board is inserted intoa live PCI slot, the supply bypasscapacitors on the board can drawhuge transient currents from the PCIpower bus as they charge. The tran-sient currents can cause permanentdamage to the connector pins andcreate glitches on the power bus, caus-ing other boards in the system toreset.
The LTC1643 is designed to turn aboard’s supply voltages on and off ina controlled manner, allowing theboard to be safely inserted into orremoved from a live PCI slot withoutcausing glitches on the system powersupplies. The chip also protects thePCI supplies from short circuits andmonitors the supply voltages.
The LTC1643H is designed formotherboard applications, whereasthe LTC1643L is designed for Com-pactPCI applications where the chipresides on the plug-in board. The partis available in the space saving 16-pinSSOP package.
LTC1643 Features❏ Allows safe insertion into and
removal from either amotherboard (LTC1643H) orCompactPCI board (LTC1643L)
❏ Controls all four PCI supplies:–12V, 12V, 3.3V and 5.0V
❏ Programmable foldback currentlimit: a programmable analogcurrent limit with a value thatdepends on the output voltage. Ifthe output is shorted to ground,the current limit drops to keeppower dissipation and supplyglitches to a minimum.
❏ Programmable circuit breaker: ifa supply remains in current limittoo long, the circuit breaker willtrip, the supplies will be turnedoff and the FAULT pin will bepulled low.
❏ Current-limit power up: thesupplies are allowed to power upin current limit. This allows thechip to power-up boards withwidely varying capacitive loadswithout tripping the circuitbreaker. The maximum allowablepower-up time is programmableusing the TIMER pin.
❏ 12V and –12V power switcheson-chip
❏ Power-good output: monitors thevoltage status of the four supplyvoltages.
PCI Power RequirementsAll PCI connectors require four powerrails: 5V, 3.3V, 12V and –12V. Sys-tems implementing the 3.3V signalingenvironment are always required toprovide all four rails in every system.Systems implementing the 5V signal-ing environment may either ship the3.3V supply with the system or pro-vide a means to add it afterward. Thetolerance of the supplies as measuredat the plug-in card is summarized inTable 1.
Power-Up SequenceThe power supplies are controlled byexternal N-channel pass transistorsQ1 and Q2 in the 3.3V and 5.0Vpower paths and internal pass tran-sistors for the 12V and –12V powerpaths (see Figure 1).
Resistors R1 and R2 providecurrent-fault detection and R7 andC1 provide current-control loop com-pensation. Resistors R5 and R6prevent high frequency oscillations inQ1 and Q2.
When the ON pin (5) is pulled high,the pass transistors are allowed toturn on and a 20µA current source isconnected to the TIMER pin (4) (seeFigure 2). The current in each passtransistor increases until it reachesthe current limit for each supply.
R20.005Ω
Q2IRF7413
Q1IRF7413
R10.007Ω
R510Ω
R610Ω
R71k
C10.047µF
5V5A
3.3V7.5A
GND
–12V100mA
12V500mA*
R310k
*R4
10k
11 3 13 12 14
16
15
4
109
812
5
6
7
3VINGND12VINVEEIN
ON
FAULT
PWRGD
3VSENSE 5VSENSE3VOUT
LTC1643H
5VOUT
12VOUT
VEEOUT
TIMER
5VINGATE
1643 TA01
SYSTEMPOWERSUPPLY
PCIPOWERSYSTEM
CONTROLLER C20.1µF
RESETLOGIC
RESET*CONNECT TO SYSTEM CONTROLLER LOGIC-LEVEL SUPPLY
liaRrewoP tnerruClacipyT
ecnaticapaC
%5±V5 A5 Fµ0003<V3.0±V3.3 A5.7 Fµ0003<
%5±V21 Am005 Fµ033<%01±V21– Am001 Fµ021<
Table 1. PCI power supply requirements
Figure 1. Typical LTC1643H application
Linear Technology Magazine • August 1998 23
DESIGN FEATURES
Each supply is then allowed to powerup at a rate controlled either by thecurrent limit into the load capaci-tance or dv/dt = 50µA/C1, whicheveris slower. Current limit faults areignored until the voltage on the TIMERpin (4) reaches 1V below the 12VIN pin(1). Once all four supply voltages arewithin tolerance, the PWRGD pin (7)will be pulled low.
Power-Down SequenceWhen the ON pin (5) is pulled low, apower-down sequence begins (seeFigure 3). Internal switches are con-nected to each of the output supplyvoltage pins to discharge the bypasscapacitors to ground. The TIMER pin(4) is immediately pulled low. Thevoltage on the GATE pin (11) is dis-charged by a 200µA current source toprevent the load currents on the 3.3Vand 5.0V supplies from going to zero
instantaneously and causing glitcheson the power supplies. When any ofthe output voltages dips below itsthreshold, the PWRGD pin (7) is pulledhigh.
TimerDuring a power-up sequence, a 20µAcurrent source is connected to theTIMER pin (4) and current limit faultsare ignored until the voltage on thispin increases to within 1V of the12VIN pin (1). This feature allows thechip to power-up PCI cards with widelyvarying capacitive loads on the sup-plies. The power-up time will be:
tON = 2 × (CSUPPLY × VSUPPLY)/(ILIMIT – ILOAD)For CSUPPLY = 2000µF, VSUPPLY = 5V,
ILIMIT = 7A, ILOAD = 5A the turn on timewill be ~10ms. The timer period shouldbe set longer than the maximum sup-ply turn-on time but short enough tonot exceed the maximum safe operat-ing area of the pass transistor duringa short circuit. The TIMER period willbe:
tTIMER = (CTIMER × 11.1V)/22µAFor CTIMER = 0.1µF, the TIMER
period will be ~50ms. The TIMER pin
ON
TIMER
GATE
SUPPLYOUTPUTS
PWRGD
FAULT
dV/dt = 20µA/C2
dV/dt = 50µA/C1 ORSET BY CURRENT LIMIT
ON
TIMER
dV/dt = 200µA/C1GATE
SUPPLYOUTPUTS
PWRGD
FAULT
ON
TIMER
GATE
SUPPLYOUTPUTS
PWRGD
FAULT
1643 F04
Figure 2. Normal power-up sequence Figure 3. Normal power-down sequence
Figure 4. Power-up into a short circuit on a 3.3V output
Linear Technology Magazine • August 199824
DESIGN FEATURES
(4) is immediately pulled low whenthe ON pin (5) goes low.
Thermal ShutdownThe internal switches for the 12V and–12V supplies are protected by aninternal current limit and thermalshutdown circuit. When the tempera-ture of the chip reaches 150°C, allswitches will be latched off and theFAULT pin (6) will be pulled low.
Short-Circuit ProtectionDuring a normal power-up sequence,if the voltage on the TIMER pin (4) hasreached its maximum and a supply isstill in current limit, all of the passtransistors will be immediately turnedoff and the voltage on the FAULT pin(6) will be a logic low, as shown inFigure 4. If a short circuit occurs afterthe supplies are powered-up, theshorted supply’s current will dropimmediately to the limit value (seeFigure 5).
If the supply remains in currentlimit for more than 15µs, all of thesupplies will be latched off. The 15µsdelay prevents quick current spikes,for example from a fan turning on,from causing false trips of the circuitbreaker. The chip will stay in thelatched-off state until the ON pin (5)
ON
TIMER
GATE
SUPPLYOUTPUTS
15µs
PWRGD
FAULT
1643 F05
1
2
R20.005Ω
VID
Q2IRF7413
Q1IRF7413
R10.007Ω
R510Ω
R610Ω R7
1k
C10.047µF
*C40.1µF
*C30.1µF
5V5A5V
3.3V
BD_SEL#
HEALTHY#
12V
–12V
GND
3.3V7.5A
–12V100mA
12V500mA
R310k
R810k
VID
R1110k
11 3 13 12 14
16
15
4
8
109
8
5
6
7
3VIN
12VIN
VEEIN
ON
FAULT
PWR