Wideband Compact Antennas for

Wireless Communication Applications

Minh-Chau Huynh

Dissertation submitted to the Faculty of the

Virginia Polytechnic Institute and State University

in partial fulfillment of the requirements for the degree of

Doctor of Philosophy

in

Electrical and Computer Engineering

Dr. Warren Stutzman, Chair

Dr. William Davis

Dr. Ahmad Safaai-Jazi

Dr. Timothy Pratt

Dr. Beate Schmittmann

November 22, 2004

Blacksburg, Virginia

Keywords: Small antenna, compact antenna, wideband antenna, radiation efficiency,

radiation effects

Copyright 2004, Minh-Chau Huynh

Wideband Compact Antennas for Wireless Communication

Applications

Minh-Chau Huynh

(ABSTRACT)

Recent technologies enable wireless communication devices to become physically

smaller in size. Antenna size is obviously a major factor that limits miniaturization. In the

past few years, new designs of low-profile antennas for handheld wireless devices have

been developed. The major drawback of many low-profile antenna designs is their

narrow impedance bandwidth. Furthermore, the market trend of personal wireless devices

is moving toward a universal system that can be used anywhere and rapid expansion of

the wireless communication industry has created a need for connectivity among various

wireless devices using short-range wireless links in the Bluetooth operating band to get

rid of the cable connections. This requires therefore multiple frequency band operation.

In summary, physically small size, wide bandwidth, and high efficiency are the desired

characteristics of antennas in mobile systems.

This dissertation presents a comprehensive analysis of a new wide-bandwidth

compact antenna, called WC J-pole antenna, covering 50 % impedance fractional

bandwidth. A set of guidelines is also provided for a bandwidth-optimized design at any

frequency. A few design variations of the proposed antenna are also presented for

existing commercial wireless applications.

Efficiency is perhaps the most important characteristic of small antennas for

mobile systems. An extension of the Wheeler cap method to moderate-length and

wideband antennas is developed to measure quickly efficiency.

The dissertation also provides a review of human operator interaction with

handset antennas. Since the proposed antenna is intended to be used in the proximity of

human body and in a casing, coupling effects of human body and casing on the antenna

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characteristics and radio frequency (RF) energy absorption into the human body are

investigated.

iv

Acknowledgements

I would like to thank Dr. Warren Stutzman who has served many years as my advisor and

committee chairman for my Masters and Ph.D. degrees. Dr. Stutzman provided

invaluable advice and support throughout the work on this dissertation and my career

development. I thank him for his encouragement, patience, generous amounts of time and

effort, countless edits of my dissertation, and especially for giving me a chance to be part

of Virginia Tech Antenna Group.

I am indebted to Dr. William Davis for numerous discussions and suggestions on

the technical aspect of the work presented in this dissertation. I would like to thank Dr.

Koichiro Takamizawa, Dr. Nathan Cummings, and Taeyoung Yang for many suggestions

and ideas on the modeling and simulation aspect of my work. I would also like to thank

other members of my advisory committee, Dr. Ahmad Safaai-Jazi, Dr. Timothy Pratt, and

Dr. Beate Schmittmann for their help on completing my Ph.D. work.

I am also thankful to my beloved friend Marion Mangin for her moral support and

for showing me that there is more than just work in life. Without her, my days would

have been just black and white. Many thanks also go to my labmates and friends Kai

Dietze, Gaurav Joshi, and Stani Licul with whom I shared many laughs, jokes, and

sandwiches at Substation.

Finally, I am grateful to my family for their overwhelming support and

encouragement throughout my studies and life. Without my parents and their dedication

on their children’s education, I would have not had any opportunity to be where I am

right now.

v

Contents

1 INTRODUCTION ................................................................................................................................... 1 1.1 INTRODUCTION................................................................................................................................ 1 1.2 OVERVIEW OF THE DISSERTATION .................................................................................................. 4 1.3 REFERENCES ................................................................................................................................... 4

2 LITERATURE REVIEW ....................................................................................................................... 5 2.1 INTRODUCTION................................................................................................................................ 5 2.2 ELECTRICALLY SMALL ANTENNAS ................................................................................................. 5 2.3 FUNDAMENTAL LIMITATIONS ON THE RADIATION Q OF SMALL ANTENNAS ................................... 7

2.3.1 Overview of Theoretical Investigations on the Fundamental Limits ...................................... 7 2.3.2 Fundamental Limitations of Electrically Small Antennas .................................................... 11

2.4 TECHNIQUES FOR REDUCING ANTENNA SIZE ................................................................................ 13 2.4.1 Use of Short Circuits and Ground Planes ............................................................................ 13 2.4.2 Optimizing the Antenna Geometry........................................................................................ 15 2.4.3 Antenna Loading with High-Dielectric Materials ................................................................ 16 2.4.4 Use of Antenna Environment ................................................................................................ 17 2.4.5 Effects on Antenna Performance Characteristics................................................................. 18 2.4.6 Examples of Practical Small Antennas for Hand-Held Wireless Communications .............. 21

2.5 TECHNIQUES FOR WIDENING IMPEDANCE BANDWIDTH FOR MSA’S............................................. 27 2.5.1 Planar Multi-Resonator Configurations............................................................................... 28 2.5.2 Multilayer Configurations .................................................................................................... 35

2.6 SUMMARY ..................................................................................................................................... 40 2.7 REFERENCES ................................................................................................................................. 41

3 ANTENNA EFFICIENCY MEASUREMENTS ................................................................................. 44 3.1 INTRODUCTION.............................................................................................................................. 44 3.2 WHEELER CAP METHOD ON SMALL ANTENNAS............................................................................ 45 3.3 EXTENSION OF WHEELER CAP METHOD TO WIDEBAND ANTENNAS ............................................. 49 3.4 EXPERIMENTAL RESULTS OF EFFICIENCY EVALUATION USING WIDEBAND WHEELER CAP.......... 53

3.4.1 Lossy Monopole on Finite Ground Plane............................................................................. 54 3.4.2 Planar Half-Disk UWB Monopole Antenna ......................................................................... 57 3.4.3 Measurement Sensitivity Tests .............................................................................................. 59

3.5 SUMMARY ..................................................................................................................................... 61 3.6 REFERENCES ................................................................................................................................. 61

4 A REVIEW OF RADIATION EFFECTS ON HUMAN OPERATORS OF HAND-HELD RADIOS ...................................................................................................................................................... 63

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4.1 BIOLOGICAL EFFECTS OF RADIO-FREQUENCY RADIATION ........................................................... 63 4.2 RF RADIATION BASICS ................................................................................................................. 64 4.3 BIOLOGICAL EFFECTS AND HEALTH ISSUES .................................................................................. 65 4.4 RF SAFETY AND REGULATIONS .................................................................................................... 67 4.5 HUMAN OPERATOR INFLUENCE ON HAND-HELD RADIO PERFORMANCE ...................................... 69

4.5.1 Computational Simulation .................................................................................................... 70 4.5.2 Electromagnetic Modeling of the Human Operator ............................................................. 71

4.6 HUMAN OPERATOR EFFECTS ON ANTENNA CHARACTERISTICS .................................................... 74 4.7 POWER ABSORPTION IN THE HEAD AND SAR................................................................................ 78 4.8 SUMMARY ..................................................................................................................................... 80 4.9 REFERENCES ................................................................................................................................. 81

5 A NEW WIDEBAND COMPACT ANTENNA .................................................................................. 85 5.1 INTRODUCTION.............................................................................................................................. 85 5.2 THE ANTENNA STRUCTURE........................................................................................................... 86 5.3 RESULTS OF PRELIMINARY NUMERICAL SIMULATION AND EXPERIMENTAL INVESTIGATIONS ...... 89 5.4 ANTENNA STRUCTURE SIMILARITIES OF WC J-POLE TO OTHER ANTENNAS ................................ 90 5.5 ANALYSIS OF THE COMPACT PLANER WIDEBAND J-POLE ANTENNA ............................................ 97

5.5.1 Parametric Analysis of the Compact WC J-pole Antenna .................................................... 97 5.5.1.1 Variation of Feed Probe Height hP ................................................................................................. 100 5.5.1.2 Variation of Feed Plate Area .......................................................................................................... 102 5.5.1.3 Variation of Feed Probe Position, dp .............................................................................................. 109 5.5.1.4 Variation of the Short Plate Height, hs ........................................................................................... 116 5.5.1.5 Variation of the Lower Plate Length, LA........................................................................................ 119 5.5.1.6 Variation of the Top Plate Length, LB............................................................................................ 122

5.5.2 Summary of the WC J-pole Parameter Variation ............................................................... 125 5.5.3 Design Case for the Largest Impedance Bandwidth........................................................... 128

5.6 SUMMARY ................................................................................................................................... 131 5.7 REFERENCES ............................................................................................................................... 131

6 VARIATIONS OF THE WC J-POLE FOR A FEW COMMERCIAL APPLICATIONS........... 133 6.1 INTRODUCTION............................................................................................................................ 133 6.2 WIDEBAND COMPACT ANTENNA FOR COVERING APPLICATIONS FROM GPS TO BLUETOOTH BANDS (WCJP #1) .................................................................................................................................. 133 6.3 DUAL-BAND COMPACT ANTENNA FOR PERSONAL WIRELESS COMMUNICATIONS, BLUETOOTH, AND U-NII BANDS (DCLA #1)................................................................................................................ 137 6.4 DUAL-BAND COMPACT ANTENNA FOR 2.45/5.25 GHZ WLAN (DCLA #2)........................... 141 6.5 SUMMARY ................................................................................................................................... 146

7 CONCLUSIONS.................................................................................................................................. 147 7.1 SUMMARY ................................................................................................................................... 147 7.2 CONTRIBUTIONS.......................................................................................................................... 148 7.3 FUTURE WORK............................................................................................................................ 149 7.4 REFERENCE ................................................................................................................................. 150

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List of Figures

Figure 2-1 Fundamental limit curves of radiation Q versus ka. The top curve (solid curve) is a plot of

(2.3) and is for efficiency, 1=re , or %100 . The dashed curve is the plot of (2.2) with

efficiency 1=re . .................................................................................................................. 10 Figure 2-2 Comparison of several practical antenna (Q,ka) values (see Table 2-1) to the fundamental

limits curve based on (2.3) with er = 100%............................................................................. 12 Figure 2-3 An example of size reduction using a ground plane................................................................ 14 Figure 2-4 Size reduction using short circuit in microstrip patch antenna................................................ 14 Figure 2-5 The effect of notches and slots in a microstrip-patch antenna for size reduction: (a) regular

microstrip patch antenna and (b) microstrip patch antennas with notches and slots. .............. 15 Figure 2-6 Meander-line printed antenna: an example of reducing antenna size by increasing its electrical

length. ..................................................................................................................................... 16 Figure 2-7 Size reduction of a monopole by dielectric loading. ............................................................... 17 Figure 2-8 The smart monobloc integrated-L antenna (SMILA) [17]. ..................................................... 18 Figure 2-9 Effects of permittivity value on the electric field intensity of a rectangular microstrip-patch

antenna with (a) εr=1.07 and (b) εr=10.2 at resonance simulated using FDTD method. The colors representing the normalized electric field go from blue (weak E-field) to green to yellow to red (strong E-field).................................................................................................. 20

Figure 2-10 Geometry of the dual-frequency PIFA with a branch-line slit; dimensions in the figure are in millimeters [20]....................................................................................................................... 22

Figure 2-11 Measured and simulated return loss of the PIFA shown in Fig. 2-10 [20]. ............................. 23 Figure 2-12 Simulated patch surface current at (a) 950 MHz and (b) 1790 MHz for the PIFA shown in Fig.

2-10 [20].................................................................................................................................. 23 Figure 2-13 (a) The branch line planar monopole in a wrapped structure for GSM/DCS/PCS multi-band

mobile phone antenna; (b) the monopole unwrapped into a planar structure [21]. ................. 25 Figure 2-14 Measured and simulated return loss for the monopole in the wrapped structure shown in Fig.

2-13a [21]................................................................................................................................ 26 Figure 2-15 Geometry of the dual-band printed inverted-F antenna [22]................................................... 27 Figure 2-16 Measured input impedance for the antenna shown in Fig. 2-15 with h1 = 10 mm, h2 = 5 mm, l1

= 21 mm, l2 = 10 mm, d = 3 mm, and w = wf =3.05 [22]......................................................... 27 Figure 2-17 Various gap-coupled multiresonator rectangular MSAs configurations: gap-coupling (a) along

the radiating edges of the fed patch, (b) along the non-radiating edges of the fed patch, and (c) along all edges of the fed patch [28]. ...................................................................................... 29

Figure 2-18 VSWR plots of two coupled resonators having (a) narrow and (b) wide bandwidth: (….) individual resonators and () overall response...................................................................... 30

Figure 2-19 A single parasitic patch element gap-coupled with one fed patch antenna: (a) geometry, (b) computed input impedance, and (c) VSWR plots of a single fed patch with no parasitic element (---) and with one parasitic patch element () [28]. ................................................. 31

Figure 2-20 Computed input impedance (a) and VSWR plots (b) of two gap-coupled patches along radiating edge for two feed point locations x of the parasitic MSA of Fig. 2-19a: (---) 0.7 cm and () 1.1 cm [28]................................................................................................................ 32

viii

Figure 2-21 Computed input impedance (a) and VSWR plots (b) of two gap-coupled patches along radiating edge for three values of L1 of the parasitic MSA of Fig. 2-19a: (---) 2.8 cm, () 2.9 cm, and (— – —) 3.0 cm [28]. ................................................................................................ 32

Figure 2-22 Computed input impedance (a) and VSWR plots (b) of two gap-coupled patches along radiating edge for three values of s for the parasitic patch antenna of Fig. 2-19a: (---) 0.05 cm, (— – —) 0.1 cm, and () 0.15 cm [28]. ................................................................................ 33

Figure 2-23 Computed radiation patterns of two gap-coupled patches along radiating edge at frequencies (a) 2.9 GHz, (b) 3.0 GHz, and (c) 3.1 GHz: () E-plane, (---) H-plane [28]. ........................ 34

Figure 2-24 Computed radiation patterns of three gap-coupled patches along radiating edges at frequencies of Fig. 2-17a: (a) 2.89 GHz and (b) 3.09 GHz: () E-plane, (---) H-plane [28]. ................... 35

Figure 2-25 Configurations of (a) Electromagnetically-coupled and (b) aperture-coupled MSAs............. 36 Figure 2-26. Geometry of a stacked microstrip patch coaxial-fed on a large ground plane: (a) top and (b)

side views................................................................................................................................ 37 Figure 2-27 Impedance and VSWR over the frequency range from 3.3 to 4.1 GHz for the stacked

microstrip patch of Fig. 2-26 simulated to examine upper patch size variation for four values of L2: (— – —) 2.0 cm, (---) 2.3 cm, (——) 2.45 cm, and (…) 2.6 cm [28]........................... 38

Figure 2-28 Impedance and VSWR over the frequency range from 3.3 to 3.9 GHz for the stacked microstrip patch of Fig. 2-26 simulated to examine lower patch size variation for three different values of L1: (— – —) 2.50 cm, (——) 2.55 cm, and (---) 2.60 cm [28]. ................ 39

Figure 2-29 Impedance and VSWR over the frequency range from 3.4 to 4.0 GHz for the stacked microstrip patch of Fig. 2-26 simulated to examine the effect of misalignment of the top patch from the lower patch for different offset values: (——) not offset, (…) oy=0.1 cm, (— – —) ox=0.1 cm, and (---) ox=-0.1 cm [27]...................................................................................... 40

Figure 3-1 Circuit model of an antenna resistance showing radiation and loss resistances...................... 46 Figure 3-2 Illustration of the method for determining radiation efficiency of wideband antennas. Instead

of inhibiting radiation, a Wideband Wheeler Cap allows the antenna to transmit and receive the reflected signal [9]: (a) Antenna radiating in free space and (b) A “Wideband Wheeler Cap,” sized to be larger than the radianshpere dimension of the traditional Wheeler cap. ..... 50

Figure 3-3 Power budget for a TX-RX pair of the wideband Wheeler cap method if all the transmitted power is available at the receive antenna. ............................................................................... 51

Figure 3-4 Power fraction link budget for an antenna inside the wideband Wheeler cap......................... 52 Figure 3-5 Antenna radiation efficiency measurement setup using a 15-cm radius wideband Wheeler cap

probed by an HP 8720 network analyzer. ............................................................................... 53 Figure 3-6 Lossy monopole on a finite ground plane (100 by 140 mm) with a 50-Ohm resistor at the tip

of the antenna to insert some dissipation loss in the antenna. ................................................. 54 Figure 3-7 Magnitude of S11 measured using the setup in Fig. 3-5 of the lossy monopole of Fig. 3-6: (a)

in free space, and (b) in the spherical cap of 15-cm radius. .................................................... 55 Figure 3-8 Processed |S11| (blue curve) of the monopole in the spherical cap by taking the largest value

for the raw data over 100 MHz span, assuming that the impedance characteristics of the antenna does not change drastically within that span.............................................................. 56

Figure 3-9 Total efficiency ηT (radiation efficiency including impedance mismatch) of the lossy monopole of Fig. 3-7 evaluated using (3.13) with experimental data (solid curve) and numerical data (asterisk). ........................................................................................................ 56

Figure 3-10 A planar half-disk UWB monopole antenna designed for operating from 3.1 to 10.6 GHz [10]. 57

Figure 3-11 Magnitude of S11 for the planar half-disk UWB monopole antenna of Fig. 3-10 measured: (a) in free space and (b) inside the spherical cap of 15-cm radius................................................ 58

Figure 3-12 Total efficiency ηT (radiation efficiency including impedance mismatch) of the planar half-disk UWB monopole antenna of Fig. 3-10 evaluated using (3.13). ........................................ 59

Figure 3-13 Measurement setup for the planar half-disk UWB monopole antenna placed away from the center of the spherical Wheeler cap. ....................................................................................... 60

Figure 3-14 Comparison of the antenna total efficiency for test antenna locations near and away (Fig. 3-13) from the center of the spherical Wheeler cap. .................................................................. 60

Figure 4-1 The electromagnetic spectrum. ............................................................................................... 65

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Figure 4-2 ANSI/IEEE and ICNIRP/CELENEC maximum permissible exposure in terms of power density for uncontrolled environment in RF frequency range [5]. .......................................... 69

Figure 4-3 Field component value locations of a Yee cell used in FDTD computations. The E-components are in the middle of the edges and the H-components are in the center of the faces [10].......................................................................................................................................... 70

Figure 4-4 Popular human head models used in numerical simulations................................................... 73 Figure 4-5 Computed radiation patterns in φ-plane (θ = 90°) for a monopole oriented along the z-axis and

mounted on a metal box. Operation is at 900 MHz and the antenna is 1.5 cm away from a human head modeled as a rectangular box of 20-cm side length and as a sphere of 20-cm diameter with three layers of dielectric material (skin-skull-brain); see Figs 4a and b........... 75

Figure 4-6 Computed return loss values of a monopole on a handset for three cases: no human operator (“monopole only”); only a hand present (“with hand”); and both an operator hand and head present (“with hand and head”). The separation distance between 20-cm diameter spherical head model (skin-skull-brain) and the antenna is 1.5 cm and the hand model (muscle-bone) is located 6.0 cm below the antenna. .......................................................................................... 77

Figure 4-7 Handset model showing the PIFA antenna location and hand model for the results in Fig. 4-8. 77

Figure 4-8 Computed return loss values of the side-mounted PIFA on a handset shown in Fig. 7 without the hand and with the hand for three different hand locations d. ............................................ 78

Figure 5-1 Geometry of the compact wideband J-pole (WC J-pole) antenna designed for operation from fL = 1.77 GHz to fU = 2.45 GHz with center frequency fc = 2.11 GHz. ...................................... 88

Figure 5-2 Geometry of a conventional planar inverted-F antenna (PIFA)................................................. 88 Figure 5-3 Impedance characteristics as function of frequency for the compact WC J-pole of Fig. 5-1

computed using the IE3D simulation code: (a) Smith chart of complex impedance from 1.5 to 3.5 GHz; (b) VSWR referenced to 50 Ohms. The measured VSWR (dashed curve) is also plotted in (b)............................................................................................................................ 89

Figure 5-4 Far-field radiation patterns of the WC J-pole antenna of Fig. 5-1 at 2.2 GHz computed using IE3D for: (a) xz-plane, (b) yz-plane, and (c) xy-plane............................................................ 91

Figure 5-5 Average current distribution of (a) the WC J-pole and (b) a conventional PIFA on a large finite ground plane. The magnitude of the current distribution goes from minimum (blue color) to maximum (red color). The results were obtained from simulation using IE3D. ...... 92

Figure 5-6 Structure of a J-pole antenna with dimensions for operation at 1.8 GHz with a wire radius of 0.635 mm. ............................................................................................................................... 93

Figure 5-7 Return loss of J-pole antenna of Fig. 5-6 matched to 50-Ohm impedance computed using IE3D................................................................................................................................................. 93

Figure 5-8 Planar J-Pole antenna compared to a wire J-pole antenna. ........................................................ 94 Figure 5-9 Impedance properties of the planar J-pole antenna of Fig. 5-8 for various antenna widths WS

computed using IE3D: (a) resistance, (b) reactance, and (c) magnitude of S11. ...................... 96 Figure 5-10 Illustration of the increase in current path length accomplished by narrowing the short plate

(WS) width of the planar J-pole antenna of Fig. 5-8. ............................................................... 96 Figure 5-11 WC J-pole antenna structure with its dimension parameters. .................................................. 99 Figure 5-12 Input impedance (a) and VSWR relative to 50 Ohms (b) of the WC J-pole antenna of Fig. 5-

11 and Table 5-2 computed using IE3D................................................................................ 100 Figure 5-13 Impedance of the WC J-pole antenna shown in Fig. 5-11 and Table 5-2 for various feed probe

height hp values: (a) real part of impedance (antenna input resistance), (b) imaginary part of impedance (antenna input reactance), and (c) |S11| of the structure matched to 50-Ohm impedance, The height of the top plate is 5.6 mm. ............................................................... 102

Figure 5-14 Feed plate area variation of the capacitive antenna structure. The square shape feed plate side length LF increases from 2 to 22 mm. ................................................................................... 104

Figure 5-15 Impedance of the antenna structure with square feed plate LF×LF area variation from LF×LF = 2x2 mm2 (case 1) to LF×LF = 10x10 mm2 to LF×LF = 22x22 mm2 (case 3), according to Fig. 5-14. 105

Figure 5-16 Geometry of the WC J-pole antenna for the study of the feed plate area not shielded by the top plate....................................................................................................................................... 107

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Figure 5-17 Input impedance for the WC J-pole antenna of Fig. 5-16 and Table 5-4 calculated using IE3D: (a) resistance, (b) reactance, and (c) return loss referenced to 50 Ohms............................... 108

Figure 5-18 Geometry of the WC J-pole antenna. dp is the probe feed position away from the short plate. .............................................................................................................................................. 110

Figure 5-19 Input impedance of WC J-pole antenna shown in Fig. 5-18 and Table 5-5 calculated for values of feed probe position dp from 4 mm to 20 mm away from the antenna short plate with all other dimensions fixed. .................................................................................................... 111

Figure 5-20 Geometry of the WC J-pole antenna. dp is the probe feed position away from the short plate and the feed plate is moved along with the probe feed. ........................................................ 113

Figure 5-21 Impedance of the antenna structure shown in Fig. 5-20 and Table 5-6 computed using IE3D for the case when the feed plate is completely shielded by the top plate. ............................. 114

Figure 5-22 Impedance of the antenna structure shown in Fig. 5-20 and Table 5-6 computed using IE3D for the case when a portion of the feed plate is unshielded by the top plate. ........................ 116

Figure 5-23 Geometry of the WC J-pole antenna for the height hs study.................................................. 117 Figure 5-24 Impedance of the antenna structure shown in Fig. 5-23 and Table 5-7 computed for various

short plate heights hs. ............................................................................................................ 119 Figure 5-25 Geometry of the WC J-pole antenna for the analysis on the variation of the lower plate length

LA. 120 Figure 5-26 Impedance of the antenna structure shown in Fig. 5-25 and Table 5-8 for lower plate length

variation LA study.................................................................................................................. 122 Figure 5-27 Geometry of the WC J-pole antenna for the analysis on the variation of the top plate length

LB........................................................................................................................................... 123 Figure 5-28 Impedance of the antenna structure shown in Fig. 5-27 and Table 5-9 for top plate length

variation LB study.................................................................................................................. 125 Figure 5-29 Dimensions of the WC J-pole antenna for the optimal impedance bandwidth of about 50 %.

128 Figure 5-30 Input Impedance (a) and return loss for a 50-Ohm input impedance match (b) of the WC J-

pole antenna of Fig. 5-29 with its impedance bandwidth optimized..................................... 129 Figure 6-1 Wideband compact J-pole antenna (WCJP #1) designed to cover frequency bands from GPS

to Bluetooth bands. ............................................................................................................... 135 Figure 6-2 VSWR values computed using IE3D for the WC J-Pole of Fig. 6-1 relative to 50-Ohms. Note

an impedance match (VSWR≤2) is achieved for the frequency bands of interest. ............... 136 Figure 6-3 Computed values of maximum gain over the WCJ-Pole operating band.............................. 136 Figure 6-4 Geometry of the dual-band compact antenna and its prototype (DCLA #1)......................... 138 Figure 6-5 Computed and measured |S11| of the dual-band compact antenna of Fig. 6-4. A 10-dB return

loss (-10 dB S11) corresponds to a 2:1 VSWR. ..................................................................... 138 Figure 6-6 Computed (solid curves) and measured (dashed curves) radiation patterns of the dual-band

compact antenna of Fig. 6-4 in the frequency bands of interest in the yz plane for both Eθ (red curves) and Eφ (blue curves) cuts: (a) 2.2 GHz and (b) 5.2 GHz. ......................................... 139

Figure 6-7 Computed and measured gain of the dual-band compact antenna of Fig. 6-4. ..................... 140 Figure 6-8 Measured radiation efficiency of the dual-band compact antenna of Fig. 6-4 using the

wideband Wheeler cap method described in Chapter 3. ....................................................... 140 Figure 6-9 Overall dimensions of the DCLA #2a structure for WLAN used in IE3D simulation.......... 142 Figure 6-10 Computed return loss of the DCLA #2 for WLAN shown in Fig. 6-9.................................. 142 Figure 6-11 Comparison of the measured |S11| values of the DCLA #2a built with a coax cable and the

numerical results of DCLA #2a without the coax simulated using IE3D. ............................ 143 Figure 6-12 Dimensions of the DCLA structure for WLAN (DCLA #2b) including the coax cable and

tuned to compensate the detuning coupling effects due to the coax. .................................... 144 Figure 6-13 Numerical and measured return loss of the dual-band antenna shown in Figure 6-12 with the

coax cable attached. .............................................................................................................. 145 Figure 6-14 Example of antenna placement for the DCLA #2 in laptop computers that are WiFi enabled.

146

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List of Tables

Table 1-1 Frequency Bands for a Few Popular Wireless Applications. ................................................ 3 Table 2-1 Characteristics of Antennas Used to Examine Bandwidth-Size Relationships..................... 12 Table 4-1 Relative Permittivity, Conductivity, and Mass Density of the Tissues in the Hand and Head

0near 900 MHz [11]. ........................................................................................................... 71 Table 4-2 Comparison of Computed Antenna Efficiency and Power Absorbed in the Head for Simple

(Rectangular and Spherical Shape) and Complex Models Developed from Yale University and from Gent University (cf. Figure 4). Transmit Power is 1 W at 915 MHz; Distance of Monopole Antenna on a Handset from Head is 1.5 cm [15]. .............................................. 72

Table 4-3 Effects of the Change in the Separation Between the Antenna and the Head Model from University of Gent, Belgium; 915 MHz, 1W [15]. .............................................................. 74

Table 4-4 Popular Standards Limits on Exposure to the General Public of Electromagnetic Radiation in the RF. ............................................................................................................................. 80

Table 5-1 Geometric parameters of the WC J-pole antenna of Fig. 5-1. ............................................... 87 Table 5-2 Geometric parameters of the preliminary WC J-pole antenna of Fig. 5-11. ......................... 98 Table 5-3 Geometric parameters of the WC J-pole antenna of Fig. 5-14 for the feed plate area study.

........................................................................................................................................... 103 Table 5-4 Geometric parameters of the WC J-pole antenna of Fig. 5-16 for the feed plate area study.

........................................................................................................................................... 106 Table 5-5 Geometric parameters of the WC J-pole antenna of Fig. 5-18 for feed probe position study.

........................................................................................................................................... 109 Table 5-6 Geometric parameters of the WC J-pole antenna of Fig. 5-20 for feed assembly position

study. ................................................................................................................................. 112 Table 5-7 Geometric parameters of the WC J-pole antenna of Fig. 5-23 for antenna height hs study.

........................................................................................................................................... 117 Table 5-8 Geometric parameters of the WC J-pole antenna of Fig. 5-25 for antenna length LA study.

........................................................................................................................................... 120 Table 5-9 Geometric parameters of the WC J-pole antenna of Fig. 5-27 for antenna length LB study.

........................................................................................................................................... 123 Table 5-10 Summary of the WC J-pole Parameter Variation Analyzed in the Previous Section. The

Geometric Parameters are shown in Fig. 5-11................................................................... 127 Table 5-11 Electrical dimensions of the optimum bandwidth (50%) WC J-pole in terms of the

wavelength λL of the lower frequency fL of the operating band. See Fig. 5-11 for geometry............................................................................................................................................ 130

Table 6-1 Frequency Bands for a Few Wireless Applications. ........................................................... 135 Table 7-1 List of the three versions of the WC J-pole antenna with their characteristics ................... 149

1

Chapter 1

Introduction 1.1 Introduction

Wireless is a term used to describe telecommunications in which electromagnetic waves

carry the signal over part or the entire communication path [1]. Common examples of

wireless equipment in use today include: cell phones, pagers, global positioning systems

(GPS), cordless computer peripherals such as wireless keyboards, cordless telephone sets,

remote garage-door openers, two-way radios, satellite televisions, wireless local area

network (WLAN) and wireless personal area network (WPAN). Wireless technology is

rapidly evolving, and is playing an increasing role in the lives of people throughout the

world. In addition, ever larger numbers of people are relying on wireless technology,

either directly or indirectly. More recent examples of wireless communications include

the following technologies:

- Global System for Mobile Communication (GSM): a digital mobile telephone

system used in Europe and other parts of the world;

- General Packet Radio Service (GPRS): a packet-based wireless communication

service that provides continuous connection to the internet for mobile phone and

computer users;

- Enhanced Data GSM Environment (EDGE): a faster version of the Global

System for Mobile (GSM) wireless service designed to deliver data at rates up

2

to 384 Kbps and enable the delivery of multimedia and other broadband

applications to mobile phone and computer users;

- Universal Mobile Telecommunications System (UMTS): a broadband packet-

based system offering a integrated set of services to mobile computer and phone

users no matter where they are located in the world;

- Wireless Application Protocol (WAP): a set of communication protocols to

standardize the way that wireless devices, such as cellular telephones and radio

transceivers, can be used for internet access.

Wireless can be divided into three categories: fixed, mobile, and portable. Fixed

wireless refers to the operation of wireless devices or systems in fixed locations such as

homes and offices. Mobile wireless applications refer to devices or systems aboard

moving vehicles. Examples include the automotive cell phone and onboard GPS system.

Portable wireless applies to the operation of autonomous, battery-powered wireless

devices or systems outside the office, home, or vehicle; examples include handheld cell

phones and personal communication system (PCS) units.

Recent technologies enable wireless communication devices to become physically

smaller in size. Antenna size is obviously a major factor that limits miniaturization.

Antenna physical size is inversely proportional to its operating frequency. However,

reducing the antenna physical size also means reducing its electrical size since the

operating frequency of these devices does not change. Electrical size is expressed as a

fraction of wavelength, λ. As an example of electrical size, a half-wave dipole antenna is

a half-wave antenna operating at 1800 MHz (λ = c/f = 16.6 cm) is 8.3 cm long because its

electrical size is 0.5 λ. If a wireless device is required to have a physically small antenna,

say half the size or 4.15 cm, and still operate at 1800 MHz, then it requires an antenna

with a physical size of 4.15 cm, corresponding to an electrical size of 0.25 λ. Many

applications at around 1800 MHz require antennas in the order of 0.25 λ or less.

Examples of antennas of quarter-wavelength electrical size that are used include

monopole antennas, helical antennas, and PIFAs (planar inverted-F antenna).

In the past few years, new designs of low-profile antennas for handheld wireless

devices have been developed. The major drawback of many low-profile antenna designs

3

is their narrow impedance bandwidth. Some designs can barely cover the bandwidth

requirement and hence, may not be used because there is no margin in the bandwidth for

potential detuning effects due, for example, to the presence of a human operator.

Furthermore, the market trend of personal wireless devices is moving toward a universal

system that can be used anywhere. Rapid expansion of the wireless communication

industry has created a need for connectivity among various wireless devices using short-

range wireless links in the Bluetooth operating band to get rid of the cable connections.

This requires therefore multiple frequency band operation. A list of a few useful wireless

applications and their operating frequencies is shown Table 1-1. Dual-band and tri-band

compact antennas have been realized to help the transition of new wireless system

generations go smoothly but the current market demand needs wireless systems to

operate in more than three bands. In summary, physically small size, wide bandwidth,

and high efficiency are the desired characteristics of antennas in mobile systems.

Table 1-1 Frequency Bands for a Few Popular Wireless Applications.

Wireless Applications Frequency Band (MHz) Bandwidth, MHz (%)

GPS 1570.42-1580.42 10 (0.7%)

DCS-1800 1710-1880 170 (10.6%)

PCS-1900 1850-1990 140 (7.3%)

IMT-2000/UMTS 1885-2200 315 (15.5%)

ISM (including WLAN) 2400-2483 83 (3.4%)

Bluetooth 2400-2500 100 (4.1%)

U-NII 5150-5350 / 5725-5825 200 (3.8%) / 100 (1.7%)

(12.3 % for both)

This dissertation presents a comprehensive investigation of a new wide

bandwidth, low-profile, compact antenna with a primary focus on the analysis of how the

antenna operates, as detailed in the next section.

4

1.2 Overview of the Dissertation

This dissertation is organized into three parts. The first two chapters review the previous

work and provide necessary background information. A review of small antennas,

techniques for reducing antenna size, and methods for widening antenna impedance

bandwidth is included in Chapter 2, providing useful information for the analysis of the

new antenna. Chapter 2 also includes a section on the fundamental limitations on

bandwidth and size of small antennas.

The second part of the dissertation, contained in Chapters 3 and 4, discusses small

antenna efficiency measurements. Efficiency is perhaps the most important characteristic

of small antennas for mobile systems. The Wheeler Cap method for measuring small

antenna efficiency is described in Chapter 3 and extension of this method to moderate-

length and wideband antennas. Chapter 4 provides a discussion on human operator

interaction. Since the proposed antenna is intended to be used in the proximity of human

body and in a casing, coupling effects of human body and casing on the antenna

characteristics and radio frequency (RF) energy absorption into the human body should

be investigated.

Chapters 5 and 6 present the main investigation of this dissertation. Analysis the

new wide-bandwidth, low-profile, compact antenna is discussed in Chapter 5. Simulation

and experimental results for the antenna are performed to support the analysis. Chapter 6

presents a few design variations of the proposed antenna for various wireless commercial

applications. Conclusions and future investigations are then presented in Chapter 7.

1.3 References

[1] IT-Specific Encyclopedia. Available at http://whatis.techtarget.com.

5

Chapter 2

Literature Review

2.1 Introduction

This chapter presents a literature review on electrically small antennas, antennas that are

a fraction of a wavelength in maximum extend. With the current trend in technology and

consumer electronics that demand smaller wireless devices, small antennas are a hot

research topic. However, there is a fundamental limitation on how much antenna size can

be reduced. In this chapter, a review of the research on fundamental limits of small

antennas is also introduced, explaining how the physical size of antennas is related to

impedance bandwidth and radiation efficiency. Several techniques for reducing antenna

size are also presented, including the use of a short-circuit, geometry optimization,

material loading, and the environment surrounding the antenna. These techniques can

reduce antenna size but with a penalty to pay, which is often the reduction of impedance

bandwidth. Therefore, a few methods for enhancing impedance bandwidth of small

antennas are reviewed.

2.2 Electrically Small Antennas

There are many reasons for using electrically small antennas, including: esthetics, safety,

cost, aerodynamics, and physical size constraints due to system considerations. There is

6

no consensus on the definition of ‘electrically small’. An electrically small antenna is one

whose size is a small fraction of a wavelength. A maximum dimension of λ/30 is often

adopted for electrically small antennas [1]. This criterion was probably first chosen

because it permits the use of two approximations: constant current for small loop and a

linear current distribution for short monopole or dipole antennas. However, according to

H.A. Wheeler [2], an electrically small antenna is one that is smaller than the

radiansphere, which is a spherical volume having a radius of λ/2π, or 0.16 λ. The surface

of the radiansphere forms the boundary between the near field and the far field. Interior to

the sphere (the near filed), the stored energy dominates over the radiating energy. In

current practice, it is popular to use a maximum extent criterion of λ/8 for electrically

small antennas.

The analysis of electrically small antennas was initiated in the mid-1940s by H.A

Wheeler [2]. Wheeler used the radiation power factor, PFRAD, to quantify the radiation of

an antenna. PFRAD is defined as the ratio of the radiated power to the stored power. This

corresponds to the ratio of the antenna resistance R to the antenna reactance X and is

proportional to the volume occupied by an antenna. Using a simple lumped circuit, he

deduced that this ratio is equivalent to the bandwidth multiplied by the efficiency, in

cases where the antenna is matched to the tuned circuit. This early paper was the first

attempt to confirm mathematically the intuition we have that the product of (efficiency ×

bandwidth) is directly related to PFRAD, hence, to the volume occupied by the antenna,

since this product is equal to the ratio of the antenna resistance to the antenna reactance.

According to Wheeler’s formulation, electrically small antennas can be characterized by

their low radiation resistance and large reactance (small PFRAD), and thus, small

impedance bandwidth and low efficiency.

The worldwide growth of personal wireless communication devices has been

tremendous. One of the trends in wireless mobile technology in the last decade has been

to dramatically decrease the size and the weight of the handset. With this progress in

mobile terminal size reduction, the design of antennas is acquiring even greater

importance. Antennas must be small, and yet achieve specified electrical performance,

such as wide bandwidth, operation in dual or triple frequency bands, diversity, and so

forth. Accordingly, antenna designers have encountered difficulty in designing antennas

7

that can maintain electrical performance characteristics while being reduced in size

because, in general, efficiency and bandwidth degrade with size decrease. The next

sections discuss how much an antenna can be reduced in size.

2.3 Fundamental Limitations on the Radiation Q of Small Antennas

The radiative properties of electrically small antennas were first investigated by Wheeler

[2]. Later, Chu [3] derived an expression for the minimum radiation quality factor Q of

an antenna enclosed inside a sphere of a given radius. The quality factor Q is defined as

the radian frequency ω times the ratio of the reactive energy stored about the antenna to

the radiated power [9]. It is a measure of the reactive energy that necessarily accompanies

real output power. In 1960, Harrington [4] extended Chu's theory to include circularly

polarized antennas. Later, Collin [5] and Fante [6] derived exact expressions for the

radiation Q based on the evanescent energy stored around an antenna. McLean [7]

corrected an error in Chu's approximate expression and re-derived the exact expression

using non-propagating energy. Most recently, Caswell and Davis [8], and Grimes and

Grimes [9] re-derived the fundamental limit on radiation Q using a time-domain

approach. Their work is summarized in the following section to show the concept of the

fundamental limit applied to practical small antennas.

2.3.1 Overview of Theoretical Investigations on the Fundamental Limits

There have been numerous theoretical investigations of antenna size and performance

over the past five decades. Reducing antenna volume generally degrades antenna

performance. It is, therefore, important to examine the fundamental limits and parameter

tradeoffs involved in size reduction. Certainly at some point, electrical performance

specifications will not be satisfied if the allocated volume for the antenna region is

reduced too much. In particular, it is well known that size reduction is obtained at the

8

expense of bandwidth and efficiency. However, this size-performance trade-off must be

quantified in order to have useful guidelines in the search for more compact antennas.

The concept of Chu's work [3] was to place an antenna inside a sphere of radius a

that it just encloses the antenna and then to represent the fields outside the sphere as a

weighted sum of spherical wave modes. Then, radiation Q can be computed using [3]

>

>=

emrad

m

merad

e

WWP

W

WWP

W

Q ω

ω

2

2

(2.1)

where We and Wm are the time-average, non-propagating, stored electric and magnetic

energy, respectively, ω denotes radian frequency, and Prad denotes radiated power. Chu

assumed that all of the stored energy is outside the sphere enclosing the antenna. This

concept leads to the minimum possible radiation Q since any stored energy inside the

sphere would increase the radiation Q of the antenna. However, the calculation of this

radiation Q is not straightforward because the total time-average stored energy outside

the sphere is infinite, just as it is for any propagating wave or combination of propagating

waves and non-propagating fields [3]. Chu addressed this by deriving an RLC equivalent

ladder network for each spherical waveguide mode and then radiation Q is computed

from the stored energy in the inductors and capacitors of the equivalent circuit network.

However, this is still a tedious computation if many modes exist. So, Chu approximated

the system as an equivalent second-order series RLC network and solved the problem

assuming that the antenna only excites the n = 1 mode. The resultant radiation Q

expression is

( )( ) ( )[ ]23

2

121

kakakaQ

++= (2.2)

where the algebraic error cited by McLean [7] has been accounted for, k is the wave

number associated with the electromagnetic field and a if the radius of the sphere

enclosing the antenna. Chu also showed that an antenna which excites only the n = 1

mode has the lowest possible radiation Q of any linearly polarized antenna.

9

MacLean [7] re-examined this fundamental limit in order to achieve the highest

possible accuracy. McLean presented an exact expression for the minimum radiation Q.

Like Chu, he assumed that the antenna radiates only one mode, in this case the n = 1

spherical mode. The fields of this mode are equivalent to the fields radiated by a short

dipole antenna. He computed the stored energy due to the total fields and then subtracted

the stored energy due to the radiated fields, leaving only the non-propagating stored

energy from which the radiation Q is determined. The resulting expression for Q is

( ) kakaQ 11

3 += (2.3)

This relationship between the radiation Q and the antenna size represented by ka is

plotted in Fig. 2-1. The family of curves includes the dependence of antenna radiation

efficiency, er. Therefore, the plot in Fig 2-1 shows the family of curves from

( )

+

kakaer

113 . The top curve is the plot of (2.3) and is for 100% radiation efficiency

(er=1). The vertical axis is logarithmic. So as ka increases, Q decreases rapidly,

indicating a strong relationship between antenna size and radiation Q. Figure 2-1 shows

also how McLean formulation in (2.3) differs from Chu’s (2.2). For electrically small

antennas with ka less the 0.5, the Q values from (2.2) and (2.3) are the same.

Bandwidth is an important parameter of interest for antennas and is related to the

radiation quality factor Q. From a terminal standpoint the antenna can be viewed as a

circuit device. Viewing the antenna as a resonant, parallel RLC circuit, fractional

bandwidth is simply the inverse of Q,

QBW dB /13 = (2.3)

BW3dB denotes the 3-dB bandwidth of the antenna and is defined as

C

LUdB f

ffBW

−=3 (2.4)

where Uf and Lf correspond to the upper and lower 3-dB points and cf is the center

frequency, ( ) 2/LUC fff += , of the frequency band of interest. The 3-dB point is where

the signal is 3 dB below its peak value I the band from fL to fU. Once bandwidth is

determined, Q can be found from (2.3).

10

0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 210-1

100

101

102

ka

QMcLeanChu

er=1

er=0.5

er=0.25 er=0.1

er=1

Figure 2-1 Fundamental limit curves of radiation Q versus ka. The top curve

(solid curve) is a plot of (2.3) and is for efficiency, 1=re , or %100 .

The dashed curve is the plot of (2.2) with efficiency 1=re .

In practice, antenna bandwidth is often defined in terms of VSWR values. Because

Q is defined by the 3 dB points of the equivalent circuit model, Q is not just the inverse

of the VSWR bandwidth, as in (2.3). The relationship between Q and VSWR is

determined as follows [11]:

VSWRBWVSWRQVSWR

1−= (2.5)

For VSWR = 2, (2.5) reduces to

221

=

=VSWRBW

Q (2.6)

The family of fundamental limit curves in Fig. 2-1 can be explained as follows. A

point for an antenna is located on the graph using the Q value for the antenna and its ka

11

value at midband, fc. The enclosing radius, a , of an antenna includes any images of the

antenna in a ground plane. Realizable antennas have Q values above the curve

corresponding to the efficiency of the antenna, re . That is, points corresponding to

realizable antennas must lie above the curves. The curves are also used to find the size

limit for a given bandwidth.

2.3.2 Fundamental Limitations of Electrically Small Antennas

The fundamental limits presented in the previous section are based on theory and several

assumptions are involved. It is therefore important to compare this theory to actual

antennas in order to see if the fundamental limits are also useful for practical antennas.

We present results of such a study in this section.

The antennas of Table 2-1 used in the comparison were found in the literature or

were fabricated at the Virginia Tech Antenna Laboratory. Figure 2-2 shows the

bandwidth-size performance values for the practical antennas listed in Table 2-1 along

with the fundamental limit curve based on (2.3) for 100 % radiation efficiency. To

achieve small size, it is important for a point on the plot to be as close to the desired

efficiency curve as possible. A point that falls on the curve indicates that the antenna has

achieved maximum bandwidth for its size. The example antennas shown in Fig. 2-2

represent a wide variety of antennas and none exceed the fundamental limits. Thus, we

can conclude that the fundamental limits appear to provide realistic limits.

12

Table 2-1 Characteristics of Antennas Used to Examine Bandwidth-Size

Relationships.

Antenna Center Frequency

(fc)

Bandwidth

(VSWR<2)

Radius of Enclosing

Sphere (a)

λ/2 Dipole [12] 300 MHz 36 MHz λ/2

Goubau [13] 300 MHz 75 MHz 0.166λ

IFA [14] 923.5 MHz 17 MHz 3.909 cm

DIFA [14] 917 MHz 30 MHz 3.926 cm

PIFA 859 MHz 70 MHz 3.774 cm

λ/2 Patch [15] 3.03 GHz 32 MHz 2.237 cm

Foursquare [16] 6.0 GHz 2.12 GHz 1.670 cm

0 0.5 1 1.5 2 2.510

-1

100

101

102

DipoleGoubauIFADIFAP IFAP atchFours quare

Q

ka0 0.5 1 1.5 2 2.5

10-1

100

101

102

DipoleGoubauIFADIFAP IFAP atchFours quare

Q

ka

Figure 2-2 Comparison of several practical antenna (Q,ka) values (see Table 2-

1) to the fundamental limits curve based on (2.3) with er = 100%.

13

2.4 Techniques for Reducing Antenna Size

As mobile communication equipment become smaller and lighter, antennas must follow

the trend. Conventional wireless handheld devices use short monopole antennas, normal-

mode helical antennas (or stubby antennas), and microstrip antennas. These solutions are

still in use but have to be improved in terms of physical size because they may be too

large for devices in consumer electronics.

Techniques for making antennas electrically smaller have been known for a long

time, and many of them are described in standard textbook. The principle behind these

techniques will be described here. The main miniaturizing tools used are optimizing the

geometry, using short circuits and ground planes, loading the antenna with high-dielectric

materials, and using the antenna environment such as the casing to reinforce the

radiation. These techniques have been used extensively in the mobile-communication

industries, where the most successful results were obtained by combining several of them

for the design of one antenna.

2.4.1 Use of Short Circuits and Ground Planes

A popular strategy for making antennas electrically smaller and lowering their profiles is

to use ground planes and short circuits. The principle can be easily explained by the well-

known example of the monopole compared to the dipole, as shown in Fig 2-3. A dipole

has a length of roughly half a wavelength at resonance. This dipole height can be halved

by replacing one dipole arm with a ground plane, which will, in turn, create a virtual

dipole arm, according to image theory [12]. The principle can also be extended to planar

antennas by adding short circuits to the ground planes, as in the case of a half-wave

microstrip patch antenna, shown in Fig 2-4, that is reduced in size by a half using a short

circuit that acts as a mirror in the middle of the antenna, forming a quarter-wave patch

antenna.

14

Dipole Monopole over ground plane

Figure 2-3 An example of size reduction using a ground plane.

e

ogaε

λλ22

=≈

e

ogaε

λλ44

' =≈

Figure 2-4 Size reduction using short circuit in microstrip patch antenna.

h = λo/4

h = λo/2

a a

a’a’

15

2.4.2 Optimizing the Antenna Geometry

A fundamental technique for reducing antenna electrical size is to lengthen the path of the

current flow on its structure. An example of a modified antenna leading to a smaller size

shown in Fig. 2-5 consists of a microstrip-patch antenna, reduced in size by inserting

slots into its structure. The slots force the surface currents to meander, thus artificially

increasing the antenna electrical length without modifying its global dimensions. Another

one of the most popular antennas for hand-held communication devices is the meander-

line antenna, as shown in Fig. 2-6. The global dimensions of the patch remain the same

but the path of the current flow is now guided by the meander line of the structure.

(a) 2

gaλ

≅

(b) 2

gaλ

<

Figure 2-5 The effect of notches and slots in a microstrip-patch antenna for size

reduction: (a) regular microstrip patch antenna and (b) microstrip

patch antennas with notches and slots.

a

Js

a

Js

a

JsJs

aa

JsJs

a

16

Figure 2-6 Meander-line printed antenna: an example of reducing antenna size

by increasing its electrical length.

2.4.3 Antenna Loading with High-Dielectric Materials

Loading the antenna can be accomplished by filling the inner space or coating the

antenna with dielectric or magnetic material. Since the wavelength is shorter in a material

with εr and µr greater than 1, where εr and µr are the relative permittivity and

permeability of the material, respectively, the antenna becomes smaller when embedded

in such a material. Embedding an antenna in a material (εr, µr) would enable size

reduction by a factor of rr µε because the phase velocity in the material is rr

cvµε

= ,

where c is the velocity of the wave in vacuum. For finite material loading, size reduction

would be much less. The size reduction depends on the shape of the dielectric material.

Figure 2-7 illustrates the principle with a simple dielectric-loaded monopole. The height

of the monopole can be reduced by a maximum factor of rr µε .

17

(a) 4

ohλ

= (b)44

o

rr

o hλ

µελ

<<

Figure 2-7 Size reduction of a monopole by dielectric loading.

2.4.4 Use of Antenna Environment

Designing an electrically small antenna reduces, in most cases, to finding the best

possible compromise among antenna dimensions and radiation characteristics. A very

efficient way to do this is to make the antenna’s environment, e.g., the casing of the

device, participate in the radiation. In extreme cases, the casing radiates most of the

power, whereas the actual antenna merely acts as a resonator to set the appropriate

working frequency. A practical example of this technique is the “Smart monobloc

integrated-L Antenna (SMILA)” [17], which is derived from the planar inverted-F

antenna concept and is placed on the narrow side of a box, as shown in Fig. 2-8. In this

case described, the box is a round “pillbox”, corresponding to the size of the casing.

Dielectric materialDielectric material

18

Ground

ConformedRadiating strip

Holes forExcitation port

Ground

ConformedRadiating strip

Holes forExcitation port

Figure 2-8 The smart monobloc integrated-L antenna (SMILA) [17].

2.4.5 Effects on Antenna Performance Characteristics

The size of the antenna for a given application is not related mainly to the technology

used, but is determined by the laws of physics. The antenna size relative to the

wavelength is the parameter that will have the dominant influence on the radiation

characteristics. This follows from the fact that an antenna is used to transform a guided

wave into a radiated wave, and vice-versa, and to perform this transformation, efficiently,

the size should be of the order of half a wavelength or larger [18]. Antennas can, of

course, be smaller, but at the expense of bandwidth, gain, and efficiency.

A miniaturized antenna will show a higher concentration of surface currents than

standard antennas, and thus, the ohmic losses will be increased. Therefore, electrically

small antennas cannot have high gain. The work of Wheeler and Chu, on the fundamental

limits of small antennas discussed in Sec. 2.3.1, was extended by Harrington [19] to

include the effect of losses. Harrington derived a very useful and simple formula, more

suitable to antenna design, that gives a practical upper limit on the gain that a small

antenna can achieve while still having a reasonable bandwidth:

19

+

<

λπ

λπ aaG 222 2

max (2.7)

This represents an upper limit for the gain, and the limit is approached for conventional

antennas. The approximation is good for standard antennas, but this limit is more difficult

to reach as the antenna becomes very small. Indeed, in those cases the losses may

increase drastically.

When reducing the antenna size by modifying its geometry or by using short

circuits and a ground plane, as described in Sections 2.4.1 and 2.4.2, there will be two

effects. First, antenna size reduction produces greater current concentrations on the

antennas (see Fig. 2-5 for example), and, therefore, ohmic losses increase and gain

decreases. Second, these techniques – the image effect, the position of short circuits, the

position of the slots – are often frequency sensitive. The antenna bandwidth is thus

reduced, compared to standard antennas. When loading the antennas with dielectric

material, the bandwidth will be reduced due to higher quality factor. This is due to the

concentration of the electric field in high-permittivity or permeability regions, which

makes the launching of the guided wave into free space more difficult. An example for

the case of a microstrip-patch antenna simulated using FDTD method is shown in Fig. 2-

9. The electric field intensity above the patch is much weaker for the case with material

of high permittivity, εr=10.2 (Fig. 2-9b), than that with low permittivity, εr=1.07 (Fig. 2-

9a) because of inherent material behavior and surface waves created within the material.

Moreover, a higher permittivity is, unfortunately, often accompanied by higher dielectric

losses, which degrades the efficiency of the antenna.

20

(a)

(b)

Figure 2-9 Effects of permittivity value on the electric field intensity of a

rectangular microstrip-patch antenna with (a) εr=1.07 and (b) εr=10.2

at resonance simulated using FDTD method. The colors representing

the normalized electric field go from blue (weak E-field) to green to

yellow to red (strong E-field)

21

2.4.6 Examples of Practical Small Antennas for Hand-Held Wireless

Communications

This section presents a few selected antennas that make use of one of the size-reduction

techniques, or a combination of techniques discussed in the previous section. Very

recently, especially after year 2000, many novel and more exotic low-profile antenna

designs have been developed and published in the open literature and may be good

candidates for the present-day mobile cellular communication systems, including the

global system for mobile communications (GSM, 890-960 MHz), the digital

communication system (DCS, 1710-1880 MHz), the personal communication system

(PCS, 1850-1990 MHz), the universal mobile telecommunication system (UMTS, 1920-

2170 MHz), and the wireless local area network systems (WLAN, 2400-2484 MHz and

5150-5350 MHz).

Patch Antenna with a Branch Line Slit [20]

The promising proposed planar inverted-F antenna (PIFA) design in Fig. 2-10 operates in

the two 900/1800 MHz bands as seen by the measured and simulated return loss of that

antenna in Fig 2-11. This design utilizes an asymmetric branch-line slit to meander the

excited surface currents in the top patch, which leads to a large reduction in the required

top patch dimension of the PIFA; the first two resonant frequencies of the meandered

resonant path are also tuned to obtain 900 and 1800 MHz operation. The top patch is

printed on a 0.8-mm FR4 substrate and short-circuited to the antenna’s ground plane with

a shorting plate of length 2 mm. A branch-line slit consisting of a main slit, a long folded

branch slit (branch slit 1 in the Fig. 2-10), and a short bent branch slit (branch slit 2 in the

Fig. 2-10) is embedded in the top patch. The main slit, which has an open end at the patch

boundary, and the long folded branch slit in the top patch are mainly for effectively

meandering the excited patch surface currents. On the other hand, the short bent branch

slit is mainly for achieving impedance matching of the first two excited resonant

frequencies to achieve the operating bandwidth. The excited surface current distributions

22

in the top patch for the two resonances are illustrated in Fig 2-12 at the two resonant

frequencies. It is seen that the current path from the feed point to the portion of the patch

encircled by the long folded slit, determines the excitation of the two desired resonant

frequencies. The patch portion with this current acts as the main radiator in this PIFA

design to resonate at a quarter and a half of a wavelength.

This design is very attractive because of its compactness (30 by 10 by 6.2 mm).

However, enhancement of the bandwidth in the 900-MHz band is needed in order to

satisfy bandwidth requirement from commercial wireless handheld applications. The

bandwidth of the lower band is only 24 MHz, as shown in Fig 2-11.

Figure 2-10 Geometry of the dual-frequency PIFA with a branch-line slit;

dimensions in the figure are in millimeters [20].

23

Figure 2-11 Measured and simulated return loss of the PIFA shown in Fig. 2-10

[20].

(a) f = 950 MHz

(b) f = 1790 MHz

Radiation Region

Radiation Region

Opposite CurrentRegion

Opposite CurrentRegion

(a) f = 950 MHz

(b) f = 1790 MHz

Radiation Region

Radiation Region

Opposite CurrentRegion

Opposite CurrentRegion

Figure 2-12 Simulated patch surface current at (a) 950 MHz and (b) 1790 MHz

for the PIFA shown in Fig. 2-10 [20].

24

Low-Profile Folded Three-Branch Compact Antenna [21]

A planar monopole consisting of three branch strips wrapped into a compact rectangular

box-like structure is illustrated in Fig. 2-13. The proposed monopole is first printed on a

thin FR4 substrate of thickness of 0.4 mm and relative permittivity of 4.4. Then, by

following the dashed lines shown in Fig. 2.13 (b), the antenna is wrapped into a three-

dimensional compact structure like a rectangular box of dimensions 6x6.5x25 mm3,

which makes it suitable for mounting on the top portion of a mobile phone with a small

distance of 3mm to the ground plane.

The three branch strips are design to operate as a quarter-wavelength structure at the

three bands: 900, 1800, and 1900 MHz. The impedance bandwidths covering the required

bandwidths of the GSM, DCS, and PCS bands are shown in Figure 2-14. Branch 1

generates a resonant mode at about 900 MHz for GSM operation. Starting from point A

(the feed point), the strip length (l1) of branch 1 is 143 mm (about 0.43 wavelength at 900

MHz), which is meandered to achieve a compact structure. Branch 2 and branch 3 are

designed to operate at about 1800 and 1900 MHz, respectively, and are about 0.30-

wavelength long. The branches are unbalanced monopole-like, similar to a stubby

monopole antenna, but in planar form.

25

(a)

(b)

Figure 2-13 (a) The branch line planar monopole in a wrapped structure for

GSM/DCS/PCS multi-band mobile phone antenna; (b) the

monopole unwrapped into a planar structure [21].

26

Figure 2-14 Measured and simulated return loss for the monopole in the

wrapped structure shown in Fig. 2-13a [21].

Printed Inverted-F Monopole Antenna [22]

Small dual-band antenna designs for 2.4/5.25 GHz bands for WLAN applications are

high in demand in today’s systems. The printed inverted-F monopole antenna shown in

Fig. 2-15 is an example for such applications. It is printed on a dielectric substrate

capable of integration with the associated circuitry on the same substrate, which can

reduce the fabrication cost and the required volume of the whole system. The antenna

consists of two stacked inverted-F metal strips. The two metal strips and the 50-Ω

microstrip feed line are all printed on the same dielectric substrate, an FR4 substrate of

thickness of 1.6 mm and relative permittivity of 4.4. The two metal strips are short-

circuited with a common shorting pin through a via-hole in the substrate to the ground

plane printed on the other side of the substrate. The larger inverted-F metal strip has

dimensions l1 and h1, and the shorter inverted-F metal strip, l2 and h2, as shown in Fig. 2-

15. Note that the two inverted-F strips are both designed to operate as a quarter-

wavelength structure. In other words, (l1 + h1) and (l2 + h2) can be determined from one

quarter of a wavelength of the lower and the upper frequency, respectively. However, the

smaller inverted-F metal strip electrical length l2 is about 0.35 of a wavelength. This is

probably due to the coupling effects between the strips. The following antenna

27

dimensions gives a return loss shown in Fig. 2-16: h1 = 10 mm, h2 = 5 mm, l1 = 21 mm, l2

= 10 mm, d = 3 mm, and w = wf =3.05.

Figure 2-15 Geometry of the dual-band printed inverted-F antenna [22].

Figure 2-16 Measured input impedance for the antenna shown in Fig. 2-15 with

h1 = 10 mm, h2 = 5 mm, l1 = 21 mm, l2 = 10 mm, d = 3 mm, and w =

wf =3.05 [22].

2.5 Techniques for Widening Impedance Bandwidth for MSA’s

Antennas such as microstrip antennas (MSA) and PIFAs have found several applications

because of their low-profile and conformal geometry. Despite the attractive features,

28

however, their narrow bandwidth (MSA and PIFA bandwidths are about 3% and 7%,

respectively) limit their use [23]. For example, the patch antenna with a branch slit line

[20] in Fig. 2-10 and described in the previous section, can be a good candidate for the

dual-band GSM/DCS for its compactness. Unfortunately, it does not satisfy the

bandwidth requirement for these bands (c.f. Table 1.1 in Chap. 1). As shown in Fig. 2-11,

the bandwidths of the two resonant bands at GSM and DCS are only 24 MHZ and 113

MHz, respectively. GSM and DCS require 70 and 170 MHz, respectively.

Fortunately, there are various techniques available for increasing the impedance

bandwidth of MSA. One method is to increase the antenna height. However, there are

some limitations on how high the antenna can be. Current applications require the

antenna volume to be small. Therefore, increasing the height of the antenna may not

satisfy the size specifications for the required bandwidth. There are also other limitations

from increasing the height that can degrade the characteristics of the antenna. For

instance, the increase in substrate thickness of MSAs is limited by the excitation of

surface waves which causes loss in radiation efficiency [24]. Also, bandwidth is

increased by lowering the dielectric constant value. Suzuki and Chiba [25] showed such a

behavior for MSA where the unloaded quality factor Q is proportional to εr, and the

bandwidth is inversely proportional to Q.

Modifying the shape of the patch of MSAs can increase impedance bandwidth.

Examples of such regular MSA configurations are the modification rectangular and

circular patches to rectangular ring [26] and circular ring [27], respectively. The

bandwidth is larger because of a reduction in the quality factor Q of the patch antenna,

which is due to less energy stored beneath the patch and higher radiation.

There are three other techniques that have been used extensively in recent antenna

designs to enhance bandwidth. They are described in the following sections.

2.5.1 Planar Multi-Resonator Configurations

Planar stagger-tuned coupled multiple resonators yield wide bandwidth in a manner

similar to multistage tuned circuits. Various parasitic narrow strips, shorted quarter-

29

wavelength rectangular patches, and rectangular resonator patches have been gap-coupled

to the central-fed rectangular patch. Three configurations are shown in Fig. 2-17: gap-

coupling along radiating edges, along non-radiating edges, and along all edges [28].

(a)

(b) (c)

Figure 2-17 Various gap-coupled multiresonator rectangular MSAs

configurations: gap-coupling (a) along the radiating edges of the fed

patch, (b) along the non-radiating edges of the fed patch, and (c)

along all edges of the fed patch [28].

The mechanism of parasitic coupling for wide bandwidth is as follows. A parasitic

patch placed close to the fed patch is excited through the coupling between the two

patches. If the resonant frequencies f1 and f2 of the two patches (when isolated) are close

to each other, then broad bandwidth is obtained, as shown in Fig. 2-18. The overall input

VSWR will be the superposition of the responses of the two resonators resulting in a

wide bandwidth. If the bandwidth is narrow for each individual patch, then the difference

between f1 and f2 should be small as shown in Fig 2-18(a). If the bandwidth of the

individual patches is large, then the difference in the two frequencies should be larger,

30

yielding an overall wide bandwidth as shown in Fig. 2-18(b). This can also be explained

in terms of input impedance plot on the Smith chart. A loop observed on the Smith chart

is due to the coupling between the two patches, as illustrated in Fig. 2-19.

(a) (b)

Figure 2-18 VSWR plots of two coupled resonators having (a) narrow and (b)

wide bandwidth: (….) individual resonators and () overall

response.

Gap-coupling effects can be studied by examining the case of one parasitic element,

as shown in Fig. 2-19(a). The dimensions of the configuration are as follows: L=3 cm,

W=4 cm, L1=2.9 cm, x=0.7 cm and s=0.1 cm. The substrate parameters are εr = 2.55,

h=0.159 cm, and tan δ= 0.001. The computed input impedance and VSWR of the

configuration simulated using IE3D, a method of moment EM software, are plotted in

Figure 2-19 (b,c). The loop is observed in the impedance plot due to coupling between

the two patches. To shift the loop closer to resonance (i.e., to increase the impedance at

resonance close to 50 Ω, assuming that the probe-feed has the impedance of 50 Ω), the

feed point is shifted to x=1.1 cm; see Fig. 2-20. Now the loop is within the VSWR=2

circle. For x=1.1 cm, the bandwidth for VSWR<2 is 207 MHz (7%). This bandwidth is

more than three times that of a single rectangular MSA.

31

(a)

VSWR = 2VSWR = 2

(b) (c)

Figure 2-19 A single parasitic patch element gap-coupled with one fed patch

antenna: (a) geometry, (b) computed input impedance, and (c)

VSWR plots of a single fed patch with no parasitic element (---) and

with one parasitic patch element () [28].

x

W

L L1s

x

W

L L1s

32

VSWR = 2VSWR = 2 (a) (b)

Figure 2-20 Computed input impedance (a) and VSWR plots (b) of two gap-

coupled patches along radiating edge for two feed point locations x

of the parasitic MSA of Fig. 2-19a: (---) 0.7 cm and () 1.1 cm

[28].

The length of the parasitic patch determines the position of the loop on the Smith

chart. The input impedance and VSWR plots are shown in Fig. 2-21 for three different

length of the parasitic patch, L1 = 2.8, 2.9, and 3.0 cm. As L1 increases, the resonance of

the parasitic patch decreases, and the loop is formed at the lower frequency of the fed

patch in the impedance plot. The position of the loop moves anti-clockwise direction on

the Smith chart.

Figure 2-21 Computed input impedance (a) and VSWR plots (b) of two gap-

coupled patches along radiating edge for three values of L1 of the

parasitic MSA of Fig. 2-19a: (---) 2.8 cm, () 2.9 cm, and (— – —

) 3.0 cm [28].

33

The gap between the two patches controls the size of the impedance loop. Figure 2-

22 shown this gap effect for three different values of gap: s = 0.05, 0.1, and 0.15 cm. As

the gap increases, the gap coupling decreases and therefore the loop size decreases.

Maximum bandwidth is obtained when the loop in the impedance plot is completely

inside the VSWR=2 circle and its size is as large as possible. For s = 0.1 cm, broader

bandwidth of 207 MHz is obtained.

(a) (b)

Figure 2-22 Computed input impedance (a) and VSWR plots (b) of two gap-

coupled patches along radiating edge for three values of s for the

parasitic patch antenna of Fig. 2-19a: (---) 0.05 cm, (— – —) 0.1

cm, and () 0.15 cm [28].

The effects of adding one parasitic element to the fed patch give rise to asymmetry

in the radiation patterns. Figure 2-23 illustrates computed radiation patterns in the E- and

H-planes at three different frequencies (2.9, 3.0, and 3.1 GHz). In the H-plane, there is

not much change in the radiation pattern at these three frequencies. However, in the E-

plane, the beam maximum shifts away from broadside. At 3.1 GHz, the dominant

radiation comes from the parasitic patch. Since the parasitic patch sees a phase delay

compared to the fed patch, the beam maxima shift away from broadside.

34

Figure 2-23 Computed radiation patterns of two gap-coupled patches along

radiating edge at frequencies (a) 2.9 GHz, (b) 3.0 GHz, and (c) 3.1

GHz: () E-plane, (---) H-plane [28].

To remedy the effect of radiation pattern asymmetry, two identical parasitic patches

can be used and are placed along the radiating edges of the fed patch, as shown in Fig. 2-

17(a). In this arrangement, both the parasitic patches are on the opposite sides of the fed

patch, one patch will shift the beam maxima in the +θ direction, while the other patch

will shift it in the -θ direction. The overall pattern of the three patches will be the

superposition of the individual pattern, and hence will remain symmetrical with broadside

direction, as shown in Fig. 2-24. Note also that there is an effect on the impedance when

adding an identical parasitic patch to the configuration. The loop size is larger compared

to the two-coupled patch configuration. This is due to the fact that the two identical

parasitic patches are resonant at the same frequency and hence the coupling increases. To

counter balance this effect, one can increase the gap between the fed and the parasitic

patches.

35

(a) (b)

Figure 2-24 Computed radiation patterns of three gap-coupled patches along

radiating edges at frequencies of Fig. 2-17a: (a) 2.89 GHz and (b)

3.09 GHz: () E-plane, (---) H-plane [28].

The bandwidth can be further increased by using unequal-length parasitic patches.

The same mechanism occurs. The overall impedance bandwidth is the superposition of

the responses of the three unequal-length resonators. However, this configuration will

give rise to asymmetrical radiation pattern.

Coupling along the non-radiating edges of fed patch as in the configuration shown

in Fig. 2-17(b) has similar effects previously discussed, except that the effects are smaller

compared to those from coupling along radiating edges because the field varies

sinusoidally along the non-radiating edges.

2.5.2 Multilayer Configurations

In some cases, the planar multi-resonator arrangement discussed in the previous section

may not be permitted due to physical size constraint because such an arrangement is too

large for many applications. In multilayer arrangement, two or more patches on different

36

layers of dielectric substrate are stacked on each other. Based on the coupling

mechanism, these configurations are categorized as electromagnetically coupled or

aperture-coupled MSAs. The two types of configurations are shown in Fig 2-25. In

electromagnetically coupled MSA, one or more patches at the different dielectric layers

are electromagnetically coupled to the feed line located at the bottom dielectric layer.

Alternatively, electromagnetic coupling can be obtained between two patches with one

fed by a coax probe. One the other hand, in the aperture-coupled MSA, the field is

coupled from the microstrip line placed on the other side of the ground plane to the

radiating patch through an electrically small aperture or slot in the ground plane, as in

Fig. 2-25(b).

Microstrip line feed Coaxial Probe feed

(a)

(b)

Figure 2-25 Configurations of (a) Electromagnetically-coupled and (b) aperture-

coupled MSAs.

37

The multilayer broadband MSAs, unlike single-layer multiresonator configurations,

show a very small degradation in radiation pattern over the complete operating frequency

band. The drawback of these antennas is the increase in height. The following section

describes briefly the mechanism of electromagnetically-coupled of coaxial-fed

multiplayer MSAs, which provides helpful information for the next chapters on the

analysis of the proposed wide-band antenna.

The geometry of the coaxial-fed electromagnetically MSA under study is shown in

Fig. 2-26. It consists of two square patches with side lengths L1 and L2 constructed on two

different substrates εr1 and εr2. The two patches are above a ground plane of size large

enough so that it can be considered infinite. There are three size parameters whose effects

need to be examined: the length of the top square patch, the length of the bottom square

patch, and misalignment of the top square patch. The moment method was used to

investigate these parameter and results are now presented.

(a) (b)

Figure 2-26. Geometry of a stacked microstrip patch coaxial-fed on a large

ground plane: (a) top and (b) side views.

Effect of Varying the Length of the Top Square Patch

The lower square patch length L1 is 2.5 cm. Both substrates have parameters εr=2.2,

h=0.159 cm, and tan δ=0.001. The resonant frequency of the lower patch in this

38

configuration is about 3.5 GHz. The top square patch length L2 is varied from 0.5 and 2.6

cm. For the feed point at x=0.85 cm, the theoretical input impedance and VSWR plots,

obtained using IE3D, for four values of L2 are shown in Fig. 2-27. As L2 is increased

from 0.5 to 2 cm, there is no change in the impedance plot in the frequency range from

3.3 GHz to 4.1 GHz. This is because the resonance frequency of the top patch is much

larger than that of the fed patch. As L2 increased to 2.3 cm, a loop appears in the

impedance plot in the above frequency range because the resonance frequency of the top

patch is now comparable with that of the fed patch. As the length of the top patch is

increased to 2.6 cm, its resonance frequency decreases further. Also, the size of the loop

increases because the coupling between the two patches increases due to a larger overlap

area. For L2 = 2.45 cm, the loop is contained within the VSWR=2 circle, giving a

bandwidth of 315 MHz (8.4%), which is larger than that of a single patch antenna of the

same overall dimensions and substrate, about 200 MHz. Thus, a two-stacked square MSA

yields 50% more bandwidth as compared to a single-square MSA with the same volume.

(a) (b)

Figure 2-27 Impedance and VSWR over the frequency range from 3.3 to 4.1

GHz for the stacked microstrip patch of Fig. 2-26 simulated to

examine upper patch size variation for four values of L2: (— – —)

2.0 cm, (---) 2.3 cm, (——) 2.45 cm, and (…) 2.6 cm [28].

39

Effect of Varying Length of the Bottom Square Patch

The upper square patch length L2 is fixed at 2.5 cm and the substrate parameters are

chosen the same as above, as well as the feed position. Figure 2-28 illustrates the input

impedance and VSWR plots for three different values of L1 (2.50, 2.55, 2.60 cm). As L1

increases from 2.5 cm to 2.6 cm, the resonance frequency of the bottom patch becomes

smaller than that of the top patch, and hence, the loop due to the parasitic patch is formed

in the higher frequency region of the fed patch. This effect is similar to the previous case

when the upper patch is smaller than the bottom patch. The increase of L1 moves the loop

clockwise in the impedance plot. It may be observed that the difference in the two patch

dimensions is more important than their actual values for influence on the input

impedance.

(a) (b)

Figure 2-28 Impedance and VSWR over the frequency range from 3.3 to 3.9

GHz for the stacked microstrip patch of Fig. 2-26 simulated to

examine lower patch size variation for three different values of L1:

(— – —) 2.50 cm, (——) 2.55 cm, and (---) 2.60 cm [28].

40

Effect of Misalignment of the Top patch

Figure 2-29 shows the numerical input impedance and VSWR plots for L1 = 2.5 cm, L2 =

2.45 cm, and x = 0.85 for a 0.1-cm misalignment of the top patch with respect to the

bottom patch along ox and oy axes. Misalignment along the width of patch, i.e, oy = 0.1

cm gives no appreciable change in the performance of the antenna. However, when the

offset is along the ox axis, i.e. ox = ±0.1 cm, the size of the impedance loop increases.

Since the increased loop size is still within the VSWR=2 circle for an offset ox along the

length of the patch, the bandwidth of the antenna increases, which is the desired effect.

(a) (b)

Figure 2-29 Impedance and VSWR over the frequency range from 3.4 to 4.0

GHz for the stacked microstrip patch of Fig. 2-26 simulated to

examine the effect of misalignment of the top patch from the lower

patch for different offset values: (——) not offset, (…) oy=0.1 cm,

(— – —) ox=0.1 cm, and (---) ox=-0.1 cm [27].

2.6 Summary

A review of literature on small antennas and related topics was presented. It was shown

that reducing antenna size degrades antenna performance, including impedance

41

bandwidth and that there is a fundamental limit on how small an antenna can be reduced

without degrading its radiation efficiency performance.

Several techniques for reducing antenna size were described, including material

loading, the use of short circuit and ground plane, antenna geometry optimization to

lengthen current flow such as meander-line technique, and the use of the environment as

a radiating element. Reducing the size is always accompanied by impedance bandwidth

degradation. To enhance such a performance degradation, a few methods were presented

using coupling effects and parasitic elements in the antenna structure with various

configurations.

2.7 References

[1] R.A. Burberry, “Electrically Small Antenna: A Review,” IEE Colloquium on

Electrically Small Antenna, Oct. 1990.

[2] H.A. Wheeler, “Fundamental Limitations of Small Antennas,” Proc. IRE, vol. 35,

Dec. 1947, pp. 1479-1484.

[3] L.J. Chu, “Physical Limitations on Omni-Directional Antennas,” J. Appl. Phys.,

vol. 19, Dec. 1948, pp.1163-1175.

[4] R.F. Harrington, “Effect of Antenna Size on Gain, Bandwidth, and Efficiency,” J.

Res. Nat. Bur. Stand., vol64-D, Jan/Feb 1960, pp. 1-12.

[5] R.E. Collin and S. Rothschild, “Evaluation of Antenna Q,” IEEE Trans. Ant.

Prop., vol. AP-12, Jan. 1964, pp.23-27.

[6] R. Fante, “Quality Factor of General Ideal Antennas,” IEEE Trans. Ant. Prop.,

vol. AP-17, March. 1969, pp. 151-155.

[7] J.S. Maclean, “A Re-Examination of the Fundamental Limits on the Radiation Q

of Electrically Small Antennas,” IEEE Trans. Ant. Prop., vol. AP-44, no. 5, May

1996, pp. 672-676.

[8] E.D. Caswell, W.A. Davis, and W.L. Stutzman, “Fundamental Limits on Antenna

Size,” Submitted to IEEE Trans. Ant. Prop., April 2000.

42

[9] D.M. Grimes, C.A. Grimes, “Radiation Q of Dipole Generated Fields,” Radio

Science, vol. 34, No. 2, pp. 281-296.

[10] C.A. Grimes, and D.M. Grimes, “Minimum Q of Electrically Small Antennas: A

Critical Review,” Microwave and Optical Technologies Letter, vol. 28, no. 3, Feb.

2001, pp. 172-177.

[11] K.R. Carver and J.W. Mink, “Microstrip Antenna Technology,” IEEE Trans. Ant.

Prop., vol. AP-29, Jan. 1981, pp. 2-24.

[12] W.L. Stutzman and G.A. Thiele, Antenna Theory and Design, 2nd Ed., Wiley,

New-York: 1998.

[13] R.C. Hansen, “Fundamental Limitations in Antennas,” Proc. IEEE, vol. 69, Feb.

1981, pp. 170-182.

[14] A.T. Gobien, Investigation of Low-Profile Antenna Designs for Use in Hand-held

Radios, Antenna Group Report 97-4, August 1997.

[15] R.C. Hansen, “Fundamental Limitations in Antennas,” Proc. IEEE, vol. 69, Feb.

1981, pp. 170-182.

[16] C. Buxton, W.L. Stutzman, and R. Nealy, “Analysis of a New Wideband Printed

Antenna Element – The Foursquare – Using FDTD Techniques,” IEEE Ant. Prop.

Int. Symp., Atlanta, June 1998.

[17] J.-F. Zürcher, A.K. Skrivervik, and O. Staub, “SMILA: A miniaturized antenna

for PCS Applications,” Ant. And Propag. Soc Internat. Symposium, vol. 3, Jul.

2000, pp. 1646-1649.

[18] O. Staub, J.-F. Zürcher, A.K. Skrivervik, and R Mosig, “PCS Antenna Design:

The Challenge of Miniaturisation,” Ant. And Propag. Soc Intern. Symposium, vol.

1, Aug. 1999, pp. 548-551.

[19] R.F. Harrington, “Effect of Antenna Size on Gain, Bandwidth, and Efficiency,”

Journal of Research of the National Bureau of Standards – D. Radio

Propagation, 64D, Jan.-Feb 1960, pp. 1-12.

[20] F.R. Hsiao, H.T. Chen, G.Y. Lee, T.W. Chiou, and K.L. Wong, “A dual-band

planar inverted-F patch antenna with a branch-line slit,” Microwave Opt. Technol.

Lett., vol.32, Feb 20, 2002, pp. 310-312.

43

[21] P.L. Teng and K.L. Wong, “Planar monopole folded into a compact structure for

very-low-profile multi-band mobile phone antenna,” Microwave Opt. Technol.

Lett., vol. 33, April 5, 2002, pp.22-25.

[22] Y.L. Kuo, T.W. Chiou, K.L. Wong, “A novel dual-band printed inverted-F

antenna,” Microwave Opt. Technol. Lett., vol. 31, Dec. 5, 2001, pp.353-355.

[23] K. Hirisawa and M. Haneishi, Analysis, Design, and Measurement of small and

Low-Profile Antennas, Artech House, Boston: 1992.

[24] C.K. Aamandan, P. Mohanan, and K.G. Nair, “Broad-Band Gap Coupled

Microstrip Antenna,” IEEE Trans. Ant. Prop., vol. AP-38, no. 10, Oct. 1990, p.

1581.

[25] Y. Suzuki and T. Chiba, “Designing Method on Microstrip Antennas Considering

Bandwidth,” Trans. IECE of Japan, vol. E67, Sept. 1984, pp.488-493.

[26] V. Palanisamy and R. Garg, “Rectangular ring and H-shaped microstrip Antennas

alternative to rectangular patch antennas,” Electronics Letter, vol. 21, no. 19,

1985, pp. 874-876.

[27] W.C. Chew, “A broadband annular ring microstrip antennas,” IEEE trans. Ant.

Propag., vol. 30, Sept. 1982, pp. 918-922.

[28] G. Kumar and K.P. Ray, Broadband Microstrip Antennas, Artech House Inc.,

Boston: 2003.

44

Chapter 3

Antenna Efficiency Measurements

3.1 Introduction

Radiation efficiency is an important characteristic of antennas for wireless portable

devices. Radiation efficiency, η, is the fraction of input power that ultimately ends up as

radiated power [1]. It decreases as dissipative losses on the antenna increase compared to

the antenna radiation resistance. For antennas that are on the order of a half-wavelength

in maximum extent or smaller, radiation efficiency is usually the limiting factor in the

antenna performance. Antenna radiation efficiency, thus, can have a great effect on the

overall system performance.

There are several methods available to measure antenna efficiency. The technique

that is considered the most accurate is called the pattern integration method. In this

method, radiation intensity obtained through measurements is integrated over a spherical

surface, usually in the far-field region, that completely encloses the antenna. The method,

however, requires good calibration of the antenna range or anechoic chamber in order to

achieve high accuracy. It is also often very time consuming, especially when the antenna

has complicated radiation pattern or polarization. If a range or anechoic chamber is not

available, this technique cannot be used.

Fortunately, there are other methods that require less measurement and computation

effort than that required for the pattern integration method, yet provide accurate results.

45

The resistance comparison method uses two identical antennas except that they are made

from two different metals, with slightly different conductivities. Measurement of their

input resistance values is performed to compute the efficiency of either antenna according

to [2]. However, this technique requires constructing an extra antenna. Another

technique, called the Q-Factor method [2], measures the Q-factor of the antenna. It is by

far the fastest and easiest method for this kind of measurement. Unfortunately, the

measured Q-factor has to be compared to the theoretical Q value of the antenna in order

to compute the efficiency. The theoretical value of Q for an ideal lossless antenna is

difficult to compute, especially when the antenna structure is not simple.

A technique that is quick and easy to use is called the Wheeler Cap method, which

was introduced by H.A. Wheeler in the late 1950’s for measuring the radiation efficiency

of small antennas [3]. Numerous papers related to this method followed with variations of

the technique such as using different size and shape of the conducting cavities for

different types of small antennas; all yield high measurement accuracy [5,6]. The

Wheeler Cap method was originally designed for electrically small single-element

antennas. An extension of this method to a moderate-size and wideband antennas is

derived in this chapter and results from simulations and measurements are presented to

validate the method.

3.2 Wheeler Cap Method on Small Antennas

The radiation efficiency, η, of an antenna is the ratio of the total power radiated to the

power accepted at its terminal and can be written in the following two forms:

LR

R

IN

R

PPP

PP

+==η (3.1)

LR

R

RRR+

=η (3.2)

where PR is the power radiated, PIN is the power supplied to the antenna, and PL the

power lost in the antenna due to mechanisms like ohmic heating. For simple antennas that

46

can be modeled as a series or parallel RLC circuit, their input resistance can be described

with two resistances in series, RR and RL, which represent the radiation resistance and the

loss resistance, respectively, as shown in Fig 3-1. The quantity (RR+RL) is the real-value

part of input impedance at the terminals of the antenna. This quantity can be easily

measured. This leaves the problem of determining how the input resistance separates into

radiation and ohmic loss portions.

Figure 3-1 Circuit model of an antenna resistance showing radiation and loss

resistances.

Wheeler [3] suggested enclosing the antenna with a hollow conductive sphere of

radius πλ

2 to eliminate the radiation resistance RR from the input impedance. This

assumes that the conducting sphere causes no change in the current distribution on the

antenna. If the assumption is correct, as will be discussed later, the real part of the input

impedance with the sphere in place is the loss resistance RL. The assumption of the

undisturbed current distribution is important because if it is not true, then the power lost

due to RL measured with the sphere in place is not the same as that measured in free

space and the method cannot be applied to measure the efficiency of the antenna. The

hollow metallic sphere of radius of πλ

2 is called the radiansphere and defines the space

in which the reactive power density exceeds the radiation power density of the antenna.

RRPR

RLPL

IIN

PREFLPIN S11

47

The role of the conducting sphere is to eliminate the antenna radiation while causing no

disturbance to the near field.

For electrically small antennas (extent <<λ), the current distribution is almost the

same at any frequency. For example, a short dipole (L<<λ) has approximately a

triangular-shape current distribution and remains nearly unchanged as frequency changes,

as long as L<<λ. Therefore, the near fields of electrically small antennas are almost

quasi-static and disturbance due to nearby metallic objects is negligible. A mathematical

proof that the near fields of electrically small antennas inside a conducting sphere remain

undisturbed using scaling and asymptotic analysis can be found in [7]. It has also been

determined that neither the size, shape, nor electrical conductivity of the cavity are

critical [2].

An attractive feature of the Wheeler cap method is that it is easy to implement in

practice, requiring only two measurements: the S11 with and without the cap in place. It is

practical to express the efficiency in terms of reflection coefficient S11 because it is easy

to measure directly using a network analyzer. The first measurement consists of

measuring the reflection coefficient of the antenna in free space, S11FS. It can be related to

the total input resistance, RIN, as we now derive. In free space, the power lost due to

radiation and ohmic loss from the circuit in Fig. 3-1 is

( )LRFS

INFS

FS RRIRIP +==22

(3.3a)

where IFS is the input current of the antenna. It can be also expressed in terms of power

available, PAVAIL, to the antenna as

AVAILFS

FS PSP

−=

2

111 (3.3b)

Thus, equating (3.3a) and (3.3b), the total resistance (RR +RL) is

−=+

2

112 1 FS

FS

AVAILLR S

I

PRR (3.4)

The second measurement of the antenna reflection coefficient is performed with the

cap in place, S11WC. The power lost in this case is only due to ohmic loss RL, since RR is

suppressed by the cap. We assume also that the loss tangent of the conducting cap is high

48

enough so that its resistive loss RS is negligible and power is not absorbed into the cap.

Then, analogous to (3.4), we have

−=

2

112 1 WC

WC

AVAILL S

I

PR (3.5)

where IWC is the input current of the antenna enclosed in the cap. The efficiency can be

computed using (3.4) and (3.5) as

2

2

2

11

2

11

1

11

WC

FS

FS

WC

LR

R

I

I

S

S

RRR

−

−−=

+=η (3.6)

The input current can be expressed in terms of incident and reflected waves [8]:

11aSbZ

baIo

=

−=

( )oZ

aSI 111−

= (3.7)

where a and b correspond to incident and reflected waves, respectively, and Zo is the

characteristic impedance. The ratio of the input current in free space and with the cap

present in terms of S11 can be found using (3.7) in (3.6) as

2

11

2

11

2

11

2

11

2

2

1

1

)1(

)1(

WC

FS

o

WC

o

FS

WC

FS

S

S

ZaS

ZaS

I

I

−

−=

−

−

= (3.8)

The efficiency in terms of the two measurements of S11 with and without the cap in place,

as found from (3.8) in (3.6), becomes

2

11

2

11

2

11

2

11

1

1

1

11

WC

FS

FS

WC

S

S

S

S

−

−

−

−−=η (3.9)

49

Equation (3.9) shows that efficiency can be evaluated directly from reflection coefficient

measurements, without any need to actually determining the resistances RR and RL.

Electrically small antennas (extent <<λ) are always non-resonant antennas that have

non-zero input reactance. However, since the antenna model used in the Wheeler cap

method, as shown in Fig. 3-1, includes no reactance, there is an implicit assumption that

the measurements must be taken at resonance, where the reactance is zero. This problem

can be solved by tuning out the reactance at a frequency of interest by adding a

transmission line to the feed of the antenna without affecting the antenna input resistance

value so that the Wheeler cap method can be used to measure their radiation efficiency

[4]. If the reactance is not tuned out, i.e. the antenna is not resonant, and Wheeler cap

method is used, there will be some inaccuracy in the results since this method does not

take into account the reactance part of the antenna input impedance. This is the major

criticism of this method and McKinzie demonstrated in [4] that the assumed model in

Wheeler method can indeed be a poor one.

3.3 Extension of Wheeler Cap Method to Wideband Antennas

The traditional Wheeler Cap method described in the previous section for evaluating

antenna radiation efficiency assumes that the antenna can be modeled as a series or

parallel RLC circuit and that the antenna operates near resonant point where the antenna

reactance is negligible. Also, it is assumed that the antenna input resistance can be

modeled as a series of resistances, namely the loss resistance RL and radiation resistance

RR because the power available is either dissipated by losses in the antenna or radiated.

Under these assumptions, the antenna efficiency can be evaluated by (3.1) or (3.2).

However, more complicated antennas, including wideband antennas, often have more

complicated loss mechanisms that cannot be modeled by either a series or parallel RLC

circuit, in which case Wheeler cap method may not be a valid technique to determine

efficiency [7]. Furthermore, even if the impedance of a wideband antenna is modeled as

purely resistive and a traditional Wheeler cap method is used for computing radiation

efficiency, it is still difficult to apply this method because the cap size depends on

operating frequency. In fact, to evaluate the radiation efficiency of a wideband antenna

50

using the traditional method, one must perform S11 measurements at each frequency of

interest and also adjust the cap size. Therefore, this method is not practical for evaluating

radiation efficiency of wideband antennas.

A new method was developed by Schantz [9] for evaluating the radiation efficiency

of ultra-wideband (UWB) antennas. Instead of inhibiting radiation by employing a

spherical cap of radius r≈πλ

2 enclosing the antenna, a larger cap is used to perform

measurements. For a cap radius larger than the radiansphere πλ

2 (the distance from the

antenna at which the reactive and radiating fields are equal), the antenna is allowed to

radiate freely and then receives its own transmitted-reflected signal, as depicted in Fig. 3-

2. For a wideband antenna, the Wheeler cap should then be larger than πλ

2 at the low

frequency end of the band of interest.

S11FS

πλ

2>r

Free Space

S11WC

Wideband Wheeler Cap

Antenna

S11FS

πλ

2>r

Free Space

S11WC

Wideband Wheeler Cap

Antenna

(a) (b)

Figure 3-2 Illustration of the method for determining radiation efficiency of

wideband antennas. Instead of inhibiting radiation, a Wideband

Wheeler Cap allows the antenna to transmit and receive the

reflected signal [9]: (a) Antenna radiating in free space and (b) A

“Wideband Wheeler Cap,” sized to be larger than the radianshpere

dimension of the traditional Wheeler cap.

51

This wideband Wheeler cap method assumes that various reflections occur inside

the cap and can be represented by power fractions. The analysis can be visualized using

the power budget link for a transmit-receive pair if all the transmitted power is available

at the receive antenna, as shown in Fig. 3-3. This assumption is valid if the sphere is

made of a conductor with high loss tangent so that there is negligible loss due to the cap.

When a signal is incident on the input port of an antenna, a fraction of its power is

reflected ( 211

ReFS

In

flected SP

Pq == ) due to impedance mismatch and the remaining fraction

(1-q) passes through the port on to the antenna. This portion is then radiated with

radiation efficiency er, representing the dissipation on the antenna. In other words, the

fraction of radiated power is (1-q)er. When the antenna receives the signal reflected from

the Wheeler cap, it is absorbed by the antenna with an efficiency η and an impedance

mismatch (1-q) at the port. However, due to the mismatch at the port, part of the received

signal is retransmitted with fraction of qer. This reradiated portion is then reflected back

to the antenna and this process of reradiation-reflection continues. The power fraction

link budget of the antenna enclosed by the UWB Wheeler cap is shown in Fig. 3-4.

1 TX

q

RXer er (1-q)2er2

(1-q)qer3

1 TX

q

RXerer erer (1-q)2er2

(1-q)qer3

Figure 3-3 Power budget for a TX-RX pair of the wideband Wheeler cap

method if all the transmitted power is available at the receive

antenna.

In terms of power fractions, the scattering coefficient inside the UWB Wheeler cap

becomes:

52

( ) ( ) ( ) ( ) ( ) ( ) ( )( ) ( ) ( )[ ]( ) ( )∑

∞

=

−+=

++++−+=

+−+−+−+−+=

0

222

3222222

3422232222222211

1

11

1111

n

n

r

rrrr

rrrrWC

qeqq

qeqeqeeqq

qeqqeqqeqeqqS

η

K

K

(3.10)

The summation in the second term of (3.10) is an infinite power series for which there is

a simple closed form: ( )∑∞

= −=

02

2

11

k r

k

r qeqe . So, (3.10) can be written as

( )qe

eqqSr

rWC 2222

11 111

−−+= (3.11)

Note that (3.11) is only applicable for q = |S11FS|2 ≠ 1 (not a perfectly reflective antenna).

Otherwise, there would be no energy going to the antenna and the method would not be

applicable. However, |S11FS| is generally not equal to unity in practice. Solving (3.11) for

re and substituting for 211FSSq = yields the following result for radiation efficiency in

terms of scattering parameters S11, one measured when the antenna under test is in free-

space, |S11FS|2, and the other when the antenna is inside the cap, |S11wc|2:

211

211

211

211

211

21 WCFSFS

FSWCr

SSS

SSe

+−

−= (3.12)

For the special case where q = |S11FS|2 = 0 (perfectly matched antenna), the radiation

efficiency WCS11=η and only one measurement is needed.

UWB Wheeler Cap

Antenna Under Test

1

q

( ) 221 req−

( ) ( ) qeq r2221−

( ) ( ) 23221 qeq r−

er

( ) req−1( ) qeq r

31−

( ) 251 qeq r−

( ) 371 qeq r−

UWB Wheeler Cap

Antenna Under Test

1

q

( ) 221 req−

( ) ( ) qeq r2221−

( ) ( ) 23221 qeq r−

erer

( ) req−1( ) qeq r

31−

( ) 251 qeq r−

( ) 371 qeq r−

Figure 3-4 Power fraction link budget for an antenna inside the wideband

Wheeler cap.

53

A more useful figure of merit is the efficiency that includes impedance mismatch

factor (1-q), because it cannot usually be separated from radiation efficiency. Impedance

mismatch can be a significant loss contribution to the total efficiency, especially for

electrically small antennas. It is, therefore, useful to evaluate the efficiency including

impedance mismatch. This contribution can easily be included by multiplying the

radiation efficiency of (3.12) by the power loss due to this mismatch factor. The total

efficiency ηT can then be computed using the following equation:

( ) 211

211

211

211

2112

1121

1)1(WCFSFS

FSWCFSrT

SSS

SSSeq

+−

−−=−=η (3.13)

3.4 Experimental Results of Efficiency Evaluation Using Wideband Wheeler Cap

The experimental setup used to measure antenna efficiency is shown in Fig. 3-5. The

Wheeler cap radius is 15 cm, permitting radiation efficiency measurement down to 300

MHz. There are two measurements to make: one set of S11 for the antenna under test in

free space and one when the antenna is enclosed in the Wheeler cap. Experimental results

are then compared to numerical ones for a few antennas.

Figure 3-5 Antenna radiation efficiency measurement setup using a 15-cm radius

wideband Wheeler cap probed by an HP 8720 network analyzer.

54

3.4.1 Lossy Monopole on Finite Ground Plane

A lossy monopole on finite ground plane is illustrated in Fig. 3-6. The monopole is 28

mm long placed near the center of a 100 by 140 mm ground plane. It is loaded with a 50-

Ohm resistance (4 mm long) at its tip so that some dissipation loss is inserted for the

purpose of antenna efficiency evaluation. The monopole including the resistor is 34 mm

long. The measured |S11| in free space and in the spherical cap are shown in Fig. 3-7. The

antenna in free space is well matched at 2.2 GHz, as indicated in Fig 3-7a. Also note the

good agreement between measured and simulated results. When the antenna is enclosed

in the cap, spherical resonant modes are excited at certain frequencies. This behavior

manifests itself as sharp, narrowband decreases in the measured return loss of the antenna

in the cavity, as illustrated in Fig. 3-7b, and ultimately, will introduce error in the

efficiency evaluation. To remove this artifact, the raw data of the return loss for the

antenna in the cavity case is smoothed by taking the largest value for the raw data over a

100 MHz span, assuming that the impedance characteristics of the antenna does not

change drastically within that span. The processed data in this fashion are shown in Fig.

3-8; note that the dropouts are removed. Radiation efficiency including impedance

mismatch is then evaluated with the processed data using (3.13) and the result is shown in

Fig. 3-9. The numerical value of the lossy monopole efficiency from simulations using

FDTD method (Fidelity software), also shown in Fig. 3-9, agrees well with the measured

values.

50-Ω resistor50-Ω resistor

Figure 3-6 Lossy monopole on a finite ground plane (100 by 140 mm) with a

50-Ohm resistor at the tip of the antenna to insert some dissipation

loss in the antenna.

55

1 2 3 4 5-25

-20

-15

-10

-5

0

Frequency, GHz

|S11

|

MeasuredSimulated (FDTD)

(a)

1 2 3 4 5-16

-14

-12

-10

-8

-6

-4

-2

0

2

Frequency, GHz

|S11

|

(b)

Figure 3-7 Magnitude of S11 measured using the setup in Fig. 3-5 of the lossy

monopole of Fig. 3-6: (a) in free space, and (b) in the spherical cap

of 15-cm radius.

56

1 2 3 4 5-16

-14

-12

-10

-8

-6

-4

-2

0

2

Frequency, GHz

|S11

|

Raw dataSmooth Peak

Figure 3-8 Processed |S11| (blue curve) of the monopole in the spherical cap by

taking the largest value for the raw data over 100 MHz span,

assuming that the impedance characteristics of the antenna does not

change drastically within that span.

1 2 3 4 50

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

Frequency, GHz

Tota

l Effi

cien

cy η

T

MeasuredSimulated (FDTD)

Figure 3-9 Total efficiency ηT (radiation efficiency including impedance

mismatch) of the lossy monopole of Fig. 3-7 evaluated using (3.13)

with experimental data (solid curve) and numerical data (asterisk).

57

3.4.2 Planar Half-Disk UWB Monopole Antenna

The planar half-disk UWB monopole antenna [10] shown in Fig. 3-10 is designed

to operate from 3.1 to 10.6 GHz, matching the FCC approved UWB communication

band. The antenna, including the half-disk element of 12.7 mm radius (Rm) and a 2.4-mm

width (Fw) microstrip line connecting the half-disk element to the feed port, is printed on

one side of a substrate with permittivity of 2.33. A ground plane is printed on the other

side of the substrate as shown in Fig. 3-10. The measured S-parameter raw data shown in

Fig. 3-11a demonstrates that the antenna has acceptable return loss (> 10 dB). In order to

suppress cavity resonance, the return loss measured when the antenna is enclosed in the

cap is smoothed by taking the largest value of the raw data over a 100 MHz span. The

resulting smoothed return loss data (solid curve in Fig. 3-11b) has the efficiency

including impedance mismatch from the measured return loss is evaluated using (3.13)

and is plotted in Fig. 3-12.

Gh = 25.4

H = 57.2

Fw = 2.4

L = 76.2

Rm = 12.7

Dimensions in mm

Gh = 25.4

H = 57.2

Fw = 2.4

L = 76.2

Rm = 12.7

Dimensions in mm

Gh = 25.4

H = 57.2

Fw = 2.4

L = 76.2

Rm = 12.7

Dimensions in mm Figure 3-10 A planar half-disk UWB monopole antenna designed for operating

from 3.1 to 10.6 GHz [10].

58

0 2 4 6 8 10 12-35

-30

-25

-20

-15

-10

-5

0

5

Frequency, GHz

|S11

|

(a)

0 2 4 6 8 10 12-20

-15

-10

-5

0

5

Frequency, GHz

|S11

|

Raw dataSmooth Peak

Figure 3-11 Magnitude of S11 for the planar half-disk UWB monopole antenna

of Fig. 3-10 measured: (a) in free space and (b) inside the spherical

cap of 15-cm radius.

59

1 2 3 4 5 6 7 8 9 10 11 120

0.2

0.4

0.6

0.8

1

Frequency, GHz

Tota

l Effi

cien

cy η

T

MeasuredSimulated (FDTD)

Figure 3-12 Total efficiency ηT (radiation efficiency including impedance

mismatch) of the planar half-disk UWB monopole antenna of Fig.

3-10 evaluated using (3.13).

3.4.3 Measurement Sensitivity Tests

In the Wheeler cap method, it was assumed that small antennas are placed at the

center of the spherical cap during testing. The phase center of moderate-size antennas

should also be located at the center of the cap. However, it is difficult not only to find the

phase center of complex antenna structures but also to place the antenna at the correct

location during measurement setup. Therefore, the following question arises: what is the

effect on the measurements if the antenna under test is not located exactly at the center of

the enclosing sphere? For this investigation, the planar half-disk monopole antenna is

displaced away off the center of the sphere, as shown in Fig. 3-13. The total efficiency

for this setup compared to that when the antenna is very near the center of the spherical

cap shown previously in Fig. 3-12 is illustrated in Figure 3-14. The results show that the

effect of antenna displacement from the cap center is negligible. Thus, during Wheeler

cap measurements, the test antenna does not have to be placed exactly at the center of the

cap.

60

Figure 3-13 Measurement setup for the planar half-disk UWB monopole

antenna placed away from the center of the spherical Wheeler cap.

1 2 3 4 5 6 7 8 9 10 11 120

0.2

0.4

0.6

0.8

1

Frequency, GHz

Effi

cien

cy

About the cap centerAway from the cap center

Figure 3-14 Comparison of the antenna total efficiency for test antenna locations

near and away (Fig. 3-13) from the center of the spherical Wheeler

cap.

61

3.5 Summary

The Wheeler cap method was described and associated quantities were derived to

evaluate radiation efficiency of small antennas. This traditional method is popular

because of its simplicity for measuring radiation efficiency. However, prior

understanding has limited its use to small narrowband antennas. Indeed, the conventional

Wheeler cap method can only be used for simple antennas that can be modeled as a series

or parallel RLC circuit. Furthermore, the method are only accurate near antenna

resonance where antenna input reactance can be assumed negligible. In this work, the

method was extended for the efficiency measurement of antennas electrically larger in

size and antennas with wider impedance bandwidth. In this method, there is no

assumption on the antenna model complexity and measurement can be performed at any

frequencies other than at resonance. Tests performed for a lossy monopole antenna and

an UWB antenna demonstrated that experimental results are comparable to the numerical

values and yet, the measurement setup remains simple and the efficiency evaluation is

still fast.

3.6 References

[1] W.L Stutzman and G.A. Thiele, Antenna Theory and Design, 2nd Ed., Wiley:

New-York, 1998.

[2] Glenn S. Smith, “An Analysis of the Wheeler Method for Measuring the

Radiating Efficiency of Antennas,” IEEE Trans. Antennas. and Propagation, pp.

552-556, July 1997.

[3] H.A. Wheeler, “The Radian Sphere Around a Small Antenna,” Proc. IRE, Vol.

47, pp. 1325-1331, August 1959.

[4] W.E. McKinzie III, “A Modified Cap Method for Measuring Antenna

Efficiency,” IEEE Antennas and Propagation Society International Symposium,

Vol. 1, July 1997, pp. 542 – 545.

62

[5] E.H. Newman, P. Bohley, and C.H. Walter, “Two Methods for the Measurement

of Antenna Efficiency,” IEEE Trans. Antennas and Propagation, Vol. AP-23, pp.

457-461, July 1975.

[6] D.M. Pozar and B. Kauffman, “Comparison of Three Methods for the

Measurements of Printed Antenna Efficiency,” IEEE Trans. Antennas and

Propagation, Vol. 36, No. 1, pp. 136-139, January 1988.

[7] Y. Huang, R.M. Narayanan, and G.R. Kadambi, “On Wheeler’s Method for

Efficiency Measurement of Small Antennas”, IEEE Trans. Antennas and

Propogation, pp. 346-349, August 2001.

[8] Test & Measurement Application Note 95-1 S-Parameter Techniques, Hewlett-

Packard, 1997. Available at http://www.sss-mag.com/pdf/an-95-1.pdf.

[9] H.G. Schantz, “Radiation efficiency of UWB Antennas,” Ultra Wideband Systems

and Technologies, 2002, Digest of Papers. 2002 IEEE Conference on, 21-23 May

2002.

[10] T. Yang and W.A. Davis, “Planar Half-Disk Antenna Structures for Ultra-

Wideband Communications,” IEEE Antennas and Propagation Symposium,

Monterey, California, June 2004.

63

Chapter 4

A Review of Radiation Effects on Human Operators of Hand-Held Radios

4.1 Biological Effects of Radio-Frequency Radiation

There has been a dramatic increase in the worldwide use of radio-frequency (RF)

technologies in the last two decades, especially in personal communications. In fact,

since the introduction of commercial systems in 1983, cellular telephone service has been

one of the fastest growing segments of the U.S. telecommunications industry. By the end

of 1984, cellular service had nearly 100,000 users. By the end of 1991, it had climbed to

over two million users [1]. Today, there are over 100 million hand-held cellphone users

in the U.S. Because of this dramatic increase in the use of RF devices, there is an

increasing concern over the biological effects of RF radiation from personal wireless

communication devices and it is prudent to have on-going studies of health effects. The

large number of cell-phone users would lead to major health problem if there was even a

small hazard rate.

This chapter presents an overview of the biological effects of RF radiation,

including safety issues and regulations. The interaction between handset antennas and

biological tissues, quantified by specific absorption rate (SAR), is explained. Discussion

of the issue of health hazards associated with RF exposure is also presented, but the

authors do not speculate whether or not low-level RF radiation is harmful to humans.

Even though it is widely believed that use of mobile phones is safe, there are still many

64

scientific studies and laboratory investigations being conducted that address potential

health hazards of RF radiation from mobile phones [2, 3].

4.2 RF Radiation Basics

Electromagnetic waves are distinguished by their wavelength (λ) and frequency f (f=

speed of light/λ). The RF portion of the electromagnetic spectrum is generally defined as

that part of the spectrum where electromagnetic waves have frequencies in the range of 3

kHz to 300 GHz, as illustrated in Fig. 4-1. The microwave portion (several hundred MHz

to several GHz) of the spectrum is popular for commercial communication devices.

Electromagnetic waves can be divided into two categories: ionizing and non-

ionizing radiation, as indicated in Fig. 4-1. “Ionization” is the process by which electrons

are stripped from atoms and molecules. This process can produce molecular changes that

damage biological tissue, including effects on DNA. This process requires high levels of

electromagnetic energy. X-rays and Gamma-rays often have an energy level sufficiently

high enough to ionize biological material. The energy levels associated with RF and

microwave radiation, on the other hand, are rarely great enough to cause the ionization of

atoms and are classified as non-ionizing radiation. Often the term “radiation” is applied to

ionizing radiation such as that associated with nuclear power plants. Ionizing radiation

should not be confused with lower-energy, non-ionizing radiation with respect to possible

biological effects, since the mechanisms are quite different [4]. Non-ionizing radiation is

discussed in this article.

65

Frequency(Hertz)

IONIZINGNON-IONIZING

Radio Frequencies

MicrowavesRadio andTelevision

105 1011 1017

IR UV X-rays Gamma RaysPower Lines

106 107 108 109 1010

Mobile Radio

Visible Light

Frequency(Hertz)

IONIZINGNON-IONIZING

Radio Frequencies

MicrowavesRadio andTelevision

105 1011 1017

IR UV X-rays Gamma RaysPower Lines

106 107 108 109 1010

Mobile Radio

Visible Light

Figure 4-1 The electromagnetic spectrum.

4.3 Biological Effects and Health Issues

The mechanisms of non-ionizing electromagnetic radiation interaction with biological

systems is grouped into two major types: “thermal effects” and “non-thermal (or

athermal) effects”, depending on whether they are attributable to deposition of heat

(thermal) or to a direct interaction of the EM field with the tissue substance without a

significant heat component (non-thermal).

Thermal effects can occur in tissues as a result of EM field absorption in the

dissipative tissue media. Radio frequencies cause water molecules and dissolved ions to

vibrate, causing absorption. Water content is then an important parameter in determining

the dielectric properties of biological tissues. An equivalent body tissue conductivity σ is

related to the imaginary part of the permittivity ε″ as:

"2 επσ f= [S/m] (4.1)

where f is the frequency of operation. Particularly in the microwave band, tissues having

high-water content show a conductivity that increases with frequency. The absorbed

power per unit volume is proportional to the electric field in the material and its

conductivity:

66

2

21 Eσ=aP [W/m3] (4.2)

where E is the amplitude of the electric field strength.

The RF energy actually absorbed in tissues is quantified using “specific absorption

rate” or SAR, the absorbed power per unit mass of tissue. SAR is the primary parameter

used when discussing the health risk due to electromagnetic power absorption in the body

and is defined as

2

2Ε

ρσ

ρ== aP

SAR [W/kg] (4.3)

where ρ is the material density in kg/m3.

Non-thermal effects, on the other hand, are not caused by heat, but are due to direct

interaction with the RF field on molecules and tissue components; the particles tend to

orient themselves along the electric fields such that potential energy is minimized [5].

Non-thermal effects are not very well understood and their human health consequences

are still under investigation.

A biological effect occurs when a biological change is measured in response to a

stimulus. However, the observation of a biological effect does not necessarily suggest the

existence of a biological hazard. A biological effect only becomes a safety hazard when it

causes a detectable impairment of individual’s health or offspring [4]. There are many

published reports in the scientific literature concerning possible biological effects

resulting from RF energy exposure. These studies can be categorized into three types: cell

studies, animal studies, and epidemiological studies. Cell studies focus on single cells,

multiple cells, or organs that are exposed to RF energy in a highly controlled exposure

environment. Animal studies observe health changes or chemical effects in animals

exposed to RF energy. Epidemiological studies examine a segment of the human

population for relationships between health effects and RF exposure. Cellular studies can

produce accurate results, but it is difficult to extrapolate conclusions from cellular

response to effects on animals. Animal and epidemiological studies are subject to

questions about the controls during the study and the extraction of specific cause-effect

relationships.

67

Thermal effects from RF radiation have been studied extensively in cells and

animals and it has been known for many years that exposure to very high levels of RF

radiation can be harmful due to the ability of RF energy to rapidly heat biological tissue.

In fact, microwave ovens cook food based on RF thermal effects. Exposure to very high

RF field intensity produces heating in biological tissue and can increase body

temperature. Fortunately, the intensity of radiation from hand-held devices is too low to

cause such effects.

Non-thermal effects due to low RF radiation levels have also been investigated over

the past two decades [6]. However, evidence of harmful biological effects is ambiguous

because results from these studies are often inconsistent or not replicated [6].

Nonetheless, studies are ongoing and key government agencies and industry

organizations continue to monitor the results of the latest scientific research on this topic

for using in and setting safety guidelines for RF exposure.

4.4 RF Safety and Regulations

There are two parameters commonly used in RF radiation regulations: SAR and

maximum permissible exposure (MPE). As discussed in the previous section, SAR

relates to the absorption of RF radiation in biological tissues. MPE is the upper limit of

RF-radiation power density (mW/cm2) exposure for biological tissues.

RF exposure standards have been developed by various organizations and countries.

These standards recommend safe levels of exposure for both the general public and

workers. The recommended safe levels have been revised downward several times in

recent years, but not all scientific bodies agree on safe levels. In the United States, the

Federal Communications Commissions (FCC) has adopted and used recognized safety

guidelines for evaluating RF environmental exposure since 1985 [4].

Exposure standards are largely based on the thermal effects and are derived from

absorption of plane waves by man or laboratory animals [4]. Standards usually

recommend RF exposure limits for two cases: “controlled environment” and

“uncontrolled environment”. For the “controlled environment” case, energy levels are

68

known and everyone in the exposure area is aware of the presence of electromagnetic

fields (EMF). Also, personnel are expected to have received instruction in radiation

safety and to have suitable approval protective clothing, or there is limited occupancy, in

terms of time and distance, of the area. For “uncontrolled environment” (or “general

population”), energy levels are not known or some people present may not be aware that

they are exposed to EMF fields, including special risk groups, such as young children and

pregnant women. The duration of exposure can also be substantially different in both

groups.

The most popular recommendations are the ANSI/IEEE [7] and the

ICNIRP/CENELEC [8] guidelines. ANSI/IEEE C95.1-1992 guidelines [7] for

recommended EMF exposure limits went into effect in 1992 and were adopted by the

FCC with some revisions. The guidelines replaced a 1982 ANSI guideline that permitted

somewhat higher exposure levels. ANSI-recommended limits before 1982 were even

higher.

For uncontrolled environments, the ANSI/IEEE C95.1-1992 standard suggests that:

(a) the power density (W/m2) in the microwave frequency band, from 300 MHz to 15

GHz, should remain under f/1500, where f is the frequency of operation in MHz; (b) the

SAR averaged over the whole body for 30 minutes or more should remain under 0.08

mW/g; and (c) the SAR averaged over any 1 gram of tissue for 30 minutes or more

should remain under 1.6 mW/g. These recommended levels of exposure would raise the

tissue temperature by no more than 1°C. However, the temperature increase rarely

reaches 1°C, due to thermoregulation in the human body, and does not present an

unacceptable thermal load, avoiding adverse effects on the functioning of the human

body. The standard was not developed to protect against possible hazards from long-term

exposure to low-level RF radiation because such hazards are not well understood.

Figure 4-2 shows the maximum permissible exposure in terms of power density

incident on a human for an uncontrolled environment in the frequency range from 100

MHz to 300 GHz recommended by ANSI/IEEE standard and ICNIRP/CENELEC

European standard. Measurements, computational analysis using models of the human

head, and other studies of SAR distribution for hand-held cellular and PCS phones have

69

shown that, in general, the 1.6 mW/g limit is unlikely to be exceeded under normal

conditions of phone use [9].

101 102 103 104 1053x102 2x103 1.5x104

0.1

1

10

Max

imum

Per

mis

sibl

e E

xpos

ure

(MPE

), m

W/c

m2

Frequency (f), MHz

10 mW/cm2

1 mW/cm2

f/1500

f/20000.2 mW/cm2

ANSI/IEEEICNIRP/CELENEC

101 102 103 104 1053x102 2x103 1.5x104

0.1

1

10

Max

imum

Per

mis

sibl

e E

xpos

ure

(MPE

), m

W/c

m2

Frequency (f), MHz

10 mW/cm2

1 mW/cm2

f/1500

f/20000.2 mW/cm2

ANSI/IEEEICNIRP/CELENEC

Figure 4-2 ANSI/IEEE and ICNIRP/CELENEC maximum permissible

exposure in terms of power density for uncontrolled environment in

RF frequency range [5].

4.5 Human Operator Influence on Hand-Held Radio Performance

There are two reasons to understand the effects of electromagnetic fields radiated from

mobile phones in the presence of the user’s head and hand. The first reason is related to

antenna/handset design because the presence of a user influences the radiation

characteristics of a mobile phone. The presence of the hand and head can be taken into

account in an early stage of mobile phone design. The second reason to understand

interaction effects is for compliance with safety standards by examining dosimetry

aspects to evaluate the SAR inside the head and inside the hand, which are then compared

to permitted levels. Interaction studies can be performed both by computation and by

measurements. There are various computational methods to perform numerical studies.

Finite difference time domain (FDTD) computational method is often used for such a

70

numerical evaluation process and its implementation is briefly discussed in the next

section.

4.5.1 Computational Simulation

The FDTD numerical method (Finite Difference Time Domain) of computational

simulation is very well suited for evaluating large and inhomogeneously-filled

computational domains, and is widely used for analyzing the coupling of mobile

communication equipment and the human head that are in close proximity. FDTD is

based directly on the differential form of time domain Maxwell’s equations, and material

parameters ε, µ, and σ which are the space-dependent permittivity, permeability, and

conductivity, respectively. The entire problem space is gridded in both spatial and

temporal dimensions, as shown in Fig. 4-3. The details of this process were originally

formulated in the 1960s by Yee [10] and are summarized in [11, chap. 9]. Once the time-

domain computation using this method is completed, a Fourier transform is performed to

obtain the desired frequency-domain quantities such as input impedance, radiation

pattern, and gain.

(i,j,k)

EX

EX

EXEX

EY

EY

EY

EY

EZEZ

EZEZ

HX

HX

HYHY

HZ

HZ

x

y

z

(i,j,k)

EX

EX

EXEX

EY

EY

EY

EY

EZEZ

EZEZ

HX

HX

HYHY

HZ

HZ

x

y

z

Figure 4-3 Field component value locations of a Yee cell used in FDTD

computations. The E-components are in the middle of the edges and

the H-components are in the center of the faces [10].

71

4.5.2 Electromagnetic Modeling of the Human Operator

Specific absorption rate (SAR) is measured directly as a temperature increase in a

localized area of tissue, as quantified later in Equation (4-4). Measuring SAR directly is

nearly impossible in practice because it is necessary to insert calorimetric probes into a

live human head tissue. As a result, head models with mathematical simulations (or

measurement using physical head models) are used for estimating SAR.

Over the past ten years, there have been many studies that model biological tissues

such as the human hand and head [12-26]. Simple models of human tissues were realized

first. For instance, the human hand was modeled as a layer of bone surrounded by a layer

of muscle. The human head was represented by a rectangular box or a sphere with

homogeneous material, as shown in Fig. 4-4a. With the advances in computational

methods and in computation speed, the human body models became more complex and

accurate by including more types of tissue layers, as shown in Fig 4-4b. With the help of

the magnetic resonance imaging, very realistic human hand and head models were finally

obtained, as shown in Fig. 4-4c. Table 4-1 presents values of relative permittivity,

conductivity, and density of tissues in the hand and head near 900 MHz [12].

Table 4-1 Relative Permittivity, Conductivity, and Mass Density of the Tissues

in the Hand and Head near 900 MHz [11].

Tissue Permittivity Conductivity (S/m) Density (g/cm3)

Bone 8.0 0.105 1.85

Skin/Fat 34.5 0.60 1.10

Muscle 58.5 1.21 1.04

Brain 55.0 1.23 1.03

Humour 73.0 1.97 1.01

Lens 44.5 0.80 1.05

Cornea 52.0 1.85 1.02

As the degree of the biological tissue model complexity increases, the computer

performance requirement also increases. It is, thus, important to employ only the level of

72

modeling complexity that is necessary for accurate results. Okoniewski and Stuchly [16]

compared various simple models with complex ones from Yale School of Medicine and

University of Gent in Belgium. Table 4-2 summarizes their computed values of antenna

efficiency, η, which is the ratio of the power radiated to the power input to the antenna,

and power absorbed, Pabs, in the head for a handset with a monopole antenna located 1.5

cm from the head. Two issues were investigated in the paper: the utility of the simplified

“canonical” models and the significance of a more realistic model of the head. The

important conclusion that can be drawn from the Table 4-2 is that spherical models

provide estimates of the antenna efficiency and total absorbed power in the head that are

in reasonably good agreement with non-homogeneous complex model predictions.

However, the very simple box model is inadequate.

Table 4-2 Comparison of Computed Antenna Efficiency and Power Absorbed in

the Head for Simple (Rectangular and Spherical Shape) and Complex

Models Developed from Yale University and from Gent University

(cf. Figure 4). Transmit Power is 1 W at 915 MHz; Distance of

Monopole Antenna on a Handset from Head is 1.5 cm [15].

Head Model η (%) Pabs (W)

Homogeneous 16 0.84

Skull-Brain 24 0.76 Box

Skin-Skull-Brain 19 0.81

Homogeneous 46 0.54

Skull-Brain 52 0.48 Sphere

Skin-Skull-Brain 45 0.55

Gent Head 51 0.49

Yale Head – with ear (5-mm resolution) 60 0.40

Yale Head – no ear (5-mm resolution) 50 0.50 Anatomical

Yale Head – with ear (3.4-mm resolution) 53 0.47

73

(a) Simple Models

(b) Multi-Layered Model [12]

(c) Complex model based on MRI data

Box Model Sphere Model

Yale Head Model(http://noodle.med.yale.edu/phant.html)

(a) Simple Models

(b) Multi-Layered Model [12]

(c) Complex model based on MRI data

Box Model Sphere Model

Yale Head Model(http://noodle.med.yale.edu/phant.html)

Figure 4-4 Popular human head models used in numerical simulations.

74

In the past decade, laboratory human head body models, called “phantoms”, have

been developed for experimental investigations. Again, there are various degrees of

complexity in the phantoms, ranging from a simple geometric shape to realistic human

body shapes where the interior is filled with one or more liquids with dielectric properties

equivalent to those of human body at the frequencies under consideration. With the help

of phantom models, experimental determination of SAR is possible.

4.6 Human Operator Effects on Antenna Characteristics

The data shown in Table 4-2 indicates that the efficiency of a monopole antenna located

1.5 cm from the head at 900 MHz is roughly 50 %. This means that the human head

absorbs about half of the power radiated from the monopole antenna. Table 4-3 illustrates

additional results for the power absorbed in the head as function of the separation

distance between the head and the antenna, giving the expected result that efficiency

increases with distance from the head and, thus, power absorbed in the head decreases.

Table 4-3 Effects of the Change in the Separation Between the Antenna and the

Head Model from University of Gent, Belgium; 915 MHz, 1W [15].

SAR in the head (W/kg)

Separation

(cm) η (%) Pabs (W)

Peak 1 g 10 g

1.5 51 0.49 11.2 8.6 4.8

2.0 57 0.43 7.1 5.6 3.3 Gent

Model 3.0 72 0.28 3.2 2.4 1.4

The effects of power absorption on the far-field radiation pattern of the antenna are

shown in Fig. 4-5. The patterns were obtained from simulations of a monopole on a

rectangular box model for the handset located 1.5 cm from the head for three levels of

model complexity using FDTD method (Fidelity from Zeland) and are similar to those

reported in [16]. Figure 4-5 demonstrates that the omni-directional pattern is radically

altered by the presence of the human head. The pattern for the sphere model case is very

75

similar to the one for the Yale anatomical head model in [15], which is considered to be

the most accurate model. It is not surprising that the box model results are less accurate

than the sphere model because the box representing the head nearly entirely blocks the

radiation in the head direction. In the case of a sphere, there is less dramatic but

significant reduction in power radiated towards the half-space in the direction of the head

(φ = 0°). The pattern for the spherical head model in Fig. 4-5 shows that radiation is

reduced by more than 10 dB over a wide angular region in the horizontal plane.

No headBox-model head

Sphere-model head

Directionof head

No headBox-model head

Sphere-model head

Directionof head

Figure 4-5 Computed radiation patterns in φ-plane (θ = 90°) for a monopole

oriented along the z-axis and mounted on a metal box. Operation is

at 900 MHz and the antenna is 1.5 cm away from a human head

modeled as a rectangular box of 20-cm side length and as a sphere

of 20-cm diameter with three layers of dielectric material (skin-

skull-brain); see Figs 4a and b.

The influence of the human operator on input impedance must also be considered in

antenna design because overall efficiency reduces with increasing impedance mismatch.

76

In fact, with a large impedance mismatch, only a small amount of power from the source

is accepted by the antenna when transmitting, and only a small portion of the power is

transferred from the antenna when receiving. The effects of biological tissues on the input

impedance for two antenna types and configurations are shown in Figs. 4-6 to 4-8.

Results of these plots are similar to those reported in [12]. In contrast to human operator

influence on the radiation pattern of a monopole, there is relatively little influence on the

antenna input impedance as illustrated in Fig. 4-6, which shows the computed return loss

of a monopole with and without biological tissues in proximity. This is a desirable

characteristic of the monopole, which partly accounts for its widespread use. For

conformal and internal antennas, such as the PIFA (planar inverted-F antenna), dramatic

detuning can occur due to hand and/or head presence. An example of such an effect can

be seen for the PIFA antenna shown in Fig. 4-7. Figure 4-8 shows the return loss for the

PIFA antenna for several operator configurations. A return loss of 10 dB (approximately

equivalent to VSWR = 2) or more is the commonly accepted limit for impedance

mismatch. The plot compares the results with no biological tissue included to those when

a hand is placed at three vertical locations on the handset. As can be seen from the figure,

the hand detunes the antenna operating frequency and introduces impedance mismatch. A

high impedance mismatch occurs when the hand begins to mask the antenna. These

results show the importance of minimizing antenna masking through proper antenna

placement on the handset.

77

Frequency (f), GHz

Ret

urn

Loss

(RL)

, dB

Frequency (f), GHz

Ret

urn

Loss

(RL)

, dB

Frequency (f), GHz

Ret

urn

Loss

(RL)

, dB

Figure 4-6 Computed return loss values of a monopole on a handset for three

cases: no human operator (“monopole only”); only a hand present

(“with hand”); and both an operator hand and head present (“with

hand and head”). The separation distance between 20-cm diameter

spherical head model (skin-skull-brain) and the antenna is 1.5 cm

and the hand model (muscle-bone) is located 6.0 cm below the

antenna.

4.0

5.0

50.020.0

75.0

60.0

Plastic Casing

Feed

Metallic boxShort Wire

145.0

Hand Model(muscle-bone)Dimensions in mm

4.0

5.0

50.020.0

75.0

60.0

Plastic Casing

Feed

Metallic boxShort Wire

145.0

Hand Model(muscle-bone)Dimensions in mm

Figure 4-7 Handset model showing the PIFA antenna location and hand model

for the results in Fig. 4-8.

78

d

Frequency (f), GHz

Ret

urn

Loss

(RL)

, dB d d

Frequency (f), GHz

Ret

urn

Loss

(RL)

, dB

Figure 4-8 Computed return loss values of the side-mounted PIFA on a handset

shown in Fig. 7 without the hand and with the hand for three

different hand locations d.

4.7 Power Absorption in the Head and SAR

As previously mentioned, radiation exposure standards are based on the thermal effects

of RF radiation, specifically the RF power absorbed in the head. SAR, defined in (3), is

expressed in units of Watts per Kilogram (W/kg) and is the fundamental parameter used

to examine health risks of electromagnetic power absorption in the head. SAR is a point

quantity that varies with location and, thus, is used to both locate and quantify regions of

high absorption in the human operator. The accuracy and reliability of a given SAR value

depends on the accuracy of three key parameters: tissue density, tissue conductivity, and

the electric field level present, the most significant of which is the induced electric field.

RF-power absorption in biological tissue causes the tissue’s kinetic energy to

increase with exposure duration. If the incident power density is sufficiently high, the

79

absorbed RF energy will produce a temperature increase. The rate of increase of

temperature is proportional to the SAR value found using the formula

tTcSAR

∆∆= [W/kg] (4.4)

where c is the specific heat of the tissue (J/kg per °C), ∆T is the transient temperature rise

(°C), and ∆t is the duration (s) of power application used for the linear portion of the

temperature rise. At the start of exposure, temperature rise is linear with time [27].

If the whole body is exposed to the radiation, the average SAR is defined as the

ratio between the total power absorbed and the mass of the whole body. When a part of

the body is exposed to radiation, the SAR is also evaluated for a reference mass (for

example, 10 g). A higher SAR level can be tolerated locally, provided that the allowed

average SAR is not exceeded. Popular standards of electromagnetic radiation exposure

limits for the general public are summarized in Table 4-4. Allowed SAR values vary with

the mass of tissue used in the standard (1 g for IEEE standards, and 10 g for CENELEC

and ICNIRP). The distribution of absorbed microwave energy varies greatly from point

to point inside the user’s head. Therefore, a larger averaging mass (such as 10 g) smooths

the computed or measured SAR distribution, tending to lower the averaged value by a

factor of two or three for the same exposure over a smaller mass (such as 1g) [28]. For

example, a 10-g SAR of 2 W/kg (mW/g) is equivalent to a 1-g SAR of 4 to 6 W/kg

(mW/g). In other words, the absorbed power averaged over a tissue mass of 10 g is low

compared to a 1-g SAR value. The 1-g SAR quantity is a more precise spatial measure of

localized microwave-power absorption, providing a detailed distribution of power

absorbed inside the head. The SAR values for mobile phone human operator exposure to

microwaves vary with simulation model sophistication and depend on individual operator

habits.

80

Table 4-4 Popular Standards Limits on Exposure to the General Public of

Electromagnetic Radiation in the RF.

ICNIRP

“Health Physics”, 4/1998

[8]

CENELEC

ENV 50166-2: 1995

[8]

ANSI/IEEE

C95.1-1991 [7]

Region of

application International Europe US

Frequency

range 100 kHz – 10 GHz 10 kHz – 300 GHz

100 kHz – 6

GHz

Average SAR

(whole body) 0.08 W/kg 0.08 W/kg 0.08 W/kg

Local SAR

Average mass

2 W/kg

10 g (contiguous tissue)

2 W/kg

10 g (cube)

1.6 W/kg

1 g (cube)

Most cellular telephones sold in North America are in compliance with FCC rules

and regulations, but at present, there does not exist a standard testing protocol for

measuring SARs for mobile phones. Some variations in measured SAR values were as

large as 100% for the same telephone [27]. In addition to the phantom head model used,

the measurement technique, and the instrumentation, SAR values also depend on the

orientation of the handset, its tilt angle from vertical, the geometry of the ear and the

handset antenna to head separation distance.

4.8 Summary

With the progress in computational electromagnetics and the advancement in biological

research, one can expect more accurate and conclusive results for biological effects on

human tissues due to low-level microwave radiation from portable wireless devices.

Unlike pollution of the environment where there is a tradeoff between public health and

commercial development, the goal for wireless communications is to improve

performance by reducing the power absorbed in the user’s body. This increases the signal

81

to distant points for the same transmitter power. Thus, both communication performance

and health hazard conditions are improved.

The study of microwave exposure requires collaboration between engineers and

medical professionals. Understanding how operator presence affects antenna performance

and the associated health hazards aids engineers to develop handheld wireless products

that minimize operator exposure and improve antenna radiation efficiency. Also,

biologists can have a better understanding of low-level electromagnetic radiation in

human tissues with the help of more accurate electromagnetic propagation models in

complex inhomogeneous media.

There is no widely accepted relationship between microwave exposure and health

effects. The preponderance of evidence suggests that there is no health hazard due to

long-term low-level exposure to handset radiation [21]. However, due to the

pervasiveness of cellphone use, it is prudent to continue investigating potential health

problems.

4.9 References

[1] U.S. Government Accounting Office, “Telecommunications: Concerns About

Competition in the Cellular Telephony Industry,” GAO/CED-92-220, July 1992.

[2] J.C. Lin, “Cell Phone Testing and Fundamental scientific Research,” IEEE Ant.

Propag. Magazine, Vol. 43, No. 4, pp. 156-158, Aug. 2001.

[3] J.C. Lin, “Cellular Telephone Radiation and Electroencephalograms (EEG),”

IEEE Ant. and Propag. Magazine, Vol. 45, No. 5, pp. 150-153, October 2003.

[4] R.F. Cleveland Jr. and J. Ulcek, “Questions and Answer about Biological Effects

and Potential Hazards of Radiofrequency Electromagnetic Fields”, OET Bulletin

56, 4th ed., FCC/OET, August 1999.

[5] A. de Salles, “Biological Effects of Microwave and RF”, SBMO/IEEE MMT-S

IMOC’99 Proceedings, pp. 51-56, April 1999.

82

[6] R. Goldberg, “Literature Resources for Understanding Biological Effects of

Electromagnetic Fields,” EMF-Link Multimedia Resource, Available:

http://infoventures.com/emf/top/lit-rev.html.

[7] American National Standards Institute (ANSI), “IEEE C95.1-1992: IEEE

Standard for Safety Levels with Respect to Human Exposure to Radio Frequency

Electromagnetic Fields, 3 kHz to 300 GHz,” The Institute of Electrical and

Electronics Engineers, Inc, 1992.

[8] ICNIRP Guidelines, “Guidelines for limiting Exposure to Time-Varying Electric,

Magnetic, and Electromagnetic Fields (up to 300 GHz),” International

Commission on Non-Ionizing Radiation Protection, “Health Physics,” Vol. 74,

No. 4, pp. 494-522, April 1998.

[9] FCC/OET, “Information on Human Exposure to Radiofrequency Fields from

Cellular and PCS Radio Transmitters”, January 1998

[10] K. S. Yee, "Numerical solution of inital boundary value problems involving

maxwell's equations in isotropic media," IEEE Trans. Antennas Propagat., Vol.

14, pp. 302 - 307, May 1966.

[11] W.L. Stutzman and G.A. Thiele, Antenna Theory and Design, 2nd edition, John

Wiley, 1998

[12] M. A. Jensen and Y. Rahmat-Samii, “EM Interaction of Handset Antennas and a

Human in Personal Communications,” Proceedings of the IEEE, Vol. 83, No. 1,

pp. 7-17, Jan. 1995.

[13] O. P. Gandhi, J. Y. Chen, D. Wu, “Electromagnetic absorption in the human head

for mobile telephones at 835 and 1900 MHz,” Proc. Int. Symp. Electromagn.

Compat., Rome, Italy, Vol. I, pp. 1 - 5, Sept. 1994.

[14] J. Toftgard, S.N. Hornsleth, and J.B. Andersen, “Effects on Portable antennas in

the Presence o a person,” IEEE Trans. On Ant. Propag., Vol. 41, No.6, pp. 739-

746, 1993.

[15] S. Watanabe, M. Taki, T. Nojima, O. Fujiwara, “Characteristics of the SAR

Distribution in the Head Exposed to Electromagnetic Field Radiated by a Hand-

Held Portable Radio,” IEEE Trans. On Microwave Theory and Techniques, Vol.

44, No. 10, pp. 1874-1883, Oct. 1996.

83

[16] M. Okoniewski and M.A. Stuchly, “A study of the handset antenna and human

body interaction,” IEEE Trans. on Microwave Theory and Techniques, Vol. 44,

No. 10, pp. 1855-1874, Oct. 1996.

[17] Le-Wei Li et al., “FDTD analysis of electromagnetic interactions between handset

antennas and the human head,” Microwave Conference Proceedings, APMC '97,

1997 Asia-Pacific, Vol. 3, pp. 1189 –1192, 2-5 Dec 1997.

[18] C.W. Trueman, “Validation of FDTD handset and head patterns by

measurement,” Antennas and Propagation for Wireless Communications, IEEE-

APS Conference, pp. 93 –96, 1-4 Nov 1998.

[19] K. Ito, Y. Okano, A. Hase, I. Ida, “A tissue-equivalent solid phantom for

estimation of interaction between human head and handset antenna,” Proceedings

of IEEE APS conference on Antennas and Propagation for Wireless

Communications, IEEE-APS Conference, pp. 89 –92, 1-4 Nov 1998.

[20] N. Kuster, “Radiation performance and evaluation of human exposure from

mobile handsets using near-field measurements,” Proceedings of Intern. Symp. on

Electromagnetic Compatibility, pp. 480-483, May 1999.

[21] K.R. Foster and J.E. Moulder, “Are mobile phones safe?,” IEEE Spectrum, Vol.

37, No. 8 , pp. 23-28, August 2000.

[22] K. Nikita et al., “A study of uncertainties in modeling the handset antenna and

human head interaction using the FDTD method,” IEEE MTT Soc. Intern.

Symposium Digest, Vol. 2, pp. 1025 –1028, Dec. 2000.

[23] A. Drossos, V.Santomaa, N. Kuster, “The dependence of electromagnetic energy

absorption upon human head tissue composition in the frequency range of 300-

3000 MHz,” IEEE Trans. on MTT, Vol. 48, No. 11, pp. 1988-1995, Nov. 2000.

[24] J. Moustafa, “Investigations of reduced SAR personal communications handset

using FDTD,” Proceedings of IEE Eleventh International Conference on

Antennas and Propagation, 2001, Vol. 1, pp. 11 –15, April 2001.

[25] M. Francavilla, A. Schiavoni, and G. Richiardi, “Effect of the hand on cellular

phone radiation,” IEE Proc. on Microw. Ant. Propag., Vol. 148, No. 4, pp. 247-

253, Aug. 2001.

84

[26] N. Chavannes et al, “Suitability of FDTD-based TCAD tools for RF design of

mobile phones,” IEEE Ant. Propag. Magazine, Vol. 45, No. 6, pp. 52-66, Dec.

2003.

[27] J.C. Lin, “Specific absorption rates (SARs) induced in head tissues by microwave

radiation from cell phones,” IEEE Antennas and Propagation Magazine, Vol. 42,

No. 5, pp. 138 –139, Oct. 2000.

[28] P. Bernardi, M. Cavagnaro, S. Pisa, E. Piuzzi, “Specific absorption rate and

temperature increases in the head of a cellular-phone user,” IEEE Trans. on MTT,

Vol. 48, No. 7, pp. 1118-1126, July 2000.

85

Chapter 5

A New Wideband Compact Antenna

5.1 Introduction

This chapter introduces a new wideband compact antenna along with results of a

preliminary investigation into its performance characteristics. The detailed analysis

includes a parametric study of the antenna. The proposed antenna structure has more than

one resonating element, providing up to 50 % impedance bandwidth. Its radiation

mechanism relies on closely spaced resonant frequencies to form a wider impedance

bandwidth, as discussed in Chapter 2. However, the coupling effects between radiating

elements are not fully understood. Understanding how these effects play a role in each of

the resonating elements in this multilayer structure not only can give deeper insights of

the antenna behavior, but also help improve its performance and ability to design it more

effectively and efficiently.

The analysis of the proposed wideband compact antenna begins by looking at its

radiation characteristics and current distribution in order to recognize some functional

similarities with other known antenna types. Parametric study results follow and provide

insight into the effects of each antenna structure geometric parameter on the impedance

performance. After the parameter effects are explained, a design model for optimal

impedance bandwidth is presented.

86

5.2 The Antenna Structure

The geometry of the proposed antenna is shown in Fig. 5-1 and is called the wideband

compact J-pole (WC J-pole) antenna. It is an extension, albeit a significant one, of the

conventional planar inverted-F antenna (PIFA) shown in Figure 5-2 [1]. The name WC J-

Pole is used due to its structural similarities to J-pole antennas; this will be discussed

further in Section 5.3. The two major differences between the WC J-pole and the PIFA

are the use of a capacitive feed instead of the conventional conducting probe feed and the

use of a small ground plane that participates in the radiation mechanism. The contacting

probe feed of the conventional PIFA limits its bandwidth to about 10 % and the PIFA’s

large ground plane makes it a moderately large structure.

The non-contacting feed of the WC J-pole is constructed by terminating the inner

conductor of a SMA connector to a conducting plate parallel to the upper plate. A small

shorting plate is placed in the center of the left edge of the upper plate (in x=0 plane) and

is attached to the center of the left edge of the ground plate, as shown in Fig. 5-1. The

upper plate, the feed plate, and the ground plate have the same width, W. The difference

between this design and the conventional PIFA is that the ground plate is part of the

whole antenna structure and participates in the antenna radiation mechanism. Table 5-1

summarizes the geometric parameters of the WC J-pole, including values referenced to

the wavelength λc of the WC J-pole shown in Fig. 5-3 where λc is the wavelength for

center operating band frequency (fc = 2.175 GHz).

87

Table 5-1 Geometric parameters of the WC J-pole antenna of Fig. 5-1.

Geometric

parameters

Dimension

in mm

Dimension in λc

(fc = 2.175 GHz) Description

LA 56.0 0.406 Length of ground plate

LB 21.5 0.156 Length of upper plate

LF 20.6 0.149 Length of feed plate

W 9.0 0.065 Width of ground, feed, and upper

plates

WS 2.0 0.014 Width of short plate

hP 3.4 0.025 Height of feed plate

hS 5.6 0.041 Height of upper plate

LP 12.0 0.087 Distance of probe feed from short

plate

s 0.9 0.006 Separation distance between short

and feed plates

88

Figure 5-1 Geometry of the compact wideband J-pole (WC J-pole) antenna

designed for operation from fL = 1.77 GHz to fU = 2.45 GHz with

center frequency fc = 2.11 GHz.

Probe feedGround plane

Short plate or pin

Upper plate

Probe feedGround plane

Short plate or pin

Upper plate

Figure 5-2 Geometry of a conventional planar inverted-F antenna (PIFA).

LB=21.5

hS=5.6

LA=56.0

W=9WS=2

hp= 3.4

Dimensions in mm

Ground plate

Feed plateLP=12.0

LF=20.6s=0.9

Side View

Top View

Upper plate

Short plate

LB=21.5

hS=5.6

LA=56.0

W=9WS=2

hp= 3.4

Dimensions in mm

Ground plate

Feed plateLP=12.0

LF=20.6s=0.9

Side View

Top View

Upper plate

Short plate

x

y

z

x

y

z

89

5.3 Results of Preliminary Numerical Simulation and Experimental Investigations

Figure 5-3 illustrates the computed impedance characteristics and VSWR of the compact

WC J-pole antenna referenced to 50 Ohms using the moment method EM package IE3D.

The antenna exhibits a bandwidth of 790 MHz (for a VSWR less than 2), from 1.77 GHz

to 2.45 GHz, or 32.2 % bandwidth relative to a center frequency of 2.11 GHz based on

the computed results in Fig. 5-3b.

The antenna with the dimensions shown Table 5-1 was also fabricated (see photo in

Fig. 5-1). Impedance measurements were performed using a HP8720C network analyzer.

Figure 5-3(b) shows the measured VSWR of the WC J-pole antenna along with the

computed values. The plot shows that there is good agreement between computed and

measured VSWR, validating the IE3D simulation approach.

(a) (b)

Figure 5-3 Impedance characteristics as function of frequency for the compact

WC J-pole of Fig. 5-1 computed using the IE3D simulation code:

(a) Smith chart of complex impedance from 1.5 to 3.5 GHz; (b)

VSWR referenced to 50 Ohms. The measured VSWR (dashed

curve) is also plotted in (b).

1.5 1.75 2 2.25 2.5 2.75 31

1.5

2

2.5

3

3.5

4

4.5

5

Frequency f, GHz

VS

WR

1.5 1.75 2 2.25 2.5 2.75 31

1.5

2

2.5

3

3.5

4

4.5

5

Frequency f, GHz

VS

WR

90

The total far-field radiation patterns of the WC J-pole computed using IE3D are

presented in Fig. 5-4 for the three principal plane cuts: x-y, x-z, and y-z planes at 2.2

GHz. For each cut, two E-field pattern components are shown, one for each polarization

Eθ and Eφ, which are perpendicular to each other. As illustrated, the E-field component

that gives maximum intensity in each cut is the one that is parallel to the x-axis when

radiation peak intensity occurs. This suggests that the main radiating mechanism is due to

currents running in directions along the long metallic dimensions; that is along the x-axis.

5.4 Antenna Structure Similarities of WC J-Pole to Other Antennas

As mentioned in Section 5.1, the WC J-pole antenna evolved from a conventional PIFA

but has a capacitive feed rather than a contacting feed probe. The use of a capacitive feed

in PIFAs is described in Rowell and Murch’s article [2]. The WC J-Pole capacitive feed

differs from that described by Rowell and Murch in size. In fact, the feed plate in Rowell

and Murch’s design is much smaller than the top plate in the WC J-Pole. The significant

advantage of the WC J-pole is that the ground plate is small compared to the operating

wavelength and therefore radiates. Thus, the WC J-pole has completely different

radiation characteristics from the conventional PIFA structure. Figure 5-5 illustrates the

average current distribution of the WC J-pole at 2.2 GHz along with the average current

distribution of a conventional PIFA to show that the ground plate of the WC J-pole

radiates while the ground plate in the conventional PIFA has low current distribution and,

thus, low radiation intensity. As illustrated in Fig. 5-4, the WC J-pole has similar

radiation characteristics of a dipole with the main radiating element being the lower plate,

i.e., the ground plate, in the structure. If one looks at the WC J-pole without the

capacitive feed, the structure is similar to that of a J-pole antenna. Therefore, it is helpful

to look at the characteristics of J-pole antennas for the analysis of the proposed wideband

antenna.

91

(a)

(b)

(c)

Eθ

EΦ

Eφ

Figure 5-4 Far-field radiation patterns of the WC J-pole antenna of Fig. 5-1 at

2.2 GHz computed using IE3D for: (a) xz-plane, (b) yz-plane, and

(c) xy-plane.

10 0 -10 -20 -30 -40 -30 -20 -10 0 10

30

210

60

240

90 270

120

300

150

330

180

0

10 0 -10 -20 -30 -40 -30 -20 -10 0 10

30

210

60

240

90 270

120

300

150

330

180

0

10 0 -10 -20 -30 -40 -30 -20 -10 0 10

30

210

60

240

90 270

120

300

150

330

180

0

z

x

z

x

z

y

z

y

x

y

x

y

xx

zzyy

xx

zzyy

92

(a) (b)

Figure 5-5 Average current distribution of (a) the WC J-pole and (b) a

conventional PIFA on a large finite ground plane. The magnitude of

the current distribution goes from minimum (blue color) to

maximum (red color). The results were obtained from simulation

using IE3D.

The J-pole antenna structure is shown in Fig. 5-6. The structure consists a wire of

half-wavelength in length fed by a quarter-wave matching stub. Effectively, the antenna

is an end-fed half-wave dipole. The quarter-wave stub acts a transformer that provides a

means of transforming the high impedance of the antenna to the impedance of the

transmission line. The antenna has an omnidirectional pattern in the x-y plane. The return

loss for a J-pole antenna designed to resonate at about 1.8 GHz is shown in Fig. 5-7. The

antenna is matched to 50-Ohm impedance. Simulations were performed using IE3D,

which is suitable for use with wire and planar structures. As shown in Fig 5-7, J-pole

antennas have poor impedance bandwidth (about 3 %).

93

fc = 1.8 GHz

LA = 120 mm

LB = 40 mm

LC = 3 mm

LD = 3 mm

Figure 5-6 Structure of a J-pole antenna with dimensions for operation at 1.8

GHz with a wire radius of 0.635 mm.

Figure 5-7 Return loss of J-pole antenna of Fig. 5-6 matched to 50-Ohm

impedance computed using IE3D.

The J-pole antenna can be extended to form a planar structure, as shown in Fig. 5-8,

which is then very similar to the WC J-pole antenna structure without a capacitive feed.

LA = ¾ λ

LB = ¼ λ

LC

LD

Z

End-fed half-wave dipole

Quarter-wave stub

LA = ¾ λ

LB = ¼ λ

LC

LD

Z

End-fed half-wave dipole

Quarter-wave stub

94

The planar geometry introduces the width parameter that can be used to control its

characteristics. Figure 5-9 presents computed impedance values for the planar J-pole of

Fig. 5-8 for several antenna width values W. The results show that overall width W

influences the antenna input impedance. As the width W increases, the real part of the

impedance, or resistance, decreases, as well as the impedance imaginary part, or

reactance. However, if only the width of the lower and upper plates, W, is increased

without changing the shorting plate width Ws, the resistance and reactance values increase

and the antenna resonant point shifts downward in frequency, as can be seen in Fig 5-9.

The fact that the resonant frequency decreases in this case is because the path of the

current flow is longer, as illustrated in Fig. 5-10.

By transforming the wire J-pole antenna structure into a planar one and making the

width WS of the shorting plate narrower than the overall antenna width W, one is able to

decrease the resonant frequency from 1.82 GHz (wire structure) to 1.64 GHz (planar

structure with overall width W = 9 mm and short plate width WS = 1 mm) and at the same

time increase the impedance bandwidth from 73.5 MHz or 4 % fractional bandwidth to

87 MHz or 5.3 %. If the same planar J-pole structure is to be designed at the frequency

of the wire counterpart (fc = 1.82 GHz), its overall length LA will be reduced by 10 %.

Therefore the planar structure with WS<W leads to a size reduction of the antenna.

LA = 120 mm

LB = 40 mm

LC = 3 mm

LD = 3 mm

W = (1,5,9) mm

WS = W

Figure 5-8 Planar J-Pole antenna compared to a wire J-pole antenna.

J-Pole Antenna Planar J-Pole Antenna

WS

W

LA

LB

LC

J-Pole Antenna Planar J-Pole Antenna

WS

W

LA

LB

LC

95

1 1.2 1.4 1.6 1.8 2 2.2 2.4 2.6 2.8 30

10

20

30

40

50

60

70

80

Frequency f, GHz

Rea

l (Z1

1)W = Ws = 1 mmW = Ws = 5 mmW = Ws =9 mmW = 9 mm, Ws = 1 mm

(a) Resistance

1 1.2 1.4 1.6 1.8 2 2.2 2.4 2.6 2.8 3-10

0

10

20

30

40

50

60

70

80

90

Frequency f, GHz

Imag

(Z11

)

W = Ws = 1 mmW = Ws = 5 mmW = Ws =9 mmW = 9 mm, Ws = 1 mm

(b) Reactance

, Ω, Ω

96

1 1.2 1.4 1.6 1.8 2 2.2 2.4 2.6 2.8 3-25

-20

-15

-10

-5

0

Frequency f, GHz

|S11

|

W = Ws = 1 mmW = Ws = 5 mmW = Ws = 9 mmW = 9 mm, Ws = 1 mm

(c) Magnitude of S11

Figure 5-9 Impedance properties of the planar J-pole antenna of Fig. 5-8 for

various antenna widths WS computed using IE3D: (a) resistance, (b)

reactance, and (c) magnitude of S11.

Shor

t Pla

te

Shor

t Pla

te

Shor

t Pla

teSh

ort P

late

Shor

t Pla

teSh

ort P

late

Figure 5-10 Illustration of the increase in current path length accomplished by

narrowing the short plate (WS) width of the planar J-pole antenna of

Fig. 5-8.

97

5.5 Analysis of the Compact Planer Wideband J-Pole Antenna

The results from the numerical study presented in the previous section indicating that a

narrow shorting plate leads to size reduction of a planar J-pole antenna suggests further

investigation of the remaining antenna size parameters: the lengths of the lower and

upper plates, the short plate height, and the feed probe position. However, the planar J-

pole antenna still has a narrow impedance bandwidth (5.3 %) and many applications

require wider operating bandwidths. For example, many commercial wireless application

bands have about 8 to 10 % bandwidth, such as DCS-1800 and PCS-1900; see Table 1.1.

Thus, this study seeks feed geometries for wideband operation.

It was shown in the Chapter 2 that one way to the increase impedance bandwidth of

planar microstrip antennas is to use a probe-feed configuration to increase the height of

the microstrip patch above the ground plane. However, doing so leads to an unavoidable

impedance mismatch due to an inductive impedance component associated with the

probe [2-4]. The inherent inductive nature of the feed probe in the planar J-pole antenna

in Fig. 5-8 would give the same impedance mismatch problem to the antenna with

increased distance between the top and lower plate of the planar J-pole. In order to solve

this problem, a capacitive-type feed is used to help cancel the inductive antenna

reactance. This increases the structural complexity slightly and introduces a few more

parameters in the analysis, i.e., the feed plate area and position, and the probe position

relative to the feed plate. The following presents the results of a parametric study of the

WC J-pole antenna structure and a design guideline for a 50%-impedance bandwidth

model.

5.5.1 Parametric Analysis of the Compact WC J-pole Antenna

The parametric studies of the WC J-pole antenna are based on the structure shown in Fig.

5-11, which shows the parameters to be examined. The influence of each parameter on

98

the antenna impedance characteristics was investigated using IE3D. The initial

dimensions of the structure are taken from the preliminary model, which were shown in

Fig. 5-1, and are shown in Table 5-2. The fractional impedance bandwidth from this

antenna is 32.2 % for a VSWR less than 2 and a 50-Ohm impedance match; see Fig 5-3.

Figure 5-12 shows the impedance characteristics for the WC J-pole of Fig. 5-11 and

Table 5-2. The goal of this study is to increase the operating bandwidth even more than

32.2 %.

Table 5-2 Geometric parameters of the preliminary WC J-pole antenna of Fig.

5-11.

Geometric parameters Dimension in mm

LA 56.0

LB 21.5

LF 20.6

W 9.0

WS 2.0

hP 3.4

hS 5.6

LP 12.0

s 0.9

Input impedance of an antenna is represented as

−+=

CLjRZ

ωω 1 (5.1)

where R, L, C, and ω are the overall antenna resistance, inductance, capacitance, and the

radian frequency, respectively. In order to match the antenna impedance Z to 50 Ohms,

its imaginary part should be zero: this occurs where the reactance curve crosses zero. For

resonant antennas, this reactance curve has zero crossings with a sharp slope, which

corresponds to a small impedance bandwidth. In order to widen the impedance bandwidth

99

wider, a structure can be designed so that the reactance value varies around zero over a

wide frequency range, while still keeping the real part of the impedance around 50 Ohms,

which is shown in Fig. 5-12(b).

LB

hS

LA

WWS

hp

dP

LFs

LB

hS

LA

WWS

hp

dP

LFs

Figure 5-11 WC J-pole antenna structure with its dimension parameters.

1500 1750 2000 2250 2500 2750 3000-50

0

50

100

150

Frequency f, MHz

Impe

danc

e Z1

1, O

hm

ResistanceReactance

(a) Impedance

100

1500 1750 2000 2250 2500 2750 30001

1.5

2

2.5

3

3.5

4

4.5

5

Frequency f, MHz

VS

WR

(b) VSWR

Figure 5-12 Input impedance (a) and VSWR relative to 50 Ohms (b) of the WC

J-pole antenna of Fig. 5-11 and Table 5-2 computed using IE3D.

5.5.1.1 Variation of Feed Probe Height, hP

In this parametric investigation, all the dimensions of the antenna of Fig. 5-11 and Table

5-2 are fixed except for hp, which varies from 3.5 mm to 5.5 mm. Variation of the probe

feed height hp affects the capacitance and inductance of the antenna as well as its

resistance. Shortening the feed probe reduces its resistance. Therefore, we should expect

the input resistance of the antenna to decrease with the feed probe height hp. At the same

time, as the feed plate is moved away from the top plate, the capacitance between the two

plates is decreased. Therefore, a decrease in the antenna input reactance is expected

because the term Cϖ1 starts to dominate over ωL for small C values according to (5.1).

These effects are shown in Fig. 5-13.

101

Note from Fig. 5-13 that the resonance dips remain centered close to 2 GHz for all

examined values of hp. Therefore, the feed probe height can be used to control the

antenna impedance without affecting its frequency characteristics.

1500 2000 2500 3000 3500 40000

50

100

150

200

250

300

Frequency f, MHz

Rea

l (Z1

1)

hp = 3.5 mmhp = 4.0 mmhp = 4.5 mmhp = 5.0 mmhp = 5.5 mm

(a) Antenna resistance

1500 2000 2500 3000 3500 4000-150

-100

-50

0

50

100

150

200

250

Frequency f, MHz

Imag

(Z11

)

hp = 3.5 mmhp = 4.0 mmhp = 4.5 mmhp = 5.0 mmhp = 5.5 mm

(b) Antenna reactance

, Ω, Ω

102

1500 2000 2500 3000 3500 4000-30

-25

-20

-15

-10

-5

0

Frequency f, MHz

|S11

|, dB

hp = 3.5 mmhp = 4.0 mmhp = 4.5 mmhp = 5.0 mmhp = 5.5 mm

(c) Magnitude of S11

Figure 5-13 Impedance of the WC J-pole antenna shown in Fig. 5-11 and Table

5-2 for various feed probe height hp values: (a) real part of

impedance (antenna input resistance), (b) imaginary part of

impedance (antenna input reactance), and (c) |S11| of the structure

matched to 50-Ohm impedance, The height of the top plate is 5.6

mm.

5.5.1.2 Variation of Feed Plate Area

The second geometric parameter investigation was on the surface area of the feed plate,

which controls the amount of capacitance between the top and feed plates. It is expected

that the reactance will increase with increasing feed plate area according to (5.1), because

the capacitance value C increases. For a square-shaped feed plate area, there are three

situations, as depicted in Fig. 5-14: (1) the feed plate is smaller than the top and bottom

plate, (2) the feed plate is wider but shorter than the top plate, and (3) the feed plate right

edge goes beyond that of the top plate. Dimensions of the WC J-pole antenna of 5-14 for

103

this study is shown in Table 5-3. Figure 5-15 shows the square feed plate area variation,

while the other structure dimensions remain fixed. As the area increases to LF×LF = 9×9

mm2, which is a case (1) example, the antenna reactance increases and its resistance

change is very small, as expected. When the area increases further to LF×LF = 14×14

mm2, the feed plate is wider but shorter than the top plate (case 2) and there is almost no

change in the impedance curves, which is expected because increasing the feed plate area

larger than that of the top plate does not increase the capacitance between the plates

much. As the feed plate area is increased even further and its right edge goes beyond the

right edge of the top plate (case 3), a small decrease in resistance value occurs.

Table 5-3 Geometric parameters of the WC J-pole antenna of Fig. 5-14 for the

feed plate area study.

Geometric parameters Dimension in mm

LA 56.0

LB 21.0

LF 2.0 to 22.0

W 9.0

WS 2.0

hP 3.4

hS 5.6

LP 13.0

s 0.9

104

LF

LF

LF

LF

(a) Tuning plate geometry

(b)

9 mm

21 mm

13 mm

56 mm

Case 1: feed plate smaller than top plate.

Case 2: feed plate wider than top plate and its right edge remains under top plate.

Case 3: feed plate wider the top plate and its right edge goes beyond top plate’s right edge.

9 mm

21 mm

13 mm

56 mm

Case 1: feed plate smaller than top plate.

Case 2: feed plate wider than top plate and its right edge remains under top plate.

Case 3: feed plate wider the top plate and its right edge goes beyond top plate’s right edge.

(b) Relative tuning plate size cases

Figure 5-14 Feed plate area variation of the capacitive antenna structure. The

square shape feed plate side length LF increases from 2 to 22 mm.

105

1 2 3 4 5 6 7 8 9 100

20

40

60

80

100

120

140

Frequency f, GHz

Rea

l (Z1

1)

2x2 mm2 (case 1a)6x6 mm2 (case 1b)10x10 mm2 (case 2a)14x14 mm2 (case 2b)18x18 mm2 (case 3a)22x22 mm2 (case 3b)

(a) Antenna resistance

1 2 3 4 5 6 7 8 9 10-1200

-1000

-800

-600

-400

-200

0

200

400

Frequency f, GHz

Imag

(Z11

)

2x2 mm2 (case 1a)6x6 mm2 (case 1b)10x10 mm2 (case 2a)14x14 mm2 (case 2b)18x18 mm2 (case 3a)22x22 mm2 (case 3b)

(b) Antenna reactance

Figure 5-15 Impedance of the antenna structure with square feed plate LF×LF

area variation from LF×LF = 2x2 mm2 (case 1) to LF×LF = 10x10

mm2 to LF×LF = 22x22 mm2 (case 3), according to Fig. 5-14.

LF×LF

, Ω, Ω

106

For the case when the feed plate right edge goes beyond the right edge of the top

plate (case 3), some perturbation in the impedance occurs in higher frequencies, around

5.5 GHz. This suggests that the portion of the feed plate that goes beyond the right edge

of the top plate radiates, leading to the perturbation in the impedance curve. This effect is

better seen in Figure 5-17 which shows calculated input impedance for an exposed feed

plate by increasing the exposed feed plate length LF by a portion Lu beyond the right edge

of the top plate, as shown in Fig. 5-16 and Table 5-4. As the length Lu of this portion

increases, the resistance value of the antenna impedance at around the 3-GHz region

decreases while the reactance at this frequency band is not affected. Lu is, therefore, a

good parameter to control the resistance of the antenna impedance without affecting its

reactance. It is noted also that the resonance in the upper frequency band around 5-6 GHz

that was created by this unshielded portion of the feed plate decreases with increasing Lu.

This feature can be helpful for developing a dual-band antenna structure.

Table 5-4 Geometric parameters of the WC J-pole antenna of Fig. 5-16 for the

feed plate area study.

Geometric parameters Dimension in mm

LA 56.0

LB 21.0

LF 20.1

Lu 0.0 to 10.0

W 9.0

WS 2.0

hP 3.4

hS 5.6

LP 12.0

s 0.9

107

Lu

0.9 20.1

9.0

56.0

Dimensions in mm

5.6 3.4

21.0

Exposed portion of feed plate

Lu

0.9 20.1

9.0

56.0

Dimensions in mm

5.6 3.4

21.0

Exposed portion of feed plate

Figure 5-16 Geometry of the WC J-pole antenna for the study of the feed plate

area not shielded by the top plate.

1 2 3 4 5 6 7 8 9 10-50

0

50

100

150

200

250

300

Frequency f, GHz

Rea

l (Z1

1)

Lu = 0 mmLu = 2 mmLu = 4 mmLu = 6 mmLu = 8 mmLu = 10 mm

(a) Antenna resistance

, Ω

108

1 2 3 4 5 6 7 8 9 10-100

-50

0

50

100

150

200

250

300

Frequency f, GHz

Imag

(Z11

)

Lu = 0 mmLu = 2 mmLu = 4 mmLu = 6 mmLu = 8 mmLu = 10 mm

(b) Antenna reactance

1 2 3 4 5 6 7 8 9 10-35

-30

-25

-20

-15

-10

-5

0

Frequency f, GHz

Mag

(S11

)

Lu = 0 mmLu = 2 mmLu = 4 mmLu = 6 mmLu = 8 mmLu = 10 mm

(c) |S11|

Figure 5-17 Input impedance for the WC J-pole antenna of Fig. 5-16 and Table

5-4 calculated using IE3D: (a) resistance, (b) reactance, and (c)

return loss referenced to 50 Ohms.

Lu

, Ω

109

5.5.1.3 Variation of Feed Probe Position, dp

The feed assembly of the WC J-pole antenna consisting of a tuning plate and a probe can

be positioned laterally, i.e., relative to the lower plate by adjusting the distance dp. Figure

5-19 shows the impedance results of the antenna structure shown in Fig. 5-18 where all

the dimensions shown in Table 5-5 remain fixed except for the probe position dp. As the

probe distance dp increases (i.e., probe moves from the short plate), the resistance of the

antenna impedance increases but the peak values are not shifted in frequency. The

reactance variations due to probe distance variation in Fig. 5-19b are less drastic. The

increase in reactance at about 2 GHz is very small and probe positioning can therefore be

useful to match antenna resistance without affecting too much its reactance around this

frequency band.

Table 5-5 Geometric parameters of the WC J-pole antenna of Fig. 5-18 for feed

probe position study.

Geometric parameters Dimension in mm

LA 56.0

LB 21.5

LF 20.6

W 9.0

WS 2.0

hP 3.4

hS 5.6

dP 4.0 to 20.0

s 0.9

110

dp

0.9 20.6

9.0

56.0

Dimensions in mm

5.6 3.4

dp

0.9 20.6

9.0

56.0

Dimensions in mm

5.6 3.4

dp

0.9 20.6

9.0

56.0

Dimensions in mm

5.6 3.4

Figure 5-18 Geometry of the WC J-pole antenna. dp is the probe feed position

away from the short plate.

1500 2000 2500 3000 3500 40000

20

40

60

80

100

120

140

160

180

200

Frequency f, MHz

Rea

l (Z1

1)

dp= 4 mmdp = 8 mmdp = 12 mmdp = 16 mmdp = 20 mm

(a) Antenna resistance

, Ω

111

1500 2000 2500 3000 3500 4000-100

-50

0

50

100

150

Frequency f, MHz

Imag

(Z11

)

dp = 4 mmdp = 8 mmdp = 12 mmdp = 16 mmdp = 20 mm

(b) Antenna reactance

1500 2000 2500 3000 3500 4000-35

-30

-25

-20

-15

-10

-5

0

Frequency f, MHz

Mag

(S11

)

Dp = 4 mmDp = 8 mmDp = 12 mmDp = 16 mmDp = 20 mm

(c) |S11|

Figure 5-19 Input impedance of WC J-pole antenna shown in Fig. 5-18 and

Table 5-5 calculated for values of feed probe position dp from 4 mm

to 20 mm away from the antenna short plate with all other

dimensions fixed.

, Ω

112

The feed assembly position within the WC J-pole dp was also examined for the

structure dimensions shown in Fig 5-20, and Table 5-6. Two cases can arise: the feed

plate is either completely or partially shielded by the top plate. As the feed assembly

distance from the short plate dp increases while still completely shielded by the top plate,

the antenna resistance peak value increases as well as the reactance peak at 1.8 GHz, with

the reactance value between 2 and 3 GHz remains almost unchanged, as shown in Fig. 5-

21. As the feed assembly distance increases further such that a portion of the feed plate

becomes unshielded by the top plate, the resistance values decrease, as discussed

previously for the case of unshielded feed plate portion. The reactance decreases as well,

which is due to the fact that the surface area of the feed plate that couples with the top

plate decreases, leading to a decrease in capacitance between the plates, hence a decrease

of the reactance as well, according to (5.1). This behavior is illustrated in Fig. 5-22.

Table 5-6 Geometric parameters of the WC J-pole antenna of Fig. 5-20 for feed

assembly position study.

Geometric parameters Dimension in mm

LA 56.0

LB 21.5

LF 10.0

W 9.0

WS 2.0

hP 3.4

hS 5.6

dP 8.0 to 20.0

s 0.9

113

dp

4.0

10.0

9.0

56.0

Dimensions in mm

5.6 3.4

21.5

dp

4.0

10.0

9.0

56.0

Dimensions in mm

5.6 3.4

21.5

Figure 5-20 Geometry of the WC J-pole antenna. dp is the probe feed position

away from the short plate and the feed plate is moved along with

the probe feed.

1500 2000 2500 3000 3500 40000

20

40

60

80

100

120

140

Frequency f, MHz

Rea

l (Z1

1)

dp= 8 mmdp = 10 mmdp = 12 mmdp = 14 mm

(a) Resistance

, Ω

114

1500 2000 2500 3000 3500 4000-90

-80

-70

-60

-50

-40

-30

-20

-10

0

10

Frequency f, MHz

Imag

(Z11

)

dp= 8 mmdp = 10 mmdp = 12 mmdp = 14 mm

(b) Reactance

1500 2000 2500 3000 3500 4000-45

-40

-35

-30

-25

-20

-15

-10

-5

0

Frequency f, MHz

Mag

(S11

)

dp = 8 mmdp = 10 mmdp = 12 mmdp = 14 mm

(c) |S11|

Figure 5-21 Impedance of the antenna structure shown in Fig. 5-20 and Table 5-

6 computed using IE3D for the case when the feed plate is

completely shielded by the top plate.

, Ω

115

1500 2000 2500 3000 3500 40000

20

40

60

80

100

120

140

Frequency f, MHz

Rea

l (Z1

1)

dp= 16 mmdp = 18 mmdp = 20 mm

(a) Antenna resistance

1500 2000 2500 3000 3500 4000-100

-80

-60

-40

-20

0

20

Frequency f, MHz

Imag

(Z11

)

dp= 16 mmdp = 18 mmdp = 20 mm

(b) Antenna reactance

, Ω, Ω

116

1500 2000 2500 3000 3500 4000-40

-35

-30

-25

-20

-15

-10

-5

0

Frequency f, MHz

Mag

(S11

)

dp = 16 mmdp = 18 mmdp = 20 mm

(c) |S11|

Figure 5-22 Impedance of the antenna structure shown in Fig. 5-20 and Table 5-

6 computed using IE3D for the case when a portion of the feed plate

is unshielded by the top plate.

5.5.1.4 Variation of the Short Plate Height, hs

One of the important antenna parameters that has influence not only in the overall

structure dimensions but also in its impedance characteristics is the height of the short

plate, hs. The height parameter effects were investigated by simulating the antenna of Fig.

5-23 and Table 5-7 for hs values from 4.0 to 7.5 mm. Figure 5-24 presents computed

impedance results for the structure of Fig. 5-23. As hs increases, the peak values of both

resistance and reactance of the antenna decrease, leaving a smoother impedance

fluctuation with frequencies, as shown in Fig 5-24 (a) and (b). The reason that the

reactance value decreases is due again to the decrease in coupling effect between to top

and feed plates, according to (5.1). A smooth variation in resistance about the specified

matching impedance, 50 Ohms in this case, and in reactance about zero Ohm over a wide

frequency band gives rise to a wideband antenna, as depicted in Fig. 5-22 (c). However,

117

when the height increases above 6.5 mm, the resistance and reactance values becomes too

small for a good impedance match around 2 GHz. Therefore, the thicker the antenna, the

wider its impedance bandwidth up to certain limit.

Table 5-7 Geometric parameters of the WC J-pole antenna of Fig. 5-23 for

antenna height hs study.

Geometric parameters Dimension in mm

LA 56.0

LB 21.5

LF 20.6

W 9.0

WS 2.0

hP 3.4

hS 4.0 to 7.5

LP 12.0

s 0.9

12.0

0.9 20.6

9.0

56.0

Dimensions in mm

hs 3.4

12.0

12.0

0.9 20.6

9.0

56.0

Dimensions in mm

hs 3.4

12.0

Figure 5-23 Geometry of the WC J-pole antenna for the height hs study.

118

1500 2000 2500 3000 3500 40000

50

100

150

200

250

300

Frequency f, MHz

Rea

l (Z1

1)

hs = 4.0 mmhs = 4.5 mmhs = 5.0 mmhs = 5.5 mmhs = 6.0 mmhs = 6.5 mmhs = 7.0 mmhs = 7.5 mm

(a) Antenna Resistance

1500 2000 2500 3000 3500 4000-150

-100

-50

0

50

100

150

Frequency f, MHz

Imag

(Z11

)

hs = 4.0 mmhs = 4.5 mmhs = 5.0 mmhs = 5.5 mmhs = 6.0 mmhs = 6.5 mmhs = 7.0 mmhs = 7.5 mm

(b) Antenna reactance

, Ω, Ω

119

1500 2000 2500 3000 3500 4000-35

-30

-25

-20

-15

-10

-5

0

Frequency f, MHz

Mag

(S11

)

hs = 4.0 mmhs = 4.5 mmhs = 5.0 mmhs = 5.5 mmhs = 6.0 mmhs = 6.5 mmhs = 7.0 mmhs = 7.5 mm

(c) |S11|

Figure 5-24 Impedance of the antenna structure shown in Fig. 5-23 and Table 5-

7 computed for various short plate heights hs.

5.5.1.5 Variation of the Lower Plate Length, LA

The length of the lower plate is an important parameter because it determines the antenna

size. When size reduction is required, it is necessary to minimize this parameter value.

Figure 5-25 illustrates the configuration of the antenna structure with dimensions listed in

Table 5-8 used to analyze its input impedance behavior influenced by the variation of the

lower plate length, as shown in Fig. 5-26. When decreasing this length, there is a shift

upward in frequency, which is expected since the structure is similar to the J-pole antenna

and this element is the main radiating part of the antenna. This parameter can be used to

adjust impedance fluctuation over the frequencies in the resonant range around 2 and 3

GHz.

120

Table 5-8 Geometric parameters of the WC J-pole antenna of Fig. 5-25 for

antenna length LA study.

Geometric parameters Dimension in mm

LA 45.0 to 60.0

LB 21.5

LF 9.0

W 9.0

WS 2.0

hP 3.4

hS 5.6

LP 12.0

s 0.9

12.0

4.5

9.0

9.0

LA

Dimensions in mm

5.6 3.4

21.5

12.0

4.5

9.0

9.0

LA

Dimensions in mm

5.6 3.4

21.5

Figure 5-25 Geometry of the WC J-pole antenna for the analysis on the variation

of the lower plate length LA.

121

1500 2000 2500 3000 3500 40000

20

40

60

80

100

120

140

160

Frequency f, GHz

Rea

l (Z1

1)

LA = 45 mmLA = 50 mmLA = 55 mmLA = 60 mm

(a) Antenna resistance

1500 2000 2500 3000 3500 4000-100

-80

-60

-40

-20

0

20

40

60

Frequency f, GHz

Imag

(Z11

)

LA = 45 mmLA = 50 mmLA = 55 mmLA = 60 mm

(b) Antenna reactance

, Ω, Ω

122

1500 2000 2500 3000 3500 4000-30

-25

-20

-15

-10

-5

0

Frequency f, MHz

|S11

|, dB

LA = 45 mmLA = 50 mmLA = 55 mmLA = 60 mm

(c) |S11|

Figure 5-26 Impedance of the antenna structure shown in Fig. 5-25 and Table 5-

8 for lower plate length variation LA study.

5.5.1.6 Variation of the Top Plate Length, LB

The top plate of the antenna structure, as shown in Fig. 5-27 and Table 5-9, influences the

input impedance in a similar fashion as with the lower plate. Its length controls the

frequency of resonance as well as the fluctuation of the antenna resistance and reactance,

as illustrated in Fig. 5-28. As the length LB increases, the resonant frequency is shifted

downward. This suggests that the top plate length plays a role in radiation and therefore,

varying this parameter not only changes the antenna input impedance values but also

affects the resonant frequency.

123

Table 5-9 Geometric parameters of the WC J-pole antenna of Fig. 5-27 for

antenna length LB study.

Geometric parameters Dimension in mm

LA 56.0

LB 15.0 to 22.5

LF 9.0

W 9.0

WS 2.0

hP 3.4

hS 5.6

LP 12.0

s 0.9

12.0

4.5

9.0

9.0

56.0

Dimensions in mm

5.6 3.4

LB

12.0

4.5

9.0

9.0

56.0

Dimensions in mm

5.6 3.4

LB

Figure 5-27 Geometry of the WC J-pole antenna for the analysis on the variation

of the top plate length LB.

124

1500 2000 2500 3000 3500 40000

20

40

60

80

100

120

140

160

180

200

Frequency f, MHz

Rea

l (Z1

1)LB = 15 mmLB = 17.5 mmLB = 20 mmLB = 22.5 mm

(a) Antenna resistance

1500 2000 2500 3000 3500 4000-120

-100

-80

-60

-40

-20

0

20

40

60

80

Frequency f, MHz

Imag

(Z11

)

LB = 15 mmLB = 17.5 mmLB = 20 mmLB = 22.5 mm

(b) Antenna reactance

, Ω, Ω

125

1500 2000 2500 3000 3500 4000-25

-20

-15

-10

-5

0

Frequency f, MHz

Mag

(S11

)

LB = 15 mmLB = 17.5 mmLB = 20 mmLB = 22.5 mm

(c) |S11|

Figure 5-28 Impedance of the antenna structure shown in Fig. 5-27 and Table 5-

9 for top plate length variation LB study.

5.5.2 Summary of the WC J-pole Parameter Variation

Table 5-10 summarizes impedance characteristics performance of the WC J-pole as

function of its geometric parameter variation previously analyzed. The dimension

parameters were shown in 5-11. The frequency band of interest is around the 2-GHz and

3-GHz region since there is resonance at each of these regions. Performance in the 5-GHz

region created by certain parameter variation cases is not shown because the focus of the

analysis is to help the antenna have wideband characteristics due to resonances in close-

proximity and resonance in the 5-GHz region is too far away from the other lower

resonances to play a role in widening antenna impedance bandwidth. However, this

resonance will be useful to design a dual-band antenna. Such an design example will be

given in Chapter 6. The definition of resonance at fo in Table 5-10 is more relaxed in the

126

sense that it is not at the frequency point where the antenna reactance is zero but rather it

is the frequency point with minimum |S11|.

127

Table 5-10 Summary of the WC J-pole Parameter Variation Analyzed in the

Previous Section. The Geometric Parameters are shown in Fig. 5-11.

Parameter Variation

Antenna

Performance

parameter

Effects in the 2-GHz Region Effects in the 3-GHz Region

Lower-plate length LA

(40 → 60 mm); see Fig. 5-26

ℜe

ℑm

fo

↓↓↓

↓↓↓

small shift down

↓↓

↑↑

large large shift down

Top-plate length LB

(15 → 22.5 mm); see Fig. 5-28

ℜe

ℑm

fo

↑

↑

small shift up

↓↓↓

↓↓↓

small shift down

Short-plate height hS

(4 → 7.5 mm); see Fig. 5-24

ℜe

ℑm

fo

↓

↓↓↓

small shift down

↑

↑↑↑

↓↓↓

GHz 3.0at GHz 2.9at constant

GHz 2.75at

large shift up

Feed-plate length LF

Feed plate < top plate (case 1)

(2x2 → 9x9 mm2); see Fig. 5-15

Feed plate > top plate (case 2 and 3)

(9x9 → 22x22 mm2); see Fig 5-15 and 5-17

ℜe

ℑm

fo

ℜe

ℑm

fo

↑

↑↑↑

negligible change

↓

negligible

negligible change

↑

↑↑↑

negligible change

↓

negligible

negligible change

A resonance appears in the 5-HGz region when the feed plate is longer than the top plate.

ℜe ↑↑↑ ↑↑↑

ℑm

↓↓↓

↑

↑↑↑

GHz 3at GHz 2.75at negligible

GHz 2.5at GHz 2at negligible

GHz 1.75at

Feed-probe position dp

(4 → 20 mm); see Figs 5-19

fo small shift up negligible change

ℜe ↑↑↑ ↑↑↑

ℑm

↓↓

↑↑↑↑↑↑

GHz 3at GHz 2.75at negligible

GHz 2.5at GHz 2at GHz 1.75at

Feed-probe height hp

(3.5 → 5.5 mm); see Fig 5-13

fo negligible change negligible change

Note: ↑ or ↓ : variation between 0 and 40 Ω; ↑↑ or ↓↓: variation between 41 and 80 Ω; ↑↑↑ or ↓↓↓: variation of more that 80 Ω.

128

5.5.3 Design Case for the Largest Impedance Bandwidth

After analyzing all the parameters of the WC J-pole antenna, the design process can be

optimized to obtain a structure with the largest impedance bandwidth for a given antenna

size. Since the length of the lower plate governs the overall size of the antenna, it is a

good idea to optimize the impedance bandwidth by adjusting the other parameters for a

given lower plate size. Figure 5-29 shows the antenna dimensions for an antenna with an

optimal impedance bandwidth of about 50 %, covering from 1595 to 2644 MHz for a

return loss of 10 dB or more. The length of the lower plate remained fixed during the

optimization process at LA = 60 mm. The width of the elements were fixed as well at W =

9 mm and Ws = 2 mm. The final antenna is shown in Fig. 5-29 for the optimized-

bandwidth WC J-pole antenna. The optimization goal was based on the return loss of 10

dB or more for a 50-Ohm impedance match, as shown in Fig. 5-30(a).

19.0

21.0

10.0

9.0

60.0

Dimensions in mm

6.0 3.5

23.0

2.0

19.0

21.0

10.0

9.0

60.0

Dimensions in mm

6.0 3.5

23.0

2.0

Figure 5-29 Dimensions of the WC J-pole antenna for the optimal impedance

bandwidth of about 50 %.

129

1500 1750 2000 2250 2500 2750 3000-60

-40

-20

0

20

40

60

80

Frequency f, MHz

Impe

danc

e, O

hm

RealImaginary

1500 1750 2000 2250 2500 2750 3000-60

-40

-20

0

20

40

60

80

Frequency f, MHz

Impe

danc

e, O

hm

RealImaginary

(a)

1500 1750 2000 2250 2500 2750 3000-25

-20

-15

-10

-5

0

Frequency f, MHz

Mag

(S11

)

S1,1

1500 1750 2000 2250 2500 2750 3000-25

-20

-15

-10

-5

0

Frequency f, MHz

Mag

(S11

)

S1,1

1500 1750 2000 2250 2500 2750 3000-25

-20

-15

-10

-5

0

Frequency f, MHz

Mag

(S11

)

S1,1

(b)

Figure 5-30 Input Impedance (a) and return loss for a 50-Ohm input impedance

match (b) of the WC J-pole antenna of Fig. 5-29 with its impedance

bandwidth optimized.

130

The results for the optimum-bandwidth WC J-pole design can be used to form a set

of design guidelines for the electrical dimensions (see Table 5-11). This permits scaling

the design to any frequency. The following antenna model has its size relative to the

wavelength λL obtained from the lower frequency fL of the antenna bandwidth. In the case

of the optimized bandwidth show in Fig. 5-30(a), fL = 1595 MHz, therefore, λL = 188

mm. Table 5-11 shows the design model size in terms of λL for a WC J-pole antenna

having an impedance bandwidth of 50 %.

Table 5-11 Electrical dimensions of the optimum bandwidth (50%) WC J-pole in

terms of the wavelength λL of the lower frequency fL of the operating

band. See Fig. 5-11 for geometry.

Structure Parameters Physical Size (mm) Electrical Size at fL

LA 60.0 0.319 λL

LB 23.0 0.122 λL

LF 21.0 0.112 λL

s 10.0 0.053 λL

dP 19.0 0.101 λL

hs 6.0 0.032 λL

hp 3.5 0.019 λL (0.583 hs)

W 9.0 0.048 λL

WS 2.0 0.011 λL (0.222 W)

131

5.6 Effects of Human Operator on WC J-Pole Antenna Performance

It is important to comment on the effects of human operator on the WC J-pole antenna

performance since the antenna can be use in proximity of a hand or a head. It was

mentioned previously that the WC J-Pole antenna is similar to a dipole antenna in terms

of radiation mechanism. It is then expected that the effects of human operator in close

proximity of the WC J-pole are similar to the effects described in Chapter 4 for the

monopole antenna.

5.7 Summary

A new wideband compact antenna was introduced. It was shown that the proposed

antenna has similar radiation characteristic of a J-pole antenna. By extending to a planar

structure and using a capacitive feed instead of a conventional probed feed, antenna size

was reduced and impedance bandwidth was widened. With a further parametric study on

the antenna structure, a design for optimal impedance-bandwidth antenna was obtained.

The new wideband compact J-pole (WC J-pole) has 50 % bandwidth.

5.8 References

[1] K. Hirisawa and M. Haneishi, Analysis, Design, and Measurement of small and

Low-Profile Antennas, Artech House, Boston: 1992.

[2] C.R. Rowell and R.D. Murch, “A capacitively Loaded PIFA for Compact Mobile

Telephone Handsets”, IEEE Trans. On Antennas and Propagation, Vol. AP-45,

No. 5, May 1997, pp. 837-841.

132

[3] G.A.E. Vandenbosch and A.R. Van de Capelle, “Study of the Capacitively Fed

Microstrip Antenna Element,” IEEE Trans. On Antennas and Propagation, Vol.

AP-42, No. 12, Dec. 1994, pp. 1648-1652.

[4] Y.T. Lo and S.W. Lee, Antenna Handbook: Antenna Theory, Vol. II, chap. 10,

Van Nostrand Reinhold Co., New-York: 1988.

133

Chapter 6

Variations of the WC J-Pole for a Few Commercial Applications

6.1 Introduction

In the previous chapter, the WC J-Pole antenna was analyzed and detailed parametric

investigations on the antenna impedance characteristics showed that the antenna

operating frequency and bandwidth can be easily tuned by adjusting the size parameters.

There are two antenna resonances that can occur close to each other, forming an overall

wide operating band that can cover applications ranging from 1500 to 2500 MHz or for

future applications in a portion of the ultra-wide band from 3 to 5 GHz [1]. The antenna

dimensions can be adjusted for dual-band applications, as we discuss in this Chapter.

6.2 Wideband Compact Antenna for Covering Applications from GPS to Bluetooth Bands (WCJP #1)

Figure 6-1 shows dimensions of a version of the WC J-Pole antenna for an operating

frequency band covering from GPS (1570 MHz) to Bluetooth (2500 MHz) bands (WCJP

#1). The impedance match and gain characteristics were calculated over the complete

band from 1500 to 2600 MHz and are plotted in Figs 6-2 and 6-3. A VSWR value below

2 is commonly accepted for determining the impedance bandwidth for antennas in

personal wireless devices. Figure 6-2 shows that the antenna has an acceptable

134

impedance bandwidth of 990 MHz (49 %) extended from 1.525 GHz to 2.515 GHz for a

2:1 VSWR, which covers from the GPS to the Bluetooth bands. The computed gain data

in Fig. 6-3 shows that the gain is very stable over its full impedance bandwidth.

There are a large number of uses for this wideband antenna in personal wireless

applications. Mobile phones, personal digital assistant devices (PDAs), and laptop

computers are the potential applications. Tremendous importance is also given in the

United States, as well as in Europe to the development of third-generation (3G) wireless

personal communication systems. These systems are known as IMT-2000, or UMTS in

Europe; the frequency bands are shown in Table 6-1. Transition from analog personal

radios using the AMPS band at 800-900 MHz to digital radios (2G) at 1900 MHz PCS

band require wireless phones to operate in both bands. Likewise, the transition from 2G

to 3G personal radios will require systems, and therefore antennas, to operate in PCS

band as well as IMT-2000 band. Therefore, the WCJP #1 is a good candidate for such a

transition.

New applications are arising that will be included in mobile phones. One prominent

example is Bluetooth. Applications using the Bluetooth (2400-2483 MHz) band include:

wireless headsets for mobile phones, synchronization of mobile phones and PDAs with

desktop computers and laptops, mobile phones in a handheld remote device to control

other Bluetooth-enabled appliances. Another prominent area of interest in the next

generation of wireless personal communication systems is the integration of GPS in

phones. This market will be bustling with activity in the next few years because of the

FCC’s mandate that requires all mobile phone manufacturers to include position location

feature in the so called E911 specification. This feature allows 911 operators to pinpoint a

mobile phone user’s location through GPS technology.

135

Table 6-1 Frequency Bands for a Few Wireless Applications.

Wireless Applications Frequency Band (MHz) Bandwidth (MHz)

GPS 1570.42-1580.42 10 (0.7%)

DCS-1800 1710-1880 170 (10.6%)

PCS-1900 1850-1990 140 (7.3%)

IMT-2000/UMTS (3G) 1885-2200 315 (15.5%)

ISM (including WLAN) 2400-2483 83 (3.4%)

Bluetooth 2400-2500 100 (4.1%)

U-NII 5150-5350 / 5725-5825 200 (3.8%) / 100 (1.7%)

z

x

y

x

26.0

6.0

62.0

10.03.2

28.7

2.8

Dimensions in mm

Plate G

Plate F

1.3 Plate R

16.2

SMA ConnectorVia

z

x

z

x

y

x

y

x

26.0

6.0

62.0

10.03.2

28.7

2.8

Dimensions in mm

Plate G

Plate F

1.3 Plate R

16.2

SMA ConnectorVia

Figure 6-1 Wideband compact J-pole antenna (WCJP #1) designed to cover

frequency bands from GPS to Bluetooth bands.

136

PCS

DCS

IMT2000

Bluetooth

ISM

GPS

Figure 6-2 VSWR values computed using IE3D for the WC J-Pole of Fig. 6-1

relative to 50-Ohms. Note an impedance match (VSWR≤2) is

achieved for the frequency bands of interest.

Gai

n, d

Bi

Frequency, GHz

Gai

n, d

Bi

Frequency, GHz

Figure 6-3 Computed values of maximum gain over the WCJ-Pole operating

band.

137

The proposed wideband antenna is a very good candidate for use in many

applications with future mobile phones, PDAs, and laptop computers because it offers a

single antenna solution.

6.3 Dual-Band Compact Antenna for Personal Wireless Communications, Bluetooth, and U-NII bands (DCLA #1)

The geometry of the proposed dual-band compact low-profile antenna (DCLA #1) is

shown in Figure 6-4. The structure is a WC J-pole except that the dimensions are

different and an extra parasitic element is added. Note that the feed plate is partially

covered by the top plate. The analysis in the previous chapter showed that a resonance

occurs in the 5-GHz region due to radiation of the feed plate portion unshielded by the

top plate. A parasitic rectangular plate is added to the structure at the same height as the

feed plate. This extra parasitic plate creates a resonance at the 5-GHz band.

Figure 6-5 depicts the computed and measured impedance of the antenna and shows

that there are two very wide bands with return loss below –10 dB (or 2:1 VSWR), one

from 1.83 GHz to 2.52 GHz (31.7 % bandwidth) and the other from 5.05 GHz to 5.39

GHz (6.5 % bandwidth). The lower band of the antenna can cover then the following

bands in use today: PCS-1900, IMT2000, UMTS, ISM, WLAN, and Bluetooth bands.

The other band of the antenna covers two of the three 5 GHz unlicensed national

information infrastructure (U-NII) bands, which spans from 5.15 GHz to 5.35 GHz.

138

Figure 6-4 Geometry of the dual-band compact antenna and its prototype

(DCLA #1).

1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6-30

-25

-20

-15

-10

-5

0

Frequency, GHz

|S11

|, dB

NumericalExperimental

Figure 6-5 Computed and measured |S11| of the dual-band compact antenna of

Fig. 6-4. A 10-dB return loss (-10 dB S11) corresponds to a 2:1

VSWR.

57.0

20.0

26.0

3.01.5

29.0 25.51.0

10.01.0

x

y

x

z

Dimensions in mm

Parasitic Plate

Upper Plate

Feed Plate

Lower Plate

Shorting Plate

SMA Connector

57.0

20.0

26.0

3.01.5

29.0 25.51.0

10.01.0

x

y

x

z

Dimensions in mm

57.0

20.0

26.0

3.01.5

29.0 25.51.0

10.01.0

x

y

x

z

Dimensions in mm

Parasitic Plate

Upper Plate

Feed Plate

Lower Plate

Shorting Plate

SMA Connector

139

Radiation patterns, gain, and radiation efficiency of the antenna were also measured

and are shown in Figures 6-6 to 6-8. The radiation patterns in Fig. 6-6 show an omni-

directional pattern behavior. Figure 6-7 shows that the gain is about 2.5 dBi in the 2-GHz

band and 6.5 dBi in the 5-GHz band. Figure 6-8 shows that the radiation efficiency in the

two bands is about 85 %. Omni-directional pattern, a gain above 2.5 dB, and a radiation

efficiency of 80 % are considered good performance for antennas used in handheld

wireless devices.

NumericalNumericalExperimentalExperimentalNumericalNumericalExperimentalExperimental

2.2 GHz2.2 GHz 5.2 GHz5.2 GHz

zz

yy

(a) (b)

Figure 6-6 Computed (solid curves) and measured (dashed curves) radiation

patterns of the dual-band compact antenna of Fig. 6-4 in the

frequency bands of interest in the yz plane for both Eθ (red curves)

and Eφ (blue curves) cuts: (a) 2.2 GHz and (b) 5.2 GHz.

xx

zzyy

xx

zzyy

140

NumericalNumericalExperimentalExperimentalNumericalNumericalExperimentalExperimental

Figure 6-7 Computed and measured gain of the dual-band compact antenna of

Fig. 6-4.

1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6 6.5 70

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

Frequency, GHz

Effi

cien

cy

Radiation efficiency including impedance mismatchRadiation efficiency

Figure 6-8 Measured radiation efficiency of the dual-band compact antenna of

Fig. 6-4 using the wideband Wheeler cap method described in

Chapter 3.

141

This dual-band, compact, low-profile antenna (DCLA) was developed to be

embedded in wireless fixed and mobile devices and is ready to be introduced into future

systems including third-generation personal radios, Bluetooth third-party, and wireless

LAN applications. The improvement of this antenna over other dual-band antennas is that

its two bands are very wide, while the size of the antenna remains small. In fact, this

antenna covers the current 2G- as well as the 3G-system frequency bands, from 1.83 to

2.52 GHz. This wide coverage of the antenna lower operating band can help the 2G-to-

3G personal radio transition go smoothly. In addition, the DCLA can also be used in the

5-GHz U-NII bands (5.15-5.25 GHz and 5.25-5.35 GHz bands), including the ISM band,

which is the useful band for the next generation of wireless LANs. Wireless devices that

cover both of these bands simultaneously are in commercial development.

6.4 Dual-Band Compact Antenna for 2.45/5.25 GHz WLAN (DCLA #2)

In this section, a variation of the previous dual-band antenna (DCLA #1) is presented for

WLAN applications. Figure 6-9 shows this WLAN dual-band version of antenna (DCLA

#2a). The dimensions of this promising design are 4.5 by 4.5 by 40 mm3. It is smaller

than the DCLA #1 shown in the previous section. This design was found from simulation

trials using IE3D method of moments code. The return loss of this simulation is show in

Fig. 6-10. However, during the experiment phase, a coax cable built to feed the antenna

because the antenna is too small to solder the SMA connector directly onto it. Therefore,

the simulation should include this coax cable in the design. Figure 6-11 shows the effect

of the coax cable that was built onto the antenna by comparing the measured return loss

of antenna with the coax cable and numerical return loss of the antenna excluding the

coax. Obviously the detuning effect due to the coax is not negligible.

142

40.0

4.5

4.5 x

y

x

z

Dimensions in mm

40.0

4.5

4.5 x

y

x

z

Dimensions in mm

Figure 6-9 Overall dimensions of the DCLA #2a structure for WLAN used in

IE3D simulation.

Frequency bands of interest (VSWR <2)Frequency bands of interest (VSWR <2)

Figure 6-10 Computed return loss of the DCLA #2 for WLAN shown in Fig. 6-9.

143

2 2.5 3 3.5 4 4.5 5 5.5 6-25

-20

-15

-10

-5

0

Frequency, GHz

S11

, dB

IE3D SimulationMeasurements Antenna detuned at bands of interest

2 2.5 3 3.5 4 4.5 5 5.5 6-25

-20

-15

-10

-5

0

Frequency, GHz

S11

, dB

IE3D SimulationMeasurements Antenna detuned at bands of interest

Figure 6-11 Comparison of the measured |S11| values of the DCLA #2a built

with a coax cable and the numerical results of DCLA #2a without

the coax simulated using IE3D.

The antenna was simulated using a second code, an FDTD simulation software

package, Fidelity, for the model including the coax cable (DCLA #2b) and was tuned to

compensate the detuning effect due to the coax. Figure 6-12 shows the dimensions of the

antenna with the coax included and Fig. 6-13 shows the measured and numerical return

loss of the antenna with the coax attached. The reasons to go through this process of

adjusting the structure of this antenna are to show the confidence between the numerical

and measured values of return loss and that the antenna can be easily adjusted to adapt to

the environment.

144

Figure 6-12 Dimensions of the DCLA structure for WLAN (DCLA #2b)

including the coax cable and tuned to compensate the detuning

coupling effects due to the coax.

40.0

4.5

4.5 x

y

x

z

Dimensions in mm

Coax Cable

13.58.0

15.5

15.0

3.5

15.0

145

2 2.5 3 3.5 4 4.5 5 5.5 6-50-45-40-35-30-25-20-15

-10-50

Frequency, GHz

S 11, d

B

NumericalExperimental

2 2.5 3 3.5 4 4.5 5 5.5 6-50-45-40-35-30-25-20-15

-10-50

Frequency, GHz

S 11, d

B

NumericalExperimental

Figure 6-13 Numerical and measured return loss of the dual-band antenna

shown in Figure 6-12 with the coax cable attached.

The DCLA #2 is a potential candidate for WiFi applications such as laptop

computers. Most of the next generation laptop computers will have WiFi capability

embedded. Because of its size and shape, the DCLA #2 can be easily embedded into the

laptop computer. Figure 6-14 illustrates a good antenna placement into the laptop. This

configuration enables spatial diversity that will improve signal reception and

communication performance.

146

Antenna placement with diversity capabilityAntenna placement with diversity capability

Figure 6-14 Example of antenna placement for the DCLA #2 in laptop

computers that are WiFi enabled.

6.5 Summary

A few versions of the WC J-Pole antenna were presented for some personal wireless

commercial applications. The antenna structure dimensions were adjusted to meet

specific commercial operating bands. One design is a wideband antenna that covers an

operating band ranging from 1525 to 2515 MHz. It can be used for wireless systems that

need GPS, personal communication service, WLAN, and third-party device interaction,

all in one device. The other two designs are dual-band antenna. One is for WLAN-

specific systems using both 2.4 and 5-GHz and the other has a large operating band from

1.83 to 2.52 GHz and another band from 5.05 to 5.39 GHz. Numerical and experimental

results were in good agreement.

147

Chapter 7

Conclusions 7.1 Summary

A review of previous work related to small antennas was presented in Chapters 1 and 2.

Chapter 2 reviewed conventional small antennas and how antenna size relates to

performance including its impedance bandwidth and efficiency. This relation was

detailed in the theory of fundamental limit on small antennas. Conventional solutions

such as short monopole, stubby antennas, and microstrip antennas may be inadequate for

handheld wireless devices of the current trend. Therefore it is imperative to reduce them

further in size. A few techniques of size reduction were also presented, including:

material loading, the use of short circuit and ground plane, geometry optimization such as

meandered line configuration, and the use of the environment as radiating element. These

techniques have to be accompanied by ways of enhancing impedance bandwidth and

efficiency, as detailed in Chapter 2, because reducing antenna size degrades the antenna

characteristics, according to the fundamental limit theory.

Chapter 3 provides a development of a technique for measuring antenna radiation

efficiency based upon the Wheeler cap method. It is a modification of the traditional

Wheeler method for wideband and larger antennas. It is based on the link budget of

transmit power from the antenna and subsequent reflected power from the spherical

conducting cap. The difference between the traditional method and this proposed

technique is in the size of the cap enclosing the antenna. The traditional method uses a

148

cap of radius λ/2π that inhibits radiation whereas this proposed technique uses a larger

cap so that radiation is allowed with the cap.

A review of biological effects of radio-frequency radiation, and radiation safety and

regulations was presented in Chapter 4. The interaction between handset antennas and

biological tissues quantified by specific absorption rate (SAR) was explained.

Understanding how operator presence affects antenna performance and the associated

health hazards aids engineers to develop handheld wireless products that minimize

operator exposure and improve antenna radiation efficiency.

A new wideband compact antenna was introduced in Chapter 5. Detailed analyses

including parametric study were performed to understand the fundamental radiation

mechanism of the antenna. It was found that the proposed antenna is similar to a J-Pole

antenna except that it is extended to a planar configuration and is fed capacitively. With

the help of the parametric study, the antenna impedance bandwidth was optimized to

50%. This wideband compact J-Pole antenna has omnidirectional radiation pattern over

the entire band, a peak average gain of 2.5 dBi, and radiation efficiency above 90 %. It is,

therefore, suitable for small handheld wireless devices.

Finally, a few designs of the wideband compact J-pole antenna analyzed in Chapter

5 were shown in Chapter 6 for specific wireless commercial applications. These include a

wideband compact antennas that can be operated from GPS band up to the unlicensed

band at 2.4 GHz, a dual-band compact antenna for personal wireless communications,

Bluetooth, and the 5-GHz unlicensed band, and a dual-band compact antenna for

2.45/5.25 GHz WLAN.

7.2 Contributions

The major contribution of this dissertation is in the analysis of a new wideband, compact,

and low-profile antenna for handheld wireless devices that led to a patent [1]. A detailed

analysis of this new antenna, called the wideband compact J-Pole antenna (WC J-Pole),

enabled the optimization of its impedance bandwidth to reach 50 %; see Fig. 5-29. This

small antenna can be considered to have one of the largest bandwidths among other small

resonant antennas intended for commercial use. Two other versions of the WC J-pole

149

were developed and can be operated in the 5 GHz frequency range. The three antennas

characteristics and performance are summarized in Table 7.1

Another major contribution of this research described in Chapter 3 was the radiation

efficiency measurement for moderate-length and wideband antennas. This method can be

further extended to measuring radiation efficiency of a whole wireless device, such as

measuring the efficiency of an antenna that is embedded in a handheld unit. This

measurement method was demonstrated by measuring a few antennas and comparing the

experimental and numerical results.

Table 7-1 List of the three versions of the WC J-pole antenna with their

characteristics

WCJP

(Sec. 6.2)

DCLA #1

(Sec. 6.3)

DCLA #2

(Sec. 6.4)

Overall

dimensions 62x10x6 mm 57x10x3 mm 40x40x4.5 mm

Bandwidth

(MHz)

for VSWR <2

1525-2515 MHz

(990 MHz, 49%)

1830-2520 MHz

(690 MHz, 31.7 %)

5050-5390 MHz

(340 MHz, 6.5 %)

2370-2530 MHz

(160 MHz, 6.5 %)

5000-5600 MHz

(600 MHz, 11.3 %)

Peak Gain (dB) ~2.5 dBi ~2.5 dBi (lower band)

~6.5 di (upper band)

~2.5 dBi (lower band)

~6.5 di (upper band)

Radiation

Efficiency ~85 % ~85 % (both bands) ~ 85 % (both bands)

Radiation Pattern Omnidirectional Omnidirectional Omnidirectional

7.3 Future Work

The research in this dissertation can be extended by investigating the possibility of

further reducing the size of the WC J-Pole antenna. The antenna is not loaded with any

dielectric material. Since its radiation efficiency and impedance bandwidth values are

150

large, the antenna could give up some bandwidth in order to achieve smaller size and still

be useful for many applications. Furthermore, adding dielectric material can enhance its

mechanical integrity, help the manufacturing process, and help integrate the antenna

directly into the system PC board.

It was mentioned that the WC J-pole antenna behaves like a dipole antenna.

Therefore, it was assumed that the effects of human operator on the WC J-pole are

similar to those of a dipole detailed in Chapter 4. However, thorough study of these

effects on the WC J-pole was not performed. Furthermore, SAR results for the WC J-pole

were not investigated. These analyses on the interaction between the human operator and

the antenna are important when integrating the WC J-pole in the wireless system device.

It is therefore an important topic that needs further investigations.

7.4 Reference

[1] M-C. Huynh and W.L. Stutzman, “The Wideband Compact PIFA,” U.S. Patent

No. 6,795,028.

151

Vita Minh-Chau T. Huynh was born in Saigon, Vietnam, and grew up in Liège, Belgium.

Growing up in a multicultural milieu, he learned and speaks fluently four languages:

Vietnamese, French, English, and Spanish. In 1993, he began studying electrical

engineering at Virginia Tech. He completed his BS, MS, and PhD in 1997, 2000, and

2004, respectively, all from Virginia Tech.

In December 1998, Minh-Chau joined Virginia Tech Antenna Group as a research

assistant. His research interests include wideband low-profile compact antennas,

ultrawideband antennas, and biological effects of RF on human operators.