Wearable Devices With Low SAR

3808 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 63, NO. 9, SEPTEMBER 2015

A Compact, Wideband Circularly PolarizedCo-designed Filtering Antenna and Its Application

for Wearable Devices With Low SARZhi Hao Jiang, Member, IEEE, and Douglas H. Werner, Fellow, IEEE

Abstract—A compact circularly polarized (CP) co-designed fil-tering antenna is reported. The device is based on a patch radiatorseamlessly integrated with a bandpass filter composed of coupledstripline open-loop resonators, which are designed together as asystem. In the proposed design, the patch functions simultaneouslyas the radiator and the last stage resonator of the filter, resulting ina low-profile integrated radiating and filtering module with a smalloverall form factor of 0.53λ0 × 0.53λ0 × 0.07λ0. It is shownthat the filtering circuit not only ensures frequency selectivitybut also provides impedance matching functionality, which servesto broaden both the impedance and axial ratio bandwidths. Thedesigned filtering antenna was fabricated and measured, exper-imentally achieving an S11 < −13.5 dB, an axial ratio of lessthan 3 dB and a gain higher than 5.2 dBi over a bandwidth from3.77 to 4.26 GHz, i.e., around 12.2%, which makes it an excel-lent candidate for integration into a variety of wireless systems. Alinearly polarized version of the integrated filtering antenna wasalso demonstrated. In addition, further full-wave simulations andexperiments were carried out to verify that the designed CP fil-tering antenna maintains its properties even when mounted ondifferent positions of the human body with various body gestures.The stable impedance and radiation properties also make it asuitable candidate as a wearable antenna for off-body wirelesscommunications.

Index Terms—Bandpass filter (BPF), circularly-polarized cou-pled resonator, filtering antenna, wearable antenna.

I. INTRODUCTION

A S WIRELESS communication technology continues itsrapid pace of development, the integration of multiple

microwave/RF components has become ever more demanding,especially those which provide size reduction and improvedoverall system efficiency [1]. The antenna and bandpass filter(BPF) are the two most essential components at the front-endof a typical RF system for wireless communications. Whilethe antenna is responsible for receiving and transmitting sig-nals, the BPF is crucial for distinguishing the desired signalfrom extraneous signals outside the targeted band(s) of interest[2], [3]. Conventionally, the BPF and the antenna are designed

Manuscript received December 31, 2014; revised June 23, 2015; acceptedJune 24, 2015. Date of publication July 06, 2015; date of current versionSeptember 01, 2015. This work was supported by the National ScienceFoundation ASSIST Nanosystems ERC under Award EEC-1160483.

The authors are with the Department of Electrical Engineering, ThePennsylvania State University, University Park, PA 16802 USA (e-mail:[email protected]; [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TAP.2015.2452942

separately and then directly connected by sharing a commonreference impedance of 50 or 75 Ω. Since the antenna and theresonators of the BPF resonate at the same frequency, inter-ference between them may occur which affects the return lossand the antenna gain response. In addition, further performancedegradation may be experienced due to the fact that the inputimpedance of the antenna may not be perfectly matched to apure resistance value, e.g., 50 or 75 Ω, over a finite bandwidthand especially at the band edges.

Recently, efforts have been made to co-design the BPF andantenna as a single module, i.e., a filtering antenna, which pro-vides an alternative and more attractive solution [4]. So far,several filtering antennas have been proposed by integratingtogether a variety of different filter and antenna structures. Insome of the reported works, the antenna is treated as a disper-sive complex load of the filter, including frequency selectivesurface (FSS) coverings for horn antennas [5], [6] and wiremonopoles [7], coupled planar resonator filter connected patchantennas [8], [9], and coupled substrate-integrated waveguide(SIW) resonator filter connected planar coaxial collinear anten-nas [10]. In other filtering antennas, the antenna element isutilized as the radiator and as the last resonator of the BPFsimultaneously, including coupled rectangular waveguide res-onator filter backed electromagnetic bandgao antennas [4],coupled SIW cavity filters cascaded behind slot antennas [11]–[13], and planar monopole antennas combined with varioustypes of coupled-line filters [14]–[16]. These demonstrationsshow that co-designed filtering antennas indeed have smalleroverall device volume and superior band selectivity. However,the filtering antennas demonstrated to date still suffer from largefootprints, high profiles, and are limited to linear polarization,making them not suitable for certain newly emerging appli-cations such as antennas for compact indoor base stations orportable/wearable devices. In addition, most of these planar fil-tering antennas possess near-omnidirectional radiation patternsdue to the employed monopole elements [7], [10], [14], [16]and/or require fabrication of complicated three-dimensional(3-D) structures [4]–[6], [10], [11]–[13].

As one of the nascent but fast growing fields in theantenna community, wearable antennas have garnered world-wide research interest due to their applications in wirelesssystems for body area networks, which hold great promisefor future medical applications, battlefield survival, and patienttracking systems [17], [18]. Since wearable antennas oper-ate in close proximity to the human body, the mutual impact

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JIANG AND WERNER: COMPACT WIDEBAND CP CO-DESIGNED FILTERING ANTENNA AND ITS APPLICATION 3809

between the antenna and human tissue makes the design ofhighly efficient wearable antennas with low specific absorp-tion rate (SAR) a challenging task. So far, various types ofwearable antennas have been proposed, such as the early workon vertical monopole [19] and inverted-F antennas [20], whichare not low profile or conformal. Planar microstrip monopoles[21]–[23], planar inverted-F antennas [24], [25], slot anten-nas [26], and stepped impedance resonator antennas [27] havesmall form factors but undesirably radiate a significant amountof power into the human body, resulting in low efficiency.Hence, they are usually positioned at a certain distance awayfrom the human body. Patch antennas [28] and cavity-backedslot antennas [29], [30] are appropriate for off-body commu-nications because of their broadside radiation patterns, butare limited by their narrow operational bandwidth. Isotropicartificial magnetic conducting (AMC) surfaces [31]–[33] andstrongly truncated anisotropic metasurfaces [34] have recentlybeen employed to achieve a high degree of isolation betweenthe antenna and human tissue while maintaining a small over-all profile. However, nearly all of these wearable antennas arelinearly polarized (LP), which may lead to unreliable wire-less links due to constantly changing human body movement.In the most severe case, a transmission null can occur due tocomplete polarization mismatch, resulting in an entirely lostlink. Circularly polarized (CP) antennas, on the other hand, aremore robust to polarization mismatch caused by human bodymotion, albeit at the expense of a ∼3 dB loss, but have been lessexploited for wearable applications. Single pin-fed CP wear-able antennas have been reported in [35] and [36]. However, the3-dB axial ratio (AR) bandwidth is very narrow, i.e., less than2.5%, making them of limited use for high-data-rate body-areanetwork systems.

In this paper, we address these two major challenges (i.e.,narrow operating bandwidth and linear polarization) by intro-ducing a new design methodology for realizing compact, co-designed CP filtering antennas, all achieved without increasingthe antenna form factor. In Section II, we describe the co-designprocess of the filtering antenna by illustrating the layout, equiv-alent circuit, structural design, and optimization of the module.A LP filtering antenna is also compared to the same patchradiator without a BPF and with a directly connected BPF.Section III gives the measured results in free space. The numer-ical and experimental investigations of its wearable applicationunder various realistic scenarios are provided in Section IV,followed by the conclusion in Section V.

II. FILTERING ANTENNA DESIGN

A. Topology of the CP Filtering Antenna

The configuration of the co-designed CP filtering antennais illustrated in Fig. 1, which consists of a square patchradiator with truncated corners on the top and a planar striplinemicrowave circuit on the bottom, including the filtering, phaseshifting, power dividing, and impedance matching sections,required to generate a left-hand CP (LHCP) radiated wave.Considering that wearable antennas operate in close proximityto the human body, the stripline configuration, which isolates

Fig. 1. (a) Three-dimensional (3-D) tilted view of the proposed co-designedCP filtering antenna using SOLRs with a stripline feed. (b) Side view ofthe antenna structure. The thickness values of the three substrate layers areh1 = 3.68, h2 = 0.79, h3 = 0.79, all in millimeters. All the substrates areRogers RT/duroid 5880 with εr = 2.2 and δtan = 0.0009. The copper layersare 18-µm thick.

the BPF resonators from the effects of human tissue loading,was adopted rather than the more commonly used microstripcircuits. Two BPFs were employed, each connecting to thepatch through a coupled stripline and a metallic pin that passesthrough the top ground plane of the stripline circuit. The feed-ing positions of the two pins were aligned on the centerlines ofsymmetry along the x- and y-axis of the patch but offset fromthe center. The top ground plane of the filter also serves as thebottom ground plane of the patch radiator. The BPF circuitconsists of three edge-coupled stripline open-loop resonators(SOLRs) [37]. The filtering antenna was fed by an SMAconnector from the side, such that the antenna can be placedvery close to the surface of a human body compared to thebottom-fed topologies which have been considered previouslyin the literature for wearable applications [35], [36]. The formfactor of the filtering antenna is 40mm× 40mm× 5.3mm(0.53λ0 × 0.53λ0 × 0.07λ0, where λ0 = 75mm), which ismore compact than many previously demonstrated filteringantennas [4]–[16]. The methodology presented here can alsobe used to design compact CP filtering antennas that operate inother frequency bands.

B. Equivalent Circuit Model

The equivalent circuit for each branch of the integrated CPfiltering antenna is shown in Fig. 2(a). A parallel RLC circuit(Rx, Lx, Cx or Ry, Ly , Cy) was used to model the orthogonalmode in the x- or y-direction, respectively. A series inductor(Lp) was employed to represent the inductance of the feedingpin. The three parallel RLC resonators represent the three


Fig. 2. (a) Original and (b) simplified equivalent circuit model for each branchof the CP filtering antenna.

SOLRs, which can be either synchronously or asynchronouslytuned. The coupling between them was modeled as admit-tance inverters (Jij , i �= j) [38]. The coupled stripline thatcapacitively couples the third resonator to the patch radiatorin each filter branch is represented as an admittance inverterwith a section of transmission line on each side [39]. At thecenter frequency, the equivalent circuit model can be furthersimplified to the one shown in Fig. 2(b), by incorporating theseries inductor Lp into the capacitor of the third resonator. Inthe original circuit

Y3 =J2341

jωLp+1/Yant

= jωLpJ234 +

J234

Yant= jωCm +

J234

Yant.

(1)

The shunt capacitor of the third resonator in the simplifiedcircuit model can thus be modified to be C ′

3 = C3 + Cm. Thecircuit in Fig. 2(b) can then be used directly to synthesize a typi-cal Chebyshev BPF, in which the antenna becomes the last stageresonator of the filter, in addition to its role as the radiator. Theinput impedance of each BPF filter was set to match a certainpurely resistive impedance of ZBPFi. The two branches wereconnected in parallel at a T-junction with a 90◦ phase differencebetween them to achieve circular polarization. A quarter-wavetransformer was used to match the input impedance at the T-junction to 50 Ω at the antenna port. It should be noted that theBPF circuit can also be considered as a lossless two port net-work that greatly enhances the bandwidth of the conventionalpatch antenna [40], which provides simultaneous filtering andimpedance matching functionalities.

C. Physical Design

The target was a filtering antenna having a fourth-orderChebyshev equal-ripple broadside antenna gain response, witha center frequency at 4 GHz, a fractional bandwidth of 12.5%,and a 50-Ω port impedance. Within this 12.5% bandwidth, theAR should remain below 3 dB. We first designed the LP filter-ing antenna with the same spectrum requirement, which willthen be used to construct the final CP filtering antenna. Byutilizing filter design tables in the literature [2], [3], the nor-malized Chebyshev low-pass filter prototype element valueswith a ripple level of 0.2 dB were found to be g0 = 1, g1 =1.303, g2 = 1.284, g3 = 1.976, g4 = 0.847, and g5 = 1.539.The resulting values for parameter extraction are Qei = Qeo =10.423, M12 = M34 = 0.097, and M23 = 0.078. The valueof ZBPFi, i.e., Z2, was set to be 80 Ω. Further considering

Fig. 3. (a) Top view of the patch radiator layer. The dimensions of the ini-tial design are p = 22.3, c = 1, m = 5, and GND = 40, all in millimeters.(b) Input impedance (ZANT ) response of the pin fed CP patch obtainedfrom the full-wave simulation and equivalent circuit model. The correspondinglumped circuit element values are Rx = 98 Ω, Cx = 4.6 pF, Lx = 0.34 nH,and Lp = 1.95 nH.

the dispersive properties of the transmission sections with anelectrical length of a quarter wavelength at the center frequencyand a characteristic impedance of Zc = 80 Ω due to the cou-pled stripline, the coupling parameters and the external qualityfactors were modified to be Qei = 9.28, Qeo = 10.02, M12 =0.106, M23 = 0.086, and M34 = 0.107 to provide the desiredfiltering performance.

The detailed geometry of the pin-fed LP radiating patch isdepicted in Fig. 3(a). The Ansoft high-frequency structure sim-ulator (HFSS) was used to simulate the pin-fed patch antennaalone. By adjusting the edge length of the patch, the cuttingsize of the corners, the thickness of the top substrate layer,and the position of the feeding pin, the input resistance andquality factor of the patch, i.e., Qeo, can be designed to meetthe target value. The simulated input impedance is shown inFig. 3(b), with the corresponding patch dimensions listed in thefigure caption. The circuit parameters were then extracted byfitting the input impedances (ZANT ) of the radiator’s equiv-alent circuit consisting of a series inductor (Lp) connectedparallel RLC circuit. The calculated lumped element values areLp = 1.95 nH, Rx = 98 Ω, Lx = 0.34 nH, and Cx = 4.6 pF.The impedance calculated using the equivalent circuit modelagrees well with the full-wave result in the 3.5–4.5 GHz fre-quency range, which encompasses the targeted passband. Itshould be noted that the ground plane size was chosen to pro-vide a reasonably high FB ratio and a low SAR value forwearable applications, while maintaining a compact footprintthat covers the stripline circuit structure.

The layout and dimensional parameters of the BPF res-onators and the coupled stripline are illustrated in Fig. 4. Thephysical dimensions of the coupled resonators were determinedbased on the design curves illustrated in Fig. 5. The resonantfrequency of the SOLRs was calculated using the HFSS eigen-mode solver. The resonant frequency as a function of the width(br) of the SOLRs, while the gap and length values were fixed,is plotted in Fig. 5(a). The SOLRs with a slightly differentvalue of ar all have a similar quality factor of around 600.The external quality factor can be controlled by properly posi-tioning a tapped stripline directly connecting to the SOLR at adistance t away from the center, as shown in Fig. 5(b). The cou-pling parameter between two adjacent SOLRs as a function oftheir distance can be found in Fig. 5(c). The coupled striplineshares one of its transmission line sections with the third SOLR.


Fig. 4. Top view of the stripline BPF layer. The dimensions of the initialdesign are ar1 = ar2 = ar3 = 7, br = 7.6, sr = 1.5, wr = 0.5, s12 =0.38, s23 = 0.52, ss = 0.21, ws = 0.5, ls1 = 3.5, ls2 = 9, ls3 = 3.7,t = 3.7, la1 = 2, la2 = 9.4, la3 = 3.4, la4 = 2.4, la5 = 7.1, lb1 = 2,lb2 = 10.5, lb3 = 1.1, lf2 = 11.2, lf3 = 2, wf1 = 1.35, wf2 = 1.47,and wf3 = 1.73, all in millimeters.

Fig. 5. (a) Resonant frequency of an isolated SOLR with br = 7.6mm, sr =1.5mm, wr = 0.5mm, where ar is the tuning variable. (b) Extracted externalquality factor. The SOLR is the same as that used in (a) with ar = 6.6mm,while t is the tuning variable. (c) Coupling parameter between resonators forvarying s.

The length of the coupled stripline should be around a quarterwavelength at the center frequency, which suggests that a goodchoice for its value is 12.6 mm. Furthermore, the separationdistance (s) was determined to be 0.21 mm, by following theequations provided in [39].

A CP filtering antenna with a wideband AR bandwidth canbe constructed using two LP filtering antennas, each connectedto a branch of the T-junction with a stripline. The two striplineshave a length difference of around 12.6 mm, which is a quarterwavelength in the substrate at the center frequency. A quarter-wave stripline is good enough to provide the desired 90◦ phaseshift over the 12.5% bandwidth with a phase deviation of lessthan ±7◦. The longer stripline was applied to the pin feedthat excites the resonant mode of the patch in the x-directionto generate LHCP radiation. Within the passband, the inputimpedance at the T-junction is around 40 Ω, which can be read-ily matched to 50 Ω using a quarter-wave transformer with acharacteristic impedance of 44.72 Ω.

D. CMA-ES Optimization of CP Filtering Antenna

An initial integrated CP filtering antenna design was obtainedby choosing the geometrical parameters for both the patch radi-ator and the BPF structure from the design curves. However,when the two BPF branches and additional phase shift-ing and impedance matching striplines were incorporated,mutual coupling among different waveguiding or resonatingcomponents becomes more complicated, which would causeminor frequency shifts and/or coupling parameter variations.A MATLAB-based covariance matrix adaptation evolutionarystrategy (CMA-ES) [41], [42] optimization code was employedand coupled with HFSS to fine tune the geometrical dimensionsfor the integrated CP filtering antenna module. The CMA-ESmethod is well suited to this type of design problem, sinceit is a real-valued optimization strategy and has been shownto be effective and efficient for optimization of designs witha large number of parameters [42]. Its operation proceeds byreshaping and moving a multivariate Normal search distribu-tion within the parameter space. The distribution aligns itself totraverse along the contour of decreasing cost, thereby enlargingor contracting during each iteration to reduce the total numberof function calls required to find the optimal cost value. Theimportant dimensional parameters to be tuned include the sizesof all SOLRs (ar1, ar2, ar3, br), the separation distances inbetween adjacent SOLRs (s12, s23), the edge length of the patchradiator (p), the length of the cutting corner of the patch (c), thefeeding position of the pins (m), the position of the tappingstripline (t), the length (ls1, ls2, ls3), width (ws), and spacing(ss) of the coupled stripline traces, as well as the length differ-ence (la1, la2, la3, la4, la5, lb1, lb2, lb3) of the two striplines con-necting each BPF branch to the T-junction. The dynamic rangeof each parameter was set to be ±20% away from the value ofthe initial design. A small population of only 16 was used con-sidering that the recommended population size should be largerthan 4 + 3 ln(N ), where N is the number of parameters to beoptimized [41].

The optimization goal was to achieve S11 below −15 dB,AR smaller than 3 dB, and gain higher than 6 dBi within thetargeted 3.75–4.25-GHz passband, and stop bands elsewhere.Specifically, eight frequency samples, denoted as fpbi, wereselected in the passband and eight frequency samples, denotedas fsbi, were considered in the stopbands outside the passband.A cost function was then defined as


Fig. 6. Simulated (a) S11, (b) gain, and AR of the originally designed andoptimized CP filtering antennas in free space. The dimensions of the opti-mized design are p = 22.5, c = 1.03, m = 4.75, ar1 = 6.88, ar2 = 7.16,ar3 = 7.22, br = 7.6, sr = 1.5, wr = 0.5, s12 = 0.38, s23 = 0.49, ss =0.19, ws = 0.4, ls1 = 3.68, ls2 = 8.78, ls3 = 3.78, t = 4.39, la1 = 2.1,la2 = 8.6, la3 = 4.5, la4 = 3, la5 = 7.84, lb1 = 2.5, lb2 = 8.9, lb3 = 2.3,lf2 = 11.25, lf3 = 2, wf1 = 1.35, wf2 = 1.45, and wf3 = 1.73, all inmillimeters.

Fcost =∑

fpbi

[max (0, |S11| − 0.18)

+max (0, |ARdb| − 3) + max (0, 6− |Gaindb|)]+∑

fsbi

(max (0, 0.8− |S11|)) . (2)

For each design candidate, MATLAB called HFSS to per-form the full-wave simulation of the antenna using two 2.6-GHzprocessors each having 12 cores. The simulation time for eachcandidate was about 45 min. After 20 generations, which tookabout 10 days, the CMA-ES converged to an optimal designthat satisfied all the goals.

E. Optimized CP Filtering Antenna Results

The S11 of the initial and optimized designs are shownin Fig. 6(a), where the optimized design has an S11 below−14.5 dB and a 12.5% fractional bandwidth in the pass-band centered at 4 GHz, with a sharp roll-off of 100.5 and105.2 dB/GHz on the band edges. The S11 calculated usingthe full circuit model, including the two LP filtering antennabranches, phase shifter, power dividing and matching circuits,is also presented in Fig. 6(a), which agrees well with the full-wave simulation result. The filtering antenna has a broadsidegain profile [see Fig. 6(b)] similar to the transmission of theChebyshev BPF with a peak total broadside gain of 6.8 dBi andin-band gain variation of 0.8 dB. The simulated average radia-tion efficiency in the passband was around 80%. The nonperfectradiation efficiency is primarily due to the insertion loss of theadded BPF and the stripline feeding network. It should be notedthat the out-of-band rejection of this filtering antenna is slightly

Fig. 7. Simulated S11 of the LP filtering antenna, the LP patch alone, and theLP patch directly connected to a separately designed BPF.

lower than that of a fourth-order Chebyshev BPF, due to the factthat the intrinsic gain response of the patch alone has relativelypoor frequency selectivity. Therefore, to be more accurate, thesynthesized filtering antenna module has a quasi-fourth-orderChebyshev response. The broadside AR displayed in Fig. 6(b)also has a window from 3.76 to 4.24 GHz, i.e., a fractionalbandwidth of 12.4%, within which the value is below 3 dB.Compared to the optimized design, the S11 < −15 dB bandfor the initial design is narrower and shifted to the frequencyrange from 3.87 to 4.28 GHz. Correspondingly, the AR bandof the initial design also becomes narrower and shifts towardhigher frequencies. Its value is also higher than 3 dB in the cen-ter portion of the targeted band, due to the in-band impedancemismatch in each LP filtering antenna branch. The mismatchedinput impedances of the two branches undergo transmissionlines with an electrical length difference of around a quarterwavelength, thereby causing unequally split power transmit-ted to the two branches at the T-junction, which leads to theundesirably high AR values in the targeted band. It should benoted that wideband CP antennas with broader bandwidth canbe achieved using other techniques such as by employing athick air substrate [43], dual feeding with a planar hybrid cou-pler [44], incorporating broadband 90◦ baluns [45], as well asother methods [46], but all require a larger footprint or thick-ness. Integrating filtering functionalities into them will furtherincrease the antenna size, thus such approaches would not becompact enough for certain applications.

F. LP Filtering Antenna and Related Comparison

As a constitutive part of the CP filtering antenna, the LPfiltering antenna was compared with the LP patch alone, andthe LP patch with a directly connected BPF that was sepa-rately designed. For all three antennas, the same patch wasused, ensuring the same quality factor for the radiating element.As shown in Fig. 7, the integrated filtering antenna exhibits asimulated S11 below −15 dB within the frequency range from3.74 to 4.26 GHz, whereas the S11 < −15 dB bandwidth of thepatch radiator alone, which only provides first order filtering, ismuch narrower (i.e., from 3.96 to 4.08 GHz). A third antennawas also considered that follows the conventional approachwhere a separately designed well-performing BPF consistingof four SOLRs was connected to the patch via a short section of50 Ω transmission line. In this case, a wide bandpass windowcan be identified in the frequency range of interest. However,the antenna input impedance is not maintained at 50 Ω when


Fig. 8. Photographs of the stripline circuit and the final module of (a) the LPand (b) CP filtering antennas.

there is a deviation from the center frequency, while the twoports of the BPF still maintain 50 Ω terminations. This mis-match results in band-edge selectivity degradation, especiallyat lower frequencies. It also leads to increased in-band S11,exhibiting a bump at around 4.17 GHz with a maximum valueof −7.9 dB. It should be noted that, in addition to the degradedperformance, the footprint of the antenna is also increased byabout 15% to accommodate the four SOLR BPF circuits. Thiscomparison illustrates the advantage of the co-design approachfor synthesizing filtering antennas, which provides a compactand integrated solution with superior performance comparedto those obtained by conventional methods. Moreover, it canbe seen that the coupled SOLR circuit not only provides thefiltering functionality to block unwanted signals outside the tar-geted band, but also serves as a reactive matching network thatgreatly broadens the S11 < −15 dB impedance bandwidth ofthe antenna by around 333% [40]. It should be noted that, thisachieved bandwidth is still well within the Chu limit on antennabandwidth [47].

III. FREE-SPACE MEASUREMENT RESULTS

Both the LP and CP integrated filtering antennas were fab-ricated by a standard printed circuit board etching method andthen assembled to form the final modules. Eight nylon screwswere used on the outer periphery of the filtering antennas toachieve a tight bonding between the layers. An SMA connec-tor was soldered to the stripline feed and the top and bottomground layers. Fig. 8(a) and (b) shows photographs of the fab-ricated stripline BPF structures and the final assembled LP andCP antenna modules, respectively.

The impedance of the filtering antennas was measured usingan Agilent E8364B network analyzer. As shown in Fig. 9(a),the measured S11 of the LP filtering antenna exhibits good fil-tering properties with S11 < −14 dB from 3.77 to 4.24 GHz.The minor difference in the simulated and measured in-bandS11 and slightly decreased roll-offs are attributed to fabricationand assembly inaccuracies. The measured broadside gain dis-played in Fig. 9(a) has a flat passband with values higher than

Fig. 9. Simulated and measured S11 and gain of the (a) LP and (b) CP filteringantennas. (c) Simulated and measured AR of the CP filtering antenna.

5.3 dBi in the band from 3.78 to 4.22 GHz and high rejectionelsewhere. The gain in the passband is on average about 1.1 dBlower than the value predicted in simulation primarily due tothe loss caused by the relatively low conductivity solder usednear the SMA and the vertical pin.

The measured S11 of the CP filtering antenna is presented inFig. 9(b), which also shows good filtering properties with S11 <−13.5 dB from 3.76 to 4.30 GHz, although slightly shifted tohigher frequencies as well as broadened. Such a minor blueshift and band broadening can also be observed in the measuredbroadside gain response [see Fig. 9(b)], which possesses a flatpassband with values higher than 5.2 dBi in the band from 3.77to 4.33 GHz and high rejection elsewhere. The gain in the pass-band is on an average about 1.2 dB lower than the simulatedresults. The measured broadside AR, as shown in Fig. 9(c), isbelow 3 dB from 3.77 to 4.26 GHz. The normalized LHCPand RHCP far-field patterns in both the x–z and y–z planeswere measured in an anechoic chamber. The patterns at 3.8,4.0, and 4.2 GHz are displayed in Fig. 10, which correspondwell to simulation predictions with only slight discrepancies.High cross-polarization discrimination is obtained within awide angular range in both planes. The main beam is slightlyshifted by 3◦ – 4◦ from broadside due to the offset pin feedswith respect to the center of the patch and the offset patch posi-tion away from the center of the ground plane. They have nearlyequal half power beam widths (HPBWs) of about 83◦ and 84◦

in the x–z and y–z planes, respectively, and front-to-back ratiosthat vary from 18 to 22 dB. The wide HPBW is favorable


Fig. 10. Simulated and measured normalized LHCP and RHCP radiation pat-terns of the CP filtering antenna at (a) 3.8 GHz, (b) 4 GHz, and (c) 4.2 GHzin both the x–z and y–z planes (black dashed line: simulated LHCP; blacksolid line: measured LHCP; blue dashed line: simulated RHCP; blue solid line:measured RHCP).

for off-body communication due to its wide angular coverage.Above all, the measurements confirmed a fairly high-fidelityimplementation of the proposed LP and CP integrated filter-ing antenna designs. To compare with the previously reportedfiltering antennas, the antenna properties including footprint,antenna profile, S11 bandwidth, polarization, and pattern typeare listed in Table I.

IV. WEARABLE APPLICATION OF THE CP FILTERING

ANTENNA

While the proposed compact CP filtering antenna can be agood candidate for various wireless communication systems,its suitability for wearable devices to enable off-body commu-nications is further investigated by evaluating its performancenumerically and experimentally when it is mounted directly ona human body.

A. Impact of the Human Body on Antenna Performance

The impact of the human body on the input impedanceand the radiation properties of the CP filtering antenna were

studied next. A homogenous full-scale human body model wasemployed which had a height of 174 cm and a chest width of46 cm [48]. In the HFSS simulation domain, the human bodymodel was assigned as an integral equation (IE) region, whichwas solved by the method of moments, whereas the filteringantenna was still assigned to the finite element region. The per-mittivity of the homogeneous human body model was chosen tobe two-thirds of the permittivity of muscle, which represents areasonable approximation as proven in the literature [17], [49].Such a homogeneous model allows for a numerically econom-ical way to provide a fairly accurate evaluation of both theimpedance and radiation properties of wearable antennas, butnot the SAR values [23].

As shown in Fig. 11, the proposed filtering antenna wasplaced at two different locations on the body, including thearm and the chest with an antenna to body distance of approx-imately 2 mm. For the on-arm case, the antenna is positionedsuch that the patch is offset toward the bottom, whereas for theantenna on-chest case, the patch is offset toward the right ofthe body. Two different body gestures in standing and runningmodes were also considered. Owing to the employed buriedstripline structure, the human body in close proximity does nothave a great impact on the electromagnetic properties of boththe patch antenna and the BPF circuit. Hence, the S11, gain,and AR of the CP filtering antenna under the four different cir-cumstances remain similar to the values when the antenna is infree space. A slight resonance shift in the S11 can be observed,but it is maintained below −13.5 dB in the targeted band for allfour cases. The broadside gain profiles are also well maintainedwith a peak gain of 6.7, 6.4, 6.6, and 6.5 dBi for the on-cheststanding, on-arm standing, on-chest running, and on-arm run-ning cases, respectively, with in-band variations of around 1.2,1.1, 1.3, and 1.1 dB. The simulated average radiation efficiencyin the passband for the antenna when employed in the wearablescenarios is around 76%. The slight drop (∼4%) compared tothe antenna in free space case indicates that the human body hasa very minor impact on the radiation efficiency of the antennawhen it is located in close proximity to human tissue, due to theemployed stripline structure. The broadside ARs are all below3 dB in the majority of the targeted band, except for small peaksthat reach 3.3 dB. The AR band slightly moves to lower fre-quency by about 0.02 GHz on average, due to the loading andshadowing effect of the human body. The simulated normalizedLHCP and RHCP radiation patterns in both the horizontal andvertical planes at 3.8, 4.0, and 4.2 GHz for all four cases arealso displayed in Fig. 11. It can be seen that the majority of theenergy is directed into the hemisphere away from the humanbody. Slight differences exist between the radiation patternsfor the antenna in free space versus on-body. Particularly, pat-tern asymmetry and radiation nulls can be observed in the backlobes, which are primarily attributed to the human body shad-owing effect. The weak LHCP back radiation, upon reflectingfrom the human body surface behind the antenna, is convertedinto RHCP and interferes with the RHCP waves directly radi-ated from the antenna, resulting in the observed multiple lobesin the half space in front of the antenna. The impact of differ-ent human body gestures can also be observed, where the liftedright arm in the running gesture causes more obvious pattern


TABLE IPROPERTY AND PERFORMANCE COMPARISON AMONG DIFFERENT FILTERING ANTENNA DESIGNS

* AR < 3 dB bandwidth in percentage.

asymmetry and a tilting of ∼10◦ in the main beam in the hor-izontal plane for the antenna on-chest case but not the antennaon-arm case. When located on the chest, the filtering antennahas LHCP HPBWs of around 78◦/79◦ and 82◦/80◦ in the y–z/x–y plane, respectively, for the two gestures. When located onthe arm, the filtering antenna has LHCP HPBWs of around80◦/83◦ and 79◦/82◦ in the x–z/x–y plane, respectively, for thetwo gestures. The beamwidths are slightly reduced from thatof the antenna in free space in both the vertical and horizon-tal planes. Importantly, the radiation patterns exhibit a highdegree of cross-polarization discrimination within a wide angu-lar range. Specifically, for all of these four cases, the LHCPwave is at least 15 dB stronger than the RHCP wave within awide angular range of about 85◦ in both the horizontal and verti-cal planes. Such simultaneous wide angular power and circularpolarization coverage indicates that stable wireless links canbe achieved for off-body communications within a wide band-width of ∼500 MHz, which are robust and tolerant to humanbody movement and multipath interference.

Measurements were carried out to validate the on-body simu-lations, where the fabricated CP filtering antenna was mountedon the arm and chest of a person in both a standing and run-ning gesture. Similar to the behavior manifested in the freespace measured results, the S11 curves are slightly wider thanthe simulated results, but in all achieve very good agreement.The maximum in-band S11 values were found to vary between−12.9 dB and −13.6 dB among the four cases. The measuredbroadside gain curves possess similar profiles compared tothose of the filtering antenna in free space, though with a0.5 dB lower peak value and a 0.4 dB higher in-band fluctua-tion on average. The broadside AR curves are smaller than 3 dBwithin the majority of the targeted band, with a slight frequency

shift to upper frequencies and band broadening, correspondingto the behavior that was also exhibited in the S11 and gaincurves. These experiments verify that the proposed antennamodule maintains its simultaneous filtering and CP radiatingfunctionalities when it is mounted on different positions ofan actual human body as well as while undergoing differ-ent gestures. Moreover, compared to commonly used backsidemicrostrip feeding techniques, the stripline configuration pro-vides much better shielding for the high-Q coupled resonatorcircuits.

B. SAR Simulations

Having evaluated the impact of the human body on theperformance of the CP filtering antenna when it is placed inclose proximity to a homogeneous human body model, theSAR values are now investigated which quantifies the effectof the antenna on the human body. A numerical 3-D volu-metric anatomical HUGO human body model produced by theNational Library of Medicine [50], which contains 40 differ-ent types of tissues at a resolution of 2× 2× 2mm mesh size,was incorporated into the CST microwave studio (MWS) soft-ware package for accurate calculation of the SAR values in thenear-field.

Similar to Fig. 11, the filtering antenna was placed at two dif-ferent positions on the HUGO human body, including the chestand the arm, as shown in Fig. 12(a) and (b), respectively. As abenchmark, a power of 100 mW accepted by the antenna waschosen to evaluate the SAR performance. To reduce the simu-lation time, only a portion of the chest or arm with a sectionalarea of larger than 10 times that of the antenna size was usedsince the SAR is a near-field effect. Fig. 12(a) and (b) show the


Fig. 11. Simulated and measured S11, gain, and AR, in addition to simulated LHCP (solid lines) and RHCP (dashed lines) radiation patterns at 3.8 GHz (bluelines), 4.0 GHz (black lines), and 4.2 GHz (red lines) in both the vertical (x–z or y–z) and horizontal (x–y) planes of the integrated CP filtering antenna mountedon the (a) chest and (b) arm with a standing gesture and (c) chest and (d) arm with a running gesture.

Fig. 12. Simulated 1-g averaged SAR values of the filtering antenna mountedon (a) the chest and (b) the arm of the HUGO human body model in CST MWSat 3.8, 4.0, and 4.2 GHz. (c) Simulated peak 1- and 10-g averaged SAR valuesin the targeted band as a function of frequency.

simulated 1-g averaged SAR values for the two cases at 3.8,4.0, and 4.2 GHz. It can be observed that the SAR distribu-tions have the strongest values in the tissue regions closest tothe radiating patch and slight variations elsewhere at differentfrequencies. For both cases, the peak 1-g averaged SAR val-ues as a function of frequency are shown in Fig. 12(c), whichare lower than 0.24 W/kg throughout the entire targeted pass-band; well below the 1.6-W/kg specification provided by theFederal Communication Commission (FCC) [51]. Further eval-uation of the 10-g averaged SAR values shows that the peakvalue is smaller than 0.066 W/kg within the targeted pass-band [see Fig. 12(c)]. Such a small value is also far below themaximum allowed value of 2 W/kg as required by the FCC.Hence, from the 1- and 10-g averaged SAR evaluations, themaximum allowable input power of the CP filtering antenna is667 mW.


V. CONCLUSION

In conclusion, we have proposed a methodology to designcompact, wideband CP filtering antennas using two LPbranches. Rather than independently designing componentsthat are matched to a common impedance, the antenna andBPF circuit are co-designed by treating the patch as the radi-ator as well as the last resonator of the BPF. A proof-of-conceptexample was designed, fabricated, and tested for operationaround 4 GHz. While maintaining a compact form factor of0.53λ0 × 0.53λ0 × 0.07λ0, the antenna is still able to exhibitan S11 < −13.5 dB, an AR smaller than 3 dB, and a gainhigher than 5.2 dBi in a 12.2% bandwidth. Comparing to aconventional patch antenna with and without a directly con-nected BPF, the proposed antenna suppresses the undesiredout-of-band signals and retains the desired flatness of the pass-band antenna gain response. The filtering antenna was shownto maintain its broadband CP performance when placed on dif-ferent positions of a human body and while undergoing diversegestures. Moreover, the filtering antenna exhibits small SARvalues throughout the band. The demonstrated compact filter-ing antennas will find potential applications in various wirelesssystems as well as for advanced wearable devices. It should benoted that the demonstrated antennas were built on rigid sub-strate, making them only wearable on parts of the human bodythat are locally flat. The effect of incorporating flexible sub-strate materials into these integrated wearable filtering antennaswill be further investigated in the future.

ACKNOWLEDGMENT

The authors would like thank Dr. M. Gregory for his helpduring the assembly of the fabricated filtering antennas.

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Zhi Hao Jiang (S’07–M’13) was born in Nanjing,China, in 1986. He received the B.S. degree in radioengineering from Southeast University, Nanjing,China, in 2008, and the Ph.D. degree in electricalengineering from The Pennsylvania State University,University Park, PA, USA, in 2013.

Currently, he is a Postdoctoral Fellow withthe Computational Electromagnetics and AntennasResearch Laboratory (CEARL), Department ofElectrical Engineering, The Pennsylvania StateUniversity. He was a Research Assistant with the

State Key Laboratory of Millimeter Waves, School of Information Scienceand Engineering, Southeast University. He was with Base Station AntennaR&D, Andrew Telecommunication, China, as an intern in 2007. He has coau-thored 3 book chapters and over 60 papers in peer reviewed internationaljournals and conference proceedings. He holds 2 Chinese patents and 4 U.S.patents (pending). His research interests include antennas, microwave circuits,metamaterials, and nanophotonics.

Dr. Jiang is as a Reviewer for Nature Materials, Nature Communications,Scientific Reports, IEEE TRANSACTIONS ON ANTENNAS AND

PROPAGATION, IEEE Antennas and Wireless Propagation Letters, IEEEMicrowave and Wireless Component Letters, IEEE Antennas and PropagationMagazine, Nanoscale, Applied Physics Letters, Journal of Applied Physics,and PIER. He is the Meritorious Winner of the 2006 Interdisciplinary Contestin Modeling funded by the National Security Agency and administrated bythe Consortium for Mathematics and its Applications. He is also the recipientof the 2007 Microsoft Young Fellow awarded by Microsoft Research Asia(MSRA), the 2007 Top Ten Outstanding Students of Jiangsu Province, and the2012 A. J. Ferraro Outstanding Doctoral Research Award in Electromagnetics,and an Honorable Mention in 2013 IEEE AP-S International Symposium onAntennas and Propagation Student Paper Contest.

Douglas H. Werner (F’05) received the B.S., M.S.,and Ph.D. degrees in electrical engineering and theM.A. degree in mathematics from The PennsylvaniaState University (Penn State), University Park, PA,USA, in 1983, 1985, 1989, and 1986, respectively.

He holds the John L. and Genevieve H.McCain Chair Professorship at the Departmentof Electrical Engineering, The Pennsylvania StateUniversity. He is the Director of the ComputationalElectromagnetics and Antennas Research Laboratoryas well as a Member of the Communications and

Space Sciences Laboratory (CSSL). He is also a Faculty Member of theMaterials Research Institute (MRI), Penn State. He holds 8 patents, hasauthored over 625 technical papers and proceedings papers, and is the authorof 14 book chapters with several additional chapters currently in preparation.He has authored several books including Frontiers in Electromagnetics(Piscataway, NJ: IEEE Press, 2000), Genetic Algorithms in Electromagnetics(Hoboken, NJ: Wiley/IEEE, 2007), and Transformation Electromagnetics andMetamaterials: Fundamental Principles and Applications (London, U.K.:Springer, 2014). He has also contributed chapters for several books includingElectromagnetic Optimization by Genetic Algorithms (New York, NY, USA:Wiley Interscience, 1999), Soft Computing in Communications (New York,NY, USA: Springer, 2004), Antenna Engineering Handbook (New York, NY,USA: McGraw-Hill, 2007), Frontiers in Antennas: Next Generation Designand Engineering (New York, NY, USA: McGraw-Hill, 2011), NumericalMethods for Metamaterial Design (New York, NY, USA: Springer, 2013), andComputational Electromagnetics (New York, NY, USA: Springer, 2014). Hisresearch interests include computational electromagnetic, antenna theory anddesign, phased arrays (including ultra-wideband arrays), microwave devices,wireless and personal communication systems (including onbody networks),wearable and e-textile antennas, RFID tag antennas, conformal antennas,reconfigurable antennas, frequency selective surfaces, electromagnetic waveinteractions with complex media, metamaterials, electromagnetic bandgapmaterials, zero and negative index materials, transformation optics, nanoscaleelectromagnetics (including nanoantennas), fractal and knot electrodynamics,and nature-inspired optimization techniques (genetic algorithms, clonalselection algorithms, particle swarm, wind driven optimization, and variousother evolutionary programming schemes).

Prof. Werner is a Fellow of the IET (formerly IEE) and the ACES. Heis a former Associate Editor of Radio Science, a former Editor of the IEEEAntennas and Propagation Magazine, a member of URSI Commissions B andG, Eta Kappa Nu, Tau Beta Pi, and Sigma Xi. He was presented with the1993 Applied Computational Electromagnetics Society (ACES) Best PaperAward and was also the recipient of a 1993 International Union of RadioScience (URSI) Young Scientist Award. In 1994, he received the PennsylvaniaState University Applied Research Laboratory Outstanding Publication Award.He also received the 2015 ACES Technical Achievement Award. He was therecipient of a College of Engineering PSES Outstanding Research Award andOutstanding Teaching Award in March 2000 and March 2002, respectively. Hewas also presented with an IEEE Central Pennsylvania Section MillenniumMedal. In March 2009, he received the PSES Premier Research Award. Hewas a coauthor (with one of his graduate students) of a paper published in theIEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION which receivedthe 2006 R. W. P. King Award. He received the inaugural IEEE Antennas andPropagation Society Edward E. Altshuler Prize Paper Award and the Harold A.Wheeler Applications Prize Paper Award, in 2011 and 2014, respectively.

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Wearable Devices With Low SAR