Voltage Controllable Power Factor Corrector

8/12/2019 Voltage Controllable Power Factor Corrector

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VOLTAGE CONTROLLABL POWER

FACTOR CORRECTOR BASED

INDUCTIVE COUPLING POWER

TRANSFER SYSTEM


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CHAPTER 1

INTRODUCTION

1.1 INTRODUCTION

ICPT power systems (also known as contactless power supplies) are known to have

significant advantages in applications such as the materials handling, lighting and

transportation industries. There are many applications in both high and low power systems in

which use of these power supplies is advantageous. ICPT systems have a primary conductive

path supplied with alternating current from a power supply. One or more secondary devices(which may be referred to as pick-ups) are provided adjacent to, but electrically isolated

from, the primary path. The pick-ups have a pick-up coil in which a voltage is induced by the

magnetic field associated with the primary path, and supply a load such as an electric motor,

a light, or a sensor for example. The pick-up coil is usually tuned using a tuning capacitor to

increase power transfer to the pick-up. A problem with existing ICPT systems is control of

the power transferred to pick-ups when they are lightly loaded, for example when a motor is

supplied by a pick-up and is idle while it awaits a command from a control system. A

solution to this control problem is the use of a shorting switch across the pick-up coil to

decouple the pick-up when required and thus prevent flow of power from the primary

conductive path to the pick-up. This project proposed a novel inductive coupling power

transfer (ICPT) topology to improve the power factor, output voltage regulation and

efficiency. The proposed ICPT is mainly constructed by a voltage controllable power factor

corrector (VC-PFC) and a LLC resonant circuit. Additionally, the series compensation and

series-parallel compensation are used in the primary and the secondary sides of the coupling

transformer to increase the coupling efficiency and the load range. Finally, the circuit


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simulation of the proposed ICPT is presented to verify the performance. Simulation results

show that under the 10 mm coupling distance, the power factor correction and output voltage

regulation can be achieved at the same time. Contactless Energy Transfer (CET) transfers

energy by contactless non metal means, such as electromagnetic coupling, capacitive

coupling, and acoustic waves (ultrasonic), and light. The CET systems allow elimination of

cables, rails, slip rings, plugs, and sockets, resulting in extended maintenance-free operation

and increased reliability and safety (no sparkling, ruggedness against dust, nor aggressive

environmental conditions). Inductive Coupling Power Transfer (ICPT) power supply uses a

magnetic field to transfer power to the load. This technology has been widely applied in

aerospace electric vehicles industrial equipments and battery charging systems The ICPT

often uses a contactless coupled transformer (CCT) for magnetic energy transfer. During the

energy transfer process, an air gap of the CCT will lower the electromagnetic coupling

efficiency and generate substantial leakage inductance. The reactive power produced by the

leakage inductance increases the system VA rating and decreases the input power factor, thus

increases the system losses and reduces the transfer efficiency. In order to solve this problem,

the resonant frequency control and the var compensation strategies had been proposed to

increase the CCT coupling efficiency.


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CHAPTER2

LITERATURE SURVEY

2.1 INDUCTIVE COUPLED POWER TRANSFER

Inductively coupled power transfer (ICPT) systems are designed to deliver power from a

stationary primary source to one or more secondary loads efficiently via magnetic coupling

[1-2]. ICPT systems have been paid more and more attention for their better performance

than traditional power transmission method, such as no sparking, maintenance free or less

maintenance, dustproof and waterproof. Recently, some high power ICPT systems have been

developed in applications that require a relatively large air gaps and consequently caused

much low coupling factors between the first and the secondary coils, such as electric vehicles,

industry rotate robot[3-4]. Normally, due to low coupling coefficient of the ICPT system, a

resonant tank or compensation capacitors are necessary to reduce the VA rate of the system

power supply and enhance power transfer performance [5,6].There are many studies on the

resonance and compensate topologies for ICPT systems[5], and power transfer capability of

ICPT system[6], and some ICPT system design methods have been proposed in [1,7].

However, though many papers have discussed the problems of different conditions of ICPT

system, include the load model, power transfer control, compensating topology and

optimizing design, there is not a general reflected load model which could determine the

precise relationship between the transfer power and the load level with different secondary

compensating topologies. In this paper, utilizing reflected load model, a boundary load value

is found for optimal power transfer performance to select secondary compensation topology

in designing an ICPT system with variable load, and VA rating requirements are investigated

for variable load resistance and variable coupling coefficient on different compensation


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topologies. Finally, some simulations are carried out to verify the theory. INDUCTIVELY

coupled power transfer (ICPT) technology involving some important components such as

high frequency electromagnetic [1][3] coupling, resonant converter [4], and power

regulation [5] has been a key technology sin cethe 1990s. With its contactless transmission

for supplying power, it can be used in special environments, such as combustible or explosive

areas and wet or under water environments. In addition, biomedical and transit equipment

with electrical power supply and other portable electronic devices are also beginning to apply

it. This rapid development of ICPT technologyis attracting expanding interest in the research

and application of this field [6][12]. For ICPT systems, parameter variations, particularly

small change in load, will cause frequency drift [13], resulting in adecrease in the efficiency

and distance for power transmission, as well as creating sine wave distortion and other issues.

It has been pointed out [14], [15]that uncertainty in the operating frequency can result in

difficulties in designing electromagnetic interference (EMI) filters with suitable bandwidths

and choosing suitable switching devices. It has also been pointed out [16] that, once the

frequency exceeds a certain range, ICPT systems will have multiple operating conditions and

system stability will be affected. As is well known, a mathematical model for the analysis of

an ICPT system [17][19] is always an approximation of the true physical reality of the

system dynamics, such as the generalized state-space averaging method which achieves an

approximate linear description of the nonlinear original system by omitting the high-order

terms of the Fourier series of the system variables [20]. The typical sources of the

discrepancy between the mathematical model and the actual ICPT system include the

dynamics of the system that have not been modelled (usually the high-frequency elements)

and the system-parameter variations due to environmental changes and wear and tear factors.

All these dynamic uncertainties will inevitably affect the performance and stability of the

corresponding control system. It is well understood that an ICPT system which includes the

inverter, the rectifier, and the resonant and magnetic coupling units becomes a typical high-

order and complex nonlinear system, which is vulnerable to interference signals and dynamic

perturbations. Therefore, robustness is of crucial importance in the design of the control of an

ICPT system. The main approaches currently are the H optimal approach and the -

synthesis approach. The H optimal approach can achieve robust stability against

unstructured system perturbations and nominal performance requirements but neglects robust

performance requirements [21], whereas the synthesis approach is based on the structured

singular value , can achieve robust stability and robust performance (RSRP) of a control

system with regard to structured perturbations [22], and quantitatively characterizes the effect


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of structured uncertainties on the RSRP of a linear dynamic system. Thus, it avoids the

unnecessary and conservative aspects of using the H optimal approach forcontrol system

design. This study uses the -type ICPT system [23] as an example. The approach to -

synthesis is based on the structured singular value for the uncertainty in the operating

frequency and involves obtaining an appropriate model and then proceeding to design a

controller for the model to restrain the effects on the system dynamics and robustness due to

frequency perturbation. A standard configuration with a perturbed feedback for -synthesis is

derived from the generalized state-space averaging model of the -type ICPT system using

linear fractional

Fig. 2.1. Circuit topology for -type resonant ICPT system

2.2 FUNDAMENTAL STRUCTURE OF AN ICPTSYSTEM

The general structure of an ICPT system is shown in Fig.2.2. AC source is commutated by a

rectifier and then feed to a high frequency inverter, high frequency alternating current

generated by the inverter is injected into the primary coil after primary compensation. The

primary coil is tuned with compensation elements to minimize the VA rating of the power

supply, and the secondary coil is tuned to enhance power transfer capability. The primary

side and the secondary side are isolated by an air gap and there is no physical contact

between them, therefore, the secondary side(load) may sliding or rotating with the first coil

with no mechanical attrition.

Figure 2.2. General structure of ICPT system

2.3 SECONDARY COMPENSATION


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describe the voltage, current and power relationship of input and output of ICPT system

accurately. According with mutual inductance model, the equivalent circuit of ICPT is shown

in fig.2.4.Fig.2.4(a) shows the equivalent circuit of series compensation for secondary coil,

while the parallel compensated secondary coil is shown in Fig.2.4(b). As is shown in fig.2.4,

where, rp, rs represent resistance values of the first coil and the secondary coil, respectively,

Zr2 represents reflected impedance of the secondary side, therefore, Voc represent the

inductive voltage from the primary current via mutual inductance. The reflected load model

can be deduced utilizing mutual inductance theory:

Where, S-SC and P-SC represent that the secondary coil is compensated in series and in

parallel, respectively.

Because the resistance rp, rs are much smaller than the load resistance R, they are generally

neglected to simplify the analysis, thereby the reflected impedance for the secondary side

can be deduced at the nominal frequency[5]:

(a) series compensation for (b) paralllel compensation for secondary secondary


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Figure 2.4. Equivalent circuit of ICPT system

Therefore, for a constant primary coil current Ip[9], let the power transferred to the load Rbe

P2, there is:

Substituting equation (5) and (6) into equation (7), the power transferred to the load at

different compensation topologies can be derived out:

From equation(8), utilizing appendix parameters, power transfer performance for ICPT

system with different secondary


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Figure 2.5. Power transfer performance for different secondary compensation

Topologies

compensation topologies is drawn in Fig.2.5, as is shown, on the rated frequency, when the

coupling coefficient is invariable, following with increasing of load resistance value, the

transferred power drastically decreased before the intersection, and then slowly decreased

with series secondary compensation; while with parallel secondary compensation, the

transferred power increased linearity following with increasing of load resistance value. As

can be seen in Fig.2.5, when the load resistance is less than the intersection resistance value,

which is defined as boundary load resistance Rb, power transfer capability of series

secondary compensation topology is better than parallel secondary compensation topology,

when the load resistance is greater than Rb, power transfer capability of parallel secondary

compensation is better than series secondary compensation. From equation(8), Rbcan be

derived out:

2.5 VARATING REQUIREMENT

In ICPT system, VA rating requirement for power supply is a major system cost, thereby the

primary coil is compensated to reduce the VA rating and hence the size and cost of the power

supply[10]. As is shown in Fig.2.6, the basic primary compensation topologies include series

compensation and parallel compensation. At the rated condition, the primary compensation

capacitor is design to satisfy zero phase angle between the current and the voltage of power

source[1, 9]. Thereby primary compensation capacitor values can be derived out with

dirrerent compensation topologies

TABLE I. PRIMARY COMPENSATION CAPACITOR VALUE


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Where SS represents series-series compensation for the primary coil and the secondary coil

respectively, and the same goes for SP, PS and PP. Because the compensation capacitors

are designed at rated load value and rating coupling coefficient, for some ICPT system, when

load level and coupling coefficient may deviate from the rated value [1, 3], then VA rating

requirement for power supply would deviate the rated value. As Fig.2.6 shows, let VA

rating requirement for ICPT system be S:


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Thereby, as shown in Table II., VA rating requirement Scan be derived:

TABLE II. VARATING REQUIREMEN FOR DIFFERENT COMPENSATION

TOPOLOGIES

According with Table II., utlizing appendix ICPT system parameters, VA rating requirements

of different compensation topologies for ICPT system are ploted. For series secondary

compensation, when coupling coefficient kRs, and the primary coil inductance is

compensated at rated load resistance Rsand rated coupling coefficient k0, the VA rating

requirement Svalues are always less than P2. However, for parallel secondary compensation,

when coupling coefficient k


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(a)parallel compensation (b) series compensationFigure 2.6. Primary compensation topologies

2.6 OPERATING PRINCIPLES OF THE -TYPE ICPT SYSTEM

A circuit topology of the -type ICPT system is shown in

Fig. 2.7. Compared to the conventional ICPT circuits, it appends a resonant network

consisting of Ln, Cn1, and Cn2. For a completely tuned network, it will be a purely

resistive network and can not only improve the input power factor of the whole resonant

network ( network and series resonant network) but also ensure the zero-voltage switching

(ZVS) condition of the full-bridge inverter. In addition, the network as an effectiveband

pass filter only works over a finite bandwidth around the nominal frequency 0 so that

harmonics will not propagate. Specific operating principles of this system can be described as

follows: The dc power supply Edc can be obtained from a buck dc chopper, and dc

inductor Ld in series keeps the dc current constant. The two main switching pairs (S 1, S4

and S2, S3) of the full-bridge inverter network alternate the direction of charging current into

the resonant circuit. The series resonant network consisting of Lp, Cp, and R Lp is used to

generate sinusoidal excitation current with low distortion, and the coil Lp as the transmitter

transfers the high-frequency resonant energy to the pickup coil Ls by the magnetic fields

coupling. The inductor Ls is completely tuned by the parallel capacitor Cs, and the ac signal v

Cs is rectified by a full-bridge uncontrolled rectifier consisting of four diodes; then, the output

voltage for a given load is directly obtained after an LC filter consisting of Lf and Cf to

reduce the output voltage ripple. For a full-bridge topology in the primary side, the resonant

ac driving voltage v p in rms is


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and the whole resonant network should be fully tuned to ensure that the input power factor is

1. At the nominal frequency 0,the impedance of each component of the network and the

series resonant network should satisfy the equations as follows:

Additionally, for a given sine sinusoidal wave inputvpat0, the excitation currentiLpin rms

can be expressed as

Moreover, this current is always 90 lagging the input voltage, and its rms value is

completely determined by the input voltage magnitude and the characteristic impedance of

the network.

2.7. LLC RESONANT CONVERTER

Resonant converter, which were been investigated intensively in the 80's, can achieve very

low switching loss thus enable resonant topologies to operate at high switching frequency. In

resonant topologies, Series Resonant Converter (SRC), Parallel Resonant Converter (PRC)

and Series Parallel Resonant Converter (SPRC, also called LCC resonant converter) are the

three most popular topologies. The analysis and design of these topologies have been studied

thoroughly. In next part, these three topologies will be investigated for front-end application.

2.7.1 LLC Resonant Half-Bridge Power Converter

While half-bridge power stages have commonly been used for isolated, medium-power

applications, converters with high-voltage inputs are often designed with resonant switching

to achieve higher efficiency, an improvement that comes with added complexity but that

nevertheless offers several performance benefits. This topic provides detailed information on

designing a resonant half-bridge converter that uses two inductors (LL) and a capacitor (C),

known as an LLC configuration. This topic also introduces a unique analysis tool called first

harmonic approximation (FHA) for controlling frequency modulation. FHA is used to define


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circuit parameters and predict performance, which is then verified through comprehensive

laboratory measurements. Higher efficiency, higher power density, and higher component

density have become common in power-supply designs and their applications. Resonant

power convertersespecially those with an LLC half-bridge configurationare receiving

renewed interest because of this trend and the potential of these converters to achieve both

higher switching frequencies and lower switching losses. However, designing such

converters presents many challenges, among them the fact that the LLC resonant half-bridge

converter performs power conversion with frequency modulation instead of pulse-width

modulation, requiring a different design approach. This topic presents a design procedure for

the LLC resonant half-bridge converter, beginning with a brief review of basic resonant-

converter operation and a description of the energy-transfer function as an essential

requirement for the design process. This energy-transfer function, presented as a voltage

ratio or voltage-gain function, is used along with resonant-circuit parameters to describe the

relationship between input voltage and output voltage. Next, a method for determining

parameter values is explained. To demonstrate how a design is created, a step-by-step

example is then presented for a converter with 300 W of output power, a 390- VDC input,

and a 12-VDC output. The topic concludes with the results of bench-tested performance

measurements.

2.7.2 Brief Review of Resonant Converters

There are many resonant-converter topologies, and they all operate in essentially the same

way: A square pulse of voltage or current generated by the power switches is applied to a

resonant circuit. Energy circulates in the resonant circuit, and some or all of it is then tapped

off to supply the output. More detailed descriptions and discussions can be found in this

topics references. Among resonant converters, two basic types are the series resonant

converter (SRC), shown in Fig. 2.7a and the parallel resonant converter (PRC), shown in

Fig. 2.7b. Both of these converters regulate their output voltage by changing the frequency of

the driving voltage such that the impedance of the resonant circuit changes. The input

voltage is split between this impedance and the load. Since the SRC works as a voltage

divider between the input and the load, the DC gain of an SRC is always


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a. Series resonant converter b. Parallel resonant converter

Fig. 2.7. Basic resonant-converter configurations.

lower than 1. Under light-load conditions, the impedance of the load is very large compared

to the impedance of the resonant circuit; so it becomes difficult to regulate the output, since

this requires the frequency to approach infinity as the load approaches zero. Even at nominal

loads, wide frequency variation is required to regulate the output when there is a large

input-voltage range. In the PRC shown in Fig. 2.7b, the load is connected in parallel with the

resonant circuit, inevitability requiring large amounts of circulating current. This makes it

difficult to apply parallel resonant topologies in applications with high power density or

large load variations.

2.7.3 LCC and LLC Resonant Converters

To solve these limitations, a converter combining the series and parallel configurations,

called a series-parallel resonant converter (SPRC), has been proposed. One version of this

structure uses one inductor and two capacitors, or an LCC configuration, as shown in Fig.

2.8a. Although this combination overcomes the drawbacks of a simple SRC or PRC by

embedding more resonant frequencies, it requires two independent physical capacitors that

are both large and expensive because of the high AC currents. To get similar characteristics

without changing the physical component count, the SPRC can be altered to use two

inductors and one capacitor, forming an LLC resonant converter (Fig. 2.8b). An advantage of

the LLC over the LCC topology is that the two physical inductors can often be integrated

into one physical component, including both the series resonant


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a. LCC configuration.

b. LLC configuration

Fig. 2.8. Two types of SPRC

inductance, Lr , and the transformers magnetizing inductance, Lm. The LLC resonant

converter has many additional benefits over conventional resonant converters. For example, it

can regulate the output over wide line and load variations with a relatively small variation of

switching frequency, while maintaining excellent efficiency. It can also achieve zerovoltage

switching (ZVS) over the entire operating range. Using the LLC resonant configuration in

an isolated half-bridge topology will be described next, followed by the procedure for

designing this topology.

2.8. LLc resonant Half bridge converter

This section describes a typical isolated LLC resonant half-bridge converter; its operation; its

circuit modelling with simplifications; and the relationship between the input and output

voltages, called the voltage-gain function. This voltage-gain function forms the basis for the

design procedure described in this topic.


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a. Typical configuration.

b. Simplified converter circuit

Fig. 2.9. LLC resonant half-bridge converter.


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2.8.1 Configuration

Fig. 2.9a shows a typical topology of an LLC resonant half-bridge converter. This circuit is

very similar to that in Fig. 2.8b. For convenience, Fig. 2.8b is copied as Fig. 2.9b with the

series elements interchanged, so that a side-by-side comparison with Fig. 2.9a can be made.

The converter configuration in Fig.2.9a has three main parts:

1. Power switches Q1 and Q2, which are usually MOSFETs, are configured to form a

squarewave generator. This generator produces a unipolar square-wave voltage, Vsq, by

driving switches Q1 and Q2, with alternating 50% duty cycles for each switch. A small dead

time is needed between the consecutive transitions, both to prevent the possibility of

crossconduction and to allow time for ZVS to be achieved.

2. The resonant circuit, also called a resonant network, consists of the resonant capacitance,

Cr , and two inductancesthe series resonant inductance, Lr , and the transformers

magnetizing inductance, Lm. The transformer turns ratio is n. The resonant network

circulates the electric current and, as a result, the energy is circulated and delivered to the

load through the transformer. The transformers primary winding receives a bipolar square

wave voltage, Vso. This voltage is transferred to the secondary side, with the transformer

providing both electrical isolation and the turns ratio to deliver the required voltage level to

the output. In Fig. 2.9b, the load RL includes the load RL of Fig. 2.9a together with the

losses from the transformer and output rectifiers.

3. On the converters secondary side, two diodes constitute a full-wave rectifier to convert

AC input to DC output and supply the load RL. The output capacitor smooths the rectified

voltage and current. The rectifier network can be implemented as a full-wave bridge or

centertapped configuration, with a capacitive output filter. The rectifiers can also be

implemented with MOSFETs forming synchronous rectification to reduce conduction

losses, especially beneficial in low-voltage and highcurrent applications.

2.8.2 Operation

This section provides a review of LLC resonant-converter operation, starting with series

resonance. Resonant Frequencies in an SRC Fundamentally, the resonant network of an SRC

presents a minimum impedance to the sinusoidal current at the resonant frequency,


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regardless of the frequency of the square-wave voltage applied at the input. This is

sometimes called the resonant circuits selective property. Away from resonance, the circuit

presents higher impedance levels. The amount of current, or associated energy, to be

circulated and delivered to the load is then mainly dependent upon the value of the resonant

circuits impedance at that frequency for a given load impedance. As the frequency of the

square-wave generator is varied, the resonant circuits impedance varies to control that

portion of energy delivered to the load. An SRC has only one resonance, the series resonant

frequency, denoted as

The circuits frequency at peak resonance, fc0, is always equal to its f0 . Because of this, an

SRC requires a wide frequency variation in order to accommodate input and output

variations.

a)fc0, f0, and fpin an LLC Circuit

However, the LLC circuit is different. After the second inductance (Lm) is added, the LLC

circuits frequency at peak resonance (fc0) becomes a function of load, moving within the

range of fp fc0 f0 as the load changes. f0 is still described by Equation (1), and the pole

frequency is described by

At no load, fc0 = fp . As the load increases, fc0 moves towards f0 . At a load short circuit,

fc0 = f0 . Hence, LLC impedance adjustment follows a family of curves with fp fc0 f0,

unlike that in SRC, where a single curve defines fc0 = f0 . This helps to reduce the

frequency range required from an LLC resonant converter but complicates the circuit

analysis. It is apparent from Fig. 2.9b that f0 as described by Equation (1) is always true

regardless of the load, but fp described by Equation (2) is true only at no load. Later it will

be shown that most of the time an LLC converter is designed to operate in the vicinity of f0 .

For this reason and others yet to be explained, f0 is a critical factor for the converters

operation and design.


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b)Operation At, Below, and Above f0

The operation of an LLC resonant converter may be characterized by the relationship of the

switching frequency, denoted as fsw, to the series resonant frequency (f0). Fig. 2.10

illustrates the typical waveforms of an LLC resonant converter with the switching frequency

at, below, or above the series resonant frequency. The graphs show, from top to bottom, the

Q1 gate (Vg_Q1), the Q2 gate (Vg_Q2), the switch-node voltage (Vsq), the resonant

circuits current (Ir), the magnetizing current (Im), and the secondary-side diode current (Is).

Note that the primary-side current is the sum of the magnetizing current and the secondary-

side current referred to the primary; but, since the magnetizing current flows only in the

primary side, it does not contribute to the power transferred from the primary-side source to

the secondary side load.

a. At f0

.


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b. Below f0

c.Above f0

Fig. 2.10. Operation of LLC resonant converter

c) Operation at Resonance (Fig. 2.10a)

In this mode the switching frequency is the same as the series resonant frequency. When

switch Q1 turns off, the resonant current falls to the value of the magnetizing current, and

there is no further transfer of power to the secondary side. By delaying the turn-on time of

switch Q2, the circuit achieves primary-side ZVS and obtains a soft commutation of the

rectifier diodes on the secondary side. The design conditions for achieving ZVS will be

discussed later. However, it is obvious that operation at series resonance produces only a


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single point of operation. To cover both input and output variations, the switching frequency

will have to be adjusted away from resonance.

D) Operation Below Resonance (Fig. 2.10b)

Here the resonant current has fallen to the value of the magnetizing current before the end of

the driving pulse width, causing the power transfer to cease even though the magnetizing

current continues. Operation below the series resonant frequency can still achieve primary

ZVS and obtain the soft commutation of the rectifier diodes on the secondary side. The

secondary-side diodes are in discontinuous current mode and require more circulating

current in the resonant circuit to deliver the same amount of energy to the load. This

additional current results in higher conduction losses in both the primary and the secondary

sides. However, one characteristic that should be noted is that the primary ZVS may be lost

if the switching frequency becomes too low. This will result in high switching losses and

several associated issues. This will be explained further later. Operation Above Resonance

(Fig. 2.10c) In this mode the primary side presents a smaller circulating current in the

resonant circuit. This reduces conduction loss because the resonant circuits current is in

continuous-current mode, resulting in less RMS current for the same amount of load. The

rectifier diodes are not softly commutated and reverse recovery losses exist, but operation

above the resonant frequency can still achieve primary ZVS. Operation above the resonant

frequency may cause significant frequency increases under light-load conditions. The

foregoing discussion has shown that the converter can be designed by using either fsw f0

or fsw f0, or by varying fsw on either side around f0. Further discussion will show that the

best operation exists in the vicinity of the series resonant frequency, where the benefits of

the LLC converter are maximized. This will be the design goal.

2.8.3 Modeling an LLC Half-Bridge Converter

To design a converter for variable-energy transfer and output-voltage regulation, a voltage

transfer function is a must. This transfer function, which in this topic is also called the input-

to output voltage gain, is the mathematical relationship between the input and output

voltages. This section will show how the gain formula is developed and what the

characteristics of the gain are. Later the gain formula obtained will be to describe the design

procedure for the LLC resonant half-bridge converter.


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Figure 2.12 Half Bridge Series Resonant Converter

2.9.2 Parallel resonant converter

The schematic of parallel resonant converter is shown in Figure 2.12 [B14]- [B17]. Its DC

characteristic is shown in Figure 2.13. For parallel resonant converter, the resonant tank is

still in series. It is called parallel resonant converter because in this case the load is in parallel

with the resonant capacitor. More accurately, this converter should be called series resonantconverter with parallel load. Since transformer primary side is a capacitor, an inductor is

added on the secondary side to math the impedance.

Figure 2.13 Half bridge parallel resonant converter


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2.9.3 Series parallel resonant converter

The schematic of series parallel resonant converter is shown in Figure 2.14 [B18]-[B20]. The

DC characteristic of SPRC is shown in Figure 2.14. Its resonant tank consists of three

resonant components: Lr, Cs and Cp. The resonant tank of SPRC can be looked as the

combination of SRC and PRC. Similar as PRC, an output filter inductor is added on

secondary side to math the impedance. For SPRC, it combines the good characteristic of PRC

and SRC. With load in series with series tank Lr and Cs, the circulating energy is smaller

compared with PRC. With the parallel capacitor Cp, SPRC can regulate the output voltage at

no load condition. The parameters of SPRC designed for front end DC/DC application are:

Figure 2.14 Half bridge series parallel resonant converter

2.10. POWER FACTOR CORRECTION

2.10.1 Power Factor Correction (PFC)

Power factor correction is the method of improving the power factor of a system by using

suitable devices. The objective of power factor correction circuits is to make the input to a

power supply behave like purely resistive or a resistor. When the ratio between the voltage

and current is a constant, then the input will be resistive hence the power factor will be 1.0.

When the ratio between voltage and current is other than one due to the presence of non-

linear loads, the input will contain phase displacement, harmonic distortion and thus, the

power factor gets degraded [5-7].


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2.10.2 Need of PFC

The rise in the industrial, commercial and residential applications of electronic equipments

has resulted in a huge variety of electronic devices requiring mains supply. These devices

have rectification circuits, which is the prominent reason of harmonic distortion. These

devices convert AC to DC power supply which causes current pulses to be drawn from the ac

network during each half cycle of the supply waveform. Even if a single device for example,

a television may not draw a lot of reactive power nor it can generate enough harmonics to

affect the supply system significantly, but within a particular phase connection, there may

exist several such devices connected to the same supply phase resulting in production of a

large amount of reactive power flow and harmonics in line current [5-7].

With improvement in the field of semiconductors, the size and weight of control circuits have

drastically reduced. This has also affected their performance and thus power electronic

converters have become increasingly popular in industrial, commercial and residential

applications. However this mismatch between power supplied and power used cannot be

detected by any kind of meter meant for charging the domestic consumers, and hence, results

in direct loss of revenues [5-7].

Moreover, since different streets are supplied with different phases, a 3-phase unbalanced

condition may also arise within a housing scheme. The unbalance current flows in the neutralline of a star connected network causing undesirable heating and burning of the conductor [5-

7].

This pulsating current contains harmonics which results in additional losses and dielectric

stresses in capacitors and cables, increasing currents in windings of rotating machinery (e.g.,

induction motors) and transformers and noise emissions in many equipments. The rectifier

used in the AC input side is the prime source of this problem. Thus, in order to decrease the

effect of this distortion, power factor correction circuits are added to the supply input side of

equipments used in industries and domestic applications to increase the efficiency of power

usage [5-7].

2.10.3 Types of Power Factor Correction (PFC)

Power Factor Correction can be classified as two types:

1. Passive Power Factor Correction

2. Active Power Factor Correction


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1) Passive Power Factor Correction

In Passive PFC, in addition to the diode bridge rectifier, passive elements are introduced to

improve the nature of the line current. By using this, power factor can be increased to a value

of 0.7 to 0.8 approximately. As the voltage level of power supply increases, the sizes of PFC

components increase. The idea of passive PFC is to filter out the harmonic currents by use of

a low pass filter and only allow the 50 Hz power frequency wave to increase the power factor

[5], [7].

Advantages of Passive PFC :

It has a simple structure. It is reliable and rugged. The cost is very low because only a filter is required. The high frequency switching losses are absent and it is not sensitive to noises and

surges.

The equipments used in this circuit dont generate high frequency EMI [5], [7].Disadvantages of Passive PFC :

For achieving better power factor the size of the filter increases.

Due to the time lag associated with the passive elements it has a poor dynamicresponse.

The voltage cannot be regulated and the efficiency is low. Due to presence of inductors and capacitors interaction may take place

between the passive elements and the system resonance may occur at different

frequencies.

Although by filtering the harmonics can be filtered out, the fundamentalcomponent may get phase shifted thus reducing the power factor

The shape of input current is dependent upon what kind of load is connected .


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2) Active Power Factor Correction

An active PFC is a power electronic device designed to control the amount of power drawn

by a load and obtains a power factor as close as possible to unity. Commonly any active PFC

design functions by controlling the input current in order to make the current waveform

follow the supply voltage waveform closely (i.e. a sine wave). A combination of the reactive

elements and some active switches increase the effectiveness of the line current shaping and

to obtain controllable output voltage [5], [7], [8].

The switching frequency differentiates the active PFC solutions into two classes.

Low frequency active PFC:

Switching takes place at low-order harmonics of the line-frequency and it is synchronized

with the line voltage.

High frequency active PFC:

The switching frequency is much higher than the line frequency.

The power factor value obtained through Active PFC technique can be more than 0.9. With a

suitable design even a power factor of 0.99 can be achieved easily. Active PFC power supply

can detect the input voltage automatically, supports 110V to 240V alternative current, its size

and weight is smaller than passive PFC power supply.

Advantages of Active PFC :

The weight of active PFC system is very less.

The size is also smaller and a power factor value of over 0.95 can be obtained throughthis method.

It reduces the harmonics present in the system.

Automatic correction of the AC input voltage can be obtained.

It is capable of operating in a full range of voltage


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Disadvantages of Active PFC :

The layout design is somewhat more complex than passive PFC.

It is very expensive since it needs PFC control IC, high voltage MOSFET, highvoltage ultra-fast choke and other circuits

2.11. Single Phase Uncontrolled Rectifier

One of the first and most widely used application of power electronic devices have been in

rectification. Rectification refers to the process of converting an ac voltage or current source

to dc voltage and current. Rectifiers specially refer to power electronic converters where the

electrical power flows from the ac side to the dc side. In many situations the same converter

circuit may carry electrical power from the dc side to the ac side where upon they are referred

to as inverters. In this lesson and subsequent ones the working principle and analysis of

several commonly used rectifier circuits supplying different types of loads (resistive,

inductive, capacitive, back emf type) will be presented. Points of interest in the analysis will

be.

Waveforms and characteristic values (average, RMS etc) of the rectified voltage and

current. Influence of the load type on the rectified voltage and current.

Harmonic content in the output.

Voltage and current ratings of the power electronic devices used in the rectifier circuit.

Reaction of the rectifier circuit upon the ac network, reactive power requirement, power

factor, harmonics etc.

Rectifier control aspects (for controlled rectifiers only) In the analysis, following

simplifying assumptions will be made.

The internal impedance of the ac source is zero.

Power electronic devices used in the rectifier are ideal switches.

The first assumption will be relaxed in a latter module. However, unless specified otherwise,

the second assumption will remain in force. Rectifiers are used in a large variety of

configurations and a method of classifying them into certain categories (based on common

characteristics) will certainly help one to gain significant insight into their operation.

Unfortunately, no consensus exists among experts regarding the criteria to be used for such


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the type of the load. In this section, operation of this rectifier with resistive, inductive and

capacitive loads will be discussed.

Fig.2.15. Single phase uncontrolled half wave rectifier with resistive load. (a). circuit

diagram, (b) wave forms

The circuit diagram and the waveforms of a single phase uncontrolled half wave

rectifier. If the switch S is closed at at t = 0, the diode D becomes forward biased in the the

interval 0 < t . If the diode is assumed to be ideal then


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b)INDUCTIVE LOAD:

The ripple factor of output current can be reduced to same extent by connecting an inductor

in series with the load resistance as shown in Fig 2.16 (a). As in the previous case, the diode

D is forward biased when the switch S is turned on. at t = 0. However, due to the load

inductance i0

increases more slowly. Eventually at t = , v0becomes zero again. However, i

0

is still positive at this point. Therefore, D continues to conduct beyond t = while thenegative supply voltage is supported by the inductor till its current becomes zero at t = .

Beyond this point, D becomes reverse biased. Both v0

and i0

remains zero till the beginning of

the next cycle where upon the same process

Fig.2.16. Single phase uncontrolled half wave rectifier with inductive load (a). circuit diagram (b).

waveforms


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2.11.2. Single phase uncontrolled full wave rectifier

Single phase uncontrolled half wave rectifiers suffer from poor output voltage and/or input

current ripple factor. In addition, the input current contains a dc component which may cause

problem (e.g. Transformer saturation etc) in the power supply system. The output dc voltage

is also relatively less. Some of these problems can be addressed using a full wave rectifier.

They use more number of diodes but provide higher average and rms output voltage. There

are two types of full wave uncontrolled rectifiers commonly in use. If a split power supply is

available (e.g. output from a split secondary transformer) only two diode will be required to

produce a full wave rectifier. These are called split secondary rectifiers and are commonly

used as the input stage of a linear dc voltage regulator. However, if no split supply is

available the bridge configuration of the full wave rectifier is used. This is the more

commonly used full wave uncontrolled rectifier configuration. Both these configurations are

analyzed next.

Fig.2.17. Single phase un controlled full wave rectifier supplying an R-L load, (a). circuit

diagram (b). Wave forms

The circuit diagram and waveforms of a single phase split supply, uncontrolled full wave

rectifier supplying an R L load. The split power supply can be thought of to have been


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obtained from the secondary of a center tapped ideal transformer (i.e. no internal impedance).

When the switch is closed at the positive going zero crossing of v1

the diode D1

is forward

biased and the load is connected to v1. The currents i

0and i

i1start rising through D

1. When v

1

reaches its negative going zero crossing both i0

and ii1

are positive which keeps D1

in

conduction. Therefore, the voltage across D2

is . Beyond the negative going zero crossing of

vCB21v=v-vi,

D2becomes forward biased and the current i

0commutates to D

2from D

1. The

load voltage v0

becomes equal to v2

and D1

starts blocking the voltage . The current

iAB12v=v-v0

however continues to increase through D2

till it reaches the steady state level

after several cycles. Steady state waveforms of the variables are shown in from t = 0

onwards. It should be noted that the current i0,

once started, always remains positive. This

mode of operation of the rectifier is called the Continuous conduction mode of operation.

This should be compared with the i0

waveform of for the half wave rectifier where i0

remains

zero for some duration of the input supply waveform. This mode is called thediscontinuous

conduction mode of operation.


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CHAPTER3

PROPOSED CONCEPT

3.1 INTRODUCTION

Contactless Energy Transfer (CET) transfers energy by contactless non metal means, such as

electromagnetic coupling, capacitive coupling, acoustic waves (ultrasonic), and light. The

CET systems allow elimination of cables, rails, slip rings, plugs, and sockets, resulting in

extended maintenance-free operation and increased reliability and safety (no sparkling,

ruggedness against dust, nor aggressive environmental conditions). Inductive Coupling

Power Transfer (ICPT) power supply uses a magnetic field to transfer power to the load. This

technology has been widely applied in aerospace [1], [2], electric vehicles [3]-[5], industrial

equipments [6],[7] and battery charging systems [8]-[11]. The ICPT often uses a contactless

coupled transformer (CCT) for magnetic energy transfer. During the energy transfer process,

an air gap of the CCT will lower the electromagnetic coupling efficiency and generate

substantial leakage inductance. The reactive power produced by the leakage inductance

increases the system VA rating and decreases the input power factor, thus increases the

system losses and reduces the transfer efficiency. In order to solve this problem, the resonant

frequency control and the var compensation strategies had been proposed to increase the CCTcoupling efficiency [12]. However, the voltage gain of the ICPT at the resonant frequency is

variable when the load is changing. Therefore, the output voltage is difficult to be regulated

to a stable and constant value. In order to regulate the output voltage, the operation frequency

of ICPTS can be shifted away the resonant frequency [13]. Unfortunately, the result is the

reducing of the CCT coupling efficiency and the increasing of the system VA rating. Using a

DC/DC converter at the secondary side to control the output voltage is also presented. But,

the multi-stage construction has more components, lower efficiency and more circuit cost

[14],[15]. Recently, switching capacitor and switching inductor were used to change the

characteristics of the resonant tank in order to regulate the output voltage. The disadvantage

is that the resonant tank current increases during the output power voltage control process,

which leads to increasing of power component current stress and conduction loss, as well as

decreases circuit transfer efficiency [16]-[18]. This paper proposed a novel ICPT topology to

improve the power factor, output voltage regulation and efficiency. The proposed ICPT is

mainly constructed by a voltage controllable power factor corrector (VC-PFC) and a LLC

resonant circuit. Using the proposed VC-PFC, the functions of power factor correction and


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output voltage regulation can be achieved at the same time. Using the LLC resonant tank with

phase-locked control, the functions of operation frequency control and zero-voltage switching

(ZVS) can also be obtained. Additionally, the series compensated and series-parallel

compensated are used in the primary and the secondary sides of the coupling transformer to

increase the coupling efficiency and the load range. Finally, a 1kW prototype for 10mm

coupling distance is designed and simulated to verify the performance of the proposed ICPT.

The simulation results also show that the input power factor is above 0.92.

3.2 ICPT TOPOLOGIES

An ICPT consists of two independent mutually coupled electrical systems. The primary side

uses a suitable resonant high frequency switching power supply with primary compensation

to minimize the VA rating of the supply. In the secondary side, compensation is required to

enhance the power transfer capability. A controller is normally used to control the operation

frequency in the primary side to achieve maximum power transfer capability. Figure 3.1

shows the frequency-fixed ICPT topology with simple, but high VA rating [18]. In order to

minimize the VA rating of the power supply, frequency-varied control is presented. The

frequency varied control ICPT construction with phase locked loop (PLL) technology is

shown as Fig. 3.2[12]. However, the ideal control point becomes difficult to determine when

more than one zero phase angle condition exists in the frequency spectrum. Additionally, the

voltage gain of the ICPT at the resonant frequency is changed when the load is changed and

the output voltage is not stable. In order to regulate the output voltage, the voltage control

oscillation (VCO) ICPT is used to control the operation frequency according to the feedback

output voltage o V[13]. The VCO ICPT construction is shown in Fig. 3.3. It indeed can

regulate the output voltage, decrease the transform energy efficiency and increase the VA

rating. The multistage construction uses a DC/DC converter in power secondary circuit as

shown in Fig.3.4. Though the multi-stage ICPT construction works fine under constant

normal loading conditions, it can cause significant conduction and switching losses at light or

no-load conditions, and make the controller unsuitable for practical applications. Recently,

switching capacitor and switching inductor are used to change the characteristics of the

resonant tank in order to regulate the output voltage [16],[17]. The switching

capacitor/inductor ICPT construction is shown as Fig. 3.5 and has an advantage of uses in

the wide-rating load. But the switching capacitor/inductor control response is slow, and the

resonant current increase when changing the characteristics of the resonant tank. It is leading

to the increasing of power component current stress and conduction loss, as well as the


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decreasing of circuit transfer efficiency. In order to overcome the above shortcomings, this

paper proposes a novel ICPT, shown in Fig. 3.6, to improve the power factor, output voltage

regulation and efficiency. The proposed ICPT is mainly constructed by a VC-PFC and a LLC

resonant circuit. Using the proposed VC-PFC, the functions of power factor correction and

output voltage regulation can be achieved at the same time.

Fig 3.1 The traditional of frequency-fixed ICPT

Fig. 3.2 The PLL ICPT


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Fig. 3.3 The VCO ICPT

Fig. 3.4 The Multi-stage ICPT

Fig. 3.5 The switching capacitor/inductor ICPT

Fig. 3.6 The propose ICPT


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3.3. SYSTEM DESCRIPTION

Figure 3.7 shows the configuration of the proposed ICPT which consists of a presented VC-

PFC, an inverter with a PLL, a CCT, a var compensation and a rectifier. The leakage

inductance LP, the magnetizing inductance m L of the CCT and a resonant capacitor CP are

consummated the LLC resonant tank, as show in Fig. 3.7. The system operation is briefly

described as following: First, the VC-PFC is to convert the input AC power source in v into a

controllable DC voltage d V for sourcing the inverter. By controlling the DC voltage d V, the

output load power voltage regulation can be achieved. Next, the inverter provides a high

frequency square wave supply to the LLC resonant tank in order to obtain a sinusoidal wave.

Notably, the operation frequency of the inverter is at the resonant

Fig. 3.7 The configuration of the proposed ICPT

Frequency of the LLC resonant tank. Then, the sinusoidal wave is coupled to the load by the

CCT. After that, the var compensation is used to make the input impedance of the CCT

secondary side as a resistor for realizing the high efficiency in electric energy transfer.

Finally, the rectifier is used to convert the AC power of the secondary side into a DC power

for the load load R . Figure 3.8 is the equivalent model of the propose ICPT. We can see that

the CCT is equivalent to the combination of a leakage inductor L p , a magnetizing inductor

Lmand an ideal transformer [19-21], where ais the turns ratio of the ideal transformer Tr .

The CCT leakage inductor Lp ,magnetizing inductor m L and resonant capacitor Cr

constitute a LLC resonant tank. The leakage inductor Ls , compensation inductor Lr and

compensation capacitor Cs constitute a var compensation. The output rectifier with

capacitive filter can represent a load resistance transformer with equivalent es R value and

written as [22]


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Fig. 3.8 The equivalent model of the propose ICPT

(3.1)

The voltage transfer function of the LLC resonant tank of the propose ICPT can be defined

as

(3.2)

,where Vp

() v and Vs() v are the LLC resonant tank input voltage of the fundamental

frequency and the output rectifier voltage. That is

(3.3)

(3.4)

(3.5)

(3.6)


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According to (5), we know that the output voltage can be regulated by controlling the

operating frequency [12],[13], capacitor or inductor values [16]-[18]. However, these

traditional methods cannot obtain a high power factor nor reduce var rating. Thus, a voltage

controllable power factor corrector (VC-PFC) is presented in this paper. Using the proposed

VC-PFC, an ICPT can control the load voltage VO and maintain high power factor at the

same time. The VCPFC is composed of a rectifier 1 BR, and a buck-boost converter. The

buck-boost converter is composed of two power MOSFETs S1, S2, two diodes D1, D2, and

an inductor Lb . The VC-PFC circuit operation can be divided into the buck and boost

modes, as shown in Fig. 3.9. When the rectifier voltage vac is smaller than the controllable

DC voltage Vd , the VC-PFC works in the boost mode. In the boost mode, the power

MOSFET S1 is turned on and the diode D1 is reversed. The VC-PFC is equal to a boost

converter, which controlling the inductor current wave by control the duty cycle of the power

MOSFET S2 . When the power MOSFET S2 is turned on, the diodes D1, D2 are reversed as

shown in Fig. 3.10(a). The inductor Lb is charged up linearly by the voltage Vac, so that the

current in inductor Lb begins to increase linearly with slope Lb Vac/Lb. When the power

MOSFET S2 is turned off, the inductor Lb is discharge across the diode D2as shown in Fig.

3.10(b). In the boost mode, the relationship between the duty cycle and the DC bus voltage of

the proposed VC-PFC is defined as

(3.7)

where d1 is the duty cycle of the power MOSFET s1,d2 is the duty cycle of the power

MOSFET s2


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Fig 3.9. Waveforms of the VC-PFC

(b)

Fig. 3.10 Operation stages of boost mode


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When the rectifier voltage Vac is over the controllable DC voltage V d, the VC-PFC starts

working in the buck mode. In the buck mode, the power MOSFET S 2 is turned off and the

diode D2is reversed. The VC-PFC is equal to a buck converter, which controls the inductor

current wave by controlling the duty cycle of the power MOSFET S1. When the power

MOSFET S1is turned on, the diode D1is reversed as shown in Fig. 3.11(a). The inductor Lb

is charged up linearly by the voltage Vac-Vd , so that the current in the inductor Lbstarts to

increase linearly with slope Vac-Vd /Lb . When the power MOSFET S1 is turned off, the

inductor Lbis discharged across the diode D1as shown in Fig. 3.11(b). In the buck mode, the

duty cycle and the DC bus voltage Vdrelationship is defined as

(3.8)

Where d1 is the duty cycle of the power MOSFET S 1 , d2 is the duty cycle of the power

MOSFET S2

(b)

Fig. 3.11 Operation stages of buck mode


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Note that the operation frequency of the inverter in the proposed ICPT always maintains at

the frequency that little smaller than the LLC tanks resonant frequency. Thus, a small

inductance is obtained and the zero-voltage-switching (ZVS) in this inverter is achieved.

Additionally, high coupling efficiency for an inductive coupling system can also be obtained

at the same time.

CHAPTER 4

MATLAB/SIMULATION RESULTS

MATLAB is a high-performance language for technical computing. It integrates

computation, visualization, and programming in an easy-to-use environment where problems

and solutions are expressed in familiar mathematical notation. Typical uses include-

Math and computation Algorithm development Data acquisition

Modeling, simulation, and prototyping Data analysis, exploration, and visualization Scientific and engineering graphics

MATLAB is an interactive system whose basic data element is an array that does not require

dimensioning. This allows solving many technical computing problems, especially those with

matrix and vector formulations, in a fraction of the time it would take to write a program in a

scalar non-interactive language such as C or FORTRAN.

The MATLAB system consists of six main parts:

(a) Development Environment

This is the set of tools and facilities that help to use MATLAB functions and files. Many of

these tools are graphical user interfaces. It includes the MATLAB desktop and Command

Window, a command history, an editor and debugger, and browsers for viewing help, the

workspace, files, and the search path.


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(b) The MATLAB Mathematical Function Library

This is a vast collection of computational algorithms ranging from elementary functions, like

sum, sine, cosine, and complex arithmetic, to more sophisticated functions like matrix

inverse, matrix Eigen values, Bessel functions, and fast Fourier transforms.

(c)The MATLAB Language

This is a high-level matrix/array language with control flow statements, functions, data

structures, input/output, and object-oriented programming features. It allows both

"programming in the small" to rapidly create quick and dirty throw-away programs, and

"programming in the large" to create large and complex application programs.

(d) Graphics

MATLAB has extensive facilities for displaying vectors and matrices as graphs, as well as

annotating and printing these graphs. It includes high-level functions for two-dimensional and

three-dimensional data visualization, image processing, animation, and presentation graphics.

It also includes low-level functions that allow to fully customize the appearance of graphics

as well as to build complete graphical user interfaces on MATLAB applications.

(e)The MATLAB Application Program Interface (API)

This is a library that allows writing in C and FORTRAN programs that interact with

MATLAB. It includes facilities for calling routines from MATLAB (dynamic linking),

calling MATLAB as a computational engine, and for reading and writing MAT-files.

(f) MATLAB Documentation

MATLAB provides extensive documentation, in both printed and online format, to help to

learn about and use all of its features. It covers all the primary MATLAB features at a high

level, including many examples. The MATLAB online help provides task-oriented and

reference information about MATLAB features. MATLAB documentation is also available

in printed form and in PDF format.

(g) Mat lab tools

(i)Three phase source block


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The Three-Phase Series RLC Load block implements a three-phase balanced load as a series

combination of RLC elements. At the specified frequency, the load exhibits constant

impedance. The active and reactive powers absorbed by the load are proportional to the

square of the applied voltage.

Three-Phase Series RLC Load

(v) Three-Phase Breaker block

The Three-Phase Breaker block implements a three-phase circuit breaker where the opening

and closing times can be controlled either from an external Simulink signal or from an

internal control signal.

Three-Phase Breaker block

(vi)Integrator:

Library: Continuous

The integrator block outputs the integral of its input at the current time step. The following

equation represents the output of the block y as a function of its input u and an initial

condition y0, where y and u are vector functions of the current simulation time t.


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(vii)Breaker:

Implement circuit breaker opening at current zero crossing.

Library: Elements

Purpose: The Breaker block implements a circuit breaker where the opening and closing

times can be controlled either from an external simulink signal (external control mode), or

from an internal control timer (internal control mode).

A series Rs-Cs snubber circuit is included in the model. It can be connected to the

circuit breaker. If the Breaker block happens to be in series with an inductive circuit, an open

circuit or a current source, you must use a snubber.

When the breaker block is set in external control mode, a Simulink input appears on

the block icon. The control signal connected to the simulink input must be either 0 or 1 (0 to

open the breaker, 1 to close it).

When the Breaker block is set in internal control mode, the switching times are

specified in the dialog box of the block.

When the breaker is closed, it is represented by a resistance Ron. The Ron value can beset as small as necessary in order to be negligible compared with external components (a

typical value is 10 mohms). When the breaker is open, it has an infinite resistance.

(viii) Three-Phase Programmable Voltage Source

Implement three-phase voltage source with programmable time variation of

amplitude, phase, frequency, and harmonics


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Library: Electrical Sources

Purpose:This block is used to generate a three-phase sinusoidal voltage with time-varying

parameters. It can be programmed with the time variation for the amplitude, phase, or

frequency of the fundamental component of the source. In addition, two harmonics can be

programmed and superimposed on the fundamental signal.

(ix)Trigonometric Function

Specified trigonometric function on input

Library: Math Operations

Purpose:The Trigonometric Function block performs common trigonometric functions

(x)Three-Phase Transformer (Two Windings)

Implement three-phase transformer with configurable winding connections

Library: Elements

Purpose:


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The Three-Phase Transformer (Two Windings) block implements a three-phase transformer

using three single-phase transformers. The saturation characteristic, when activated, is the

same as the one described for the saturable Transformer block, and the icon of the block is

automatically updated. If the fluxes are not specified, the initial values are automatically

adjusted so that the simulation starts in steady state.

(xi)Three-Phase Transformer 12 Terminals

Implement three single-phase, two-winding transformers where all terminals are accessible

Library: Elements

Purpose:The Three-Phase Transformer 12 Terminals block implements three single-phase,

two-winding linear transformers where all the twelve winding connectors are accessible. The

block can be used in place of the Three-Phase Transformer (Two Windings) block to

implement a three-phase transformer when primary and secondary are not necessarily

connected in Star or Delta.

(xii)IGBT/Diode

Implements ideal IGBT, GTO, or MOSFET and antiparallel diode

Library: Power Electronics

Purpose: The IGBT/Diode block is a simplified mode of an IGBT (or GTO or

MOSFET)/Diode pair where the forward voltages of the forced-commutated device and

diode are ignored.


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4.1)MATLAB/SIMULINK RESULTS:


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CHAPTER5

CONCLUSION AND FUTURE SCOPE

CONCLUSIONS

This paper proposed a novel ICPT topology to improve the power factor, output voltage

regulation and efficiency. The proposed ICPT is mainly constructed by a VC-PFC and a LLC

resonant circuit. Additionally, the series compensated and series-parallel compensated are

used in the primary and the secondary of the coupling transformer to increase the coupling

efficiency and load range. Finally, a 1kW prototype is designed and simulated to assess the

feasibility and excellent performance of the proposed ICPT. The experimental results show

that under the 10 mm coupling distance, the power factor correction and output voltage

regulation can be achieved at the same time.

FUTURE SCOPE:


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