Three-Port Triple-Half-Bridge Bidirectional Converter With Zero-Voltage Switching

782 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 23, NO. 2, MARCH 2008

Three-Port Triple-Half-Bridge BidirectionalConverter With Zero-Voltage Switching

Haimin Tao, Jorge L. Duarte, Member, IEEE, and Marcel A. M. Hendrix, Member, IEEE

Abstract—A three-port triple-half-bridge bidirectional dc–dcconverter topology is proposed in this paper. The topology com-prises a high-frequency three-winding transformer and threehalf-bridges, one of which is a boost half-bridge interfacing apower port with a wide operating voltage. The three half-bridgesare coupled by the transformer, thereby providing galvanic iso-lation for all the power ports. The converter is controlled byphase shift, which achieves the primary power flow control, incombination with pulsewidth modulation (PWM). Because ofthe particular structure of the boost half-bridge, voltage varia-tions at the port can be compensated for by operating the boosthalf-bridge, together with the other two half-bridges, at an appro-priate duty cycle to keep a constant voltage across the half-bridge.The resulting waveforms applied to the transformer windings areasymmetrical due to the automatic volt-seconds balancing of thehalf-bridges. With the PWM control it is possible to reduce therms loss and to extend the zero-voltage switching operating rangeto the entire phase shift region. A fuel cell and supercapacitorgeneration system is presented as an embodiment of the proposedmultiport topology. The theoretical considerations are verified bysimulation and with experimental results from a 1 kW prototype.

Index Terms—Bidirectional dc–dc converters, multiport con-verters, soft-switching, three-port converters, triple-half-bridge(THB).

I. INTRODUCTION

ALTERNATIVE energy generators such as fuel cells haveslow dynamics. Energy supplied by sustainable sources

like solar and wind energy has an intermittent nature, whichnecessitates a battery-type storage capable of long-term energybuffering. A three-port energy management system can accom-modate a primary source and a storage, and combines their ad-vantages by utilizing a single power stage that has multiple in-terfacing ports. Having the two energy inputs, the instantaneouspower can be redistributed in the system in a controlled manner,which improves system dynamics and increases reliability. Asecond advantage of using a three-port system is that the pri-mary source only needs to be sized according to the averagepower consumed by the load, not necessarily to the peak power.Such operation is economically beneficial since per watt cost ofthe primary source is usually high, and thus it makes sense tooperate the primary source at the maximum power.

Although multiport converters are increasingly finding appli-cations in various systems like alternative generation [1], [2],

Manuscript received June 8, 2007; revised August 17, 2007. Recommendedfor publication by Associate Editor K. Ngo.

The authors are with the Electrical Engineering Department, Eindhoven Uni-versity of Technology, Eindhoven 5600 MB, The Netherlands (e-mail: [email protected]; [email protected]; [email protected]).

Digital Object Identifier 10.1109/TPEL.2007.915023

Fig. 1. Three-port energy management system–the storage smooths the powerflow of the primary source.

electric vehicles [3], uninterruptible power supply (UPS) sys-tems [4], and hybrid energy storage systems [5], limited workon multiport topologies has been reported due to the compli-cated structure and control [6]. One simple way to implement amultiport system is to use a transformer with multiple secondarywindings, thus providing multiple inputs [7] or outputs [8]. Mul-tiple-input converters based on the flyback topology [9] or fluxadditivity in a multiwinding transformer [10] are also reportedin the literature. An obvious limitation of these topologies is thatthey are not bidirectional in power. Hence, they are not suitedfor interfacing storage elements.

The converter presented in [3] uses a common dc bus to in-terconnect a main source and storage elements through indi-vidual buck/boost bidirectional switching cells. Furthermore, athree-switch boost converter is proposed in [11]. The topologyonly needs three power switches while providing bidirectionalinterfacing of two dc voltage sources with a dc-link. However,these two topologies do not provide electrical isolation.

Derived from the dual-active-bridge (DAB) topology [12],the three-port triple-active-bridge topology (TAB) reported in[4], [13], [14] and [15] has the property of being bidirectionalin power due to the active bridges at all the ports. The powerflow between the three ports can be managed by phase shiftingthe bridges. Using a -model of the transformer network fa-cilitates the system analysis [13]. The major drawback of thethree-port TAB converter, however, is that it is not able to main-tain soft-switching in case of wide operating voltages at theports, for example, when using supercapacitors and weak pri-mary sources like fuel cells and solar cells. In order to improvethe switching conditions, a method referred to as volt-secondsbalance control has been presented in [1] and [16], where a fur-ther degree of freedom in control is gained by shifting the gatesignals between the legs of the full-bridge to generate a duty

0885-8993/$25.00 © 2008 IEEE

TAO et al.: THREE-PORT TRIPLE-HALF-BRIDGE BIDIRECTIONAL CONVERTER 783

Fig. 2. Proposed triple-half-bridge (THB) bidirectional dc–dc converter topology for a three-port interface.

ratio controlled rectangular-pulse wave voltage. With this ap-proach soft-switching is achievable for all the switches. How-ever, in the case of a small duty ratio, i.e., when the voltage at theport (a supercapacitor) is high, the current stress of the converterincreases, especially in the case in which the supercapacitor isdesigned to support a considerable amount of transient power.

The three-port concept has also been recognized in recentlypublished works. In [5], a three-port converter having twocurrent-fed ports is used to interface with multiple energystorage elements; however, this converter cannot operate withzero-voltage switching (ZVS) when port voltages vary widely.In [17], a so-called tri-modal half-bridge converter is presentedbased on an isolated half-bridge converter topology. It has onlyone bidirectional port and thus does not support a regenerativeload.

This paper is a reviewed version of our previous work [18].Presented in the following is a triple-half-bridge (THB) bidirec-tional dc–dc converter consisting of three half-bridges, one ofwhich is a boost half-bridge [19] interfacing a power port withwide input range. A method to handle voltage variations at theport is proposed. Voltage variations can be compensated for byoperating the boost half-bridge and the other two half-bridgesat an appropriate duty cycle. A similar application of this tech-nique is found in [20] where a two-port dc–dc converter usingasymmetrical half-bridges was analyzed. Operating principlesand the control scheme are presented in the following sections.The topology is verified by simulation and with experimentalresults from a fuel cell and supercapacitor generation system.The converter topology is also of interest in three-port applica-tions like solar and battery energy generation and hybrid energystorage systems.

II. TOPOLOGY DESCRIPTION AND ANALYSIS

A. System Description

A fuel cell and supercapacitor generation system is used asan example to present the converter topology. Low-power fuelcells (around 1 kW) for residential power generation are recentlybecoming attractive [1], [2]. Due to the slow transient responseof fuel cells, to meet load changes and improve system dynamics

energy storage like batteries and/or supercapacitors is required.Such a fuel cell system needs multiple power ports.

Fig. 1 shows the system structure, where a three-port bidirec-tional converter interfaces the primary source, load and storage,and manages the power flow in the system. To be capable of inte-grating the three power ports, the power converter should havethe ability of matching different dc voltage levels, have bidi-rectional power flow, and enable galvanic isolation. In additionto this, soft-switching, preferably realized without auxiliary cir-cuits, is desirable in order to increase efficiency.

B. Triple-Half-Bridge Topology and Operating Principles

Due to the diversity of fuel cells and storage elements, theiroperating voltages can be very different. In addition, the acoutput, e.g., 50 Hz, 230 V, normally requires a 400 V dc voltageto feed an inverter, while fuel cell generators and superca-pacitor cells are low-voltage devices. Hence, a transformerto boost the low voltage to match a 400 V output voltage isdesirable. Besides, depending on applicable local standards,systems may require isolation between the fuel cell generatorand the load (inverter) in domestic applications. Taking theserequirements into account, as shown in Fig. 2, the proposedconverter consists of two half-bridges (HB1 and HB2) couplingthe fuel cell and load, and a boost half-bridge (HB3) interfacingthe supercapacitor. Note that the boost half-bridge includesthe inductor . A three-winding high-frequency transformeris used to link the three half-bridges. Being multifunctional,the transformer electrically isolates the three ports, and booststhe source and storage side low voltages to the load side highvoltage. In addition to these, the leakage inductances,and , are utilized as energy transfer elements.

The proposed topology inherits the advantages of thethree-port TAB converter in [13]. The operation of HB1,HB2, and HB3 is bidirectional. This implies that power canflow in any direction for all the ports. Moreover, low currentripple can be achieved for the supercapacitor port due to thecurrent-fed structure of HB3. Each bridge generates a phaseangle controlled high-frequency voltage and applies it to thecorresponding winding of the transformer. Fig. 3 shows thesimplified circuit model with the bridges replaced by equivalent


Fig. 3. Primary-referred �-model representation of the THB converter withthe bridges replaced by equivalent voltage sources (the magnetizing inductanceis neglected), where i ; i ; v and v are referred to the primary and denotedby i ; i ; v and v , respectively.

voltage sources, where the transformer and inductor networkare transformed from a T-model to a -model [15] (the mag-netizing inductance, which does not contribute to the powerflow, is neglected to simplify the analysis). Conceptually, theconverter can be viewed as a network of inductors driven byvoltage sources with controlled phase shifts between each other.The inductors in Fig. 2 represent the sums of the transformerleakage inductances and external inductors. As shown in Fig. 3,the voltages are shifted by and with respect to as thereference, and have the same duty cycle . The phase shift ispositive when the voltage lags the reference and negative whenit leads the reference.

Note, however, that the half-bridge configuration of the con-verter has limitations from the viewpoint of power rating. High-frequency current has to pass through the half-bridge capacitorsin full. In practice, a considerable number of high-frequencyfilm capacitors has to be connected in parallel with the elec-trolytic capacitors. This is of concern in low-voltage high-cur-rent applications. Therefore, the converter is generally not suit-able for high-power (say, higher than a few kilowatts) applica-tions.

C. PWM Control

For the proposed THB converter, as long as the dc voltageacross each bridge ( and as indicated in Fig. 2) iskept constant, the operation of the three-port converter is op-timal with respect to losses. It is possible to match the differentvoltage levels of the ports by choosing an appropriate numbersof turns for the windings, i.e.

(1)

where and are the numbers of turns of the windings;denotes the duty cycle of the upper switches and ;

and and are the voltages of the fuel cell, su-percapacitor and load, respectively. It is preferable that the fuelcell operates at the maximum power in order to achieve the max-imum utilization of the fuel, whereas the load side voltage is reg-ulated. Thus, we can assume that and keep a constantoperating voltage. The operating voltage of the supercapacitor,however, varies dynamically in a wide range. This variation canbe compensated for by adjusting the duty cycle while keeping

constant

(2)

The boost half-bridge structure of HB3 plays an essential rolein accommodating the variation of the supercapacitor voltage.The pole voltage formed by the half-bridge capacitors driftsin response to the duty cycle (automatic volt-seconds balanceof half-bridges, not possible with a full-bridge). However, fora proper operation of the converter, all the three half-bridgesshould be operated at the same duty cycle. As a result, in order tokeep all the three voltages the same wave shape, HB1 and HB2have to be implemented as half-bridges, not full-bridges. Theresulting high-frequency voltages applied to the transformer areasymmetrical when is not equal to 0.5. The waveforms areillustrated in Fig. 3, where the shadowed areas represent thevolt-seconds and are all equal. The peak values of all the threevoltages ( and ) are also equal to each other (both thepositive and negative peaks).

D. Soft-Switching Principle

In the -model representation, because of the direct powerflow link between any two of the three ports, the three-portmodel is decomposed into three two-port models. Fig. 4 plotsthe idealized operating waveforms [ will be defined in (8)].The voltages applied to the inductor network determine the cur-rents through the three inductors ( and ), which in turndetermine the current for each power port

(3)

Therefore, for a given set of phase shifts and voltages the cur-rents in the system can be determined.

As shown in Fig. 4, HB1 is switched at and , HB2 atand , and HB3 at and . The ZVS condition simply

says that a positive current should flow through the switch thatis going to be switched off [19]. For the half-bridges HB1 andHB2, the turn-off currents of the switches are only determinedby the half-bridge output current. Therefore, ZVS conditions forthe switches of HB1 and HB2 can be summarized as

(4)

The currents and can be solved analytically at theswitching instants similar to the results for the two-port systemin [20], and then one can judge whether they are positive or


Fig. 4. Idealized operating waveforms of the triple-half-bridge converter(' > ' > 0; ' = 0.5' ;L = L = L ).

negative. For example, for the case shown in Fig. 4holds

(5)

(6)

(7)

where represents all the primary-referred dc voltages acrossthe bridges, i.e.,

(8)

Provided that the current through each path satisfies the ZVScondition, certainly this is true for the combination of the cur-rent in two paths. For instance, by referring to Fig. 4 at theswitching instant (turn-off) of , we have [see

(5)] and 0 [see (7)], both satisfying the ZVS con-dition. The sum of these two currents yields (that is,

0), which obviously complies withthe ZVS condition. Similarly, the ZVS conditions for HB1 andHB2 can be verified for all the switching instants (also refer toFig. 4). For operating points different from the one in Fig. 4,ZVS conditions can also be proved in the same way.

For HB3, due to the boost half-bridge configuration, theturn-off currents of the switches should be examined sepa-rately. The turn-off currents are determined by and . Byusing (3), (6), and (7), we can calculate the turn-off current for

and

(9)

where is the average current supplied by the supercapacitorover one switching period, and is the amplitude of theripple current in , given by

(10)

where is the switching frequency. Obviously, the amplitudeof increases with the average current because more poweris transferred from or to the supercapacitor port. For the boosthalf-bridge HB3, ZVS conditions are achievable over the entirephase shift region in either direction of power flow when oper-ated in square-wave mode (i.e., 0.5) [19]. However, theturn-off currents of and are not equal because the boosthalf-bridge is not symmetrical [19]. As can be inferred from (9),a high ripple current in improves the switching conditionsfor both and . This can be easily realized by choosing alower inductance for . In fact, if is sufficiently large,ZVS is achievable for all operating conditions even if consid-erably differs from 0.5 (at least in steady-state).

By making and equal when referred to the primaryand keeping them constant, the peak values of the primary-re-ferred voltages ( and ) are equal as shown in Fig. 4, thepositive peaks all being equal to and the negativepeaks all being equal to . Therefore, the waveform of thecurrent through each inductor in the -model is trapezoidal, i.e.,has a flat top. In this optimized case, the peak current and rmsloss are reduced and soft-switching is achieved over the entireoperating region .

For comparison, Fig. 5 illustrates the operating waveformsfor different control methods when the dc voltages at the twoports do not match, i.e., the primary-referred voltages are notequal . As shown in Fig. 5(a), when controlled onlyby the phase shift as shown in Fig. 5(a), the current has a largepeak value and hard-switching also occurs [16]. As shown inFig. 5(b), with volt-seconds balance control (through duty ratioadjustment of the rectangular-pulse-wave) [1], soft-switching


Fig. 5. Comparison of different control methods in a two-port converter: (a)with only phase shift control, (b) with phase shift and volt-seconds balance con-trol (i.e., duty ratio control of the rectangular-pulse-wave) [1], and (c) with phaseshift and PWM control (asymmetrical wave control).

can be achieved and the peak current is also decreased. It is evi-dent that with the phase shift and PWM control method shown inFig. 5(c), both the peak and the rms values of the current are re-duced when transferring the same amount of power in an equiv-alent situation.

However, for a three-port application, to maintainsoft-switching, with method (c) only one port voltage isallowed to vary because all the three half-bridges have tooperate at the same duty cycle, whereas with method (b), all theport voltages may vary by applying the volt-seconds balancecontrol (through duty ratio adjustment) to those ports [16]. Inaddition, method (b) is more suitable for high-power applica-tions because it can use full-bridges for all the ports, therebyreducing the current stress to a half when compared with thehalf-bridge configuration, although the current waveform doesnot have a flat top. This method may have less high-frequencyharmonics and thus may have less magnetic losses. Method(c) is more suited to low-current or low-power applicationsbecause the half-bridge capacitors are the limiting factor. Bothmethods have their own advantages and disadvantages.

In the case of a battery-backed fuel cell system, one more de-gree of freedom is obtained because of the near constant oper-ating voltage of batteries. In this case, HB3 can be used to inter-face the fuel cell, allowing it to operate at different voltage levels

(fuel cells are weak voltage sources and have a droop charac-teristic) and thereby different output powers while maintainingsoft-switching.

E. Power Flow Calculation

The power flow in the three-port system has been investigatedin [4] and [15]. The power flow in the system is fully control-lable by the phase shifts between the half-bridges. The leakageand external inductances, and , act as energy transferelements. By using the volt-seconds balance of the inductors andamp-seconds balance of the capacitors [20], after some manip-ulations the power flow in the system is calculated to be

(11)

where 2 ; the phase shifts and are in radians;and are the powers delivered by the primary source,

load and storage, respectively. A positive value means supplyingthe power, whereas a negative value suggests consuming thepower. Note that for a lossless three-port system we have

; therefore is redundant. According to the aboveequations, when the switching frequency is fixed, the power flowis related to the phase shift, leakage inductance and duty cycle.In a phase shift controlled three-port system, the power flow ofthe source port and load port can be controlled simul-taneously by the two phase shift commands [15], while the re-maining power is balanced by the storage port–supplying or ab-sorbing the power (with a seamless transition) according to thesystem’s needs.

Due to the asymmetrical voltages applied to the transformerwindings, there is a limit on the maximum phase shift for thephase shift between any two ports, which is found to be

(12)

On the other hand, for a designed maximum phase shift ,the range of duty cycle variation (thereby the operating range ofthe supercapacitor voltage) is limited to

(13)

In practice, the operating phase shift is usually chosen to beless than 4 in order to limit the amount of reactive power (adrawback resulting from the phase shift controlled active-bridgetopology [12]). This can be done by choosing a smaller induc-tance. In addition, the variation of the supercapacitor voltage isusually within 50% to 100% of the rated voltage. Therefore, theconstraint imposed by (13) will not be a problem for a practicaldesign.


Fig. 6. Power flow between two ports of the THB converter versus phase shiftat different duty cycles D.

Fig. 6 plots the power flow as a function of the phase shift andduty cycle in a two-port system. The power flow in the figureis expressed in per unit (p.u.) with the base defined as

, where is the inductance between the two ports. Itclearly shows that as the duty cycle becomes either larger orsmaller, the maximum phase shift is constrained to a smallervalue and the power flow also becomes less.

F. Design Guidelines

Because of the -model representation, the THB convertercan be analyzed qualitatively based on a two-port system. Fig. 7shows the calculated peak (the absolute value of the positive ornegative peak, taking the larger one) and rms current throughthe transformer, and also the average source current in the two-port system when transferring the same amount of power (theaverage source current stays the same, as shown in Fig. 7).The parameters used for the calculation are 54 V,20 kHz and 3.5 H. It can be seen that both rms currentand peak current are minimized when operating in square-wavemode 0.5 . Therefore, limiting the dynamic range of theduty cycle (i.e., the operating voltage range of the supercapac-itor) would increase the overall system efficiency. However, thiswould compromise the utilization of the energy storage capacityof the supercapacitor.

Regarding the inductor sizing, for the same amount of trans-ferred power a lower inductance requires a smaller operatingphase shift, which results in a lower circulating current. There-fore, the chosen inductances of the THB converter should besmall. However, this will increase the sensitivity of the powerflow to the phase shift and thus requires a higher accuracy/res-olution of the control circuit. To allow for a desired maximumpower flow in the two-port system, the maximum inductanceis determined by the maximum/minimum operating duty cycle( or ) as shown in Fig. 8 (choose the smaller induc-tance value), where the inductance is expressed in p.u. with thebase defined as . With phase shift and PWMcontrol, a lower inductance leads to a lower current stress [20].The inductor sizing is a trade-off between the system loss andthe controllability. An operating phase shift of 15 to 45 is rec-ommended.

Fig. 7. Peak (I ) and rms (I ) current through the transformer and av-erage source current (I ) versus D when transferring the same amount ofpower (500 W) in the two-port system (parameters used for the calculation:V = 54 V, f = 20 kHz and L = 3.5� H).

Fig. 8. Maximum allowed inductance as a function of the the maximum/min-imum operating duty cycle (D or D ) in the two-port system.

III. CONTROL SCHEME AND POWER FLOW MANAGEMENT

As stated earlier, the power flow in the system is fully control-lable by the two phase shifts. Since in most cases it is not prac-tical to regulate the power of the load port, the load voltage isregulated instead. The control system shows a typical two-inputtwo-output plant. Two degrees of freedom (i.e., and ) areavailable for controlling the power flow in the system. The con-trol strategy aims at regulating the output voltage and the fuelcell power simultaneously. The control objectives are to keepa near-constant load voltage and to operate the primary source,the fuel cell, at a near-constant power (for example, the max-imum power of the source). The control system decouples theprimary source dynamically from the load–the power/current ofthe source is controlled to a specified value no matter what therest of the system is doing.

Three variables can be used as the control inputs, namely thephase shifts and the duty cycle . The power flowat the three ports can be fully controlled by the phase shiftcommands, whereas the control of is only for achieving


Fig. 9. Proposed multiloop control scheme for the triple-half-bridge converter.

soft-switching. Since the operating voltage of the supercapac-itor varies very slowly compared to other system variables, theadjustment of has little impact on the system dynamics (thepower flow control).

The modeling of the three-port converter has been described in[16]. The same method can be applied to the THB converter. Thethree-port converter can be modeled, using an averaging method,as three controlled dc current sources whose amplitudes are de-termined by the two phase shifts [16]. The current at each port canbe averaged over one switching cycle, being a function of the twophase shifts. We can assume that the voltages at the ports are keptconstant. Then average current is the power divided by the portvoltage. Thereby, the current source functions can be derived byusing the power flow equations in (11). They are nonlinear func-tions of the two phase shifts and should be linearized at the oper-ating point for a control-oriented model [16].

As shown in Fig. 9, the proposed control scheme employstwo PI controllers that are devoted to the regulation ofand , respectively. This control strategy is similar to the oneused in [1]. The controllers are implemented as

(14)

where 80 and 1 are the proportional gains and0.2 ms and 0.5 ms are the time constants. A low-pass filter(LPF) having a time constant of 1 ms is included in thecontrol scheme for smoothing the measured current .

The PWM controller can be an open-loop feedforward using(2). Alternatively, a third PI controller with narrow bandwidth(since changes very slowly) can be used to regulate insteadof the open-loop feedforward, or a combination of a PI controland a feedforward. The control scheme is fully digitalized andimplemented with a digital signal processor (DSP).

To manage the state-of-charge (SOC) of the supercapacitor,a SOC manager is integrated in this control scheme. Thanks tothe coupling between the SOC and the supercapacitor voltage,it only needs to monitor [21].

The storage acts as a power filter to smooth the power flow ofthe primary source. In this control scheme the energy deliveredby the supercapacitor is not controlled directly. The supercapac-itor sinks or sources the power difference between the load andfuel cell automatically. This is an autonomous system matchingthe load variations while the power of the primary source is

TABLE ICIRCUIT PARAMETERS OF THE THB CONVERTER

Fig. 10. Simulation results at steady state of the THB converter with ' =

0.1�; ' = 0.05�;D = 0.6, showing the voltagesv ; v and v generated bythe three half-bridges and the corresponding currents i ; i and i through thetransformer windings.

kept at the same level–leaving the storage compensating for thepower balance.


Fig. 11. Simulation results at steady state of the THB converter with ' =

0.1�; ' = 0.05�;D = 0.4, showing the voltages v ; v and v generatedby the three half-bridges and the corresponding currents i ; i and i throughthe transformer windings.

IV. SIMULATION AND EXPERIMENTAL VERIFICATIONS

A. Simulation Results

To verify the proposed THB topology and our theoretical con-siderations, the converter and control scheme were simulatedwith PSIM7.0. The simulation and design parameters are listedin Table I. The fuel cell has a nominal operating voltage of 54 V.A 400 V dc output is desired in order to supply a single-phaseinverter (230 V, 50 Hz ac output). The dc voltage across HB3,

, is chosen to be 42 V. To match these three voltages, thetransformer winding turns were selected as

, which is the closest match for the particular trans-former used in the prototype (the turn numbers should be inte-gers). The inductances and can be designed in sucha way that the power rating of each port is identical, i.e., in the

-model (primary-referred values). In orderto limit the rms and peak current through the transformer whenoperating at a small or large duty cycle (see Fig. 7), the max-imum and minimum limiting duty cycles are set to be0.7 (for maximum ) and 0.3 (for minimum ),respectively. For a supercapacitor, a simple calculation showsthat such an operating range would utilize over 80% of the en-ergy storage capacity of the supercapacitor.

Fig. 10 shows the simulated results of the steady-state opera-tion at 0.1 0.05 and 0.6 25.2 V).Fig. 11 presents the results at 0.1 0.05 and

0.4 16.8 V). The parasitic parameters of the cir-cuit (including the ON-resistance of the switches, ESR of thecapacitors, winding resistances, etc.) were also taken into ac-

Fig. 12. Simulation results of the power flow control in the THB converter,showing the current waveforms i ; i and i : (a) in response to a stepincrease of 250 W in the load, and (b) in response to a step decrease of 250 Win the load.

count for the simulation. The waveforms shown in both figuresare asymmetrical and are consistent with the theoretical analysis(see, e.g., Fig. 4).

To show the effectiveness of the control scheme, Fig. 12 dis-plays the simulated results of closed-loop power flow controlunder step load changes. The changes were carried out by sud-denly switching on and off an extra resistor in parallel with theload. In Fig. 12(a) the converter was operated at 1 kW and astep-increase of 250 W in the load (transient) demand was ap-plied at 30 ms, whereas Fig. 12(b) shows the step-decrease inthe load demand from 1 kW to 750 W initiated at 30 ms. As canbe seen, in both simulated cases the power delivered by the fuelcell remains unchanged at 1 kW after some transients. The sur-plus and deficiency in the load power demand are managed bythe supercapacitor. Since the voltages at the three ports remainconstant during the load variation, the currents are representa-tive for the powers. Note that the high-frequency ripples inand are already filtered out.

B. Experimental Results

A laboratory prototype was constructed, laid out for 1 kWmaximum power at 20 kHz switching frequency. The switching


Fig. 13. Experimental results at steady-state operation of the THB converterwith ' = 0.1�; ' = 0.05�;D = 0.6, showing (a) voltages generatedby the half-bridges, and (b) currents through the transformer windings (in bothfigures Ch.1: fuel cell side, Ch.2: load side, Ch.3: supercapacitor side).

devices were implemented with power MOSFETs for all thethree bridges. The three-winding transformer was designed andbuilt using a E65 ferrite core. High-frequency litz wires wereused for all the windings. The leakage inductances of the woundtransformer were too small so that external inductors were de-signed and connected in series with the windings. It would alsobe possible to integrate the external inductors into the trans-former–a transformer having a relatively large leakage induc-tance. However, such a transformer is difficult to build and thesize would not be smaller than a normal transformer. Problemslike localized over-heating of the magnetic core caused by theleakage flux have to be carefully considered. For this reason,external inductors were used in the implementation. In the ac-tual experimental prototype, the inductances are not exactly thesame as the ones in the simulated circuit.

A polymer electrolyte membrane (PEM) fuel cell (maximum1 kW) was used as the generator. The fuel cell has a built-indc-dc converter and provides a roughly regulated output. It wasoriginally used as a battery charger for a 48 V system. A 145 Fsupercapacitor was used as the storage. The control scheme wasdigitally implemented with a TMS320F2812 DSP controllerfrom Texas Instruments.

To illustrate the steady-state operating waveforms, measure-ment results of the voltages generated by the three half-bridges

and and the corresponding current waveforms

Fig. 14. Experimental results at steady-state operation of the THB converterwith ' = 0.1�; ' = 0.05�;D = 0.4, showing (a) voltages generatedby the half-bridges, and (b) currents through the transformer windings (in bothfigures Ch.1: fuel cell side, Ch.2: load side, Ch.3: supercapacitor side).

and are shown in Fig. 13 for the operating point of0.1 0.05 0.6. In Fig. 14, the converter was op-erated at 0.1 0.05 0.4. As shown, thewaveforms are asymmetrical. The volt-seconds are automati-cally balanced by the half-bridges. A comparison between thesimulation and experimental waveforms shows the consistencybetween the simulated and measured results.

The power flow control in response to step load variationsunder closed-loop operation is shown in Fig. 15, where (a)shows the currents of the three ports when a step increasein the load occurs, and (b) shows the response waveformswhen the load is step-reduced. The experiments were done byshifting between two set points of an electronic load operatingin constant resistance mode. The output voltage is regulatedto a constant value. As can be seen, the desired power flowmanagement is achieved, i.e., drawing a constant power fromthe fuel cell while variations in the load demand take place.The high-frequency ripples in the fuel cell side and load sidecurrents and were filtered out by the capacitors inparallel with the fuel cell and load bridges (see Fig. 2) andthe inductances of the connecting wires between the powerconverter and the sources.

The efficiency of the THB converter at different (thus dif-ferent ) is shown in Fig. 16. With phase shift and PWM con-trol, a substantial increment in the efficiency can be observed.


Fig. 15. Experimental results of the power flow management in the THB con-verter with closed-loop control, showing (a) response to a step increase in theload, and (b) response to a step decrease in the load (in both figures Ch.1: fuelcell side, Ch.2: load side, Ch.3: supercapacitor side).

Fig. 16. Efficiency of the power stage of the THB converter with phase shift andPWM control and only with phase shift control at different operating voltagesof the supercapacitor when the fuel cell power P = 410 W and the loadR = 420.

In all the measured operating points, the power of the primarysource is kept at around 415 W and the load is a 420 re-sistor. The converter was operated with closed-loop control. Theefficiency is calculated as %,where is the power consumed by the load, and and

are the powers delivered by the fuel cell and the supercapac-itor, respectively. In the measured cases in Fig. 16, the average

power from the storage port is low and stays positive. Underthis operating condition, the PWM control is especially impor-tant because it can keep rms value of the current of the storageport very low (as shown in Fig. 4), thus significantly reducingthe rms loss (as the switch voltages are quite low, rms currentbased losses usually dominate). If the PWM control is not ap-plied, will have a near-triangular shape when the storage portvoltage is either high or low (far from the matched value, seealso Fig. 5). This results in a very high rms value of (thoughthe real power of is almost zero), leading to the decrease ofthe efficiency as shown in Fig. 16.

V. CONCLUSION

A transformer-coupled three-port bidirectional converter im-plemented with three half-bridges has been proposed in thispaper. A control method has been presented for achieving soft-switching over a wide input range. In addition to the phase shiftcontrol, a PWM control method is applied to the triple-half-bridge converter. The particular structure of a boost half-bridge,which interfaces the port having a wide operating voltage (e.g., asupercapacitor), makes it possible to handle voltage variations atthis port by adjusting the duty cycle of all the three half-bridges.With this approach, the operation of the converter is optimizedwith both current stress and rms loss being reduced. Moreover,soft-switching conditions for all switches are achievable overthe entire phase shift region. A control scheme based on multiplePI regulators manages the power flow, regulates the output, andadjusts the duty cycle in response to the varying voltage on theport. Simulation and experimental results were presented, vali-dating the effectiveness of the proposed converter and its controlscheme.

REFERENCES

[1] H. Tao, A. Kotsopoulos, J. L. Duarte, and M. A. M. Hendrix, “Asoft-switched three-port bidirectional converter for fuel cell and su-percapacitor applications,” in Proc. IEEE Power Electron. Spec. Conf.(PESC’05), Recife, Brazil, Jun. 2005, pp. 2487–2493.

[2] H. Tao, A. Kotsopoulos, J. L. Duarte, and M. A. M. Hendrix,“Multi-input bidirectional DC-DC converter combining DC-link andmagnetic-coupling for fuel cell systems,” in Proc. IEEE Ind. Appl.Soc. Conf. Annu. Meeting (IAS’05), Hong Kong, China, Oct. 2005,pp. 2021–2028.

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[14] H. Tao, A. Kotsopoulos, J. L. Duarte, and M. A. M. Hendrix, “De-sign of a soft-switched three-port converter with DSP control for powerflow management in hybrid fuel cell systems,” in Proc. 11th Eur. Conf.Power Electron. Appl. (EPE’05), Dresden, Germany, Sep. 2005, pp.1–10.

[15] J. L. Duarte, M. Hendrix, and M. G. Simoes, “Three-port bidirectionalconverter for hybrid fuel cell systems,” IEEE Trans. Power Electron.,vol. 22, no. 2, pp. 480–487, Mar. 2007.

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Haimin Tao was born in China in 1976. He receivedthe B.S. degree in electrical engineering and theM.S. degree in power electronics from ZhejiangUniversity, Hangzhou, China, in 2000 and 2003, re-spectively, and the Ph.D. degree from the EindhovenUniversity of Technology (TU/e), Eindhoven, theNetherlands, in 2008.

From 2003 to 2004, he worked for Philips LightingElectronics, Shanghai, China. His current researchfields of interest include multiport and bidirectionalconverters, grid-connected inverters, digital control,

soft-switching techniques, and sustainable energy systems.

Jorge L. Duarte (M’00) received the M.Sc. degreefrom the University of Rio de Janeiro, Rio de Janeiro,Brazil, in 1980 and the Dr.-Ing. degree from theInstitut National Polytechnique de Lorraine (INPL),Nancy, France, in 1985.

He has been with the Electromechanics and PowerElectronics Group, Technical University of Eind-hoven, Eindhoven, The Netherlands, as a member ofthe scientific staff, since 1990. During 1989, he wasappointed a Research Engineer at Philips LightingCentral Development Laboratory, and since October

2000 he has also been a consultant Engineer at Philips Power Solutions,Eindhoven. His teaching and research interests include modeling, simulationand design optimization of power electronic systems.

Marcel A. M. Hendrix (M’98) received the M.S.degree in electronic circuit design from the Eind-hoven University of Technology (TU Eindhoven),Eindhoven, The Netherlands, in 1981.

He is a Senior Principal Engineer at PhilipsLighting, Eindhoven. In 1983, he joined PhilipsLighting, Eindhoven, and started to work in thePre-Development Laboratory, Business GroupLighting Electronics and Gear (BGLE&G). Sincethat time he has been involved in the design andspecification of switched power supplies for both

low and high pressure gas-discharge lamps. In July 1998, he was appointed apart-time Professor (UHD) with the Electromechanics and Power ElectronicsGroup, TU Eindhoven, where he teaches design-oriented courses in power elec-tronics below 2000 W. His professional interests are with cost function basedsimulation and sampled-data, nonlinear modeling, real-time programming, andembedded control.

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