Soft Switching Bidirectional Converter for Battery charging discharging

7/27/2019 Soft Switching Bidirectional Converter for Battery charging discharging

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Soft Switching Bidirectional Converter for Battery

Discharging-Charging

E. Sanchis-Kilders(1), A. Ferreres(1), E. Maset(1), J.B. Ejea(1), V. Esteve(1), J. Jordán(1), A. Garrigós(2), J. Calvente(3)

(1) Dpt. Ingeniería Electrónica

E.T.S.E. / Universitat de València

Dr. Moliner, 50,E-46100 Burjassot, SPAIN

e-mail: [email protected]

(2) División Tecnología Electrónica

E.P.S.E. / Uni. Miguel Hernández

Avda. del Ferrocarril s/n,E-03202 Elche, SPAIN


(3)Dpt. d’Enginyeria Electrònica,

Elèctrica i Automàtica

E.T.S.E. / Uni. Rovira i Virgili,E-43007 Tarragona, SPAIN


Abstract — This paper presents the results of a project that

looked after a high efficiency bidirectional converter which

could be used as a battery discharge/charge regulator when the

bus voltage is above the battery voltage. High efficiency, high

stability and simplicity are the main goals, no galvanic isolation

is required. Taking into account all these parameters, our

proposed solution has been a new topology based on a Boost

converter with coupled inductors. The use of a bidirectional

converter reduces the mass of the overall charge/discharge

subsystem and lowers cost and component count. In the project,

its use is intended for space applications, but telecom,

automotive or similar applications can also benefit of this new

concept.

I. I NTRODUCTION

High power buses used nowadays are normally backed up

with batteries, which have to be charged and dischargeddepending on the bus power demand. These systems are

common in telecommunications applications, space platforms

and automotive electrical buses. In space applications the power source is usually a DC voltage and therefore DC-to-DCconverters are needed. Telecom applications need normally an

input rectifier (AC-to-DC) to feed the bus. Modularity also

requires the ability of connecting modules in parallel without

complex additional circuitry. The easiest way to comply withthis requirement is to provide the converter with current

regulation. Current regulated converters with common voltage

loop can be immediately parallelized, short circuit protections

are inherent to the control loop and current sharing is alsoguaranteed. The use of a bidirectional unit saves the need of

one additional converter for the charge operation. This

translates into less mass and volume and finally less cost [1].

The prototype designed and built complies with thefollowing specifications:

Input voltage (battery voltage) Vi = 82V..100V

Output voltage (bus voltage) Vo = 120V±0.5%(no galvanic isolation required)

Switching frequency f s = 100kHz

Output power Po = 1kW

Efficiency η>95%

To achieve a high MTBF, two basic principles have to be

observed: all components must have a low temperature riseand the circuit must be as simple as possible. One way to

assure a high MTBF is to apply strict derating rules [2].

II. THE COUPLED I NDUCTOR BOOST CONVERTER

The selected topology was already introduced in [3] and [4].This topology is a boost and therefore a step up converter,

with output filter and with its two inductors (input inductor

and output filter inductor) coupled together. Bidirectionality

was achieved by using bidirectional switches.The use of the coupled inductors assures that the converter

behaves, from the control point of view, as a minimum phase

system. This means that it does not have a right half plane

zero, which can create stability problems. These systems can be easily stabilized like the buck converter. Conductance

Control [5] can be easily applied and parallelizing is then

straightforward thank to the current control loop.

Figure 1. Bidirectional Coupled Inductors’ Boost.

Fig. 1 shows the scheme of the converter. It already

includes the two MOSFET which permit bidirectional flow of

power. Bidirectional behavior is immediate and does not need

additional control circuitry. As soon as the output voltageincreases, due to an additional flow of power to the load from

another source (photovoltaic panels or fuel cell, for example;not shown in Fig. 1), the exceeding power will flow from

output to input of the converter and charge the battery (hereVin). If M2 is substituted by a diode pointing to the output

like a boost diode, the converter would be unidirectional but

retain its other advantages (minimum phase system).

6030-7803-9547-6/06/$20.00 ©2006 IEEE.



The design of the coupled inductances is explained in [3].

The coupled inductors have been substituted by two real

components which are a transformer with a given magnetizinginductance and leakage inductance. As the value of the

leakage inductance (54µH) is of the same order of magnitude

as the value of the magnetizing inductance (108µH) an

additional inductance, L b, has to be placed in series with the

transformer “to increase its leakage inductance”. We couldcall the transformer a “flyback-forward-transformer” because

it is like a classical one but with a small magnetizing

inductance which is discharged to the output. Energy transfer is done in an inductive and direct way as for example in the

buck converter or the two-inductor boost [4]. This can be seen

by studying its operation mode. The two switching states in

continuous conduction mode are shown in Fig. 2. It can beseen that current is always flowing from the input to the

output.

Figure 2. The two operating stages of the bidirectional boost converter

with coupled inductors. Please note that there is always a direct current path

from input to output.

During the ON state we are charging the magnetizinginductance of the transformer and also transmitting current tothe output (ILb) and during the OFF state we are discharging

the magnetizing inductance to the output and again also

transmitting current to the output (ILb).

The DC levels of the currents are determined by the power balance of the converter and the current ripple (AC) depends

on the value of the inductances. The peak currents at the input

and the output are described by Eq. 1 and Eq. 2. Please note

that Eq. 2 has two terms for the ripple, the one from themagnetizing inductance and the one from the output

inductance, L b, reflected to primary.

_ 0max

1

2

s i Lb pk ON

p b

N V I I t

N L

= + (1)

2

_ max

1 1

2 2

s i iin pk i ON ON

p b mag

N V V I I t t

N L L

⎛ ⎞= + +⎜ ⎟⎜ ⎟

⎝ ⎠

(2)

This behavior leads us to the only drawback of the circuit

which is the mass (and volume) of its magnetic elements. The

mass of the transformer is larger than the one of an inductor because of its additional secondary (it is a transformer!) and

the primary needs additional copper because of the not

negligible magnetizing current (large magnetizing current!).The only comparable topology would be the two-inductor-

boost [6] which does not couple both inductors and therefore

saves the secondary of the transformer. It is also a minimum

phase system and has a step up transfer function. But thetransistor, in case of unidirectional operation is floating and

this requires additional driving circuitry. By comparing the

energy stored in the magnetic elements of both topologies

designed for the same specifications and with the same currentripple, we found out that the two inductor boost stores about

8mJ and the coupled inductor boost stores 5mJ. We think that

this data means that the mass of the magnetic elements will beat least similar, because the coupled inductors boost stores less

energy but needs a kind of transformer. But more detailed data

must be compared.

The circuit itself shows also that the DC voltage at thecapacitor C is equal to the output voltage. In fact Co and C are

connected together by a short seen from the DC point of view.Fig. 3 shows the simulated waveforms of the converter. We

can see how the current through C confirms that it isdischarged during the ON state and charged during the OFF

state.

Figure 3. Simulated converter waveforms at full power: Upper

waveforms: Iin (=Ibat; red), ILb (blue), IC (green), ICd (magenta) Lower

waveforms: Vo (red) and VC (blue). The names are after Fig. 1. The DC value

of VC is Vo.

III. SMALL SIGNAL A NALYSIS

The coupled inductor boost converter has the same DC

transfer function as the classical boost converter and turn ratiodoes not affect the DC transfer ratio.

1

1o i

V V D

=−

(3)

604



The damping (R d-Cd) shown in Fig. 1 is needed for stability

purposes and does not imply appreciable losses (PRd = 0.5W at

P0 = 1kW). Without damping, the minimum phase behavior isgiven only under a given condition [3] that is:

1

1

s

p

N

N D>

−(4)

When damping the resonance properly, then the aboveinequality can be overridden. In fact, damping provides us

with smaller inductors by avoiding the fore mentioned

condition (Eq. (4)). By using the design equations given in [3]

a perfect and stable behavior can be expected under all loadand input voltage range. Fig. 4 shows the frequency response

and how damping assures stability.

Figure 4. Calculated frequency response of the converter ( Lbi

d

) with

(solid) and without (dashed) damping.

IV. BIDIRECTIONAL SOFT SWITCHING

Although theoretically no reverse recovery is present in the

circuit due to the use of two MOSFET as switches, soft

switching has been investigated due to the fact that the bodydiodes of both transistors can conduct during short periods of

time. This happens because we have to provide some dead

time before switching on each switch to avoid short circuit.

This could force the body diode on during these transitionsand can lead to reverse recovery losses.

In order to apply soft switching to the converter a new

bidirectional soft switching circuit has been proposed. Based

on a classical soft switching circuit [7], bidirectionality has been achieved by blending the soft switching circuits of the

boost topology with the one of the buck topology and of course using transistors instead of diodes. It has to be taken

into account that the topology of Fig. 1 is a buck converter with input filter when working in reverse mode and a boost

with output filter in direct mode. The design of the circuits is

as explained in the references [7] and [8].

Figure 5. In red soft switching cells for both transitors based on

classical soft switching cells (resonant capacitor, connected between drain

and source of each MOSFET, is not shown for simplicity).

Fig. 5 shows the two soft-switching cells like in the classical

design for each active switch. In order to simplify the circuit

we propose a new single circuit (Fig. 6) taking into accountthat: a) the connection points are the same, b) the diodes

correspond to the body diodes of the MOSFET and c) one

single inductor can be used. Resonant inductor Lr , resonates

with the parasitic drain source capacitance of the MOSFET

which can be increased by adding external capacitance (tosimplify not shown in Fig. 5, but shown in Fig. 7).

Figure 6. The new proposed soft switching bidirectional cell.

When adding the bidirectional soft switching cell to the bidirectional boost circuit we end up with the circuit shown inFig. 7. Note that we have already added the resonant

capacitors in parallel with the drain-source capacitance. A

single resonant inductor is needed and two auxiliary MOSFET

are used. Therefore one additional floating driver is neededfor Maux2.

Figure 7. The bidirectional BOOST with the new proposed soft

switching bidirectional cell added.

This circuit can be driven with two different strategies. The

one described in [7] and a much newer one proposed in [8]

which simplifies the driving timing and avoids the thermal

605



run-away of active components. Both strategies provide

similar (small) efficiency enhancements and both have been

developed and tested with high voltage, low current circuits.The classical strategy has been chosen because the power

layout does not change. This means that the auxiliary

transistors are driven with a short on time corresponding to thetime required by Lr to charge up to Iin plus a quarter of the

resonant period. This happens just before the main switches

are switched on (see [7]). After that, the inductor discharges

linearly.In the following figure (Fig. 8) we show the basic

waveforms of the converter with the soft switching cell. We

can see that the auxiliary transistors are switched on just

before the main transistors are switched on and therefore asmall time gap ([t1-t2] and [t4-t5]) is left between both main

transistors. The more detailed discussion on this waveform can

be found in [7].The main problem encountered in our circuit is shown by

the red trace. Due to the reverse recovery of the body diodes

of the auxiliary transistors and the resonance of Lr and the

parasitic capacitors of the circuit, the current through Lr doesnot stay at 0A, but reaches a relatively large value and remains

at this constant level during the time intervals of [t0-t1] and [t3-t5]. This current is circulating through closed loops (M1-Maux1-

Lr and M2-Maux2-Lr ) and is not providing energy to the output.Therefore it is only dissipating energy and contributing to the

overall losses of the circuit. In our driving scheme we also

drive Maux2 during direct operation of the converter for controlsimplicity purposes, although this action does not provide soft

switching to M2. From the “ideal behavior” point of view

(demonstrated in simulation where no recovery currents were

present) no current change should happen through Lr by thisadditional on switching of Maux2. But real circuit shows how

the current changes its sign and unfortunately reaches a new

constant level different form 0A (see [t1-t2] in Fig. 8).

Figure 8. Main waveforms of the soft switching boost with directcurrent flow (boost mode). The red trace shows the real current flowing

through the resonant inductor.

These circulating currents which happen due to the reverse

recovery of the body diodes of the auxiliary MOSFET appear to be relatively large and high losses are induced in these

transistors. Experimental waveforms are shown in Fig. 8. A

heatsink becomes necessary for the auxiliary transistors. This

problem has been already mentioned in [7] and [8]. In [7] it issolved with an additional diode in series with L r to block the

inverse current, but in our case this makes the cell

unidirectional. In [8] they do not add the additional diode butreduce the di/dt through Lr to reduce the reverse recovery

through the diodes. However the experimental waveforms

presented in [8] still show this circulating current.

Figure 9. Switching waveforms of bidirectional soft switching cell.

Please observe how the current through Lr stays on an unacceptable level

instead to return to zero after switching OFF. Ch1: VDS (50V/div), Ch2: ID

(5A/div), Ch3 ILr (5A/div), Ch4: VIsens (5V/div). P0=1kW, Vi=82V.

To solve this problem, several solutions have been tried.

First is to go back to the unidirectional circuit what means that

the soft switching cell has to bee duplicated (one for eachtransistor). Unidirectionality is achieved with an additional

diode put in series with the resonant inductor. The circulating

current disappears completely.

Figure 10. Switching waveforms of unidirectional soft switching cell due

to added diode. Ch1: VDS (50V/div), Ch2 ILr (10A/div), Ch3: ID (10A/div),Ch4: VIsens (5V/div). P0=1kW, Vi=82V.

Of course bidirectionality could be preserved if the diode isreplaced by a switch capable of blocking current in one of the

two directions depending on if we are working in buck mode

606



or boost mode. Complexity of this scheme is unacceptable for

our application and this solution has been discarded.

Another solution is to connect a saturable inductor in serieswith the resonant inductor. This solution improves the

efficiency but still does not avoid completely the circulating

current through the circuit (see Fig. 11).

Figure 11. Switching waveforms of bidirectional soft switching cell.Saturable inductor has been added in series with Lr . Please observe how the

circulating current through Lr has been reduced. Ch1: VDS (50V/div), Ch2: ID (5A/div), Ch3 ILr (5A/div), Ch4: VIsens (5V/div). P0=1kW, Vi=82V.

No efficiency improvement is observed with any of these

solutions as far as we have measured in our lab (see Fig. 12).

Efficiency

93,0%

94,0%

95,0%

96,0%

97,0%

98,0%

99,0%

200,0 300,0 400,0 500,0 600,0 700,0 800,0 900,0 1000,0 1100,0

Po [W]

η

Vi=85V HSW

Vi=100V HSW

Vi=85 SSW w/diode

Vi=100 SSW w/diode

Vi=85 SSW w/Lsat

Vi=100 SSW w/Lsat

Figure 12. Efficiency of the converter under hard switching conditions

(HSW), unidirectional soft switching conditions (SSW w/diode) and

bidirectional soft switching conditions (SSW w/Lsat).

From Fig. 12 we see that the hard switching circuit is the

most efficient. At high power levels efficiency is increased

about 0.5% by the classical soft switching cell. The saturableinductor leaves efficiency below 96% in any case.

Just for comparison purposes we present also the measured

efficiency of a 5kW interleaved boost converter with passive

soft switching [9]. We see how the efficiency is just a little

better (+0.5%) with soft switching under high power conditions, but otherwise we reach a very high efficiency

under hard switching conditions. In this case we were using a

passive circuit, but if an active circuit is required, thecomplexity is not always worth the efficiency increase due to

the loss of reliability.

Efficiency Interleaved Boost

96,0%

96,5%

97,0%

97,5%

98,0%

98,5%

99,0%

500,0 1.000,0 1.500,0 2.000,0 2.500,0 3.000,0 3.500,0 4.000,0 4.500,0 5.000,0

Po [W]

η

Vi=82V

Vi=82V sw1D

Vi=82 sw2D

Figure 13. Efficiency of an interleaved BOOST converter under hard

switching conditions (Vi=82V), under soft switching conditions with a single

diode (Vi=82V sw1D) and under soft switching conditions with two diodes(Vi=82V sw2D). The second diode was placed only to increase reliability.

These experimental results have made us think of the realuse soft switching circuits, which add complexity with a small

efficiency benefit. Our experience confirms that with high

current applications, diodes always behave “badly” in real live

and it is very difficult to overcome the losses introduced bythis behavior. In our application, due to the fact that we had a

low voltage (200V) we could use Schottky diodes which have

a much better reverse recovery behavior, but for higher

voltages the reverse recovery losses become important. As allsoft switching circuits need somewhere a diode which usually

has to switch in a very fast manner, the problem is normally

shifted from the main switch to the auxiliary switch.

Efficiency improvements are therefore quite small. In somevery specific cases soft switching can be used but normally it

is only useful under determined load and line conditions.

In our application we have concluded that the bidirectional

soft switching cell does not provide the necessary practical benefits in order to be implemented in our power circuit. Only

one cell for each transistor provides a very small improvement

but increases complexity of the circuit a lot. Following this

criteria we could also place one passive soft switching cell for each transistor like the one used in the interleaved boost [9].

But only 0.5% efficiency improvement can be expected.

If we control the di/dt of the transistor (here slow turn on

and fast turn off) we can optimize the losses due to the reverserecovery of the diodes in the circuit and no additional element

must be added to the circuit. This provides us with high

efficiency, a simple circuit and fewer components. Further

measurements are going to be conduct in order to find theoptimum switching times.

V. EXPERIMENTAL R ESULTS

The built prototype is shown in Fig. 14. The system

includes the main transistors on the left side attached to a

heatsink and the auxiliary transistors of the soft switching

circuit and the resonant inductor in lower left area of the PCB.

An additional heatsink for these two transistors to avoid

overheating was required in the final version. On the upper side we find the transformer with the designed magnetizing

inductance and the additional inductor at its right side.

607



Figure 14. Power section of the prototype. Additional heatsink had to be

added to the auxiliary switches if no diode was placed in series with L r . Main

power switches (TO-247) are at the left, auxiliary switches are at the bottom-

left (TO-220AB). At the top-left is the transformer and at its right the

additional inductor, L b. The rectangular blue shape between the two yellow

MKT capacitors at the lower right is a LEM sensor.

Experimental results show that the converter is bidirectionalas expected and that switching occurs in a soft manner in both

directions. We tested the unit with the soft switching circuit in

direct and reverse mode of operation observing that both waysof operation performed correctly. But unfortunately efficiency

was not improved. The oscillograms show (Fig. 15) that the

soft switching circuit actuates during the turn on of the devices

achieving a zero-current-transition. The turn off is not assmooth but still reduces switching noise.

Figure 15. Switching waveforms of M1 in direct mode and reverse mode

with bidirectional soft switching cell. VDS 50V/div; ID 5A/div. In direct mode

P0=1kW and in reverse mode P0=410W

Safe operating area of the main transistor under hard and

soft switching conditions is shown in Fig. 16. It is interesting

to see that the on transition is not too hard in the hard

switching mode except for the current spike. Of course under

soft switching operation the on transition is loss less. The off transition is hard in hard switching mode and a little bit

smoother in soft switching mode.

Figure 16. Safe Operating Area (SOA) of the main transistor under direct

operation and handling P0=1kW. On the left we see the hard switching and onthe right the soft switching transition (y-axis 50V/div; x-axis 5A/div).

VI. CONCLUSION

A new bidirectional step up converter with coupled

inductors has been built and tested. When coupling its two

inductors and damping the resonance, a minimum phasesystem results. Conductance control can be applied and

bidirectional behavior is naturally achieved with this

converter. At system level, this topology and thank to its

bidirectionality saves volume, mass and cost. A new soft

switching cell have been designed and tested but no efficiency

improvements have been gained. But the new bidirectionalsoft switching cell works as expected and provides soft

switching to the converter in either direction of power flow

softening the on transitions. Only when using two separate

unidirectional soft switching cells, one for each transistor, wecan expect a slight efficiency increase at the higher power end.

We have finally chosen to not use any soft switching cell andhard switch the circuit. Reverse recovery problems can be

minimized by controlling the di/dt of the transistors (here slow

on transition and fast off transition).

ACKNOWLEDGMENT

The authors would like to thank:

• Spanish Ministry of Science and Technology

(SubD.G.P.I.) which has supported this researchwith the project ref: ESP 2003-08905-C03-03.

• LEM and Mr. Stefan Lüscher for providing uswith samples for the prototype and good advices

for the right selection of the current transducer.

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Soft Switching Bidirectional Converter for Battery charging discharging