S W R F -E B L E T by Anith Selvakumar · 2019-03-12 · Anith Selvakumar Master of Applied Science Graduate Department of Electrical and Computer Engineering University of Toronto

SUB-MW RECEIVER FRONT-END FOR BLUETOOTH LOW ENERGY IN

130NM TECHNOLOGY

by

Anith Selvakumar

A thesis submitted in conformity with the requirements

for the degree of Master of Applied Science

Graduate Department of Electrical and Computer Engineering

University of Toronto

© Copyright 2015 by Anith Selvakumar

II

Abstract

Sub-mW Receiver Front-End for Bluetooth Low Energy in 130nm Technology

Anith Selvakumar

Master of Applied Science

Graduate Department of Electrical and Computer Engineering

University of Toronto

2015

Bluetooth Low Energy (BLE) is the new operative mode introduced in the fourth release of

the Bluetooth wireless technology standard. It is primarily designed for portable devices

powered by coin cell batteries, including sensors and wearable biomedical devices. Under

this standard, minimal power consumption is essential for the long-term functionality that

these devices require.

In this thesis, a sub-mW BLE receiver front-end operating under a 0.8 V supply is presented.

Low power consumption is obtained sharing the same bias current among all the RF front-

end building blocks and the first stage of the base-band. With an NF of 15.8 dB, an IIP3 of -

17dBm at the maximum gain and an image rejection above 30 dB the receiver is fully

compliant with BLE specifications with a minimum sensitivity of -84.2 dBm.

III

Acknowledgments

I would like to first express my gratitude to my supervisor, Professor Antonio Liscidini, for

all his guidance and support. He has truly made my time under his supervision an insightful

and enjoyable learning experience.

I would also like to thank the committee members, Professor Liscidini, Professor Chan

Carusone, Professor Ng, and Professor Broucke for their time, feedback, and constructive

criticism on the development of this thesis.

I would like to thank Meysam Zargham who helped lay out some high-frequency blocks and

also provided CAD support during tape-out time.

To the folks at BA5000, you have been a tremendous support (both moral and technical) for

this thesis. Special thanks to Michal Fulmyk who has always been of assistance these past

two years.

To my fellow comrades, Janahan Ramanan and Rosanah Murugesu, our regular

misadventures (and tea breaks) over the most ridiculous of topics will be remembered for

years to come. These past eight years have flown by and I have no doubt that we will

continue to laugh at each other’s misfortunes together.

Finally, I would like to thank my family, whom have continually encouraged me to pursue

the path of academia.

IV

Contents

Abstract ........................................................................................................................................... II

Acknowledgments......................................................................................................................... III

Contents ........................................................................................................................................ IV

List of Figures and Tables............................................................................................................. VI

List of Acronyms ....................................................................................................................... VIII

Chapter 1: Introduction .................................................................................................................. 1

1.1 Motivation ........................................................................................................................ 1

1.2 Structure: .......................................................................................................................... 3

Chapter 2 ......................................................................................................................................... 4

Bluetooth Low Energy Standards ................................................................................................... 4

2.1 System Overview ............................................................................................................. 4

2.2 Radio Interface ................................................................................................................. 5

2.3 Modulation ....................................................................................................................... 7

2.4 Radio Specifications ....................................................................................................... 10

2.4.1 Receiver Sensitivity ................................................................................................ 11

2.4.2 Linearity and Filtering ............................................................................................ 13

2.4.3 Summary of Requirements ..................................................................................... 16

2.5 Typical Architectures ..................................................................................................... 17

2.5.1 Superheterodyne Receiver: ..................................................................................... 18

2.5.2 Low-IF .................................................................................................................... 22

2.5.3 Direct-Conversion ................................................................................................... 28

2.5.4 Table of Comparison ............................................................................................... 30

2.6 Challenges ...................................................................................................................... 31

2.6.1 Portability:............................................................................................................... 31

Chapter 3 ....................................................................................................................................... 33

V

Front End Receiver ....................................................................................................................... 33

3.1 Starting Point: LMV Cell ............................................................................................... 33

3.1.1 LMV Cell Overview ............................................................................................... 34

3.1.2 Challenges: .............................................................................................................. 36

3.1.2.1 N-Type VCO: ...................................................................................................... 36

3.1.2.2 Quadrature Generation: ....................................................................................... 38

3.1.2.3 Loss Mechanisms: ............................................................................................... 38

3.2 Design Solutions ............................................................................................................ 41

3.2.1 P-N VCO:................................................................................................................ 41

3.2.2 Low Voltage Drop LNA with Quadrature Generation (QLNA): ........................... 45

3.2.3 TIA : ........................................................................................................................ 49

3.2.4 Channel Selection and Image Reject Filter: ............................................................ 51

3.3 Summary ........................................................................................................................ 61

Chapter 4 ....................................................................................................................................... 63

Measurements ............................................................................................................................... 63

4.1 Silicon Die and Test Board ............................................................................................ 63

4.1.1. Measurement Equipment ........................................................................................ 65

4.2 Chip Measurements ........................................................................................................ 66

4.2.1. Input Matching: S11 ............................................................................................... 66

4.2.2. AC Transfer Function ............................................................................................. 68

4.2.3. Noise Figure ............................................................................................................ 69

4.2.4. Linearity .................................................................................................................. 71

4.3 Performance Summary and Comparison ........................................................................ 73

Chapter 5 ....................................................................................................................................... 75

Conclusion .................................................................................................................................... 75

5.1 Summary ........................................................................................................................ 75

5.2 Future Work ................................................................................................................... 76

Bibliography ................................................................................................................................. 78

VI

List of Figures and Tables

Figure 1: Example Bluetooth Network: Showing an example of a scatternet with 3 separate

Piconets P1, P2, and P3. Each Picnonet has its own master and slave. P3 shows that although

D is a slave to P1, it is also the master for P3 [2]. .......................................................................... 5

Figure 2: Spectrum channel distribution for Bluetooth Low Energy. ............................................ 6

Figure 3: Available BLE channels under Wi-Fi interference. ........................................................ 7

Figure 4: Time and frequency representations of the rect and sinc function. ................................. 8

Figure 5: Spectrum of a GFSK spectrum for Bluetooth [4]. ........................................................... 9

Figure 6: Zero intersymbolic interference between time samples [6]. ......................................... 10

Figure 7: Intermodulation products formed by blockers f1 and f2. .............................................. 14

Figure 8: Bluetooth image rejection requirements. ....................................................................... 16

Figure 9: System level diagram of a superheterodyne receiver architecture ................................ 19

Figure 10: System level and frequency representation of signal downconversion. ...................... 20

Figure 11: Image corruption into the signal band as result of downconversion. .......................... 20

Figure 12: System level diagram of a Low-IF receiver architecture. ........................................... 22

Figure 13: Image rejection using quadrature signals. ................................................................... 23

Figure 14: Phase and amplitude imbalance impact on maximum image rejection achievable

[14]. ............................................................................................................................................... 25

Figure 15: Alternative IQ generation scheme for improved robustness with respect to PVT

variations. ...................................................................................................................................... 26

Figure 16: Polyphase filter topology and signal representation [17]. ........................................... 27

Figure 17: LMV Cell topology. .................................................................................................... 34

Figure 18: Differences between the common mode and the differential LC tank. ....................... 37

Figure 19: Decomposition of an RF input signal into a differential and common mode

component. .................................................................................................................................... 39

Figure 20: Traditional N type VCO and its equivalent small signal model. ................................. 41

Figure 21: Complementary Cross-Coupled VCO and its equivalent small signal model. ........... 42

Figure 22: Single-voltage overdrive QLNA. ................................................................................ 45

Figure 23: Phase and magnitude relationship for I and Q paths. .................................................. 46

Figure 24: Complete LMV topology with I-Q channels ............................................................... 48

Figure 25: TIA Topology to reduce conversion gain losses at LMV output. ............................... 49

Figure 26: Active inductor to resonate with LMV cell parasitic output capacitor, Cpar. ............. 50

Figure 27: Small signal analysis for the derivation of the active inductor impedance ................. 50

Figure 28: Gm-C Complex-Filter Topology ................................................................................. 53

Figure 29: Proposed current reuse gm-C topology by Lin et al. .................................................. 54

Figure 30: Positive feedback loop which must be maintained stable by keeping Q<1. ............... 55

Figure 31: Filter topology implemented. ...................................................................................... 56

VII

Figure 32: Contour plot showing the set of possible α and Q’ where the positive feedback

loop is stable; ................................................................................................................................ 59

Figure 33: A flatband response was created by cascading two complex poles with different

shifts to realize the complex bandpass with Q=0.7 about 2MHz. ................................................ 61

Figure 34: Top-level Diagram of BLE Front-End. ....................................................................... 62

Figure 35: Die micrograph (left) with its wirebonded implementation on PCB (right). Filter

layout is also shown (below)......................................................................................................... 64

Figure 36: PCB Fabricated for chip measurements. ..................................................................... 65

Figure 37: Matching circuit and component values. ..................................................................... 67

Figure 38: S11 Measurements. ..................................................................................................... 67

Figure 39: AC Transfer Function. ................................................................................................. 68

Figure 40: Gain and Image Rejection for each BLE Channel. ..................................................... 69

Figure 41: Noise Figure Measurement for each BLE Channel. .................................................... 70

Figure 42: IIP3 Measurement for each BLE Channel. ................................................................. 72

Figure 43: Scope of this thesis. ..................................................................................................... 76

Table 1 Design requirements for the BLE Receiver. .................................................................... 16

Table 2: Comparison between Superheterodyne, Low-IF, and Direct-Conversion topologies .... 30

Table 3: Test Equipment ............................................................................................................... 65

Table 4: Performance Summary and Comparison. ....................................................................... 73

VIII

List of Acronyms

ADC Analog to Digital Converter

AFH Adaptive Frequency Hopping

BLE Bluetooth Low Energy

BER Bit Error Rate

CMOS Complementary Metal-Oxide Semiconductor

DCR Direct Conversion Receiver

GFSK Gaussian Frequency Shift Keying

IF Intermediate Frequency

IIP3 Third-order Intercept Point

IM Intermodulation Product

ISM Industrial, Scientific, Medical Radio Band

LNA Low Noise Amplifier

LO Local Oscillator

NF Noise Figure

P2MP Point to Multi-Point

P2P Point to Point

PCB Printed Circuit Board

SNR Signal-to-Noise Ratio

TIA Transimpedance Amplifier

VCO Voltage-Controlled Oscillator

1

Chapter 1:

Introduction

1.1 Motivation

Over the past few decades, continual progress in microelectronic technologies has given

rise to the wireless world we live in today. Factors including the scaling of VLSI CMOS

processes as well as developments in power-saving architectures have made it possible to

turn wireless communication devices to wireless autonomous devices, which have now

become almost a necessity to our daily lives. The invention of the cell phone for voice

transmission is only one example of this phenomenon that has revolutionized wireless

communications. It has now translated even smaller devices which store and transmit smaller

packet data on a regular basis, such as biosensors on wearable electronics, where longevity

and portability are key requirements to its use. Data transmission has now begun to taken the

world by storm to make our lives easier through a vast array of applications.

For wireless autonomous devices to be feasible, it is important that power consumption

be kept minimal to avoid large batteries that reduce the portability of the device, or otherwise

limit autonomy by requiring frequent recharging. In addition, the recent development of

alternative power harvesting technologies, such as photovoltaic cells, has not matured

2

sufficiently to provide the voltage required by many CMOS technology processes. The

typical solar cell can only provide between 0.5V to 0.8V of voltage, while many technology

nodes require a supply of 1.2V. Although additional solar cells can be added to achieve

higher voltage supplies, this solution takes away from the compactness (and hence,

portability) of the device. This new need for small and high efficiency communication

devices beyond traditional uses has opened up an area within microelectronic circuits which

is now at its early stages of being explored: low-power electronics for wireless autonomous

communication applications.

Bluetooth Low Energy (BLE) is the new operative mode introduced in the fourth release

of the Bluetooth wireless technology standard. BLE operates in the same 2.4GHz ISM radio

band as Classic Bluetooth, however is tailored towards ultra-low power devices powered by

coin-cell batteries, such as wireless sensor networks for indoor localization, wireless payment

tags, and wearable devices. In such applications, performance can be sacrificed in favour of

an extended battery life obtained by minimizing the overall power consumption of the radio.

Its relaxed specifications make BLE-based transceivers ideal for reducing the cost and

autonomy of the up and coming market for short-range communication devices.

Modern transmitter/receiver architectures require several blocks, that when traditionally

integrated, consume large amounts of current to collectively deteriorate power consumption.

Consequently, the strategy of current recycling, whereby blocks are stacked to share the same

bias current, can present considerable power savings. The LMV cell developed by Liscidini

et al. is an excellent example of this power reduction technique, where the Low-Noise

Amplifier, Mixer, and VCO cells are stacked to recycle its bias current [1]. The extension the

3

LMV cell for improved performance and adoption to the BLE wireless technology standard

is the focus of this thesis.

1.2 Structure

This thesis describes the design and implementation of a sub-mW receiver for Bluetooth Low

Energy applications in IBM 130nm technology, and is organized into 5 chapters:

Chapter 2 gives an overview of the Bluetooth Low Energy wireless technology

standard and provides a comparison of traditional receiver architectures. Specifically, the

structure of the BLE operating scheme will be discussed as well as its technical

specifications. This chapter also provides a summary of traditional receiver architectures

including an analysis of their merits and demerits. Finally, the challenges associated of

designing a receiver that complies with the specifications and overall vision of the BLE

standard will be presented.

Chapter 3 describes the implementation of the front-end receiver. The core block

which forms the starting point of this research, namely the LMV cell, will be presented, as

well as a summary of the benefits and challenges associated with it. Next, the proposed

modifications to improve performance and compliancy to the BLE standard will be described

and any circuit implementations will be presented.

Chapter 4 provides a full characterization of the front-end receiver, based on

simulation and measurement results obtained in the lab.

Finally, in Chapter 5 a summary of the accomplishments of this thesis will be

described. Future work will also be discussed.

Chapter 2

Bluetooth Low Energy Standards

2.1 System Overview

Bluetooth technology was designed to allow for wireless communication between a wide

array of devices through short-range, ad hoc networks known as piconets. Piconets are

created dynamically when two or more Bluetooth-enabled devices are within radio proximity

range to allow for immediate connectivity and transfer of information. Therefore, any device

with a Bluetooth enabled radio can form a piconet to instantly transfer information wirelessly

to a connected device. [2]

Bluetooth devices form piconets for both point-to-point (P2P) and point-to-multipoint

peer (P2MP) connections. Upon initiating a P2P or P2MP connection, where multiple

devices share the same channel, a Master and Slave relationship is established to avoid data

transfer collisions within the piconet [2]. In Bluetooth, a piconet can support a maximum of 1

Master and 7 Slave devices, however, each device itself can also be a part of several other

piconets simultaneously in what is known as a scatternet. Scatternets are piconets that share

the same coverage area. To minimize unwanted interference between scatternets, Bluetooth

CHAPTER 2: BLUETOOTH LOW ENERGY STANDARDS

5

employs an adaptive frequency-hopping spread (AFH) spectrum scheme to reduce the

chances of two transmitters occupying the same transmit channel [2]. This scheme will be

discussed further in the next section.

A

DC

E

B

F G

H

P1

P2

P3

Figure 1: Example Bluetooth Network: Showing an example of a scatternet with 3 separate

Piconets P1, P2, and P3. Each Picnonet has its own master and slave. P3 shows that although

D is a slave to P1, it is also the master for P3 [2].

2.2 Radio Interface

Bluetooth Low Energy, a variant of classic Bluetooth, operates in the 2.4 GHz ISM

spectrum internationally reserved for the use of industrial, scientific and medical purposes.

Its operating band of 2400 - 2483.5 MHz is divided into 40 channels (37 data communication

channels and 3 advertising channels for data discovery) spaced 2 MHz apart:

(2.1)


6

This is in contrast to Classic Bluetooth, which can support 78 channels within the same

operating band. Larger channel spacing is one of the trade-offs that Bluetooth Low Energy

has adopted in efforts to improve power savings. Specifically, increasing channel spacing

leads to relaxed demands on RF filtering and consequently, reduced device power

consumption.

Figure 2: Spectrum channel distribution for Bluetooth Low Energy.

Bluetooth’s Adaptive Frequency Hopping (AFH) mechanism improves its immunity

to interference. Transmitted data is divided into packets and transmitted on one of 37

transmit channels. Since the physical channels used to transmit information are constantly

changing, the probability of two devices in the piconet transmitting on the same channel is

quite low. In BLE frequency hopping occurs at a rate of 1600 hops/sec. The pseudo-random

frequency hopping sequence is determined by the master and must be followed by the slaves

of the piconet. In AFH, the rate of collision is further reduced by identifying fixed sources of

interference in the environment and avoiding their use [3].


7

Figure 3: Available BLE channels under Wi-Fi interference.

Figure 3 visually describes the Bluetooth Adaptive Frequency Hopping strategy. In

this case, of the 37 available channels that can be used for BLE data transmission, 9 channels

can be utilized because of interference from Wi-Fi channels. The BLE AFH mechanism will

identify this and avoid any possible collisions.

2.3 Modulation

Bluetooth Low Energy uses a Gaussian Frequency Shift Keying (GFSK) modulation

scheme with a modulation index between 0.45 and 0.55 [4]. GFSK is a variation of

Continuous Phase Frequency Shift Keying (CPFSK), where digital information is impressed

into discrete frequency changes of a sinusoidal carrier signal, typically by adjusting the

voltage applied to a voltage controlled oscillator (VCO). This implementation reduces the

spurious emissions generated from phase discontinuities from fast changing bit sequences

when two independent oscillators are employed. The choice of pulse applied to the VCO is

also crucial to reduce unwanted emissions and can be better understood by examining the

frequency-domain of the traditional rectangular pulse. [5]


8

Time Domain Frequency Domain

A(t)

t

f

A(f)

sinc(t) rect(f)

t

A(t)

f

A(f)

rect(t) sinc(f)

Figure 4: Time and frequency representations of the rect and sinc function.

As shown in Figure 4 the frequency content (spectrum) of the rect signal (time

domain) is infinite, potentially causing significant interference issues to other channels on the

radio spectrum. Additional power is also burned when transmitting a signal with a wider

frequency range. It is therefore favourable to transmit band-limited signals to reduce

interference and increase power efficiency of the transmitter/receiver. Contrarily, the sinc

function in (2.2) has a bandlimited frequency spectrum (rect) that would not pose any

interference to nearby channels and would not require additional filtering.


9

(2.2)

Unfortunately, since the sinc function it is a non-causal pulse it not realizable in

hardware [6]. Instead, BLE’s GFSK modulation scheme employs a compromise between the

band-limited sinc and easy-to-implement rect through performing Gaussian filtering to the

traditional rect function.

Figure 5: Spectrum of a GFSK spectrum for Bluetooth [4].

The tradeoff that arises when sending a band-limited (filtered) signal is intersymbolic

interference, which refers to the corruption by one symbol with subsequent signals in the

time domain. This issue can be clearly seen in Figure 6, where all symbols occur for a finite

amount of time and do not distort future symbols. In order for there to be no interference with

future data when transmitting a symbol, the following criterion (Nyquist Criterion for Zero


10

ISI) must hold:

(2.3)

Figure 6: Zero intersymbolic interference between time samples [6].

By applying a sharp band-limiting filter to the traditional rect pulse, the rect becomes

“wider” and no longer satisfies the Nyquist ISI criterion, leading to distortion in the time

domain. ISI reduces the overall noise margin, but can be tolerated in low amounts depending

on the acceptable error rate.

2.4 Radio Specifications

In order for the receiver to be BLE compliant, it must satisfy a set of criterion as dictated by

the Bluetooth Low Energy Standard. These constraints form the major specifications for the

design of the BLE Receiver front end:


11

2.4.1 Receiver Sensitivity

Receiver sensitivity is defined as the minimum signal power that can be detected based

on a specified accuracy. This accuracy primarily refers to the bit error rate (BER), the

percentage of incorrectly measured bits over the transmission system. Incorrectly measured

bits can be the result of a weakly transmitted signal in the presence of large noise

interferences in the air link or at the circuit level. A high BER signifies a low Signal to Noise

ratio (SNR), since the signal power and noise power levels are comparable in magnitude.

This level of interference can make it difficult for the receiver to decode the bit and

consequently increase the probability of making an incorrect decision. Contrarily, a high

signal to noise ratio results in a low BER, as the signal power level is sufficiently higher than

any interference that can corrupt the decision [6]. According BER to SNR curves for the

GFSK Modulation scheme with a maximum BER of 10E-3 (as per the BLE Standard), a

minimum SNR of approximately 12-14 dB is required at the output of the receiver,

depending on the demodulation algorithm chosen [7] . This information is crucial to the

understanding of the SNR degradation that can be tolerated from the input to output of the

receiver, also known as the noise figure:

(2.4)

where PRS is the noise power generated from the source resistance (typically 50 Ohm), and

Psig is with respect to a unit bandwidth. In order to obtain the total power of the signal, it

must be integrated over its total bandwidth:


12

(2.5)

which can be simplified further based on PRS = -174 dBm, assuming a matched source

resistance:

(2.6)

where K is the Boltzmann constant and T is temperature in Kelvin. By taking the logarithm

to convert units into dBm

(2.7)

Finally, the noise figure can be written as a function of the sensitivity of the receiver, the

power of the source resistance, the SNR at the output, the bandwidth of the signal, and an

insertion loss factor to account for the losses present between antenna and the input to the

analog processing:

(2.8)

According to the Bluetooth Low Energy standard, the minimum sensitivity of a BLE receiver

must be > -70 dB [2]. The following expression in (2.9) also takes into account the minimum

SNR at the output that must be achieved. An SNR value of 12 dB was used based on a

typical demodulator and the requirement of BER > 0.1% [7]. An insertion loss of 2 dB was

also accounted for. The bandwidth of the BLE signal is 1 MHz. Based on these specifications

the maximum noise figure can be calculated as,

(2.9)


13

yielding a maximum noise figure of 30 dB. In other words, the receiver can tolerate a

maximum of 30 dB signal degradation (or noise interference) before losing compliancy to

the Bluetooth Low Energy standard. However for improved accuracy in demodulation, a

lower BER (and thus a higher SNR) would be desired for a high-quality BLE receiver. To

achieve this, the noise figure of the receiver must be minimized to suppress any noise

generation that contribute to the SNR degradation in the receiver chain.

2.4.2 Linearity and Filtering

A receiver’s susceptibility to interference can be described by linearity performance. All

semiconductor devices experience some level of non-linearity by generating unwanted

harmonics at the output from a pure tonal input [8]. When two or more signals tones are input

to a non-linear device, intermodulation (IM) products are generated that can fall in-band to

deteriorate the SNR. The source of these tones can be nearby signals that are picked up by

the radio antenna. The frequencies of IM products are located at:

, (2.10)

where M, N = 0, 1, 2, 3;

and the order of distortion is the sum of M+N.


14

Blockers

Signal

IM3 IM3 IM2IM2 IM2IM2

Figure 7: Intermodulation products formed by blockers f1 and f2.

For example, third order intermodulation products occur at f1 + 2f2, f1 – 2f2, 2f1 +

f2, 2f1 – f2. The third order IM products, in particular, are of special interest because they

often fall very close to or within the desired channel. For this reason the linearity of a

receiver is often characterized by the Third Order Intercept Point (IP3). The IP3 relates the

IM3 (third order IM product) to the fundamental signal. Specifically, the input power level at

which the IM3 intersects the fundamental tone is known as the IIP3, and its corresponding

output, OIP3. This is often tested by the two-tone test, where two blockers are fed into a

system to create the IM in the in-band of the signal [9]. Higher IIP3 signifies weak

intermodulation products with respect to the fundamental tone, and hence a more linear

system.

The Bluetooth Low Energy standard requires that a BER < 0.1% is maintained in the two

tone test, given the following requirements:

The wanted signal is 6 dB over the reference sensitivity level at frequency fo


15

Two interferers are of power level -50 dBm at frequency f1 and f2

Frequencies fo, f1, f2, are chosen such that fo = 2*f1-f2 and |f2-f1| = n*1MHz, where n=3, 4 or

5. The system must satisfy at least one of the three conditions. [2]

The IIP3 that must be met in order to meet the Bluetooth Low Energy standard can be

calculated as:

[10] (2.11)

(2.12)

Based on a required SNR of 12 dB the minimum IIP3 of the Bluetooth Low Energy Receiver

must be -37 dBm.

With respect to image rejection, the Bluetooth Low Energy standard requires that a

BER<0.1% be maintained for the image interference case shown below.


16

-67 dBm

-58 dBm

ωIM-ωIM

SignalInterferer

Figure 8: Bluetooth image rejection requirements.

In order to maintain the 12 dB required at the output of the receiver, the image reject

filter must be able to suppress the -58 dBm signal to a value of -79 dBm, such that the signal

is 12dB larger [2]. This would require an image rejection of 21 dB.

2.4.3 Summary of Requirements

The following table summarized the key performance specifications required by the BLE

receiver:

Table 1 Design requirements for the BLE Receiver.

Specification Requirement

Noise Figure < 30 dB

IIP3 > -37 dB

Image Rejection > 21 dB

SNR > 12 dB


17

Compliance to the performance specifications of the Bluetooth Low Energy standard is

not particularly a challenge. This is because the particular standard is catered towards

applications where performance can be sacrificed in favour of power savings. This can be

easily seen when comparing BLE requirements with the Bluetooth Classic standard, which

specifies a minimum IIP3 greater than -16.5 dBm and image rejection greater than 29 dB

[11]. Instead, the main emphasis for a good BLE receiver is in its efficiency. While high

performance solutions can be developed, it is usually at the cost of power consumption. A

high noise figure can be obtained by increasing the gain of the receiver blocks. Increased

image rejection can be obtained by sharp cut-off filters through higher orders. Increased

linearity can be obtained by raising the voltage supply or increasing the bias current [12]. In

each of these cases, the trade-off is increased power consumption. Considerable research has

been conducted in maintaining high performance designs without considerably sacrificing the

power consumption. In this thesis, the aim is to obtain a sub-mW power consumption while

still meeting and exceeding all requirements of the Bluetooth Low Energy standard. Typical

receiver architectures will be first discussed in order to determine the best topology given the

requirements for BLE.

2.5 Typical Architectures

The analog signal picked up by the antenna at the receiver is an RF signal (2.4GHz) with

considerable attenuation and noise. Before it can be successfully decoded into a bitstream of

1’s and 0’s it must undergo analog and baseband processing to down-convert the signal to a

lower frequency and apply filtering for channel selection and image rejection. Converting


18

the signal to the baseband frequency for processing eases the requirements for the filter

(lower Q) and reduces the power consumption as the circuits can operate at the lower

baseband frequency (MHz range) instead of RF (GHz range) [10]. Unfortunately, the process

of downconversion can also lead to the corruption of the desired channel with the image.

This will be described in greater detail in Section 2.5.2. The image is dealt with in the

receiver chain and can sometimes be completely eliminated. In other cases, image

suppression is often sufficient enough to meet the needs of the wireless standard.

Considerable research has gone into the development of Transmitter and Receiver

architectures for different standards, since they play a pertinent role in the overall size,

performance, cost, and power-consumption of the receiver. As different wireless standards

have their own requirements, certain architectures may not be ideal for some applications. In

this section, three typical receiver architectures will be described and analyzed for their

compatibility for the Bluetooth Low Energy wireless standard. They are: Heterodyne, Direct-

Conversion, Low-IF.

2.5.1 Superheterodyne Receiver

The heterodyne receiver architecture has been one of the most widely used topologies

over the past century since its inception in the early 1900’s. Heterodyne receivers use the

technique of down converting the desired channel to a lower frequency known as the

intermediate frequency (IF) through one or more stages. As mentioned above, down

conversion serves many purposes, including power savings and channel selection-filtering

with a reasonable quality factor [10]. This is done by using a mixer, which requires a local

oscillator input to perform the frequency translation.


19

LNA

~ ~

Preselection Filtering

LO1 LO2

Low Pass FilterDownconversionRF Input

Figure 9: System level diagram of a superheterodyne receiver architecture

The frequency domain representation shown in Figure 10 illustrates this

downconversion, where the input signal is convolved with the tones of the local oscillator.

The result is the shift of the channel to ± (ωin ± ωLO). The higher frequency components are

usually not of interest and are often removed through low pass filtering. Mixing also

introduces an unwanted signal known as the image into the signal band. Specifically, this

operation translates both signals fLO + fIF and fLO – fIF to the same fIF band. This can

significantly corrupt the desired signal if not handled carefully, as the two signals which lie

in the same band are indistinguishable. This effect is shown in Figure 11.

ωRF

~ Acos(ωLOt)

ωIF

Low Pass Filter

ωIF = ωRF - ωLO


20

ωRF-ωRF

ωLO-ωLO

*

-ωRF-ωLO -ωRF+ωLO ωRF-ωLO ωRF+ωLO

=

Figure 10: System level and frequency representation of signal downconversion.

ωRF-ωRF ωLOωIMωLO ωIM

ωIFωIF ωIFωIF

ωIFωIF

Before Downconversion:

Desired Signal

ImageImage

Desired Signal

After Downconversion:

Figure 11: Image corruption into the signal band as result of downconversion.

The superheterodyne topology down-converts the RF signal in multiple stages,

applying filters at each stage for image suppression and blocker interference. The weak

signal picked up by the receiver is first filtered by a RF pre-selection filter and amplified with

a Low-Nose Amplifier (LNA). As one of the first blocks in the receiver, the low noise


21

amplifier plays a pertinent role in the overall performance of the receiver chain. Specifically,

the first block in a cascade typically dominates the overall noise and linearity performance of

the receiver. This is based on Frris formula, which relates the total noise factor of a cascade

of stages based on each stages gain and noise factor.

[10] (2.13)

,where Fn and Gn are the noise factor and gain for the n-th stage

Based on (2.13) it is important for the LNA to provide high gain and have a low noise

figure. Despite the RF Filter, an image signal of some magnitude will still exist. This image

and the desired signal will be amplified after the LNA stage. Before the RF signal is

translated to the first IF, IF1, it goes through another preselection filter to further suppress the

image before mixing. This is sometimes repeated through another stage of preselection and

mixing to IF2 which is then low-pass filtered and processed by an Analog-to-Digital

Converter (ADC). Breaking the down-conversion into partial conversion stages with

filtering improves the image rejection and also relaxes the requirements for each filter [13].

Channel selection can be done through tuning the frequency local oscillator such that it

falls within the bandwidth of the fixed channel select filter at the IF. In other

implementations, the local oscillator is fixed and the filter is tuned to select only the channel

of interest. The latter is less frequent as it is easier to tune the LO than design a tuneable filter

with constant passband gain.


22

2.5.2 Low-IF

The Low-IF receiver architecture employs a similar working principle as the

superheterodyne receiver, however being unique in the choice of the IF frequency. The Low-

IF architecture uses a very low intermediate frequency, on the order of a few megahertz. By

doing so, a major challenge arises in terms of the image. Since the LO is placed very close to

the signal of interest, the image will also be closely placed to the signal, making preselection

and channel filtering particularly difficult. This problem gives rise to a trigonometric solution

to the image problem.

∑LNA ~0o

90o

LO1RF Input

Polyphase Filter &Channel Selection

Quadrature Downconversion

Figure 12: System level diagram of a Low-IF receiver architecture.

The Low-IF receiver topology is able to suppress the image by creating two phase shifted

copies of the image-corrupted signal which can then be used to reconstruct the original

signal. This is known as quadrature down conversion or complex mixing. The two phase

shifted-signals are known as quadrature signals, in-phase (I) and quadrature (Q). Consider the

following signal with image:

(2.14)


23

After complex mixing, the two outputs on the I and Q paths are:

(2.15)

(2.16)

(2.17)

(2.18)

In 2.18 for the quadrature channel, the image and desired signal are out of 180 degrees out of

phase. This makes it possible to completely suppress the image upon recombining with the

in-phase channel after an additional 90 degree phase shift:

(2.19)

This can be understood clearly in the following graphical representation:

1/j1

1/j1

2

In-Phase Quadrature

IF Output

Ideal Complex

Filter

-

Figure 13: Image rejection using quadrature signals.


24

Special care must be taken to ensure that the gain and phase error is minimized (I and Q

paths are perfectly matched). Any phase mismatch between the I and Q paths can result in

poor image rejection.

To show the effect of mismatch on gain and phase between the I and Q phases, consider a

gain and phase mismatch in signals xa(t) and xb’(t). After summation the result is:

(2.20)

(2.21)

An Image Reflection Ratio, IRR, can be defined as the ratio of signal power to image power

at the output and is a measure of image rejection:

(2.22)

The plot in Figure 14 shows how a small mismatch and limit the total image rejection that

can be achieved.


25

Figure 14: Phase and amplitude imbalance impact on maximum image rejection achievable

[14].

The image remains a major drawback in the Low-IF architecture as it requires

excellent matching and relies on ideal phase shifts for image cancellation [13].

Unfortunately, on a circuit point of view, it is often difficult to realize perfect 90 degree shifts

across all PVT corners. To help matching, the 90 degree phase shift in one path is replaced

by a +45 degree phase shift in one path and -45 degree phase shift in the other. This way, if

there are any PVT variations, it affects both paths equally. It can be realized using a simple

RC-CR network as shown in Figure 15.


26

∑~0o

90o

LO1

IFOutput

RF Input

R1

C1

C1

R1

Figure 15: Alternative IQ generation scheme for improved robustness with respect to PVT

variations.

However, this implementation also adds room for mismatch, since the phase and gain is not

constant and will vary across the band of the channel. As result image rejection may degrade

at the outer bounds of the channel. This problem is further exacerbated as the IF is in the

same range as the channel bandwidth, making the channel especially sensitive to the non-

linear characteristic of the RC-CR network. An alternative solution is to employ polyphase

filters for high image rejection [15].

Polyphase filters can essentially be described as a symmetric RC network with inputs

and outputs at equidistant phases – 0o

, 90o, 180

o, 270

o. Using a differential scheme, the

polyphase filter performs two different phase rotations at the RC cut-off frequency, one

clockwise and the other counter clockwise, which can pass the wanted signal and null the

image [16]. Consider the following circuit scheme and its equivalent output transfer function,

obtained via superposition:


27

R1 C1

V1 V2Vout

+

_

(2.23)

MixerOutput

Vsig + Vim

-Vsig - Vim

jVsig - jVim

-jVsig + jVim

Vout1 Vout2 Vout3 Vout4

+Vsig-Vsig

+jVsig

-jVsig

+Vim-Vim

+jVim

-jVimVout1

Vout2

Vout4

Vout3

Vout1=Vout2=Vout3=Vou4 = 0

Figure 16: Polyphase filter topology and signal representation [17].

Note that when V2 is –jV1, it can be shown through Eq. 2.23 that Vout = V1(1-j). If V2 =

+jV1, the result is Vout = 0. By exploiting the asymmetrical nature of the quadrature signals,

Vsig+Vim, -Vsig-Vim, jVsig – jVim, -jVsig + jVim, the desired signal will be simply shifted

45 degrees, while the image can be completely cancelled. This can also be referred to as a

complex filter, as it produces a band pass frequency response that is asymmetrical about the

LO frequency, shifted by the frequency 1/RC.


28

2.5.3 Direct-Conversion

Direct conversion receivers, also known as homodyne or zero-IF receivers can be

considered a variant of the Low-IF, with a different choice of IF. In direct conversion

receivers, the RF signal is downconverted directly to baseband. This presents a unique case

where the signal itself becomes the image and is known as self-corruption. To rectify this,

quadrature downconversion can be performed to create a phase shift that can then be used to

reconstruct the original signal, as in Low-IF receivers [10]. Since the signal is translated

directly down to baseband frequency, channel selection can be realized through on-chip low

pass filters with sharp cut off frequencies. This reason may make DCRs seem an attractive

option for receiver implementations. However, there are a number of issues that arise when

performing downconversion to baseband which is not as severe in other topologies, including

I/Q mismatches, LO leakages, DC offset, and 1/f noise.

As in the case of the Low-IF topology, I/Q gain and phase mismatches can lead to

distortion from poor image rejection. In DCRs, since the image is the signal itself, there are

no preselection or channel select filters making it especially easy for the signal to be

corrupted if the I/Q paths are not perfectly matched. The “image” is at the same power level

as the desired signal. This is not the case for superheterodyne topologies, where the image is

usually suppressed prior to mixing [18].

Another problem that affects direct conversion receivers more than other topologies is

LO leakage, which leads to DC offsets. Parasitic pathways (capacitances and resistances)

from the LO input to the mixer can allow a signal to make its way to the front of the receiver

and be emitted by the antenna [10]. Since the LO frequency is ωRF, the LO leakage which

finds its way to the front of the receiver, can also go through the entire receiver chain. Now


29

instead of just the input signal, the additional LO leakage signal becomes downconverted by

the mixer and is seen as a DC offset. As typical receiver chain produces a gain on the range

of 30-100 dB, the DC component also undergoes amplification to potentially saturate the

receiver. The DC offset issue does not only affect the direct conversion receiver, as LO

leakage can occur in all receiver architectures which employ a mixer. However, in DCR the

DC offset becomes a severe matter solely because of the choice of zero-IF. That is, the DC

offset occurs in-band to the signal of interest and thus cannot be easily filtered out. In other

topologies, such as Low-IF or Superheterodyne, AC coupling, by means of a high pass filter,

can be used for offset cancellation without interfering with the signal. This is not feasible for

the direct conversion receiver because high pass filters do not have a sharp response to block

DC but allow signals near zero frequency to pass without some attenuation [10].

Flicker noise, also known as 1/f noise is a type of noise with a frequency-dependant

power spectrum. Unlike white noise, whose power spectral density is constant over all

frequency, flicker noise produces large noise components at low frequencies (falling off at a

rate of 1/f), until it reaches the corner frequency, after which thermal noise dominates the

power spectrum. This phenomena makes the choice of IF crucial to minimize the receivers

noise figure. Unfortunately, for the direct-conversion receiver flicker noise directly affects

the signal band, since it is downconverted to zero-frequency. Similar to the DC offset

problem, other topologies are not significantly affected by this, as it is avoided through the

choice of IF (avoiding the flicker noise band) or strong filtering prior to baseband

downconversion.

Another major drawback associated with DC-receivers is the more demanding

requirement of the ADC used for image rejection. A direct conversion receiver produces I


30

and Q outputs at baseband, both the signal and the image. It relies on the ADC to perform I/Q

processing, namely image rejection and demodulation. Unfortunately, this produces

demanding requirements for the ADC, as it would require a very high dynamic range. This is

because the signals at the input to the ADC can cover a large dynamic range (depending on

the image interference power that is unknown). It would be required to process small

interferences in order to achieve perfect image rejection.

2.5.4 Table of Comparison

Each of the above topologies has their own merits and demerits, making them ideal for

some applications and unideal for others. For the case of Bluetooth Low Energy, it was

important that the design be, first and foremost, power efficient. Since the requirements for

noise are more relaxed in BLE, noise figure of the receiver can be compromised slightly in

favour of a low-power implementation. However, noise figure is an important aspect to the

quality of a receiver, and should not be overlooked.

The table below summarizes the advantages and disadvantages of the three topologies listed.

Table 2: Comparison between Superheterodyne, Low-IF, and Direct-Conversion topologies

Superheterodyne Low-IF Direct Conversion

Advantages High Selectivity

High Sensitivity

Flicker Noise

Power consumption

High Selectivity

Power Consumption

Disadvantages Power consumption

Area

Image Rejection

I-Q Generation

DC-Offset

Flicker Noise

High performance ADC


31

Considering Table 1, the Low-IF topology remained a promising architecture for the BLE

receiver front end. The superheterodyne topology, although having excellent selectivity and

sensitivity, its complexity makes it excessive for the BLE standard. Although multiple stages

of filtering would be required in some applications, the typical applications of BLE do not

require such high performance. Furthermore, this will be at the cost of higher power

consumption and area. On the other end of the spectrum, the direct conversion receiver

would be excellent for power savings in BLE. However, the DC-offset and flicker noise

combined with a low voltage supply could potentially deteriorate the overall performance of

the receiver from a large noise figure. The Low-IF receiver is an excellent compromise

between the two aforementioned topologies, with respect to power consumption, noise

performance, and complexity.

2.6 Challenges

2.6.1 Portability

The Bluetooth Low Energy standard has been designed specifically for portable, battery-

powered devices. As such, power consumption must be minimized in order to maintain

portability of the device. Unlike cellular phones which carry large bulky batteries, devices

designed specifically for the Bluetooth Low Energy standard use small, coin cell batteries

which are intended to last for weeks under normal usage. Since portability is a major factor

for BLE devices, the option to also operate the device under alternative energy forms is a

feature that is in-line with Bluetooth’s vision for the wireless standard. Energy harvesting

solutions, including photovoltaics, remains a feasible source of energy for many low power


32

applications and has proven to be a welcomed feature in scenarios where ambient light is

available.

Unfortunately, research in photovoltaics has not yet matured enough to provide high

efficiency and open-circuit voltages across solar cells to power most CMOS technologies.

The typical CMOS technology operates on voltages of 1V, 1.2V, 1.8V or even 3.3V, while

the voltage generated by a typical photovoltaic cell is less than 1V [19]. In order to facilitate

dual-mode operation, the proposed receiver front end must also operate on a sub-voltage

supply. The IBM 130nm CMOS technology chosen for this project operates on a maximum

VDD of 1.2V. This can be reduced, however, at the cost of voltage swing. For BLE

applications this trade-off of power consumption to SNR can be tolerated. As such,

operation of a 0.8V supply proves to be an ideal compromise between facilitating energy

harvesting solutions and also providing sufficient headroom for transistors to be safely

stacked and in the correct region of operation.

Chapter 3

Front End Receiver

3.1 Starting Point: LMV Cell

A novel receiver front end topology known as the LMV was first introduced by Liscidini

et al. in 2006 [1]. The LMV cell takes advantage of current recycling techniques in order to

merge three fundamental blocks in a Low-IF receiver (LNA, Mixer, VCO) into a single,

power efficient block. Stacking circuit blocks can facilitate the recycling of bias current

across multiple blocks to reduce the overall current consumption and consequently result in

power savings. A typical example of this is in LNA and Gilbert cell implementations, where

the LNA is cascoded with the Gilbert mixer [10]. In this way the bias current for the LNA is

shared by the two branches of the Gilbert cell. For the LMV cell, the mixing block is merged

with LO generation in what is known as a self-oscillating mixer (SoM) and stacked with the

LNA. When cascading blocks for current recycling, a low supply voltage can severely limit

the number of blocks that can be stacked. The LMV cell is able to avoid this by exploiting

similarities the blocks and implement device sharing. For example, as the differential pair in

an oscillator behave like switches that operate at the LO frequency, much like in a mixer,

these two can be merged into a single block. As power consumption is priority in the design

CHAPTER 3: FRONT END RECEIVER

34

of the front-end receiver for Bluetooth Low Energy, the LMV cell promises to be an

excellent platform, due to its compact, power efficient design. This chapter focuses on the

design of a Low-IF Receiver for Bluetooth Low Energy, based on the LMV cell, and also

following its core philosophy of current recycling and device merging techniques for an

overall power efficient design. In the next few sections, the operation of the LMV cell will be

described along with the areas of which it can be improved. Consequently, the design

solutions which address these issues will be presented, along with the design of a novel low

power complex filter that follows the BLE-based LMV cell.

3.1.1 LMV Cell Overview

Figure 17: LMV Cell topology.


35

As shown in Figure 17 is the LMV cell with double switching pair self-oscillating

mixer. The operation is as follows: In each half period of the LO frequency (dependant on

the LC tank values), the cross coupled transistors M1 and M2 alternate between open and

closed positions. When M1 is open (and consequently M4), M2 and M3 are closed. Two

conversions take place: an upconversion of the DC current and downconversion of the

superimposed RF signal [1].

The DC upconversion occurs as result of DC current used to sustain the tank oscillation

alternating between the switching pair M1 and M2 at the tank frequency. Capacitor, Cdiff,

which presents a low impedance path for RF and high impedance for IF, allows the up-

converted “DC” current to pass through transistors M4 and M1 during one half of the tank

period and through M3 and M2 during the second half. To avoid degeneration of the M1 and

M2 leading to an insufficient current to sustain the tank oscillation, the following condition

must be met:

, [1] (3.24)

where gm refers to the transconductance of transistors M1, M2.

Contrarily, RF downconversion occurs due to the mixing functionality of the Self-Oscillating

Mixer. Switching pair M1 and M2, and correspondingly M4 and M3, downconverts any RF

signal at Vin to the IF frequency. Recall that Cdiff presents a high impedance for low

frequencies, forcing the downconverted signal to pass through the IF load instead, producing

a differential voltage that can be sensed [1].


36

3.1.2 Challenges

The LMV cell with its stacked architecture, although an excellent strategy for power

savings, is not without limitations. Namely, in a Low-IF topology where quadrature paths are

required for image rejection the LMV cell would require the usage of two tanks, drastically

increasing the size of the design. Furthermore, parasitic capacitances at the IF output node of

the LMV cell limit the conversion gain (ratio of output voltage at the output frequency to the

input voltage at the input frequency) of the LMV cell for both current and voltage mode

solutions. Finally, the originally proposed LMV cell only allows for the equivalent of half the

bias current producing oscillations at the tank, also playing a factor in reducing the

conversion gain of the cell. These limitations will be briefly described next, for motivation

for a modified design.

3.1.2.1 N-Type VCO

Typically the current required to sustain the oscillation in the tank is often the limiting

factor for current consumption based on the quality factor of the tank [20][21]. While a low

quality factor tank would require biasing larger biasing current to counteract tank losses, it

was shown to negatively affect the conversion gain of the LMV cell:

[22] (3.25)

where LT and QT is the tank inductance and quality factor, respectively. A poor conversion

gain could have an impact on the overall SNR of the RX architecture and may require

additional stages to compensate for the loss in gain. To counteract this, a variation of the


37

original LMV cell adopting a differential LC tank has been implemented in [22] and has

showed considerable power consumption savings and conversion gain improvement over the

original scheme. By adopting a differential tank and N-type active devices, Tedeschi et al.

has shown that the quality factor of the inductor plays a negligible effect on the conversion

gain of the SoM, allowing the use of a high quality factor inductor while consuming less

current.

LO-LO+LO-LO+

2CT2CT LTLT

LT LT

CT

CMCM

Common-mode LC Tank Differential LC Tank

Figure 18: Differences between the common mode and the differential LC tank.

There is still however, an ultimate bound to the current that can be used to sustain the

oscillation. In both configurations of [1] and [22], the negative conductance that is required

to sustain the oscillation can only be generated by a single NMOS or PMOS transistor at a

time. Tanks with a poor quality factor would therefore require a considerable amount of bias

current in order to sustain oscillation. This disadvantage typically plays a dominant role in

the power consumption of most receivers.


38

3.1.2.2 Quadrature Generation

In typical receiver architectures, the phase shift required for quadrature outputs is

generated from an RC phase shift from a single LO. This implementation, however, is highly

sensitive to PVT variations leading to a poor quadrature (imperfect 90o shift) and LO

amplitude losses from the RC filter cut off [23] [1]. The original LMV cell utilises a

solution presented by Rofougaran et al, where an additional set of cross-coupled transistors

are added in parallel to the VCO switching pair to absorb the current from the negative

resistance to shut off the oscillator when the I and Q tanks are in-phase or anti-phase [23]. In

this implementation, oscillation in the two tanks can only co-exist if they are in quadrature.

While this is a clever solution to producing accurate RF quadrature generation, it requires a

large amount of space owing to the 4 LC tanks proposed in [1] for this operation. While

using an external inductor could be a potential solution, it is less attractive to an integrated

alternative because of the additional headroom that is required by external components as

opposed to a silicon chip.

3.1.2.3 Loss Mechanisms

The LMV cell suffers from a parasitic capacitance Cpar at the output of the cell which

limits the conversion gain of the block. In an ideal mixer the conversion gain, the voltage

gain between the input RF signal and output IF signal, is 2/pi, assuming hard switching of all

transistors (i.e. square waveform) [10]. In the case of the LMV cell, however, parasitic

capacitances at source of M1/M2, drain of M3/M4, and input to the TIA produce an

unwanted load resistance at the IF output node in parallel to the tank.


39

To analyze the effect of Cpar for high frequency and low frequency signal components,

the drain currents of M3 and M4 was decomposed into two components, a common-mode

current and a differential current. This is based upon the simplification of drain currents of

the transistor pair, showing the switching behaviour of the transistors:

(3.26a)

(3.26b)

Note that when the sign of the LO voltage is positive, all current passes though M3

while M4 is off, and vice versa. For calculating conversion gain, it was shown that for an RF

signal imposed on the input bias current will similarly decompose into a CM and differential

component, I(ωRF) and +/- I((ωRF- ωLO)t). This is shown in Figure 19.

Figure 19: Decomposition of an RF input signal into a differential and common mode

component.

CM Differential


40

For the low frequency differential component, Cpar heeds no affect for the current-based

LMV cell, as all current goes into the virtual ground. For the high frequency common-mode

component, the impedance presented by Cpar in parallel to the tank effectively reduces the

conversion gain of the cell to:

[22] (3.27)

Notice that, when Cpar = 0, the conversion gate equals to the ideal 2/pi as expected and for

large Cpar the conversion gain reduces to 1/pi. This capacitance must be limited to values

less than 200 fF in order to not drastically reduce the gain of the LMV, based on simulation

results shown in [1]. Overall, this ultimate bound poses several trade-offs for the designer.

Assuming that the critical node would be dominated by the transimpedance amplifier (in the

case of a current-mode LMV solution), special care must be taken to limit the input

capacitance which may or may not be at the cost of the TIA gain. Limiting the input

capacitance of a TIA may require smaller sized transistors and in turn a lower gain for the

TIA stage. Ultimately, the presence of this parasitic capacitance produces unwanted affects

which reduce the versatility of the LMV cell.

In the next section, design solutions will be presented which address these fundamental

drawbacks of the previous LMV architecture, namely the number of coils, parasitic losses,

tank power consumption.


41

3.2 Design Solutions

3.2.1 P-N VCO

In addition to the power savings of using a differential LC tank for both I and Q paths,

by adopting a complementary cross-coupled pair the tail current required to bias the LMV

cell for sustaining the oscillations can essentially by cut in half. Consider first a single a

traditional N or P-type VCO with differential LC tank shown in Figure 20, with total

inductance 2L and capacitance C/2. The resistive losses of the tank can be denoted as 2Rp.

To ensure continuous operation of the oscillator, the negative differential resistance of the

NMOS pair must be greater than the losses R present in the tank which cause the oscillation

to die out.

C/2

IBias

2L

To Mixer and LNA

-2 gm

2L

C/2

Cross Coupled Pair

2Rp

Figure 20: Traditional N type VCO and its equivalent small signal model.


42

At resonant frequency the tank behaves as an infinite impedance, leaving only tank

losses to deteriorate the oscillations. In this cross-coupled topology, an active device

replenishes the energy lost in the tank (2Rp) in order to maintain oscillations by presenting

negative resistance. This can be shown to be equal to -2/gm. For oscillations to occur, the

negative resistance must be able to compensate for tank losses. That is,

(3.28)

Adopting a complementary cross-coupled pair shown in Figure 21, however, can lead to a

more power-efficient solution.

2L

C/2

IBias

To Mixer and LNA

-2 gm

2L

C/2

N - Cross Coupled Pair

2Rp

-2 gm

P - Cross Coupled Pair

Figure 21: Complementary Cross-Coupled VCO and its equivalent small signal model.


43

Following the same analysis and using the same tank, it can be seen that there are now two

cross-coupled pairs to compensate the same loss 2Rp. That is, for the same LC tank and bias

current,

(3.29)

Therefore in the complementary cross-coupled pair, since there are two active elements

which produce twice the negative resistance to counteract the same losses, each active

element can be biased with half the current of the original scheme, leading to power savings

of up to four times the traditional N or P type topology.

Keeping the same bias current constant, the complementary cross-coupled pair VCO

also allows for increased output amplitude. Consider the first case of a traditional NMOS or

PMOS VCO with differential LC tank. For one half period, one NMOS is completely off

while the other carries the entire tail current. This tail current then exits through only one of

the LC tanks through the centre tap of the inductor. Since it only passes through one of the

inductors, of which has an equivalent resistance loss of R, the peak voltage is limited to (2/π)

R*Ibias.

Now consider the P-N structure. In this topology, for a single switched period the

entire tail current passes through transistor M1, flows entirely into the LC tank, and out

through transistor M3. Here, the current is able to pass through both inductors and therefore

produces a peak voltage of (4/π) R*Ibias. By having twice the output voltage, the phase noise

of the overall block can be improved by a maximum of 6 dBc/Hz and is therefore very

desirable when designing for strict wireless standards.


44

The main benefit sought out by adopting this particular topology is the savings that

can be realized by cutting the current consumption in half, while still achieving the same

peak voltage for the same LC tank. Typically, the ultimate bound of the power consumption

is from the consumption of the VCO, based on the current required to sustain oscillation in

the tank. Since integrated tanks are preferred to maintain a compact design, the cost is that of

a lower quality factor tank than that one using external components. Lower quality factor

tanks, however, experience much higher losses and therefore require larger tail currents to

counteract the tank losses, resulting in higher power consumption. Therefore any savings

that can be achieved in the LO generation stage can directly lead to significant savings for the

overall power consumption of the receiver. In this direction, the P-N VCO configuration was

adopted to replace that of the one in [22].

Each design solution typically invokes a cost to pay for benefits that are reaped. In this

case, the cost is an extra overdrive voltage from the additional cross-coupled pair. While this

may be solved by increasing the supply rail voltage, this is not ideal for two reasons. The first

being that higher rail voltage directly means increased power consumption. The second is to

maintain compatibility with low voltage supply systems. As mentioned previously, the

Bluetooth Low Energy standard was developed for autonomous wireless devices. Towards

this direction of portability, it would require efficient operation through coin batteries and

energy-harvesting solutions such as photovoltaic cells, which are limited in the voltage

supply that they can produce. In favour of maintaining this compatibility, the additional

overdrive voltage that is lost from the additional cross-coupled pair must be recovered

through some other means. In this design, it was done recovered through adopting a low-

voltage drop LNA with quadrature generation, and will be described in the next section.


45

3.2.2 Low Voltage Drop LNA with Quadrature Generation (QLNA)

In order to maintain a strong quadrature relationship across all channels of the Bluetooth

Low Energy band, quadrature generation occurs at the RF stage with the single voltage

overdrive QLNA. The quadrature is generated by an RC network that works as a low pass

filter for the VGS of M0I and as a high pass filter for the VGS of M0Q. By choosing C0 much

larger than gate-source capacitances of M0I/M0Q, a wideband quadrature relationship is

obtained between the two output currents (equal in magnitude around the filter cut-off

frequency 1/RC).

Figure 22: Single-voltage overdrive QLNA.

The two currents IRF-I and IRF-Q are proportional to the gate-source voltages which clearly

show the quadrature relationship:


46

(3.30)

(3.31)

At the tuned frequency the quadrature relationship is obvious:

(3.32)

Figure 23: Phase and magnitude relationship for I and Q paths.

This particular topology raises some concern that since it is a tuned network, the

quadrature relationship can only be valid at the tuned frequency 1/R0C0, and otherwise re-

tuning would need to occur for each respective Bluetooth Channel. This, however, is not true.


47

Since the filter cut off frequency 1/RC (which should be tuned to the center frequency of the

BLE band 2.44175 GHz) is much greater than the entire bandwidth of the Bluetooth Low

Energy spectrum, it can be shown that retuning of the capacitor is not necessary to maintain

strong amplitude matching and quadrature in channels that are even farthest from the cut-off

frequency:

Using the result from (3.33) we can define the amplitude mismatch as

(3.33

)

By choosing R0C0 such that ωo = 2.44175 GHz, we can see the amplitude mismatch in its

worst case is +/- 3.2% at extreme ends of the spectrum (2.402GHz, and 2.48GHz). Although

image rejection can be highly sensitive to amplitude and phase mismatches, the 3.2% gain

mismatch (or equivalently 0.1368 dB) can still result in strong image rejection, as shown in

Figure 14 of Chapter 2, given there is little phase mismatch. This will be analyzed next.

With respect to the quadrature, the phase response of both current pathways can be shown to

be constant for all frequencies:

(3.34)

(3.35)

(3.36)


48

Therefore the single voltage overdrive QLNA is able to provide excellent amplitude

matching across all Bluetooth Low Energy channels, with a constant 90 degree phase

relationship. It also requires only one voltage overdrive, which facilitates the usage of the

complementary cross-coupled VCO as described previously.

The design solutions presented in 3.2.1 and 3.2.2 were used in order to redesign the

original LMV cell. Specifically, the complementary cross-coupled VCO of 3.2.1 replaced the

N-Type VCO in [22] in order to cut the power consumption of the cell in half. This change,

however, reduced the voltage headroom by one voltage overdrive, which was compensated

for by adopting the single-voltage overdrive QLNA shown in 3.2.2. The result is shown in

Figure 24.

M1I M2I M1Q M1Q

M3I M4I M4Q

M5 M6

RFIN

IRFI IRFQ

Cin

Ls

C0

R0

M0I M0Q

BIAS

Pad

M3Q

Out I Out Q

Complementary Cross-Coupled VCO

Mixer

QLNA

Figure 24: Complete LMV topology with I-Q channels


49

3.2.3 TIA

As mentioned in Section 3.1, the LMV cell suffers from parasitic capacitances at the

output node, which limit the conversion gain of the cell. Special care must be taken to ensure

that the input capacitance of the TIA the follows the LMV cell be limited to below 200 fF

such that the conversion gain is not greatly reduced. The design implemented in this thesis is

able to mitigate these losses by adopting the Common-Gate TIA shown in Figure 25.

FromLMV Cell

BIASL

IIFIIF

CGS CGS

IBIAS IBIAS

MX MY

R

Vo+ Vo-

Figure 25: TIA Topology to reduce conversion gain losses at LMV output.

Note the resistor R that is added in series to the gate of the Common Gate TIA. The result is

the creation of an active inductor, with impedance 1/gm(MX,MY) + s(RCGS/gm(MX,MY)), where

gm is the transconductance of transistors MX/MY.


50

Vbias

CGS

=

Zin(CM) ≈ 1 s RC

gm gm +

R

Figure 26: Active inductor to resonate with LMV cell parasitic output capacitor, Cpar.

RG D

rds

Vtest

CGS gmVGS

S

Figure 27: Small signal analysis for the derivation of the active inductor impedance

Recall in Section 3.1, where Cpar was shown to limit the conversion gain of the

current-mode LMV cell due to the parasitic load for the common mode RF signal. Using the

topology of Figure 25, the active inductor can be tuned to resonate with Cpar at the output of

the LMV cell at the LO frequency. By doing so, the effect of Cpar can be eliminated as result


51

of the high-impedance it presents at resonance and the full conversion gain of 2/pi can be

restored.

For the differential signal the input impedance to the TIA remains 2/gm and is unaffected by

the resistance that is added to the gate of the transistor. The TIA is able to maintain a low

input impedance to absorb the current from the LMV cell’s output.

For a Cgs of 100 fF, an R gate resistance value of 670 Ohms was chosen to form an inductor

that resonated with the output node capacitance of 85 fF at the LO frequency.

3.2.4 Channel Selection and Image Reject Filter

As described earlier, one of the downfalls of adopting a Low-IF topology is that it

requires a baseband filter to perform image rejection and channel selection. Since this

processing is performed at low frequencies, it often requires very little power consumption

compared to the core LMV cell which operates at RF. Image rejection is performed through

a complex filter, where an asymmetry between positive and negative frequency offsets with

respect to the LO produce the attenuation at the image frequency to prevent any corruption of

the wanted signal [24][25]. Conventionally, complex filters can be implemented in both

passive and active realizations. However due to limited operating frequencies ranges and

poor gain, active implementations are instead favoured at the cost of power consumption.

Amongst active topologies, the active-RC and gm-C architectures are popular choices for

low-IF channel select applications, as they require little area and can provide large gain and

image rejection. The active-RC topology, however, has shown to typically consume more

power than the gm-C architecture while achieving poorer performance [26]. This can be


52

attributed to its operational amplifier requiring an operational bandwidth larger than the cut-

off frequency. In efforts to maintain high power efficiency, a current-recycle baseband

complex gm-C filter was adopted similar to the one proposed by Lin et al [27].

The filter proposed by Lin et al., however, suffers from a limitation in the choice of

complex pole that can be synthesized, due to a feedback loop that must be limited by

choosing a passband that is smaller than the center frequency of the pole. This will be

described in the coming section.

Gm-C filters are continuous time filters which uses an OTA (operational

transconductance amplifier) and capacitor as its basic building blocks. The transconductor

produces a current proportional to the input voltage which is then integrated on a shunt

capacitor to produce an output voltage. This produces a first-order low pass response.

However, for application of BLE channel selection the general first-order transfer function is

insufficient. Further, in order to reject the image adequately there must be an asymmetry in

the positive and negative frequency offset of the LO. This is performed by transforming the

filter above into a “complex” filter, by shifting the pole along the imaginary axis to form a

complex pole, as shown in Figure 28. This shift transformation (𝑠 → 𝑠 + 𝑗𝜔𝑜) can be

realized by cross coupling between the real and imaginary signal paths, such that the

frequency response is no longer centered about the zero frequency [17][25].


53

Figure 28: Gm-C Complex-Filter Topology

The real component of the pole is synthesized through the transconductances gmRE in shunt

with the capacitance C having:

(3.37)

while for the imaginary part, the transconductances gmIM with cross-connected I and Q paths

are used. The frequency shift is proportional to gmIM and is given by:

ωP = ωRE

ωRE

ωP = ωRE + jωSHIFT

ωSHIFT

2ωRE

Real Filter Complex Filter

ω ω0 0

GainGain

C/2

C/2+ −

− +

+ −

− +

+ −

− + +

−

−

+

Vo,IP

Vo,IN

Vo,QP

Vo,QN

gmRE

gmRE

gmIM gmIM

gmin

gmin

Vin

jVin

Vout

jVout

Complex Load


54

(3.38)

These two components combined lead to the complex pole

(3.39)

In order to save power, Lin et al. have proposed the solution reported in Figure 29, where the

bias current is recycled among the transconductance stages [27]. In this topology, the bias

current of the input transconductor gmIN is re-used to implement both gmRE and gmIM. MIM

and MRE can be sized according to the frequency shift and bandwidth desired by the designer.

Unfortunately, the solution proposed by Lin et al. has an intrinsic limit in the achievable ratio

Q between the frequency shift and the filter bandwidth. This limit is due to a stability issue

that will be now analyzed.

C/2 C/2

Vo,Ip Vo,In Vo,Qp Vo,Qn

Vo,Qp Vo,Qn Vo,In Vo,Ip

MIM MRE MRE MIM MIM MRE MRE MIM

IN,Ip IN,In

Ibias

IN,Qp IN,Qn

Ibias

Ibias2

Ibias2

Ibias2

Ibias2

Figure 29: Proposed current reuse gm-C topology by Lin et al.


55

Vo,Qp Vo,Ip Vo,Qn Vo,In

Loop Gain L(s)

MIM MRE MIM MRE MIM MRE MIM MRE

Q

Q Q Q Q

Figure 30: Positive feedback loop which must be maintained stable by keeping Q<1.

Analyzing the structure in Figure 29 it is possible to verify that the cascade of the

transconductances gmRE and gmIM creates the positive feedback loop shown in Figure 30.

The positive loop is formed by the four common-source amplifier stages MIM loaded by the

diode-connected transistor MRE. It can be easily seen that the gain of each stage is equal to Q

while the overall loop gain is equal to:.

(3.40)

Since the stability of this positive loop is guaranteed only when |L(s)| ≤ 1, Q must be lower

than one, and consequently,

(3.41)


56

Since gmIM cannot exceed gmRE, the frequency shift ωshift must be lower than the filter

bandwidth preventing the synthesis of a bandpass filter. This constraint can particularly

become an issue in low-IF receivers, where flicker noise can corrupt the signal band unless a

sufficiently high IF frequency is chosen. An example is the channel selection filter for a

Bluetooth receiver generally centered at 2MHz with a bandwidth of 1MHz.

In order to safely synthesize any complex pole without violating stability conditions

the original topology proposed by Lin was modified, introducing an additional cross-coupled

pair as shown in Figure 31. Contrary to the original complex gm-C cell, with this new

topology there exists an infinite number of combinations that can be used to synthesize any

kind of ratio Q while preserving the stability.

The key idea is to add an additional degree of freedom by having a common mode gain that

differs from the differential one.

IN,Ip IN,In

Ibias

C/2

Vo,Ip Vo,In

Vo,Qp Vo,Qn

MIM MRE MIMMNEG

C/2

Vo,Qp Vo,Qn

Vo,In Vo,Ip

MIM MRE MIMMNEGMRE MRE

IN,Qp IN,Qn

Ibias

Ibias2

Ibias2

Ibias2

Ibias2

Figure 31: Filter topology implemented.


57

For differential signals, the cross coupling of transistor MNEG creates a negative conductance

in parallel with the diode-connected transistor MRE, leading to an effective transconductance

equal to:

(3.42)

The reduction of the transconductance gmRE results in a new location of the complex pole

given by:

(3.43)

(3.44)

(3.45)

The filter bandwidth is now determined by gmRE(eff) (i.e. the difference between the

transconductances gmRE and gmNEG). The new ratio Q’ can be written as:

(3.46)

Notice that the effect of the negative transconductance is to increase the quality factor

of the filter. It does this in the same way a negative resistance in parallel to an LC bandpass

filter would, by injecting current to offset the loss resistance associated with the tank which

dampen the oscillation.

For the common mode signals, as the cross-coupled transistors no longer acts like a

negative conductance, the gain of each stage differs from (3.45) and thus L(s) is no longer


58

equal to Q4. The additional transconductance reduces the common mode gain of the stage

(and with it the positive loop gain). The new expression of the loop gain is given by:

(3.47)

Once again, the positive feedback loop described by (11) reaches instability when L(s)>1.

However in this case, this occurs when:

(3.48)

If the gain of each stage is not equal to 1, MNEG experiences an attenuation or boost of its

transconductance (gmNEG) proportional to the gain across the transistor.

The relation between the stability and the complex pole position can be studied by

rewriting (3.46) as a function of Q’ and a new parameter α, defined as the ratio between the

negative and positive transconductances (i.e. α = gmNEG/gmRE).

(3.49)

The contour plot of (3.48) is drawn in Figure 32 and clearly shows all possible sets

(α, Q’) that yield L<1. A higher stability margin is obtained by choosing α in the darker

areas. Notice that for higher shift-to-bandwidth ratios (Q’ increasing), the available set of

solutions to guarantee highest stability slowly decreases and requires the use of larger α’s.

This can be unfavourable for two major reasons: stability and noise.


59

L(s)|DC:

Figure 32: Contour plot showing the set of possible α and Q’ where the positive feedback

loop is stable.

As the ratio α approaches 1, this means that the negative transconductance used

becomes closer and closer to the value of the positive transconductance. Recall that when

these two are equal the filter becomes unstable, since the effective transconductance

approaches 0 and the Q’ in (3.45) reaches an infinite value. As the contour plot experiences a

narrowing in the solution set, it also translates to greater sensitivity. The plot shows steep

transitions in loop gain surrounding even the most stable of regions for large Q’ values. This

may produce certain harmful effects during fabrication where mismatch and process

variations can cause deviations from the desired transconductance values. While theoretically

it is possible to achieve any Q’ using this topology, the imperfections described above make

it advisable to limit the Q’ to a choice of <10, based on limiting the ratio α to 0.8 for a

considerable margin before the filter becomes unstable. This limitation, however, does not


60

take away from the versatility of the filter. A Q’ of 10 means a bandpass that is centered at a

frequency ten times greater than its width, an application that is typically not used in favour

of achieving higher gain and lower power consumption by using a lower IF.

The second reason why larger α is unfavourable is simply because it translates to

larger transistors and a thus a larger noise contribution at the output. Since the negative

transconductance produces still a positive noise component, when the α tends to 1 the noise

contribution of the two combined transistors becomes closer to the double of MRE. One

redeeming factor with regards to this, however, is that the noise contribution of the filter is

dominated by the input transconductance, especially when a high gain is designed for the

cell. Therefore, the increased noise contribution from the larger load transistors is

overpowered by the noise from the larger input transistor, reducing its significance to the

overall noise performance of the cell.

When comparing the stability of the proposed solution to that of Lin et al, the plot

also shows that when no negative transconductance is used (i.e. α=0), stability can be

guaranteed only for Q’<1 [2]. From this comparison, the flexibility of the proposed solution

is obvious. Specifically, this is the freedom to synthesize any complex pole desired, beyond

the original limitation of Q’<1.

According to the topology of Figure 31, two complex poles (with two different

frequency shifts) were synthesized and cascaded in order to realize a flat band response with

center frequency 2MHz and 1MHz bandwidth. The complex frequency response will be

shown in Section 4.


61

INI

+

_

INQ

+

_

POLE 1 POLE 2

BIAS

BIAS

OUTI

OUTQ

To Combiner

Figure 33: A flatband response was created by cascading two complex poles with different

shifts to realize the complex bandpass with Q=0.7 about 2MHz.

3.3 Summary

The LMV solution presented by Tedeschi et al. was able to achieve good performance

with respect to a low power wireless solution. In order to improve its performance for the

Bluetooth Low Energy wireless standard, a number of changes were proposed to improve

the power efficiency and overall performance of the system. These changes include adopting

a complementary cross-coupled VCO and introducing a single-voltage overdrive LNA to

maintain using a low voltage supply. Furthermore, loss mechanisms of the LMV cell are

reduced through implementing a novel TIA at the baseband output. Finally, in the philosophy

of power efficiency, a current re-use gm-C complex filter was used for channel selection and

image rejection. A block diagram depicting the top level architecture employed is shown in

Figure 34, illustrating the quadrature generation at the LNA and the use of a shared LC tank

for downconversion in both I and Q paths. The results of the above design solutions are

summarized in Chapter 4.


62

Figure 34: Top-level Diagram of BLE Front-End.

Chapter 4

Measurements

4.1 Silicon Die and Test Board

The sub-mW receiver front-end design described in Chapter 3 was fabricated in IBM

CMOS 130nm and is shown in Figure 35. The chip occupies a footprint of 1.0mm by 1.2mm,

with an active area of 0.25mm2.

In order to test the functionality of the IC, a printed circuit test board was developed to

provide the chip with an external voltage supply source and biasing control voltages. Voltage

regulation was provided by the Semtech SC4215H, providing a tuneable output voltage of

0.8V from a 3.3V-5V source. The National Instruments RIO USB-7856R programmable

FPGA was used to generate bias voltages according to the pin summary above. For the

purpose of biasing, the RIO’s analog and digital I/O ports were programmed and interfaced

with the IC via a 68 pin D-type port. The use of an external FPGA controller such as the

RIO allowed for increased flexibility when adjusting for variations between the actual silicon

die and chip simulations, providing 12-bit resolution between +/- 1V. For reduced parasitic

CHAPTER 4: MEASUREMENTS

64

effects, the IC die was wirebonded directly onto the board by Silitronics Assembly in San

Jose, California.

LMV

+ TIA

FILTER

1.2

mm

1 mm

Pole 1

Pole 2

Figure 35: Die micrograph (left) with its wirebonded implementation on PCB (right). Filter

layout is also shown (below)


65

RFIN

BIAS TUNING

IC

OUTPUT

VOLTAGE REGULATION

FPGACONT’L

5"

5.5"

Figure 36: PCB Fabricated for chip measurements.

4.1.1. Measurement Equipment

The following table summarizes the equipment used to obtain the measurements that are

described in this section:

Table 3: Test Equipment

Equipment Type Model

DC Power Supply Agilent E3648A

Network Analyzer Rohde & Schwarz ZVH8

Vector Signal Generator Rohde & Schwarz SMW200A

Signal and Spectrum Analyzer Rohde & Schwarz FSW


66

4.2 Chip Measurements

4.2.1. Input Matching: S11

Input matching to 50 Ohms was obtained through an external inductor and capacitor

preceding the LNA block to reduce transmission losses and maximize power transfer. By

using external components, matching can be optimized for each chip, which may have

unique impedances due to imperfections in PCB routing, wirebonds, etc. Similar uses of

external components have also been employed in other BLE (or equivalent) receivers in light

of component accuracy and ability to achieve strong matching [22].

Mathematical modelling of the receiver on the Wolfram Mathematica Suite showed an

input impedance of:

(4.50)

By setting Re[Zin] = 50 Ohm and Im[Zin] = 0, the following initial component values were

obtained:

(4.51)

(4.52)

Actual component values were later tweaked in-lab through S11 measurements on the Rohde


67

& Schwarz ZVH8 Network Analyzer in order to re-calibrate the impedance to 50 Ohms

around 2.44175 GHz.

RFIN

Cin

Ls

Pad

To LNA

Cin = 900 fFLs = 3.9 nH

Figure 37: Matching circuit and component values.

BLE Standard Bandwidth

Figure 38: S11 Measurements.

Measurement results show that good matching (<-15 dB) is obtained over the BLE standard

bandwidth of 83.5 MHz.


68

4.2.2. AC Transfer Function

The AC Transfer Function shown in Figure 39 was obtained by inputting an RF signal of

power -70 dBm into the receiver and sweeping the LO frequency. The transfer function

shown corresponds to the first channel of Bluetooth Low Energy at 2.402 GHz.

Measurements report a maximum gain of 55.5 dB and a minimum image rejection of 30.5 dB

across the 2 MHz channel bandwidth. Notice the asymmetry about the center fLO axis, which

is as result of the complex filter response used to suppress the image from corrupting the

wanted signal. This measurement is representative of the worst case measure in terms of

maximum gain and image rejection.

Figure 39: AC Transfer Function.


69

Further measurements were taken to evaluate the gain and image rejection across all

channels in the Bluetooth Low Energy Standard. While gain remains relatively constant over

the entire band, image rejection varies up to 7 dB. This could be attributed to imperfections

in the quadrature at the input of the filter. Regardless, the minimum 30 dB exceeds the 21 dB

demanded by the Bluetooth Low Energy standard.

Figure 40: Gain and Image Rejection for each BLE Channel.

4.2.3. Noise Figure

The noise figure was measured using the Rohde & Schwarz ZVH8, by analyzing the

noise at the input and output. This relation can be seen by analyzing the equation shown in

Chapter 2, and rewritten in terms of the noise at the output and the output referred noise at

the input. This can be clearly seen by the equation shown in Chapter 2, but rewritten in terms

of the noise at the input and the output:


70

(4.53)

By terminating the input of the receiver with 50 Ohm source, the output referred input noise

can be calculated as -174 dBm/Hz + Gain of the Receiver. The output noise was measured by

powering on the receiver and measuring the marker noise at the output. This measurement

was repeated for each channel of the Bluetooth Low Energy standard, and is shown in Figure

41.

Figure 41: Noise Figure Measurement for each BLE Channel.

Notice that the Noise Figure is relatively constant over the Bluetooth Low Energy spectrum,

ranging from a minimum value of 15.1 dB and maximum of 15.8 dB. For the BLE

application, a maximum of 30 dB is tolerable.


71

The noise figure can also provide insight into the sensitivity of the receiver, another

metric to measure the performance of a receiver. Recall that the sensitivity refers to the

minimum signal power at the input that is required to obtain a given SNR at the output. The

noise figure is the only attribute that is defined by the design of the receiver and directly

impacts the sensitivity of the system. Based on the worst case measure of noise figure, 15.8

dB, the sensitivity of the receiver is -84.2 dBm. This exceeds the minimum sensitivity of -70

dBm required by the Bluetooth Low Energy standard.

4.2.4. Linearity

The linearity performance of the receiver was measured by the third-order

intermodulation product (IIP3). Using the Rohde & Schwarz SMW200A, two tones were

input into the receiver at a 5MHz and 8MHz offset from the LO, in order to produce an in-

band intermodulation product at 2MHz. By recording the input tone power at the 5MHz and

8MHz offset, along with the third-order intercept power at 2MHz, the IIP3 can be calculated

according to the following expression:

[10] (4.54)

IIP3 measurements were recorded for each channel and is shown in Figure 42. Notice that the

IIP3 remains relatively constant over the entire Bluetooth Low Energy spectrum, having a

minimum value of -16.75 dB and maximum value of -15.2 dB.


72

Figure 42: IIP3 Measurement for each BLE Channel.


73

4.3 Performance Summary and Comparison

A summary of receiver performance is reported in Table 4, along with a comparison to

the current state of the state of the art in Bluetooth and ZigBee receivers.

Table 4: Performance Summary and Comparison.

* Baseband filter not included in measurement ** Measured at moderate or minimum gain

[28] [20] [22] [27] [21] This work

Power Consumption [mW] 1.1 1.8 3.6 2.7 1.6 0.6

RF Input Freq [MHz] 2400 2400 2400 2400 2400 2400

Voltage Gain [dB] (min/max)

- 57.8 75 55 83 55.5/56.1

Sensitivity [dBm] (min/max) -81.4 -88 - - -94 -84.9/-84.2

NF [dB] (min/max) 16*/16.6* 15.7 9 9 6.1 15.1/15.8

IIP3 [dBm] (min/max) -2.9** -18.5** -12.5 -6 -21.5 -15.8/-16.8

IRR [dB] (min/max) - 37/40 35 28 - 30.5/37.3

Supply Voltage [V] 1 1 1.2 0.6/1.2 0.3 0.8

Technology 130nm 65nm 90nm 65nm 65nm 130nm

Active Area [mm2] - 0.45 0.35 0.26 2.496 0.25

Compared to the solutions present in literature with similar performances [28] [20]

the proposed design consumes only a fraction of the power. The other designs reported in the

table have better noise figures but also report much higher power consumption. Although in

this case a fair comparison is difficult to make, it can be verified that the power consumption

required to generate the local oscillator in [22], [27], [21] is either higher or comparable to

the power consumption of the entire receiver front-end proposed in this paper. Therefore,

even if the performances of the other RX chains were significantly compromised in favor of


74

a lower consumption (by scaling the power consumption of LNA, mixers, and base-band

section), it would be difficult to achieve an overall power consumption below the one

reported here since the power necessary for LO generation cannot be arbitrary scaled.

Despite using a 130nm technology node, the proposed design has demonstrated the lowest

power consumption, occupies one of the smallest areas, and is fully compliant with BLE

requirements.

Chapter 5

Conclusion

5.1 Summary

This thesis describes the development of a sub-mW Receiver front-end for the Bluetooth

Low Energy standard suitable for energy-harvesting supplies. The Bluetooth Low Energy

wireless standard was designed specifically for low power applications, where performance

can be sacrificed in favour of longer battery life between charges. In these applications,

typically for use in closed-environments with its respective signal source nearby, blocker

tolerance, and noise performance need not be rigorous, allowing the power consumption of

the receiver to be reduced considerably. In this direction, the sub-mW receiver front-end was

designed, fabricated, and tested to show full compliance to the BLE standard. The receiver

achieves a gain of 55.5dB with a minimum 30.5dB rejection of the adjacent channel in the

worst-case channel. The noise figure at the worst-case channel is 15.8dB, resulting in a

minimum receive sensitivity of -84.2dBm and IIP3 greater than -17dBm.

CHAPTER 5: CONCLUSION

76

5.2 Future Work

The sub-mW receiver front-end for Bluetooth Low Energy presented in this thesis

showed good conformity to the BLE specifications such as blocker tolerance, gain, and noise

figure. It is still, however, only one building block amongst others in a complete receiver

chain. Further work is required to expand from the scope of this thesis (receiver front-end) to

realize a fully functional receiver.

RX Front End ADC

100110...01010

TX Front EndDAC100110...01010

TRANSMITTER RECEIVER

THIS THESIS

Figure 43: Scope of this thesis.

The RX front end demonstrated the ability to amplify and down-convert an RF

sinusoid centered at 2.4-2.5 GHz to low-IF, but was not tested with a GFSK signal as per the

BLE standard. Future work would involve producing a modulated BLE signal as input to the

RX front-end and performing demodulation, as the ADC would, to recover the transmitted

bits. This testing can be performed using R&S FSW and SMW200A models and should

eventually result in the development of an ADC block to complete the receiver chain.

For a more robust design in the presence of LO non-idealities such as frequency-drift,

a phased-lock loop (PLL) should be designed to regulate the voltage at the varactor to

maintain a fixed LO frequency. Since the LMV cell resembles a traditional cross-coupled

CHAPTER 5: CONCLUSION

77

VCO at RF it can be easily inserted to a typical PLL cell without further modification to the

cell. Research should also go into producing a power-efficient design, maintaining the

philosophy of current recycling.

Finally, since the transmitter and receiver perform reciprocal functions (that is, one

upconverts a signal and other other downconverts it), it would be also beneficial to

investigate the possibility of modifying the presented topology for performing transmitting

capabilities. This would require introducing a power amplifier at the output stage and

removal of the LNA. By doing so, a complete transceiver can be developed.

78

Bibliography

[1] A. Liscidini, A. Mazzanti, R. Tonietto, L. Vandi, P. Andreani, and R. Castello,

“Single-Stage Low-Power Quadrature RF Receiver Front-End: The LMV Cell,” vol.

41, no. 12, pp. 2832–2841, 2006.

[2] “Bluetooth Specification Version 4.1,” Bluetooth, 2013. [Online]. Available:

http://www.bluetooth.com.

[3] N. Cho, L. Yan, J. Bae, and H. Yoo, “A 60 kb / s – 10 Mb / s Adaptive Frequency

Hopping Transceiver for Interference-Resilient Body Channel,” J. Solid State Circuits,

vol. 44, no. 3, pp. 708–717, 2009.

[4] A. Chi, W. Wong, M. Dawkins, G. Devita, N. Kasparidis, A. Katsiamis, O. King, F.

Lauria, J. Schiff, A. J. Burdett, and S. Member, “A 1 V 5 mA Multimode IEEE

802.15.6/Bluetooth Low-Energy WBAN Transceiver for Biotelemetry Applications,”

J. Solid State Circuits, vol. 48, no. 1, pp. 186–198, 2013.

[5] H. Darabi, B. Ibrahim, and A. Rofougaran, “An Analog GFSK Modulator in 0 . 35um

CMOS,” J. Solid State Circuits, vol. 39, no. 12, pp. 2292–2296, 2004.

[6] J. Proakis and M. Salehi, Digital Communications. New York, NY: McGraw-Hill,

2008.

[7] B. Xia, C. Xin, W. Sheng, and A. Y. Valero-lopez, “A GFSK Demodulator for Low-IF

Bluetooth Receiver,” J. Solid State Circuits, vol. 38, no. 8, pp. 1397–1400, 2003.

[8] S. Otaka, M. Ashida, M. Ishii, and T. Itakura, “A +10-dBm IIP3 SiGe Mixer With

IM3 Cancellation Technique,” J. Solid State Circuits, vol. 39, no. 12, pp. 2333–2341,

2004.

[9] T. C. Carusone, D. Johns, and K. Martin, Analog Integrated Circuit Design, 2nd ed.

John Wiley & Sons, Inc., 2011.

[10] B. Razavi, RF Microelectronics, 2nd ed. Upper Saddle River, NJ: Prentice Hall, 2011.

[11] H. Darabi, S. Khorram, H. E. Chien, M. Pan, S. Wu, S. Moloudi, J. C. Leete, J. J.

Rael, S. Member, M. Syed, R. Lee, and B. Ibrahim, “A 2.4-GHz CMOS Transceiver

for Bluetooth,” J. Solid State Circuits, vol. 36, no. 12, pp. 2016–2024, 2016.

[12] M. P. Van Der Heijden, S. Member, and H. C. De Graaff, “A Novel Frequency-

Independent Third-Order Intermodulation Distortion Cancellation Technique for BJT

Amplifiers,” J. Solid State Circuits, vol. 37, no. 9, pp. 1176–1183, 2002.

79

[13] W. Kluge, F. Poegel, H. Roller, M. Lange, T. Ferchland, and L. Dathe, “A Fully

Integrated 2.4-GHz IEEE 802.15.4-Compliant Transceiver for ZigBee TM

Applications,” J. Solid State Circuits, vol. 41, no. 12, pp. 2767–2775, 2006.

[14] Q. Gu, RF System Design of Tranceivers for Wireless Communication. Springer, 2006.

[15] A. A. Emira and S. Member, “A Pseudo Differential Complex Filter for Bluetooth

With Frequency Tuning,” Trans. Circuits Syst. - II, vol. 50, no. 10, pp. 742–754, 2003.

[16] F. Behbahani, Y. Kishigami, J. Leete, and A. A. Abidi, “CMOS Mixers and Polyphase

Filters for Large Image Rejection,” J. Solid State Circuits, vol. 36, no. 6, pp. 873–887,

2001.

[17] K. W. Martin, “Complex Signal Processing is Not Complex,” Trans. Circuits Syst. - I,

vol. 51, no. 9, pp. 1823–1836, 2004.

[18] J. Crols, M. S. J. Steyaert, and S. Member, “Low-IF Topologies for High-Performance

Analog Front Ends of Fully Integrated Receivers,” Trans. Circuits Syst. - II, vol. 45,

no. 3, pp. 269–282, 1998.

[19] M. Danesh, S. Member, and J. R. Long, “Photovoltaic Antennas for Autonomous

Wireless Systems,” Trans. Circuits Syst. - II, vol. 58, no. 12, pp. 807–811, 2011.

[20] K. Wang, J. Koo, R. Ruby, and B. Otis, “A 1.8mW PLL-Free Channelized 2.4GHz

ZigBee Receiver Utilizing Fixed-LO Temperature- Compensated FBAR Resonator,”

ISSCC Dig. Tech. Pap., vol. 3, pp. 372–374, 2014.

[21] F. Zhang, K. Wang, J. Koo, Y. Miyahara, and B. Otis, “A 1.6mW 300mV-Supply

2.4GHz Receiver with -94dBm Sensitivity for Energy-Harvesting Applications,” pp.

456–458, 2013.

[22] M. Tedeschi, A. Liscidini, and R. Castello, “Low-Power Quadrature Receivers for

ZigBee (IEEE 802.15.4) Applications,” J. Solid State Circuits, vol. 45, no. 9, pp.

1710–1719, 2010.

[23] A. Rofougaran, J. Rael, M. Rofougaran, and A. Abidi, “A 900 MHz CMOS LC-

Oscillator with Quadrature Outputs,” ISSCC Dig. Tech. Pap., vol. 14, no. 5, pp. 392–

393, 1996.

[24] C. Laoudias and C. Psychalinos, “1.5-V Complex Filters Using Current Mirrors,”

Trans. Circuits Syst. - II, vol. 58, no. 9, pp. 575–579, 2011.

[25] B. Guthrie, J. Hughes, T. Sayers, and A. Spencer, “A CMOS Gyrator Low-IF Filter for

a Dual-Mode Bluetooth / ZigBee Transceiver,” J. Solid State Circuits, vol. 40, no. 9,

pp. 1872–1879, 2005.

80

[26] H. A. Alzaher, N. Tasadduq, and F. S. Al-ammari, “Optimal Low Power Complex

Filters,” Trans. Circuits Syst. - I, vol. 60, no. 4, pp. 885–895, 2013.

[27] Z. Lin, P. Mak, and R. Martins, “A 1.7mW 0.22mm2 2.4 GHz ZigBee RX Exploiting

a Current-Reuse Blixer + Hybrid Filter Topology in 65nm CMOS,” ISSCC Dig. Tech.

Pap., no. 25.5, pp. 448–450, 2013.

[28] J. Masuch and M. Delgado-restituto, “A 1.1-mW-RX -81.4 dBm Sensitivity CMOS

Transceiver for Bluetooth Low Energy,” Trans. Microw. Theory Tech., vol. 61, no. 4,

pp. 1660–1673, 2013.

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S W R F -E B L E T by Anith Selvakumar · 2019-03-12 · Anith Selvakumar Master of Applied Science Graduate Department of Electrical and Computer Engineering University of Toronto