RF Amplifiers · 2020. 4. 8. · Kampouris, Richard Barber, Milena Khazak, Farhang Vessal, Mathew Atekwana Amberetu, Dusan Suvakovic, Mehrada Ramezani, Sotoudeh Hamedi-Hagh, John

Multiharmonic Tuning Behavior of

MOSFET RF Power Amplifiers

by Yucai Zhang

A thesis submitted in conformity with the requirements for the degree of Master of Applied Science

Edward S. Rogers Sr. Department of Electrical and Computer Engineering University of Toronto

2001

O Copyright by Yucai Zhang 2001

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Multiharmonic Tuning Behavior of MOSFET RF Power Amplifiers

Master of Applied Science, 200 1

Yucai Zhang

Edward S. Rogers Sr. Department of Electrical and Cornputer Engineering

University of Toronto

Abstract

This thesis investigates multiharmonic tuning of RF power amplifiers using power

MOSFETs implemented in bulk silicon CMOS technology. The use of this technique may

lead to the low-cost implementation of the RF power amplifier integrated on the sarne chip

as the rest of the wireless transceiver.

The work proposes a complete classification of multiharmonic tuning into fow basic

modes: both odd/even harmonics SHORT (SS), odd harmonics SHORT and even harmonics

OPEN (SO), odd harmonics OPEN and even harmonics SHORT (OS), and both oddeven

harmonics OPEN (00) . Conventional power amplifiers c m then be characterized using

these modes of operation in so far as multiharmonic tuning is concerned. A systematic

multiharmonic tuning optimization procedure is introduced to find the optimal harmonic

terrninations.

The newly proposed 00 mode features a sinusoidal drain curent waveform containing

no harmonics, resulting in little or no energy wasted at harmonic fiequencies and yielding

high eficiency.

To study the multiharmonic tuning behavior of MOSFET RF power amplifiers, power

MOSFETs were implemented in a 0.25pm silicon CMOS process. For power amplifiers

using these MOSFETs, at 1.88GH2, the 00 mode yields the highest efficiency (PAE4lYo)

with a 23.3dBm output power at a 12dBm input power and at a 2.OV supply voltage.

Acknowledgrnents

1 would like to express my sincere gratitude to Professor C.A.T. Salama for his

insightful guidance and invaluable assistance throughout the course of this work.

1 am indebted to Mr. J. Illowski, Mr. P. Watson and Mr. M. Stubbs from Nortel

Networks for their technical advice and assistance with load-pull measurements.

My appreciation extends to al1 the staff and students in the Microelectronic Research

Laboratory. 1 am specially grateful to Jaro Pristupa for his assistance with CAD tools and

Dana Reem for her technical assistance during the chip testing. Thanks go to Anthoula

Kampouris, Richard Barber, Milena Khazak, Farhang Vessal, Mathew Atekwana Amberetu,

Dusan Suvakovic, Mehrada Ramezani, Sotoudeh Hamedi-Hagh, John Ren, Namdar Saniei

for al1 their help.

Thanks also to my wonderfùl fiiends who made my Iife at Uofï pleasant and

unforgettable. Especially, 1 would like to express my appreciation to Song Ye, Zhixian Jiao,

Rick Kubowicz and Wei An, for valuable discussion both technically and personally, and

the rest of my fî-iends: Hongfei Lu, I-leng Jin, Shuo Chen, Wei Yang, Mike Sheng, Edward

Chun Keung Yu, 1-Shan Michael Sun for constructive discussions and cheerfbl chats.

Special thanks to my other fnends, Ting Lu, Jun Zhang, Tingju Zhu, Yajuan Su,

Mengsi You, Jian Yang, for sharing both my hard tirne and good time.

My deepest appreciation goes to my parents and sister for their constant support and

encouragement.

This work was supported by the Natural Sciences and Engineering Research Council of

Canada, Micronet, CITO, Gennwn, Mitel, Nortel Networks and PMC Sierra.

Table of Contents

Page

............................................................................................... CHAPTER 1 Introduction 1

I . I RF Power Amplifiers for Wireless Communications ............................................. 2 .............................................................................................. 1.2 RF Power Amplifiers 4

.................................................................... 1.3 Objectives and Outline of the Thesis 1 1

CHAPTER 2 Theoretical Multiharmonic Tuning Behavior of MOSFET RF Power Amplifiers .....~..~.........b........................................ 14

2.1 Introduction ........................................................................................................... 14 ................................................................... 2.2 Analysis of Power Amplifier Modes 1 4

....................................................... 2.2.1 Conventional Power Amplifier Modes 1 4 ....................................... 2.2.2 Classification of Multiharmonic Tuning Behavior 21

................................................... 2.3 Multiharmonic Tuning Optimization Procedure 23 2.4 Multiharmonic Tuning Behavior of MOSFET PAS .............................................. 25

............................................................................................... 2.4.1 Device Design 25 2.4.2 MHT Optimization for PAE ........................................................................ 28

2.5 Summary ................................................................................................................ 34

CHAPTER 3 Experimental Results ........................................................................... *..38

3.1 Power MOSFET Implementation ......................................................................... 38 ............................................................................... 3.2 Power Device Characteristics 41

................................................................................. 3.3 MOSFET PA Characteristics 45 3 -4 Summary ............................................................................................................... -49

........................................................................................... CHAPTER 4 Conclusions ...51

APPENDIX A Harmonic Load Pull Measurement ................................................ ....*53

iii

List of Figures

Page

............................................... . Fig 1 . 1. Block diagram of a generic digital RF transceiver 2 ............................. Fig . 1.2. A generic schematic of Class A.B. AB or C Power Amplifier 6

Fig . 1.3. RF power and drain eficiency as a lùnction of conduction angle ....................... 7 Fig . 1.4. Class F power amplifier with even harmonic trap ............................................... 8 Fig . 1.5. Power and Efficiency contours vs . phases of TL of the MESFET PA ................. 9

................................................................................. . Fig 1.6. Inverse Class F wavefoms 10 ............................................................................... Fig . 2.1 : Waveforms of Class A/B/C 1 5

Fig . 2.2. Waveforms of ideal Class F mode ..................................................................... 17 ....................................................... Fig . 2.3. Conceptual "target" waveforms of Class E 1 9

Fig . 2.4: (a) Schematic of basic (low-order) Class E power amplifier ...................................................................................................... (b) Actual waveforms -20

.............................................. . Fig 2.5 : Classification of multiharmonic tuning behavior -22 ............................................................ . Fig 2.6. Basic circuit used in MHT optimization 24

............................ . Fig 2.7. Drain current for different gate width (gate length=0.25pm) 26 ........................................................ Fig . 2.8. Modified BSIM3V3 CMOS device mode1 -27

Fig . 2.9. Schematic used in ADS simulator ...................................................................... 28 ........................................ Fig . 2.10. PAE and Pout contours for SS mode (Pin=12dBm) -30 ........................................ Fig . 2.1 1 : PAE and Pout contours for OS mode (Pin= 12dBm) 30 ........................................ . Fig 2.12. PAE and Pout contours for SO mode (Pin= 12dBm) 31 ....................................... Fig . 2.1 3: PAE and Pout contours for 00 mode (Pin= 12dBm) 31

Fig . 2.14. PAE. Pout and Gain vs . Pin of the MOSFET ................................................... 32 .................................... Fig . 2.15. Drain voltage and current wavefoms in the 00 mode 32

.... Fig . 2.16. PAE and Pout contours vs . phases of TL of the MOSFET power amplifier 33 ...................................................... Fig . 3.1 : RF MOS ce11 layout with substrate contacts 39

....................................................................................... Fig . 3.2. RF input pad structure -39 ..................................................................................... . Fig 3.3. Layout of test structures 40

............................................................................... Fig . 3.4. Micrograph of the test chip -40 ....... Fig . 3.5. Ids-Vgs transfer characteristics of the MOSFET (W/L=200Opm/0.25pm) 41

....................................................... Fig . 3.6. Breakdown characteristics of the MOSFET 42 ................... Fig . 3.7. IDs-VDs characteristics of the MOSFET (W/L=200qim/0.25pm) 42

.............................................................. . Fig 3.8. fT andf,, of the CMOS power device 44 ................................. Fig . 3.9. Harmonic on-wafer load-pull measurement system setup 46

............................. . Fig 3.10. Measured characteristics of the MOSFET power amplifier 48 . . ......................................................... Fig A 1 : Harmonic Load Pull Measurement Setup -54

List of Tables

Page

............................................................... Table 1 . 1 : Characteristics of Wireless Standards 4 Table 2.1 : Optimal ZL(oO) of the MOSFET

..................................................... optimized for maximum PAE (Pin= 12dBm) 29 ............... Table 2.2. Simulated characteristics of the power device and power amplifier 35

Table 3.1 : Measured and Simulated fT and f,, ............................................................... 44 ............................. Table 3.2. Measured and simulated performance of each MHT mode 47

.................. Table 3.3 : Measured and Simulated Parameters of power MOSFET and PA 49

Chapter 1: Introduction 1

Introduction

In addition to persona1 communication products such as pagers and cellular phones,

wireless technology has impacted many other rapidly growing markets, for instance,

wireless local area networks (WLANs), global positioning systems (GPS), and RF

identification systems (RFIDs). A wide variety of system standards have been adopted to

support these applications. The explosive market of wireless communications is motivating

extensive research and design effort to develop communications devices with increasingly

higher performance, lower cost and low power consurnption.

Wherever there are wireless communications, there are transmitters; wherever there are

transmitters, there are RF power amplifiers. Power amplifiers (PAS) are used to amplie the

signal being transmitted to the necessary level needed to drive the antema at a particular

power output level, so that it can be received and decoded by the receiver within a certain

geographical area. Power amplifiers typically dominate the power consumption of the

transmitters (or transceivers), thus have critical impact on system performance and cost,

especially in low-voltage, low-power portable applications.

Advances in conventional CMOS technology have made this technology a promising

alternative for low-cost, low-voltage implementation and integration of wireless transceiver

building blocks, such as Digital Signal Process (DSP) cores, Low Noise Amplifiers,

Mixers, and other front-end ICs. However, RF power amplifiers, the bottleneck of wireless

Multiharmonic Tuning Behavior of MOSFET RF Power Amplifiers University of Toronto


transceivers, are still being implemented in expensive GaAs technologies or specific RF

LDMOS technologies, which prevent the integration of the power amplifiers into

transceivers. To enable a single-chip transceiver implementation and furiher reduce system

costs, it is highly desirable to develop RF power amplifiers h l ly compatible with

conventional CMOS technologies.

1.1 RF Power Amplifiers for Wireless Communications

Power Amplifiers in Wireless Transceivers

Digital Voice Modulator

Voice Coding Pulse O) ,* cornPressior * Interieaving * Shaping

I 1 Amplifier

Carrier

Down Converter

Ampl ifter Carrier

De-interleaving Voice 1

Decoding * Decompression - DAC - i Audio Speaker Amplifier

(b)

Fig. 1 . 1 : Block diagram of a generic digital RF transceiver (a) transmitter, (b) receiver

A generic digital RF transceiver is s h o w in Fig. 1 .la. On the transmitter side (Fig.

l.la), the voice signal is first digitized by an analog-to-digital converter (ADC) and



compressed to reduce the bit rate and hence the required bandwidth. Then, the data

undergoes "coding" and "interleaving" to format the data such that the receiver can detect

and minimize errors by performing the reverse process. Since rectangular pulses are usually

not optimum for modulation, the data is "shaped and modulated by the RF carrier

fiequency. After filtering, the signal is applied to the power amplifier which drives the

antenna. As illustrated in Fig. 1.1 b, on the receiver side, the signal received by the antenna

is amplified, downconverted, and digitized. Subsequently, demodulation, equalization,

decoding, de-interleaving, and decompression are performed in the digital domain. The

resulting data is then converted to an analog signal by a digital-to-analog converter (DAC),

amplified, and applied to the speaker.

Wireless Communication Modulation Schemes

The type of power amplifier in a wireless system depends on the type of modulation

standard used in the system. Digital modulation with binary baseband waveforms can be

performed by one of the following methods [ 1 ] : Amplitude Shift Keying (ASK), Frequency

Shift Key ing (FSK), Phase Shi fi Key ing (PSK), or Quadrature Amplitude Modulation

(QAW

In many applications, "quadrature modulation" is used to reduce the bandwidth

requirement [2]. Quadrature modulation includes two broad categories: quadrature phase

shift keying (QPSK) and minimum shift keying (MSK). QPSK includes specific options

such as Offset QPSK (OQPSK) and d4-QPSK. MSK has a widely used subset known as

Gaussian MSK (GMSK).

These modulation schemes fa11 into two general categories: linear modulation and

constant envelope modulation, depending on the envelope shape of the modulated

waveforms. In linear modulation schemes, such as QPSK and QAM, the abrupt phase

changes in the modulated waveform result in envelope variations if a filtcr limits the


Chapter 1: introduction 4 - - -

bandwidth. Such variations in turn require a linear power amplifier to avoid spectral

degradation. On the other hand, constant envelope modulation signals, such as FSK and

GMSK signals, c m be processed by high efficiency noniinear power arnplifiers. Table 1.1

lists some of the characteristics of several wireless standards such as Digital European

Cordless Telephone (DECT), Personal Handyphone System (PHs), Personal

Communications Services at l9OOMHz (PCS 1900) and Universal Digital Portable

Communications (UDPC). The power amplifiers to be studied in this thesis target non-

linear power arnplifiers used in conjunction with constant envelope modulation schemes.

Table 1.1 : Charactenstics of Wireless Standards [3,4]

1 Standard II DECT 1 P H s 1 PCS1900 1 UDPC 1

1.2 RF Power Amplifiers

Modulation

Envelope

Power Amp

Power Amplifier Metrics

The most commonly used metric to characterize the efficiency of a power amplifier is

the Power Added Eficiency (PAE), which is defined as

FSK

constant

non-linear

PAE = POUT -Pm p~~

where Pour is the RF power delivered to the load, PIN is the available input power and PDc

is the total power taken fiom the DC supply.

d4-QPSK

variable

linear

Another metric of efficiency is the drain (or collector) efficiency given by

- - - -


GMSK

constant

nonlinear

d4-QPSK

variable

linear

Chapter 1: Introduction 5 - - -- -

Linearity is another concem in power arnplifier design. When the input power is small,

the gain (POUdPIlv) of a power arnplifier is almost constant. When the input power

increases, the output power is a compressive function of the input due to the nonlinearity of

the amplifier, that is, the gain approaches zero for sufficiently high input levels. The

nonlinearity of a power arnplifier can be characterized by the "1-dB compression point",

defined as the input signal power level that causes the smail-signal gain to drop by 1 dB.

Conventional Power Amplifiers (PAS)

Power amplifiers have been traditionally categorized as Class A, B, AB, C, D, E, F and

S 151. They cm be classified into three families:

Unsaturated PA: The output power is a function of the input power. This family

includes Class A, B, AB, C, and F. The power transistor in these PAS operates as a

current source. Class A, AB, and B PAS may be used as linear PAS, whereas Class C

power amplifiers are more nonlinear in nature.

Saturated PA (or switching-mode PA): This family includes Class D, E, and S. In

these classes, the power transistor operates as a switch. The transistor "on" voltage

is usually as close to zero as possible.

Mixed-mode PA: This family includes Class AB, B or C where the power transistors

are over driven into gain compression at fûll output to improve efficiency. The

transistor saturates during part of the "on" portion of the RF cycle, acting as a

switch; for the rest period of the "on" portion, the transistor operates as a current

source.

Fig. 1.2 shows a generic schematic of Class A, B, AB, or C power arnplifier using a

MOSFET and a tuned load. The primary distinction between these classes is the gate bias

voltage of the transistor that determines the fraction (conduction angle a) of the RF cycle

for which the transistor conducts. For Class A power amplifiers, the transistor is on for the



entire cycle (a=2n, device biased far above threshold VI), whereas it is on for half the cycle

for Class B PAS (a-, device biased ai threshold), is on for greater than half the cycle for

Class AB PAS (a-, device biased slighily above threshold), and is on for less than half the

cycle for Class C PAS (a


1.5 Po

Po(C1ass A) - -

1 .& +-- RF Power -

- - -

0.5 0% 27~ x Conduction angle

Class A AB B C

Fig. 1.3: RF power and drain efficiency as a function of conduction angle. (optimum load, harmonic short and zero-Vsat assumed)

classical PAS such that the voltage or current waveform of the transistor is clipped

significantly at both ends [SI. Although undesirable distortion is introduced, there can be

usefiil trade-off between eficiency enhancement and linearity that can be utilized in low or

intermediate envelope amplitude applications. Appropriate harmonic termination at the

output can also help enhancing the performance. An example involves replacing the

sinewave voltage at the device output with a flatter, square-like periodic waveform. This

can result in important benefits in both power and efficiency, both of which can be traded

effectively for linearity. This method results in Class F power amplifiers [ 5 ] and other

subtypes [6]. An implementation of Class F is show in Fig. 1.4, where the basic structure is

that of a Class B power amplifier but with a quarterwave SCSS (short-circuit shunt stub)

used as an even harmonic trap. A maximum esciency of 88.4% can be achieved for such a

power amplifier using an ideal transistor.

In Class D, E, and S, the transistor acts as a switch. An ideal switch has either zero

voltage across it or zero current through it, thus it consumes no power. Therefore switching



RF Shon Vdc

l ' t l

Even Harmonic Trap W4) (SCSS) Series LC @Io

Input Match

RL

I

Fig. 1.4: Class F power amplifier with even harmonic trap

type power amplifiers have ideally 100940 eff~ciency. Class D power amplifiers need a high-

side device which makes them unsuitable for RF applications, both in terms of parasitic

reactances and drive requirements. Class E power amplifiers are attracting attention in RF

communications applications [7,8]. Theoretical efficiency of around 90% can be achieved,

but with an RF output power lower than that obtainable from the same device in a

conventional Class AB power amplifier, and with a peak voltage as high as 3.6 times the

DC supply. Class S power amplifiers are similar to a pulse-width modulator with a low pass

filter at its output and are not suitable for high frequency operation.

Multiharmonic Tuning of Power Amplifiers

The conventional power amplifier theones are generally based on waveforrn analysis

for ideal power devices. However, real devices exhibit parasitics such as finite on-state

resistance and output capacitance. Conventional PA theories cannot be used to analysis the

effects of these parasitics on the overall PA performance, thus preventing improvements in

PA performance.



Multiharmonic tuning techniques, combined with harmonic balance simulations, have

recently been proposed for GaAs MESFET and HEMT power amplifiers to provide M e r

insights in improving power amplifier performance[9-141. The key idea is to control the

shape and overlap of the drain voltage ancilor current waveforms using appropriate muhi-

harmonic terminations. Two kinds of multihannonic tuning behavior were reported:

The first one is the widely known Class F. The drain voltage waveform is square-

like. Staudinger et al. [9,10] studied the multiharmonic tuning effect on a set of

GaAs MESFET power amplifiers. Fig. 1.5 shows the output power and efficiency

contours versus the phases of TL at 200 and 3a0. The variations of eficiency and

Po,, are insensitive to the phase of TL(3mn); in another words, the results exhibit a

strong dependence of efficiency and output power on the second harmonic

termination and a weaker dependence on the third and higher hannonic

terminations. These results suggest that significant improvements in linearity,

eficiency and output power c m be achieved with proper harmonic terminations.

Specifically, the best performance can be obtained when the 2nd harmonic is SHORT

and the 3rd harmonic is OPEN (Class F operation).

ma(r,@ - D.B AM@@ =O) - ~.t( (a) (b)

Fig. 1.5: Power and Eniciency contours vs. phases of load reflection coefficients (rL) of the MESFET PA [IO]



The second type is the opposite to the first one, Le., the 2nd harmonic is OPEN and

the 3rd harmonic is SHORT. Or in generai, the drain current waveform is square-

like, containing only odd harmonics, and the drain voltage waveform containing

only even harmonics, as shown in Fig. 1.6. This type is referred as Inverse Class F.

Drain Voltage

Fig. 1.6: Inverse Class F waveforms

This mode was first reported in power amplifiers using GaAs MESFET or HEMT

power devices [11,12]. A more complete analysis of this type of behavior was

performed recently [13,14]. The eficiency of power amplifiers in this mode can be

expectcd to be higher than Class F. However, the peak drain voltage swing is larger

than that in Class F and hence power devices with higher breakdown voltages are

required.

Muitiharmonic Tuning Behavior of MOSFET RF Power Amplifiers University of Toronto


Unresolved Issues

Even though the multiharmonic tuning technique has been proposed and successfully

used in power amplifier design, there are still some unresolved issues. First, there is no

complete classification of multiharmonic tuning behavior. Second, the relationship between

conventional power amplifier modes (Class AB, E, F, etc.) and multiharmonic w i n g modes

is vague. Finally, there is no systematic procedure to find the best mode and corresponding

optimal multiharmonic impedances for power amplifier design. These issues restrict the

practical design of power amplifiers using multiharmonic tuning techniques.

1.3 Objectives and Outline of the Thesis

As the channel length of a conventional MOS device is reduced into the deep

submicron range, its high frequency performance improves and it becomes attractive for

low-cost integrated implementations of RF power ampli fiers [16- 191.

The objective of this thesis is to study the multihmonic tuning behavior of RF power

amplifiers using power MOSFETs implemented in conventional bulk silicon CMOS

technologies. The MOSFET power amplifiers are targeted to operate at 1.88GHz fiom a 2V

supply with an output power of 2OOmW suitable for wireless applications.

In Chapter 2 the multiharmonic tuning behavior is analyzed and classified into four

basic modes and conventional power amplifier modes are characterized using this

classification. A systematic multiharmonic tuning optimization procedure is proposed to

find the best mode and corresponding optimal harmonic terminations. The multiharmonic

tuning behavior of bulk silicon MOSFET RF power amplifiers is then studied by

simulation. The power devices used are implemented in a 0.25pm CMOS process. Device

characterization and measured power amplifier performance are then presented in Chapter 3

[20]. Finally, Chapter 4 summarizes the results and outlines fbture work.



References

V. K. Garg and J. E. Wilkes, "Wireless and Persona1 Communications Systems", Pren- tice-Hall, New Jersey, 1996.

B. Razavi, "RF Microelectronics", Prentice-Hall, New Jersey, 1998.

R. Pandya, "Mobile and Persona1 Communication Services and Systems", IEEE Press, New York, 2000.

L. E. Larson, "RF and Microwave Circuit Design for Wireless Communications", Artech House, Boston, 1996.

S. C. Cripps, "RF Power Amplifiers for Wireless Communications", Artech House, Boston, 1999.

B. Ingruber, W. Pritzl, D. Smely, M-Wachutka and G Magerl, "High-Eficiency Har- monic-Control Amplifier", IEEE Tram. Microwave Theory and Techniques, Vo1.46, pp.857-862, 1998.

N. O. Sokal, "Class-E switching-mode high-eficiency tuned RFImicrowave power amplifier improved design equations," IEEE MTT-S Int. Microwave Symp. Dig., vol. 2, pp. 779-782,2000.

T. Sowlati, C.A.T. Salama, J. Sitch, G. Rabjohn, and D. Smith, "Low Voltage, High Eficiency GaAs Class E Power Amplifiers for Wireless Transmitters", IEEE Journal of Solid-State Circuits, Vo1.30, pp. 1074- 1085, 1995.

J. Staudinger, "Multiharmonic load termination effects on GaAs MESFET power amplifiers," Microwave Journal, pp.60-77, April 1996.

1101 J. Staudinger and G Norris, "The effect of harmonic load terminations on RF power amplifier linearity for sinusoidal and pi/4 DQPSK stimuli," IEEE MTT-S Tech. for Wireless Applications Dig., pp.23-28, 1997.

11 13 J. C. Pedro, L. R. Gomes and N. B. Carvalho, "Design techniques for highly efficient Class-F amplifiers driven by low voltage supplies," IEEE Int. Cod. on Electronics, Circuits and System, vol. 3, pp.201-204, 1998.

[12] Y. Tkachenko, A. Klimashov, C. J. Wei, Y. Zhao and D. Bartle, "E-PHEMT for single supply, no drain switch and high eficiency cellular telephone power amplifiers," GaAs IC Symp. Proc., pp. 127-1 30, 1999.

[13] C. J. Wei, P. DiCarlo, Y. A. Tkachenko, R. McMorrow and D. Bartle, "Analysis and experimental waveform study on Inverse Class-F mode of microwave power FETs," IEEE MTT-S Int. Microwave Syrnp. Dig., vol. 1, pp. 525-528,2000.

[14] A. houe, T. Heirna, A. Ohta, R. Hattori and Y. Mitsui, "Analysis of Class-F and Inverse Class-F mode amplifiers," IEEE MTT-S Int. Microwave Symp. Dig., vol. 2, pp. 775-778,2000.

-- - - - -

Multiharrnonic Tuning Behavior of MOSFET RF Power Amplifiers University of Toronto


[15] E. Morifuji, H.S. Momose, T. Ohguro, T. Yoshitomi, H. Kimijima, F. Matsuoka, M. Kinugawa, Y. Katsumata and H. Iwai, "Future Perspective and Scaling Down Roadmap for RF CMOS," IEEE Symposium on VLSI Technology, Proceedings, pp. 163 - 164, 1999.

1161 G Hefiman, "GaAs, MOS, Bipolar Vie For Power Applications," Microwave & RF, pp.3 1-39, July 1998.

[17] D. Su and W. McFarland, "A 2.5-V 1-W Monolithic CMOS RF Power Amplifier," IEEE Custom Integrated Circuits Conference, Proceedings, pp. 1 89-1 92, 1997.

[18] T. Ohguro, M. Saito, E. Morifuji, K. Murakami, K. Matsuzaki, T. Yoshitomi, T. Morim- oto, H.S. Momose, Y. Katsumata and H. Iwai, "High eficiency 2 GHz power Si-MOS- FET design under low supply voltage down to IV," IEEE Int. Electron Devices Meeting, Technical Digest, pp.83-86, 1996.

[19] 1. Yoshida, M. Katsueda, Y. Maruyarna, and 1. Kohjiro, "A Highly Escient 1.9-GHz Si High-Power MOS Amplifier," IEEE Trans. Electron Devices, Vo1.45, pp.953-956, 1998.

[20] Y.-C. Zhang and C. A. T. Salama, "Multiharmonic Tuning Behavior of MOSFET RF Power Amplifiers," IEEE MTT-S [nt. Microwave Symp., May 2001.


Chapter 2: Theoretical Multiharrnonic Tuning Behavior of MOSFET RF Power Amplifiers 14

CHAPTER 2

Theoretical Multiharmonic Tuning Behavior of MOSFET RF Power Amplifiers

2.1 Introduction

In this chapter the conventional power amplifier modes and multiharmonic tuning

modes are analyzed. Multiharmonic tuning is then classified into four basic modes and

conventional power arnplifiers are characterized using this classification. A systematic

multiharmonic tuning optimization procedure is proposed to find the optimal fundamental

and harrnonic terminations. The multiharmonic tuning behavior of MOSFET RF power

arnplifiers is then studied using the proposed multiharmonic tuning optimization procedure.

2.2 Analysis of Power Amplifier Modes

2.2.1 Conventional Power Amplifier Modes

In this section each power amplifier mode is first analyzed in a traditionai way and then

analyzed fiom a multiharmonic tuning point of view and the requirements of load

impedance at the harmonic fiequencies are given.

Class A, B, AB and C

Class A, B, AB and C power amplifiers are the conventional power arnplifiers. In these

PAS the drain voltage waveform is assumed to be sinusoidal. The drain current can be


Chapter 2: Theoretical Multiharmonic Tuning Behavior of MOSFET RF Power Amplifiers 15

modeled as a complete or truncated sinewave of amplitude Im and with a DC component Iq

as shown in Fig. 2.1.

Fig. 2.1 : Wavefonns of Class A/B/C (The value of a de fines the class of operation)

In general, the drain current I (8 ) can be expressed as

where a ( O 5 a < 2n ) is the conduction angle (as defined on page 5). The Fourier series

expansion o f this waveform is given by

where

is the DC component and


C hapter 2: Theoretical Multiharmonic Tuning Behavior of MOSFET RF Pow er Amplifiers 16 --

where i l is the fundamental fiequency current and in (n>l) is the n-th order harmonic

component.

The DC component of the drain current can be used to calculate the average power

drawn fiom the power supply. With an appropnately designed load network, only the

fiindamental component of the drain current will be fed to the load resistor.

In order to obtain the maximum voltage swing at the drain, an optimal load resistance

RL must be present at the drain. For a given a, RL is given by

where Vdd is the drain supply voltage, V, is the MOSFET saturation (knee) voltage and

iI(a) is the amplitude of the fundamental fiequency current for the specified a.

The output power is

Given the above, the drain efficiency is

Vdd - Vsat , a - sina '1 =

V d d +in a - a cos 7 a>

Thus ideal Class AB power amplifiers have drain efficiency ranging fiom 50% to 78% and

ideal Class C power ampli fiers have an efficiency ranging from 78% to 100%.

As stated previously, the classical PA modes assume a sinusoidal drain voltage

waveform at the fundamental fiequency. The drain voltage V(0) can be expressed as



where ZL is the load impedance, ZLO is the DC load resistance, ZLi is the load

impedance at the fundamental fiequency, and ZLn ( n N ) is the load impedance at the n-th

order harrnonic fiequency. The load impedance at al1 harmonic fiequencies (ZLn (n>l))

MUST be ZERO (short circuit) in order to produce the required sinusoidal waveform.

Class F

The sinusoidal drain voltage wavefom assurnption in the above PA modes restricts the

efficiency and output power. If the drain voltage waveform is shaped to reduce the overlap

of the voltage and current waveforms, a significant improvement in efficiency and power

can be achieved [ 1 1.

In the Class F mode, improvements of both efficiency and output power are

accomplished by using odd harmonics to cause the drain voltage waveform to approximate

a square waveform and using even harmonics to cause the drain current to approximate a

half sinusoid.

Drain Voltage

Fig. 2.2: Waveforms of ideal Class F mode

The ideal wavefoms are depicted in Fig. 2.2. The total peak-to-peak drain voltage is

twice the supply voltage Vdd. The amplitude of the fundamental component is given by

Fourier analysis as



The load network is designed so that only the fundamental fiequency power can be

delivered to the load. Thus the output power is given by

where RL is the fundamental load resistance presented at the drain. Note that the value of Vi

exceeds Vdd which indicates higher output power can be delivered to the sarne fundamental

load RL compared to Class B mode. The eficiency of the ideal Class F mode is 100%.

The presence of harmonics in the voltage/current waveforms requires the correct load

impedances at the harmonic fiequencies. For ideal Class F mode, since only odd harmonics

exist in the voltage waveform and only even harmonics exist in the current waveform, the

load impedance should be M F N T E (open circuit) at ODD harmonics and be ZERO (short

circuit) at EVEN harmonics.

It is practically diKicult to realize the waveforms of ideal Class F mode. Other

alternatives of Class F mode utilize only the third harmonic frequency (and the fifih

harmonic frequency) to maximally flatten the drain voltage waveform to approximate a

square waveform [Il]. They require INFMTE load impedance at the THlRD (and FIFTH)

harmonics and ZERO load impedance at EVEN harmonics. The eficiency ranges from 88.4%

(for using only the third harmonic) to 92% (for using both the third and fifth harrnonics).

Class E

Power amplifier eficiency is maximized by minimizing power dissipation, while

providing a desired output power. The largest power dissipation occurs usually in the RF

power transistor and is determined by the average integration of the product of transistor

Multiharmonic Tuning Behavior of MOSFET RF Power Amplifiers Universi@ of Toronto


voltage and transistor current over the RF period. This power dissipation can be minimized

by avoiding simultaneous occurrence of high voltage and high current. Fig. 2.3 shows the

conceptual transistor voltage and current waveforms needed to meet the high eficiency

requirements. For this case,

Drain Current

7 ; switch : a I I

I on state ; I I I Drain I I I I

1 I 1 Voltage I I I

1 I I I I I I I I ~ i m e D

0 x 2n 3x 4n

Fig. 2.3: Conceptual "target" waveforms of Class E

(a) The voltage across the switch (transistor) at the turn on time must be zero. (b) the

slope of the voltage across the switch (transistor) at the turn on time must be minimized,

and (c) the rise of the voltage across the switch at turn-off must be delayed till the current

drops to zero.

Thus the power delivered to the load RL equals the power provided by the power

supply source, achieving 100% efficiency. A Class E power amplifier containing a switch

and a load network meets the above critena [IO].

Fig. 2.4(a) shows the basic Class E circuit implementation (named "low-order Class

E") which consists of an ideal switch (transistor) shunted by a capacitor C I , a senes LC

circuit Lr-C2a resonating at the fundamental frequency fo, an additional reactance Ca to

adjust the phase of the voltage of capacitor C I , and the RF load resistor RL [9]. An RF choke

LI provides power supply to the switch. The wavefonns generated by such circuit

approximate the conceptual waveforms shown in Fig. 2.4(b). Note that these actual



waveforms meet al1 three criteria listed

is part of a sinusoid, the current through

'A' I C2a C2b

above. Even though the current through the switch

the load resistor is sinusoidal.

Switch Voltage

r Load Current . -

Fig. 2.4: (a) Schematic of basic (low-order) Class E power amplifier (b) Actual waveforrns

The design equations for the Class E amplifier obtained by time domain analysis are

surnrnarized below [9,1 O]

RFC with X usually > 1 OXc

Q$L RnfO

where fo is the fundamental frequency, QL (>1.7879) is the network loaded Q factor

(chosen by the designer as a trade-off), CI includes Co, of the switch transistor, C2 includes

in series with C2b.

The harmonic impedances do not play a direct part in the conventional time domain

analysis of Class E mode. However, the network topology itself provides the hannonic

terminations, which is analyzed below. In Fig. 2.4(a), if the impedance of the RF choke LI



is assumed to be infinite and Cl is treated as the output capacitance of the switch, the load

impedances ZL at reference plane A-A' is then given by

Substituting (2.15) into (2.16), ZL at the fündarnental frequency fo is

which is equivalent to a resistor RL in series with an inductor L less than L2,

And ZL at the harmonic frequencies nf, (m 1) is

zL( nfo) = j2nnf,L2 - \ 1 - 1 }+RL (2.18) n2[ 1 + 1.1 10 /(QL - 1.7879)]

which is equivalent to a resistor RL in series with an inductor L approximately equal to L,.

At radio frequencies, the absolute value of ZL(nfo) is very large and can be assurned to be

INFNTE (open circuit).

As a conclusion, the load irnpedances of Class E mode at al1 harmonic fiequencies are

INFINITE (open circuit).

2.2.2 Classification of Multiharmonic Tuning Behavior

Based on the analysis of conventional power amplifier mdoes and the characteristics of

the multiharmonic tuning power amplifiers, this section proposes a complete classification

of multiharmonic tuning behavior. As shown in Fig. 2.5, this behavior can be classified into

four basic types according to the phases of the harmonic load reflection coefficients rL (the magnitudes are assumed to be close to 1 .O). The four basic modes are:

SS mode: Soth odd and even hannonics are SHORT. The drain VOLTAGE waveform

is sinusoidal, containing no harmonic components. This type includes conventional

Class A, AB, B and C.

Multiharrnonic Tuning Behavior of MOSFET RF Power Amplifiers University of Toronto


180" 1) (short) .- OS mode SS mode (Class F) (Clé ss A, B,C)

00 mode SO mode (Class E)

f \ \ J

-1 80" 0" 180" (short) (open) (short)

Phase of TL at odd harmonics

Fie. 2.5: Classification of multiharmonic tuning behavior

OS mode: Odd harmonics are OPEN and even hannonics are SHORT. The drain

VOLTAGE waveform is square-like, containing only odd harmonic components. This

type corresponds to the Class F [2,3].

SO mode: Odd hannonics are SHORT and even harmonics are OPEN. The drain

CURRENT waveform is square-like, containing only odd harmonics. The peak

voltage is higher than twice the power supply. This type corresponds to the Inverse

Class F 16-71.

00 mode: Both odd and even harmonics are OPEN. The drain CURRENT waveforrn

is sinusoidal, containing no harmonics, resulting in little or no energy wasted at

harmonic frequencies and yielding high eficiency. The peak voltage is also higher

than twice the power supply. This type includes the low-order Class E [9]. This

mode has not been reported in previous work on multiharmonic tuning.

Compared to the low-order Class E, the 00 mode has a few advantages. First, it

does not require rectangular gate driving waveforms, easing the design of the



dnving stage; second, it exhibits lower peak drain voltage, relieving the voltage

stress on the power device; third, the power device can be biased at a conduction

angle other than 180°, enabling trade-offs between efficiency and output power.

However, the 00 mdoe may result in a Iower eficiency than the low-order Class E.

In Fig. 2.5 it is interesting to note that the OS mode (Inverse Class F) is the dual of the SO

mode (Class F), and the 00 mode (Class E) is the dual of the SS mode (Class A, B, C).

This classification reveals that there are other possible high efficiency multiharmonic

tuning modes besides the reported Class F and Inverse Class F. Furthemore, the

relationship between the conventional power amplifier modes and multiharmonic tuning

modes is also shown in Fig. 2.5'.

2.3 Multiharmonic h i n g Optimization Procedure

Traditionally, the load-pull measurements have been used to find the optimal load

terminations (at fundamental and harrnonic fiequencies) for power amplifier design. But the

experiments are time consuming. The proposed simulation procedure [8] c m help find the

optimal load impedances. However, this procedure does not consider the effect of harmonic

teminations on the optimal fundamental load impedance.

This section describes a systematic optimization procedure to find the optimal load

tenninations. Fig. 2.6 shows the basic circuit used in harmonic balance (HB) simulations

using an HB simulator such as Agilent Technologies' ADS [16]. On the load side, only the

2nd and 3rd harmonics are considered and the magnitudes of their reflection coefficients are

set to be 0.99. Because higher order harmonics have smaller amplitudes and hence do not

have a significant effect, they can be ignored by terminating them to 5 M l On the source

side, al1 harmonics are also terminated to 5CKZ The optimization takes into account the

1. The SS mode can be further classified into Clam A, AB, B or C, according to the conduction angle. Furthermore, all multiharmonic tuning modes can be biased at a conduction angle between O" and 360".


Chapter 2: Theoretical Multiharmonic Tuning Behavior of MOSFET RF Power Ampliaers 24

mutual effects of the harmonic terminations on the choice of the optimal fundamental load

impedance.

Fig. 2.6: Basic circuit used in MHT optimization

DUT

The procedure used to find the optimal harmonic terminations is as follows:

Tuner 2

Estimate the load resistance RL at the fundarnental frequency (oo) according to the

available power supply Vdd and the required output power Pour:

where V,, is the saturation voltage, usually estimated to be O. 1 VdK0.2 Vdd.

Then determine the fundamental source impedance Z'(on) for good input match.

I w r

For each multiharmonic tuning mode, set the phases of rL(20d and rL(3a/9,

perforrn load-pull simulations to find the optimal fundarnental load impedance

ZLo(mo) Retuning Zs(oo) may be necessary.

2s

For each ZLo(wo) obtained in Step 2, simultaneously sweep the phases of TL(2qJ

and rL(3wd. Compare the results and chose the best mode and corresponding

load impedance.

Z-param ZL block

TL

- -

-- -


-

Chapter 2: Theoreticai Multiharmonic Tuning Behavior of MOSFET RF Power Amplifiers 25

2.4 Multiharmonic Tuning Behavior of MOSFET PAS

As previously mentioned, Class F (OS mode) or Inverse Class F (SO mode) have been

found to achieve the highest efficiency for GaAs MESFET or HEMT power amplifiers[2-

71. An important difference between GaAs devices and silicon MOSFETs for power

amplifier applications is that GaAs devices have smaller drain-source capacitances (Cd*)

because of the semi-insulating substrate. For devices with large drain-source capacitance,

such as power MOSFETs implemented on bulk silicon substrate, the question is which

multiharmonic tuning mode yields the highest eficiency.

In order to investigate this issue, power MOSFETs implemented in a conventional

0.25pm bulk silicon CMOS technology were used. The technology features single poly,

five metal layers and a minimum drawn channel length of 0.25pm7 suitable for RF and high

speed mixed signal circuits.

2.4.1 Device Design

Device Sizing

Determining the size of the power MOSFET is an iterative process where the initial

guess originated from the 1-V characteristic of the power transistor and is based on the

traditional load line theory [ I l .

The first step is to calculate the maximum drain current according to the power supply.

The drain supply voltage Vdd was chosen to be 2V for low voltage applications. (The

maximum Vdd is limited by the drain-gate oxide breakdown voltage BVgd of the MOSFET.

BVgd for the 0.25pm CMOS technology was estimated to be above SV) To obtain an output

power of 250mW, the optimum load resistance ROpl was firstly estimated to be



And the peak current through the load was then given by

According to the load line theory, the maximum drain current Idm is about twice I& i.e.,

Idm = 2 Ipk = 580mA (2.22)

Ih occurs when the gate voltage reaches its peak value VP. The maximum amplitude of

the driving (gate) signal was around Vdd and thus the peak gate voltage was given by

Vg,, Va + Vdd = 2.7V (2.23)

where Vgg is the gate biasing voltage (which equals 0.7V as described in Section 2.4.2).

The next step is to obtain an initiai guess of the transistor size. Fig. 2.7 shows the drain

current for 0.25pm N-MOSFETs with different gate width at a gate voltage of 2.7V. It can

be seen that a gate width of 900pm can provide 600mA drain current. Therefore the initial

guess of the transistor size was obtained as

Gate Width (urn)

Fig. 2.7: Drain current for different gate width (gate length=0.25pm)



Once the initial size was seîected the more accurate RF behavior of the MOSFET was

then found by load-pull simulations which indicated that the initial size is not sufficient to

obtain the required output power. Afier a few steps of iteration, the device size was finally

chosen to be

W / ' = 2 0 0 0 ~ m f l . 2 5 ~ m

Device Modeling

The BSIM3V3 device model was used in simulations. It was reported that BSIM3V3

model is basically adequate as a nonlinear physical large signal model for power amplifier

design [12,13]. However, to improve simulation accuracy, the BSIM3V3 model was further

enhanced to take parasitics into account, as shown in Fig. 2.8 and described below.

Fig. 2.8: Modified BSIM3V3 CMOS device model

The gate resistance and substrate resistances were added to the model as shown in

Fig. 2.8 [14]. The gate resistance Rg was calculated to be 0.34Q from the

interdigitated transistor layout and the resistivity of the gate material. The substrate

resistances Kubd was estimated to be 3.5R frorn simulations [14,15]. Cjd and Cjs are the drain and source junction capacitance, respectively, and were set to be the

same corresponding values in the BSIM3V3 model.

Multiharmonic huiing Behavior of MOSFET RF Power Amplifiers University of Toronto


The layout parasitics such as pad capacitances and in te rco~ec t overlap

capacitances were added in p s t layout simulations.

2.4.2 MHT Optimization for P A .

Before MHT optimization, DC and S-parameter simulations were performed to obtain

some key device parameters as follows: W/L=200ûpm/0.25pm, Vth=O.6V, BVd,=7.0Vy

%,=0.55*, gm=330mS/mrn, Cd,=2.8pFy f ~ 3 1 GHz, fm,=28GHz.

The final circuit schematic used in multiharmonic tuning optimization is shown in Fig.

2.9. The device was biased at Vdd=2.0V and Vgg=0.7V through RF chokes. The RF choke

parasitic resistances (032) were measured and included in the schematic. The DC blocks

separate the RF input/output signals to DC supplies. The probe-pad contact resistance Rpp

(OSSI) (especially the contact resistance at the drain terminal) in the on-wafer load-pull test

setup was also included in the schematic. This value is small but must be included to match

the simulation and experimental results.

DC block

- -

Fig. 2.9: Schematic used in A D S simulator

2-param Tuner 2

model

Multiharmonic Tunïng Behavior of MOSFET RF Power Amplifiers University of Toronto


Using the MHT optimization procedure described in Section 2.3, the fhdamental load

resistance RL to obtain over 200mW output power at 1.88GHz was estirnated to be 6 0 and

Zs(od was determined to be 6+j14.8R The optimal fundamental impedances obtained

from load-pull simulations for each harrnonic tuning mode are shown in Fig. 2.10, Fig. 2.1 1,

Fig. 2.12, Fig. 2.13 and compared in Table 2.1. The highest eficiency was achieved in the

00 mode where ZL(od=5.6+j7.8Q Since the corresponding output power was only

22.4dBm, further trade-off between output power and efficiency must be made to achieve

high eficiency and the required power simultaneously. By inspecting the PAE and power

contours shown in Fig. 2.13, the optimal ZL(Wd was finally chosen to be 6.5+j4.OR

Table 2.1 : Optimal ZL'd of the MOSFET optimized for maximum PAE (Pi,=12dBm)

Fig. 2.14 shows the power, gain and eficiency variation as a function of the input

power for the various modes considered. At low gain compression, the 00 mode and OS

mode (Class F) exhibit higher efficiency than the SO mode (Inverse Class F) while at higher

gain compression, the 00 mode achieves the highest eficiency of 64% with an output

power of 23.6dBm (230rnW) at an input power of 12dBm.

OS mode

SO mode

00 mode

Fig. 2.15 displays the drain voltage and current wavefonns of the 00 mode at

Pi,=12dBm. The peak drain voltage is 4.4V, greater than 2*V& (4V); the drain current

wavefonn is sinusoidal, implying little or no energy wasted at hannonic frequencies.

Fig. 2.16 shows the PAE and Peut contours versus the phases of TL at 2a0 and 3a0 for

Pi,=12dBm. The variations of PAE and P, are insensitive to the phase of TL(3ao)

59.1

68.1

68.9

Multihamonic Tuning Behavior of MOSFET RF Power Amplifiers University of Toronto

6.3+j3.7

5.8+j7.7

5.6-ej7.8

23.1

22.2

22.4

Chapter 2: Theoretical Multihannonic Tuning Behavior of MOSFET RF Power Amplifiers 30

Fig. 2.10: PAE and Po,, contours for SS mode (Pi,= 12dBm)

Fig. 2.1 1 : PAE and Po,, contours for OS mode (Pi,= 12dBm)



Fig. 2.12: PAE and Po,, contours for SO mode (Pin=l 2dBm)

Fig. 2.13: PAE and Po,, contours for 00 mode (Pin=12dBm)

- --



7 0 ~ 1 - 00 mode 1 I 4 I

-5 O 5 10 15

Pin (dBm)

Fig. 2.14: PAE, Pout and Gain vs. Pin of the MOSFET

Fig. 2.15: Drain voltage and current waveforrns in the 00 mode

Furthemore, In Fig. 2.16(a) the white region is the region of the highest efficiency and is

somewhat independent of the phases of rL(2mO) and ïL(300), enabling easy

implementation of the load network.

The above simulation results demonstrate that the 00 mode yields the highest

efficiency at high gain compression for bulk silicon MOSFET RF power amplifiers.

Multiharmonic Tuning Behavior of MOSFET RF' Power Amplifiers University of Toronto


PAE scale :%)

51-53 53-55 57-59

EZI 59-61 O 61-63 O 63-65 7

-150 -100 -50 O 50 100 150

Phase of ïL(3a0)

(a) PAE contours

Phase of rL(3m0)

(b) POM contours

Fig. 2.16: PAE and Po,, contours vs. phases of ïL of the MOSFET power amplifier (ZL(wd=6.5+j4.m Pin=l 2dBm)

Multiharmonic Tuning Behavior of MOSFET RF Power Amplifiers UniversiQ of Toronto


It is worthy to compare previous results to the multiharmonic tuning behavior of GaAs

MESFET or HEMT power amplifiers[2-71. As mentioned before, an important difference

between GaAs devices and silicon MOSFETs for power amplifier applications is that GaAs

devices have smaller drain-source capacitances (CdJ because of the semi-insulating

substrate. The cornparison suggests that Cds may play an important role for the 00 mode to

achieve higher efficiency than other modes.

The conventional power amplifier modes and multiharmonic tuning modes were

analyzed in this chapter. A complete classification of multiharmonic tuning behavior was

proposed. Multiharrnonic tuning was classified into four basic modes and conventional

power amplifier modes can also be characterized using this classification. A systematic

multiharmonic tuning optirnization procedure was proposed to find the optimal harmonic

tenninations. The multiharmonic tuning behavior of bulk silicon MOSFET RF power

amplifiers was studied. The results showed that the 00 mode yields the highest efficiency

at high gain compression for the bulk silicon MOSFET RF power amplifiers. Table 2.2

sumrnarizes the simulated characteristics of the power device and power amplifier.



Table 2.2: Simulated characteristics of the power device and power amplifier

-

Operating Voltage (V) 11 2.0

Device Chawteristics

Size WiL (pxdpm)

Threshold Voltage (V)

Vduës

2000/0.25

Ron at Vgs=2.5V (Cl) (1 0.55

Best mode for highest PAE 11 O 0 mode Peak PAE (%) 11 64 Corresponding Po,, (am) 11 23.6 Corresponding Pi, (dBm) II 12


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