Research Article Illustration of the Impedance Behaviour ...downloads.hindawi.com/journals/ijap/2014/712790.pdf · indeed, worse properties in the UHF frequency band. u s, AMC surfaces

Research ArticleIllustration of the Impedance Behaviour of ExtremelyLow-Profile Coupled Shorted-Patches Antennas for UHF RFIDof People

Milan Svanda and Milan Polivka

Department of Electromagnetic Field, Faculty of Electrical Engineering, Czech Technical University in Prague,Technicka 2, 166 27 Prague 6, Czech Republic

Correspondence should be addressed to Milan Svanda; [email protected]

Received 4 July 2013; Revised 30 April 2014; Accepted 2 May 2014; Published 28 May 2014

Academic Editor: Francisco Falcone

Copyright © 2014 M. Svanda and M. Polivka. This is an open access article distributed under the Creative Commons AttributionLicense, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properlycited.

The recently introduced coupled shorted-patches technique for the design of extremely low-profile UHF RFID tag antennas is usedto illustrate the flexibility of selected feeding methods for tuning the antenna input impedance for the complex values required formatching with typical RFID chips.We present parametric studies of the impedance behaviour of dipole-excited and directly excitedantennas designed for radiofrequency identification of people in the European UHF frequency band. Our study can significantlyfacilitate the design of this class of on-body tag antennas.

1. Introduction

Small, low-profile, lightweight, and cheap: these are themost frequently used words in connection with antennas forwearable applications. If we add to this sufficient radiationefficiency and immunity of the electrical parameter againstthe presence of a human body, we have a specification of thebasic requirements for wearable antennas. Unfortunately, anumber of these demands conflict with each other. Miniatur-ization of radiator size is limited by fundamental laws [1–3],while frequency bandwidth and radiation efficiency declinewith size reduction.

Immunity of the wearable antenna against the influenceof the human body is often dealt with by inserting a thinmetallic plane that provides additional screening for a singleradiator (e.g., a dipole antenna), or it forms an inherentpart of the antenna (e.g., patch-type antennas) [4]. In bothcases, however, this type of solution significantly affects theelectrical properties of the antenna, as will be discussedbelow. Another requirement for a wearable antenna for useas a UHF RFID tag antenna is that the inductive inputimpedance complex conjugate to the chip impedance must

have a strong capacitive component [5]. A special radiatorarrangement, including various more or less sophisticatedfeeding techniques, must therefore be used.

The coupled shorted-patches technique that has beenpresented recently in [6–9] enables the design of extremelylow-profile antennas (typically below 0.01𝜆

0). These have

much better immunity from the influence of a human bodysituated in the close vicinity than for the same substrateheight standard half- and quarter-wavelength microstrippatch antennas. At the same time, the radiation efficiency ofthe structure is satisfactory—typically better than 50%. Thetwo ways of excitation also enable the input impedance to betuned for complex values, which is an essential condition forfeeding the antenna in the case of RFID chips.

This paper begins by providing a brief summary of theimpedance properties and the radiation efficiency of low-profile wire and patch antennas operating closely spacedabove or inherently above metallic planes. The benefits anddrawbacks of wire- and patch-type antennas are outlined anddiscussed.

Further, we illustrate the impedance flexibility of cou-pled shorted patches for achieving the required complex

Hindawi Publishing CorporationInternational Journal of Antennas and PropagationVolume 2014, Article ID 712790, 10 pageshttp://dx.doi.org/10.1155/2014/712790

2 International Journal of Antennas and Propagation

value using parametric changes of the antenna, excitationdipole, and/or tuning slot geometry. This enables the inputimpedance to be set for typical complex values of UHF RFIDchips, that is, approximately 5Ω to 50Ω for the real part and100Ω to 200Ω for the imaginary part.

2. Benefits and Drawbacks of Wire- andPatch-Type Antennas Closely Spaced aboveMetallic Planes

2.1. Wire-Type Antennas. A dipole-type antenna closelyspaced above a perfect electric conductor (PEC) may betreated as dipole pair with opposite currents using the mirrorprinciple. As mentioned above, the immunity against nega-tive objects is significantly improved if the primary radiatoris placed at a relative distance of approximately 𝜆

𝑔/4 above

the PEC plane. In this case, the direct and reflected wavesinterfere constructively, and in an ideal case the antenna gainmay be improved on as far as 3 dB.

However, destructive interferences appear if the distanceof radiator ℎ from the plane is a very small fraction ofthe wavelength. The image currents (𝐼

2) exhibit a direction

opposite to the direction of the primary dipole currents (𝐼1).

The dipole pair input impedance 𝑍in is then expressed as𝑍in = 𝑍11 +𝑍12𝐼1/𝐼2, where, for a close metallic plane mirror,the current is 𝐼

2≈ 𝐼1. 𝑍11

stands for the self-impedance ofthe single dipole, while𝑍

12represents the mutual impedance

of the dipole and its image [10]. For ℎ/𝜆 → 0 : 𝑍11∼

𝑍12, therefore 𝑍in → 0; consequently the dominant part

of the input resistance, radiation resistance 𝑅rad → 0.As a result, both the radiation and the antenna efficienciesdecrease significantly. A detailed analysis can be found in [4].

It is simple to apply a distance dielectric pad between thewire radiator (e.g., a planar dipole) and the metallic plane,but this is not a sufficient improvement of the problem withthe decrease in radiation efficiency, which was discussedabove. Unfortunately, the pad needs to be relatively thick(0.01–0.1𝜆

0); see [11–13].

Unlike when a perfect electric conductor (PEC) is used,a perfect magnetic conductor (PMC) reflects the electromag-netic wave without a phase shift; that is, the image current(𝐼2) exhibits a direction that is the same as the direction of the

primary dipole currents (𝐼1). This phenomenon theoretically

enables us to place the primary radiator at an arbitrarydistance from the shielding PMC plane, without a rise in thedestructive interferences.

Artificial magnetic conductors (AMC) or highimpedance surfaces (HIS), usually based on mushroomstructures as a particular realization of PMC planes, aresometimes considered for use for low-profile antennas.However, narrow frequency bandwidth and complexity ofthe structure are a major limitation for the use of AMCsurfaces [14–18] in the UHF band.

These limitations can be explained on the basis of theparallel resonant circuit model of the cell of a mushroomstructure with the resonant frequency, which can be reducedby a rise in the capacity or the inductance values. A rise incapacity can be achieved either by reducing the gap between

0

0.06 0.080.040.02 0.1

Radi

atio

n effi

cien

cy (%

)

h/𝜆0

0

20

40

60

80

100

𝜀r = 1

𝜀r = 3

𝜀r = 10

Figure 1: Radiation efficiency of a rectangular patch antenna as afunction of the relative substrate height; see [27].

the metallic patches or by increasing the patch size. Anotherway is to increase the shunt inductance via an increase in sub-strate height. Increasing the capacity has a negative influenceon the bandwidth, according to the relation. Increasing theinductance, on the other hand, sets high requirements on thecomplexity of the structure. Unfortunately, due to the neces-sity to manufacture shunt inductive vias several millimetersin length in theUHFband [19], their implementation, to date,does not allow us to construct antennas with sufficiently lowprofiles (ℎ/𝜆

0< 0.01) [4]. A number of modifications and

a property analysis of AMC surfaces have been published,for example, [15, 16, 20–23]. However, they have similar or,indeed, worse properties in the UHF frequency band. Thus,AMC surfaces can be used successfully for improving theperformance of the antennas in the high rf and microwavefrequency bands, above approximately 2GHz.

2.2. Patch and PIFA Antennas. The application of antennasbased on a metallic ground plane, that is, patch and PIFAantennas [24–26], which can prevent undesirable effects of anearby object in the proximity of the antenna, is a differentapproach to these problems. However, at relatively lowoperational frequencies (hundreds ofMHz), several potentialdifficulties must be taken seriously into account. First, whenthe substrate is less than approximately 0.01𝜆

0in height,

and the relative permittivity 𝜀𝑟is higher than that of the

air or foam substrate, see Figure 1, their radiation efficiencydecreases significantly [27]. Second, the basic patch resonantfrequency corresponds to 𝜆

𝑔/2 or 𝜆

𝑔/4 and, therefore, at

UHF frequencies, patch or PIFA antennas may not be smallenough for the intended application.

2.3. Coupled Shorted-Patches Technique. As mentioned inSection 1, the employment of coupled shorted patches enablesthe design of extremely low-profile antennas, lower than0.01𝜆

0, with very good immunity from the influence of a

human body located in the close vicinity. The structure isderived from the standard shorted-patches antenna. Despiteits many virtues, the standard shorted-patches antenna suf-fers from a significant fall in its radiation efficiency (below50%) if the relative thickness of the substrate drops belowapproximately 0.01𝜆

0; see [27]. This phenomenon can be

International Journal of Antennas and Propagation 3

Resonating cavityFeeding point

∼𝜆0/4

(a)

Resonating cavityFeeding point

∼𝜆0/2

(b)

(c)

(d)

Figure 2: Cross section ((a) and (b)) and electric field distribution ((c) and (d)) of the standard shorted patch and coupled shorted patches,respectively.

eliminated if there are two quarter-wavelength patches thatare strongly coupled by a narrow gap; see Figure 2(b).

The radiation properties of this coupled structure are sig-nificantly enhanced even in the case of low-profile substrateswith thicknesses below 0.01𝜆

0and are, to a large extent,

insensitive to the width of the coupling gap. The electricfield distributions of the standard shorted patch and coupledshorted patches are demonstrated in Figure 2.

Considering the analogy of the coupled shorted-patchesradiator with the common patch antenna, the radiationmight be explained on the basis of the transmission linemodel (TLM) [28].The radiation of the common rectangularpatch antenna is attributed to the electromagnetic fieldson the opposite edges along the resonance length of thepatch. The phase shift of the normal components of thefield to the ground plane is 180∘ and, consequently, the farfield from these components is minimized. By contrast, thephase shift of the tangential components of the field is 0∘and consequently the far field from these components ismaximized. However, the normal E-field component passingthrough the substrate causes additional dielectric losses. Onthe other hand, coupled shorted patches concentrate theelectric field into the coupling slot going partially through theair above the substrate, and they thus minimize the dielectriclosses in the only one coupled radiation slot.

Figure 3 compares the simulated radiation efficiencies ofboth the standard half-lambda and quarter-lambda patch andcoupled shorted-patches antennas, which use the same sub-strate and have the same footprint size (60 × 100 × 0.76mm).

820

0

10

20

30

40

50

60

70

Radi

atio

n effi

cien

cy (%

)

830 840 850 860 870 880

Frequency (MHz)

Dipole-excited antennaDirectly excited antenna

Standard half-lambda patchStandard quarter-lambda patch

Figure 3: A comparison of the radiation efficiency of standard patchand coupled shorted-patches antennas using the same substrate andfootprint size (60 × 100 × 0.76mm).

Standard patches exhibit very low radiation efficiency ofabout 10% for ℎ/𝜆 ∼ 0.005, while coupled shorted patchesexhibit efficiency better than 50% in the European frequencyband (865–869MHz).

The coupled shorted-patches antenna can be excited bya linear planar radiator etched on a very thin superstrate, ordirectly by a symmetrical feeder (e.g., an RFID chip), insertedinto the coupling slot between the inner patch edges. Ways of


Upper substrate Lower substrate Ground planePatch

Excitation dipole

Coupling slotChip

(a) (b)

Figure 4: A sketch and a photograph of a dipole-excited coupled shorted-patches antenna.

0.8 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 0.98 1

0

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500

Frequency (GHz)

Resis

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e (O

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50

Patches length = 91.7mmPatches length = 94.1mm


Patches length = 96.5mm

(a)

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400

0.8 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 0.98 1

Frequency (GHz)

−100

−200



Patches length = 96.5mm

Reac

tanc

e (O

hm)

(b)

Figure 5: Tuning the real and imaginary parts of the input impedance by varying the patch length.

0

0.8 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 0.98 1

Frequency (GHz)

100

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500

Gap width = 0.25mm Gap width = 0.75mmGap width = 2mm

50

Gap width = 0.5mm

Resis

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(a)

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Frequency (GHz)

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Gap width = 0.25mm Gap width = 0.75mmGap width = 2mm

50

−50

−150

−100

Reac

tanc

e (O

hm)

Gap width = 0.5mm

(b)

Figure 6: Tuning the real and imaginary parts of the input impedance by varying the width of the coupling gap.


0.8 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 0.98 1

Frequency (GHz)

Resis

tanc

e (O

hm)

0

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50

Dipole length = 63mmDipole length = 66mm

Dipole length = 60mm

Dipole length = 54mmDipole length = 57mm

(a)

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Frequency (GHz)

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300

400

−100

−200

Dipole length = 63mmDipole length = 66mmDipole length = 57mm



Reac

tanc

e (O

hm)

(b)

Figure 7: Tuning the real and imaginary parts of the input impedance by varying the length of the excitation dipole.

0.8 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 0.98 1

Frequency (GHz)

0

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Dipole substrate thickness = 0.05mmDipole substrate thickness = 0.25mmDipole substrate thickness = 0.45mm

Resis

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hm)

(a)

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Frequency (GHz)

100

150

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250

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350

50

−50

−150

−100

0

Dipole substrate thickness = 0.05mmDipole substrate thickness = 0.25mmDipole substrate thickness = 0.45mm

Reac

tanc

e (O

hm)

(b)

Figure 8: Tuning the real and imaginary parts of the input impedance by varying the height of the superstrate layer.

tuning and matching the impedance to the chip impedanceusing the two types of feeding are described in the followingsection.

3. Parametric Study of Tuning the InputImpedance of Coupled Shorted Patches

3.1. Dipole-Excited Coupled Shorted Patches. The total size ofthe dipole-excited coupled shorted-patches antenna consid-ered here is 60 × 95 × 1mm (the relative size is equal to0.17 × 0.28 × 0.003𝜆

0at 869MHz). The dielectric constant

accounts for 𝜀𝑟= 3.2, while its loss tangent reaches tan

𝛿 = 0.002. A planar double meander folded dipole 1 mm inwidth (see Figure 4) and 58 × 16mm in outer size is used asan excitation element. However, other radiator shapes mightbe considered, such as a meander dipole, a folded dipole, anda loop antenna; see [6, 29–31].

The principle for setting the antenna impedance forthe required complex value at the specified frequency ofthe European UHF RFID band is based on the change inthe current distribution on the excitation meander dipoleby modifying its geometry and size in conjunction withmodifying the size of the coupled-patches structure. Theparametric study employs impedance-sensitive dimensionssuch as patch length, coupling gap width, excitation dipolelength, and superstrate height to tune the input impedancefor a complex conjugate value to the RFID chip impedance𝑍chip = 76 − 𝑗340Ω at observation frequency 869MHz.The changes are applied just to this one parameter, whilethe other parameters remain the same as mentionedabove.

The first study presents the sensitivity of the antennaimpedance to the patch length; see Figure 5. The real partvaries between a few and 400Ω, and the imaginary part varies


0.8 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 0.98 1

Frequency (GHz)

0

10

20

30

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60

70

Substrate thickness = 0.3mmSubstrate thickness = 0.53mmSubstrate thickness = 1.22mm

Substrate thickness = 0.99mmSubstrate thickness = 0.76mm

Radi

atio

n effi

cien

cy (%

)

Figure 9: Radiation efficiency as a function of patch substrateheight.

between −100Ω and 350Ω at observation frequency for apatch size between 94 and 96.5mm.

Significant sensitivity of the input impedance depends onthe width of the coupling gap, where changes between 0.75and 2.0mm cause a variation of the input resistance between50Ω and 420Ω and input reactance between −100Ω and320Ω; see Figure 6.

Further, the length of an excitation dipole placed on asuperstrate layer effectively changes just the input reactancebetween −50Ω and 50Ω for dipole length varying between54 and 66mm; see Figure 7.

Finally, varying the height of the superstrate layerbetween 0.05 and 0.45mm changes the input resistancebetween approximately 25 and 100Ω and changes the reac-tance between −60Ω and 60Ω, respectively; see Figure 8.

As can be seen from the previous study, basic tuning ofthe antenna input impedance is performed by changing thepatch length, whereby the radiation efficiency in the vicinityof the maximum is ensured simultaneously. The patch lengthshould be set to a quarter-wavelength on the substrate thatis used. Then finer tuning can be achieved by means of thedipole length and the gap width.

The radiation efficiency is dominantly affected by theheight of the patch substrate; see Figure 9. The other trackedparameters do not have a significant influence.

3.2. Directly Excited Coupled Shorted Patches with TuningSlots. Another way to simplify the structure of the radiatoris by removing the upper substrate and exciting directly bythe symmetrical feeder, inserted into the slot situated betweenthe inner patch edges. Unfortunately, this structure does notsuffice the capability of the impedance tuning. However, thiscould be solved by inserting two tuning slots as reactiveelements placed symmetrically on both sides of the couplingslot; see Figure 10.

The total size of the directly excited coupled-patchesantenna is 60 × 100 × 0.76mm (the relative size is 0.17×0.29×0.0022𝜆

0at 869MHz). The dimensions of the impedance

tuning slots are as follows: length 40mm, width 6mm,and distance from the coupling gap 7mm. The antenna isperformed on a low-permittivity substrate 𝜀

𝑟= 3.2, while its

loss tangent is tan 𝛿 = 0.002.The principle for setting the antenna impedance for the

required complex values at the specified frequency is basedon the change in the field distribution of the inner part ofthe coupled patches using the tuning slots and consequentlythe overall antenna size. The parametric study again employsimpedance-sensitive dimensions such as patch length andwidth, coupling gap width, tuning slot length, and theirdistance from the coupling gap to tune the input impedancefor the complex conjugate value to the RFID chip impedance𝑍chip = 76 − 𝑗340Ω at observation frequency 869MHz. Thechanges are again applied just to the one parameter, while theother parameters remain the same, as mentioned above.

The first study of a directly fed shorted-patches antennapresents the sensitivity of the antenna impedance to the patchlength and width; see Figure 11. The input resistance variesbetween a few and approximately 50Ω, and the imaginarypart varies between 90Ω and 230Ω at observation frequencyfor a patch length between 95 and 105mm.This extent is notsufficient for matching the intended RFID chip. In addition,the input resistance varies between approximately 20 and1100Ω, and the imaginary part varies between 200 and 500Ωat observation frequency for a patch width between 42 and78mm.

There is significant sensitivity of the input impedance tothe width of the coupling gap, where changes between 0.3 and5.7mm cause variation of the input resistance between a fewΩ and 600 Ω and variation in input reactance between 120and 400Ω; see Figure 12.

Further, the length of the tuning slots effectively changesboth the input resistance between a few and 600Ω and thereactance between −280Ω and 380Ω for length varying from160 to 168mm; see Figure 13.

Finally, varying the distance of the tuning slot from acoupling slot between 1.5 and 4.5mm changes the inputresistance between a few and 120Ω and changes the reactancebetween 140Ω and 340Ω; see Figure 14.

Similarly, as for the dipole-excited case, basic tuning ofthe antenna input impedance can be performed by changingthe patch length, whereby the radiation efficiency in thevicinity of themaximum is ensured simultaneously.Thepatchlength should be closer to the quarter-wavelength on thesubstrate that is used. Then finer tuning can be achieved bymeans of the gap width and by tuning the position and thewidth of the slots.

The radiation efficiency is again dominantly affected bythe height of the patch substrate; see Figure 15. The othertracked parameters have no significant influence.

4. Measurement and Identification Tests

Both antenna prototypes discussed above were manu-factured and their performance properties were verifiedby measurement. The simulated and measured reflectioncoefficient, the total efficiency in free space and in the


SubstrateGround plane

PatchCoupling slot

Tuning slots

Chip

(a) (b)

Figure 10: Sketch and photograph of the directly excited coupled shorted-patches antenna with tuning slots.

Resis

tanc

e (O

hm)

0.8 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 0.98 1

Frequency (GHz)

Patches length = 95mmPatches length = 105mm

Patches length = 100mm


800

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100

0

(a)

0.8 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 0.98 1

Frequency (GHz)

Patches length = 95mmPatches length = 105mm

Patches length = 100mm


0

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−100Reac

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hm)

(b)

Resis

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Frequency (GHz)

1200

1000

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0

Patch width = 42mm

Patch width = 78mmPatch width = 60mm

(c)

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Frequency (GHz)

800

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200

0

−200

−400

−600

Reac

tanc

e (O

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Patch width = 42mm

Patch width = 78mmPatch width = 60mm

(d)

Figure 11: Tuning the (a) real and (b) imaginary parts of the input impedance by varying the patch length, and the same ((c) and (d)) forpatch width.


0.8 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 0.98 1

Frequency (GHz)

Gap width = 0.3mmGap width = 1.5mm


Gap width = 3mm

Resis

tanc

e (O

hm)

800

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100

0

(a)

0.8 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 0.98 1

Frequency (GHz)

Gap width = 0.3mmGap width = 1.5mmGap width = 3mm

400

300

200

100

0

−200

−100

−300

−400

Reac

tanc

e (O

hm)


(b)

Figure 12: Tuning the real and imaginary parts of the input impedance by varying the width of the coupling gap.

0.8 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 0.98 1

Frequency (GHz)

Turning slot length = 160mmTurning slot length = 163.2mmTurning slot length = 164mm

Turning slot length = 164.8mmTurning slot length = 168.1mm

Resis

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e (O

hm)

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(a)

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Frequency (GHz)

Turning slot length = 160mmTurning slot length = 163.2mmTurning slot length = 164mm

Turning slot length = 164.8mmTurning slot length = 168.1mm

400

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100

0

−200

−100

−300

−400

Reac

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(b)

Figure 13: Tuning the real and imaginary parts of the input impedance by varying the tuning slot length.

Table 1: Simulated and measured electrical parameters of coupled-patch antennas.

Antenna 𝑆11,free (dB) 𝑆

11,body (dB) 𝜂𝑡,free (%) 𝜂

𝑡,body (%) 𝐺free (dBi) 𝐺body (dBi)Dipole-excited simulation −28 −15 54 57 1 1.1Dipole-excited measurement −13 −16 47 64 1.3 1.7Directly excited simulation −26 −18 56 58 1.1 0.9Directly excited measurement −23 −14 52 55 1.0 1.6

Table 2: Measured maximum identification distance in the corridor of building 4m in width and 3.5m in height.

Tested configuration Reader and tag antenna axis offset [m] Reach of correct identification 𝑑max [m]

Dipole-excited antenna on the chest of a person 0 7.51.8 7

Directly excited antenna on the chest of a person 0 71.8 6


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Frequency (GHz)

Resis

tanc

e (O

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Turning slot position = 1.5mm

Turning slot position = 4.5mmTurning slot position = 3mm

800

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100

0

(a)

0.8 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 0.98 1

Frequency (GHz)

Turning slot position = 1.5mm

Turning slot position = 4.5mmTurning slot position = 3mm

400

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0

−200

−100

−300

−400

Reac

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(b)

Figure 14: Tuning the real and imaginary parts of the input impedance by changing the distance of the tuning slot from the coupling gap.

0.8 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 0.98 1

Frequency (GHz)

Radi

atio

n effi

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cy (%

)

Substrate thickness = 1.22mmSubstrate thickness = 0.99mmSubstrate thickness = 0.76mm

Substrate thickness = 0.53mmSubstrate thickness = 0.3mm

90

80

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60

50

40

30

20

10

0

Figure 15: Radiation efficiency of directly fed coupled shortedpatches with tuning slots as a function of the height of the patchsubstrate.

close vicinity of a human body of both coupled shorted-patches antennas and the power gain are compared inTable 1. Good immunity against the influence of a humanbody can be seen for the final matched prototypes.

The measurement of the designed antenna was carriedout in the monopole arrangement, in order to avoid theuse of a balun situated between the antenna and the coaxialconnector. The monopole input impedance then accountsfor a half of the value compared to the dipole impedance[8]. Consequently, 𝑍monopole = 𝑍dipole/2 is considered forfurther evaluation (where 𝑍dipole = 𝑍chip

∗= 76 + 𝑗340Ω).

The radiation efficiency was evaluated from the impedancemeasurement performed using the Wheeler cap method[32]. The cap size was equal to 122 × 122 × 122mm. Themeasurementwas performedwith andwithout a humanbodyphantom (manufactured from agar with 𝜀

𝑟∼ 55 and tan

𝛿 ∼ 0.5 of 80 × 110 × 15mm size), which was enclosed intothe back of the antenna.

In order to evaluate the performance of the tag antennasin real system conditions, a read range test was performed in a4m wide building corridor. Tag antennas with the chip werefixed at a height of 1.25m on the chest of a person. The testresults are included in Table 2.

5. Conclusion

An illustration has been provided of the flexibility of thetuning of coupled shorted-patches RFID tag antennas forthe required complex input impedance using two differentfeeding techniques.The parametric studies of the influence ofgeometrical modifications on the input impedance behaviourcan significantly facilitate the design of this class of on-bodytag antennas.

The prototypes presented here were matched to RFIDchip impedance 𝑍chip = 76 + 𝑗340Ω at frequency 869MHzwith concurrent very good immunity from the influence of ahuman body situated in the direct vicinity of the antenna.Themeasured radiation efficiency of the structure was sufficient,typically better than 50%.

Conflict of Interests

The authors declare that there is no conflict of interestsregarding the publication of this paper.

Acknowledgments

This research was undertaken at the Department of Electro-magnetic Field at the Czech Technical University in Prague.It was supported by the Czech Science Foundation Projectno. P102/12/P863 “Electromagnetic Properties of RadiatingStructures and Artificial Screening Surfaces in the CloseVicinity of the Human Body.”


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