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7/29/2019 Ref[1] Refered Circuit
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IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMSI: REGULAR PAPERS, VOL. 59, NO. 6, JUNE 2012 1215
A 1-V 5-GHz Self-Bias Folded-Switch Mixer in90-nm CMOS for WLAN Receiver
Hwann-Kaeo Chiou, Member, IEEE, Kuei-Cheng Lin, Wei-Hsien Chen, and Ying-Zong Juang
AbstractA 5 GHz double balanced mixer (DBM) is imple-mented in standard 90 nm CMOS low-power technology. Anovel low-voltage self-bias current reuse technique is proposedin the RF transconductance stage to obtain better third-order
intermodulation intercept point (IIP ) and conversion gain (CG)when considering the process variations. The DBM achieves a CGof 12 dB, a noise figure (NF) of 10.6 dB and port-to-port isolationsof better than 50 dB. The input second-order (IIP ) and IIP
are 48 dBm and 4 dBm, respectively. Two I/Q DBMs are thenintegrated with a differential low-noise amplifier (DLNA) and a
poly-phase filter, to from a direct-conversion receiver (DCR). TheDCR achieves a CG of 26 dB with an NF of 2.7 dB at 21 mWpower consumption from a 1 V supply voltage. The port-to-port
isolations are better than 50 dB. The IIP and the IIP of the DCRare 33 dBm and dBm, respectively.
Index TermsAC coupling, CMOS, current reuse, direct-con-version receiver, double balanced mixer, folded-switch mixer, lowvoltage.
I. INTRODUCTION
C MOS technology satisfies the requirements of the lowpower consumption, compact size, and high integrationlevel in multi-GHz radio frequency (RF) systems-on-a-chip
(SoC) designs. In particular, the direct-conversion receiver
(DCR) architecture has been chosen for the use in RF SoCsbecause of its relative simplicity and low cost [1], [2]. The
critical impacts of DCR are dc offset, IQ mismatch, even-order
intermodulation, LO leakage and radiation, flicker noise, and
dc power consumption. The dc offset which originates from
the self-mixing of the LO signal leakage, strong in-band in-
terferences and device mismatch, is the most critical design
issue in DCR. The dc offset associated with self-mixing effect
is mostly due to insufficient isolation between the LO and
RF ports [3], [4]. Therefore, increasing the isolation of the
mixer is a simple method of mitigating self-mixing effect. The
DBM naturally has high port-to-port isolations and directly
relieves the dc offset. The conventional Gilbert cell DBM
provides high port-to-port isolations with high CG, and is thusfavored for integrated circuit applications [5], [6]. However,
Manuscript received February 11, 2011; revised May 03, 2011; acceptedSeptember 13, 2011. Date of publication December 14, 2011; date of currentversion May 23, 2012. This work was supported by the National ScienceCouncil under Grant NSC 99-2221-E-008-100-MY3. This paper was recom-mended by Associate Editor P.-I. Mak.
H. K. Chiou and K. C. Lin are with the Department of Electrical Engineering,National Central University, Jhongli 320, Taiwan (e-mail: [email protected]).
W.-H. Chen and Y.-Z Juang are with the National Chip ImplementationCenter (CIC) of National Applied Research Laboratories, Hsinchu 300, Taiwan.
Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TCSI.2011.2173399
the LO/RF and RF/LO isolations of conventional Gilbert cell
DBM are about 4050 dB, which are constrained by the device
mismatching and parasitic coupling from the lossy substrate.
Moreover, the Gilbert cell DBM needs three stacked transistors,
which wastes the voltage headroom and compresses the output
voltage amplitude and inevitably influences the linearity and
power consumption of the mixer. Since the input signal to the
mixer is higher than that to the LNA and no filter is placed
before LNA/mixer stages, a highly linear mixer is important
in DCR (i.e., high IIP and IIP ) to decrease the high order
distortions. This drawback becomes increasingly severe with
the shrinkage of the feature size in CMOS technology. Thelow-voltage operation requires fewer stacked battery cells to
reduce the power consumption and the battery weight. In the
past few years, several mixer designs have been published
using stacked cascode configuration. These designs achieve
low-power and low-voltage performance despite their relative
low CG and IIP due to the limited voltage swing across the
IF load [7][16]. Therefore, a folded-switch topology has been
proposed to reduce the supply voltage and power consump-
tion while keeps high CG, IIP and port-to-port isolations,
[17][23]. However, few studies discussed these important
features related to low-voltage operation.
Thefl
icker noise of the transistors strongly affects the base-band signal because the down-converted spectrum of DCR is
around zero frequency. Meanwhile, the flicker noise due to the
LO switch core must be considered in determining NF perfor-
mance. Therefore, a PMOS switch core is often preferred in low
flicker noise mixer [24][28].
This work proposes a low-voltage topology to achieve a
DBM with high CG, IIP , IIP , isolations and low NF simul-
taneously. A 5 GHz CMOS DBM operating at 0.7 and 1.0 V
supply voltages is designed to investigate the above mentioned
performance. Fig. 1 shows the block diagram of the designed
90 nm CMOS DCR. Two designed mixers are combined with
a differential low noise amplifier (DLNA), a poly-phase filter,
and two buffer amplifiers to form a 4.75.7 GHz DCR. The
DLNA amplifies the input RF signal which is then down-con-
verted to zero intermediate frequency (IF) by two quadrature
LO signals.
II. CIRCUIT DESIGNS FOR LOW VOLTAGE OPERATION
A. Self-Bias Current Reuse Technique and AC-Coupling
Folded-Switch Mixer
Fig. 2 shows the conceptual diagram of the proposed DBM
for low-voltage operation. The newly developed self-bias
current reuse technique has two advantages: 1) improves the
CG and IIP in the RF transconductance stage, and 2) uses
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Fig. 1. The block diagram of direct-conversion receiver.
Fig. 2. The conceptual diagram of the proposed ac-coupling, PMOS folded-switch DBM for low-voltage operation.
single PMOS folded-switch transistor with LC tank to allow
low-voltage operation. The resonant frequency of the LC tank
is designed around the RF and LO frequencies to prevent the
signals leakage through the power supply. Since the LC tank
requires no voltage headroom between the supply voltage rail
and ground, the voltage across the IF load can be maximized
and effectively increases the CG and IIP . Furthermore, the
PMOS transistor has lowerflicker noise than its NMOS coun-
terpart, and thus the NF is improved. The PMOS switch core
also has high linearity and low NF under low LO power at low
bias current. Fig. 3 shows the schematic diagram of the pro-posed mixer, which consists of an RF transconductance stage
, a self-bias current reuse bias circuit , a
PMOS LO folded-switch core , two ac-coupling
capacitors, two LC tanks , and two IF loads .
B. Conversion Gain, Noise Figure and Linearity
The ideal mixer multiplies the RF input signal by the LO
pumped signal to produce the sum or difference signal, i .e., in-
termediate-frequency (IF). If the LO signal is an ideal square
wave, then the CG of the mixer is given by (1).
(1)
Fig. 3. Schematic diagram of the proposed mixer.
The values of and are 210 and 0.032 (A/V)
in this design, the calculated CG is 12.7 dB. If the LO waveform
is a sinusoidal, then the CG is given by (2) [24].
(2)
where is the time interval of the positive and negative period
turning on simultaneously. isthe LO frequency.At high LO
amplitude, the LO waveform resembles a square waveform and
its fundamental coefficient of the power series is approximately
. At low LO power level, the higher LO power corresponds
to the higher CG. However, when the mixer is driven by too
much LO power, the harmonic distortion reduces the CG. In
practice, the optimum LO power level is set to maximize CG.
Fig. 4 shows the CG with respect to the LO power. The optimalLO power levels for the NMOS and the PMOS switch cores
are 0 and dBm, respectively. Although the PMOS switch
core has a lower CG than its NMOS counterpart, the CG of the
PMOS switch core exhibits a plateau response over an extended
LO power [24].
The PMOS switch core has a lower sensitivity to the LO
power level than the NMOS one. Thus, a mixer with a PMOS
switch core requires less LO power and relaxes the required
output power from the VCO design.
Flicker noise in CMOS mixer has been analyzed in detail
elsewhere [24][28]. Fig. 5 shows the simulated flicker noise of
the PMOS and the NMOS transistors use for the LO switch core
in tsmc 90 nm CMOS process. The PMOS transistor has a
lowerflicker noise than that in the NMOS transistor[24]. Hence,
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Fig. 4. The conversion gain versus the LO power using the PMOS and theNMOS switch cores.
the PMOS switch core exceeds NMOS in the flicker noise per-
formance. The total noisesof the direct-conversion mixer are the
sum of the flicker noise and the thermal noise associated with
the RF transconductance, the LO switch core, and IF load. The
MOSFET transistor is known to have significant flicker noise.
Equations (3)(6) show the thermal and flicker noise contribu-
tions to the NF of the proposed mixer.
(3)
(4)
(5)
(6)
where , and are the flicker noise, drain current of tran-
sistor, and the peak-to-peak voltage at the LO switch
core. and are the process-dependent constant, bias-de-
pendent factor and source resistor. is the drain current of the
RF transconductance stage. is the power gain of the mixer.
The flicker noise and thermal noise dominate the noise perfor-
mance of the mixer. According to (3) and (4), increasing the LOamplitude of the switch core improves the noise performance.
The PMOS LO switch core has better NF performance than the
NMOS switch core because the of PMOS transistor is
low. The process parameters in tsmc 90 nm CMOS process
are listed as follows: the of PMOS is 0.44 nV at 10 MHz,
and . The value for a short
channel device is about 1 [29]. The power gain is 12 dB. In
this design, the NF is 10.1 dB according to (6).
Equation (7) shows the power-series expansion of the voltage
transfer function that is adopted to evaluate the linearity [3],
[30].
(7)
Fig. 5. Comparison of the flicker noise of the PMOS and the NMOS switchcores.
The first four terms in (7) are sufficient to characterize the
nonlinearity in practical mixer. In tsmc 0.18 m CMOS
process, the short channel model of drain current is approx-
imately as (8) [31].
(8)
(9)
where is the field-limited electron mobility, is the gate
oxide capacitance per unit area, and is the velocity satu-
ration field strength. Equations (11) and (12) indicate the IIP
of the self-bias current reuse with ac-coupling folded-switch
mixer. is the overdrive voltage of the RF transconductance
stage. The terms of and are the (A/V ) of the NMOSand PMOS, respectively. The terms of and are the (V)
of the NMOS and PMOS, respectively. The values of and
in tsmc 90 nm CMOS process are 0.067 and 0.045. The
values of and in tsmc 90 nm CMOS process are 0.43
and 2. In this design, the IIP is 2.2 dBm according to (11)(12).
(10)
(11)
(12)
C. RF Transconductance for Low Voltage Operation
According to (2), the RF transconductance stage is mainly
responsible for the CG. Fig. 6 presents six possible topologies
of the RF transconductance stage for low-voltage operation.
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Fig. 6(a) shows a common source (CS) amplifier with a resis-
tive load for the RF transconductance stage. The resistive load
not only degrades the NF performance but also wastes the
voltage headroom. This drawback can be mitigated by using an
active current source load instead of the resistive load, as shown
in Fig. 6(b). Since the output impedance of the PMOS transistor
is sufficiently high, the NF and CG are improved. The PMOS
current source can be further used to amplify the RF signal and
the amplifier becomes a CMOS inverter, as shown in Fig. 6(c).
This inverter results in high gain and low-power performance
because it combines the transconductances of the parallel-con-
nected NP MOS transistors with current reuse topology [9].
When the input signal has positive-going swing, the PMOS tran-
sistor moves toward the cut-off region and provides an equiva-
lent load resistance forthe NMOS CS amplifier. On the contrary,
when the input signal has negative-going swing, the operation
region of the transistors is exchanged. The PMOS transistor is
used as a CS amplifier with an NMOS resistive load. The gain
of the CMOS inverter equals the sum of the gain of the NMOS
and PMOS CS amplifiers. The total transconductance of the RFtransconductance stage equals . The terms and
are the transconductances of the NMOS and PMOS, re-
spectively. Equation (13) presents the minimal required voltage
of the topology in Fig. 6(c).
(13)
where
and are the overdrive voltage of the NMOS and the
PMOS transistors, respectively. and are the threshold
voltage of the NMOS and the PMOS transistors. is
gate-source voltage of the PMOS transistor. and are
the bias voltages at gate of the NMOS and PMOS transistors.
As seen in (13), the overdrive voltage of the NMOS and PMOS
transistors determines the minimal required voltage of the
CMOS inverter. Fig. 6(d) shows a resistive feedback inverter as
the fourth topology for the RF transconductance stage. When
the feedback resistance exceeds 3 k , the gain is close to that
of the inverter without feedback. , is the same as (13).
In general, the value of in tsmc 90 nm CMOS process is
typically 210 mV. If the supply voltage is 1 V, then theresulting IIP is approximately dBm, which value limits
the dynamic range of the receiver. To overcome this limitation,
the topology of the CMOS inverter is modified as shown in
Fig. 6(e) [18]. The parallel-connected NP MOS transistors can
be separately biased by adding a capacitor at the gate node. The
of this topology is expressed as (14).
(14)
If is chosen to be greater than , then can
be reduced. In this way, the value of the IIP in tsmc 90
nm CMOS process is approximately 1 dBm, giving a 4.5 dB
in IIP improvement over the conventional CMOS inverter as
presented in Fig. 6(d). Although this biasing scheme reduces the
required supply voltage and increases IIP , it requires two inde-
pendent biases. Fig. 6(f) shows the proposed self-bias scheme
for the PMOS transistor to overcome this drawback. This self-
bias scheme enables the gate voltage of the PMOS transistor is
directly biased from the dc output which saves one bias path
compared with [18]. The self-bias condition of the PMOS tran-
sistor has less one bias in bias path. The minimal required
voltage can be set by selecting the value of according to
(15).
(15)
The transconductance stage of [18], as shown in Fig. 6(e),
requires two independent biases ( and ) for NMOS
and PMOS transistors. Fig. 6(f) shows the self-biased topology
in the RF transconductance stage which is a typical shunt-shunt
feedback configuration. Since the drain and the gate of the
PMOS transistor have the same voltage, the PMOS transistor
always operates in saturation. The advantages of this topology
compared with [18] are in four folds. First, the gate voltage ofthe PMOS transistor is directly biased from the output which
saves one bias path compared with [18]. Second, a smaller
feedback resistor is used for broad-band matching which
achieved the input return loss better than 10 dB. Third, the
proposed transconductance stage uses the resistive feedback
topology that improves the linearity and resists the PVT vari-
ations. The IIP of the transconductance stage compared with
that of [18] has less sensitivity to the process variations. Fourth,
the self-biased topology of the PMOS transistor always oper-
ates in saturation to achieve better IIP and transconductance
compared with those in [18] when considering the process
variations.The device mismatch model was applied in the simulations
to evaluate all performance of the proposed mixer, including
dc bias conditions, CG, NF, IIP , IIP , and isolations. The
mismatch model provides the device size-dependent mismatch
model to reflect the size-dependent mismatch behavior. The
random variations in Gaussian distribution of the process
parameters including / and of the transistors are
included in the model to account for the mismatch performance
of and parasitic capacitor effect.
Take tsmc 90 nm CMOS process as an example. Figs. 7
and 8 present the simulated transconductance and IIP of six
RF transconductance stages. As observed, the self-bias current
reuse topology achieves the smallest error of transconductance
and IIP performance among all possible topologies with de-
vice mismatch effect. As indicated in simulations, the proposed
topology decreases the errors in transconductance of 2.6 mS and
IIP of 3.5 dBm as compared with the topology in Fig. 6(e).
D. Isolation
The dc offset caused by self-mixing effect mostly comes from
insufficient LO/RF and RF/LO isolations of the mixer.
In a DBM as shown in Fig. 9(a), the non-ideal effects such
as device mismatch, layout asymmetry, substrate leakage,
radiation, and parasitic capacitor loading at the source of the
switching pair provide several leakage paths
from LO to RF port and thus reduce the isolations. Fig. 9(b)
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Fig. 6. Six possible topologies of the RF transconductance for low-voltage operation. (a) CS stage with the resistive load. (b) CS stage with the current source asactive load. (c) CMOS inverter. (d) CMOS inverter with the resistive feedback. (e) PMOS and NMOS transistors are separately biased. (f) CMOS inverter withthe PMOS self-bias current reuse technique.
Fig. 7. Simulated transconductance of six possible different topologies for theRF transconductance stage.
shows the equivalent half circuit around the RF feed-node
of the proposed mixer. The LO-to-RF leakage produced by
the drain-to-gate capacitor of the RF stage is partially
attenuated by the shunt feedback resistor that improves
LO-RF isolation.
(16)
Fig. 8. Simulated IIP of six possible different topologies for the RF transcon-ductance stages.
The value of is15 fF. Fig.10 shows the simulated LO-to-RF
isolation versus different value of . If is set to open cir-
cuit, thecircuit can be taken as a conventional Gilbert cell mixer.
When the value is increased gradually, the LO-to-RF isola-
tion is improved. As can be seen, when is 300 that im-
proves a 12-dB LO-to-RF isolation. Fig. 11 shows the simulated
LO-to-RF isolation of the conventional and proposed mixers
driven by the differential LO with a phase imbalance of 0 to
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Fig. 9. The ideal and actual differential models. (a) Double balanced. (b) LC tank ac-coupling folded-switch with self-bias current reuse technique.
Fig. 10. LO-to-RF isolation versus .
. As can be seen, the proposed mixer improves a 15 dB
LO-to-RF isolation.
E. Differential Variable Gain Low Noise Amplifier
Fig. 12 shows the schematic diagram of the DLNA with vari-
able gain control. The input match network uses a capacitor in
series with a bond-wire inductor. , and are
chosen to simultaneously match the minimal reflection and NF.
The inductors are realized by using on-chip spiral inductors.
The inductance with its parasitic capacitance is designed for
output matching to maximize the gain at the frequency band of
interest. A current-splitting technique is applied here for vari-
able gain control. The current-splitting transistors and
Fig. 11. LO-to-RF isolation.
are controlled by varying the control voltage . When
is set to 0 and are OFF and the DLNA
operates in high gain mode. When is set to 1
and are ON and the DLNA operates in low gain mode.
F. Poly-Phase Filter
The poly-phase filter is a symmetric RC network whose
outputs are with phase differences. The differential input
voltage is converted to quadrature phase of the LO signal by
a multi-order poly-phase circuit [32], [33]. Since unwanted
phase difference occurs due to the asymmetric interconnection
through the first stage, the capacitor value is optimized in
the second stage to achieve the best quadrature accuracy of
the ployphase filter. The optimized is 0.165 pF. Fig. 13(a)
presents the schematic diagram of the 5.19 GHz poly-phase
filter and output buffer amplifiers with their detail design
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Fig. 12. Schematic diagram of the fully differential LNA with variable gaincontrol.
Fig. 13. (a) Schematic diagram of the poly-phase filter. (pF, pF). (b) Output buffer (
m/0.1 m, pF, nH).
parameters. The output buffer followed by the second stage of
poly-phase filter is used to avoid the loading effect. Fig. 13(b)
shows the schematic diagram of the output buffer which adopts
a single stage differential cascode topology. The inductance
with its parasitic capacitance is designed for the output
matching to maximize the power to drive the next mixer stage.
III. MEASUREMENT RESULTS
A. Low-Voltage AC-Coupling Folded-Switch DBM
The circuit was simulated using Agilent Advanced Design
System (ADS) simulator. The self-bias current reuse mixer
and direct-conversion receiver were designed and fabricated
Fig. 14. Chip photo of the fabricated mixer.
using tsmc 90 nm CMOS 1P9M low-power technology.
The RF model included standard NMOS/PMOS, an MIM
capacitor with or without an underground metal, an MOS
varactor, a junction varactor, a resistor, an RTMOM and nine
metallization layers with a deep N-well (DNW)/P-substrate.
The model included three scalable inductor models, standard,
symmetric and symmetric with center tap. The and ofthe NMOS in the transconductance stage are 102 GHz and 85
GHz, respectively.
Fig. 14 shows the photo of the fabricated mixer. The chip size
is 0.545 mm . Fig. 15 shows the measured input return losses
of the RF and LO stages. The RF stage achieves the return loss
better than 10 dB from 4.7 to 5.8 GHz. Fig. 16 shows the CG
with respect to the LO power. The peak CG is obtained at an LO
power of dBm. When the LO power is less than dBm, the
CG declines because the switch core operates in the triode re-
gion and cannot perform good current commutation. When the
LO power is higher than dBm, the harmonic distortion re-
duces the CG. As can be seen, the simulated and the measuredresults agree closely with each other. Figs. 17 and 18 plot the
simulated, calculated and measured of CG and NF of the mixer,
respectively. Fig. 17 shows the frequency responses of CG and
the maximum CG is 12.3 dB at 4.8 GHz with 4.6-mW dc power
consumption. The CG is dB from 4 to 6.6 GHz. The
correspondent RF bandwidth is 2.6 GHz. The measured NF is
below 10.6 dB in Fig. 18. The measurements were performed at
a fixed IF frequency of 10 MHz. Fig. 19 shows the 1-dB com-
pression point measurement. The measured CG and the
input are 12 dB and dBm, respectively. Fig. 20 shows
the measured isolations at LO/RF power levels of and
dBm, respectively. The LO-to-IF and LO-to-RF isolations ex-ceed 52 dB at the LO frequencies from 4.5 to 8.5 GHz. The high
impedance of the LC tank at the RF and the LO frequencies pro-
vides good LO-to-RF isolation. The RF-to-IF isolation exceeds
60 dB.
Fig. 21(a), (b) plot the simulated and measured of IIP and
IIP , respectively. The two-tone intermodulation measurements
are performed using two signals with equal power at 5.189 and
5.191 GHz. The measured IIP and IIP are 48 dBm and 4
dBm, respectively. Fig. 22 plots the simulated and measured
CG and IIP as varied from 0.7 to 1.0 V. The pro-
posed topology functions properly at the of 0.7 V and
achieves a CG of 9 dB and an IIP of 0 dBm.
Table I summarizes the simulated and the measured perfor-
mance of the proposed mixer. Since the IIP of a balanced mixer
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Fig. 21. (a) The simulated and measured results of the IIP . (b) The simulatedand measured results of the IIP .
Fig. 22. The gain and IIP measurements with various .
Fig. 23. Benchmark of the low-voltage mixers.
TABLE IPERFORMANCE SUMMARY OF THE PROPOSED MIXER
B. Front-End Receiver
Fig. 24 shows the photo of the fabricated DCR. The chip area
is 2.24 mm . The dc pads are wire-bonded to the PCB to pro-
vide the dc bias. TheLO/RF signals are fed by on-wafer GSGSGprobes. The differential IF signal is buffered by a source fol-
lower output stage to match 50 termination. The receiver con-
sumes a dc power of 21 mW from a 1 V supply voltage. Fig. 25
shows that the measured input return loss is 16 dB at 5.2 GHz,
and remains better than 10 dB from 4.7 to 5.7 GHz. Fig. 26
plots the CG versus the LO power. The DCR obtains a flat CG
of 26 dB at LO power between to 0 dBm which is attributed
by the LO power insensitivity of the PMOS switch core. No-
tably, if the LO power is generated by an silicon-based VCO
at 1 V operation, the output power of VCO is unlikely to ex-
ceed 0 dBm, which power level corresponds to a of 633
mV at 50 load. This value of is very difficult to achievein low-voltage operation. Therefore the PMOS switch core is
adopted to relax the power requirement of VCO.
Fig. 27 shows the CG frequency responses of the receiver and
the maximum CG is 26.2 dB at 5.4 GHz. The CG is
dB from 4.6 to 6 GHz. The correspondent RF bandwidth is 1.4
GHz. Fig. 28 plots the simulated and the measured input
at RF and IF frequencies of 5.2 GHz and 10 MHz, respectively.
The measured input is dBm. Since the NF of DCR
is affected by its high noise power spectral density near dc fre-
quency, Fig. 29 shows the simulated and measuredNF as a func-
tion of the IF frequency to evaluate the flicker noise corner. As
seen, the corner frequency is about 600 KHz. The NF is 2.7 dBat 5.2 GHz RF frequency (IF = 10 MHz).
Fig. 30 shows the measured IIP and IIP based on two tone
test which is performed by applying two RF signals with the
equal power level at 5.189 and 5.191 GHz. The measured IIP
and IIP are 33 dBm and dBm, respectively. Since the
maximum signal handled by the receiver is dBm for both
WLAN and WiMAX standards, the obtained IIP meets the
specification. When the LO power is 0 dBm, the dc offset is
measured according the method developed in [40]. The induced
dc offset of the DCR are 2.2 mV and 0.5 mV at high gain and
low gain modes, respectively. Table III summaries the simu-
lated and the measured results of the DCR at high CG and low
CG mode, respectively. According to these experimental re-
sults, the designed DCR satisfies the requirements of WLAN
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TABLE IIPERFORMANCE SUMMARY OF THE RECENT CMOS MIXER DESIGNS
Fig. 24. Chip photo of the fabricated DCR.
Fig. 25. Measured the input return loss of the DCR.
and WiMAX standardsin 4.7 to 5.7GHz frequency band. Works
[41][43] adopted current driven passive mixer in direct conver-
sion receiver to obtain high linearity and low noise performance.
Table IV benchmarks the overall performance of recent works
and our proposed mixer [40][44].
Fig. 26. The conversion gain versus the LO power ( GHz,GHz).
Fig. 27. The simulated and measured results of the conversion gain of the re-ceiver ( MHz, RF dBm, LO dBm).
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Fig. 28. The simulated and the measured of the DCR ( MHz,LO dBm).
Fig. 29. Simulated and measured DCR NF versus IF bandwidth (GHz, LO dBm).
Fig. 30. Measurements of IIP and IIP .
IV. CONCLUSION
This work demonstrates a low-voltage self-bias folded-
switch mixer design, which maintains the performance of CG,
NF, linearity, and port-to-port isolations at low supply voltage.
In the RF transconductance stage, the self-bias current reuse
topology makes the supply voltage as low as possible and fa-
vors the CG and IIP . In the LO switch core design, the PMOS
folded-switch with the LC tank allows low-voltage operation,
and simultaneously provides low NF and high linearity. The
proposed mixer outperforms the conventional Gilbert cell
mixer by 3 dB in CG, 1.3 dB in NF, 7 dB in IIP and 10
dB in LO-to-RF isolation at the same power consumption of
4.6 mW. The measured mixer achieves very good FOM, as
compared with the overall performance of the CG, NF, IIP ,
TABLE IIIPERFORMANCE SUMMARY OF THE RECEIVER
TABLE IVPERFORMANCE SUMMARY OF THE RECENT CMOS RECEIVERDESIGNS
IIP , and power dissipation. The measured results for variouscharacteristics are all agreed with the simulations. This mixer
topology is suitable for low-voltage applications. A 1 V 4.7 to
5.7 GHz DCR has been successfully integrated and fabricated
in tsmc 90 nm CMOS low power technology. The DCR
performs very well and can be adopted in 4.7 to 5.7 GHz band
wireless products.
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Hwann-Kaeo Chiou (M05) was born in Taichung,Taiwan, in 1959. He received the B.S. degree in elec-trical physics from National Chiao Tung University,Hsinchu, Taiwan, in 1982, and the M.S.E.E. degreeand Ph.D. degrees in electrical engineering from Na-tional Taiwan University, Taipei, in 1985, and 1997,respectively.
From 1985 to 2000, he was an Associate Re-searcher with the Chung-Shan Institute of Scienceand Technology (CSIST), where he was in chargeof the development of microwave integrated circuits
(MICs), MMICs and microwave subsystems for mobile communication. From2000 to 2002, he was an Associate Vice President at BenQ Inc., where he was
in charge of the development of broadband wireless access technology. InAugust 2002, he joined the faculty of the Department of Electrical Engineering,
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CHIOU et al.: A 1-V 5-GHZ SELF-BIAS FOLDED-SWITCH MIXER IN 90-NM CMOS FOR WLAN RECEIVER 1227
National Central University, Jhongli, Taiwan, where he is currently a Professorand Chair. His current research interests include RF integrated circuits (RFICs),MMICs, and millimeter-wave integrated circuits.
Kuei-Cheng Lin was born in Kaohsiung, Taiwan,in 1981. He received the M.S. degree from the De-
partment of Electrical Engineering, National CentralUniversity, Jhongli, Taiwan, in 2006. He is currentlyworking toward the Ph.D. degree in electrical engi-neering at National Central University, Jhongli.
In October 2006, he joined National Chip Imple-mentation Center, Hsinchu, Taiwan, as an AssociateResearcher, responsible for RF front-end transceivercircuit design. His current research interests are lowpower and high linearity design for RF front-end
transceiver circuit in CMOS, SiGe BiCMOS, and compound semiconductors.
Wei-Hsien Chen received the B.S. degree in elec-trical engineering from National Taiwan OceanUniversity, Keelung, in 2003, and the M.S. degree in
electronics engineering from National Chiao-TungUniversity (NCTU), Hsinchu, Taiwan, in 2005.In 2006, he joined the System-on-Chip (SoC)
Technology Center (STC), Industrial TechnologyResearch Institute (ITRI), Hsinchu. He was an RFdesign engineer and designed an ultra-low-powerRF receiver for hearing aids. From 2007 to 2008,he designed analog baseband circuits for ultra-wide-
band and DVB-H system. In 2009, he was designing a neural sensor appliedfor biotechnology in the human body. His research interests include analogcircuit design for wireless transceivers and biotechnology applications. In2010, he joined National Chip Implementation Center (CIC), Hsinchu, wherehe is currently a Researcher and a Deputy Department Manager in charge ofthe development of system-on-chip design environment and technologies.
Ying-Zong Juang received the M.S. and Ph.D.degrees in electrical engineering from the NationalCheng Kung University, Taiwan, in 1992 and 1998,respectively.
He joined the Institute of Chip ImplementationCenter, Science-Based Industrial Park, Hsinchu,Taiwan, in October 1998. At CIC, he has majoredin the RF circuit design and device modelingworks, furthermore, from 2001 to 2004, he joineda project to develop the CMOS MEMS platform.Now, he is the Researcher and Department M anager
of CISD/CIC and his interested topics include the technologies of wirelessmicro-sensing system. Therefore, he organized several projects including RFtop-down design, RFIP methodology for frequency synthesizer, RF SiP, and0.35 m/0.18 m CMOS MEMS/BioMEMS design environment.