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Wide Input Wide Output Dc-Dc Converter
Abstract
WIWO(wide input wide output) presents a new wide-input–wide-output dc–dc
converter, which is an integration of buck and boost converters via a tapped inductor,
Coherent transition between step-down and step-up modes is achieved by a proper
control scheme that is by applying proper control to the two active switches, the
converter exhibits both buck and boost features]. This paper presents theoretical
concepts and experimental results.
EEE DEPT. HKBKCE 2012 Page 1
Wide Input Wide Output Dc-Dc Converter
Chapter 1.Introduction
The BUCK, boost, buck–boost, and Cu´ k converters are the four basic dc–dc
non-isolating converters that have found wide applications in industry. The buck
converter can step down the dc voltage, whereas the boost converter is capable to
perform a step-up function. In applications where both step-up and step-down
conversion ratios are required, the buck–boost and Cu´k converters can be used.
Simplicity and robustness are among the advantages of the buck–boost converter.
However, the pulsating input and output currents cause high conduction losses, and
thus, impair the efficiency of buck–boost. Furthermore, the buck–boost converter
uses the inductor to store the energy from the input source, and then, release the
stored energy to the output. For this reason, the magnetic components of buck– boost
are subjected to a significant stress. These disadvantages limit the applications of the
buck–boost converter mainly to low power level. The isolated version of buck–boost,
referred to aas the flyback converter, can achieve greater step-up or step-down
conversion ratio utilizing a transformer, possibly, with multiple outputs. As compared
with the buck–boost converter, the Cu´k converter has higher efficiency and smaller
ripples in input and output currents.
A significant improvement of the Cu´k converter performance can be achieved
by applying the zero ripple concept. The Cu´k converter can be found in many high-
performance power applications. In theory buck and boost converters can generate
almost any voltage, in practice, the output voltage range is limited by component
stresses that increase at the extreme duty cycle. Consequently, buck converter losses
mount at low duty cycle, whereas boost converter efficiency deteriorates when the
duty cycle tends to unity. Accordingly, voltage conversion range of the buck
converter below 0.1–0.15 becomes impractical whereas that of the boost converters’
is limited to below 8–10. Additional problems associated with narrow duty cycle are
caused by MOSFET drivers rise and fall times as well as pulse width-modulated
(PWM) controllers that have maximum pulse width limitations. These problems
become even more severe at higher voltages and higher frequencies.
Introducing a transformer helps attaining large step-up or step-down voltage
conversion ratio. Transformers turn ratio should be chosen as to provide the
EEE DEPT. HKBKCE 2012 Page 2
Wide Input Wide Output Dc-Dc Converter
desired voltage gain while keeping the duty cycle within a reasonable range for
higher efficiency. The transformer, however, brings in a whole new set of
problems associated with the magnetizing and leakage inductances, which cause
voltage spikes and ringing, increased core and cooper losses as well as increased
volume and cost.
In a quest for converters with wide conversion range, quite a few authors
proposed using converters with nonlinear characteristics. Single-transistor
converter topologies, with quadratic conversion ratios, were proposed in [1] and
demonstrated large step-down conversion ratio. This method has successfully
achieved wide conversion range in the step down direction. A different approach
to obtain wide conversion range utilizing coupled inductors was proposed in [2].
With only minor modification of the tapped-inductor buck, [2] shows low
component count and solves the gate-drive problem by exchanging the position of
the second winding and the top switch. The problem of a high turn-OFF voltage
spike on the top switch was solved by applying a lossless clamp circuit. Due to the
coupled inductor action, the converter demonstrated high step-down dc–dc
conversion ratio, whereas the converter’s efficiency was improved by the
extended duty cycle. A tapped-inductor buck with soft switching was introduced
in [3]
.
3.Tapped-Inductor Buck
Derivations of the tapped-inductor buck were also suggested in [4] and [5].
An- other modification of the tapped-buck converter was realized in [6] for power
factor correction (PFC) application. With the addition of a line-frequency
commutated switch and a diode, both flyback and buck characteristics were
achieved and large step-down was demonstrated.
EEE DEPT. HKBKCE 2012 Page 3
Wide Input Wide Output Dc-Dc Converter
Tapped-Inductor Boost Converters
Some applications, especially battery-operated equipment, require high
voltage boosting. To attain very large voltage step- up, cascaded boost converters that
implement the output voltage increasing in geometric progression were introduced in
[7]. These converters effectively enhance the voltage transfer ratio; however, their
circuits are quite complex. In comparison, tapped-inductor boost converters proposed
in [8] and [9] attain a comparable voltage step-up preserving relative circuit
simplicity.In [10], the boost converter output terminal and fly- back converter output
terminal are connected in series to increase the output voltage gain with the coupled
inductor. The boost converter also functions as an active clamp circuit to recycle the
snubber energy.
EEE DEPT. HKBKCE 2012 Page 4
Wide Input Wide Output Dc-Dc Converter
CHAPTER 2.LITERATURE SURVEY
Brief Survey of Converters
There are two types of converters: Non-Isolated Converters and Isolated
Converters.
2.1 Non-isolated converters
The non-isolating type of converter is generally used where the voltage needs
to be stepped up or down by a relatively small ratio (say less than 4:1), and there is no
problem with the output and input having no dielectric isolation. Examples are
24V/12V voltage reducers, 5V/3V reducers and 1.5V/5V step-up converters. There
are five main types of converter in this non-isolating group.
i) Buck converters
ii) Boost converters
iii) Buck-boost converters
iv) Cuk converters
v) Charge-Pump converter
The buck converter is used for voltage step-down/reduction, while the boost
converter is used for voltage step-up. The buck-boost and Cuk converters can be used
for either step-down or step-up, but are essentially voltage polarity reversers or
‘inverters’ as well. (The Cuk converter is named after its originator, Slobodan Cuk of
Cal Tech university in California.) The charge-pump converter is used for either
voltage step-up or voltage inversion, but only in relatively low power applications.
2.1.1 Buck converter:
The basic circuit configuration used in the buck converter is shown in Fig.1.
As you can see there are only four main components: switching power MOSFET Q1,
flywheel diode D1, inductor L and output filter capacitor C1. A control circuit (often a
single IC) monitors the output voltage, and maintains it at the desired level by
switching Q1 on and off at a fixed rate (the converter’s operating frequency), but with
a varying duty cycle.
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Wide Input Wide Output Dc-Dc Converter
Fig.2.1.1: The basic circuit for a Buck type of DC-DC converter
When Q1 is turned on, current begins flowing from the input source through
Q1 and L, and then into C1 and the load. The magnetic field in L therefore builds up,
storing energy in the inductor - with the voltage drop across L opposing or ‘bucking’
part of the input voltage. Then when Q1 is turned off, the inductor opposes any drop
in current by suddenly reversing its EMF, and now supplies current to the load itself
via D1.
The DC output voltage which appears across the load is a fraction of the input
voltage, and this fraction turns out to be equal to the duty cycle. So we can write:
Vout/Vin = D,
or Vout = Vin x D
where D is the duty cycle, and equal to Ton/T, where T is the inverse of the
operating frequency.
So by varying the switching duty cycle, the buck Converter’s output voltage
can be varied as a fraction of the input voltage. A duty cycle of 50% gives a step
down ratio of 2:1, for example, as needed for a 24/12V step-down converter. The
current ratio between output and input will be the reciprocal of the voltage ratio;
ignoring losses for a moment, and assuming our converter is perfectly efficient. So
Iout/In = Vin/Vout
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Wide Input Wide Output Dc-Dc Converter
So when we are stepping down the voltage by 2:1, the input current is only
half the value of the output current. Or it would be, if it were not for the converter’s
losses. Because real-world converters aren’t perfect the input current is typically at
least 10% higher than this.
2.1.2 Boost converter:
The basic boost converter is no more complicated than the buck converter, but
has the components arranged differently (Fig.2.1.2) in order to step up the voltage.
Again the operation consists of using Q1 as a high speed switch, with output voltage
control by varying the switching duty cycle. When Q1 is switched on, current flows
from the input source through L and Q1, and energy is stored in the inductor’s
magnetic field. There is no current through D1, and the load current is supplied by the
charge in C1. Then when Q1 is turned off, L opposes any drop in current by
immediately reversing its EMF - so that the inductor voltage adds to (i.e., ‘boosts’)
the source voltage, and current due to this boosted voltage now flows from the source
through L, D1 and the load, recharging C1 as well.
a
b
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Wide Input Wide Output Dc-Dc Converter
Fig.2.1.2: A non-ideal boost converter: (a) schematic, (b) inductor voltage and
capacitor current waveforms.
The output voltage is therefore higher than the input voltage, and it turns out that the
voltage step-up ratio is equal to:
Vout/Vin = 1/(1-D)
where 1-D is actually the proportion of the switching cycle that Q1 is off, rather than
on. So the step-up ratio is also equal to:
Vout/Vin = T/Toff
Again, if we assume that the converter is 100% efficient the ratio of output
current to input current is just the reciprocal of the voltage ratio:
Iin/Iout = Vout/Vin
So if we step up the voltage by a factor of 2, the input current will be twice the
output current. Of course in a real converter with losses, it will be higher
2.1.3 Buck-boost converter
The main components in a buck-boost converter are again much the same as
in the buck and boost types, but they are configured in a different way (Fig.2.1.3).
Fig.2.1.3: The Buck-Boost converter.
This allows the voltage to be stepped either up or down, depending on the
duty cycle. Here when MOSFET Q1 is turned on, inductor L is again connected
directly across the source voltage and current flows through it, storing energy in the
magnetic field. No current can flow through D1 to the load, because this time the
diode is connected so that it is reverse biased. Capacitor C1 must supply the load
current in this ‘Ton’ phase. But when Q1 is turned off, L is disconnected from the
source. Needless to say L again opposes any tendency for the current to drop, and
instantly reverses it’s EMF. This generates a voltage which forward biases D1, and
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Wide Input Wide Output Dc-Dc Converter
current flows into the load and to recharge C1. With this configuration the ratio
between the output and input voltages turns out to be:
Vout/Vin = -D/(1-D)
which again equates to
Vout/Vin = -Ton/Toff
So the buck-boost converter steps the voltage down when the duty cycle is
less than 50% (i.e., Ton < Toff), and steps it up when the duty cycle is greater than
50% (Ton > Toff). But the output voltage is always reversed in polarity with respect
to the input . so the buck-boost converter is also a voltage inverter.
When the duty cycle is exactly 50%, for example, Vout is essentially the same
as Vin, except with the opposite polarity. So even when it’s not being used to step the
voltage up or down, the buck-boost converter may be used to generate a negative
voltage rail in equipment operating from a single battery. As before, the ratio between
output and input currents is simply the reciprocal of the voltage ratio, if we ignore
losses.
2.1.4 CUK CONVERTER:
The basic circuit of a Cuk converter is shown in Fig.2.1.4, it has an additional
inductor and capacitor. The circuit configuration is in some ways like a combination
of the buck and boost converters, although like the buck-boost circuit it delivers an
inverted output. Virtually all of the output current must pass through C1, and as ripple
current so C1 is usually a large electrolytic with a high ripple current rating and low
ESR (equivalent series resistance), to minimize losses.
L1 C1 L2
Fig.2.1.4: The Cuk converter
When Q1 is turned on, current flows from the input source through L1 and Q1,
storing energy in L1’s magnetic field. Then when Q1 is turned off, the voltage across
L1 reverses to maintain current flow. As in the boost converter current then flows
from the input source, through L1 and D1, charging up C1 to a voltage somewhat
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Wide Input Wide Output Dc-Dc Converter
higher than Vin and transferring to it some of the energy that was stored in L 1. Then
when Q1 is turned on again, C1 discharges through via L2 into the load, with L2 and C2
acting as a smoothing filter. Meanwhile energy is being stored again in L1, ready for
the next cycle. As with the buck-boost converter, the ratio between the output voltage
and the input voltage again turns out to be:
Vout/Vin = -D/(1-D)
= -Ton/Toff
where the minus sign again indicates voltage inversion. So like the buck-boost
converter, the Cuk converter can step the voltage either up or down, depending on the
switching duty cycle. The main difference between the two is that because of the
series inductors at both input and output, the Cuk converter has much lower current
ripple in both circuits. In fact by careful adjustment of the inductor values, the ripple
in either input or output can be nulled completely.
2.1.5 Charge-pump converter
All of the converters we’ve looked at so far have depended for their
operation on storing energy in the magnetic field of an inductor. However there’s
another type of converter which operates by storing energy as electric charge in
a capacitor, instead. Converters of this type are usually called charge-pump
converters, and they’re a development from traditional voltage doubling and
‘voltage multiplying’ rectifier circuits.
The basic circuit for a voltage doubling charge-pump converter is shown
in Fig.2.1.5, and as you can see, it mainly uses four MOSFET switches and a
capacitor C1 — usually called the ‘charge bucket’ capacitor.
Operation is fairly simple. First Q1 and Q4 are turned on, connecting C1
across the input source and allowing it to charge to Vin. Then these switches
are turned off, and Q2 and Q3 are turned on instead. C1 is now connected in
series with the input voltage source, across output reservoir capacitor C2. As a
result some of the charge in C1 is transferred to C2, which charges to twice
the input voltage. This cycle is repeated at a fairly high frequency, with C2
providing the load current during the part of the cycle when Q2 and Q3 are
turned off.
As you can see all of the energy supplied to the load in this type of
converter flows through C1, and as ripple current. So again this capacitor needs
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Wide Input Wide Output Dc-Dc Converter
to have a relatively high value, have low ESR (to minimise losses) and be able to
cope with a heavy ripple current.
A slightly different circuit configuration from that shown in Fig.2.1.5 can
be used to deliver an inverted voltage of the same value as Vin, instead of a
doubled voltage. This type of converter finds use in generating a negative supply
rail for electronic circuits running from a single battery.
On the whole, though, the fact that charge-pump converters rely for their
operation on charge stored in a capacitor tends to limit them to relatively low
current applications. However for this type of operation they’re often cheaper
and more compact than inductor-type converters.
Fig 2.1.5: A basic Charge-Pump converter which doubles the input voltage.
2.2 ISOLATED CONVERTERS:
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Wide Input Wide Output Dc-Dc Converter
All of the converters above have virtually no electrical isolation between the
input and output circuits; in fact they share a common connection. This is fine for
many applications, but it can make these converters quite unsuitable for other
applications where the output needs to be completely isolated from the input. Here is
where a different type of inverter tends to be used - the isolating type. There are two
main types of isolating inverter in common use: the ‘flyback’ type and the ‘forward’
type. Like most of the non-isolating converters, both types depend for their operation
on energy stored in the magnetic field of an inductor; or in this case, a transformer.
2.2.1 Flyback converter:
The basic circuit for a flyback type converter is shown in Fig.2.2.1. In many
ways it operates like the buck-boost converter of Fig.2.1.3, but using a transformer to
store the energy instead of a single inductor.
Fig.2.2.1: The Flyback converter
When MOSFET Q1 is switched on, current flows from the source through
primary winding L1 and energy is stored in the transformer’s magnetic field. Then
when Q1 is turned off, the transformer tries to maintain the current flow through L1 by
suddenly reversing the voltage across it, generating a ‘flyback’ pulse of back-EMF.
Q1 is chosen to have a very high breakdown voltage, though, so current simply can’t
be maintained in the primary circuit. But because of transformer action an even
higher flyback pulse is induced in secondary winding L2.
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Wide Input Wide Output Dc-Dc Converter
And here diode D1 is able to conduct during the pulse, delivering current to
the load and recharging filter capacitor C1 (which provides load current between
pulses). So as you can see, the flyback converter again has two distinct phases in its
switching cycle. During the first phase Q1 conducts and energy is stored in the
transformer core via the primary winding L1. Then in the second phase when Q1 is
turned off, the stored energy is transferred into the load and C1 via secondary winding
L2. The ratio between output and input voltage of a flyback converter is not simply a
matter of the turns ratio between L2 and L1, because the back-EMF voltage in both
windings is determined by the amount of energy stored in the magnetic field, and
hence depends on the winding inductance, the length of time that Q1 is turned on, etc.
However the ratio between L2 and L1 certainly plays an important role, and most
flyback converters have a fairly high turns ratio to allow a high voltage step-up ratio.
Because of the way the flyback converter works, the magnetic flux in its
transformer core never reverses in polarity. As a result the core needs to be fairly
large for a given power level, to avoid magnetic saturation. Because of this flyback
converters tend to be used for relatively low power applications, like generating high
voltages for insulation testers, Geiger counter tubes, cathode ray tubes and similar
devices drawing relatively low current. Although it’s not shown in Fig.2.2.1, a third
small winding can be added to the flyback transformer to allow sensing of the flyback
pulse amplitude (which is reasonably close to the output voltage Vout). This voltage
can be then fed back to the MOSFET switching control circuit, to allow it to
automatically adjust the switching to regulate the output voltage.
2.2.2 Forward converter
In contrast with the flyback converter, where there are two distinct phases for
energy storage and delivery to the output, the forward converter uses the transformer
in a more traditional manner, to transfer the energy directly between input and output
in the one step. The most common type of forward converter is the push-pull type,
and the basic circuit for this type is shown in Fig.2.2.2.1.
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Wide Input Wide Output Dc-Dc Converter
Fig.2.2.2.1: The basic circuit for a Forward converter
Forward converter is another popular switched mode power supply (SMPS)
circuit that is used for producing isolated and controlled dc voltage from the
unregulated dc input supply. As in the case of fly-back converter the input dc supply
is often derived after rectifying (and little filtering) of the utility ac voltage. The
forward converter, when compared with the fly-back circuit, is generally more energy
efficient and is used for applications requiring little higher power output (in the range
of 100 watts to 200 watts). However the circuit topology, especially the output
filtering circuit is not as simple as in the fly-back converter.
Fig. shows the basic topology of the forward converter. It consists of a fast
switching device ‘S’ along with its control circuitry, a transformer with its primary
winding connected in series with switch ‘S’ to the input supply and a rectification and
filtering circuit for the transformer secondary winding. The load is connected across
the rectified output of the transformer-secondary.
The transformer used in the forward converter is desired to be an ideal
transformer with no leakage fluxes, zero magnetizing current and no losses. The basic
operation of the circuit is explained here assuming ideal circuit elements and later the
non-ideal characteristics of the devices are taken care of by suitable modification in
the circuit design. In fact, due to the presence of finite magnetizing current in a
EEE DEPT. HKBKCE 2012 Page 14
Wide Input Wide Output Dc-Dc Converter
practical transformer, a tertiary winding needs to be introduced in the transformer and
the circuit topology changes slightly.
2.3 Resonant Converters
Resonant converters use a resonant circuit for switching the transistors when
they are at the zero current or zero voltage point; this reduces the stress on the
switching transistors and the radio interference. We distinguish between ZVS- and
ZCS-resonant converters (ZVS: Zero Voltage Switching, ZCS: Zero Current
Switching). To control the output voltage, resonant converters are driven with
constant pulse duration at a variable frequency. The pulse duration is required to be
equal to half of the resonant period time for switching at the zero-crossing points of
current or voltage. There are many different types of resonant converters. For
example the resonant circuit can be placed at the primary or secondary side of the
transformer. Another alternative is that a serial r parallel resonant circuit can be used,
depending on whether it is required to turn off the transistor, when the current is zero
or the voltage is zero.
Future renewable energy systems will need to interface several energy sources
such as fuel cells, photovoltaic (PV) array with the load along with battery backup. A
three-port converter finds applications in such systems since it has advantages of
reduced conversion stages, high-frequency ac-link, multiwinding transformer in a
single core and centralized control. This has been described [1]. Some of the
applications are in fuel-cell systems, automobiles, and stand-alone self-sufficient
residential buildings.
A three-port bidirectional converter has been proposed in [2] for a fuel-cell
and battery system to improve its transient response and also ensure constant power
output from fuel-cell source. The circuit uses phase-shift control of three active
bridges connected through a three-winding transformer and a network of inductors.
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Fig. 2.3.1. System overview: a power electronic converter regulates the energy flow
between the fuel cell generator, an energy storage device, and the load.
To extend the soft-switching operation range in case of port voltage
variations, duty-ratio control is added in [3]. Another method to solve port voltage
variations is to use a front-end boost converter, as suggested in [3] for ultra-capacitor
applications. This topology comprises a high-frequency three-winding transformer
and three half-bridges, one of which is a boost half-bridge interfacing a power port
with a wide operating voltage. The three half-bridges are coupled by the transformer,
thereby providing galvanic isolation for all the power ports. The converter is
controlled by phase shift, which achieves the primary power flow control, in
combination with pulse width modulation (PWM).
Fig. 2.3.2. Three-port energy management system
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Because of the particular structure of the boost half-bridge, voltage variations
at the port can be compensated for by operating the boost half-bridge, together with
the other two half-bridges, at an appropriate duty cycle to keep a constant voltage
across the half-bridge. The resulting waveforms applied to the transformer windings
are asymmetrical due to the automatic volt-seconds balancing of the half-bridges.
With the PWM control it is possible to reduce the rms loss and to extend the zero-
voltage switching operating range to the entire phase shift region.
To increase the power-handling capacity of the converter, three-phase version
of the converter is proposed in [5]. A high-power converter to interface batteries and
ultracapacitors to a high voltage dc bus has been demonstrated in [6] using half
bridges, a battery and an ultracapacitor. The converter consists of three half-bridges
and a high-frequency multi-winding transformer as shown below.
Fig:2.3.3. multiple-input ZVS bi-directional dc–dc converter.
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Bi-directional power flow between input and output is achieved by adjusting
the phase-shift angles of the voltages across the two sides of the transformer. Soft-
switching is implemented naturally by snubber capacitors and transformer leakage
inductance.
Since the power flow between ports is inversely proportional to the impedance
offered by the leakage inductance and the external inductance, impedance has to be
low at high power levels. To get realizable inductance values equal to or more than
the leakage inductance of the transformer, the switching frequency has to be reduced.
Hence, the selection of switching frequency is not independent of the value of
inductance. A series-resonant converter has more freedom in choosing realizable
inductance values and the switching frequency, independent of each other. Such a
converter can operate at higher switching frequencies for medium and high-power
converters.
A three-port series resonant converter operating at constant switching
frequency and retaining all the advantages of a three-port structure is proposed in [1].
Other circuit topologies [7]–[12] are suggested in the literature for a three-port
converter such as the current-fed topologies [11] that have more number of magnetic
components and flyback converter topologies [12] that are not bidirectional. A
constant-frequency phase-controlled parallel-resonant converter was proposed in [13],
which uses phase shift between input bridges to control the ac-link bus voltage, and
also between input and output bridge to control the output dc voltage. Such high-
frequency ac-link systems using resonant converters have been extensively explored
for space applications and telecommunications applications. The series-resonant
three-port converter proposed in this paper uses a similar phase shift control but
between two different sources. The phase shifts can be both positive and negative,
and are extended to all bridges, including the load-side bridge along with
bidirectional power flow. A resonant converter topology is suggested in [16] but
phase-shift control is not utilized for control of power flow; instead, the converters
are operated separately based on the power flow direction.
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2.4 ENERGY RECOVERY SNUBBER
Snubbers are an essential part of power electronics. Snubbers are small
networks of parts in the switching circuit whose function is to control the effects of
circuit reactances.
Snubbers enhance the performance of the switching circuits and result in
higher reliability, higher efficiency, higher switching frequency, smaller size, lower
weight and lower EMI. The basic intent of a snubber is to absorb energy from the
reactive elements in the circuit. The benefits of this may include circuit damping,
controlling the rate of change of voltage or current or clamping voltage over shoot.
In performing these functions a snubber limits the amount of stress which the
switch must endure and this increases the reliability of the switch. When a snubber is
properly designed and implemented the switch will have lower average power
dissipation much lower peak power dissipation, lower peak operating voltage and
lower peak operating current.
Since the WIWO operates a coupled inductor, the energy stored in the
leakage inductances becomes a problem to deal with. Besides increased switching
losses, discharge of the leak- age inductance energy causes oscillations and increased
voltage spikes across the switches. The resulting voltage stress becomes intolerable
at higher voltages and higher power. If not snubbed, overvoltage breakdown of the
MOSFET devices may occur.
The proposed lossless snubber is comprised of a snubber capacitor CS and a
pair of fast diodes DS1 and DS2 . The snubber is fitted to WIWO, as shown in Fig
5.7.5. The snubber is effective both in buck mode and in boost mode.
Detailed description of the snubber operation is out of scope of this paper; in
brief, however, the operation is as follows. With WIWO in the buck mode, at the
instant when the S2 switch is turned off, the snubber diode DS1 conducts L1 leakage
current and allows the stored energy to be discharged into the snubber capacitance
CS and to the output of the circuit. This takes one- half resonant cycle dictated by
the leakage inductance and the snubber capacitance. CS will remain charged until
the S2 switch is turned ON again at the onset of the subsequent switching cycle. With
S2 turned ON, the energy stored in the snubber capacitor is discharged into L2
winding via DS2 and recycled.
In the boost mode, the snubber operation is similar. However, here, S1
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Wide Input Wide Output Dc-Dc Converter
interrupts the current and is subject to the voltage spike while S2 switch is constantly
ON with zero voltage VD S2 across.
Fig 2.4 Energy recovering snubber for WIWO power stage.
EEE DEPT. HKBKCE 2012 Page 20
DC Input
DC-DC Converter
MOSFET Driver/ Pulse
Driver
PIC / PWM Controller
Supply
Load
Wide Input Wide Output Dc-Dc Converter
Chapter3. BLOCK DIAGRAM
3.1 Motivation in the search for new switching topology
Fig 3.1.0 Block diagram
Fig 3.1.1 Buck-derived converters with tapped inductors
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Fig 3.1.2 Voltage conversion ratio of buck-derived converters with tapped
inductors. (a) 0 < n < ∞. (b) 0 < n < ∞. (c) n > 1. (d) n > 1.
Fig 3.1.3 Boost-derived converters with tapped
inductors.
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Fig 3.1.4 Voltage conversion ratio of boost-derived converters with tapped
inductors. (a) 0 < n < ∞. (b) 0 < n < ∞. (c) 0 < n < 1. (d) 0 < n < 1.
The basic buck and boost converters can be transformed into a number of new
topologies by bringing in the tapped inductor. The proposed tapped-inductor buck-
derived converters are shown in Fig 3.1.1, with their corresponding voltage
conversion ratios plotted in Fig 3.1.2. The proposed tapped-inductor boost- derived
topologies and their corresponding voltage conversion ratios are given in Figs. 3.1.3
and 3.1.4. Here, D is the duty ratio of switch S, M is the voltage conversion ratio,
and n is the turn ratio of the tapped inductors, which is defined as n = n2 : n1 . As
the turn ratio n tends to infinity, the conversion ratio of the buck-derived converters
approach the characteristic of a basic buck topology. On the other hand, as the turn
ratio n goes to zero, the conversion ratio of the boost-derived converters approach the
characteristic of a basic boost topology. Inspection of the conversion ratio plots, as
given in Fig. 3.1.1(a), reveals that the proposed buck-derived converter achieves wider
voltage step- down than a basic buck converter. Also, by examining Fig 3.1.3(a), it
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becomes evident that the suggested boost-derived converter attains a wider voltage
step-up than a basic boost converter.
The converter topologies shown in Figs. 3.1.1(a) and 3.1.3(a) are strikingly
similar. The idea proposed here is that these two topologies may be combined to
form a new two-switch topology, with an extended conversion range. Same
conclusion can be reached comparing the converters given in Figs. 3.1.1(c) and 3.1.3(c).
The proposed WIWO range converter topology is described in the next section.
3.2. WIWO DC–DC CONVERTER TOPOLOGY
Fig. 3.2. WIWO dc–dc converter topology.
3.2.1 Proposed WIWO DC–DC Converter Topology
The proposed WIWO dc–dc converter is illustrated in Fig. 3.2. The converter is
comprised of two active switches S1 and S2, tapped inductors L1 and L2 with turns
ratio n = n2 : n1 , diode D, and capacitive output filter C.
Specifically, note that the tapped inductor in Figs. 3.1.1 and 3.1.3 is
reconfigured into a pair of coupled inductors in Fig. 3.2.1. Being equivalent
electrically, this reconfiguration is beneficial from a practical point of view. In Fig.
3.2, S1 and S2 are connected to a common junction or midpoint. The midpoint is
periodically switched by S1 to ground, which allows recharging the bootstrap power
supply and reliable operation of the flying driver of the top switch S2.
Consequently, a standard half-bridge driver chip can be used with the low-side driver
operating the bottom switch S1 and the bootstrap high-side driver activating the top
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switch S2.
WIWO can operate either in the step-down or the buck mode or in the step-up
or the boost mode. To operate the WIWO in the buck mode, the switch S1 is
assigned a high-frequency switching signal with a predetermined duty cycle D,
whereas S2 is switched complementarily to S1. The diode D is kept ON by the
inductor L2 current, which is assumed to be continuous. To operate WIWO in the
boost mode, the controller keeps S2 switch continuously ON and issues the
required duty cycle signal for the S1 switch. Thus, the diode D is forced to switch
on and off complementarily to S1.In both modes, the capacitor C filters the
pulsating current and provides a smoothed output voltage for the load R.
3.3 Control Scheme
Fig. 3.3. Proposed WIWO dc–dc converter and PWM control circuitry.
For the proper operation of WIWO, a modified PWM control circuitry is
required. The implementation is not unique. One possible realization of the
modulator is shown in Fig. 3.3. Here, a window comparator is employed to derive
the required switching signals for S1 and S2 by comparing the sawtooth ramp with
amplitude of Vm to the two control voltages VC and V . The control voltage VC
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for the upper comparator is delivered by an external source, whereas the lower
comparator input signal is derived by the PWM circuitry, downshifting the control
voltage VC by Vm: V = VC−Vm. The relationship between the control voltage VC
and the sawtooth ramp amplitude Vm can be expressed by means of a variable m
as VC=mVm . WIWO operates in the buck mode when 0 <VC <Vm, i.e., when
0 ≤ m < 1. Here, the upper comparator generates the required duty cycle for the
S2 switch, whereas the lower comparator is in “1” state and commands the NAND
gate to provide the complimentary duty cycle for the S1 switch. Therefore, WIWO
operates similarly to a synchronous buck converter. On the other hand, for
Vm<VC<2Vm ,or 1≤m<2, the upper comparator is in “1” state and keeps S2
continuously ON, whereas the lower comparator and the NAND gate provide the
required duty cycle for the S1 switch. Thus, the converter enters the boost mode.
3.4 Operating Principle of the WIWO Converter
In the following, the steady-state operation of the proposed WIWO converter
is described. The analysis is performed assuming that the circuit is comprised of
ideal components. The coupling coefficient of the tapped inductor is assumed to be
unity. Under continuous inductor current (CCM) condition, the proposed WIWO
converter exhibits four topological states, as shown in Fig3.4.1. Here, the large
output filter capacitor is replaced by an ideal voltage source. The waveforms and
timing of WIWO for both buck and boost modes are illustrated in Fig3.4.2.
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Fig3.4 .1 Four topological states of the WIWO converter. (a) Buck-mode
charging state. (b) Buck-mode discharging state. (c) Boost-mode charging state.
(d) Boost-mode discharging state.
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Fig 3.4.2. Waveforms of the WIWO dc–dc converter. (a) Buck mode. (b) Boost
mode.
Buck Mode: State 1 (t0 ≤ t < t1 ) is the buck-mode charging state [see
Figs. 3 .4.1(a) and 3 .4 .2 (a)]. Here, the switch S2 is turned on and S1 is turned off.
The diode D conducts and the coupled inductors L1 and L2 are charged. The energy
is also transferred from dc source to load. State 2 (t1 ≤ t ≤ t2 ) is the buck-mode
discharging state [see Figs3.4.1(b) and 3.4.2(a)]. Here, the switch S2 is turned off also
cutting off the current in the L1 winding, whereas S1 is turned on and the diode D
conducts L2 current to the load.
Boost Mode: State 3 (t0 ≤ t < t1
) is the boost-mode charging state [see
Figs. 3.4.1(c) and 3.4.2(b)]. Here, the switches S1 and S2 are turned on charging the
L1 inductor. The diode D is cut off by the negative voltage induced inL2 winding.
The output voltage is supported by the capacitor C. State 4 (t1 ≤ t ≤ t2 ) is the
boost-mode discharging state [see Figs. 3.4.1(d) and 3.4.2(b)]. Here, the switch S2 is
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still ON whereas S1 is turned off. Both windings L1 and L2 conduct through the
diode D and discharge the stored energy to the output.
3.5 Steady-State Analysis
The steady-state models of the proposed WIWO converter are shown in Fig
3.4.1. These models preserve the tapped-inductor symbol. More suitable for analysis
purposes, however, are the models of Fig 3.5. Here, the role of the magnetizing
inductance Lm is clearly shown. The detailed analysis was carried out in [11]
using state-space averaging technique. WIWO voltage conversion ratio, output
voltage ripple, voltage stresses, etc., were obtained. The characteristics of WIWO
are summarized in Table I(appendix) for a general case of n and separately for the
special case of n = 1.
Fig. 3.5. Switched circuit models. (a) State 1. (b) State 2. (c) State 3. (d) State 4.
WIWO voltage transfer characteristics M (n, m) are plotted in Fig3.6.1.
Clearly, for n=1, the voltage transfer ratio is smooth at the vicinity of the buck to
boost switchover point m=1, whereas for other values of n, the curves exhibit a
slope change. This statement can be verified analytically by calculating the
derivatives of M (m) at m = 1. Using the expressions for voltage conversion ratio
given in Table I, the result is ((n + 1)/n) for the buck mode and (n + 1) for the
boost mode. Obviously, the slope of WIWO dc–dc characteristic becomes
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continuous for n = 1. Table I also presents the line-to-output and control-to-output
transfer functions. The small-signal transfer functions of the WIWO converter
were derived by linearizing the state-space equations around the operating point
[11]. The line-to-output and control-to-output transfer functions reveal strong
dependence on the operating point and a right-half-plane (RHP) zero. This is
also the case in other tapped-inductor t opo log i e s [13], [14]. These
characteristics make the WIWO compensation network design somewhat difficult.
3.6. RIPPLE AND EFFECTS OF RIPPLE
The most common meaning of ripple in electrical science is the small
unwanted residual periodic variation of the direct current (dc) output of a power
supply which has been derived from an alternating current (ac) source. This ripple is
due to incomplete suppression of the alternating waveform within the power supply.
As well as this time-varying phenomenon, there is a frequency domain ripple
that arises in some classes of filter and other signal processing networks. In this case
the periodic variation is a variation in the insertion loss of the network against
increasing frequency. The variation may not be strictly linearly periodic. In this
meaning also, ripple is usually to be considered an unwanted effect, its existence
being a compromise between the amount of ripple and other design parameters.
3.6.1 Time-domain ripple
Fig 3.6.1.1: Full-wave rectifier with a smoothing capacitor
Ripple factor (γ) may be defined as the ratio of the root mean square (rms)
value of the ripple voltage to the absolute value of the dc component of the output
voltage, usually expressed as a percentage. However, ripple voltage is also commonly
expressed as the peak-to-peak value. This is largely because peak-to-peak is both
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easier to measure on an oscilloscope and is simpler to calculate theoretically. Filter
circuits intended for the reduction of ripple are usually called smoothing circuits.
The simplest scenario in ac to dc conversion is a rectifier without any
smoothing circuitry at all. The ripple voltage is very large in this situation; the peak-
to-peak ripple voltage is equal to the peak ac voltage. A more common arrangement
is to allow the rectifier to work into a large smoothing capacitor which acts as a
reservoir. After a peak in output voltage the capacitor (C) supplies the current to the
load (R) and continues to do so until the capacitor voltage has fallen to the value of
the now rising next half-cycle of rectified voltage. At that point the rectifiers turn on
again and deliver current to the reservoir until peak voltage is again reached. If the
time constant, CR, is large in comparison to the period of the ac waveform, then a
reasonably accurate approximation can be made by assuming that the capacitor
voltage falls linearly. A further useful assumption can be made if the ripple is small
compared to the dc voltage. In this case the phase angle through which the rectifiers
conduct will be small and it can be assumed that the capacitor is discharging all the
way from one peak to the next with little loss of accuracy.
Fig 3.6.1.2: smoothed Ripple voltage from a full-wave rectifier
With the above assumptions the peak-to-peak ripple voltage can be calculated
as:
For a full-wave rectifier:
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For a half-wave rectification:
where
Vpp is the peak-to-peak ripple voltage and I is the current in the circuit
F is the frequency of the ac power and C is the capacitance
For the rms value of the ripple voltage, the calculation is more involved as the
shape of the ripple waveform has a bearing on the result. Assuming a sawtooth
waveform is a similar assumption to the ones above and yields the result
where
γ is the ripple factor and R is the resistance of the load
Another approach to reducing ripple is to use a series choke. A choke has a
filtering action and consequently produces a smoother waveform with less high-order
harmonics. Against this, the dc output is close to the average input voltage as opposed
to the higher voltage with the reservoir capacitor which is close to the peak input
voltage. With suitable approximations, the ripple factor is given by
Where ω is the angular frequency 2πf and L is the inductance of the choke
More complex arrangements are possible; the filter can be an LC ladder rather
than a simple choke or the filter and the reservoir capacitor can both be used to gain
the benefits of both. The most commonly seen of these is a low-pass Π-filter
consisting of a reservoir capacitor followed by a series choke followed by a further
shunt capacitor. However, use of chokes is deprecated in contemporary designs for
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economic reasons. A more common solution where good ripple rejection is required
is to use a reservoir capacitor to reduce the ripple to something manageable and then
pass through a voltage regulator circuit. The regulator circuit, as well as regulating
the output, will incidentally filter out nearly all of the ripple as long as the minimum
level of the ripple waveform does not go below the voltage being regulated to.
The majority of power supplies are now switched mode. The filtering
requirements for such power supplies are much easier to meet owing to the frequency
of the ripple waveform being very high. In traditional power supply designs the ripple
frequency is either equal to (half-wave), or twice (full-wave) the ac line frequency.
With switched mode power supplies the ripple frequency is not related to the line
frequency, but is instead related to the frequency of the chopper circuit.
3.6.2 Effects of ripple
Ripple is undesirable in many electronic applications for a variety of
reasons,the ripple frequency and its harmonics are within the audio band and will
therefore be audible on equipment such as radio receivers, equipment for playing
recordings and professional studio equipment.
The ripple frequency is within television video bandwidth. Analogue TV
receivers will exhibit a pattern of moving wavy lines if too much ripple is present.
The presence of ripple can reduce the resolution of electronic test and
measurement instruments. On an oscilloscope it will manifest itself as a visible
pattern on screen.
Within digital circuits, it reduces the threshold, as does any form of supply rail
noise, at which logic circuits give incorrect outputs and data is corrupted. High
amplitude ripple currents shorten the life of electrolytic capacitors.
3.7 PULSE WIDTH MODULATION( PWM)
Pulse Width Modulation, or PWM, is a technique for getting analog results with digital means. Digital control is used to create a square wave, a signal switched between on and off. This on-off pattern can simulate voltages in between full on (5 Volts) and off (0 Volts) by changing the portion of the time the signal spends on versus the time that the signal spends off. The duration of "on time" is called the
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pulse width. To get varying analog values, you change, or modulate, that pulse width. If you repeat this on-off pattern fast enough with an LED for example, the result is as if the signal is a steady voltage between 0 and 5v controlling the brightness of the LED.
In the graphic below, the green lines represent a regular time period. This duration or period is the inverse of the PWM frequency. In other words, with PWM frequency at about 500Hz, the green lines would measure 2 milliseconds each. A call to analogWrite is on a scale of 0 - 255, such that analogWrite(255) requests a 100% duty cycle (always on), and analogWrite(127) is a 50% duty cycle (on half the time) for example.
A 100-W prototype WIWO converter was designed for input voltage range of
12–48 Vdc and a constant output voltage of 28 Vdc . The turn ratio of the tapped
inductor was set to n =1 with a total inductance of 400 µH. The switching
frequency of 200 kHz was chosen. The tapped inductors were wound on C058548A2
toroidal powder core, chosen for its low leakage, with 50 turns of AWG20 wire for
both windings. The design yielded 400 µH inductance with only 560 nH leakage
inductance. Two FDD2572 MOSFETs were paralleled to comprise the top switch
and two IRFR3518 were used for the low switch providing low Rds−ON and low
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gate capacitance. Schottky diode 20CTQ150 was selected due to superior reverse
recovery characteristics.
Fig.3.6.1. Voltage transfer characteristics M (n, m) of the WIWO dc–dc
converter.
Experimental waveforms of WIWO converting 48 V input to 28 V output
(buck mode) are shown in Fig.3.6.2. In the buck mode, S2 is the leading switch,
gated by the duty cycle command shown as the bottom trace in Fig. 3.6.2, whereas
the bottom switch S1 is switched complementarily, similarly to a synchronous
converter. Switch voltages (see Fig.3.3 for definition) are shown as top two
waveforms in Fig.3.6.2. The middle traces show the winding currents. These were
measured by ac probe, so only the ripple components could be observed. As could be
seen, as the S2 switch conducts, both windings carry the same current. At the S2 is
turned off, the input current ceases whereas the output current is doubled in
amplitude, consistent with WIWO models in Fig.3.5(a) and (b). The ramp portion
of the current is hardly noticeable due to the relatively high frequency and
sufficiently large inductance value. The leakage inductance of L1 developed a turn-
OFF voltage spike across S1 that is smoothed by the snubber circuitry. The snubber is
used to clamp the voltage spike, as described later.
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Fig.3.6.2. Experimental waveforms of the WIWO converter in the buck mode.
Top trace: drain voltage V1 of S1 switch (50 V/division, 2 µs/division); second
top trace: drain voltage V2 of S2 switch (50 V/division, 2 µs/division); middle
trace: input current Ii (0.2 A/division, 2 µs/division); second bottom trace: output
current Io (0.2 A/division, 2 µs/division); bottom trace: S2 switch gating voltage
(20 V/ division, 2 µs/division).
The experimental waveforms of WIWO in the boost mode with 12 V input
and 28 V output, under full-load condition, are shown in Fig 3.6.3. To supply the
power requirements of the load at lower input voltage range, WIWO calls for greater
input current, and therefore, turn-OFF voltage spike on S1 is observed. In the boost
mode, the S1 switch is the leading switch that is issued the duty cycle command,
shown as the bottom trace in Fig.3.6.3. Since in the buck mode the S2 switch is
constantly ON, the drain voltage of S2 and the drain voltage of S1 are almost
identical. The winding currents were measured by a high-frequency ac probe, and
therefore, only ac current components are shown as two middle traces in Fig.3.6.3.
As S1 switch conducts, the input winding carries the input current and is charging,
whereas the output current is cut off. As the S1 switch
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Fig.3.6.3 Experimental waveforms of WIWO in the boost mode (see Fig.3.3 for
designation of variables). Top trace: drain voltage of S1 switch (20 V/division,2
µs/division); second top trace: drain voltage of S2 switch (20 V/division,2
µs/division); middle trace: input current Ii (0.5 A/division, 2 µs/division); second
bottom trace: output current Io (0.5 A/division, 2 µs/division); bottom trace: S1
switch gating voltage (20 V/division, 2 µs/division).
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Fig.3.6.4 Comparison of K with Kcr i t for n =
1.
is cut off, both windings carry the same current and are discharging into the output
capacitor and feeding the load. For this reason, the currents ripple components appear
in antiphase, as predicted by WIWO models in Fig. 3.5(c) and (d). Also could be
seen is the snubber circuit resonant discharge as the snubber recycles the stored energy.
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Fig 3.6.5.Comparison of the experimental and theoretical voltage conversion ratio
under different loading conditions. (a) K = 2. (b) K = 0.2. (c) K = 0.02.
With decreased load, the converter enters the discontinuous conduction mode
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(DCM). To measure the tendency of the converter to operate in DCM, the parameter
K = (2Lm /RTs ) is defined as suggested in [15]. The critical value of K for n = 1 is
compared with K = 2, 0.2, 0.02 in Fig.3.6.4. The experimental voltage conversion
ratio M as function of m for different values of K plotted on top of the theoretical
curve is given in Fig.3.6.5 (a)–(c). Due to the parasitic resistances in the circuit, the
experimental voltage conversion ratio M is slightly lower than theoretical prediction.
For very same reason, the experimental M cannot become infinite and drops as m
approaches 2. A narrow buck to boost-mode transition can be observed on the WIWO
characteristic in the vicinity of m = 1. The conversion ratio in DCM is higher than that
in CCM, as shown in Fig. 3.6.5(b) and (c).
The efficiency of the experimental WIWO dc–dc converter for different dc
input voltages versus the load current is plotted in Fig.3.7.1. The output voltage was
kept at the nominal value of 28 Vdc. No attempt was made to optimize the
preliminary design, still the converter demonstrated high efficiency.
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CHAPTER 4. HARDWARE ANDSOFTWARE IMPLEMENTATION
4.1 Introduction to Matlab
4.1.1 SimPowerSystems
SimPowerSystems and SimMechanics of the Physical Modeling product family
work together with Simulink to model electrical, mechanical and control systems.
4.1.2 Role of Simulation in Design
Electrical power systems are combinations of electrical and electromechanical
devices like motors and generators. Engineers working in this discipline are constantly
improving the performance of the systems. Requirements for drastically improved
efficiency have forced power system designers to use power electronic devices and
sophisticate control system concepts that tax traditional analysis tools and techniques.
Further complicating the analyst’s role is the fact that the system is often so nonlinear
that the only way to understand it is through simulation.
Land based power generation from hydroelectric, steam or other devices are not
the only use of power systems. A common attribute of these systems is their use of
power electronics and control systems to achieve their performance objectives.
SimPowerSystems is a modern tool that allows scientists and engineers to
rapidly and easily build models that simulate power systems. SimPowerSystems uses
the Simulink environment, allowing to build a model using simple click and drag
procedures. Not only can draw the circuit topology rapidly, but analysis of the circuit
can include its interactions with mechanical, thermal, control, and other disciplines.
This is possible because all the electrical parts of the simulation interact with the
extensive Simulink modeling library. Since Simulink uses MATLAB as its
computational engine, designers can also use MATLAB toolboxes and Simulink block
sets. SimPowerSystems and SimMechanics share a special Physical Modeling block
and connection line interface.
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4.1.3 SimPowerSystems Libraries
SimPowerSystems can be made to work rapidly. The libraries contain models of
typical power equipment such as transformers, lines, machines and power electronics.
These models are proven ones coming from textbooks and their validity is based on the
experience of the Power Systems testing and Simulation Laboratory of Hydro Quebec, a
large North American utility located in Canada and also on the experience of Ecole de
Technologies superieure and Universite Laval.
The capabilities of SimPowerSystems for modeling typical electrical systems are
illustrated in demonstration files. And for users who want to refresh their knowledge of
power system theory, there are also self-learning case studies.
4.2 Design and Simulating of a Simple Circuit
SimPower Systems allows building and simulating of electrical circuits
containing linear and nonlinear elements.
In this section it is possible to
1 Explore the powerlib library of SimPowerSystems
2 Learn how to build a simple circuit from the powerlib library
3 Interconnect Simulink blocks with your circuit
This section contains discussion of the following topics:
1 Building the Electrical Circuit with powerlib Library
2 Interfacing the Electrical Circuit with Simulink
3 Measuring Voltages and Currents
4 Basic Principles of Connecting Capacitors and Inductors
5 Using the Powerlib Block to Simulate SimPowerSystems Models
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4.3 Designing a Electrical Circuit with Powerlib Library
The graphical user interface makes use of the Simulink functionality to
interconnect various electrical components. The electrical components are grouped in a
special library called powerlib.
SimPowerSystems library is opened by entering the following command at the
MATLAB prompt.
Powerlib
This command displays a Simulink window showing icons of different block libraries.
Fig 4.1 Powerlib Library
It is possible to open these libraries to produce the windows containing the
blocks to be copied into given circuit. Each component is represented by a special icon
having one or several inputs and outputs corresponding to the different terminals of the
component.
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4.4 Interfacing Electrical Circuit with Simulink
The Voltage Measurement block acts as an interface between the
SimPowerSystems blocks and the Simulink blocks. The Voltage Measurement block
converts the measured voltages into Simulink signals.The Current Measurement block
from the Measurements library of powerlib can also be used to convert any measured
current into a Simulink signal.
It is also possible to interface from Simulink blocks to the electrical system. For
example, it is possible to use the Controlled Voltage Source block to inject a voltage in
an electrical circuit, as shown in the following figure.
Fig 4.2 Interfacing electrical circuit with simulink
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4.5 Measuring Voltages and Currents
To measure a current using a Current Measurement block, the positive direction
of current is indicated on the block icon (positive current flowing from positive terminal
to negative terminal). Similarly, when to measure a voltage using a Voltage
Measurement block, the measured voltage is the voltage of the +ve terminal with
respect to the -ve terminal. However, when voltages and currents of blocks from the
Elements library are measured using the Multi-meter block, the voltage and current
polarities are not immediately obvious because blocks might have been rotated and
there are no signs indicating polarities on the block icons. Unlike Simulink signal lines
and input and output ports, the Physical Modeling connection lines and terminal ports of
SimPowerSystems lack intrinsic directionality. The voltage and current polarities are
determined, not by line direction, but instead by block orientation.
4.5.1 Power Supply
Single phase AC supply is given to bridge rectifier. It converts AC into DC. The
present chapter introduces the operation of power supply circuits built using filters,
rectifiers, and then voltage regulators. Starting with an AC voltage, a steady DC voltage
is obtained by rectifying the AC voltage, then filtering to a DC level, and finally,
regulating to obtain a desired fixed DC voltage.
4.5.2 Converter
Converter is a device which convert AC to DC since high voltage dc supply is
required at the input of the inverter.
4.5.3 Filtering Unit
Filter circuits which is usually a capacitor acting as a surge arrester always
follows the rectifier unit. This capacitor is also called as a decoupling capacitor or a
bypassing capacitor. It passes only low frequency signals and bypasses high frequency
signals.
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4.5.4 Driver Circuit
The function of driver circuit is to amplify the voltage generated in the
microcontroller. In the above circuit the microcontroller generates 2v to 5v. This
voltage is not sufficient for driving the MOSFET. Therefore this voltage is amplified
using the driver circuit which will be discussed in detail in the next chapter.
4.5.5 Microcontroller
Microcontroller (AT89C51) is one of the most commonly used microcontrollers
especially in automotive, industrial appliances and consumer applications. However, as
the functionality of the components such as timers, A/D converters, I/O Ports are
explained in detail in Chapter 7.0.1 and the reader can flashback to this section to view
the schematics and the specifications.
4.6 POWER SUPPLY UNIT
All electronic circuits works only in low DC voltage, so a power supply unit
is required to provide the appropriate voltage supply for their proper functioning.
This unit consists of transformer, rectifier, filter and regulator. AC voltage of typically
230V RMS is connected to a transformer which step down the voltage to the desired
AC voltage. A diode rectifier that provides the full wave rectified voltage that is
initially filtered by a simple capacitor filter to produce a DC voltage. This resulting
DC voltage usually has some ripple or AC voltage variation. A regulator circuit can
use this DC input to provide DC voltage that not only has much less ripple voltage but
also remains at the same DC value, even when the input DC voltage varies somewhat or
the load connected to the output DC voltage changes.
Fig 4.3 General Block of Power Supply Unit
4.6.1 Transformer
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A transformer is a static device in which electric power in one circuit is
transferred into electric power of same frequency in another circuit .It can raise or
lower the voltage in the circuit, but with a corresponding decrease or increase in
current. It works with the principle of mutual induction. In this project a step-down
transformer is used to provide necessary supply of 12 V for the electronic circuits.
4.6.2 Rectifier
A DC level obtained from a sinusoidal input can be improved 100% using a
process called full wave rectification. Here in this project for full wave rectification a
bridge rectifier is used.
In the bridge rectifier the diodes may be of variable types like 1N4001, 1N4003,
1N4004, 1N4005, IN4007 etc can be used. But in this project 1N4007 is used because it
can withstand up to 1000V.
4.6.3 Filters
In order to obtain a dc voltage of 0 Hz, a low pass capacitive filter circuit is used
where a capacitor is connected at the rectifier output and a DC voltage without ripples is
obtained across it. The filtered waveform is essentially a DC voltage with negligible ripples
and it is ultimately fed to the load.
4.6.4 Regulators
The filtered output voltage from the capacitor is finally regulated. The voltage
regulator is a device, which maintains the output voltage constant irrespective of the change
in supply variations, load variations and temperature changes. Here a fixed voltage
regulator namely LM7805 is used. The IC LM7805 is a +5V regulator which is used for
microcontroller.
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Features and Description of Regulators
1. Output Current up to 1A
2. Output Voltages of 5, 6, 8, 9, 10, 12, 15, 18, 24V
3. Thermal Overload Protection
4. Short Circuit Protection
5. Output Transistor Safe Operating Area Protection
The KA78XX/KA78XXA series of three-terminal positive regulator are
available in the TO-220/D-PAK package and with several fixed output voltages,
making them useful in a wide range of applications. Each type employs internal
current limiting, thermal shutdown and safe operating area protection, making it
essentially indestructable. If adequate heat sinking is provided, they can deliver over
1A output current. Although designed primarily as fixed voltage regulators, these
devices can be used with external components to obtain adjustable voltages and
currents.
Electrical Characteristics of KA7805A
Load and line regulation are specified at constant junction temperature. Change
in Vo due to heating effects must be taken into account separately. Pulse testing with
low duty is used.
Electrical Characteristics of KA7805A
Load and line regulation are specified at constant junction temperature. Change
in Vo due to heating effects must be taken into account separately. Pulse testing with
low duty is used.
This circuit can give +5V output at about 150 mA current, but it can be
increased to 1 A when good cooling is added to 7805 regulator chip. The circuit has
over overload and thermal protection.
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Figure 4.4: Circuit diagram of the power supply.
The capacitors must have enough high voltage rating to safely handle the input
voltage feed to circuit. The circuit is very easy to build for example into a piece of
overboard.
Figure 4.5: Pin out of the 7805 regulator IC.
1. Unregulated voltage in
2. Ground
3. Regulated voltage out
Component list
1. 7805 regulator IC
2. 100 uF electrolytic capacitor, at least 25V voltage rating
3. 10 uF electrolytic capacitor, at least 6V voltage rating
4. 100 nF ceramic or polyester capacitor
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Figure 4.6: circuit diagram of power supply
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4.7 DRIVER CIRCUIT
The main purpose of driver circuit is to enhance the switching voltage for
the MOSFET or any switching device and also to isolate the power circuit from the
microcontroller circuit. Since the power circuit current must not enter into the
microcontroller circuit, MCT2E which is the opto coupler will be connected to the
buffer CD4050 which send pulse signals of 5V from microcontroller to the driver
circuit. MCT2E is the device which isolates the power circuit with the
microcontroller circuit. After it gets the signal from the microcontroller it will get
enhanced using the 2N2222 transistor to higher level of voltage.After this voltage
gets regulated by the use of darlington pair. The darlington is made of 2N2222 (NPN)
and SK100 (PNP) transistor.330 OHM
MCT2E
1 K22 K
100 OHM
100 OHM
100 OHM
1 K
1000 mF/25 A
G
GROUND
330 OHM
MCT2E
1 K22 K
100 OHM
100 OHM
100 OHM
1 K
1000 mF/25 A
G
GROUND
330 OHM
MCT2E
1 K22 K
100 OHM
100 OHM
100 OHM
1 K
1000 mF/25 A
G
GROUND
330 OHM
MCT2E
1 K22 K
100 OHM
100 OHM
100 OHM
1 K
1000 mF/25 A
G
GROUND
330 OHM
MCT2E
22 K
100 OHM
1 K
100 OHM
1 K
100 OHM
100 OHM
G
GROUND
100 OHM
GROUND
1000 mF/25 A
G
330 OHM
100 OHM
1000 mF/25 A
1 K
22 K
MCT2E
1 K
Fig 4.7: Circuit diagram of driver circuit unit
Components used:
1. MOSFET
IRFP460
2. Diode
1N4007
3. Capacitors
1000µF/50V
1000µF/25V
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4. Opto coupler
MCT2E
5. Transistors
2N2222
SK100
6. Resistors
1K
100Ω
7. Transformers
230V/12V
4.8 Opto coupler
4.8.1Description
Opto coupler or opto isolator is a combination of light source and light detector
in the same package. They are used to couple signal from one point to the other
optically, by providing a complete electrical isolation between them. This kind of
isolation is provided between a low power control circuit and high power output
circuit, to protect the control circuit.
Fig 4.8 Opto coupler schematic diagram
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Depending on the type of light sources and detector used it is possible to get a
variety of optocouplers, they are as follows:
1. LED LDR optocoupler.
2. LED photodiode optocoupler.
3. LED phototransistor optocoupler.
4.8.2 Characteristics
1. Current transfer ratio (CTR).
2. Isolation voltage.
3. Response time.
4. Common mode rejection.
4.8.3 LED Phototransistor Opto coupler
The LED phototransistor opto coupler is an infrared LED acts as the high
source and the phototransistor acts as a photo detector. This is the most popularly used
optocoupler, because it does not need any additional amplification. When the pulse at
the input goes high, the LED turns ON.
The light emitted by the LED is focused on the CB junction of the
phototransistor. In response to this light, photo current starts flowing which acts as
base current for the phototransistor. The collector current of phototransistor starts
flowing. As soon as the input pulse reduces to zero the LED turns OFF and the
collector current of phototransistor reduces to zero.
4.8.4 Applications
1. AC to DC converters used for DC motor speed control.
2. High power choppers.
3. High power inverters.
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4.9 MOS Transistors
1. Two primary types:
a. MOSFET (Metal-Oxide-Semiconductor FET).
b. JFET( Junction FET)
2. MOS transistors can be:
a. N-Channel
i. Enhancement mode
ii. Depletion mode
b. P-Channel
i. Enhancement mode
ii. Depletion mode
Fig 4.9 Diagram of MOSFET
MOSFET
Fig 4.9 Symbolic representation of MOSFET
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5.0.Pulse Generator
5.0.1MICROCONTROLLER (AT89C51)
Introduction to 8051 Microcomputer
Fig 5.0.1 shows a functional block of the internal operation of an 8051
microcomputer. The internal components of the chip are shown within the broken line
box.
Fig 5.0.1 8051 functional block diagram.
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Fig 5.0.2 shows the external code memory and data memory connected to the 8051
chip.
Note – part of the external code memory can be located within the chip but
ignore this feature for now. Also, variants of the chip will allow a lot more memory
devices and I/O devices to be accommodate within the chip but such enhanced
features will not be considered right now.
Fig 5.0.2 8051 chip with external memory
Fig 5.0.3 Simplified diagram of a Pentium processor
A modern PC is powered by a Pentium processor (or equivalent), which is really
a very powerful microprocessor. Where the 8051 microcontroller represents the low end
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of the market in terms of processing power, the Pentium processor is one of the most
complex processors in the world. Fig 5.0.3 shows a simplified block diagram of the
Pentium processor and a simple comparison between the 8051 and the Pentium is given
in the table 5.0.4 below.
5.0.4 Simple comparison of Pentium Vs 8051
FEATURE 8051 PENTIUM COMMENT
Clock Speed
12MHz typical
but 60MHz ICs
available
1,000 MHz
(1GHz.)
8051 internally divides clock by 12 so for
12MHz. clock effective clock rate is just
1MHz.
Address bus 16 bits 32 bits
8051 can address 216 or 64Kbytes of memory.
Pentium can address 232 or 4 Giga Bytes of
memory.
Data bus 8 bits 64 bits
Pentium’s wide bus allows very fast data
transfers.
ALU width 8 bits 32 bits
But Pentium has multiple 32 bit ALUs along
with floating-point units.
Applications
Domestic
appliances,
Peripherals,
automotive etc.
Personal
Computers
And other high
performance areas.
Power
consumption
Small fraction of
a watt
Tens of watts
Pentium runs hot as power consumption
increases with frequency.
Cost of chip
About 2 Euros.
In value
About 200 Euros –
Depending on
spec.
Table 7.0.4 Comparison of Pentium Vs 8051
The basic 8051 chip includes a number of peripheral I/O devices including two
Timer/Counters, 8-bit I/O ports and a UART (Universal Asynchronous Receiver
Transmitter). The inclusion of such devices on the 8051 chip is shown in Fig 8.4.
These I/O devices will be described later.
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Fig 5.0.5 8051 showing the on-chip I/O devices
5.0.2Memory and Register Organization
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The 8051 has a separate memory space for code (programs) and data. In an
actual implementation the external memory may, in fact, be contained within the
microcomputer chip. However, the definitions of internal and external memory to be
consistent with 8051 instructions which operate on memory is used. Note the
separation of the code and data memory in the 8051 architecture is a little unusual.
5.0.2.1 External Code Memory
The executable program code is stored in this code memory. The code memory
size is limited to 64KBytes (in a standard 8051). The code memory is read-only in
normal operation and is programmed under special conditions e.g. it is a PROM or a
Flash RAM type of memory.
5.0.2.2 External RAM Data Memory
This is read-write memory and is available for storage of data up to 64Kbytes
(in a standard 8051).
Internal Memory
The 8051’s on-chip memory consists of 256 memory bytes organized as
follows:
First 128 bytes: 00h to 1Fh Register Banks
20h to 2Fh Bit Addressable RAM
30 to 7Fh General Purpose RAM
Next 128 bytes: 80h to FFh Special Function Registers
The first 128 bytes of internal memory is organized as shown in Fig 8.2 and is
referred to as Internal RAM or IRAM.
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7.0.3 Pin Description
P 89C 51R C 2
2930
40
20
3 1
19
18
9
3938373635343332
12345678
2122232425262728
1011121314151617
PSENA LE
VC C
GND
E A
X1
X2
R ST
P 0 . 0 /AD 0P 0 . 1 /AD 1P 0 . 2 /AD 2P 0 . 3 /AD 3P 0 . 4 /AD 4P 0 . 5 /AD 5P 0 . 6 /AD 6P 0 . 7 /AD 7
P 1 . 0 / T2P 1 . 1 / T2E XP 1 . 2 /EC I
P 1 . 3 /C EX0P 1 . 4 /C EX1P 1 . 5 /C EX2P 1 . 6 /C EX3P 1 . 7 /C EX4
P 2 . 0 /A 8P 2 . 1 /A 9P 2 . 2 /A 10P 2 . 3 /A 11P 2 . 4 /A 12P 2 . 5 /A 13P 2 . 6 /A 14P 2 . 7 /A 15
P 3 . 0 /R XDP 3 . 1 / TXDP 3 . 2 / IN T0P 3 . 3 / IN T1P 3 . 4 / T0P 3 . 5 / T1P 3 . 6 /W RP 3 . 7 /R D
Fig 5.0.3 Pin description of 89C51
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VCC
Supply voltage.
GND
Ground.
Port 0
Port 0 is an 8-bit open-drain bi-directional I/O port. As an output port, each pin
can sink eight TTL (Transistor Transistor Logic) inputs. When 1s are written to port 0
pins, the pins can be used as high impedance inputs. Port 0 may also be configured to
be the multiplexed low order address/data bus during accesses to external program and
data memory. In this mode P0 has internal pull-ups. Port 0 also receives the code bytes
during Flash programming, and outputs the code bytes during program verification.
External pull-ups are required during program verification.
Port 1
Port 1 is an 8-bit bi-directional I/O port with internal pull-ups. The Port 1 output
buffers can sink/source four TTL (Transistor Transistor Logic) inputs. When 1s are
written to Port 1 pins they are pulled high by the internal pull-ups and can be used as
inputs. As inputs, Port 1 pins that are externally being pulled low will source current
because of the internal pull-ups. Port 1 also receives the low-order address bytes during
Flash programming and verification.
Port 2
Port 2 is an 8-bit bi-directional I/O port with internal pull-ups. The Port 2 output
buffers can sink/source four TTL(Transistor Transistor Logic) inputs. When 1s are
written to Port 2 pins they are pulled high by the internal pull-ups and can be used as
inputs. As inputs Port 2 pins that are externally being pulled low will source current
because of the internal pull-ups. Port 2 emits the high-order address byte during fetches
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from external program memory and during accesses to external data memory that use
16-bit addresses.
Port 3
Port 3 is an 8-bit bi-directional I/O port with internal pull-ups. The Port 3 output
buffers can sink/source four TTL inputs. When 1s are written to Port 3 pins they are
pulled high by the internal pull-ups and can be used as inputs. As inputs, Port 3 pins that
are externally being pulled low will source current because of the pull-ups.
RST
Reset input. A high on this pin for two machine cycles while the oscillator is
running resets the device.
ALE/PROG
Address Latch Enable output pulse for latching the low byte of the address
during accesses to external memory. This pin is also the program pulse input (PROG)
during Flash programming.
In normal operation ALE is emitted at a constant rate of 1/6 the oscillator
frequency, and may be used for external timing or clocking purposes. Note, however,
that one ALE pulse is skipped during each access to external Data Memory. If desired,
ALE operation can be disabled by setting bit 0 of SFR location 8EH. With the bit set,
ALE is active only during a MOVX or MOVC instruction. Otherwise, the pin is weakly
pulled high. Setting the ALE-disable bit has no effect if the microcontroller is in
external execution mode.
PSEN
Program Store Enable is the read strobe to external program memory. When the
AT89C51 is executing code from external program memory, PSEN is activated twice
each machine cycle, except that two PSEN activations are skipped during each access to
external data memory.
EA/VPP
External Access Enable. EA must be strapped to GND in order to enable the
device to fetch code from external program memory locations starting at 0000H up to
FFFFH. Note, however, that if lock bit 1 is programmed, EA will be internally latched
on reset. EA should be strapped to VCC for internal program executions. This pin also
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receives the 12V programming enable voltage (VPP) during Flash programming, for
parts that require 12V VPP.
XTAL1
Input to the inverting oscillator amplifier and input to the internal clock
operating circuit.
XTAL2
Output from the inverting oscillator amplifier.
5.0.4 Features of AT89C51
1. Compatible with MCS-51™ Products
2. 4K Bytes of In-System Reprogrammable Flash Memory
3. Endurance: 1,000 Write/Erase Cycles
4. Fully Static Operation: 0 Hz to 24 MHz
5. Three-level Program Memory Lock
6. 128 x 8-bit Internal RAM
7. 32 Programmable I/O Lines
8. Two 16-bit Timer/Counters
9. Six Interrupt Sources
10. Programmable Serial Channel
11. Low-power Idle and Power-down Modes
The AT89C51 is a low-power, high-performance CMOS 8-bit microcomputer with 4K
bytes of Flash programmable and erasable read only memory (PEROM). The device is
manufactured using Atmel’s high-density nonvolatile memory technology and is
compatible with the industry-standard MCS-51 instruction set and pin out. The on-chip
Flash allows the program memory to be reprogrammed in-system or by a conventional
nonvolatile memory programmer. By combining a versatile 8-bit CPU with Flash on a
monolithic chip, the Atmel AT89C51 is a powerful microcomputer which provides a
highly-flexible and cost-effective solution to many embedded control applications.
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5.1 APPLICATIONS
A continuously conducting diode D has a considerable forward voltage drop.
This is not desirable for low-output- voltage applications. The voltage drop can be
reduced using a synchronous rectifier with low Rds -ON instead of the diode, as
shown in Fig 7.0.1.
Fig 5.1.1. Experimental WIWO converter efficiency.
Fig5.1.2. WIWO dc–dc converter with the synchronous rectifier.
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Fig 5.1.3. Bidirectional WIWO dc–dc converter.
Interchanging the position of the inductor L2 and switch S3, as shown in
Fig 5.1.3, the WIWO topology becomes symmetrical. This also allows driving the top
switch S3 with another flying driver. An additional advantage of the circuit in Fig
5.1.3 is the ability to sustain a bidirectional power flow. The direction of the power
flow can be controlled applying a single-pole double-throw switch, which may be
controlled manually or automatically, as illustrated in Fig5.1.3. This WIWO can be
used in a battery charging and discharging application. With the switch in position 1,
the power flows from the left port to the right port, whereas with the switch in
position 2, the power flows in a reverse direction from the right port to the left port.
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Fig 5.1.4. WIWO PFC ac–dc converter.
The WIWO dc–dc converter can also be used for PFC application (see
Fig5.1.4). Here, a sinusoidal line voltage is fed into the rectifier input. The WIWO dc–
dc converter can accept the rectified voltage and directly produce the required low dc
out- put. With the line voltage greater than the output, the converter works in the buck
mode. As the line drops below the output voltage, WIWO enters the boost mode.
Fig. 5.1.5 Hardware Implementation
5.2 SOFTWARE CODINGS
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5.2..1 Introduction to Keil Microvision2
Keil μVision2 features include
1. Project Setup for the Make and Build Process
2. Editor facilities for Modifying and Correcting Source Code
3. Program Debugging and Additional Test Utilities
4. The Device Database makes it easy to start writing programs for a particular
CPU. Just select the microcontroller to be used and μVision2 sets the necessary
options automatically.
5. New devices may be added to the database as the need arises. μVision2 provides
a Books tab in the Project window where extensive on-line manuals for the tool
chain and selected CPU are found. Double-click on a book title to open the on-
line manual.
6. Most dialogs have a help button which provides detailed information about the
dialog controls. To get help on menu items, select the item and press F1.
7. μVision2 lets us set the options for all files in a target, a group, or even a single
source file.
8. The options dialog opens via the local menu in the Project window.
9. In the Target page of this dialog, the CPU and memory parameters of the target
system may be specified.
10. μVision2 uses this information to configure basic tool options including the
linker/locater settings and the simulator driver.
11. The output page defines the output files generated by the assembler, compiler,
and linker.
Build project
1. Start compiling and assembling target application with the build target button on
the toolbar.
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2. The tool chain stores include and source file dependencies. This information is
used during the make process to build only those files that have changed.
3. Optionally, incremental retranslations are performed when global optimizing is
enabled.
4. The Build page of the output window lists tool information during the code
generation. Double-click on error messages to correct syntax errors in the
program. Errors are correctly located even after insert or delete source lines.
Break points
1. μVision2 allows to set program breakpoints while writing source text.
Simply the buttons on the editor toolbar are used to mark the breakpoints
on source lines.
2. After making the program, the Debugger with the debug toolbar button.
3. Breakpoints that are set while editing are activated in our debugging
session. μVision2 marks the status of the source lines in the attributes
column of the editor window. This provides a quick overview of the
current breakpoint settings.
Utilities
μVision2 contains many powerful functions that helps to complete projects on
time. For example, the Find in files dialog performs a text search in all specified files.
The search results are displayed in the Find in Files page of the Output window. This
feature is used to locate all uses of a function or variable.
Code Execution
1. The buttons on the toolbar are used to step through application program.
2. The run button executes code until a breakpoint is reached.
3. When Trace Recording is enabled, the Show traced records button lists the last
1024 instructions that were executed. Trace recording allows analyzing the
program flow prior to a breakpoint.
Simulator
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Peripheral Simulation
μVision2 simulates the on-chip peripherals of numerous microcontrollers. When
a CPU is selected from the device database to configure the project, μVision2
automatically configures the peripheral simulator.
With its logical and timing simulation, it is possible to test an application before
the target hardware is even available. The simulator makes it easy to test hardware
defects and critical situations which are difficult to debug with real hardware.
CHAPTER 5 . CONCLUSION
This paper has presented a new WIWO dc–dc converter, which is an integration
of buck and boost converters with coupled inductors. The paper described WIWO
principles of operation and offers a comprehensive summary of WIWO analytical
characteristics. Simulation and experimental results were also reported. A modified
PWM modulator scheme required to make the converter work coherently was also
suggested. A prototype WIWO dc–dc converter was built and tested. The converter
demonstrated in practice the WIWO dc–dc conversion ratio.
The new converter topology has several advantages. The WIWO retains the
features of both the buck and the boost converters; however, it achieves wider step-up
and wider step-down dc–dc conversion range. The WIWO converter can operate with
an input source with broadly varying voltage or, alternatively, feed loads with variable
operating voltage such as dc motors. The converter has a simple structure and
moderate component count. The advantageous buck feature allows turning off the
output voltage on demand. WIWO is also inherently capable of limiting the inrush
current and can protect the output in the case of a short circuit. Due to the nonlinear
characteristics, WIWO can avoid operating at extreme duty cycle. As a result, WIWO
efficiency remains high even throughout large input voltage swing. The transition
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between the operating modes is inherently smooth, and causes no transient disturbance
in the average current. Among the disadvantages of WIWO is the coupled inductor
whose leakage causes oscillation and high voltage spike across the switches. Clamp
circuits are needed to clamp voltage spikes upon switches, so as to recycle the leakage
energy. Another disadvantage of WIWOs is that small-signal transfer functions include
an RHP zero, and therefore, WIWO is some- what difficult to stabilize using a single
voltage loop. To resolve the dynamic problem, current loop should be employed,
which is a good practice in any case. An additional disadvantage is that WIWO does
not provide isolation. This, however, may not be much of a problem in systems with
multiple stages.
Modifications of the WIWO to synchronous WIWO dc–dc converter, bidirectional
WIWO dc–dc converter, and WIWO dc–dc converter for PFC are possible. Numerous
advantages indicate WIWO as a viable candidate for many industrial applications.
BIBLOGRAPHY
[1] D. Maksimovic and S. Cuk, “Switching converter with wide dc conversion
range,”
IEEE Trans. Power Electron., vol. 6, no. 1, pp. 151–157, Jan. 1991.
[2] K. Yao, M. Ye, M. Xu, and F. C. Lee, “Tapped-inductor buck converter for high-
step-down dc–dc conversion,” IEEE Trans. Power Electron., vol. 20, no. 4, pp.
775–780, Jul. 2005.
[3] J.-H. Park and B.-H. Cho, “Nonisolation soft-switching buck converter with
tapped-inductor for wide-input extreme step-down applications,” IEEE Trans.
Circuits Syst. I, Reg. Papers, vol. 54, no. 8, pp. 1809–1818, Aug. 2007.
[4] K. Yao, Y. Ren, J. Wei, M. Xu, and F. Lee, “A family of buck type dc–dc
converters with autotransformers,” in Proc. Appl. Power Electron. Conf. Expo.
(APEC 2003), pp. 114–120.
[5] K. Nishijima, K. Abe, D. Ishida, T. Nakano, T. Nabeshima, T. Sato, and K.
Harada, “A novel tapped-inductor buck converter for divided power distribution
system,” in Proc. IEEE PESC Conf. (PESC 2006), Jun., 18–22, pp. 1–6.
[6] G. Spiazzi and S. Buso, “Power factor preregulator based on modified tapped-
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inductor buck converter,” in Proc. IEEE PESC Conf., 1998, vol. 2, pp. 873–879.
[7] F. L. Luo and H. Ye, “Positive output cascade boost converters,” Proc.Inst.
Electr. Eng. Electr. Power Appl., vol. 151, no. 5, pp. 590–606, Sep.2004.
[8] Q. Zhao and F. C. Lee, “High efficiency, high step-up dc–dc converters,”IEEE
Trans. Power Electron., vol. 18, no. 1, pp. 65–73, Jan. 2003.
[9] N. Vazquez, L. Estrada, C. Hernandez, and E. Rodriguez, “The tapped- inductor
boost converter,” in Proc. IEEE Int. Symp. Ind. Electron., Jun.4–7, 2007, pp.
538–543.
[10] K. C. Tseng and T. J. Liang, “Novel high efficiency step-up converter,” Proc.
Inst. Electr. Eng. Electr. Power Appl., vol. 151, no. 2, pp. 182–190, Mar. 2004.
[11] H. Cheng, “Wide input wide output (WIWO) dc–dc converter,” Master’s thesis,
Univ. California, Irvine, Dec. 2007.
[12] H. Cheng and K. Smedley, “Wide input wide output (WIWO) dc–dc converter,”
in Proc. IEEE Appl. Power Electron. Conf. Expo., 2008, pp. 1562–1568.
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