Practical surface acoustic wave devices

582 PROCEEDINGS OF THE IEEE, VOL. 62, NO. 5 , MAY 1974

Practical Surface Acoustic W a v e Devices

MELVIN G. HOLLAND, SENIOR MEMBER, IEEE, AND LEWIS T. CLAIBORNE, MEDEB, IEEE

Invited Paper

Abstract-A tutorial discussion is presented on selected surface acoustic wave (SAW) devices for the use of engineers and electronic systems designers. Currently practical components for use in radar systems, communications systems, and as frequency domain filters are described. Emphasis is placed on nondispersive and dispersive delay lines, devices for generating and detecting various radar waveforms, devices for generating and detecting iixed and programmable biphase and mdtiphase codes, and broad-band and narrow-band filters. The device possibilities of several new approaches are described, including multistrip couplers, nonlinear convolvers, reflective structures, and overlay films.

It is concluded that, while SAW devices have found many applications in large radar systems because of their size, cost effectiveness, and reliability, wide-scale applications, for example as frequency domain filters, will be needed to justify continued research in this field. Wide-scale applications will come as systems engineers realize the potential of these devices and confidently design them into their systems.

I. INTRODUCTION

T HIS PAPER, which wil l describe selected surface acoustic wave (SAW) devices, is directed not to the surface wave expert but to the engineer and systems

designer. I t will not be a general review of SAW phenomena; there are several excellent such reviews [1]-[SI. We will not describe the details of how these devices are designed or pre- pared. Rather we hope to show the systems engineer what the devices can do, introduce him to some of the phonomena which influence and limit the device response, and make him more familiar with the jargon used by the SAW device designer.

We believe surface wave devices will be the basis of the components used in a variety of military and commercial systems in the next few years, and as such should no longer be considered for their potential usefulness but should be evaluated for current uses by comparing their performance with that of other types of devices. We hope that comparison will show why so much emphasis is being placed on SAW components at a large number of industrial and university laboratories throughout the world. However, we will em- phasize devices which are available for systems use, not research-stage devices.

The main applications we will consider will be Radar Systems Components (Section 111), Communications Sys- tems Components (Section IV), Frequency Domain Filters (Section V), and New Techniques and Approaches (Section VI). Radar Systems Components will include nondispersive and dispersive delay lines and devices used for generation and detection of various waveforms. Communications Systems

The Edilol. This invited paper is m e of a scricsplanncd 011 topics of genera l inkrest-

Manuscript received July 19, 1973; revised January 3, 1974. M . G. Holland is with Raytheon Research Division, Waltham, Mass.

L. T. Claiborne is with Texas Instruments, Inc., Dallas, Tex. 75222. 02154.

L Piuoekdric uMho*

Fig. 1. Schematic of simple SAW delay line. (Velocity w, -paration t, and delay time t = L/o) .

Components will include devices for generating and correlating fixed and programmable codes and delay lines for some miscellaneous uses. Frequency Domain Filters will cover the separate problems of broad-band and narrow-band filters. Finally, in New Approaches, we will consider specifically multistrip couplers, nonlinear convolvers, reflective structures, and overlay films. These are new techniques which, in our opinion, will be important to improved device performance in the near future.

We will not discuss acoustoelectric amplifiers or any acoustooptic devices. Widespread use of acoustoelectric amplifiers is still not imminent, even in the surface wave implementation. Acoustooptics is a subject which could not be properly handled in a paper of this length.

11. FUNDAMENTALS OF DEVICE OPERATION Most of the devices we will discuss are essentially tapped

delay lines with a potentially very high tap density and with the amplitude and phase a t each tap capable of precise control. This gives the devices unique signal processing capabilities. The devices usually are operated a t frequencies between 30 and 300 MHz, although higher and lower frequencies have been used. Surface wave devices can have high fractional bandwidths, in theory about an octave. They are usually produced by the same photolithographic processing which is currently used for metallization in integrated-circuit manu- facturing, and thus are compact, reliable, reproducible, and- in principle-economical.

Nondispersive delay lines are the simplest devices concep- tually, yet they have great use throughout electronic systems. Their description will serve to introduce the fundamentals of surface wave structures. The simplest device [6], [7], shown in Fig. 1, is a delay line made up of two interdigita! transducer arrays separated by a distance L, with time delay t = L / v , on a piezoelectric substrate. The properties of a number of substrate materials are listed in Table I. The most commonly used substrates are LiNbOs, ST quartz, and BilsGeOso (BGO). Each has material parameters which make i t useful under certain conditions. For example, ST quartz h a s the lowest

HOLLAND AND CLAIBORNE : SURFACE ACOUSTIC WAVE DEVICES 583

TABLE I CONSTANTS FOR VARIOUS SURFACE WAVE SUESTRATE ORIENTATIONS

Delay Time Temperature

Coefficient Material and Orientation (x105 cYm/sec) Dercent in oornl.C

I 8 1 19 I kz

LiNM3 (YZI 3.48 4.5 91 f 1 ' l o ]

Quartz WX) 3. 17 0 .23 -22f 1 [ I 1 ]

Quar tz (ST, X) 3.15 0.16 Ill 1

LiTa03 NZ) 3.22 0.74 3 1 f 3 ' 1 2 '

LiTa03 o(Z) 3.22 0.69 36f 2

LiTa03 IZY) 3.31 0 .93 67 f 2 '"' BilZGeOZO (111) ( 0 1 3 1 . 6 5 1.7 128 f 2 'I3 ' ZnO IC. any1 .-

2. IO 1.0 40 f 5 'I3 ' and the direction of wave propagation

a Symbols in parentheses are the orientation of the surface no1 *mal,

z

I O+ I

€kc. port I I

'0 I

Equivalent circuit

Fig. 2. Interdigital transducer and equivalent circuit formed by cascading. The 2's are acoustic impedances. Each box is a complex electrical impedance.

temperature coefficient of delay, LiNbO- has the highest coupling constant, and B i 1 2 G e 0 2 0 has the lowest acoustic velocity.

The operation of the simple device can be understood physically by considering that an ac voltage .applied to the input comb structure causes the material between the fingers to distort due to the piezoelectric effect. This periodic strain leads to an acoustic wave propagating away from the input transducer in both directions with a frequency equal to that of the applied signal. In one direction, the acoustic wave is absorbed in a lossy medium. In the other direction, the wave

l b l

Fig. 3. Approximate equivalent arcuits near resonance. (a) Model which best describes a transducer with only a few fingers on a quartz substrate. (b) Model which best describes a transducer with many fingers on LiNm.fo=o/Z; x=N*V-jo)/jo.

propagates to the output transducer and is detected by the inverse piezoelectric effect.

The electrical behavior of the device can be explained quantitatively through the use of a n approximate equivalent circuit [6]. This circuit, derived by Mason [14] to describe the operation of bulk resonant transducers in the thickness mode, is adequately discussed in the literature. The equivalent circuits are cascaded to form a complicated transducer, as shown in Fig. 2. The circuit tends to have a resonant frequency given by

f o = v / l

where 1 is the periodic length of the transducer (Z=Z1+/2+/3

+la in Fig. 2) . Further, the capacitance is a function of the finger length, thus the circuit can be altered by changing the finger spacings and lengths. The acoustic medium is coupled to the electrical circuit by the transformer which represents the piezoelectric coupling.

The response functions of SAW devices can be obtained by computer analysis using the appropriate equivalent circuit models. Agreement with experiment is found to be very good.

For a delay line with a few finger pairs, all with the same spacing, the approximate equivalent circuit of each transducer might look like Fig. 3. Now the frequency response of the device can be calculated and compared with experiment. Further, circuit theory can be used to obtain a matching structure to couple the device to the electrical network with a minimum loss of energy. The simplest matching network would be a series or shunt inductor to resonate the capacitive reactance at the resonant frequency.

A good example, [IS] of a more complicated matching structure used to maintain broad-bandwidth in a comb with only a few finger pairs is shown in Fig. 4. In this case an impedance inverting network was used to match the impedance of the devices to the generator and the load. These results are impressive, but there are two problems. 1) Spurious signals are often generated when there are only a few finger pairs [16]. These spurious signals arise from bulk and plate modes which can be excited at higher frequencies. The bulk modes generally propagate away from the surface into the bulk and can sometimes be eliminated with careful shaping or by treat- ing the bottom surface. 2) The matching network may take up more volume than the device, thus eliminating one of the advantages of the SAW devices.

584 PROCEEDINGS OF THE IEEE, MAY 1974

Fig. 6. Artwork of apodized stagger-tuned combs. The overlap is varied as cos* x in this design.

‘ O t 1 0 I I I I I I I 1 I 60 70 eo 90 KK) 110 1 2 0 1 3 0 1 4 0 I w )

FREQUENCY (writ)

lb l

Fig. 4. (a) Equivalent arcuit using an inverter-coupled transducer to

inverter-coupled arcuit and a delay line made using six-fingered obtain broad-band operation. (b) Insertion 1- obtained with the

transducers on LiNbO:. (After Reeder et d. [15].)

Fig. 7. Artwork of apcdized stagger-tuned combs with dummy fingers.

Fig. 5. Schematic of stagger-tuned combs in a nondispersive configuration. (After Tanaell and Holland [17].)

A . Stagger Tuning While matching schemes are useful, there is another tech-

nique which can be used to obtain broad-band devices [17]. This is shown in Fig. 5. This stagger-tuned device can be thought of as generating different frequencies a t different por- tions of the comb (where the fingers are resonant to the particular frequency). But the geometry is such that the distance between the “resonant” finger on each comb is the same for all frequencies.

I t can be shown [ I71 that the transfer function of such a comb has a phase term which is linear in frequency so that this combination of combs is always (to first order) nondispersive regardless of the function used to position the fingers.

The equivalent circuit of this device can be computed, but i t is clearly much more complicated than that of the “few- fingered-pair” comb. A useful approximation [18] to the impedance characteristics of this type of device is to use the circuits of Fig. 3 with an effective number of fingers:

where T is the delay time for an acoustic wave to traverse a

single comb, and A j is the bandwidth of the comb. The time- bandwidth product d j i s the familiar “compression ratio” of a dispersive device. I t is appropriate here because each comb alone would generate an FM acoustic waveform. I t is the symmetrical positioning of the two combs which makes the structure electrically nondispersive.

Each structure shown in Fig. 5 is usually modified by varying the finger overlap, as shown in Fig. 6. This is called “apodization” [17] or weighting. The amount of finger overlap controls the amplitude of the wave generated by a finger pair. The time (impulse) response of a n interdigital comb will re- semble the physical appearance of the comb.

B . Second-Order Effects There are effects which prevent smooth operation of even

apodized devices if they have large numbers of fingers. These can all be classified as “second-order” effects and are due to the interaction of the SAW with the interdigital fingers as the wave passes under the fingers [19]. In transducers with large numbers of fingers, a whole host of effects appear, the most prominent being acoustic wave reflections and the regeneration of electrical signals in inappropriate fingers. These effects lead to distortion in the acoustic wave and nonsmooth amplitude and phase response in the devices, but can be avoided by a number of approaches.

The first one is to use dummy fingers [20], [21], as shown

HOLLAND AND CLAIBORNE: SURFACE ACOUSTIC WAVE DEVICES 5 85

Fig. 8. Wavefront distortion as an acoustic wave incident from the left pa- under (a) an apodized

Scale is phaee lag in degrees. (After Tancrell and Williamson [20].) stagger-tuned transducer, and (b) an apodized stagger-tuned transducer with dummy fingem.

in Fig. 7. With these, an acoustic wave traveling under any section of the comb 'sees" the same amount of metal. This was not the case for a simple apodized comb since the center of the transducer had a higher density of metal than the edge (near the bus bars). ,4 graphic illustration of the difference in wave distortion between a simply apodized structure and the same structure with dummy fingers is shown in Fig. 8.

A second approach which has been very useful in avoiding acoustic reflections is to use split fingers [22], [23], as shown in Fig. 9. As an acoustic wave travels under standard fingers, it is reflected by the periodic acoustic discontinuity due to the metal. With split fingers the acoustic discontinuity has half the period and thus acoustic reflections occur a t twice the resonant frequency. This is usually outside the frequency band of interest. Various magnitudes of splittings [24] have been used, but in all cases, split fingers imply a need for higher resolution (the fingers are obviously smaller), and so would impose a limit on our ability to make devices for high- frequency operation.

One solution to this might be to use split fingers, but to operate at a third harmonic [23]. This possibility is still in the experimental stage. At present split fingers can certainly be used a t frequencies up to about 200 MHz.

Another approach which is most useful is that of "tilted

Conventional surface wave transducer geometry

Split finger transducer geometry

586

Array R r d v u

Fig. 10. Schematic of tilted stagger-tuned transducers in a nondispersive configuration. (After Tancrell and Meyer 1251.)

TABLE I1

STRUCTURE ADVANTAGE DLSADVANTAGE

Fcr finger combs Easy to make Luge possible bandwidth

Generates bulk mode spurious when used for large bamdwidth. Needs many fingers to get low bandwidth

Stagger-tuned Large bandwidth possible Ripple in amplitude.

number of fingers Good phase response Yield decreases with large

Apodired L o w ripple Wave distortions when N large on certain substrates. Possible diffraction for small overlap

Dummy Fingers Removes wave distortion Does not remove acoustic due to uneqvnl loading of reflections fingers

Split fingers Removes acoustic reflections

Reduces high frequency limit.

Tilted Removes acoustic reflections Needs l u g e r mas& md Removes electrical substrates regeneration

and few disconcerting interactions occur. The problem with this approach is that wider substrates (and masks) must be used. Distortions due to waves traveling under the bus bar have been found to be unimportant experimentally.

Several other novel geometries [27], [28] can be used for nondispersive devices and many more will of course be developed. None of these appears yet to be ready for device applications.

Since the various approaches described here will be presented again in discussing other devices, they are summarized in Table 11.

C. Temperature Efects Temperature has two major effects on a SAW device. The

coefficient of expansion of the substrate material causes a change in the path length of an acoustic wave. The elastic constants of the material are temperature sensitive so that acoustic wave velocity changes. The temperature coefficient of delay includes both these effects.

Materials such as ST-cut quartz have low or zero temperature coefficient of delay, but, as was shown in Table I , materials with low temperature coefficient have low coupling con-

PROCEEDINGS OF THE IEEE, MAY 1974

stant so that a tradeoff of high insertion loss versus low temperature coefficient must be made. The effects of temperature and of crystal orientation will be discussed more fully in the radar and the communication sections of this paper.

111. RADAR SYSTEMS COMPONENTS The present trend in radar system design is to higher

bandwidths, longer radar range, faster acquisition (multitar- get capabilities), and the development of special-purpose radars such as all-solid-state systems, all-weather landing systems, and missile guidance systems.

The fact is that currently, except for commercial marine radars, very few radar systems are being procured in large numbers. T h u s the emphasis in radar systems studies is on technological improvement rather than production economies. For this reason, the overall cost advantage of the photolithographic production of SAW devices is not necessarily an important advantage. However, the unique signal processing capabilities of the devices offer a large advantage.

The radar system designer must first decide on the waveform that is most appropriate for the function that is to be performed by the radar, and then compromise his choice by the hardware available to generate the signal and to process the return signal. I t is in matched filtering and to some extent signal generation that surface wave devices are most appropriate.

The waveforms most commonly [29] used beside an unmodulated CW signal or an unmodulated pulse of energy are 1) a frequency-modulated pulse, 2) a group of phase-modulated pulses, or 3) a group of time-modulated bursts. The relative merits of each will not be discussed since they depend on the radar function; rather we will examine the surface wave implementation of each.

A . Frequency Modulation The most extensive use of SAW'S to the present has been

as linear dispersive delay lines for the generation and detection of chirp waveforms [SO]. There have been a number of devices used for this purpose in the past and these are listed and evaluated in Table 111. Also listed is the digital approach which is another recent trend in waveform generation and detection.

What can be deduced from this table is that the SAW device offers a large timebandwidth product with low loss, although it cannot simultaneously provide ultrahigh bandwidth and exceptionally long dispersive delays available with other techniques. Further advantages, not pointed out in Table 111, are the smaller size, reproducibility, and cost advantage of surface wave devices, although (as noted previously) these latter two are presently not necessarily large advantages. Further ease of design gives the systems engineer a greater variety of chirp waveforms to work with.

1) Dispersioe Surface Wave Deoices: If a surface wave device is constructed with two stagger-tuned combs configured as mirror images [31] [Fig. l l ( a ) ] or with one stagger-tuned comb and one comb consisting of only a few finger pairs [X!] [Fig. l l (b ) ] , t he device will not have linear phase. In fact, if the finger width and spacing of each finger are adjusted cor- rectly, any arbitrary phase versus frequency response can be obtained. This will be discussed more fully in Section V where we discuss filters, but can be understood physically by noting t h a t the. acoustic waves generated a t each finger edge will add to a total traveling acoustic wave with a frequency and ampli-

HOLLAND AND CLAIBORNE: SURFACE ACOUSTIC WAVE DEVICES 587

TABLE I11 CHARACTERISTICS OF PASSIVE LINEAR FM DEVICES [30 I

Bandwidth Time f o Typical Typical (max) (max) BT (avg) (max) Loss spurious Comments

(MHz) ocs) (MHz) (dB) (dB)

Aluminum-strip delay line 1 500 200 5 15 -60 Steel-strip delay l i ~ 20 350 500 45 70 - 55) ~ e d careful temperature control

All-paas network 40 1000 300 25 25 - 40 large size Perpendicular daraction delay line 40 75 1000 100 30 - 45 expensive, nonrepmduable Wedge-type delay line 250 65 1000 500 50 - 50 expensive, nonreprcduable Folded-tape meander line 1000 1.5 1000 2000 25 Waveguide operated near CUM 1000 3 l o 0 0 5000 60 - 2SJ Y IG crystal 1000 10 2000 2000 70 - 20 low dynamic range Surface wave delay line 100 100 1000 300 < 50 - 50 Digital delay line 10 500 1000 20 low low programmable

nonlinear

Clearly, arbitrarily complicated transfer functions can be constructed by altering the values of A , and x,. Another important point is that the impulse response of the device is simply the Fourier transform of the transfer function [ 1 7 ] . Specifically, the amplitude versus frequency characteristic of this type of surface wave configuration is the Fourier transform of the function which describes the overlap of the finger positions. This is true only to first order and for the devices like that of Fig. 1 l(b), but is a useful tool i n initial device design.

2) Linear Chirp: The dispersive device most often designed with SAW’S uses a linear chirp. Devices with the ar- rangements of Fig. 11 are used. When one stagger-tuned comb is used with a few-fingered broad-band comb, the insertion loss of the overall device is usually higher than when two stagger-tuned combs are used. If the number of fingers on the few-fingered comb is increased, then the bandwidth is decreased and there is some roll-off at the band edges. Thus the advantage of using two stagger-tuned combs is that better overall loss characteristics can be obtained. The advantage of one stagger-tuned comb plus one few-fingered broad-band comb is that the design equations are easier, and for the cases where excessive dispersive bandwidth is not needed this is a simpler device to design and prepare.

The exact impedance of these devices is obtained by computer analysis [17 ] , [33] (after a model is chosen), but the equations shown in Fig. 3 with Neff= N / d z j provide a very useful approximation [ 181. Now the term TAj is recognized as the compression ratio of the dispersive device.

One of the early linear chirp devices made consisted of two stagger-tuned combs, each apodized to reduce Fresnel rimle. The results obtained are shown i n Fig. 12. If onlv one

tude dependence determined by how the contribution from each edge adds in phase and amplitude. The transfer function for a device made with one comb, with N fingers at positions x,, and with length proportional to An, and a second comb with only a few fingers is

The finger positions i n this type of design are determined by the phase response of the waveform to be generated (or received). Thus, again, high frequencies are generated (or received) a t that par t of the comb which has fingers close together, and low frequencies are generated (or received) where the fingers are further apart. The bandwidth is determined bv ~~~ - the variation in finger separation that is possible. One limitation is due to the smallest incremental motion in the equipment used to make the photomasks. A further limitation is

where y is the (assumed flat) frequency response of the broad- due to the fact that any pair of fingers, when impulsed, will band few-fingered comb and u is the SAW velocity. Vout and generate a signal at the fundamental frequency (i.e., the veloc- Vi, are the output and the input voltage, respectively. ity divided by the finger separation) and will also generate a

-,

By) =-- v ~ n t y ) - [ A . exp (jz. :9] VinCf)

588 PROCEEDINGS OF TEE IEEE, MAY 1974

FREWENCY ( M H z 1

Fig. 12. Phase and amplitude characteristics of the dispersive device of Fig. l l (a) . (After Tancrell el ul. [31].)

signal a t odd harmonics. T h u s a total bandwidth of 100 percent is a fundamental limit. Generation at certain odd harmonics can be avoided by varying the finger-width-to-separation ratio [34], [35], but this complication will not be considered here. This implies that the higher the center frequency of the device, the higher the absolute bandwidth- within the limitation set by the incremental positioning of the fingers.

The dispersive time delay is determined solely by the length of the combs, independent of the frequency. Thus to get a similar dispersive time delay at 60-MHz center frequency and a t 3W-MHz center frequency, the high-frequency comb will be the same length but will have five times as many fingers, since the wavelength is five times smaller and the resonant fingers are thus five times closer together.

To obtain higher compression ratios, either high frequencies or longer combs are necessary. I n both cases the number of fingers in the comb is increased and this leads to complications. Second-order effects, due to interactions as the acoustic wave passes under the comb, disturb the response. For lower coupling materials, such as quartz, more fingers can be tolerated; in general, the product Nk* (number of fingers times coupling constant) must be less t h a n 1 for second-order effects to be negligible.

Several designs can be used to help avoid second-order effects and obtain large timebandwidth products. Most of these were described in the section on nondispersive devices.

The first is to include dummy fingers. The improvements obtained in the impulse response of a pulse compressor made with dummy fingers, or one made without [21], are shown in Fig. 13. The type of wave distortion prevented with the use of dummy fingers is shown in Fig. 8. The main point is that dummy fingers are easy to implement, can only help the response, and should be used in general.

A scheme used to reduce acoustic reflections in the frequency band of interest is to use split fingers. While this technique will be seen to be more useful in phase-coded devices, i t does find some use in linear chirps. Again, as with nondispersive structures, splitting the fingers means that the fingers are thinner, and this limits the upper frequency operation of the device.

One technique that is useful for obtaining high time- bandwidth product devices is the tilted geometry. Direct comparisons of devices made using “in-line” and tilted geometries have been made. The results of such a comparison [26] are shown in Fig. 14.

The range of delay times, bandwidths, and compression ratios obtained with SAW linear dispersive delay lines can be evaluated from the listing in Table IV.

3) Other F M Waveforms: Several other schemes useful in frequency-modulated radar waveforms should be mentioned here. The first is the use of cascaded devices [39], [40], to obtain systems of higher compression ratio. The problems are no different using SAW devices than in pther types of devices

HOLLAND AND CLAIBORNE : SURFACE ACOUSTIC WAVE DEVICES 5 a9

- 4 l.Ops/div

(OY

l.OFs/div

(b)’ Fig. 13. Measured impulse response of dispersive filters. Filters (a) and

matched pulse mmpressors but incorporate internal Hamming (b) are pulse expanders for a linear chirp. Filters (a)’ and (b)’ are

weighting. Filters (a) and (a)‘ have no dummy fingers. Filters (b) and fb)‘ have dummy fingers. Each filter had 5-MHz bandwidth at 30 MHz with a dispersive time delay of 4 p. (After Gerard el al. [2l] .J

and examples exist in the literature. The transfer function of the overall system is the product of the transfer functions of each device, and this product function usually contains excess Fresnel ripple in the time domain leading to serious distortions [39]. The advantage of surface wave devices is that it is possible to design the separate elements to eliminate distortion problems. As an example, FM predistortion [29], [39] can readily be built into a SAW device by simply rear- ranging finger spacings near the edges of the combs. By arranging the ripple on each section of the composite pulse compressor so that t h e total is not directly additive, better performance results.

A further extension of FM waveforms is to a nonlinear chirp. As in the case of the FM predistortion, a change i n finger positions is used to realize this waveform. Devices of this type, for example the “tangent FM” [47] and the ‘hyper- bolic FM” [48], have been made and their general use should be more extensive in the next few years.

A second simple extension of the linear FM is the V-shaped F M (291 or the quaternary code [49]. This is a combination of an up- chirp followed by a down chirp, and might also include a group of such signals with a phase control between segments. The advantage of the V-shaped waveform [29] (up chirp followed by down chirp or vice versa) is that i t is the most elemental FM pulse-compression function which does not have a linear range-velocity ambiguity. That is, a Doppler shift causes the output pulse of the matched filter to split into two peaks. The separation of the peaks in time is proportional to the Doppler frequency. The advantage of the quaternary- code device is that very large timebandwidth products can be obtained by modularly adding nondispersive delay lines to a set of chirp devices.

The SAb’ chirp device is useful for these waveforms because both up and down chirps can be made. The up chirp is

-3Ok

-‘O t (4

Fig. 14. Comparison of pulse compressors. (a) In-line single dispersion.

shown. In-line single dispersion is usually limited in bandwidth. The (b) In-line double dispersion. (c) Tilted. Compressed pulses are

tilted device has much better sidelobes than the in-line double dispersion. T is the dispersive time delay. The curves of the compressed pulse are symmetrical about T = O . (After Maines el al. [26].)

harder to make with good phase and amplitude characteristics using an in-line design, but can readily be made using the tilted devices. In other types of pulse compressors, both directions of chirp cannot often be made.

4 ) A-oninterdigital Techniques: There are two other techniques which are useful in obtaining dispersive surface wave device>. One is the use of reflective structure [SO] . This appears to be a very important approach and will be discussed in Section \‘I.

The second technique involves the use of dispersive films [ j l ] and waveguides [52]. Although large time-bandwidth product devices have been made using t h i n overlay films, we feel that this approach does not yet offer the versatilit). and reproducibility of the interdigital technique. The difficulties in reproducing thin films are well known. The lack of versatility is due to the fact that the dispersion cur\-e is fixed by the materials and only a small part of the curve is usually linear.

590 PROCEEDINGS OF THE IEEE, YAY 1974

TABLE I V SURFACE WAVE DISPERSNE DEVICES

Center Dispersive Frequency Bandwidth Delay

(MHz) (MHz) (wee ) References - (dB) Comments

30 5 2 36 In a radar system

40 5.6 50 37 30-40

Sidelobes

60 20 60 12 60 20

50 37 12 37

1 31

25 30 28 In production

60 28 2 38 Breadboard met specs

60 25 5 26 30 T i l t e d 60 6 25 25 34 Tilted

60 12.5 10 38 60 8 12.5 39

60 2 25 39 26 "3 For the RASSR system

25 - 60 10 100 41 30 10 10 41 99 39 2.16 41 2*1 3o Off the shelf

100 35 5 42 35 SiOz overlay film

112 40 12 37 30

150 50 2.5 43 25 In-line

150 50 2 41 28 3 00 140 1 43

300 100 10 44 27 Dummy fingers Third harmonic 1250 500 -0.50 45 23

1300 500 0.50 46 25 t o 30 Split fingers, electron beam fob.

Thus the designer is forced to operate it at fixed center frequency, bandwidth, or dispersive delay.

5 ) Other Uses of Linear Dispersion in Radar Systems: While linear chirps are known for their use in radar pulse expansion-compression systems, they have other potential uses as spectrum analyzers or in compression receivers [53]. That is, the linear timefrequency characteristic is used to identify the frequency components of an unknown signal by its time position as i t traverses the device. For this application, the important characteristic of the device is its linearity. SAW devices have been operated successfully in this application [54]. Another possible use for a pair of conjugate dispersive delay lines is in a dispersive constant false-alarm ratio (CFAR) device [ 5 5 ] . These devices, with a limiter between, will help derive weak signals in a noisy background and might find some application even in commercial radars.

6 ) Internal Weighting: An advantage to the use of SAW devices in pulse-compression systems is their capacity to include sidelobe weighting directly in the device design [17], [32]. The output of a pulse compressor which has been used to recompress a linear chirp is a sin x / x function in time. In order to reduce the. 13-dB sidelobes obtained with this function, separate weighting filters are usually included in the system. With SAW devices it is possible to include the weighting function in the design. This is usually done by further varying the finger overlap (on top of that needed to avoid Fresnel ripple). This is readily done and analyzed in a device with one stagger-tuned comb and one broad-band comb, but is harder to design into a device consisting of two stagger- tuned combs. However, it has been done successfully, and the equivalents of Hamming functions and Gaussian weighting functions have been designed into the pulse-compression filter.

7) Temperature Efects: In a pulse expansion-compression

system there are different effects depending on whether both expander and compressor change with temperature together [ 5 6 ] , [s f ] . I n heavily weighted systems, i t normally takes a large difference in temperature between expander and compressor before the main compressed pulse distorts or the sidelobes increase [IS]. However, one important effect is that the center-frequency time delay, sometimes called the pedestal delay, can change with temperature. This can have serious consequences for a radar system since this time delay is usually built into the software of the radar signal processor.

As was noted, in using SAW devices, low temperature co- efficients can be obtained but usually at the cost of insertion loss in the system.

B. Non-FM Waveforms While linear chirps have been used extensively in radar

systems, other non-FM waveforms are more appropriate in some situations. Historically, linear chirps were more easily implemented and have become a habit with some radar system designers. The versatility of the SAW device allows a large number of waveforms to be generated and match filtered, relieving the system designer of some of the limitations of the chirp waveform.

I ) Phase-Modulated Waveforms: An entirely different type of waveform is that made using a phase code. Phase-coded waveforms are readily made using SAW devices and will be discussed in detail in Section IV since their use is more extensive as communications systems components. Their use in radar systems was limited by the availability of good devices. A recent comparison [58] of chirps and phase codes is shown in Table V.

Both code waveform generators and matched filters can be made using SAW devices. However, i t is also possible to generate phase-coded waveforms by phase modulating the

HOLLAND AND CLAIBORNE: SURFACE hCOUSTIC WAVE DEVICES 591

TABLE V PWLSE-COMPRESSION TECHNIQUE COMPARISON [58 ] 3 t 48k

R u e sidelobes Good sidelobe levels possible with easily

Sidelobes on the order of 1 I UcomDresslon ratto.

networks -implemented weigh-ting Not 'easily im-proved

Doppler behavior "Doppler Tolerant". No doppler tolerance. for doopler shift up to Needs multiple c h m e L

signal bandwidth I large fraction of the

Hatched fi lter Readily done by receiver

Readily done by putt- frequency inversion another input termirul

at opposite end of line

I b l

Fig. 17. (a) Envelope of a simple pulse-burst signal. (b) Matched filter output of (a).

t

Waveform

( 0 )

Matched filter

output (b)

Fig. 15. Pulse compression using an actively generated biphase code (Barker code) and a SAW filter. (a) Actively generated signal. Lower

verses the phase of the transmitter. 0.5 */div. (b) Compressed Pulse. trace is amplitude versus time; top trace is the gate signal which re-

1 ,,es/div. Sidelobes are 21 dB below the peak. (After Hieber (59 1.)

Burst procastor

( b l

output

(C)

2 8 t

---.L

Fig. 18. (a) Envelope of a nonperidic pulse-burst signal. (b) Schematic of the delay line matched filter of (a). (c) Matched filter output of (a) through (b).

IMPULSE RESPONSE CORELATON

FIG. 16. &bit Frank plyphase d e on ST-cut quartz with 120-MHz center frequency and 7.5-MHz chip rate. (After Bell et al. [60].)

radar transmitter, and the SAW device is used only as a matched filter. 13-bit Barker codes have been used in this way, with typical results shown in Fig. 15 [ S I . The advantage of this waveform is that unusually good sidelobes (-21 dB) are obtained with no weighting in the system.

Extension to polyphase codes is straightforward. If the waveforms are useful in radar systems, there are now practical ways to implement them. For example, Fig. 16 shows the impulse response and correlation for a 64-bit Frank polyphase code [60]. The expansion of this impulse response in the lower trace illustrates clearly that only near the center of the code are the phase changes 180'. This class of codes provides a limited number of waveforms which yield low sidelobes.

Temperature effects important to phase-coded devices will be described in Section IV-B2).

2) Burst Processor Waveforms: A simple periodic burst of R F energy, such as shown in Fig. 17(a), can be used as a radar signal [29]. If this burst is processed through a matched filter, the output is the series of triangular pulses shown in Fig. 17(b). LVhile the center pulse is a maximum, there is little difference in the height of adjacent peaks; to the radar opera- tor it looks like many targets.

To avoid this ambiguity, a nonperiodic burst such as shown in Fig. 18 can be used. If this type of radar signal is processed through a matched filter such as the tapped delay line of Fig. 18(b) and if the original burst was carefully designed, then the output is one main pulse with very low sidelobes such as shown in Fig. 18(c). If the code has 1z pulse ar- ranged such that no more than two pulses ever overlap on


TABLE VI SPREAD SPBCTRVM CORRELATION DETECTION (SYNCHRONIZATION) [62 ]

IF gain

Analog biphase modulator Reference phase detector

Digital received storage

A/D converter

Correlator output

Code comparison

S I N loas

Acquisition time

Processing gain

Maximum bit rate Minimum bit rate RF frequenaes

Frequency &set

Wide-Band Bit Detection

(Half-Bit Intervals) Q Bit Samples

all wide-band

moderate not needed wide-band, low S / N hard/expensive high-speed multibit register

high speed, Q bit moderate/expensive

hard/expensive baseband multibit digital

easy one high-speed adderaccumu-

lator, multibit buffers, gates hard/expensive excessive slight short

lo00 hard/expensive 100 Mbits/a

unlimited none

no problem

Analog Despread

mostly narrow-band

easy

narrow-band. high S / N moderate

moderate/expensive

not needed simple threshold

baseband analog

easy

not needed

slight 2 N times longer

unlimited easy l o 0 0 Mbits/s none unlimited

no problem

correlation cxce+t when they all overlap, the processed signal has a peak-to-sidelobe ratio of n to 1 and the sidelobes stretch out for a time equal to the pulse length on each side of the correlation peak.

SAW devices are advantageous for this type of application because very accurate, compact, broad-band delay lines can be made. The timing pulses for generating the waveform can be made using a SAW tapped delay line, and the matched filter can be most readily made using a SAW device. U'ave- forms up to 100 ps long with SO-MHz bandwidth pulses are under design [61].

IV. COMMUNICATIONS SYSTEMS COMPONENTS A . Introduction

SAW devices offer several advantages over conventional signal generators and matched filters in communications systems applications [SI, [60], [62], [63]. A device used as a signal generator can convert a video impulse to an R F biphase-modulated signal with a code consisting of up to many hundred chips, with bit rates of over 50 Mbits/s. Devices with slightly different finger spacings can be used to provide slightly altered operating frequencies. A multitude of codes can be easily obtained. Detection is by the conjugate device and thus needs no clock for synchronization. Very high signal fidelity can be obtained.

A comparison [62] of the SAW approach and two conventional approaches to spread spectrum synchronization is shown in Table VI. The conventional approaches are 1) post- detection digital parallel correlator, and 2) predetection serial correlator. Switched and fixed code surface wave parallel cor- relators are compared.

The conclusion drawn from the table is that serial despread is not very complex but requires excessive synchronization-

Switchable Code Surface Wave

Correlator

wide-band, extra gain needed for SW insertion

moderate lOSS

not needed

not needed low-speed storage register n

moderate simple threshold

I F analog, multiport, switchable

hard/expensive I F analog

hard/expensive

slight

short 1000 hard/expensive 100 Mbits/s 1 Mbit/s lo00 MHz

10 kHz

bits and I F analog

(FCC reg.)

(10-ps device)

Fixed Code Surface Wave

Correlator

wide-band, extra gain needed for SW insertion

moderate losS

not needed

not needed

not needed simple threshold

I F analog, single-port

moderate

not needed

slight

short lo00 hard 200 Mbits/s 1 Mbit/s lo00 MHz

10 kHz (FCC reg.)

(10-pa device)

Bandpass filter Phase coded transducer

Fig. 19. Schematic of interdigital transducer design for biphase coded surface wave device.

acquisition time. However, within parameter limitations, SAW devices offer an efficient and economical solution to the synchronization of spread spectrum sequences.

A schematic representation of a typical SAW biphase de- .vice is shown in Fig. 19. Impulsing the input transducer causes a SAW to propagate toward the taps. Each tap is spaced one chip-length apart and, as the surface wave passes under the tap, an RF signal is produced on the output bus bar, The sense of phase of the output signal is determined by the relative finger position; for example, tap 1 and tap 4 produce signals which are 180" out of phase. Each output tap pulse is actually the convolution of the impulse response of the input and tap transducer. Thus the design usually has an input transducer with enough fingers to generate a fu l l chip in time, and the tap transducer has only a few fingers so as not to distort the pulse [64].

The impedance of the input transducer [I81 is of the same form as the impedance shown in Fig. 3 with the number of effective fingers Neff= N , i.e., rAf-1. The output transducer, on the other hand, has an impedance which reflects the fact that the excited waves from each tap do not all add in phase. The impedance can be approximated by using


Impulse generator

Surface wave

I

Conjugate Surface wave

device

Oscil lorcope

Impulse response of 127 bit surface wave device &s/div

Autacorrelotion 2ps/div

Fig. 20. Results obtained with a 127-bit SAW device at 60 MHz.

where n is the number of fingers per tap, N' the number of taps, s, = 1 for a tap with zero phase, and s,= - 1 for a tap with 180' phase.

Since it is likely that a b o u t half of the taps will have opposite polarity, the conductance of the output transducer can be quite low. The capacitive reactance, on the other hand, will be like that due to a comb with n X N' fingers.

B. Fixed Code Filters A fixed code device, similar to the schematic representa-

tion of Fig. 19, can consist of any number of code types, from a Barker code with a few bits to a 1000-bit pseudorandom code. The code is picked to satisfy the system requirements. For a nominal number of bits, at say 60 MHz, the design is straightforward, and the results are reasonable. An example [65] of a generated and correlated code obtained using a quartz substrate is shown in Fig. 20.

As the number of bits increases, the number of interdigital fingers increases, and soon the same problems of second-order effects [63] and interactions under the combs dominate the response. These can be avoided to some extent by making devices on quartz, which has a lower coupling constant; however, a higher conversion loss will result. There are several helpful techniques.

The use of split fingers [22], [23] is the most obvious technique for improving the response of a coded transducer. As in a tapped delay line, the split fingers move the stopband due to reflections from the individual fingers in the taps to the second harmonic of the operating frequency of the device and thus out of the range of interference. A good example 1661 of a device made using split fingers is shown in Fig. 21. However,

Fig. 21. Effect of split lingers. In each case the computed amplitude of the output pulse as a function of time is shown for a 13-bit Barker code device operating at 60 MHz with a 20-Mbitls chip rate. (a)

at each electrode. (c) Device using split fingers plus dummy filling Conventional in-line device as in Fig. 19. (b) Device using split fingers

taps. Use of split fingers clearly improves the sharp= of the nulls between bits and decreases the spurious signal due to d - t i m e reflections which appear after the initial 650-0s signal. Dummy filling taps reduce the magnitude of the spurious signal even more. (After Sandy [MI.)

itself. The taps are spaced periodically, by the chip rate, and thus there will be reflections, and thus stopbands, at harmonics of half the chip rate. A technique useful in removing

~~

there is a second reflection problem due to the tap separation these stopbands is to fill the spaces between the taps with


Fig. 22. Artwork of improved tapping transducer geometry developed for high coupling materials. (After Judd et al. [67].)

0 2 4 6 8 10 12 14 16 18 2 0 22 24 26 28 TIME pS

0 2 4 6 E 10 12 14 16 18 20 22 24 26 28

TIME US

Fig. 23. Effect of dummy filling electrodes. Biphase code device with 127 taps. (a) Conventional taps on L iNW. (b) Dummy filling taps. (After Judd el al. [67].)

metal strips [67], connected as shown i n Fig. 22 and not electrically important to the tap output. Some results obtained using this type of structure are shown in Fig. 23.

An interesting technique used to reduce the effective coupling constant of the tap structure, and thus reduce the regenerated signal, is to overlay the tap transducer with a thin dielectric film [68]. Acoustic reflections still can occur with this scheme. Reducing the coupling constant implies that the gain-bandwidth product of the material is decreased, so that the tradeoff is between insertion loss and spurious (second-time) echo level. In many applications i t is more important to reduce the spurious signal, and it has been shown that the reflections are reduced more rapidly than the coupling constant with film thickness.

The other technique which has been highly successful in eliminating regenerated and acoustically reflected signals is

the tilted geometry [25], [69]. The portion of the acoustic wave which impinges on any tap will be reflected, but that reflected portion never passes under any other tap and t h u s causes no spurious signal at any other tap. Comparison of results obtained on LiNbOa using an in-line and a tilted geometry are shown in Fig. 24. Note the absence of spurious signals and the clean nulls obtained with the tilted geometry.

I f a tilted comb were excited with an electrical impulse, or a CW signal, then an acoustic wave would be generated over a broad wavefront. Even though -each section of the broad wavefront might have a different phase, the radiation conductance [66] of this tilted tapped transducer is

G = 8joPC,(nN)Z

where n N is the total number of fingers and is now Neff. The disadvantage of the tilt is that i t uses more surface area.

HOLLAND AND CLAIBORNE: SCRFACE ACOUSTIC WAVE DEVICES 595

SEQuOlcE LENGTH (ps) Fig. 25. Current fabrication capabilities for coded devices. LN is

L i N a ; QZ is 2-cut quartz. (After Bell et al. [@I.) Fig. 24. Comparison of results obtained using an in-line device on

LiNbOa and a tilted device on LiNbOa. In both cases , the device is for generating (or correlating) a 13-bit Barker biphase code. The artwork of each device is shown as well as a correlated pulse. For the in- line model there are a large number of sidelobes following the peak. For the tilted model there are the correct number of sidelobes, six before and six following the peak. The spurious signals are very low for the tilted c a s e . (After Schulz [65].)

1) Limilations: The limitations in the preparation of these phase-coded devices, besides the availability of good codes for the system needs, are in high-frequency applications and code length [57].

The upper frequency limit is again set by the minimum size of the metal finger which can be produced. Split fingers impose a limit lower than that set by simple fingers unless third-harmonic operation is utilized. The periodic nature of the tap placement in these devices is somewhat helpful since step-repeat techniques can be used in mask-making or even in photolithographic exposing.

Available sample length is another limit. About 10 in of quartz or LiNbOs or 5 in of BGO (half the velocity of the other two) is available. This is about 75 ps of signal. I n several cases the acoustic wave has been made to travel around the sample edge to the other side, thus doubling the total signal length. The use of reflective techniques, to be discussed in Section V, might also increase the length of the code which can be generated.

Fig. 25 is a plot [60] which indicates the range of time- bandwidth-product devices available currently and under development using standard interdigital transducer techniques. A maximum chip rate of 25-percent center frequency was assumed arbitrarily, and the effects of limiting propagation losses to 3 dB and 10 dB, respectively, are shown.

2) Phase Error Efec f s Due to Temperature Limitations and CryLtal Orientation: The effect of a temperature difference between the generating and the receiving phase-coded devices is equivalent to a frequency shift imposed between the two devices [64], [70]. The correlation peak decreases and the sidelobes change as the temperature difference increases. The

8 ,

! 1 I ! I 1 1 1 1 1 1 1 1oD lm 1110 11OK) 1laoU

NsSIWIALLMCMICY~~

Fig. 26. Effect of phase errors on correlation peak (for PSK signab). (After Bell [57].)

effect depends on the particular code used and, of course, on the material. The interesting technique of varying the frequency between the generating and the receiving devices to compensate for small temperature shifts has been used in several laboratories.

A generalized determination can be made for the effect of phase errors on the correlation peak for phase-shift keyed (PSK) signals [57]. In Fig. 26, the loss in correlation peak is given as a function of phase error and signal length. Tempera- ture is only one source of phase error between devices. A second source is that due to crystal misorientation in two dimen- sions and the relative alignment of the crystal and photomask to achieve propagation along the appropriate crystal direction. The fractional change in velocity on phase error is shown

596 PROCEEDINGS OF THE IEEB, MAY 1974

I I

/ +o- 42.75" IST-CUT)

I,

I

Fig. 27. Effect of orientation in surface wave velocity (After Bell [57].)

Fig. 28. Diode switching techniques for surface wave tapped delay lines. (a) SPDT phase switching. (b) SPST on-off switching. (After Staples and Claiborne [74].)

in Fig. 27 for ST-cut quartz. The most critical angle in this case is that for rotation about the X axis.

C. Programmable Matched Filters I t is clear that i t is desirable for many applications in

identification, system synchronization, and spread spectrum communications to develop surface wave matched -filters which can be readily programmed [70]. Except for a few experimental devices, the programmable filters have not been developed to the extent of the conventional fixed code devices. A discussion of these more complex filters is of interest here primarily because the techniques have been demonstrated,

Fig. 29. Quad-diode phase switch as applied to single tap. (b) Correct equivalent circuit. (c) Simplified equivalent circuit of transduoer, switch, and 5 0 4 outuut load. (After Staples and Ckiborne [74].)

and programmable filters will be available for systems applications in the 1975-1980 period. There are three basic classes of programmable filters which have been demonstrated and the nature of the application will. determine which technique is the best choice. The three classes are 1) piezoelectric taps with external diode switching to reverse phase [71], 2) active device taps such as the MOS field-effect transistor in silicon (MOSFET) 1721, and 3) an integrated materials scheme which uses piezoelectric taps controlled by silicon-on-sapphire (SOS) diodes [ i s ] .

Two configurations [74] for the diode switching technique are shown in Fig. 28. The arrangement in Fig. 28(a) uses single-pole double-throw (SPDT) switches to control the phase of each tap. The switches are operated in pairs, with one per pair in the grounded state a t a time. I n the second configuration, a more complex dual-acoustic-beam delay line is required but the switch i s a simple SPST type. The latter approach is suited to processing a large number of chips in multiphased delay lines for which a simple switching circuit is desirable.

For the SPDT phase switching, the phase may be switched 180' by the quad-diode technique shown in Fig. 29(a). The

approximate equivalent circuit is shown in Fig. 29(c) for one polarity, where R, is the forward resistance, Rf<<l/wC, (with C, being the reversed-bias capacitance of a single diode), and

HOLLAND AND CLAIBORKE : SURFACE ACOUSTIC WAVE DEVICES 597

I I I , , I J 0.1 0.2 0.4 a6 0.8 1.0 2.0 4.0

SWITCHIN6 D W E CAPACmwcE - )F

CR

Fig. 30. Tap degradation as a function of reversed-bias diode capaa- tance for SPDT phase switching a 32-chip L i N m tapped delay line. (After Staples and Claiborne [74].)

I I I I I 9.0 6 ATE

- 3.0 2.0

1.0

0 I I I I I I Y 0 5 IO 15 20 25 30

tool> e,, (DEGREES)

Fig. 31. Comparison of pchannel and n-channel MOSFET pienore- sistant deteetor efficiency for (100) plane of silicon versus propagation direction. (After Staples and Claiborne [74].)

R>>R, (with R representing the isolation resistor through which the switching voltage is supplied). Fig. 29(b) is the correct electrical equivalent circuit. The output load is normally the 50-0 termination in parallel with the parasitic effects of other switching networks in a delay line with N taps. Fig. 30 shows the effect of diode capacitance for a 32-chip LiNbOs programmable filter. The parallel summation of the switch parasitics appears to be the limitation of this type of switching configuration.

I t has been demonstrated that the piezoresistance effect i n the inversion layer of a Si MOSFET can be used for surface wave detection. This is the same phenomenon that has been utilized i n the past to produce filament Si strain gauges [75] . This technique is intriguing in that standard large-scale integrated (LSI) techniques can be employed to produce both the surface wave detectors and the control circuitry with standard semiconductor fabrication techniques. The output of the surface wave detector can be shown to be given by

The RF current i is determined by the bias current lo in the transistor, the gauge function G which includes an appropriate normalization for direction of current and propagation, and

(b) Fig. 32. Programmable filter implemented with MOSFET taps. (After

Staples and Claiborne [74].) (a) Building blocks of filter. (b) Hybrid test assembly.

the amplitude of the surface wave. Fig. 31 illustrates the anisotropic nature of the gauge function.

Since silicon is nonpiezoelectric, the surface wave must be introduced either by means of mode conversion or by use of an overlay transducer (see Section VI). Fig. 32 is a composite showing an input transducer using a thin film of ZnO, the MOSFET taps, and a static MOS static shift register which can be used to control the gate voltage of each tap. A hybrid test assembly is also shown.

The filter can be programmed by clocking in a series of ‘ones” and ‘zeros” to turn on the appropriate taps. One draw- back to the MOSFET approach is that the piezoresistance effect is not strong, and hence the insertion loss can be appreciable. In Fig. 33 the insertion loss for several impedance matching configurations is illustrated. The performance of a MOSFET

filter is demonstrated in Fig. 34 for two simple cases-Fig. 34(a) as a tapped delay line, and Fig. 34(b) as a matched filter for a simple 4-bit Barker sequence.

The integrated materials approach is more complicated

598 PROCEEDIKGS OF THE IEEE, MAY 1974

CASE 1

ID - 5omA I D * mmA OW Tap i.1. -%dB 0ncTapl.L -35dB

ly 1 m A

[email protected] &dB

Fig. 33. Single tap insertion loss. (After Staples and Claiborne 1741.)

( a 1 (b)

Fig. 34. MOSFET performance. (a) Top trace is input; other traces show tapped delay line behavior. (b) Correlation of a 4-bit Barker code. Lower traces show detected signal at each of four taps. Top trace is correlated output. (After Claiborne et al. 1721.)

but can be used to combine the low capacitance of an SOS diode with piezoelectric taps for good efficiency.

To summarize the results to date, one must conclude that the performance of hybrid integration techniques for programmable surface wave filters with chip rates in excesb of 10 MHz and sequence lengths greater than 100 is questionable due to the large number of bonds required within the space available. Crosstalk a t high frequencies can also be a problem. The MOSFET approach is convenient but the insertion loss and the high velocity in silicon (-4.5. lo5 cm/s) limit this technique. Finally, the complexity of the integrated materials fabrication makes costly devices inevitable. The programmable filter can be achieved but a considerable amount of development is anticipated before LSI programmable filters are a practical reality.

D. Other Needs and Applications Other devices which might be used in communications

systems applications, besides frequency domain filters which will be discussed i n Section V, are straight delay lines and possibly devices which could be used for multiplexing [76] or for storage.

Sondispersive delay lines have been discussed previously, but the special problems in devices to be used for communications purposes might include very flat amplitude response over nominal frequency bands and very low ripple. I n an application 5uch as a recirculating fading-channel modem, the requirement could be 0.3 dB of ripple over a i-,MHz band. To obtain such a device, the transducer design i.: straightforward,

but great care must be exercised in keeping the electric circuit feedthrough, triple transist, and spurious signals well below the main signal. This offers no fundamental difficulty, just expert engineering practices.

Storage devices are delay lines which operate with pulse- in-pulse-out response and thus need very high bandwidth transducers. Some initial work was done using Golay comple- mentary codes which can provide zero-sidelobe output pulses with video input pulses, but necessitate the use of at least two acoustic paths. Equivalent results have been obtained by simply using the stagger-tuned broad-band transducers described in Section 11.

There is potential use of SAW devices for simple commercial identification schemes. Such use would be gratifying to the SAW device designer. When integrated with a simple video pulse, small devices can generate an R F code which will be unique to each device. Considering the size, cost, and reliability of such a device, schemes for use as identifiers should be forthcoming. Unfortunately, while the optimum frequency of SAW device operation is above 30 MHz and below 500 MHz, the lowest practical chip rate is of the order of 0.1 to 1 Mbit/s. Below 1 GHz, commercial bandwidths of 1 MHz are generally not used.

V. FREQUEXCY DOMAIN FILTERS Frequency domain filters n a y prove to be the most ex-

tensive utilization of SA\?: devices [77 ] , [78]. Since SAW devices operate in the VHF and UHF range, are small, and are batch produced, they offer several advantages over present filters.

The filter properties are based on the fact that the impulse response of a surface wave transducer is the Fourier transform of the function which describes the geometric shape of the finger overlap oi the transducer. Another simplistic way of seeing the potential use of the SA\V devices as filter5 is by reference to the equivalent circuit of Fig. 3. Since each pair of fingers can be thought of as having an RLC equivalent circuit (with R and C dependent on the overlap) and the resonance condition determined by the finger separation, i t is logical that parallel combinations of pairs of fingers (i.e., full transducers) can provide a variety of filter transfer functions.

LVhile the basic understanding [ l i ] . [18] of how to design filters has been obtained, the experimental details and approaches necessary to achieve the very good transfer functions expected from conventional filters are still not always achieved. Filters need very flat ripple-free in-band characteristics, deep and steep falloff outside the band, and very low response outside the band. The tradeoff of in-band ripple for steepness of the roll-off is made i n the design. However, added ripple and sidelobes are due to spuriouh signals such as bulk waves, triple transit, and electromagnetic feedthrough. These are the problems which must be attacked in real filter design.

-4. Surface Wave Bandpass Filters I t is easy to visualize the relationship between the time

and the frequency domains for a few simple Fourier transforms as illustrated i n Fig. 35. A uniformly overlapped transducer which is synchronized a t f o has a n impulse response which is a rectangular envelope pulse a t frequency fo. The Fourier transform is well known to be of a sin x / x form as in Fig. 35. The flexibility of the SAW implementation allows the designer the freedom to choose time domain impulse responses which have desirable frequency transforms. For ex-

HOLLASD AND CLAIBORSE : SURFACE ACOUSTIC IVAVE LIEVICES

IMPULSE RESPONSE ENVELOPE I A t A f e! 1 FREQUENCY RESPONSE

599

Fig. 35. Surface wave bandpass filter characteristics.

TABLE VI1 SURFACE WAVE BANDPASS FILTER CAPABILITIES

PARAMETERS CURRENT PRORCED

Center Frequency 10MHI-l.OGtb IhVh-ZGHI

Bdnhridth %KHz-0.41, 2 0 K H r - 0 . 8 1 ,

Minimum Insertion Loss 6 dB 2 - 3 6 8

Minimum Shape Factor 1.2 1.2

Sidelobe Rejection 4 dB 65 dB

Ultimate Rejectim 60 dB 80 dB

Deviation From Linear PhdY t1.5, il.00

Amplitude Ripple 4.25 dB io. 1 dB

TABLE VI11 SURFACE WAVE BANDPASS FILTERS [ 7 6 ]

Lndplrs Filter Center Frequency Bndriw, Insertion Loss Raeivar

Applications M I W l (dB1

' P R C - 9 5 la 7 a4 m FM - I F la 7 P a 14 P u i s - IF 21 4 am 8 FM - [)Ita 25.0 0 . 4 . 8 I F 4 0 aa 8 N-IF 4 . 0 3.5 25 Spread Spectrum l a D KO 14 - Fmnt E n d D l 0 25 6

* Described in text.

ample, the sin x / x could be implemented in the time domain directly for a rectangular transform. However, for most applications, too many sidelobes would be required, and hence too much substrate material would be used for this to be a practical type of weighting. The Gaussian weighting transforms to a Gaussian and is useful for some applications not requiring a good shape factor. Finally, dispersive arrays with proper design can be used for good shape factors. Some of the considerations necessary for SALT bandpass filter design will be given below simply to help the systems engineer better understand the tradeoffs available.

Some current and projected bandpass performance characteristics are listed in Table VII. Center frequencies below 10 MHz require long devices and better techniques for filter implementation exist. At 1 GHz the transducer linewidths require special fabrication techniques, and to go to 2 GHz the electron-beam lithography must be used. Devices a t these frequencies can be made in limited quantities but are not yet in mass production. Bandwidths below 50 kHz require long

Fig. 36. L-band transponder. (After Hartmann et al. [ f f ] . )

0.4 MHzldiv Horizontal

Fig. 37. Bandpass characteristic of transponder shown in Fig. 36. 20-dB Midband insertion loss; 380-kHz 3-dB bandwidth; 5 M B adjacent channel rejection at f 1 MHz. (After Hartmann et al. [77].)

substrates also, and the limit on bandwidth due to bulk modes has been discussed elsewhere. The minimum insertion loss achievable with two transducers is 6 dB due to bidirectionality; however, lower insertion loss has been achieved with a three-transducer configuration. Broad-band unidirectional transducers which could lead to 1 dB or less insertion loss are under development and will be mentioned briefly below. Shape factors of 1.2 have been achieved and probably repre- sent about the limit for practical devices. Sidelobe rejection, ultimate rejection, phase linearity, and amplitude ripple are all expected to improve in the near future.

Several representative bandpass filters which have actually been developed to the point where they can be manu- factured in quantity and meet all performance requirements are listed in Table VI11 [77] . The filters denoted by an as- terisk will be used here as examples.

Fig. 36 shows a n L-band transponder which has a SA\V IF filter which can be seen in the upper left quadrant. The bandpass characteristics for this filter can be seen on a loga- rithmic scale in Fig. 37. This device is on a rotated Y-cut quartz substrate and has a 20-dB rnidband insertion loss a t 10.7 MHz. The frequency sweep is 0.4 MHz/div. A 3-dB bandwidth of 380 kHz was obtained with 50 dB of adjacent channel rejection a t 1 MHz. Although the performance requirements for this filter did not push the state of the art, the filter solved a major systems problem and added side benefits. The primary advantage of the SAW filter in this application is the stability of the bandshape and time delay with changing impedance, thus allowing a wide dynamic range for the system. The attendant benefits include size reduction, the elimination of precision electrical components and critical tuning steps, enhanced received reproducibility, an order-of-magnitude improvement in temperature stability, superior amplitude and phase response, and greatly reduced cost.


lo1 (M

Fig. 38. Low shape factor VHF bandpass filter. (a) Frequency response;

impulse response; linear phase. (After Hartmann et al. [77].) 12-MHz (7 percent) bandwidth; 14-dB insertion loss. (b) sin x / %

-10 dB

(b) Fig. 39. Frequency response of a -MHz filter. (a) Matched fo= 300

dB/div. (b) Mismatched for phaee stability. lo= 300 MHz; Ref - 10 MHz; Ref = - 6 dB. Vertical scale= 10 dB/div. Horizontal scale=2.5

dB. Vertical scale= 10 dB/div. Horizontal scale= 3 MHz/div. (After Hartmann et al. 1771.)

The frequency and impulse response for a filter [77] with a shape factor of 1.2 is shown in Fig. 38. This filter is used in a transmitter with a frequency multiplier chain to select the desired harmonic. The center frequency is 168 MHz, the bandwidth 12 MHz, and the insertion l o s s 14dB. Again this filter is on quartz for its good temperature properties. As a final example of the range of SAW filters, Fig. 39 shows the frequency response of a relatively narrow-bandwidth front- end filter centered at 300 MHz. The fractional bandwidth is 0.41 percent and this three-transducer filter has been tuned to have less than 6-dB insertion loss even though it is on quartz.

The tradeoff between insertion loss and temperature stabil- i t y ! ~ often caused by the requirement for a relatively wide fractional bandwidth. As was discussed earlier, the radiation conductance depends directly on the square of the electromechanical coupling coefficient and the number of electrodes. For narrow fractional bandwidths there are enough electrodes to make simple electrical matching to the transducer very good and, in the bidirectional case, a 6-dB insertion loss on quartz can be achieved. Fig. 40 shows the minimum matched insertion loss which can be achieved with various substrates without distortion of the passband [18]. The HC quartz

QW 10 Im F n d k n r l Bmd*dh t p I m n l l

Fig. 40. Minimum achievable imrtion loss for various substrates. (After Hartmann et al. [18].)

UNIDIRECTIONAL D M Y LlwE

Fig. 41. Unidirectional transducers. (After Hartmann et 01. 1781.)

refers to a cut which has higher coupling than the normal S T cut but less desirable temperature characteristics.

The bidirectionality of the surface wave transducers leads to two problems 1) There is an inherent 3-dB loss for each transducer even when perfectly matched electrically, and 2) triple transit leads to phase and amplitude distortions. The ideal solution to this problem would be a unidirectional transducer [79] as indicated in Fig. 41. In order to achieve a broad- band unidirectional transducer it can be shown that the optimum structure is three electrodes per wavelength with a three- phase driving network as indicated in Fig. 42. Using multi- level fabrications technology, as is standard in the semiconductor industry today, makes this configuration readily achievable. Early work on this type of transducer indicated that indeed 20 dB and more of directionality can be achieved as shown in Fig. 43. The insertion loss was not 0 dB in this case due to losses in the phase shift and matching networks. However, continuing research has led to the conclusion that the unidirectional transducer can be made low loss and can be used to achieve triple-transit suppression.

B . Resonators as Filters ?jarrow-band filters can be implemented by using a large

number of interdigital fingers in a transducer. The acoustic bandwidth of a transducer is inversely proportional to the number of fingers (171. However, obtaining very high Q’s , lo4 or lo5 for example, would require many more fingers than is feasible. Thus other techniques to obtain narrow-band filters should be considered.

One such approach is to use a long transducer, such as one which would have a large number of fingers, and a given nar-


Fig. 42. Multilayer three-phase transducer. (After Hartmann et al. [78].)

7 I I

REVERSE -- -- FORWARD

- RESPONSE

MATCHING hlATCHlNG R PHASE SHIFT NETVJORK

I d A - UNIDIRECTIONAL TRANSDUCER

INPUT .~

B - B I D I R E C T I O N A L TRANSDUCER

FORWARD & REVERSE CHARACTERISTICS

5.7 10.1 15.7 FREQUENCY IhiHzl

Fig. 43. Unidirectional transducer characteristics. (After Hartmann et QZ. [781.)

(b) Beginnings of a comb filter response as fingers are removed. (c) Comb filter. Fig. 44. Schematic of narrow-band filters. (a) Response due to many fingers.

row bandwidth, but to omit large groups of fingers from the given by 1/rt, the total time for a wave to travel the interdigi- transducer [SO], [81]. An example is shown in Fig. 44(a). tal transducer. As fingers are removed [Fig. 44(b)], more \Vhen there are 2404 fingers in each of the combs, a narrow peaks appear in the frequency response, and the frequency in- frequency response with a percentage bandwidth of about terval between peaks is determined by the reciprocal of the 0.08 percent is obtained. The bandwidth can be thought of as propagation time between the elementary combs (11~~) . Thus,

602 PROCEEDINGS OF TEE IEEE, MAY 1974

I 1 A second type of resonating structure is one in which the

r- 25 KHz r50 KHz

t 100 MHz

(a1

t 1 0 0 MHz

(b)

Fig. 46. Schematic of wraparound delay lines as a comb filter. (a) Simple transducer gives a comb response. (b) Several transducers. properly spaced, eliminate part of the comb response.

with only four fingers, the peaks are each within 0.8 MHz (= 1/rP= 1 / ~ ~ ) of each other.

I n the response of Fig. 44(c) there will also be an overall envelope. The envelope is determined by the frequency response of each elemental group of fingers and hap a width given by (1/rC).

By using two transducers with different spacings between the elemental groups, it is possible to obtain a frequency response with one or a t least only a few peaks [82]. Results obtained using two different combs are shown i n Fig. 45.

A physically similar situation is obtained when a recirculating or resonating structure is used [83]. For example, a wraparound delay line can be used as a comb filter as shown in Fig. 46. Here the frequency envelope is determined by the elementary comb ( I / T ~ ) , and the frequency separation between peaks by the propagation time ( 1 / ~ ~ ) . The peak width is determined by the losses as the wave travels around the device and is approximately given by Af-l/r(, where 71 is the relaxation time, or the time it takes the acoustic wave to lose l / e of its energy. 7 1 is similar to the 7t of the comb in Fig. 44.

The operation of the filter of Fig. 46(a) is the same as that of the filter of Fig. 44(c). By adding a second transducer as in Fig. 46(b), some of the resonances can be removed, a situation like that of the filter of Fig. 44(b). Various combinations of transducers can be used to isolate response peaks and provide a narrow-band filter.

acoustic wave is made to bounce back and forth between reflectors [84]. This type of structure, shown in Fig. 47, also leads to comb filters. In all these cases, clever combinations of elemental transducer placement can isolate one or a few peaks and give a narrow-band filter.

C. Selectible Bandpass Filters The use of discrete taps summed in parallel as discussed in

Section IV can be extended to a relatively new technique for obtaining a selectible bandpass filter with many channels from a small number of filters. Consider the Fourier transform of a square wave in the time domain which is a series of discrete frequency components with varying amplitudes. The inverse situation is an infinite series of impulses in the time domain which produces a series of square pass- and stopbands in the frequency domain. As illustrated in Fig. 48, finite-length surface wave taps can lead to a series of passbands and, if these taps can be alternately switched in phase, the pass- and stopbands can be exchanged. The two transducer configurations are shown in Fig. 49. Obviously, the phase reversal can be ac- complished with diode switches.

The utility of this concept comes from the fact that the selectible bandpass filter can be cascaded to provide multi- channel selectivity. For example, in Fig. 50 the cascading of a three-state filter is shown in a conceptual representation. UHF selectible bandpass filters are under development for receiver applications.

VI. NEW APPROACHES In this section we will consider four new approaches which

we believe will be important in components in the next few years. There are undoubtedly other new techniques which could have been included, but they have been omitted by personal preference.

-4. Multistrip Coupler (dlSC) Throughout this paper we have discussed the second-order

effects introduced by interactions under the combs. The multistrip coupler (RISC), shown schematically in Fig. 51, utilizes these interactions and couples energy from one area of the surface to another [86], [87].

The operation of the device is shown in Fig. 52. A SAW incident on track .4 of an MSC of length LT has its energy transmitted to track B. The coupling is due to the alternating potential difference set up between the metal strips as the acoustoelectric wave passes beneath the strips in track A . These potentials are conducted along the strips and thus must also appear along track B where a new wave is generated.

If the hlSC is only &/2 long, then half the energy from track A appears in track B while half remains i n track A . The phase of the signal in track B is 90" ahead of the signal in track B .

The optimum length of the coupler & is proportional to the wavelength of the surface wave and inversely proportional to k2. the electromechanical coupling constant:

LT a X/k2.

T h u s the higher the coupling constant, the less length an hlSC takes up on a sample.


ill Fig. 47. Schematic of resonator structure as a comb filter. ZL is adjusted

to give low-loas reflections. (After Auld et d. [MI.)

Time Wrmeform I Freguency Spectrum

4 t t l t t (a) .. .-. I .. - 5 -3 - 1

5 6 4 Cl, --- . ..

Fig. 48. Selectible bandpasa filter impulse response. (After Hartmann [85 ] . )


INPUT

4

Transducer Geometries I I I I I Frequency Responses

I nnnn I C

nnnn 'C

Fig. 49. Surface wave implementation of selectible filter. (After Hartmann [ 8 5 ] . )

FREQUENCY RESPONSE

Substrate

Fig. 50. Functional description of selectible bandpass filter. (After Hartmann [85 ] . )

Fig. 51. Schematic of multistrip coupler as a track changer. The signal is sent along track A to the right-hand side of the grating, where it is displaced to the left-hand side of the grating (track B, .

A more exact expression for the number of fingers is given immediate application. Other applications will become evident by as the device is used.

where F is an experimental "filling factor" (-1) and 0 = 2mf(dI ' r , ) = 2 r a ( d / X ) . Here a d represents the proportion of the M S C which is active, d is the finger separation, and z* is the SAR' velocity.

The important point is that N t has a broad flat minimum as a function of frequency. There is obviously a stopband at 0 = 2 r , and there are also other stopbands (not evident from the foregoing equation) due to bulk wave excitation: still the device is very broad-band. For example, if the finger spacing d is kept below ( X h / 2 ) on LiSbOs, where X* is the wavelength of the highest frequency to be used in the device, no stopbands should be encountered during operation.

Some of the devices made using the M S C are shown with

B. Convolrers One of the relatively new device concepts which affords

great flexibility is the use of nonlinear interaction between two oppositely traveling surface or bulk waves to achieve convolution [90]-[93]. In general, surface wave devices are quite linear and can operate over a wide dynamic range. How- ever, with some materials and with large-amplitude signals, the nonlinear effects can be used to make acceptable "convolvers." These devices have not been developed in industry as extensively as the more conventional SAW devices, and a brief description is included here only as a forecast of new device? \ e t to come.

Functionally, the flexibility comes from the fact that a signal can be convolved with any reference waveform which can be generated. However, there can be a synchronization problem since the desired correlation will be obtained only if

their responses in Figs. 52-55. The track change, mirror, uni- the signal arrives within a relatively small time period while directional transducer, and reflecting track change5 all have the reference waveform is in the device. I t has been shown

HOLLAND AND CLAIBORNE : SURFACE ACOUSTIC WAVE DEVICES 605

-

Fig. 52. Multistrip couplers. (a) Track changing using Lt. (b) 3-dB coupler using Ltl2. (c) Transmission loss for a track change wave going from track A to track A and wave going from track A to track B . (After Marshall and Paige [ t i t i ] . )

[94] that the convolution can be obtained if the reference is made repetitive and the convolver output is repetitively gated. This technique should be practical for many applications.

Investigations have been made using many different configurations for the actual device. Bulk modes have been used with metal electrodes perpendicular to the direction of propagation for the convolution [95]. At microwave frequencies, bulk modes have also been used with a microwave cavity coupler [95]. Surface wave propagation has been used with either plates or interdigital transducers for the pickup. En- hancement of the nonlinear effect has been achieved for both bulk and surface waves by the use of nonlinearities in piezo-

Fig. 53. Multistrip mirror. (a) Configuration. (b) Reflection loss. (After Marshall et d. [87].)

Fig. 54. Unidirectional transducer. (a) Configuration. (b) Reflection loss. Note the stopband at 122 MHz. Unidirectional pair (curve i) compared to bidirectional pair (curve i i ) . (After Marshall et a[. [ 8 9 ] . )

electric semiconductors and adjacent semiconductors in the case of surface waves. The latter seems a t this time to be the best way to obtain efficient devices.

The convolver offers flexibility which cannot be equaled with other filters, even a more conventional programmable filter. However, the output signals which can be achieved are usually weak. For example, convolvers which utilize nonlinear acoustic coupling have been reported with an output level of -41 dBm from input reference levels of f 20 dBm. Improve- ments of the order of 30 dB have been obtained with devices that use nonlinear semiconductor coupling [96] and even better performance may be obtained in the future with this or other techniques [97]-[99]. Systems problems may be encountered due to synchronization difficulties [94], but the convolver will also stimulate new systems concepts. This device will undoubtedly find application in future systems.


Ol I I . 1 I I I 50 60 70 80 90 100 110

frequency. MHz

( b)

Fig. 55. Reflection track changer. (a) Configuration. (b) Loss. (After Marshall e1 al. [87].)

tramduwr Output

Fig. 56. Schematic of reflective dispersive structure. (After Williamson and Smith [lo2 1.1

C. Reflective Structures Some of the earliest SAW devices were conceived of as

using acoustic reflections [loo]. A periodic array of acoustic discontinuities was shown to be a n efficient reflector of acoustic energy. Devices were made using metal strips as acoustic discontinuities, based on the fact that either 1) the metal shorted out the piezoelectricity and thus changed the velocity of the wave under the metal, or 2) the metal loaded the surface, changing both the effective density and the velocity. Unfortunately, the reflection coefficient cannot be well controlled by the use of metal strips.

The use of etched grooves as reflectors and, equally important, the use of a chevron geometry [ lol l have led to a new class of reflective devices [102], shown sch-ematically in Fig. 56. An interdigital (ID) transducer is used to launch a SAIV which travels along the surface and i s scattered by the periodic grooves i n the etched grating. The grooves scatter different frequencies preferentially, the efficiency being great- est when the separation of adjacent grooves is exactly one acoustic wavelength along the direction of propagation. The reflection coefficient can be controlled by varying the groove

i

Fig. 57 . Insertion loss versus frequency for the reAective dispersive delay line of Fig. 56. (After Williamson and Smith [lo2 I.)

TABLE IX PERFORNANCE OF REFLECTIVE DEVICES

C m k r f r w c q B d M h Tine Delay Disprsin

l M H 2 L I M I I / I i r s e c l Rdcrences COillmmtr

A simple U B Taylor filter was used for weighting in these experi- Gating sidelobes of - 33 dB were expected.

ments.

depth. In the reflective-array compressor (RAC) geometry shown in Fig. 56, the wave is scattered twice and received a t the output ID transducer.

If the spacing of the etched grating is varied, then different frequency waves can be scattered from different positions in the grating, and dispersive structures can be made. The equations which lead to a linear FM waveform generator and correlator are exactly the same as those for I D devices except that the spacing must be a wavelength rather than a half- wavelength.

Further design considerations are that the angle of the grooves (from the direction of wave propagation) is not quite 45' but must take into account the anisotropy of the surface wave velocity of the material. The grooves cannot be too deep or the surface wave velocity will be altered as the wave propagates under the grooves.

In analyzing the impedance of the device, only the straightforward equations of the broad-band input and output transducers need be considered, rather than the complicated calculations for stagger-tuned transducers of the I D device.

There are four major advantages to these devices when used as dispersive devices. First, the reflective device works on the basis of a simple mechanical reflection of a surface wave


Fig. 58. Insertion loas versus frequency for ZnO on oxidized Si. (After Hickernell et a[. [112].)

from a groove. AI! the second-order electroacoustic effects that plague large-compression-ratio I D designs, such as wavefront distortion and electrical regeneration, are absent.

The second major advantage is the comparative ease of fabrication. A device made only with I D transducers would have fingers that are X/4 wide. On the other hand, with reflective devices, the grooves would be A,’(2&) wide, the number of grooves would be half that required with the I D device, and the resolution requirement would be reduced. Since the propagation path is ‘folded,” the substrate is half the length required by the interdigital approach. This is important since some I D devices require long crystals, which are not currently available.

Third, the operation of the reflective device is much more tolerant of defects; shorts or opens affect the reflection of the acoustic wave only locally and do not short out the entire comb or open an entire finger as would happen in an I D device. I n a device which requires low phase and amplitude distortion, this is a major consideration.

Fourth, reflective devices tend to have spurious levels much lower t h a n I D devices with similar parameters. The reflection angles of the grating are correct only for the Ray- leigh waves. Other modes have different velocities, and thus are reflected a t t he wrong angle and are not completely detected by the output transducer.

Some of the results obtained with this type of device are shown in Fig. 57. These devices are currently being designed for use in radar systems. Some of the results obtained to date are listed i n Table I X [103]. The possibility of internally weighting these devices by systematically varying the groove depth, and thus the reflection coefficient along the device, is also currently under study and should be perfected in the near future.

There may be some advantage in using these devices for frequency domain filters as well [105]. The elimination of spurious signal, especially bulk wave excitations, would be the useful property in filter design.

D. Overlay Films This is not so much a new approach as a revitalization of

an old approach [106], [1O7]. Overlay films can provide versatility in material parameters for SAW devices. Revitalization is due to 1) a new set of calculations [log], [109], which show

(b) Time .-a

200 nsldiv.

Fig. 59. Results obtained from Barker code generator made with ZnO on Si. (a) and (b) are simply the conjugate codes from the same device. The droop is due to losses in the device. (After Holland et al. [1181.)

t h a t this versatility can be obtained with thin films, and 2) improved ZnO film production a t several laboratories [110]-[115]. The reason that overlay films are not yet a well-developed approach is that reproducible high-quality thin films have not been easy to obtain.

Calculations have been performed which show that the piezoelectric coupling constant and the SAU’ velocity can both be influenced by the presence of a th in overlay film on many substrates. For example, a thin (half-wavelength) film piezoelectric ZnO on weakly piezoelectric quartz can improve the piezoelectric coupling to a value higher t h a n is obtained on either material alone [lll]. A film of nonpiezoelectric SiOl, on the other hand, can reduce the coupling constant of LiNbOa [117]. Thus the coupling constant can possibly be tailored to meet various device needs.

Results have been obtained on several devices. Fig. 58 shows the insertion loss for a nondispersive surface wave delay line made on (nonpiezoelectric) silicon using a ZnO film [112]. The device is not very broad-band since the bandwidth was limited by the number of fingers in the transducer, but clearly a low-loss device was obtained on this very useful material.

A second type of device made on silicon is a phase-coded delay line. Results obtained with a 13-bit Barker code made using I D transducers on silicon are shown in Fig. 59 [118]. The problem here is that, at low frequencies, the loss due to the acoustic-wave-charge-carrier interaction is too high. However, the ZnO overlay certainly can be used to generate surface waves on Si.

The final example of the usefulness of overlay films is as a means of increasing the coupling constant in MSC structures [119]. The higher kZ, the fewer fingers are needed in an MSC. Some results obtained with ZnO on LiNbOs are shown in Fig. 60. These show clearly the potential advantage of overlay films in MSC designs.


TABLE X LIMITATIONS

Limiting Variable Solution Parameter

F’ossible 120

Y 80

LO

Upper frequency

Bandwidth

Finger width a d spacing New mask-making techniques, e .g . , scanning electron microscope

Fundamental limit of 100 percent Trade loss for bandwidth Operate at higher frequencies

New materials, e.g.. BCO, overlay f i i on

Wraparound glass

Reflectors Multistrip

Better lenses Ruling engines Reflective techniques

Third harmonic generation

t Delay Line Sample sire - O O 20 LO 60 80

frequency, MHz

Fig. 60. Effect of ZnO on a multistrip coupler on L i N m . As the ZnO thickness is increased. the coupling constant is increased and fewer fingers (Nt) are needed for energy transfer between tracks. (After Armstrong cf d. [119].)

Dispersion

Code length

Second order effects Mask s ire and sample size

Sample size Mask size Second order effects

New materials, e.g., BCO. overlay films on glass Wraparound Reflectors Multistrip Film overlays have potential use in waveguide structureS

as well [ 5 2 ] . The possible applications of guided surface waveS have been described elsewhere. Basically they could be applied analogously to microwave waveguide circuits. Thin film guides have been disappointing because the Rayleigh wave is not tightly confined, and the characteristics of the propagation are strongly dependent on film quality. A new mode of propagation which is a transverse wave in a ridge on a substrate has better confinement and may provide a solution to the guided wave devices. This subject will not be covered here.

VII. CONCLUSIONS Practical surface wave devices have been developed for a

number of systems applications. The S.4W is an ideal linear transversal filter which permits great design flexibility. I t is generally true that these devices offer a more cost-effective solution to the specific functions for which they are suited than other available techniques, e.g., the pulse-compression filter replaces either a digital computer, or a cumbersome bulk compressor which is much larger, or an L-band meander line compressor. The surface wave bandpass filter at UHF solves a problem for which there is no other practical solution. Frac- tional bandwidths of the order of 1 percent in the VHF with good shape factors can be obtained. Devices with these char- teristics fill a need which exists between the performance of high-Q crystal filters and the relatively wide-bandwidth multi- pole L-C networks. Again, the flexibility of design for the surface wave filters, the generally good phase, dynamic range, and temperature characteristics, and the reasonable insertion loss make these devices useful components for many applications.

The basic I D transducer for piezoelectric surface wave filters has been developed to the point where good design pro- cedures have been obtained. Since much R F equipment op- erates at either VHF or U H F or has I F processing in this frequency range, there are many applications for these filters. In Table X a summary of the present practical device limitations is presented. As has been discussed in the text, there are gray areas where research devices are becoming practical. This table should serve as a guide to those making decisions on the applicability of a surface wave device for a specific job, but it should not be regarded as setting absolute limits.

The programmable filters must be regarded as being still in the development phase. Programmable matched filters for phase-coded waveforms have been demonstrated using several

Filter response Number of fingers Second order effects

Better designs. e . & , digital filter concepts resonator concepts

Educate users Imrrove devices

Acceptance and wide use Ncsd

Engineer

Radar Systems

la,e I tew ita1 + Fixed Codes

Fig. 61. The future of the surface acoustic wave (SAW) industry.

techniques, but certainly would not be available-in”taige quantities a t this time. The selectible bandpass filter has.also been demonstrated and will probably be availablebefore the more complex programmable PSK matched filteps..

The newer concepts such as the MSC and the reflective structures are currently, being used- in developing research devices. These techniques will extend the range of device performance and offer some new and exciting possibilities for the future. The convolver affords a flexibility which may lead the way to new systems concepts while the ridge guide work may lead to much more compact devices compatible with integrated circuits.

HOLLAND AND CLAIBORKE : SCRFACE ACOUSTIC WAVE DEVICES 609

while SAL$’ devices have found systems applications, the I251 R. H. Tancrell and P. C. Meyer, “Operation of long surface acoustic

needs have been for small numbers of devices. This is espe- interdigital transducers,” presented a t the 1971 IEEE Ultrasonics Symp., Miami, Fla., Dec. 1971, Paper J-7.

cially true in the radar and communications fields. If SALT [26] J. D. Maines, G. L. Moule, N. R. Ogg, and G. Rich, “Improvements devices are to become more than specialty items, it will be due in surface acoustic wave pulse-compression filters,” Electron. Lett.,

to volume needs. The most obvious area in which these devices 1271 A. J. Bahr, R. E, k, and A. F. podnell, “The grating array: A vol. 9, pp. 156-157, Apr. 5, 1973.

can fill large volume needs is in the area of frequency domain new surface acoustic wave transducer, Proc. IEEE (Lett.), vol.

these filters as illustrated in Fig. 61. Without this strong ducer: A new technique for generating surface waves on nonpiez- economic base which filters can supply, the future of the field electric substrates,” presented at the 1969 IEEE Ultrasonics

as an industry is not extremely promising. 1291 C. Cook and M. Bernfeld. Radar Signds-An Introduction la

Thus we see the industry On [28] A. p. van den Heuvel, D. B. Owen, and S . G, Jmhi, “Hybrid 60, pp. 443-444, Apr. 1972.

Symp., St. Louis, Mo., Sept. 1969, Paper C-8.

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Digital Coding of Speech Waveforms: PCM, DPCM, and DM Quantizers

NUGGEHALLY S. JAYANT, MEMBER, IEEE

Incited Paper

Abstract-A study is presented on the digital coding of speech by means of a straightforward approximation of the time waveform. In particular, the closely related discrete-time discrete-amplitude signal representations that are rather well known as pnlse-code modulation (PCM), differential pulse-code modulation (DPCM), and delta modulation (DM) are discussed.

Speech is recognized as a nonstationary signal, and emphasis is therefore placed on “compandit@ and “adaptive” strategies for waveform quantization and prediction. With signal-to-quantization- error ratio SNR as a performance measure, techniques are suggested which are most likely to be appropriate for given specifications of information rate.

It is pointed out that error waveforms in speech quantization cannot be regarded as additive white noise, in general. This means that for h e r assessments of speech coders, either relative or absolute, one needs to supplement SNR-based observations with corrections for subjective and perceptual factors. The latter seem to defy quantification as a d e . Invaluable, therefore, are explicit preference tests for direct comparisons of coders from a perceptual standpoint, and notions such as isopreference and multidimensional scaling are naturally appropriate in interpreting the results of such tests.

Final points of concern are communication questions such as multiple encodings of speech by tandem coder-decoder pairs; con- versions among different digital code formats; and the effects of additive and multiplicative noise in the communication channel, as mani- fest in the erroneous reception of speech-carrying bits. Information on these topics tends to be heterogeneous and nontheoretical, and the present digression into the subject is cursory by intent.

The gramophone record accompanying this paper demonstrates some of the manipulations of speech that are discussed.

The Editor. This invited paper i s one of a series planned on fopics of genera l infer&-

Manuscript received September 10, 1973; revised January 28, 1974. The author is with Bell Laboratories, Murray Hill, N. J. 07974.

I. INTRODUCTION H E ADVANTAGES of coding a signal digitally are well known and are widely discussed in the literature. Briefly, digital representation offers ruggedness, effi-

cient signal regeneration, easy encryption, the possibility of combining transmission and switching functions, and the advantage of a uniform format for different types of signals. The price paid for these benefits is the need for increased bandwidth. Futuristic carriers, as well as some recently utilized channels, are characterized by rapidly increasing bandwidth availability. This, together with the emergence of the requisite device technology, is the reason why digital techniques today are so promising. Papers by Flanagan [ l ] and Schindler [2] include illustrative sections on digital coding as applied to speech transmission, synthesis, and storage.

The treatment in the present paper will be confined to the encoding of speech by means of a straightforward recon- struction of the acoustic time waveform: specifically, by means of the closely related discrete-time, discrete-amplitude representations known as pulse-code modulation (PCM), differential pulse-code modulation (DPCM), and delta modulation (DM). b’e are interested in straightforward approxi- mations of the time waveform in the sense that we will not consider questions on the reproduction of speech in terms of features such as excitation descriptions, vocal-tract resonances, or articulatory parameters. Many dramatic results in speech compression have indeed depended on the utilization of these fundamental features [ 11. However, straightforward waveform coding still proves to be an important subject on two counts. First, from the point of view of coder complexity,

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Practical surface acoustic wave devices