Power Supply Modulation for RF Applications

A. Bräckle1, L. Rathgeber

1, F. Siegert

1, S. Heck

1, M. Berroth

1

1Institute of Electrical and Optical Communications Engineering, University of Stuttgart, Stuttgart, Germany,

alexander.braeckle@int.uni-stuttgart.de

Abstract — Envelope tracking is a promising means to

enhance radio frequency power amplifier efficiency for

signals with non-constant envelope. The adaption of the

power amplifier supply voltage is performed by a voltage

modulator circuit, which is mainly based upon power

electronics circuits, but requires bandwidths up to the MHz

region. This paper introduces the benefits and challenges of

envelope tracking and gives an overview of several voltage

modulator architectures. Current measurement data of a

class-G modulator is presented as well.

Keywords — Envelope tracking, high efficiency, RF power

amplifier.

I. INTRODUCTION

The power amplifier (PA), the final amplifier in front of the antenna in a wireless transmitter, is the most power-consuming component in radio frequency (RF) transmitters. Therefore, in order to reduce energy costs as well as the cooling effort, a high efficiency of PAs is aspired.

In communication systems, linearity is an important issue as well because signals must not be distorted during transmission. Therefore, mainly linear power amplifiers of classes A, AB, and B are in use. In theory, they achieve efficiencies up to 50 % in the case of a class-A amplifier and 78 % for a class-B amplifier. However, high efficiencies are only reached when operating the amplifier near its maximum output power. For reduced output power, it rapidly drops. This is illustrated in Fig. 1, which shows efficiency versus output power measured for a GaN class-AB amplifier designed at our institute.

This is no problem for communication systems solely based on phase or frequency modulation, such as the Global System for Mobile Communications (GSM). However, in modern communication systems, such as Long Term Evolution (LTE), modulation schemes which also change the amplitude of the carrier are used in order to achieve higher data rates, and the signals are highly dynamic with peak-to-average power ratios (PAPR) up to 12 dB. Fig. 1 also shows the power density function (PDF) of a typical LTE signal. It can clearly be seen that the amplifier is operated in the low-efficiency region most of the time.

Therefore, efficiency enhancement techniques for RF PAs are intensively investigated. There are several concepts to increase PA efficiency without affecting linearity, for example the Doherty architecture [1], the Chireix architecture [2] and the class-S architecture [3]. The first two architectures strongly limit bandwidth whereas the latter one necessitates high power switch-mode transistors which can be operated in the GHz region, which poses a technological challenge.

Another possibility to enhance PA efficiency is

envelope tracking (ET) [4]. This concept is presented in more detail in this paper. It is structured as follows: in section II, the basic concept of ET is presented and the general requirements and challenges for voltage modulators, which are the crucial components in ET systems, are highlighted. They can be realized in various ways. Some of them are introduced in section III. Section IV gives an example of a voltage modulator designed at our institute. Finally, the paper is summarized in section V.

II. ENVELOPE TRACKING CONCEPT

The basic idea of ET is to dynamically adapt the RF PA supply voltage to the instantaneous output power level. By this, the DC power consumption is reduced without affecting RF output power, which results in a higher efficiency. This is visualized in Fig. 2 showing the normalized drain-source voltage of a transistor. In Fig. 2 (a), a constant supply voltage UDC is assumed, which results in low efficiency for reduced output power. The losses of the circuit can be represented by the area between the envelope of the RF signal and the time axis. When varying the supply voltage uenv(t) according to the envelope A(t) of the signal, this area can be eliminated, as it can be seen in Fig. 2 (b).

The effect of ET is also illustrated in Fig. 3 showing the efficiency measurements of the class-AB amplifier for several supply voltages. It can be seen that for lower supply voltages, the point of maximum efficiency is shifted towards lower output powers.

In theory, the supply voltage can be reduced down to 0 V. However, in practical RF PAs power gain drops with decreasing supply voltage. Therefore, there is a lower boundary for the supply voltage. As shown in Fig. 4, for the class-AB amplifier considered in this paper, the

Fig. 1. Measured drain efficiency versus output power of a class-AB power amplifier at a frequency of 2.75 GHz with a supply voltage of

28 V (solid) and PDF of an LTE signal (dashed).

15 20 25 30 35 dBm 450

10

20

30

40

%

60

Output Power

Pow

er-

added e

ffic

iency

15 20 25 30 35 dBm 450

2

4

6

8

%

12

Output Power

PD

F

15th International Power Electronics and Motion Control Conference, EPE-PEMC 2012 ECCE Europe, Novi Sad, Serbia

978-1-4673-1972-0/12/$31.00 ©2012 IEEE LS8d.3-1

supply voltage can be varied between 10 V and 28 V to maintain a power gain greater than 10 dB.

In Fig. 5, a block diagram of an ET system is depicted. As in conventional PA designs, a linear RF PA with efficiency ηRF is driven by a modulated input signal. In addition, there is a voltage modulator dynamically providing the supply voltage uenv appropriate to the instantaneous output power Pout such that the PA is operated in its high efficiency region. However, in order to profit from the efficiency enhancement of the RF PA, the voltage modulator efficiency mod should be as high as possible. The overall system efficiency can be computed by

.modRFtot (1)

For this reason, a high efficiency of the voltage modulator is of paramount importance to profit from the

benefits of ET. Therefore, the voltage modulator efficiency should be at least 80 %.

Furthermore, the modulator bandwidth needs to be sufficiently large to follow the envelope of the RF signal. For an LTE signal, the required bandwidth is 20 MHz.

The requirements for the voltage modulator in this work are a bandwidth of 20 MHz, a supply voltage range between 10 V and 28 V and a maximum modulator output current of 1 A for the RF PA. Furthermore, the modulator needs to be able to drive a capacitive load of about 50 pF, resulting by the bias decoupling of the RF circuit.

III. VOLTAGE MODULATOR CONCEPTS

Due to the high efficiency requirements, most voltage modulator concepts are based on switch-mode voltage regulators known from power electronics. In principle, linear regulators are possible, too, but due to the high modulator losses, overall system efficiency cannot be enhanced.

The main challenge in designing voltage modulators for RF PAs is to adapt well-known concepts of power electronics to fulfill the high bandwidth requirements and to maintain a high efficiency at the same time.

This section introduces several voltage modulator concepts which can be used for ET applications.

A. Buck Modulator

The buck modulator is based on the step-down converter. Its basic circuit diagram is shown in Fig. 6.

Fig. 5. Block diagram of an ET system.

10 15 20 25 30 35 dBm 458

9

10

11

12

13

14

dB

16

Output Power Pout

Gain

10 V ... 28 V

RL

uenv

UDC,M

uoutRF PA

Voltage

modulator

Baseband

Predistortion

(modulator and

RF amplifier)

Time Alignment

I

Q

Up-conversion

LO

Fig. 4. Measured power gains versus output power of a class-AB

power amplifier at a frequency of 2.75 GHz for supply voltages between 10 V and 28 V.

10 15 20 25 30 35 dBm 450

10

20

30

40

%

60

Output Power Pout

Pow

er-

added e

ffic

iency

10 V ... 28 V

Fig. 3. Measured drain efficiencies versus output power of a class-AB power amplifier at a frequency of 2.75 GHz for supply voltages

between 10 V and 28 V.

0 1 µs 30

0.5

1

1.5

2

Time t

Voltage u

/UD

C

uDS

(t)

UDC

(a)

0 1 µs 30

0.5

1

1.5

2

Time t

Vo

lta

ge

u/U

DC

uDS

(t)

uenv

(t)

(b)

Fig. 2. Drain-source voltage of a power transistor excited by a non-

constant envelope signal (a) for a constant supply voltage UDC and

(b) for a time-varying supply voltage uenv(t).

LS8d.3-2

The input envelope signal A(t) is compared to a saw-tooth signal to generate a pulse-width modulated control sequence with switching frequency fs which drives the buck modulator. At the modulator output, the voltage is filtered to attain a continuous-valued signal uenv(t).

As explained in the previous section, uenv(t) needs not to be as low as 0 V. Therefore, the modulator switches between two positive supply voltages UDC,M+ and UDC,M- in order to increase efficiency [5].

In theory, the transistor and the diode are assumed to behave as ideal switches and losses of the inductor and capacitor are neglected, resulting in an efficiency of 100 %. In practical designs, however, efficiency reduces due to several non-idealities. The switching frequency of the voltage modulator needs to be at least seven times the envelope bandwidth [6], so fs becomes very high and switching losses are predominant. As they are directly related to transistor capacitance, there is a contradiction between a large bandwidth, which requires fast devices with small dimensions, and high-power devices with a large geometry. Therefore, high-power buck modulators are only reported for low bandwidths in the kHz range [7] – [9], whereas high switching frequencies are only possible with low output powers below 2 W [6], [10].

B. Switch-Mode Assisted Linear Modulator

The main drawback of buck modulators is the required oversampling to produce a continuous-valued output voltage. This can be avoided by the use of switch-mode assisted linear modulators (SMALM). The circuit diagram is depicted in Fig. 7.

The modulator was first proposed in [11] and consists of a switch-mode amplifier and a linear amplifier connected in parallel to the RF PA. The state of the switch-mode amplifier is changed when the linear stage output current reaches a positive or negative threshold. Therefore, the modulator is a self-oscillating system. The circuit combines the advantages of the two amplifier types: Switch-mode amplifiers achieve high efficiencies at low frequencies but have a low bandwidth, whereas linear amplifiers have high losses but can attain high operating frequencies. It can be shown that 80 % and more of the envelope power are situated at low frequencies and can be generated very efficiently by the switch-mode amplifier by using a – compared to the total bandwidth – small switching frequency. For a properly designed circuit, only a small portion of the modulator output power needs to be generated by the linear stage, resulting in a high overall efficiency.

Due to the advantages, SMALMs are widely used when uenv has a high bandwidth. Several systems in the power class of 2 W and more and bandwidths in the MHz

range have been published [12] – [15].

C. Class-G Modulator

For very high power applications, design of SMALMs becomes challenging as well. Even if the relative power provided by the linear stage is small, the absolute value can become too high for devices with bandwidths in the MHz range.

Therefore, another modulator concept without a linear amplifier is considered. It is based on the fact that ET not necessarily requires the supply voltage to exactly track the shape of the envelope signal. Efficiency enhancement can also be achieved when switching the supply voltage between two constant values. This is referred to as class-G architecture and was first proposed in [16] for audio amplifiers and in [17] for RF applications.

As shown in Fig. 8, the modulator is the same circuit as the buck modulator, but without an output filter. If the input signal is below a threshold voltage Uref, the low supply voltage UDC,M- is connected to the output. For high input signals, uenv is set to UDC,M+. Efficiency enhancement of the RF PA achievable in this approach is lower than with continuous-valued modulator output voltages, however due to lower modulator losses overall system efficiency can be comparable to the use of other modulator concepts.

Several examples of class-G modulators have been presented in literature [18] – [21], including the ET system with the highest output power presented so far [20]. The concept is not limited to switching between two supply voltages, multilevel operation is possible as well [21].

IV. MEASUREMENT RESULTS

To demonstrate the potential of the ET concept, a

class-G modulator designed at our institute is presented.

The two supply voltages provided by the circuit are 10 V

and 28 V, respectively, at an output current of 1 A. In

A

UDC,M+

uenvC

L

Comparator

Driver

to RF PA

UDC,M-

A

UDC,M+

uenv

L

Comparator

Driver

to RF PA

UDC,M-

Rsense

A

UDC,M+

uenv

Comparator

Driver

to RF PA

UDC,M-

Uref

Fig. 6. Circuit diagram of a buck modulator.

Fig. 8. Circuit diagram of a class G modulator.

Fig. 7. Circuit diagram of a switch-mode assisted linear modulator.

LS8d.3-3

order to assure an optimum gate-source voltage all the

time, a bootstrap circuit is included in the gate-driver.

The requirements for the circuit derived from the RF

PA characteristics result in a minimum load impedance

seen by the modulator of 27 . Therefore, the class-G

modulator is first measured with a constant load resistor.

Fig. 9 shows efficiency versus duty cycle for several

switching frequencies. For high switching frequencies,

efficiency decreases due to increasing switching losses.

However, efficiency is above 80 % for switching

frequencies up to 10 MHz.

Next, the modulator is connected to the RF PA and the

entire ET system is measured. In Fig. 10, a photograph of

the setup is depicted, showing the class-G modulator as

well as the RF PA. The RF PA is driven by an

unmodulated sinusoidal input signal. The modulator

output voltage is kept constant and is 10 V for output

powers below 34 dBm. For higher values of Pout, uenv is

28 V. The measured power-added efficiency (PAE) is

plotted in Fig. 11. For comparison, also the efficiency of

the RF PA with a constant supply voltage of 28 V is

included. For high output powers, efficiency slightly

decreases due to modulator losses. However, for a 10 V

drain bias, efficiency is significantly increased and

reaches more than 60 % at an output power of 34 dBm,

which is twice as high a without envelope tracking.

Finally, the modulator is driven by an LTE signal. Two

testmodels, which are defined in the LTE standard [22],

are used to characterize the amplifier with predefined

signal sequences. Testmodel 1.1 has to be used for

adjacent-channel leakage ratio (ACLR) conformance

testing, testmodel 3.1 is required for measuring error

vector magnitude (EVM). However, ACLR and EVM are

evaluated for both testmodels in this paper. A measured

sequence of the modulator output voltage is shown in

Fig. 12. Pulse widths as short as 100 ns can be produced

by the modulator.

As the PAPR is different for the two testmodels, the

input power of the RF PA without ET is chosen to attain

the same peak input power in the both cases. Therefore,

the input power for testmodel 1.1 is about 1 dB above the

input power of testmodel 3.1. However, memory effects

slightly change the PA behaviour for the different

testmodels, resulting in a 2 dB difference in the output

power. When measuring the system with ET, the input

power is set to get the same output power as without ET.

Table 1 summarizes the measurement results for LTE

signals with ET and for a constant supply voltage. The

potential of ET becomes clear when looking at the

efficiency. Depending on the chosen test signal,

efficiency increases to around 38 %, which is an

improvement by almost a factor of two. To the best of the

0.2 0.3 0.4 0.5 0.6 0.7 0.880

85

90

%

100

Duty cycle

Effic

iency

mod

f = 1 MHz

f = 4 MHz

f = 8 MHz

f = 10 MHz

Fig. 9. Efficiency of the class G modulator with constant load resistor.

0 0.5 1 µs 25

10

15

20

25

V

35

Time t

Voltage u

env

Fig. 12. Measured class-G modulator output voltage when driven by a real-world LTE signal.

Fig. 11. Measured power-added efficiency versus output power of the

ET system using a class-G modulator (solid) and the PA with constant supply voltage (dashed) driven by a sinusoidal input signal.

10 15 20 25 30 dBm 400

10

20

30

40

50

60

%

80

Output Power Pout

Pow

er-

added e

ffic

iency

ET

w/o ET

Fig. 10. Class G modulator (top right) and RF PA (bottom left).

LS8d.3-4

author’s knowledge, this is the highest efficiency

achieved by an ET system using a class-G modulator with

output powers above 1 W. The increase in efficiency

comes at the cost of lower linearity. ACLR decreases by

about 10 to 15 dB in the first and second sideband. EVM

degrades as well. This is mainly due to the drain bias

dependency of the RF PA gain visible in Fig. 4. As for

testmodel 3.1 significantly fewer transitions between

supply voltages occur due to the signal statistics, EVM is

better than for testmodel 1.1. In any case, for operation in

a communications system, the input signal needs to be

predistorted, depending on whether uenv is 10 V or 28 V.

However, this is not investigated in this publication.

V. CONCLUSION

This paper has given an overview of several voltage modulator concepts for ET operation. The potential of ET is shown by presenting measured data of a system using a GaN RF PA and a class-G voltage modulator. For a sinusoidal input signal, backoff efficiency is doubled and achieves up to 60 % at an output power of 34 dBm, compared to 30 % without ET. When driven by an LTE signal, PAE reaches about 38 %, which is an efficiency enhancement by almost the factor of two compared to a constant supply voltage.

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TABLE I

MEASUREMENT RESULTS WITH LTE SIGNALS

Testmodel 1.1 Testmodel 3.1

with ET w/o ET with ET w/o ET

PAPR 11.5 dB 12.5 dB

Pout 29.5 dBm 27.7 dBm

PAE 38.1 % 21.2 % 32.2 % 17.8 %

ACLR1

upper -18.1 dB -27.2 dB -16.4 dB -28.0 dB

ACLR1

lower -18.3 dB -27.8 dB -17.0 dB -28.3 dB

ACLR2

upper -27.7 dB -43.2 dB -26.3 dB -44.8 dB

ACLR2

lower -28.5 dB -44.0 dB -27.0 dB -45.5 dB

EVM 33.0 % 10.5 % 31.6 % 8.8 %

LS8d.3-5