N.E.D. University of Engineering and Technology, Pakistan

PEAK POWER TRACKING SYSTEM

WITH OVERLOAD PROTECTION AND

CHARGE CONTROLLER FOR WIND

TURBINES

DEPARTMENT of ELECTRONIC ENGINEERING

ABDUL RAFAY (EL-020)

MUHAMMAD UMAIR SALEEM (EL-032)

BILAL RIASAT ALI KHAN (EL-057)

ZAHID HUSAIN TURI (EL-006)

Bachelor of Engineering – Thesis for final year project

[Results, programming code and observation plus some crucial data

has been removed for the sake of maintaining the original design,

however will be furnished upon request : for online display only]

Objective:

In this project, we developed a fully functional system for small scale stand alone wind

turbines in which we have implemented peak-power-tracking functionality, targeted at

more efficient power capture and battery charging at low wind speeds. The system in effect

converts a fixed speed wind turbine into a variable speed one. The system also includes an

electronic braking system for overload protection and a simple inverter to directly plug-in

the ac load. During this project, we strived to identify the best possible strategies to

integrate such functionalities at the lowest possible cost through the application of 'Value

Engineering'. Our research aimed at simpler yet reliable control techniques and their

implementation, sensor-less control and efficiency improvements. The integration of all

these features will result in a new class of small but 'smart' wind turbines targeting the

unmet need of efficient and cheap electricity generation by the general public throughout

the world.

Wind Energy Basics:

A wind turbine extracts kinetic energy from the swept area of the blades. The power in the

airflow is given by

Pair = ½ρAν3

where

ρ = air density, kg/m3

A = swept area of rotor, m2

ν = upwind free wind speed, ms-1

the power transferred to the wind turbine rotor is reduced by the power coefficient,

CP = Pwind turbine/Pair

A maximum value of CP is defined by the Betz limit, which states that a turbine can never

extract more than 59.3% of the power from an air stream. In reality, wind turbine rotors

have maximum CP values in the range 25–45%. It is also conventional to define a tip-speed

ratio, λ, as

λ = ωR/ν

where

ω = rotational speed of rotor

R = radius to tip of rotor

ν = upwind free wind speed, ms-1.

Figure below shows that the maximum power coefficient is only achieved at a single tip-

speed ratio and for a fixed rotational speed of the wind turbine this only occurs at a single

wind speed. Hence, by operating a wind turbine at variable rotational speed, it is possible to

operate at maximum CP over a range of wind speeds.

Variable-speed operation of wind turbines is desirable because below rated wind speed, the

wind turbine rotor can extract the most energy if the tip speed ratio can be kept constant,

requiring that the rotor speed vary with the wind speed. Variable-speed operation of the

turbine’s rotor can result in reduced fluctuating stresses, and hence reduced fatigue, of the

components of the drive train. If the speed of the rotor can be continuously adjusted such

that the tip speed ratio remains constant at the level which gives the maximum CP then the

efficiency of the turbine will be significantly increased [7].

The power output of a wind turbine at various wind speeds is conventionally described by

its power curve, shown below:

The speed control requirement of the rotor has five separate regions:

1. The cut-in speed at which the turbine starts producing power. Below this speed, it is not

efficient to turn on the turbine.

2. The constant maximum Cp region where the rotor speed varies with the wind-speed

variation to operate at the constant TSR corresponding to the maximum Cp value.

3. During high winds, the rotor speed is limited to an upper constant limit based on the

design limit of the system components. In the constant speed region, the Cp is lower than

the maximum Cp, and the power increases at a lower rate than that in the first region.

4. At still higher wind speeds, such as during a gust, the machine is operated at constant

power to protect the generator and the power electronics from overloading. This can be

achieved by lowering the rotor speed. If the speed is decreased by increasing electrical

load, the generator will be overloaded, defeating the purpose. To avoid the generator

overloading, some sort of brake, eddy current, or other type, must be installed on the rotor.

5. The cut-out speed. Beyond certain wind speed, the rotor is shut off producing power in

order to protect the blades, the electrical generator, and other components of the

systems.[8]

Maximum Power Operation

As seen earlier, operating the wind turbine at a constant tip-speed ratio corresponding to the

maximum power point at all times can generate 20 to 30 percent more electricity per year.

However, this requires a control scheme to operate with variable speed. Two possible

schemes used with the variable speed operation are:

1. Constant Tip-Speed Ratio Scheme:

This scheme is based on the fact that the maximum energy is extracted when the optimum

tip-speed ratio is maintained constantly at all wind speeds. The optimum TSR is a

characteristic of the given wind turbine. This optimum value is stored as the reference TSR

in the control computer This scheme has a disadvantage of requiring the local wind speed

measurements, which could have significant error particularly in a large wind farm with

shadow effects. Being sensitive to the changes in the blade surface, the optimum TSR

gradually changes. The computer reference TSR must be changed accordingly many times

over the life.

2. Peak Power Tracking Scheme(preferred)

The power versus speed curve has a single well-defined peak. If we operate at the peak

point, a small increase or decrease in the turbine speed would result in no change in the

power output, as the peak point locally lies in a flat neighborhood. Therefore, a necessary

condition for the speed to be at the maximum power point is as follows:

dP/dω = 0

The speed is increased or decreased in small increments, the power is continuously

measured, And ΔP/Δω is continuously evaluated. If this ratio is positive, meaning we get

more power by increasing the speed, the speed is further increased. On the other hand, if

the ratio is negative, the power generation will reduce if we change the speed any further.

The speed is maintained at the level where ΔP/Δω is close to zero. This method is

insensitive to the errors in local wind speed measurement, and also to the wind turbine

design. It is, therefore, the preferred method. This principle is used in the control scheme

below [1]:

Proposed System:

1. Hardware Description:

The proposed system is a peak-power-tracker for small scale wind turbine, a feature commonly

found only in state-of-the-art MW-class wind turbines. It is intended to be a plug-and-play

module that includes an electronic braking system for overload protection and a simple inverter,

all integrated in a single package. The block diagram is shown overleaf.

1.1. Theory of Operation:

A MPPT (maximum power point tracker) is an electronic DC to DC converter that optimizes the

match between the output of the wind turbine and the battery bank. It converts a higher voltage

DC output from a wind generator down to the lower voltage needed to charge batteries. This is

because at a high voltage, only small current can be obtained from the fixed power of a turbine at

a certain speed. The controller looks at the output of the turbine, and compares it to the battery

voltage. It then figures out what is the best power that the turbine can put out to charge the

battery. It takes this and converts it to a voltage level to get maximum current into the battery,

thus increasing efficiency. The output of the MPPT charge controller might vary continually to

adjust for getting the maximum amperes into the battery.

The DC/DC converter used is a buck converter which means it takes a higher input voltage and

converts it to a lower output voltage. Since this a switching converter topology it doesn’t

dissipate any power internally (except for some small resistive losses). That means the output

power is equal to the input power. So if the watts stay the same and the voltage drops then output

current must be greater than the input current. NMOS is used for the MOSFET switch because of

a lower Rdson which means there is less resistive losses in the switch. However, this causes

problems with driving the high side MOSFET. To fully turn on an NMOS, we must have a gate

to source voltage (Vgs) of roughly 8V or greater. The high side switch in the DC/DC converter

has its source pin tied to the input voltage so we have to generate a gate drive voltage of at least

8V higher than the input voltage. This is taken care of by the MOSFET gate driver IR2110, one

of the most challenging potions of the hardware!

The microcontroller, a PIC16F877A, controls the conversion ratio of the of the DC/DC

converter. The PIC generates a 200kHz PWM signal with its internal PWM circuit. The duty

cycle of the PWM signal sets the ratio of the on time for the high side MOSFET switch versus

the on time of the low side MOSFET switch. The ratio of the on time of the switches sets the

conversion ratio of the input to the output voltage of the DC/DC converter. The PIC tries to set

the conversion ratio of the DC/DC converter to allow the wind turbine to operate at the

Maximum Power Point. The microcontroller does this using an iterative algorithm to maximize

the output power of the turbine. The output power is calculated by measuring the voltage and

current with the PIC’s A/D inputs and multiplying internally to get the power. The output voltage

runs through a resistor divider network to get it down in the 5v range of the PIC’s A/D converter.

The output current is measured with a current sense resistor and difference amplifier to condition

the signal before it is read by the PIC’s A/D.

Finally, to drive loads directly, we needed an inverter. The inverter used is a simple BJT based

half-bridge square wave inverter using CD4047 for PWM generation. The output is connected to

a 24V to 240V step-up transformer.

1.2. Circuit Description:

1.2.1. Transformer, Rectifier and Filter:

The need for a step-down transformer arose for effective isolation as well as to reduce the

voltage ratings of all the components. Initially, we started with a 240V-12V step-down

transformer keeping in mind a single battery of 12V to charge but later, to reduce the current

ratings and the hazards associated with high currents, we switched to a 240V-48V step-down

version with adequate power ratings.

A rectifier had to be used before the DC/DC converter stage along with a low-pass filter to

smooth the dc input to the Buck regulator. A full-bridge rectifier and capacitor filter were used

due to the following reasons:

1. A bridge rectifier will rectify both halves of a sine wave and give "continuous output" through

360 degrees of the input and you don't need a center-tapped transformer to use it.

2. Because its PIV rating is less than half wave rectifier.

3. It is more preferable over center tapped rectifiers because you don’t have to use special centre

tapped transformer that has larger secondary windings thereby reducing the size and cost.

4. Because it is small in size and cost effective.

5. Inductors block high-frequency signals and conduct low-frequency signals, while capacitive

filter do the reverse.

For a small ripple in voltage and due to low frequency (60Hz), high current and small load, three

4700uF capacitors were used in parallel for a total filter capacitance of 14.1mF.

1.2.2. DC/DC Converter:

The PPT circuit is divided into two basic parts: the DC/DC converter and the microcontroller

control. The high side MOSFET is IRFZ44N. The inductor value is 42uH and it is sized to

handle 11A. The capacitor is 330uF and the diode used is Schottky diode for low power losses.

NMOS is driven by the IR2110 gate driver. It drives the high side MOSFET using the PWM

signal from the microcontroller. A diode is used to keep the battery current from flowing into the

system.

Initially, we designed the buck converter for a nominal output of 12v but later shifted to 24v.

Also, the PWM frequency was initially set to 5kHz, then increased to 10kHz and later to 100kHz

to improve efficiency and reduce component size. Finally, due to the limitation of inductor size

and value, the value was set at 200kHz. Accurate calculations were done using relations from

[11] for critical values of inductor and capacitor for continuous conduction mode, nominal,

minimum and maximum duty cycle and power dissipations for an assumed efficiency of 90%.

The whole design was simulated and experimented with by varying various parameters in

PSPICE.

Deciding and finding the power MOS, schottky diode and capacitor was not hard but the

inductor proved to be real challenge. For the frequency of 5kHz, the value of inductor was

calculated to be really high and was even raised further to smooth out the output current (14mH).

No inductor of this value that could handle 20A (at 12v output) was available. We started off

with winding an air-core inductor ourselves but the wire gauge to be used was low (thick wire).

Winding it was really hard and at the end we got only a few milli-henrys. We then looked for a

non-air core inductor. We bought a transformer which gave us approximately 14mH but it was

expensive and very bulky. Later, when we decided to increase the frequency, and reduce the

current, the requirements of the inductor were relaxed to 33uH and 10A. We found a torroidal

ferrite core and wound an inductor with parallel wires to reduce skin effect and ease of winding

(we still needed 10A rating so the wire was still thick). The inductor wound had an inductance of

42uH.

1.2.3. Gate Driver:

When an NMOS is used as the high side switch the gate voltage to drive it must be at 10V

greater than the source voltage. Since the source voltage is the input voltage on the high side

MOSFET, a high-side gate driver IC was required. The gate driver chip must generate a higher

voltage that the input voltage to turn on the high side switch. This is achieved by the bootstrap

circuit that increases the input voltage so the high switch can be driven on. The gate driver IC we

finally used was IR2110. It is powered by a 7812 linear regulator while the PWM is provided by

the PIC at TTL logic level. The minimum bootstrap capacitance for the operating frequency was

calculated and a 1uF capacitor was used.

Running the gate driver was the biggest challenge in the project. Everything worked well in

simulations but the same circuit didn’t appear to work physically. After many attempts, we tried

using PMOS along with a totem-pole BJT configuration in place of NMOS and thus eliminating

the need for the gate driver IC. We even tried a pulse-transformer but rejected it because of its

poor performance at high frequency. We switched back to the gate driver IC and after a lot of

study and changes, we were able to successfully operate it at 200kHz. The biggest mistake that

we were making was not connecting the gate driver and power grounds together.

1.2.4. The Microcontroller, overload protection and feedback circuitry:

The DC/DC converter is controlled by the PIC16F877A microcontroller. The microcontroller is

clocked at 20Mhz by a crystal. The 5V to power the PIC16F876 is generated by a 7805 linear

voltage regulator. The PWM output of the PIC is used to control the duty cycle of the DC/DC

converter which sets its voltage conversion ratio. The frequency of the PWM is set to 200Khz by

the PIC software. The PWM duty cycle is controlled by the PIC software to optimize the power

output from the wind turbine.

The PIC calculates the power output generated by reading the voltage and current of the battery

through its ADC using a voltage divider used to drop the input voltage into the 5v range and a

combination of current sense resistors. The small voltage generated by current sense resistors is

amplified and buffered by an LM358 op-amp. The op-amp is configured as a difference

amplifier.

The controller also operates the overload protection circuitry by turning on a relay through a BJT

and connects the transformer directly to dummy loads to load the turbine and reduce its output

voltage, thus preventing damage to the electronic circuitry. The controller displays duty cycle

and voltage, current values on a 16x2 LCD display.

1.2.5. Charge Controller:

One of the major proposed elements of the project that was incorporated since the inception of

the project, early this year, was the charge controller circuitry.

From uC

Vi

n

The charge controller is a block that maintains a desirable voltage at the battery terminals and

allows for the current from the previous block of the circuit to pass on successfully to the battery.

The charge controller is a combination of circuit elements that provide a current steering or

current sinking application from the preceding stage onto the final dumping ground that is the

battery or battery bank. In our criterion we had many aspects of charge controller topologies to

work and think about but ultimately we made two choices that proved them to be redundant and

excessive to the idea of Maximum Peak Power Generation. It is only fair that when describing

the charge controller circuitry we give significant knowledge about the type of battery utilized

for our project.

Following is the detailed description of work by work and phase by phase adaptation of the

project’s charge controller circuitry along with ample information about the batteries it was

meant to charge which will ultimately revert the discussion to the final conclusion regarding this

block.

BATTERIES…….. The charge banks

Although batteries include a huge variety of choices to work upon, we needed to choose at least

one that fits our suit of operation and is easy to gets hands on. Possible candidates for the battery

choices included deep cycle batteries and short cycle starter batteries.

The batteries described above have two or three differences but altogether their performance is

the same if their criterion of work is considered.

Starting Batteries, sometimes called SLI, for starting lighting ignition batteries, are commonly

used to start and run engines. Engine starters need a very large starting current for a very short

time. Starting batteries have a large number of thin plates for maximum surface area. The plates

are porous, similar in appearance to a very fine foam sponge. If deep cycled, this sponge will

quickly be consumed. Automotive batteries will generally fail after 30-150 deep cycles if deep

cycled, while they may last for thousands of cycles in normal starting use (2-5% discharge),

which makes their normal workable life to about 12 months, and for a industrial price of $80 -

$90

Deep cycle batteries are designed to be discharged down as much as 80% time after time, and

have much thicker plates. The major difference between a true deep cycle battery and others is

that the plates are SOLID Lead plates. This gives less surface area, thus less "instant" power like

starting batteries need. Although they can be cycled down to 20% charge, the best lifespan

against cost method is to keep the average cycle at about 50% discharge, which is unlikely for

many usage norms. They can be bought for a price of $317

The conclusion drawn from the above statements is that, except for the dis-charging and Cold

Crank Current delivery, the batteries exhibit the same performance in terms of charging and

charge delivery upon usage. So the team decided to go with the starter batteries and build a

circuitry over the principles that ensure the same working behavior for either starter or deep

cycle batteries. The workable prospect of our project was the ease of availability and usage for

the user, this was a strong influencer for us to incline towards the starter batteries that are used

on a day to day basis by everyone who owns an automobile.

Charging

Charging the battery required to build a topology that allows for a preset value of voltage to

appear on the output at the battery terminals which is generally higher than the terminal potential

at any stage, and allows for a path of current into the battery achieved by a very low internal

resistance of the battery (order of mili ohms)

The charge time of a sealed lead acid battery is 12–16 hours and up to 36–48 hours for large

stationary batteries. With higher charge currents and multi-stage charge methods, the charge time

can be reduced to 10 hours or less; however, the topping charge may not be complete. The three

possible charging techniques for a lead acid battery are [1] constant-current charge, [2]topping

charge and [3] float charge. The constant-current charge applies the bulk of the charge and takes

up roughly half of the required charge time; the topping charge continues at a lower charge

current and provides saturation, and the float charge compensates for the loss caused by self-

discharge. A figure showing the effect of each charge topology on the battery cell is shown

below: Figure(1.1)

During the constant-current charge, the battery charges to 70 percent in about 5–8 hours; the

remaining 30 percent is filled with the slower topping charge that lasts another 7–10 hours. If

deprived, the battery will eventually lose the ability to accept a full charge and the performance

will decrease due to sulfation. The float charge in the third stage maintains the battery at full

charge.

The switch from Stage 1 to 2 occurs seamlessly and happens when the battery reaches the set

voltage limit. The current begins to drop as the battery starts to saturate, and full charge is

reached when the current decreases to the three percent level of the rated current. A battery with

high leakage may never attain this low saturation current, and a plateau timer takes over to

initialize the charge termination.

The correct setting of the charge voltage is critical and ranges from 2.30 to 2.45V per cell. On

one hand, the battery wants to be fully charged to get maximum capacity and avoid sulfation on

the negative plate; on the other hand, an over-saturated condition causes grid corrosion on the

positive plate and induces gassing.

The discussion above leads to the idea of smart charge controller, one that modifies its output in

between operation and provides for a maximum battery life. The available topologies in smart

charger applications are

Constant voltage variable current

Constant current variable voltage

Float charge

The selected methodology of charging selected at the initial stage was constant voltage variable

current in addition with float charger.

This led to a circuit design which basically took in an input voltage of 12 volts (bare minimum

for one battery at least) to 35 volts, with current completely dependent upon the value of the

voltage fed in from the output of the BUCK regulator block. As the duty cycle is varied by the

means of Pulse Width Modulation Controller, the gate driver acts suit and opens the buck

regulator as according to the signal received on its gate. This allowed for a varying signal to be

fed across the battery terminals for the charging to occur.

Two of the major designs adopted for the charge controller are defined below:

Make-shift smart charge controller using linear voltage regulator

Smart charger (3 stages of charge) using UC3906

Charge Controller

To justify the use of two stage battery charging we adopted for the charger in the initial stage of

constant voltage variable current we must define the state of charge of the battery against the

terminal potential difference at every state of charge.

The graphs show a variety of recharge and discharge rates from C/5 to C/100. This C/XX

number is actually a rate of charge or discharge in Amperes proportioned to the capacity of the

battery. In our design system we adopted the C/10 factor. This ‘C’ is actually the rated battery

capacity and this factor determines the current pumped into the battery without damaging its

terminals. The battery modeled for use in our prototype is a 200Ah battery that makes the C/10

value to be 20 Amps. Although, this value was a subject to revision for testing purpose, so we

scaled down the value to approximately 22% and used the new value of 44Ah. Further

modifications in the battery bank will be discussed later.

Another significant assumption in the build up to the charge controller was the battery state of

charge against the internal resistance of the flooded lead acid battery. In lead acid cells, the

electrolyte (sulphuric acid) participates in the cell’s normal charge/discharge reactions. As the

cells are discharged, the sulfate ions are bonded to the plates — sulfuric acid leaves the

electrolyte. The process is reversed when the cell is recharged. A fully charged lead-acid cell has

an electrolyte that is a 25% solution of sulfuric acid in water (specific gravity about 1.26). A

fully discharged lead-acid cell has virtually no sulphuric acid in it but almost pure water

electrolyte (specific gravity about 1.00). As the sulfuric acid concentration in the electrolyte

changes so does the electrical resistance of the electrolyte, which in turn changes the internal

resistance of the entire cell. The bottom line is that the internal resistance of all lead-acid cells

changes with the cell’s state of charge. This character gives the lead-acid reaction its particular

signature on the voltage vs. SOC graphs. This signature is unique — very different from alkaline

cells whose electrolyte resistance remains constant regardless of SOC.

Hence ultimately the final conclusion drawn from the article stated above is that as a battery

charges, its internal resistance increases and as the battery discharges, its internal resistance

lessens, which creates a huge dumping ground or current sink for the incoming signal from the

battery charger.

1) Constant Voltage Variable Current and Float Voltage Charging

For the principle of constant voltage usage, the linear regulator felt like a suitable choice for the

proposed scheme of charge controller

As described from the cell voltage figure above Fig(1.1), the constant voltage variable current

scheme sees the charger maintaining a 13.X volts (X values need to greater than 5) and steering

available current from the buck regulator output into the battery through the charger circuit. On

the other hand the float voltage charger scheme will require the charger to maintain a constant

14.1 volt to 14.3 volt and limit the current as the battery charges to near 100 percent state of

charge.

The very first choice for this design circuit was the adjustable voltage regulator LM317. But this

idea was soon dropped because of its hardware limitations and operation skew when a variable

resistor at the adjust pin of the IC was replaced with a fixed resistor. This behavior was the main

reason for abandoning this idea, because apart from that it worked very well with the two stages

of smart charging applications namely constant voltage variable current and float charging, but to

even achieve these two stages, this work needed to be done manually with a turn dial on the

variable resistor. Hence the word “smart”, lost its meaning.

The next immediate option was the adaptation of the linear fixed voltage regulator for the charge

controller scheme. The second design comprised of an LM7815 IC. This fixed linear voltage

regulator provided a desired fixed value of voltage against current that needed to be fed to the

battery for charging. A simple applied logic is shown as under

Q1 allows for a high current steer path from the input onto the output and the voltage regulator in

addition with this topology provides a constant voltage at the battery terminals. Considering this

way, the smart charger was to be formed with two dedicated LM7815 voltage regulators that

work in parallel to each other. One circuit works on the constant voltage variable current and the

other circuit works on the basis of float voltage. First Circuit provided 13.5 to 13.8 volts on the

output to the battery terminals and the second one provides a float voltage of 14.1 volts to the

battery terminals. These two charge circuits were to controlled via smart switching which

depended upon the time sharing principle. The interrupt generator was 555 timer, forming a

check status pulse at near 50 percent duty cycle, where the entire work cycle was defined for 10

minutes. As the switched were altered in their opening stands, the battery was monitored via a

comparator. This comparator formed a reference voltage for itself and determined what switch

truth table will be depending upon the open circuit voltage of the battery terminals.

The bulk charger provided a C/10 current (in this case around 20 Amps), which for test criterion

was scaled down to 5 Amps. And the trickle charger provided a float voltage at the terminals and

drives a current of near 1Amp (10% of C/10). The fold-back circuitry of the trickle charger

accounts for a decreased current into the load attached through a resistor capacitor bank.

A statistical chart of the readings taken from the constant voltage variable current charger on a

scaled down load model is given as under:

Terminal voltage

Current

(Amps)

10 5.65

10.5 5.12

11 5.05

11.5 4.85

12 4.85

12.5 4.78

13 4.5

13.xx 4.5

The above measurements were taken on a 70 percent state of charge battery and the updated

reading of the terminal voltage was noted for every 1 hour or so as a 0.5 volt increment in the

terminal voltage was noted.

Problem Statement

The major erroneous nature of the reading was observed when we came across the skewing

behavior of our design. As studied from the literature, it was noted that as the battery gets

charged the internal resistance alters or more specifically increases, this results in greater

terminal voltage drop and low current sinking abilities. However this suit is not followed right

down the very detail in actual practice. The battery was supposed to sink to draw a current of 4.4

Amps (in accordance with the scaled down model of 44Ah) but it was impervious to this, and the

value of current was maintained at a random. Hence we concluded that the only relation between

current drawn into the circuit should not be based upon the internal resistance of the battery, as

no two are the same.

Another problem that we encountered with the charger was that the use of the linear voltage

regulator outputted a fixed value on the battery terminals. This resulted in very sturdy 13.xx volts

on the charger regardless what value of voltage was input to it. This was a problem because this

clashed directly with our Maximum Peak Power Tracking theory. If the voltage is regulated at

the output always, then due to constant voltage at the output, the current will never vary (increase

or decrease) with varying input to the charger. Plus the high current steering transistors were

only working as an uncontrolled sink for the current produced with no control what so ever. This

was a highly undesired trait which was rectified later on.

Smart charger (3 stages of charge) using UC3906

The UC3906 works with varying the Pulse Width as opposed to the feedback it takes from the

battery itself. This active feedback which determines what state the battery is in results in a very

precise and ‘need to get basis’ current for the battery without damaging it. The comparators with

in the IC provide this functionality with the proper calculation of resistances attached with it on

the basis of required value of voltage thresholds, float values and continue charging current.

Defined with a preset of

voltages for three states of operation, except for this property, this regulator is no different.

Problem Statement

This regulator suffered from the same problem as its predecessor. Although it monitored the

battery through a feedback and determined the state of charge for every magnitude of current

required to be put in, this still does not compensate the fact that with the input voltage varying

the current will never be provided to the battery as per change in output energy. Since it only

drove the base of the drive transistor through pin 16 depending upon the feedback taken from the

battery, it never fully utilized the available output power and was not efficiency friendly.

Conclusion of Chargers

Due to the problem which restricted the full use of available output power, the charger topologies

were dropped out of the project and it was established that the BUCK regulator will provide the

batteries with a direct output. With the varying voltage at BUCK’s output which never goes

below the least value of the battery bank, up till the final value as achieved through varying duty

cycles, the battery bank will be able to utilize a full output power with no regulation what so ever

on the terminal voltage.

1.2.6. Inverter:

In the last progress meeting with the committee, the added work of inverter block was given to

us as a part of our final year project. This block greatly increases the user proposition value for

the entire system.

Inverter topologies were studied and analyzed and the most suitable configuration as per time

restraints and simplified logic was implemented.

Topology Selection

The first step in the inverter design was the selection of the design criterion. The strongest

contenders for this try out were

MOSFET based Inverter

BJT based inverter

Although to be noted here that both the above mentioned topologies make the use of “Half

Bridge” technique for inverters. It should be declared that the mosfet based inverters provide

superior performance in terms of less consumed power and good switching frequency handling

with proper gate operation for putting the mosfet into saturation and cutoff depending on what

input is applied. Although there some reasons which led to the dropping of this topology and

adaptation of the simple BJT based inverter design.

The first design prototype for the inverter comprised of the MOSFET based inverter design. A

block diagram is given as under:

The switching frequency module was composed of many different alterations of calculation at

different times. Most commonly adapted ICs were LM555 timer, CD4017, and CD4047.

A switching frequency thus set this way was responsible for driving the MOSFET inverter load.

Problem Statement

One of the biggest problems we came across in this topology was the MOSFET driving through

the gate driver circuitry. The MOSFETs were chosen to be IRFZ44N with Id=49 Amps and a

power of approximately 150 watt. The gate driver IC that was chosen for this purpose was the

IR2110, as chosen the same by the BUCK regulator. The IR2110 was to work in its push pull

configuration, which will allow for a ground path onto the load via the drain tapping of the

MOSFET onto the transformer. A push pull configuration is given as under:

the push pull configuration was implemented for testing stages but the results were satisfactory.

The IR2110 is very sensitive to its component ratings that enable the push-pull mode. Even the

slightest tolerance in capacitance and resistance throws the results off. The times when it even

did work, the inverter couldn’t bear the output load and was subject to a short circuit.

This was a major problem considering the time constraints we were already in regarding the fyp

timeline and the sudden inclusion of the inverter block into the project. So the team decided to

work with the time tested and simple configuration of BJT inverters.

BJT Topology

The bjt inverter design comprises of a switching frequency component and a a transistor

cascading as shown in the block diagram below, providing a ground path onto the load via the

collector tapings of the transistors with the center tapped transformer.

The only major difference between this inverter criterion and the MOSFET based inverts was

that the part of the gate driver IC was secluded in case of BJT and the circuitry and greatly

simplified.

Basically what is happening in the circuit is that the IC CD 4047 wired as an astable

multivibrator produces two 180 degree out of phase 1/50 Hz pulse trains. These pulse trains then

are pre amplified by the two D313 transistors. The outputs of the D313 transistors are input to

the 2N3055 transistors to drive the transformer’s ground switch path. The 220~240V AC will be

available at the secondary of the transformer

Component Details

IC CD 4047

This IC is the heart of the circuit. It generates square-wave at the output. The square-wave is

always at 50% of duty ratio. The frequency of this IC can be set by vary the resistor and the

capacitor. Pin 1 is connecting the IC and external capacitor. This external capacitor as known as

timing capacitor should be connected between pin 1 and RC common terminal, pin 3. The

external resistor also known as timing resistor, should be connected between pin 2 and RC

common terminal, pin 3.

This IC has 2 mode, astable and monostable. In this project the mode used is astable. To apply

this mode pin 4 should ground or pin 5 is connected to supply voltage. Pin 14 is the supply

voltage for the IC and pin 7 is for ground. The output is attained at pin 10 and 11, where at pin

11 the output is 180° from the output at pin 10. Pin 9, 12 and 13 is the special feature of this IC.

If the input is high at pin 9, no output pulse at pin 10 and 11 during ON power condition. For

UPS this feature enable this inverter to turn off during normal condition. When fault happen the

control unit should sending low input to this pin to permit this inverter to work. Pin 12 is for

retriggering the waveform. In UPS system it not really compulsory. Pin 13 the connection of this

IC to other IC to synchronize.

D313 and 2N3055

The operation of these two transistors is not very different from each other. The D313 transistor

makes use of the fact that the input signal from the CD4047 is pre-amplified before putting into

the base of the switching ground transistors on 2N3055. D313 outputs a signal suitable enough

for proper saturation of the base of 2N3055 in conduction mode.

A figure showing the output of D313 transistor is given as under

<CRO diagram>

Transformer

The final output of the inverter is a square wave AC signal taken across the secondary windings

of the 12-0-12 center tapped transformer. The windings of the transformer are made with

accordance of the power in : power out ratio. As the power input to the transformer is 250 watt,

because of the total power content of the system. This power drives a current of 250 watt/24

volts which comes out to be approximately 11 Amps. For safety measures the AWG wire chosen

is of greater power wattage. The wire chosen is AWG 14.

Conclusion of the Inverter

The inverter design selected proves to be a very simple yet powerful topology over the MOSFET

based design. However the BJT inverter provides some sort of advantage as well. The major

advantage it gives is in the form of avoiding shoot-through fault and less losses due to low

switching frequency. Shoot through fault happens in the half bridge configuration when both the

transistors are simultaneously conducting, which shows a short across the DC rail.

The inverter circuit though successfully built for UPS application, but for UPS application there

are many other aspects needed to be considered. The circuit built is nothing when compared with

inverter using PWM. PWM based inverters show great decrease in the Total Harmonic

Distortion factor and produce high quality sine wave.

The circuit built has certain advantages. First it avoided the dead-through fault since it is working

on lesser switching frequency. Second, the IC has certain functions that allow the circuit to

operate in UPS system. Like synchronizing with other controller and the output can be controlled

on and off.

2. Software Description:

[bilal]

3. Simulation Results:

3.1. Changing duty cycle from 45 to 65% at Vin=48V, Rload=0.5ohms:

3.2. Changing Rload from 0.25 to 0.75ohms at Vin=48V and 55% duty cycle:

3.3. Excellent Line Regulation! (Vin from 36 to 60V, 55% duty cycle, Rload=0.5ohms):

3.4. Load Regulation (Vin=48V, 55% duty cycle):

From graph below:

Change in Vo = 25.584 - 23.164 = 2.42

Change in Io = 9.673 - 4.437 = 5.236

Load regulation = 2.42/5.236 = 0.46 V/A

3.5. Step change in input (Vin = 60 to 48V at t = 50ms):

4. Test Results:

5. Conclusion:

Bibliography

Web:

1. http://earthtrends.wri.org/updates/node/277

2. http://englishabc.wordpress.com/2008/03/11/the-leading-wind-energy-trade-fair-2008/

3. http://wings.buffalo.edu/ubgreen/energyforthefuture/technologies/wind.htm

4. http://environmentengineering.blogspot.com/2008/04/wind-energy-renewable-energy-

by.html

5. http://saferenvironment.wordpress.com/2008/11/03/wind-energy-renewable-energy-

harnesses-natural-wind-power-%E2%80%93-effective-answer-for-emission-problem-

towards-cleaner-safer-and-greener-environment/

6. http://science.howstuffworks.com/wind-power.htm/printable

7. http://www.otherpower.com/bottom_line.shtml

8. http://en.wikipedia.org/wiki/List_of_countries_by_carbon_dioxide_emissions

9. http://en.wikipedia.org/wiki/List_of_countries_by_ratio_of_GDP_to_carbon_dioxide_

emissions

Respective websites of:

A. Proven Energy

B. Skystream

C. Wind Energy Solutions

D. Bergery Energy Systems

E. Whisper Energy

F. International Energy Commission

G. Wikipedia

H. World energy Report 2008, 2009

I. Wind Statistics Journal Pakistan

J. GOOGLE!

Papers:

1. Maximum Power Point Tracking Based Optimal Control Wind Energy Conversion

System - Ahmad Nadhir & Takashi Hiyama - Kumamoto University, JAPAN

2. Design and Analysis of a Microprocessor-Controlled Peak-Power-Tracking System

by P. Huynh and B.H. Cho, Virginia Polytechnic Institute

3. Voltage Controller of a Single Phase Self-Excited Induction Generator

4. R. Ahshan and M.T. Iqbal - Faculty of Engineering and Applied Science, Memorial

University of Newfoundland, Canada

5. Voltage Controller For Stand-Alone Induction Generator Using Instantaneous Power

Control - G.V.Jayaramaiah & B.G.Fernandes - Indian Institute of Technology

Bombay, INDIA

6. Microprocessor-Based Voltage Controller for Wind-Driven Induction Generators -

N. AMMASAIGOUNDEN AND M. SUBBIAH

7. Cage Induction Generators for Wind Turbines with Power Electronics

8. Converters in the Light of the New Grid Codes - Marta Molinas, Bjarne Naess,

William Gullvik, Tore Undeland - NORWEGIAN UNIVERSITY OF SCIENCE

AND TECHNOLOGY

9. DYNAMIC ANALYSIS OF A VARIABLE SPEED WIND TURBINE EQUIPPED

WITH A VOLTAGE SOURCE AC/DC/AC CONVERTER INTERFACE - F.D.

Kanellos S.A. Papathanassiou N.D. Hatziargyriou M.P. Papadopoulos - National

Technical University of Athens

10. Output Voltage Control of PWM Inverters for Stand-Alone Wind Power

Generation Systems Using Feedback Linearization - Dong-Choon Lee & Jeong-Ik

Jang - Yeungnam University, Korea

Texts:

[1] Wind and Solar Power Systems by Mukund R. Patel

[2] Wind Power in Power Systems - Thomas Ackermann

[3] Power Electronics Handbook - M. Haroon Rashid

[5] WIND ENERGY EXPLAINED Theory, Design and Application - Second

Edition by J. F. Manwell, J. G. McGowan and A. L. Rogers

[6] Wind Turbine Technology by A.R. Jha

[7] WIND ENERGY HANDBOOK - Tony Burton, David Sharpe, Nick Jenkins,

Ervin Bossanyi

[8] WIND ENERGY GENERATION Modelling and Control - Olimpo Anaya,

Nick Jenkins, Janaka Ekanayake, Phill Cartwright, Mike Hughes

[9] POWER ELECTRONICS FOR MODERN WIND TURBINES by Frede

Blaabjerg and Zhe Chen

[10] THE WIND TURBINE COMPONENTS AND OPERATION by Henrik

Stiesdal

[11] Pulse-width Modulated DC–DC Power Converters, MARIAN K.

KAZIMIERCZUK