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OPTICAL MAGNETIC FIELD PROBE WITHLIGHT EMITTING DIODE SENSOR (RADIO
FREQUENCY, FARADAY'S LAW, INCANDESCENT,TEMPERATURE COMPENSATION, INDUCTRON)
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T T-\yf-T Dissertation U1V11 Information Service University Microfilms International A Bell & Howell Information Company 300 N. Zeeb Road, Ann Arbor, Michigan 48106
1328508
Gross, Eugene Joseph
OPTICAL MAGNETIC FIELD PROBE WITH LIGHT EMITTING DIODE SENSOR
The University of Arizona M.S. 1986
University Microfilms
International 300 N. Zeeb Road, Ann Arbor, Ml 48106
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OPTICAL MAGNETIC FIELD PROBE WITH LIGHT EMITTING DIODE SENSOR
by
Eugene Joseph Gross
A Thesis Submitted to the Faculty of the
DEPARTMENT OF ELECTRICAL AND COMPUTER ENGINEERING
In Partial Fulfillment of the Requirements For the Degree of
MASTER OF SCIENCE WITH A MAJOR IN ELECTRICAL ENGINEERING
In the Graduate College
THE UNIVERSITY OF ARIZONA
1 9 8 6
STATEMENT BY AUTHOR
This thesis has been submitted in partial fulfillment of requirements for an advanced degree at The University of Arizona and is deposited in the University Library to be made available to borrowers under rules of the Library.
Brief quotations from this thesis are allowable without special permission, provided that accurate acknowledgment of source is made. Requests for permission for extended quotation from or reproduction of this manuscript in whole or in part may be granted by the head of the major department or the Dean of the Graduate College when in his or her judgment the proposed use of the material is in the interests of scholarship. In all other instances, however, permission must be obtained from the author.
SIGNED:
APPROVAL BY THESIS DIRECTOR
This thesis has been approved on the date shown below:
CKoger C.fOatfes Professor of Electrical\md Computer Engineering
ACKNOWLEDGEMENTS
The author would like to express sincere thanks and
appreciation to Dr. Roger C. Jones, Dr. Thomas C. Cetas, and Dr. J.
Bach Andersen for their support and advice. Additional thanks to
Sally Anderson for secretarial assistance and to Anne Fletcher for
help with the illustrations.
i i i
TABLE OF CONTENTS
Page
LIST OF ILLUSTRATIONS vi
LIST OF TABLES ix
ABSTRACT x
1. INTRODUCTION 1
Optical RF Magnetic Field Measurement Systems 2
2. FUNDAMENTALS OF RF MAGNETIC FIELD MEASUREMENT SYSTEMS AND CALIBRATION * 9
Bench Calibration Procedures 11 Response of Sensing Loops to RF Magnetic Fields. . 11 DC Calibration of Incandescent Lamp Sensor .... 12 Attempt at DC Calibration of LED Sensor 17 Calibration with RF Current 21
3. PRACTICAL CONSIDERATIONS 28
Probe Sensor Construction 28 LED Sensor Protection 28 LED Temperature Sensitivity/Compensation 29 Details of Probe Sensor Construction 40
Optical Link and Photoamplifiers 45
4. CALIBRATION RESULTS AND SYSTEM CALIBRATION 52
Calibration Verification 52 Solenoidal Field Generation and Measurements ... 53 Comparison Between Probe Systems in a Magnetrode™
Applicator 57 Magnetic Field Measurements in a Current Strap . . 58
System Characteristics 65 Probe Linearity 68 Sensitivity and Dynamic Range 70 Probe Bandwidth and Impedance 73 Pulse Response 78 Thermal Stability and Optical Connection
Repeatability 81
iv
TABLE OF CONTENTS—Continued
v
Page
5. CONCLUSION 84
Summary 84 Future Considerations 86
APPENDIX A: THE MODEL EQUATION 88
APPENDIX B: TRIAL FORMULATION OF A DIRECT CURRENT CALIBRATION PROCEDURE FOR THE LED RF MAGNETIC FIELD PROBE 95
APPENDIX C: TEMPERATURE COMPENSATION PROGRAM 107
LIST OF REFERENCES 109
LIST OF ILLUSTRATIONS
Figure Page
1. Oleson's RF Magnetic Field Probe with Incandescent Field Lamp Sensor (Oleson, 1982) 3
2. System Diagram of RF Magnetic Field Probe with Light Emitting Diode (LED) 5
3. Response of LED to DC Excitation Currents 6
4. Non-optical RF Magnetic Field Measurement Sensors (Kanda et al., 1982) 10
5. Ideal Representation of the LED Sensor 13
6. Calibration Set-up for the Incandescent Probe System .... 15
7. Calibration Curve at 13.56 MHz for the Incandescent Probe System 16
8. Calibration of the LED probe at 13.56 MHz Using a DC Calibration Technique 19
9. RF Calibration Setup for Calibrating Both LED and Incandescent Versions of RF Magnetic Field Probe Systems . . 23
10. 13.56 MHz RF Calibration of the LED Probe System 26
11. 14.25 MHz RF Calibration of the LED Probe System 27
12. Experimental Setup for Determining Temperature Sensitivity of an LED 31
13. Graph Showing Temeprature Sensitivity of the LED. Output Versus Applied Loop Voltage 32
14. Linearization Circuit Used to Compensate LED Sensors .... 35
15. Early Sensor Schematic 41
16. Photograph of Early Sensors 42
vi
vi i
LIST OF ILLUSTRATIONS—Continued
Figure Page
17. Schematic of the Sensor with Empirically Arranged Temperature Compensation Components (17a). Component Placement for Mathematically Obtained Compensation (17b) . . * 43
18. Details of Component Placement for the LED Sensor 44
19. Schematic of CW Amplifier with Chopper Stabilized Amplifiers 47
20. Schematic of the Pulse Amplifier 49 -
21. Printed Circuit Foil Pattern for the Pulse Amplifier .... 50
22. Schematic of Solenoid and Resonating Components for Standard Field Generation 54
23. Physical Charactristies of the Standard Solenoid 55
24. Relative Field Strengths Internal to a Saline Loaded Magnetrode™ Hyperthermia Thigh Coil Applicator 60
25. Smith Chart to Calculate 450 ohm Transmission Line Length and Shorting Stub Length 62 '
26. Diagram of Current Strap Excitation Apparatus 63
27. Photograph of Standard Current Strap and Resonating Capacitor 64
28. Field Points Measured in the Current Strap 66
29. 13.56 RF Probe Calibration of the LED System with Linear Approximation Represented by the Solid Line 69
30. Output of the Pulse Amplifier Viewed on Oscilliscope .... 79
31. RF Excitation Waveform of the Current Strap Used to Determine LED Pulse Response Characteristics 80
32. Temperature Stability of the Empirically Compensated LED RF Magnetic Field Sensor 82
LIST OF ILLUSTRATIONS—Continued
vi
Figure Page
33. I-V Characteristics of a GaHlAs LED (Stanley ESBR 5701). . . 91
34. Agreement Between LED Data and the Curve Fit Model Equation for Currents up to 1.6 mA 92
35. Agreement Between LED Data and the Curve Fit Model Equation at Low Current Levels (below .16 mA) 93
36. Comparison Between LED Data and the Equation That Includes Temperature Sensitivity of the I-V Characteristics of the Junction 94
37. Setup with Hewlett Packard Data Acquisition System (HP DAS) to Test Accuracy of Calibration Integration .... 99
38. Schematic of Junction Block Shown in Figure 37 100
39. Prototype Probe Schematic with Various Test Points Used for Calibration and Sensor Testing 101
40. Currents Stepped Through with HP DAS to Determine the Accuracy of the Integration of the Transcendental Integral Equation (10 volt sinewave) 102
41. Currents Stepped Through with HP DAS to Determine the Accuracy of the Integration of the Transcendental Integral Equation (3.0 volt sinewave) 103
42. Currents Stepped Through with HP DAS to Determine the Accuracy of the Integration of the Transcendental Integral Equation (1.8 volt sinewave) 104
43. DC Calibration Currents and RF Magnetic Fields Predicted for the DC Calibration Attempt 105
44. More Complete RF Network Analysis of the Sensor Circuitry. . 106
LIST OF TABLES
Table Page
1. Comparison of the DC Calibrated LED Probe System and Incandescent Probe System 20
2. Comparison of the RF and DC Calibration on Oleson's Incandescent Probe System 24
3. Calculation of Average Temperature Coefficient From Figure 13 34
4. Component Values for Compensation of a -.0105/°C LED With -3.9%/°C Thermistors 39
5. Comparison of Calculated Solenoid Fields to Those Measured by the LED System and Oleson's Incandescent System 56
6. Comparison Between Oleson's Incandescent Probe System and the LED Probe System Within a Saline Loaded Magnetrode™ Hyperthermia Applicator 59
7. Comparison of Measured and Calculated Field Points Within the RF Current Strap 67
8. Probe Impedance as a Function of Frequency Measured Between Test Points 0 and 2 with a DC Bias Current of 3.5 Mi Hi amperes 75
9. Computed RMS LED Sensor Current and Associated Field Perturbation for Some Hypothetical Values of Magnetic Fields 77
10. Repeatability of Optical Fiber Connection on Amplifier Chassis 83
11. Agreement Between Measured and Calculated Average Currents . 98
ix
ABSTRACT
An isolated radio frequency (RF) magnetic field probe system
using fiber optics has been developed for both pulsed and continuous
wave (CW) fields, based on the principles demonstrated by Oleson's
earlier optical magnetic field probe system (Oleson, 1982). Improved
linearity and rise time are a consequence of incorporating a light
emitting diode in place of an incandescent source. For the pulsed
case, high slew rate pulse amplifiers were used, while chopper
stabilized direct current (DC) amplifiers were employed for the CW
case. Rather than using a standard field, the probe system can be
calibrated using bench instrumentation with an RF calibration
technique that has been developed and thoroughly tested. In addition,
the calibration technique developed can be used to more accurately
calibrate the earlier probe system design of Oleson.
x
CHAPTER 1
INTRODUCTION
With increasing applications for radio frequency (RF) electro
magnetic technology, and especially its use in cancer research and
hyperthermia (Cetas and Roemer, 1984; Oleson, 1984), the need arose
for an accurate means to measure RF magnetic fields. For certain
research applications, pulsed RF magnetic fields are also of interest.
Additionally, a standardized calibration system to establish and to
maintain the accuracy of such devices is essential. We have developed
a fiber optic coupled RF magnetic field probe system which incorporates
features that satisfy these requirements. Additionally, sensors can
be tailored to a specific frequency and dynamic range of interest.
While the specific application of concern for the field measurement
system here relates to RF magnetic induction for hyperthermic cancer
therapy, other applications in medicine, engineering, and industry
include magnetic resonance imaging (MRI), induction heating, and
surveys for safety (International Non-ionizing Radiation Committee,
1985). By following the design principles discussed here, it is
straight forward to construct probes for other frequencies and appli
cations. By sealing a sensor of reduced physical size within a
biocompatible potting compound, the probe can also be implanted for
in vivo biological experiments.
1
2
Optical RF Magnetic Field Measurement Systems
The previous RF magnetic field measurement probe system of
Oleson's is shown in Figure 1 (Oleson, 1982). The system includes a
sensor that produces an optical signal which is conveyed via optical
fibers to a photodector and amplifier. The sensor consists of a
single turn inductive loop that excites an incandescent lamp when the
loop is in the presence of an RF magnetic field. Bundled optical
fibers then are used to couple the optical information to a photodector
and amplifier. The purpose of the transconductance amplifier is to
convert the low level photodetector current to a reasonably robust
output voltage. This output voltage then is measured easily and
calibrated in terms of the magnitude of the RF magnetic field at the
sensor. The use of an incandescent filament at the sensor enables
Oleson to perform a calibration of the probe with direct current
excitation of the lamp. Fundamental concepts of electromagnetics and
circuit theory then are used to relate this DC excitation to the RF
magnetic field that produces the same lamp intensity. Oleson's
calibration procedure is described in detail in Chapter 2 under Bench
Calibration Procedures.
GE 715 AS 15 CORNING 5010 LAMP o ° ^
| FIBEROPTIC CABLE
WIRE LOOP
SENSOR
UDT Pin-040A
ZOOpF
200pF j ||Ma
Figure 1. Oleson's RF Magnetic Field Probe with Incandescent Field Lamp Sensor (Oleson, 1982).
4
Although the incandescent lamp enables a first principle
calibration of the probe, the performance characteristics of such a
probe are undesirable in certain applications. The sluggish response
(i.e., long time constant) of an incandescent lamp to burst or modu
lated RF energy prohibits the use of the incandescent filament sensor
in measuring such magnetic fields. Even if the modulation of such an
RF field fell within the response time of the incandescent probe, the
nonlinear aspects of the probe would distort the measurement of such
dynamic fields. A linear optical RF magnetic field measurement system
capable of responding to such varying fields would be beneficial.
Another feature that would enhance the operation of such a system
would be the ability of the system to maintain a calibration over a
longer duration (a heated filament tends to degrade due to evaporation
of the filament which changes both the electrical and optical proper
ties of the device).
The development of the system presented here was directed
towards producing a stable, linear and responsive optical RF magnetic
field measurement system. An overall system diagram is shown in
Figure 2. Basic operation of the system is similar to that of the
incandescent version of the probe. In order to obtain a linear and
responsive sensor, the incandescent lamp of the sensor is replaced
with an extra super bright light emitting diode (LED). The response
of the LED to an excitation current is nearly linear as is shown in
Figure 3.
PROBE WITH SENSOR
BUNDLED OPTIC FIBER
CW FIELDS
PULSE FIELDS OSCILLOSCOPE
DC VOLTMETER DETECTOR/
PHOTO AMPLIFIER
Figure 2. System Diagram of RF Magnetic Field Probe with Light Emitting Diode (LED).
I
. 006
. 0 0 5
. 0 0 - 4 Ld q: q: D (J
P QL CE Z QL O L.
P U
VI <U i. QJ Q. e
CE
. 0 0 3
0 0 2
. 00 1
0 0
I I L 4 B S
RELATIVE INTENSITY
1 0 1 3
Figure 3. Response of LED to DC Excitation Currents. CT>
7
In addition, t/ie LED is quite responsive, having a typical
rise time of one nanosecond (Stanley Technical Notes). However, the
use of the LED for improving the probe characteristics greatly com
plicates the analysis of the sensor circuitry. This will become
apparent when bench top calibration techniques are discussed in
Chapter 2. Along with making sensor modifications to enhance the
operating charateristies, modifications to the detector and amplifi
cation circuitry must be made. To achieve amplification of the
modulated or pulsed optical signal conveyed via the bundled fibers,
a transconductance amplifier must be constructed with high slew rate
operational amplifiers. This, however, adds instability to the
measurement of stable CW RF magnetic fields. To augment the stability
of the field measurement system in the continuous wave mode of opera
tion (no modulation of the RF signal being measured), a separate DC
amplifier similar to that of the incandescent probe (but chopper
stabilized) is included. Connection of the optical fibers to the
appropriate photodetector amplifier will determine the signal type (CW
or modulated) that can be measured. The now robust signal from the
transconductance amplifiers can be viewed with basic laboratory
instrumentation. As with the incandescent probe of Oleson's, the CW
probe system is monitored by a digital voltmeter. The pulse measure
ment system, however, must be monitored by an oscilloscope in order
to resolve the pulse characteristics.
In order to develop the aforementioned sensor measurement
system, a rudimentary knowledge of magnetic field measurements,
including the theoretical response of the probe, is discussed first.
From these considerations the development and testing of calibration
procedures for the probe constructed are discussed. A discussion of
the specifics and practicalities of the system and system component
construction follows, with the resulting system characteristics for
the prototype given under Results of Calibration. Finally, the
accuracy and sources responsible for inaccuracies are examined, with
future probe improvements proposed.
CHAPTER 2
FUNDAMENTALS OF RF MAGNETIC FIELD MEASUREMENT
SYSTEMS AND CALIBRATION
The direct measurement of RF magnetic fields with sensing
loops or coils is based on Faraday's law of induction. In other
systems, (Henrichsen, 1983; Nahman et al., 1985) the voltage induced
in the pickup coil is measured directly and calibrated against the
field present at the sensor. The essence of such a system is shown
in Figure 4. Measurement systems of this nature have some inherent
problems that can be remedied through the use of an optical link
between the sensor and the signal processing instrumentation. One
such difficulty alleviated is the shielding of this sensitive instru
mentation from the strong fields to be measured. Another difficulty
is the presence of induced voltages on the connecting leads from the
sensor to the instrumentation. This can produce artifacts in the
field measurement. Additionally, currents along these leads actually
can perturb the field under measure. Problems such as these are
eliminated when optical fibers are used to couple the probe to the
instrumentation. Unlike Figure 4, in an optically coupled system the
voltage induced in the pickup coil is not measured directly but must
be converted to an optical signal that can be transmitted down the
optical fiber link.
9
10
OPEN CIRCUIT SENSOR
LOADED CIRCUIT SENSOR
Non-optical RF Magnetic Field Measurement Sensors (Kanda et al., 1982)„
11
Some optical field probe systems utilize an active optical modulator
(external energy source to power the modulator) at the sensor to
generate the optical signal that is later processed (Wyss et al.,
1982; Munter, 1982). The system described here, however, employs a
completely passive sensor to monitor strong RF magnetic fields. In
this optical system the induced voltage drives a transducer (here an
LED or previously an incandescent filament) which emits an optical
signal. The relationship between the optical intensity of the signals
produced and the induced RF voltage in the sensing coil then can be
obtained. For the incandescent element this relation can be realized
from simple power considerations in the filament, whereas the complex
ity of the LED sensor demands an empirical formulation. The response
of the sensing coils to the electromagnetic fields is discussed first,
and then the sensors are considered, along with the laboratory cali
bration procedures that were developed for each system.
Bench Calibration Procedures
Response of Sensing Loops to RF Magnetic Fields
Measurements of time varying magnetic fields utilizing photo-
amplifier techniques have been performed and are described by Oleson
(Oleson, 1982). Incorporation of a light emitting diode (LED), rather
than the incandescent bulb of Oleson's version, dramatically alters
the characteristics of the sensor, as well as the calibration tech
niques. However, both sensors rely on the induced voltage developed
across a pickup coil that is subjected to the RF magnetic fields.
12
By Faraday's Law this potential is,
V =J E • dl = -iwNj'u0H . ndS (1)
(Ramo, Whinnery and Van Duzer, 1984).
The potential difference, V, is developed across the pickup coil to
produce a current that is used then to drive the incandescent lamp or
LED sensor, which is depicted ideally in Figure 5. This schematic
representation of Figure 5 is deceptively simple, because the LED
consists of a complicated gallium-aluminum-arsenic junction, which
will be discussed in detail in later chapters and in Appendices A and
B. However, the LED is favored over the incandescent version because
of its improved linearity, as was shown in Chapter 1 (Figure 1).
DC Calibration of Incandescent Lamp Sensor
The calibration procedure employed by Oleson for his earlier
version of the optical RF magnetic field probe is based on the power
dissipation in the filament of the GE 715 AS 15 incandescent lamp. The
chief underlying assumption of this technique is that the detectable
light emitted from the filament is the same whether the filament is
excited with DC or the equivalent Root Mean Square (RMS) value of RF
power. From Faraday's law, and knowing the impedance (Z) of the loop,
we can compute the RF current phasor, I(u»)= Z-l^E" • dT
I(u>) = -Z~^ia)UQ^~HzdS = -Z-^iajp0 <HZ> irr^ (2)
By computing the RMS power and equating to the DC power dissipated in
the loop we have:
l2(o)R(o) = I(o))I*(tu)R(o))/2
= [u0u <HZ> Trr2]2R(a))/ZZ* (3)
13
I
AAA/-
H f t R 2 ; ' L E D
Figure 5. Ideal Representation of the LED Sensor.
14
Rearranging terms, the calibration equation becomes,
|Z|I(o) 2R(o) <Hz> = (4)
R(w)
where, <Hz>= peak magnetic fields z component averaged over sensor loop
Z = loop and filament impedance Mag|Z|/ij>
r = loop radius
u0= 4 ttXIO-7
R(u)= RF loop resistance R(u>) = |Z| cos <|> , R(o) = DC value
I(a>) = peak RF current in filament
With these equations, calibration of the probe becomes simply
a matter of measuring the various parameters required and correlating
them to the output of the detector-amplifier. Measurement of these
values however is not a trivial task. The set up is shown in Figure
6. A 2.5 millihenry RF choke was necessary to provide the DC volt
meters with a high RF input impedance, permitting measurement of the
loop/filament impedance under biased conditions. The value of these
inductors is chosen such that ojL>>Z(o)), thereby minimizing errors in
the measurement of Z(w). (Because Z(w) is also a function of the bias
current, I, Z(<o,I) is a more correct description of this parameter;
however Z(w) will be used, with this implicitly understood and with
u)L>>Z(w)max). In addition, a measurement is taken with the vector
impedance bridge leads shorted at the loop to correct for the lead
impedance between the bridge and the sensor. Figure 7 is a compilation
of calibration data at 13.56 MHz, with the values in Root Mean Square
(RMS) Hz (the z component of the magnetic field) computed from the
calibration equation of Oleson (1982).
D C
POWER
SUPPLY
-<3>
2.5mH
2.5 mH •S f Filament
HP48I5A
Figure 6. Calibration Set-up for the Incandescent Probe System.
2 5 0
a 1 _J
u t—i L.
U I-1 1— u z (J e a: \ T. CE
w
01 a. H u QL i-
Ld Z
u \ a: CE UJ o QL a U cn Q.
r z CE CE U s:
i-o o QL
2 0 0 —
0 —
1 0 0 —
0
0
PHOTODECTECTOR-RMPLIFIER OUTPUT (Volts)
Figure 7. Calibration Curve at 13.56 MHz for the Incandescent Probe System.
17
Attempt at DC Calibration of LED Sensor
Unlike Oleson's probe, the LED versions response cannot be
correlated directly to the power dissipation in the LED junction
because of the presence of other power consuming elements in the
circuit (i.e., the current limiting resistance and temperature compen
sating network). However, the ease of generating and accurately
measuring direct currents in the laboratory makes this calibrating
scheme particularly appealing. The following was an attempt to
formulate a DC calibration procedure which relied on the following
assumptions: 1) the response time of the LED and associated circuitry
is fast enough to follow the RF sinusoidal excitation, 2) stray
capacitances and couplings minimally effect the model, 3) the impedance
of the sensor is sufficiently resisitive to permit the approximation
of the impedance as a pure resistance, and 4) the detector responds to
the average illumination of the LED. In an effort to calibrate with
DC, a value of current must be formulated that produces the same
illumination as the RF current during field measurements.
From the stated assumptions, it was determined that DC excita
tion of the probe would produce the same intensity of light in the LED
as the average value of forward RF current through the LED 's junction.
The procedure then was to formulate an equation for averaging this
forward current through the LED junction. If the reactive effects in
the sensor can be neglected (assumption 3, and possibly the most
18
deleterious of all the assumptions), then the forward current flowing
through the LED can be represented from Kirchoff's laws as,
Vosin(u)t)-Vd id (5)
R
Vo = peak value of induced voltage
Vd = forward voltage of diode
R = resistive approximation of loop impedance
Averaging of this equation then is performed by integating the expres
sion from wt = 0 to 2 I T with successive division by the period (2 I R ) .
The integration of this equation was performed and tested with a
slowly varying DC signal, and was found to model this situation rather
accurately. Appendix A gives the forward diode characteristics
represented in equation 5 as Vd, and Appendix B utilizes this infor
mation to provide the complete calibration algorithm, along with the
aforementioned testing. Figure 8 shows the resulting calibration of
the LED field probe system with this technique. This DC calibration
then was used to measure the axial RF magnetic field in a Magnetrode
hyperthermia applicator, and was compared to measurements made with
Oleson's probe system. The LED system calibrated with DC gave values
that differed vastly from Oleson's probe system. This is shown in
Table 1, where a comparison between the probe systems was made in a
Magnetrode™ hyperthermia applicator. The discrepancy between Oleson's
incandescent version and the LED is as high as 69%, totally unaccept
able. This discrepency between probe systems places much suspicion
on the DC calibration of the LED version.
3 0
B 0
CE 7" 0
a _J u B 0 M Li.
U 5 0 M
h-
U Z
m 3 0
a:
2 0
l 0
PHOTODECTECTOR-flMPLIFIER OUTPUT (Volts)
Figure 8. Calibration of the LED Probe at 13.56 MHz Using a DC Calibration Technique.
20
Table 1. Comparison of the DC Calibrated LED Probe System and Incandescent Probe System
LED Probe
(RMS A/m)
Incandescent Probe
(RMS A/m)
Applied Power
(watts)
19.9 65.5 20
24.6 73.8 60
30.0 82.5 80
34.9 89.8 100
39.2 96.6 120
43.2 102.6 140
48.0 109.7 160
51.3 115.4 180
55.0 120.1 200
To compare the DC calibrated LED probe and an incandescent probe system, a test was made at 13.56 MHz at a single point within a Magnetrode™ thigh applicator.
The sources for error in the DC calibration procedure are numerous;
however, assumptions (3) and (4) are the most suspect. It is possible
that reactive RF currents are travelling through the capacitance of
the LED's junction and not contributing to the luminence of the LED.
Such an effect would cause the DC calibration to give field values
somewhat below the actual field values. Appendix B suggests an
approach to a DC calibration procedure in which the reactive components
of the probe would be included in the formulation of a differential
equation. As for assumption (4), the exact response of the detector
and amplifier circuitry to the RF modulated light signal is also an
unknown. This response depends on many factors within the amplifier
system, such as slew rate of the amplifiers, decay time and inherent
capacitance of the photodetector. An accurate characterization of the
amplifiers and detectors in combination, over all frequencies of
interest, would be required. A more appropriate solution to this
problem, as well as other improvements to the system are included in
the conclusions. Because of the complexity of the LED junction and
the inability to model its RF behavior accurately, the idea of a DC
calibration procedure for the LED sensor was abandoned. The next
approach was to calibrate the output of the photodetector/amplifier
by applying an RF excitation to the sensor.
Calibration with RF Current
By injection of RF current into the sensing loop, all of the
frequency dependent terms and possible hidden phenomena of the probe
22
system are taken into account and become transparent in the calibra
tion. The procedure is to break the connection at the sensing loop
and apply a known RF voltage to this point. Referring to a circuit
theory analysis of the sensor, the sensing loop can be modeled as an
ideal voltage source, with its inductance represented as a series
impedance (the Thevinen representation). The magnitude of the ideal
voltage source then becomes the open circuit value of the sensor's
induced voltage in the-presence of the magnetic field. To calibrate
the probe, an RF current is injected, with the voltage measured at
the point of injection. The output of the photodetector/amplifier
can be related to the applied RF voltage and, from equation 1, to
the incident H-field this voltage simulates. Accuracy of the calibra
tions is dependent only on the ability to predict induced voltage on
the loop and the ability to measure the applied RF voltage. The
calibration arrangement is shown in Figure 9 and was tested first on
the incandescent version of Oleson's. The RF source is, of course,
not an ideal voltage source, having an RF (as well as DC) impedance
of 50 ohms. This should present no problems, as far as Oleson's probe
is concerned, because the voltage impressed on the sensor is measured
at the point of injection on the coil and not at the generator. A
comparison between this calibration procedure and the DC calibration
of Oleson's was made at 13.56MHz. Table 2 shows this comparison of
calibration procedures. Excellent agreement is obtained between the
two procedures and their predicted field values (within 2 percent).
INCANDESCENT PROBE
RF GENERATOR© 50 SI
LO AD : © RF VOLTMETER OR OSCILLOSCOPE
OPTIC FIBERS
TPO OPTIC FIBERS
TP2
LED PROBE
Figure 9. RF Calibration Setup for Calibrating Both LED and Incandescent Versions of RF Magnetic Field Probe Systems. no
CO
24
Table 2. Comparison of the RF and DC Calibrations on Oleson's Incandescent Probe System
Photodetector-Amplifier RF Calibration DC Calibration Output (Volts) (RMS A/m) (RMS A/m)
.010 36.3 37.2
.045 49.5 49.9
.157 66.0 67.4
.350 82.6 84.1
.639 99.1 100.0
1.061 115.6 116.4
1.646 132.1 134.1
2.216 148.6 148.3
2.999 165.1 164.2
3.952 181.6 180.7
5.053 198.1 198.2
25
Calibration of the new generation probe was achieved in a
simi1ar manner. In addition, the possibility of bias currents
resulting from the nonlinear nature of both the LED and the protection
diode was considered. These bias currents could result in undesired
excitation of the LED that could upset the calibration. If the
currents generated during the calibration are not the same as those
during an actual measurement, the calibration is invalid. To clarify
the situation, a calibration was performed with calibration RF source
impedances of one to fifty ohms. If bias currents are present their
magnitudes would differ with the different source impedances and the
calibrations would not correlate to one another. There was no
discernable difference observed between calibrations with different
source impedances. Even if bias currents should occur, their effects
on the calibration would be small, because the calibration source
impedance is small compared to the loop/sensor impedance (i.e., 50
ohms compared to 1500 ohms for the sensor). RF calibration of the
LED version of the probe for 13.56 MHz and 14.25 MHz are shown in
Figures 10 and 11, respectively.
e \ CE
Q _1 U n u.
u
u z (J (r 2:
cn 2: a
13 0 1—
1 2 0 —
1 1 0
1 00 —
3 0 —
B 0 —
7" 0
S 0 —
5 0 —
4 0
3 0
PHOTODECTECTOR-RMPLIFIEF? OUTPUT (Volts)
Figure 10. 13.56 MHz RF Calibration of the LED Probe System.
e \ cl
a _i Ld M Li_
u l-t h-U z (J a; T.
01 z
1 4 0
1 2 0
1 0 0 —
B 0 —
e 0 —
4 0
2 0
PHOTODECTECTOR-RMPLIFIER OUTPUT CVolts)
Figure 11. 14.25 MHz RF Calibration of the LED Probe System. ro
CHAPTER 3
PRACTICAL CONSIDERATIONS
Integrating an LED into an RF magnetic field probe system
produces numerous engineering complications. The system constructed
must be stable and reliable, as well as easily operated and modified
for varying applications. The system developed has these features,
but it is only a prototype, with future developments suggested in the
conclusions. The development of such a prototype must encompass the
practical considerations, from sensor element to final signal process
ing instrumentation, without compromising the integrity of the complete
measurement system. Sensor stablility is of primary importance, and
a description of stabilization techniques and details of sensor
construction are given. Lastly, the optical coupling and processing
instrumentation are described.
Probe Sensor Construction
LED Sensor Protection
In order to provide an accurate and stable probe system for
monitoring magnetic fields, it is necessary to understand the sensing
element, together with all of its anomalies. One such anomaly was the
unreliability of early versions of the LED sensors. Reliability is a
desirable characteristic in any measurement system. In earlier
28
29
versions of the probe system, the LED sensors were unable to with
stand the measured fields for any length of time; additionally, field
measurements with these earlier systems were seldom repeatable.
Apparently, the LED would undergo some sort of premature degradation
that either severely reduced the optical efficiency or, in some
instances, resulted in catastrophic failure of the device. Increasing
the series resistance to limit further the current through the LED
produced only moderate success. Furthermore, the forward currents
through the LED were well under the maximum junction rating of 100
mi 11iamperes, as published by the manufacturer (Stanley). The possi
bility of damaging reverse bias currents through the GaAlAs junction
was suspected next. After sacrificing similar LEDs, it was found that
the avalanche voltage of different LEDs varied considerably (from 7
to about 25 volts, Williams and Hall, 1978), and that reverse bias
operation led to performance degradation, with currents in excess of
about 20 milliamperes resulting, eventually, in catastrophic failure.
To alleviate the problem, a protection diode was mounted with reverse
polarity across the LED. This limits the reverse bias voltages to a
value of about 0.7 volts, which is well below the reverse breakdown on
the LED.
LED Temperature Sensitivity/Compensation
Stable operation of the probe in varing environments is another
prime factor for reliable operation; particularly, the ability of the
probe to perform well over a reasonable temperature range. Due to the
30
complex nature of this sensitivity to temperature (e.g., variations in
injection efficiency, band gap dependence, and radiative and non-
radiative mechanisms (Williams and Hall, 1978; Henisch, 1984), an
empirical determination of the thermal behavior of the LED is required.
As the LED's intensity provides the measurement signal, temperature
drifts of this intensity during the measurement of a stable RF magnetic
field should give an indication of the LED's sensitivity to tempera
ture. Because providing a stable RF magnetic field for performing
these thermal studies seemed out of the question (providing an RF
magnetic field internal to the copper thermal block, that was used to
control the probe's temperature, would be next to impossible), an
alternate means for exciting the LED was sought. Assuming that the
temperature variations produce only negligible changes in the AC
impedance of the sensing loop, the induced voltage of the sensing
coil could be replaced by a fixed DC voltage. With the aid of an
HP9836 based data aquisition system (HP DAS) and a component oven,
temperature sensitivities were determined. Figure 12 illustrates the
setup involved, with Figure 13 showing a graph of the output as a
function of voltage, (different voltages correspond to different
relative magnetic field strengths), with varing oven temperatures of
54, 38, 28 and 24°C. Note that a 1000 ohm series resistance was used
as a nominal resistance value for these studies. Ideally, for a zero
temperature coefficient, these curves should coincide. This is the
goal sought via temperature compensation.
Temperature compensation in semiconductor junctions can be
achieved by the use of linearizing thermistor networks (Jaffe, 1984).
THERMOCOUPLE
VOLTAGE D/A
INSULATING CAPS 3-LEAD
TEST CABLE A/D
THERMISTOR PROBE
CW OUTPUT OVEN
OPTIC FIBER
HP9836
INTERFACE
JUNCTION
BLOCK
HEATER
CONTROLLER
OPTICAL
DETECTOR /AMPLI FLIER
Figure 12. Experimental Setup for Determining Temperature Sensitivity of an LED.
The junction block is described in Appendix A.
1 2
1 13
Z3 0. H D O oc u
Q. r a: i
CK o H U u H U a o i-o X a.
o
— 2 I _L _L
2 4 . 8 2 8 . B
, 3 4 . 5 4 1 . i
...51 .5
TEMPERATURE
C C e 1 s i u s )
_L 2 3 4 5 S
RPPLIED DC EXCITRTION VOLTRGES CVolts)
Figure 13. Graph Showing Temperature Sensitivity of the LED. Output Versus Applied Loop Voltage.
Different voltages correspond to different relative magnetic field strengths. Each dot indicates a single data point taken. PO
33
In order to properly compensate the semiconductor junction, the
temperature coefficients of the junction need be determined. This
can be accomplished by refering to Figure 13. We choose a fixed loop
voltage and graphically determine the percent change in the output:
J_^Vo V0 At ( 6 )
Va = Vi, V2, V3...
V0 = the average output between temperatures, T1 and T2
AV0 = output voltage difference
AT = temperature interval T1-T2
Va = applied loop voltage parameter
By taking various combinations of applied voltages and temperatures,
an average temperature coefficient over the region of interest was
obtained. Table 3 shows the result of 15 such points, with an average
value of -1.05 %/°C taken as that necessary for compensation over a
temperature range of 24 to 51°C.
Temperature compensation was accomplished with negative
temperature coefficient thermistors that were properly linearized.
Initially, the probes were compensated by a trial and error technique
that involved adjusting the value of precision resistors in parallel
and series with the thermistor until compensation was achieved. The
prototype probe constructed was compensated by this technique. This
proved to be a tedious and time consuming task, which consequently
motivated a more mathematical approach. The method incorporated is
that of Jaffe (1984), where thermistors are linearized to compensate
the more linear nature of semiconductor drifts. The linearization
circuit is shown schematically in Figure 14.
3
4
5
6
7
3
4
5
6
7
3
4
5
6
7
Calculation of Average Temperature Coefficient From Figure 13.
°C °C Volts T2 Tj Average %/°C
51.45 41.10 1.75 1.32
51.45 41.10 3.35 1.24
51.45 41.10 4.85 1.33
51.45 41.10 6.55 1.34
51.45 41.10 8.22 1.34
41.10 28.60 1.90 0.8
41.10 28.60 3.70 0.74
41.10 28.60 5.47 0.84
41.10 28.60 7.45 0.87
41.10 28.60 9.40 0.85
34.50 24.75 1.95 1.00
34.50 24.75 3.80 1.03
34.50 24.75 5.57 1.05
34.50 24.75 7.70 1.02
34.50 24.75 9.65 1.01
THERMISTOR
J
Figure 14. Linearization Circuit Used to Compensate LED Sensors.
36
From this circuit, Jaffe uses a matched slope criteria to arrive at a
value he denotes as S, given by the following expression:
Tz'/fTTD - Tyf tm S = , - =— (7)
Trr(TiWr(T2) - T2-r(T2)-jKh)
From this parameter, relations for the resistor in parallel (Rp) with
the thermistor, the 25°C resistance of the thermistor, R(To) and the
value of the series resistor (Rs) may be obtained from Jaffe. Expres
sions for these elements are as follows: A.(T2-T1)
Rp = {S2 r(Ti)r(To) + SOCM + r(T2)] + 1> S • (r(T2)-r(Ti))
R(To) = S Rp (thermistor valve to select at 25°C) (8)
R(T0)Rp Rs = Rd - Rd = desired total resistance
R(T0)+Rp ar
where A needed for proper compensation. at
In order to apply this linearization scheme to the LED and its
associated circuitry, the temperature coefficient of the circuit must
be reformulated in terms of a change in the circuit resistance with
respect to temperature. From the graph in Figure 13, the variations
in the optical intensity of the LED were determined and then were
expressed in terms of a relative percent variation of the intensity.
By taking advantage of the linear relationship between forward LED
current and LED intensity, the temperature effects in the luminescence
of the junction then could be compensated directly by proportional
variations of the drive current. Equation 5 in Chapter 2 is a suitable
approximation for the LED forward current, and can be used to represent
37
the diode current for temperature compensation. It now becomes a
simple task to compute the temperature derivative of this equation, and
thereby to determine the relationship between current sensitivity and
resistance sensitivity (Gray and Meyer, 1977, p. 247): let
Va=V0 sin(a>t)-Vd, then id=Va/R from eq 5,
1 did 1 dVa 1 dR 1 dVa — - «— «— where,— 0 (9)
id dT Va dT R dT Va dT
Note: dVa/dT = K (from the temperature derivative of the model equation in Appendix A, K = -.0019 V/°C) and for operation in the linear region id(minimum) = 1mA (see Chapter 4 under discussion of probe linearity). Thus, for a probe with a 2000 ohm resistance element (typical value), Va(min) = id(min) X 2000 = 2 volts. It follows that for the worst case (1/Va) dVa/dT = K/Va(min) = -.0019/2 = -.095%/°C, which is considerably less than the 1.05%/°C ((l/R)dR/dT) used for compensation and thus the approximation that (l/Va)dVa/dT can be neglected is valid for compensation over the linear range of the probe.
As shown in the foregoing equations, the parts per million (ppm) or
percentile variation in the resistance with temperature is exactly
that of the current in the LED, which in turn is the same as that of
the luminesence! If for example, the circuit resistance R is 2
kohms, then for a (1/1)dl/dT of -1.05% and a (l/i)di/dT of +1.05% (a
plus sign is indicated because we are trying to cancel the intensity
effects) we have a (l/R)dR/dT of -1.05% or -2.10 ohms/°C. From here
the temperature compensation technique can be applied directly to
compensate the junction for linear temperature drifts.
Jaffe's linearization techniques and equations were programmed
into an HP 41CX calculator, for which a program listing is given in
Appendix C. Input data for the thermistors was acquired from Fenwal
thermistor/temperature conversion charts (Fenwal, 1974). From this
38
data and numerous trial runs of the program, it was found that thermis
tors with temperature coefficients of at least -3.9%/°C were required
if the percent change is above about 1.3%/°C for the circuit to be com
pensated. For example, values of -3.1%/°C and -3.4%/°C gave negative
resistance values for the series element of Figure 14 (if the
(1/I)dl/dt > 1.3%/°C), with the implication that compensation with
these thermistor values is unrealistic. The changes of the thermistor
resistance with respect to temperature were insufficient to allow
linearization of the thermistor and simultaneously permit compensation
of the junction. An example of such a situation follows: A -1.3%/°C
LED with a desired series resistance element of 1000 ohmsat 25°C is
to be compensated with a -3.4%/°C thermistor. From Fenwal thermistor
data:
r(10°)=1.70
r(50°)=.454
Linearization yields:
Rp=1698.0 ohms
R(25°)( thermistor resistance at 25°C)=2942.2 ohms
Rp//R(25°)=1076.7
Rs=1000.0-1076.7=-76.7 ohms
which is, of course, impossible. Table 4 gives values required to
properly compensate an LED with a temperature coefficient of (l/I)dI/dT
of -.0105/°C, using a -3.9%/°C thermistor. The values correspond to
the components of the schematic in Figure 14. A sample run of the
program is also included in Appendix C.
39
Table 4. Component Values for Compensation of a -.0105/°C LED With -3.9%/°C Thermistors.
Sensor Loop R
DR/DT =R x-.0105/°C
R (25°C) Thermistor R Parallel Rs
100 - 1.05 200.8 119.7 25.0
200 - 2.10 401.5 239.5 50.0
400 - 4.20 803.1 478.9 100.0
500 - 5.25 1003.8 598.6 125.0
800 - 8.40 1606.1 957.8 200.0
1000 -10.50 2007.6 1197.3 250.0
1200 -12.60 2409.2 1436.7 300.0
1500 -15.75 3011.4 1795.9 375.0
1700 -17.85 3413.0 2035.4 425.0
1800 -18.90 3613.7 2155.1 450.0
2000 -21.00 4015.3 2394.6 500.0
2200 -23.10 4416.8 2634.0 550.0
2500 -26.25 5019.1 2993.2 625.0
3000 -31.50 6022.9 3591.9 750.0
3500 -36.75 7026.7 4190.5 875.0
The 25°C loop resistance (R) is chosen first and the other values are computed via Jaffe's compensation technique.
40
Details of Probe Sensor Construction
Initial sensor designs were patterned after Oleson and
consisted of a single turn loop, with the LED substituting for the
incandescent lamp. The use of a single turn loop was not practical
when an LED was employed. The threshold voltage of the LED required
that the single-turn loops have large physical dimensions, thereby
limiting their usefulness and increasing their suseptibility to
strong electric fields. Figure 15 is a schematic of one of these
earlier loop designs, while Figure 16 is a photograph of these proto
types. The protection diode is as described in Chapter 2, with the
resistor R in the loop to provide the current limiting, as well as a
variable to alter the dynamic range and sensitivity of the probe.
These early versions of the probe did not incorporate any of the
temperature compensations that were later found to be neccesary. The
final version of the probe is shown schematically in Figures 17a and
17b, with a detailed diagram of component placement in Figure 18.
Figure 17a shows the probe with the empirical temperature compensation,
while Figure 17b implements Jaffe's compensation technique. Tempera
ture compensation of the LED was employed by potting the thermistor
against the semiconductor junction. The multi-turn loop design
performed satisfactorily, while enabling the construction of a
physically small probe.
J
41
PROTECTION DIODE
Figure 15. Early Sensor Schematic.
Figure 16. Photograph of Early Sensors.
43
SENSOR CKT DIAGRAM
3038 av>LED Wi—i ESBR 5701
-vw 1951
to)
-±\LED dfc/ESBR 5701 IN9I4 I THERMISTOR
AAAr-y
Figure 17. Schematic of the Sensor with Empirically Arranged Temperature Compensation Components (17a). Component Placement for Mathematically Obtained Compensation (17b).
44
•12 cm-
E o N-cvj
IN9I4
PICKUP COIL
BULKHEAD
CONNECTOR
WITH SENSOR
-.7 cm-
Figure 18. Details of Component Placement for the LED Sensor.
45
The entire field sensor was packaged in an Amphenol optical
bulkhead receptacle that was turned round on a miniature jeweler's
lathe. The LED was first drilled from the lead side with a number
60 drill until the bit contacted the metallic base of the junction.
This procedure prevents thermal gradients from degrading the thermal
stability of the temperature compensated probe. The ridge present
on the Stanley ESBR 5701 LED is filed flush with the surface to allow
the LED to fit snugly into the plastic Amphenol 530564-1 bulkhead
receptacle. Additionally, the LED's emitting lens is filed and then
polished flat up to the catwisker contact of the LED's anode. Care
must be taken not to sever this connection, or the LED will become
inoperative. The LED then is pushed firmly into the bulkhead recep
tacle until it contacts the receptacle's internal stop. The coil
is wound on a small teflon coil form of the appropriate diameter
(depending on what the desired probe dimensions and sensitivity) and
placed over the leads of the LED. The thermistor and associated
compensation resitors are then inserted (the thermistor is placed in
the predrilled hole) and wired as per Figures 17 and 18. Note that
certain leads are brought out to provide for calibration test points.
The entire probe is then potted in a mixture of opaque epoxy.
Optical Link and Photoamplifiers
Galite 2000P bundled optic fiber is used to convey the light
energy to the photodetector. The Amphenol connectors are assembled
to the fiber as per Amphenol's instructions and subsequently are
46
polished accordingly (Amphenol, 1982). The fiber then is attached to
the sensor with the application of a small amount of optical coupling
compound on the tip of the fiber. The photo-detectors were mounted
in metallic Amphenol 905 117 5000 bulkhead receptacles. The use of
the metal receptacles at this point aids in the isolation of the
photo-detectors and amplifiers from RF interference.
Two separate amplifiers were employed, with separate photo-
detectors to drive each of them. A DC chopper stabilized amplifier
is used for CW fields, and a high slew rate pulse amplifier is used
for measuring modulated or pulsed fields.
The CW amplifier is shown schematically in Figure 19. The
first stage amplifier is implemented in a transconductance mode with
the output voltage proportional to the input current (Swindell,
1978). More explicitly, the following relationship holds,
Vout = -i-Rf. (10)
where i is the PIN diode current. The photodetector is used in an
unbiased mode, so there is a threshold value of light before operation
in the linear region begins. The second stage is the standard invert
ing amplifier configuration, in which the gain is computed as,
Vout = (-Rf/Ri)Vin (11)
This stage serves as a buffer between the transconductance amplifier
and the output, as well as being an adjustable gain stage for sensi
tivity control. The chopper components are the same as those recom
mended by the manufacture and are included with the manufacturer's
specifications (Datel-Intersi1, 1982).
£0.0015 0.0015
AM-^
490-2 PIN 020 OUT
-V V
CW AMPLIFIER
Figure 19. Schematic of CW Amplifier with Chopper Stabilized Amplifiers.
48
A circuit board was constructed for the amplifier. The pulsed ampli
fier was constructed with the same basic techniques as the chopper
circuitry, Figure 20. A transconductance first stage is followed by
a secondary gain-buffer stage. The amplifiers used were Burr Brown
3554 operational amplifiers. Because of the fast rise times and high
gain of the circuitry, the amplifiers proved to be inherently unstable.
Care needs to be taken when the layout for the printed circuit board
is constructed. Ground loop currents in the circuit pattern can cause
unwanted feedback that can generate spurious oscillations of the
amplifiers. In addition, proper compensation of the amplifiers is
required to maximize the response characteristics, while minimizing
the instability of the circuit (Burr Brown Technical Notes, 1984).
Three circuit foil patterns were tried until a successful, stable
design was achieved. Figure 21 shows the final circuit pattern, with
the components as viewed from above.
' Both amplifier boards were housed in a 3 X 7 X 9 inch metal
circuit box, which is not radiofrequency interference( RFI) proof.
An electromagnetically shielded box is recommended if the amplifiers
themselves are to be subjected to the fields. Field measurements
made with the described prototype were made with the electronics
isolated from the measured fields. Nevertheless, with the optical
fiber link between the sensor and the electronics, RFI proofing of
the circuit becomes a simple task.
4.7 Meg SI
3554
+V
PULSE AMPLIFIER
Figure 20. Schematic of the Pulse Amplifier.
50K&
3554
+V
+V
v©
50
Figure 21. Printed Circuit Foil Pattern for the Pulse Amplifier.
51
Output from the pulsed amplifier was brought to the front
panel BNC connector for attachment of a laboratory oscilliscope. The
CW amplifier has a Datel 3-1/2 digit multimeter on the front panel,
with an additional BNC connector on the back panel for connection to
a data aquisition system or external voltmeter. In Chapter 5 there
will be further discussion on the modification of the amplifier
circuitry-modifications that can be used to enhance the operation of
the LED probe system.
CHAPTER 4
CALIBRATION RESULTS AND SYSTEM CALIBRATION
After the LED RF magnetic field probe system was calibrated
with RF current, system calibration checks were performed and system
characteristics were determined. In the first section of this Chapter,
the RF calibration of the LED field probe is verified by comparisons
to known fields and against a reliably calibrated field probe. The
final section of the Chapter is devoted to a discussion of the overall
system characteristics. Linearity, sensitivity, thermal stability,
dynamic range, frequency and pulse response, and optical connection
repeatability are all discussed.
Calibration Verification
Comparisons with both standard solenoidal fields and the
calibrated incandescent probe system were performed. Initially, a
solenoid was constructed to generate a known stable magnetic field
that could be measured with both the LED probe system and the incan
descent probe system. Comparisons were made between the calculated
fields at the center of the solenoid and those measured by both probe
systems. To further enhance the credibility of the LED probe system
with RF calibration, the comparison between the incandescent probe
and the LED probe system within the Magnetrode™ was repeated. (It
was this comparison that led to the demise of the DC calibration
52
53
technique in Chapter 2.) Lastly, a single turn strap inductor is
excited at 14.25 MHz. The fields internal to the strap are calculated
and compared to those measured with the LED probe system.
Solenoidal Field Generation and Measurements
In an effort to check the probe calibrations, standard RF
Magnetic fields were generated and measured with both probes. The
measurements were compared between probes and to the computed field
values. The technique to generate a standard field discussed here
involved a series tuned LCR circuit with a solenoid of known dimen
sions. The solenoid was wound with number 10 gauge wire and placed
in series with an air variable capacitor and a 50 ohm RF termination,
as shown in Figure 22. By measuring the voltage across the termina
tion, the magnitude of the phasor current is determined. Because we
have a series tuned circuit, this is also the phasor current in the
solenoid. For finite solenoids, the expression, H=i*n where i is the
current, and n the turns per unit length, can be used to approximate
the fields at the center of the coil (Halladay and Resnick, 1978). In
addition, this equation assumes a uniform current distribution across
the windings of the solenoid. To assure this uniform distribution,
the length of the wire used in the solenoid must be electrically
short. The physical properties of the solenoid are given in Figure 23.
Table 5 compares the computed fields to those measured by Oleson's and
the LED version of the probe system.
RESONATING CAPACITOR
© ~ ) R F G E N E R A T O R
SOLENOID
50 & LOAD RESISTOR
X
Figure 22. Schematic of Solenoid and Resonating Components for Standard Field Generation.
ooooooooooooooooooooooooo h 13,3 cm H
Figure 23. Physical Characteristics of the Standard Solenoid.
25 turns number 10 guage household electrical wire, 187.83 turns/meter.
56
Table 5. Comparison of Calculated Solenoid Fields to Those Measured by the LED System and Oleson's Incandescent System
LED Probe Oleson's System RMS A/m Calculated RMS A/m RMS A/m
46.0 47.5 54.4
57.5 56.3 66.8
69.0 67.4 77.9
80.6 79.5 88.5
92.1 92.4 99.3
103.6 104.0 109.9
115.1 118.1 121.7
126.6 132.2 134.2
138.1 146.2 147.2
57
Agreement between the solenodial calculations and the LED probe system
measurements is within 6% over the range of H-fields generated by this
system. The incandescent probe system, on the other hand, seems to
deviate from the LED system at the low end (18.7% at 56.3 A/m on the
LED system) with better agreement (within 1%) at the higher end of the
generated fields. This is probably due to the fact that the sensitivity
of the incandescent probe system is greatly reduced at these lower
field values. Due to the nonlinearities in the incandescent probe
system, the slope of the calibration curve (H vs voltage) for values
of H-field below about 100 Amperes/meter(A/m) is nearly vertical.
Therefore, large changes in the magnetic field only produce minor
changes in the output voltage of the system, i.e., low sensitivity.
This can be seen more easily by looking at the calibration curve for
the incandescent probe system (Figure 7, Chapter 2).
Comparison Between Probe Systems in a Magnetrode™ Applicator
After the calibration was verified in a standard field, a
comparison between probe systems in a clinical hyperthermia applicator
was performed. Measurements of the RF magnetic fields at 13.56 MHz
in a Henry Radio Magnetrode™ Thigh applicator for hyperthermia were
conducted. A comparison was first made between the incandescent probe
system and the LED version at one field location and then relative
fields across the center of the coil were mapped.
The applicator was first loaded with a one liter beaker of
saline solution and then excited with the Magnetrode™ generator. A
field point was chosen external to the saline load but still within
58
the applicator itself. The applicator was excited with RF powers
varying from 100 to about 200 watts (the lower limit of 100 watts was
chosen to obtain the higher accuracy with the incandescent system),
with the field point measured with both probes at 20 watt power incre
ments. Table 6 compares the values measured with both probe systems
at each of these applied powers. Agreement between probes was well
within 10% for this set of measurements with agreement to 2% again at
the higher field values (accurate and repeatability of probe placement
during this test could degrade the accuracy of these data). Once the
credibility of the LED probe system had been established, the fields
across the thigh applicator were mapped at the z equal to zero plane
(centrally between the ends of the cylindrical applicator). To map
these fields, the entire applicator was lined with a plastic membrane
and filled to the rim with saline solution. The LED probe then was
encapsulated in a small plastic vial and attached to a 60 centimeter
fiberglass rod. This rod then could be manipulated with a stepper motor
and a computer to record the magnitude of the field at each point.
Figure 24 shows the relative field strengths at the z equal to zero
plane within a 15 X 15 centimeter area mapped within the applicator.
Magnetic Field Measurements in a Current Strap
This final method used for generating a standard magnetic field
involves the excitation of a single turn strap inductance, for which
numerical integration has generated data for the internal fields. This
coil was resonated with a vacuum capacitor in a parallel configuration
and then matched to the exciter with a transmission line system.
59
Table 6. Comparison Between Oleson's Incandescent Probe System and the LED Probe System Within a Saline Loaded Magnetrode™ Hyperthermia Applicator.
LED System Incandescent System Applied Power (RMS A/m) (RMS A/m) (Watts)
96.1 105.6 100
107.4 115.3 120
114.9 121.7 140
125.1 130.3 160
133.9 137.9 180
143.0 145.8 200
RELRTIVE ( n o r m a l i
FIELD zed to
STRENGTH u n i t y )
1 UNIT OF DISTANCE = .75 cm
Figure 24. Relative Field Strengths Internal to a Saline Loaded Magnetrode™ Hyperthermia ^ Thigh Coil Applicator. o
61
The transmission line was constructed to match the 5400 ohm balanced
impedance to the 50 ohm output of the Heath SB-220 linear amplifier. A
half wavelength unbalanced to balanced coaxial balun was used to bring
the 50 ohm exciter impedance to a balanced 200 ohms. From here, a 450
transmission line consisting of 1/4 inch copper tubing was constructed
to match to the 5400 ohm parallel tuned impedance of the coil. The
Smith Chart of Figure 25 shows the line length and shorting stub calcu
lations. Figure 26 and the photo in Figure 27 show the measurement
apparatus. Knowing the dimensions of the coil and the current distribu
tion across the strap a computaion of the fields internal to the strap
was obtained. The Biot-Savart expression for the fields is as follows:
^oToa r + b i r z * . er(z-z') - k(r cos(<j>-«j,')-a) B = ———- / dz J d<f> . ; •— (12)
4ir2b -b o 2 (r2+a-2ra cos()+(z-z )2)
with the current distribution described by Butler (1985) as:
/ J o J 0 dx (13)
• o J r 7 W
Integration of this expression for the current distribution over the
width of the strap gives the magnitude of the phasor current required
to generate a specific value of magnetic field. A computer program
was written by Y. Li in which the expression for the internal fields
was evaluted for a I0 of unity. Integration of the current distribu
tion over the width of the strap yields,
62
MATCHING SECTION
mmimim on' coJouc'tamcc
SHORTING STUB!
Figure 25. Smith Chart to Calculate 450 ohm Transmission Line Length and Shorting Stub Length.
50:20012 BALUN 450SI TRANSMISSION LINE
MATCHING SECTION 4.74M
CURRENT STRAP
5 4 0 0 L O A D
INPUT
I SHORTED STRAP 0.55 M
Figure 26. Diagram of Current Strap Excitation Apparatus.
Figure 27. Photograph of Standard Current Strap and Resonating Capacitor.
65
where bn becomes a proportionality constant between the measured phasor
current and the field values (I0 was normalized for the field calcula
tions). Figure 28 diagrams a set of field points that were measured
and are presented in Table 7. The uniformity of the field calculations
agree with the uniformity of the measured field; however, the absolute
magnitude of the measured fields is considerably lower than that cal
culated. This is thought to be due to radiative emissions and losses
in the transmission line reducing the actual power reaching the strap
thereby causing an error in the determination of the strap's current
(Io). Nevertheless, this strap was excited by pulsed magnetic fields
to characterize the response of the field probe system in this domain
as well. Results of this test will be presented under System Charac
teristics, Pulse Response.
System Charateristies
The optical field probe system that was constructed exhibited
both desirable and undesirable characteristics. The desirable charac
teristics include linearity, adjustable dynamic range and sensitivity,
high probe impedance, and good pulse response. Detracting from these
favorable factors are the relatively narrow bandwidth of the probe and
its sensitivity to frequency. Depending on the application, the probe
can be designed to have selected sensitivity and dynamic range. However,
its impedance, bandwidth, and pulse behavior are dependent upon the
component selection, which in turn, is dependent upon the desired sen
sitivity and dynamic range. It is the purpose of this Chapter to review
the system characteristics, including those of the fiber optic link.
Figure 28. Field Points Measured in the Current Strap.
z = 0 for all points (central plane of strap).
Table 7. Comparison of Measured and Calculated Field Points Within the RF Current Strap.
Point Measured Value Computed Value RMS A/m RMS A/m
1 79.9 108.5 2 81.2 108.5 3 81.6 108.4 4 80.0 108.5 5 82.7 108.4 6 78.5 108.5 7 79.3 108.5 8 79.9 108.6 9 79.3 108.0
10 79.8 108.6 11 77.8 108.2 12 78.9 108.2 13 79.1 108.6 14 78.1 107.9 15 78.5 108.6 16 78.3 108.3 17 77.7 108.3 18 78.8 108.5 19 77.9 108.4 20 79.3 108.5 21 78.5 108.6 22 79.1 108.6 23 79.2 107.4 24 78.1 108.4 25 79.1 107.4
The point numbers correspond to Figure 28.
68
Alterations of these system parameters to give desired operation for
future prototype probes are also discussed.
The prototype probe system constructed here was designed
empirically to cover fields of the order of 100 A/m at 14.25 MHz.
This probe system was then calibrated at 13.56 MHz, for measurement of
fields in the Magnetrode™ hyperthermia unit. The specific construction
of the probe is described in the Construction Chapter 3; it consists
of a 15 turn coil with a 1.2 cm diameter. The Thermistor is a Fenwal
6B32J2 with a resistor network to provide a resistance of 1710 ohms at
25°C. The following is an evaluation of this probe's characteristics
with design considerations included.
Probe Linearity
Linear response of the new version, as shown in Figure 29,
results in uniform accuracy across the entire range of the probe.
From the 13.56 MHz calibration, a linear curve fit was obtained and
is represented as the solid line on Figure 29. The values of the
fitted parameters are 42.188 for the y-intercept and 14.256 for the
slope (fitted in the linear region above about one volt output for
the probe amplifier). The fitted equation becomes,
|H field|(A/m) = 14.256X(amplifier output voltage) + 42.188 (15)
In addition, the linear response of LED probe also enables the
measurement of pulsed fields, as shown under the section entitled
Pulse Response.
1 1 0
1 OB
3 G3
5 <3
7* E3
S B
^ 0 (3 a . * -5
PHOTODETECTOR-RMPLIFIER OUTPUT (Volts)
Figure 29. 13.56 RF Probe Calibration of the LED System with Linear Approximation Represented by the Solid Line. VO
70
Sensitivity and Dynamic Range
Sensitivity and dynamic range are not separable quantities
with the optical H-field probe system. Altering the sensitivity of
the sensor will affect the useful range of the probe system. The
two, sensitivity and dynamic range, are coupled through the series
resistance (the temperature compensating network, see Figure 17) and
sensing coil of the sensor. Varying the overall series resistance of
the probe will change both the sensitivity of the probe system and its
useful range. The sensitivity and range of the probe system can be
altered by two separate techniques. Method one involves the physical
alteration of the sensor coil, while method two involves an adjustment
to the amplfier gain. Caution should be exercised when adjusting
sensitivity and dynamic range of the sensor to keep the sensor currents
below about 10 milliamperes in order to prevent self-heating of the
temperature compensating thermistor used here. Self-heating of the
thermistor will degrade the temperature compensation of the LED
junction (Fenwal Electronic, 1974).
As far as an adjustment to the sensor circuit, a linear
approximation of the circuit with Ohm's law can be used to arrive at
a desired sensor. If, for instance, the dynamic range of the probe
is to be doubled, then the series resistance of the probe can be
doubled (this assumes that operation in the nonlinear region is accept
able, this will be clarified in the example. Also, this halves the
sensitivity of the probe system.). Values for the temperature compen
sating network are then calculated via the equations of Chapter 3. The
same linear relationship can be used to alter a probe's sensitivity;
71
i.e., doubling the turns in the coil (or doubling its surface area)
will roughly double the sensitivity (and roughly halve the range,
again assuming nonlinear operation is permissible).
Similarly, the sensitivity can be increased through a linear
adjustment of the amplifier gain. Gain adjustments are made to the
second stage of the photodetector amplifier. Note, however, that at
minimum amplifier gain, full scale output of the amplifier should
correspond to no more than 10 milliamperes of current through the
probe. The use of a range switch on the amplifier will allow the use
of a probe with wide dynamic range to be of use at sensitivities down
to the noise level of the amplifiers. When initially designing a
probe sensor, the threshold voltage of the LED must be considered, as
well as the maximum permissible current for the sensor (limited by
thermistor choice). The threshold voltage for the LED is approximately
1.7 volts for the Stanley ESBR5701 LEDs used (Stanley Applications
Notes). To ensure operation of the probe system in the linear region
of the LED, the probes should be designed to have an induced voltage
in the pickup coil of at least double this value at the minimum
detectable field strength. (This is only a starting place for the
design of the probe sensor. The actual linearity of the sensor is
determined by a minimum forward current of about 1 mA; but, to
determine the actual current at the minimum field strength requires
some insight into the role of resistance associated with the probe
sensor.) Realizing these restrictions, a suitable probe sensor can be
designed as follows.
72
Example: A probe sensor is to be constructed with a 2 centi
meter diameter that will measure field strength from 100 A/m to 3000
A/m at a frequency of 100 KHz. In order to achieve linear operation
of the LED at the minimum field strengths the number of turns in the
loop is calculated as,
Vmin (min.loop voltage) N =
biu H/ • \ 0 ( m n ) (16)
3.4 volts = 137 turns
2*( 100x103) (4^x10-7) (lOOA/m) (it) (.01m) 2
At 3000 A/m the voltage in the loop will be,
3.4 X 3000 = 102.0 volts. (17) Toff
Thus to prevent the current from exceeding 10 mA, a series resistance
of at least 102.0/10 mA = 10.2 k 8 is required. From Chapter 3, the
thermistor and resistor values are computed as,
Rp = 12.21 k O
R thermistor = 20.48 k 8
R series = 2351 ft
From these calculations, a prototype probe can be developed that will
approximately cover the desired requirements given (the extreme low end
of the probe sensor's response will have some nonlinearities, because
of the LED's I-V curve at currents less than 1 mA. These nonlinearities
show up between about 255 A/m and threshold for this probe sensor.)
As can be seen by the limitations on current for linearity of the LED
(between 1 mA and 10 mA), the dynamic range of the sensors becomes
73
restricted to a single decade of magnetic field strength if linear
operation of the probe system is to be achieved. However, the upper
limit of 10 mA can be increased slightly with the proper thermistor
choice. The maximum permissible current for the LED is 100 mA. If
greater dynamic range is desired, operation of the system in the
nonlinear region must be accepted.
Probe Bandwidth and Impedance
Probe system bandwidth and impedance are directly related to
the component quality and values. For high frequency operation of
the probe, low inductance resistors should be used in an effort to
minimize possible circuit resonances. The LED alone has junction
capacitances that affect both the probe's impedance and effectively
limit the bandwidth of the probe system by introducing a resonance
into the sensor. In addition to the component effects, the proximity
of components to one another can cause unwanted coupling that can
degrade the performance of the sensor. All of these cumulative
effects are too cumbersome to accurately model by any simple small
signal analysis, but they are secondary to the operation of the
calibrated probe. Therefore, the frequency, response and impedance
characteristics are directly measured for the prototype probe con
structed. As stated before this probe was more of an empirical
design than should be necessary for future probes; however, the
operation and construction of this probe provides the necessary
insight to formulate design criteria.
74
First to be considered is the resulting impedance of the LED
optical H-field probe. Table 8 shows the impedance of the probe
measured at the calibration connections shown in Figure 39, Appendix
B, (connections made to test points 0 and 2 with the indicated break
in the sensor loop). The impedance was measured over the frequency
range of 0.5 MHz to 68 MHz with a bias current of 3.5 milliamperes.
An HP 4815A Vector Impedance Bridge was used to measure the impedance,
with the bias supply isolated from the bridge with two 2.5 millihenry
RF chokes. The bias current of 3.5 mA was chosen because this provided
a reasonable mean value of the current for the probe. Furthermore,
it was determined that the impedance was independent of the sensor
current, provided that the LED was excited beyond 0.19 mA. From
Table 8 the useable frequency range of this particular probe can be
determined. Up to 55MHz the probe exhibits a relatively smoothly
varying impedance, at which point the phase angle of the impedance
begins to change rapidly. The probe is operational to this frequency,
provided the length of wire used to wind the coil is appreciably less
than a wavelength. (If the wire is too long the uniform current argu
ment for the solenoid fails, and the accuracy of the probe becomes
questionable.) Note also that the sensitivity of the probe is fre
quency dependent by equation 1.
From these impedances of the probe sensor, some concept of the
field loading (field perturbation) can be obtained. At some value of
applied H-field, the open circuit voltage of the coil is found from
equation 1.
75
Table 8. Probe Impedance as a Function of Frequency Measured Between Test Points 0 and 2 With a DC Bias Current of 3.5 Milli-amperes
Frequency Magnitude Phase Frequency Magnitude Phase (MHz) (MHz)
.5 1700 +12 35.1 910 -26 1.0 1725 + 4 36.1 900 -26 2.0 1700 - 3 37.2 900 -26 3.0 1700 - 7 38.3 900 -26 4.0 1650 -10 39.1 880 -26 5.0 1600 -13 40.2 870 -27 6.0 1550 -16 41.2 880 -26 7.0 1525 -18 42.2 880 -26 8.0 1475 -20 42.5 900 -25 9.0 1450 -22 43.4 960 -26
10.0 1400 -24 44.6 1000 -27 11.0 1350 -25 45.4 1020 -29 12.0 1325 -27 46.3 1050 -31 13.0 1275 -28 47.2 1050 -34 14.0 1225 -28 48.2 1050 -36 15.0 1200 -29 49.2 1025 -35 16.0 1175 -30 50.4 1025 -35 17.0 1125 -32 51.3 1050 -37 18.0 1075 -32 52.2 1025 -38 19.0 1050 -31 53.3 1000 -36 20.0 1050 -31 54.2 1050 -32 21.0 1025 -32 55.3 1225 -32 22.0 1000 -32 56.3 1375 -40 23.0 970 -31 57.2 1400 -47 24.0 960 -31 58.3 1375 -55 25.0 960 -31 59.2 1325 -60 26.0 940 -31 60.2 1250 -64 27.0 920 -31 61.2 1200 -67 28.0 900 -32 62.3 1150 -71 29.0 850 -33 63.2 1075 -74 30.1 790 -27 64.0 1025 -76 31.0 850 -20 65.1 950 -78 32.1 900 -24 66.3 900 -79 33.0 910 -25 67.3 850 -80 34.0 920 -26 68.2 820 -81
Bias current of 3.5 mi 11 amperes.
76
From this and the impedance of the loop, the loop's current can be
calculated as,
N(oy0<Hzm>Trr2
Iloop (approximate) = (18) m
<^zm> = measured Hz field at sensor (averaged over sensor loop area)
Z = approximate loop impedance
By considering the average perturbation of the field over the area of
the sensor the perturbation can be calculated from this induced
current as follows,
LIloop = N <j> " « No<Hzp>,,r , ,
(19) LIloop Lai
<Hzp> = <H2m> (combining with 18)
N u07rr2 |Z|
L = loop inductance (3.21 uH)
N = 15 (number of turns in sensor) turns
r = .62 cm (loop radius)
<HZp> = average perturbation over the area of the sensor for the <Hzm>
field measured
where the inductance value of 3.21 uH was both measured with an HP
vector impedance bridge and checked against standard single layer
solenoid equations (Grover, 1981). Note: This is only a perturbation
of the field during the measurement process; the calibration of the
sensor yields the field as if the sensor were absent during the
measurement. This is the beauty of the network analysis. Table 9
shows the computed perturbation of the field due to the sensor at
13.56 MHz, for the field probe constructed.
77
Table 9. Computed RMS LED Sensor Current and Associated Field Perturbation for Some Hypothetical Values of Magnetic Fields.
Measured H-field Ma Perturbation A/m Perturbation
45.7 7.14 10.0 52.1 8.11 11.4 54.4 8.45 11.9 58.5 9.09 12.8 59.9 9.32 13.1 65.0 10.07 14.2 67.6 10.55 14.8 73.1 11.35 16.0 78.2 12.17 17.1 83.7 12.99 18.3 88.7 13.79 19.4 93.7 14.61 20.5 99.2 15.41 21.7
104.2 16.23 22.8 109.2 17.03 23.9 114.7 17.85 25.1 119.7 18.67 26.2 125.2 19.47 27.4 130.3 20.29 28.5 135.3 21.09 29.6 140.8 21.91 30.8 145.8 22.73 31.9 156.3 24.34 34.2 182.4 28.40 39.9
This is the average perturbation over the internal area of the sensing coil.
Pulse Response
The response of the probe system to a pulsed RF electromagnetic
field was also determined. The ability of the probe to measure such
fields is crucial in certain high power athermal bio-electromagnetic
experiments. The set-up used to generate the fields is similar to
that used for some earlier pulsed athermal experiments on a RAT LA24
cell line (Jones et al., 1983). This experimental arrangement, and
the set-up used for the strap field generation, are the same, except
for the addition of a solid state pulse generator used to modulate
the RF signal. Photographs and schematics of the standard coil and
matching network are shown in Chapter 4. Figure 30 is an oscilloscope
trace of the pulse optical detector/amplifier output that shows the
response of probe system to the pulsed RF signal, illustrated by the
oscillogram of Figure 31. The response of the probe system appears
to be sufficient to follow the envelope of the waveform, but not the
14.25 MHz carrier. As can be seen from the oscilloscope trace, the
10 to 90% rise time of the probe system exceeds 5 microseconds. It
is unclear as to where exactly the limitation exists that prevents
the sensor from displaying the actual 14.25 MHz carrier. Only with
more testing can the answer be determined. It appears, however, that
the limitation is probably in the detector amplifier unit, because
the LED's rise time is about 1 nanosecond (Stanley LED characteristics
notes), and the L/R time constant of the sensor is only about 2.6
nanoseconds. Nevertheless, furthur testing is needed to determine
the exact location of the performance flaw.
Figure 30. Output of the Pulse Amplifer Viewed on Oscilliscope.
The RF excitation envelope is shown in Figure 31. Sweep = 10 us/division, vertical scale = 5 volts/division.
80
Figure 31. RF Excitation Waveform of the Current Strap Used to Determine LED Pulse Response Characteristics.
Sweep = 10 us/division, vertical scale = .02 volts/division.
81
Thermal Stability and Optical Connection Repeatiblity
The thermal stability of the empirically temperature compen
sated LED sensor was determined with the HP DAS by the same set up
used to determine the temperature drift of the LED. Figure 32 shows
the resulting system sensitivity, with the sensor excited by a stable
DC voltage that produced a median output on the system of 4.225 volts
at 25°C. As can be seen, the probe is stable to within +1% with this
compensation over the range of temperatures from 25° to 50°C.
The ability to swap from one sensor to another and maintain
system calibration of each sensor is a desirable characteristic of a
stable system. Therefore, a test of the repeatability of the optical
fiber connection at the photodetector was performed. The LED was
excited with a stable DC current to produce an output voltage of about
4.2 volts. The Amphenol fiber then was disconnected and reconnected
multiple times at 90 degree rotational increments. The results are
shown in Table 10, where repeatability of the optical connection was
withi n +3%.
4 . 2 G W +J
O 4 . 2 5 >
Z) CL
4 .
g 4 . 23
a: u i - t 4 . 2 2 u.
_l CL z CE I
&.
O I-u u H-u a o i-0 1 a.
4 . 2 1
1 3
1 B
1 7" 2 5 3 0 3 5 4 0 4 5 5 0
SENSOR TEMPERRTURE (Celsius)
5 5
Figure 32. Temperature Stability of the Empirically Compensated LED RF Magnetic Field Sensor.
83
Table 10. Repeatability of Optical Fiber Connection on Amplifer Chassis.
Trial No.
Rotation of Connector
Trial No. 0° O O
o>
1—» 00
o o 270°
1 4.173 4.196 4.280 4.205
2 4.175 4.201 4.265 4.196
3 4.202 4.208 4.242 4.186
4 4.180 4.195 4.265 4.156
5 4.177 4.193 4.268 4.187
Five connections and disconnections were made at each of the 90° increments of rotation.
CHAPTER 5
CONCLUSION
Summary
A linear, easily calibrated, accurate, and stable RF magnetic
field probe system was developed and described that enables accurate
mapping of strong fields in both the CW and pulse modulated domains.
Although the probe system is considerably more complex than the prior
system of Oleson, the improved characteristics more than compensate
for this inconvenience.
Linearity in magnetic field measurements was the main objective
obtained with the LED version of the optical magnetic field probe. As
is shown in Figure 29 of Chapter 4, the probe system can be calibrated
and the output voltage fitted to a linear equation over the nearly
linear region of the probe (above about one volt output on the LED
probe system). Over this region of the probe, the linear fit can
provide agreement to the calibration data. The linearity of the
probe not only allows this simple calibration equation, but also
enables the measurement of modulated fields without nonlinear distor
tions. In addition, the use of an LED rather than an incandescent
lamp at the sensor allows the measurement of these dynamic fields
when they are rapidly varying. The prototype probe system developed
can measure fields that have rise times of at least 5 microseconds.
84
85
The added features of linearity and system response could
conceivably complicate the system's calibration. However, a simple
bench top calibration procedure was developed for the LED version of
the optical magnetic field probe system. Initially, a rigorous
description of the sensor and its behavior was attempted in an effort
to calibrate the probe with a stable direct current. Unlike Oleson's
incandescent probe, a first principle analysis of the LED probe
appeared to be quite complex. Even so, a simple and accurate calibra
tion of the sensor eventually was obtained by an alternative method,
involving the excitation of the sensors with RF current. Before
calibration of the new probe, the RF calibration technique was tested
on the earlier incandescent version of the probe system and then used
with confidence to calibrate the LED version. In addition, the
simplicity of the procedure enhances the accuracy of the prior probe
by limiting the chances of measurement errors during the calibration
procedure.
Good system accuracy was also achieved with the calibrated LED
probe system. A comparison of the accuracy of the two probe systems
(incandescent and LED versions) is not reasonable at this point,
because the two probes were constructed to cover a different range of
field strengths. However, accuracy of the new probe as compared to
solenoidally generated fields is +_ 5% between probe threshold and 100
A/m at 13.56 MHz (computed from Table 5, Chapter 4). At the low end
of the old version (where the new probe system works quite well), the
incandescent probe system accuracy is somewhat less than this. Note,
86
however, that the old probe was not designed for sensitivity in this
region. The nonlinear response of the incandescent filament, i.e.,
the slope of the calibration curve (H-field versus output), is nearly
vertical in this region. Linear response obtained by the new version,
as shown in Figure 29, results in a constant accuracy across the
entire range of the LED probe system. A comparison between probe
systems at levels of H-fields above 130 A/m (where both the incan
descent probe system and the LED system are known to be accurate)
shows agreement to within 2% in both the standard solenoid measurements
and the measurements made in the Magnetrode™ applicator.
Stability, i.e., the ability of the system to maintain accuracy
over varying environmental factors, is also an important factor to be
considered. As shown in Chapter 4, the thermal stability of the
sensor with temperature compensation provided stability of the output
of the system to within + 1% over temperatures ranging from 25 to
50°C. In addition to thermal stability, the system was tested for
sensor connection repeatiblity. Here, the sensor of the probe system
was repeatedly connected and disconnected at the amplifier to determine
the feasibility of making interchangeable sensors. The sensor recon
nect!' on error was determined to be only about +3%.
Future Considerations
The probe still has many disadvantages. The response of the
probe is, as is the previous version, frequency sensitive (accurate
measurements of r.f. magnetic fields rely on the presense of only
87
negligible amounts of harmonic energy in the measured field), and
directional. In addition, the suseptibility of the LED probe system
to strong electric fields is not known. The directionality of the
probe can be improved upon by the addition of orthogonal sensors
combined in a single package. Remedies to the frequency sensitivity
of the sensor is not trivial and will be mentioned only briefly, along
with a few simple and easily made system modifications.
Future generations of the LED field probe can be improved
through the addition of a few simple features. The ability to make
quick and repeatable connections of the optic fiber will allow the
use of many probes to cover various frequency and magnitude windows.
One modification would only require the use of oversize photode-
tectors to reduce alignment problems (if less than +3% is desired)
between the optic fibers and the detectors. The addition of the range
selector switch to allow the use of sensors with a greater dynamic
range is also a simple yet useful modification. Probably the most
significant modification would be the development of a frequency
compensated broadband sensor. Introduction of the proper components
to the sensors could enable this feature (Kanda et al., 1982; Nahman
et al, 1985). Finally, the response time of the amplifiers for the
pulsed field measurements needs improvement, if sharp transient
measurements are to be realized.
APPENDIX A
THE MODEL EQUATION
In Chapter 3, an attempt to calibrate with a direct current
procedure is mentioned with a more complete description of the attempt
in Appendix B. For that DC attempt at calibration, a model for the
terminal characteristics of the LED was developed. Although the
modeling of the junction is of little importance to the present
operation of the probe, much work was involved in its development and
it is therefore included in this appendix.
For GaAlAs junctions, the following relationship for the I-V
characteristics holds:
Constants Jdo and Jrgo are to be determined empirically for the
junction. By the use of an HP 9836 data acquisition system, the I-V
characteristics were found to be as in the graph of Figure 33. It is
worthwhile to note that these data should be taken at some known
junction temperature if extreme accuracy is desired. These data were
then used in an HP curve fitting routine where values of Jdo and Jrgo
were determined. However, because equation 20 gives a relationship for
current as a function of voltage and the parameter for the calibration
(20)
(William and Hall, 1978).
88
89
equation 5 is the forward diode voltage (Vd), it was necessary to
rewrite equation 20 with voltage as a function of the LED current,
and thereby the loop current. To rewrite the equation, a substitution
was made with subsequent application of the quadratic formula:
(qv \ 1 ( 2 1 )
2kTI
solving for y ,
-Jrgo +yjrgo2 + 4 • Jdo • id v = — (22)
2Jdo
solving for v
2kT v In (M) (23)
q
This equation then was modified by the addition of a junction resistance
term Rj, and curfit to the original data once again but allowing only
Rj to vary.
2kT V = In u +Rj • id (24)
q
To obtain greater confidence in the determination of the empirical
constants, Jdo and Jrgo, an iteration with the curve fitting program
on these parameters was performed again. Taking the equation that
includes the series junction resistance, Rj, and solving in terms of
id (as a function of Vd), and allowing only the exponential values
Jdo and Jrgo to vary, we obtain more suitable values for Jdo and Jrgo
with a model that includes the series junction resistance, Rj. A
final iteration was performed (as before on Rj) in an effort to
90
converge on a more accurate value of Rj. As can be seen in the
graphs of Figures 34 and 35 (low LED currents in 35), excellent
agreement with experimental data was achieved after only one iteration.
The values obtained for Jdo, Jrgo, and Rj were 7.860E-31, 4.821E-18,
and 6.6 ohms, respectively. This model equation is, however, only
valid at the temperature in which the data was taken (25°C in this
case). The empirically determined constants Jdo and Jrgo are
temperature sensitive and either need to be recomputed at other
temperatures or compensated for by some other means. Because modeling
the LED's terminal characteristics (and not the physics of the
junction) is all that is of concern for calibration, the temperature
sensitivity can be approximated by the addition of a voltage transla
tion term:
Vd = 2kT°ln(u) + Rj id + K(T-298) (T° = 298° Kelvin) (25)
where K is an empirical constant equal to -.0019 V/°Kelvin. The
inclusion of K provides a complete description of the junction
characteristics as shown in Figure 36, where the model is compared to
the measured LED characteristics at various temperatures.
. 006 r~
. 005
. 004
. 003
. 0 0 2
0 0 1
3 5 -wL 1 . 4 1 . <4 1 . S 1 . S ! l . 7
LED FORWARD VOLTAGE (Volts)
Figure 33. I-V Characteristics of a GaAlAs LED (Stanley ESBR 5701) KO
. 00 1 G
M . 00 1 4 OJ i . (D Q. e . 0 0 is
z . 0 0 1 u o: k. D (J .0005
P o: CE 2 . 000G — QL O b. Q . 0 0 0 *4 U _l
. 0 0 0 2 —
0
1.44 1.4G 1.48 1.50 1.52 1.54 1.5G 1.58 1.60 1 .62
LED FORWRRD VOLTRGE (Volts)
Figure 34. Agreement Between LED Data and the Curve Fit Model Equation for Currents up to 1.6 mA.
Solid line is model equation. ro
. 000 1 S I—
. 0 0 0 1 4 — W <L) 4 . (D . 000 1 2
cc
t r . 0 0 0 l —
u QL a.
3 b - e - 5
Q o: CE B . EI —5 Z CK o u.
Q U _J
4 . E I -5 —
2 . E I -5 —
0. 1 . 3 1 . 3 1 . 4 1 . 4 1 . 5 1 . 5
LED FORWRRD VOLTAGE (Volts)
Figure 35. Agreement Between LED Data and the Curve Fit Model Equation at Low Current Levels (below .16 mA).
TEMPERRTURE (Celsius)
Q. 8
CE
Z u a. QL D u Q d: <x 2
idea u _i
3 5
LED FORWARD VOLTRGE (Volts)
Figure 36. Comparison Between LED Data and the Equation that Includes Temperature Sensitivity of the I-V Characteristics of the Junction.
Solid line is model equation.
APPENDIX B
TRIAL FORMULATION OF A DIRECT CURRENT CALIBRATION PROCEDURE
FOR THE LED RF MAGNETIC FIELD PROBE
Unlike Oleson's probe, the LED versions response cannot be
directly correlated to the power dissipation in the LED junction
because of the presence of other power consuming elements in the
circuit (i.e., the current limiting resistance and temperature compen
sating network). However, the ease of generating and accurately
measuring direct currents in the laboratory makes calibration by DC
particularly appealing. The following is an attempt to formulate a
DC calibration procedure that relies on the following assumptions:
(1) the response time of the LED and associated circuitry is fast
enough to follow the rf sinusoidal excitation, (2) stray capacitances
and couplings minimally effect the model, and (3) the the detector
responds to the average illumination of the LED. In an effort to
calibrate with DC, a value of current must be formulated that produces
the same illumination as the RF current during field measurements.
By combining equation 1 and the model equation for the LED's forward
characteristics, equation 25, an equation relating the loop voltage
to loop current is obtained,
( 2 6 )
95
96
where y, Rj, and K are given in Appendix A
R(to) = the approximate sensor resistance (this assumes no reactive loop components)
Vloop = sin (wt)
By replacing V]0op with VOsin(u)t) this expression represents the
circuits current in the time domain. The linear response of the
LED's intensity to it's forward current in conjunction with assumption
3, the assumed detectors response, are the keys to a DC calibration.
In essence, a direct current equivalent to the time average of the
current from relation 4.2 will result in the same illumination of the
LED as the RF current induced in the loop. From the analysis of
Chapter 2, expression (1) and V0> a value of the H-field can be
computed. It follows that averaging of equation 26 results in the
evaluation of the integral,
1 IT idave =— I ^ d0 over one period. (27)
2 TT 0
Replacing id by equation 26 we have,
1 JZ -n V0 sin 0 - (q~~*)ln(y)+ Rj*id + K*(T-248) idave = | do (28)
2Tr JQ R(a>)
This expression is a transcendental integral equation (TIE). The
evaluation of this equation is not closed form obtainable and must be
solved by an iterative technique. The solution was obtained through
the use of a Newton-Raphson iteration of the integrand with the
simultaneous application of a Simpson integration (Stark, 1970). To
verify the accuracy of the integration the loop is excited with a DC
97
voltage that is stepped through a sinusoidal waveform via an HP data
acquisition system (DAS). The set up is shown in Figure 37, with a
schematic of the junction block in Figure 38. Figure 39 represents
the prototype probe complete with temperature compensation,protection
diodes, and test points. The junction block enables the monitoring of
voltage at strategic locations within the sensor, as well as the DC
current in the sensor. (In addition to verification of the calibra
tion, this system was used to obtain the data to fit the model LED
equation and to obtain the temperature sensitivity data.) Figures 40,
41 and 42 correspond to values of current measured in the loop, with
10.0, 3.0 and 1.8 volt peak amplitudes of the stepped sinusoid,
respectively. A computation of the area under these curves averaged
over their period (2 IT) should correlate with the integration of the
TIE. Table 11 shows the agreement between the measured and calculated
average.values of current for these cases. By injection of this DC
value of current into the loop, a calibration of the photodetector-
amplifier is obtained. By knowing the predicted H-field for each DC
current and the output voltage of the photodetector that this injected
current produces, the calibration graph of Figure 43 was formed at
13.56 MHz. Figure 8, Chapter 3, shows the completed DC calibration
of RF magnetic field related to the output of the photodetector
amplifer that these DC injected currents produced. As shown in
Chapter 3, the DC calibration by this static approach to the solution
of the sensor's response to RF currents is incorrect. Although the
modeling of the LED junction and average loop current are accurate in
98
the static or nearly static case, their performance in the RF domain
is vastly different. The neglection of junction capacitances and
stray loop or component capacitances in the time domain analysis could
account for the discrepancy. In addition, the response of the
photodetector-amplifier to RF rather than DC is also an unknown.
Table 11. Agreement Between Measured and Calculated Average Currents
Vo Peak Measured Computed
10 1.28 ma 1.30 ma
3 1.56 x 10~1 ma 1.63 x 10"1 ma
1.8 1.76 x 10-5 ma 1.77 x iq-5 ma
However, a more rigorous approach to the modeling of the sensor, along
with well defined photodectors and photoamplifiers (possibly the use
of peak detectors) could result in a DC calibration technique. Modeling
of the sensor would have to include all of the time dependent terms.
Figure 44 suggests a possible model that includes the loop inductances
and junction capacitances of the sensor. The set of state equations
to be solved for this situation would be as follows (Huelsman, 1972):
di V(+) -l iR - Vd(ii) = 0 (29)
dt
11 = i - i2 (3°)
dVd 12 = C (31)
dt
D/A
A/D 3-LEAD TEST
I CABLE
PROBE
JUNCTION
BLOCK
OPTIC FIBER TO DETECTOR
Figure 37. Setup with Hewlett Packard Data Acquisition System (HP DAS) to Test Accuracy of Calibration Integration.
This is also the setup used for determining I-V Characteristics and Temperature sensitivities.
FROM VOLTAGE D/A
l&R •Wr
U <i
14 1 PROBE CURRENT MEASUREMENT
VOLTAGE ACROSS TEMPERATURE COMPENSATION NETWORK
LED FORWARD VOLTAGE
APPLIED LOOP VOLTAGE
PROBE OUTPUT VOLTAGE
9 BNC TO OPTICAL DETECTOR/AMPLIFIER
Figure 38. Schematic of Junction Block Shown in Figure 37. O o
7*T 0 2
3038
•wv
1951
I IN9I4 LED ESBR 5701
!
Figure 39. Prototype Probe Schematic with Various Test Points used for Calibration and Sensor Testing.
W <D (-01 a e a:
i-2 U Q£ QL D U Q. O O _J
a. o m z u
. 001 5 i—
. B04 —
. 003 5
. 003
. 0025 —
0 0 2 —
. 00 1 5
. 0 0 1
. 0005 —
0 3 . 5
HNGULRR INCREMENT (radians)
Figure 40. Currents Stepped Through with HP DAS to Determine the Accuracy of the Integration of the Transcendental Integral Equation.
Here a 10 volt sinewave was stepped through one half period. O ro
• 0 0 0 6
a
01 2
ANGULAR INCREMENT (radians)
Figure 41. Currents Stepped Through with HP DAS to Determine the Accuracy of the Integration of the Transcendental Integral Equation.
Here a 3.0 volt sinewave was stepped through one half period. o
M <U i. <1) Q. E a:
u o: en Z) u Q. O o
QL O cn z Ld
. 0 0 0 1 4
.. 000 12 —
. 0 0 0 i —
B . EI-5 —
G . E — 5 —
4 . EI-5 —
2 . e: —;
0 J 1 . 5 2 " 2 . 5
RNGULRR INCREMENT (radians)
3 . 5
Figure 42. Currents Stepped Through with HP DAS to Determine the Accuracy of the Integration of the Transcendental Integral Equation.
Here a 1.8 volt sinewave was stepped through one half period.
e \ CE
U
U M H-U z (J CE
L_ 0
UJ Z) _J CE >
01 z: o:
Q U I-u t-1 p u q: Q.
3 0
B 0
7> 0
S 0
50
4 0
3 0
2 0
1 0
0 0
1 1 I I I I I 1 L 1 1 . 5 2 2 . 5 3 3 . 5 4 < 3 . 5 5
DC CflLIBRRTION CURRENT ( m i 1 1 i Flmpe r es )
5 .
Figure 43. DC Calibration Currents and RF Magnetic Fields Predicted for the DC Calibration Attempt. 0 01
I •
ve t ) Q | i2(t)T tC dVd(t)
dt
Figure 44. More Complete RF Network Analysis of the Sensor Circuitry.
APPENDIX C
TEMPERATURE COMPENSATION PROGRAM
This appendix contains a program listing and sample run for
the computation of thermistor and resistor values for temperature
compensation of the Light Emitting Diode sensor. The program was
written for a Hewlett Packard 41CX hand held computer.
01 LBL 'TX 31 STO 05 61 RCL 04 02 'R AT Tl? 32 RCL 01 62 RCL 03 03 PROMPT 33 SQRT 63 -
04 STO 01 34 RCL 04 64 RCL 07 05 T R AT T2? 35 * 65 *
06 PROMPT 36 RCL 02 66 *
07 STO 02 37 SQRT 67 RCL 02 08 T Tl? 38 RCL 03 68 RCL 01 09 PROMPT 39 * 69 -
10 273.15 40 - 70 RCL 06 11 + 41 RCL 05 71 *
12 STO 03 42 / 72 / 13 T T2? 43 STO 06 73 STO 08 14 PROMPT 44 T DELTA 74 T RP = 15 273.15 45 PROMPT 75 ARCL x 16 + 46 STO 07 76 AVIEW 17 STO 04 47 RCL 01 77 STOP 18 RCL 02 48 RCL 02 78 RCL 06 19 SQRT 49 + 79 *
20 RCL 01 50 RCL 06 80 T R at 25C= 21 * 51 * 81 ARCL x 22 RCL 03 52 RCL 06 82 AVIEW 23 * 53 x + 2 83 .END.REG 173 24 RCL 01 54 RCL 01 25 SQRT 55 *
26 RCL 02 56 RCL 02 27 * 57 *
28 RCL 04 58 + 29 * 59 1 30 - 60 +
107
108
Sample program run:
Hit R/S
Response "R AT Tl?"
Enter the normalized thermistor resistence at the lower temperature
limit for compensation and hit the R/S key.
Response "R AT 12?"
Enter the normalized thermistor resaistence at the upper temperature
limit and hit R/S.
Response "Tl?"
Enter the lower temperature limit and hit R/S.
Response "T2?"
Enter the upper temperature limit and hit R/S.
Response "DELTA?"
Enter the product of the desired loop resistance (ohms) and the
percent variation to be compensated (%/degree Celius) (the product is
the change in resistance per degree Celcius).
Response "Rp=XXXX" (XXXX is the resistance value in ohms).
Hit R/S.
Response "R AT 25C = XXXX" (XXXX is the thermistor resistor
resistance in ohms at 25°C).
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Burr Brown. (1984) Product Data Book, Burr Brown Corporation, Tucson Arizona, pp. 1, 159-166.
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Cetas, T.C. and Roemer, R.B. (1984) Status and Future Developments in the Physical Aspects of Hyperthermia. Cancer Res. (suppl 44) 4894s-4901s.
Datel-Intersil Manufacturers (1982) Specifications of the AM490-2 Chopper Stabilized Operational Amplifier, 11 Cabot Blvd., Mansfield, MA 02048.
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Gray, P.R. and Meyer, R.G. (1977) Analysis and Design of Analog Integrated Circuits, John WiTey and Sons, Inc., New York.
Grover, F.W. (1981) Inductance Calculations. Working Formulas and Tables, D. Van Nostrand Co., Inc., New York.
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Jaffe, S. (1984) Temperature Compensation Using Thermistors, Microwaves and RF, 24, 101-104.
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