Octave Bandwidth Orthomode Transducers for the EVLA

1910 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 59, NO. 6, JUNE 2011

Octave Bandwidth Orthomode Transducers for theExpanded Very Large Array

Gordon M. Coutts, Member, IEEE

Abstract—Quadruple-ridge orthomode transducers (OMTs)have been designed to operate over a full octave bandwidth for theexpanded very large array (EVLA) project. The OMT separateslinearly polarized signal components by matching a circular wave-guide input to two orthogonal coaxial outputs. The OMT is used inconjunction with a quadrature hybrid to detect circularly polar-ized signal components. This paper focuses on the 1 GHz–2 GHzL-Band OMT design, which has better than 18.8 dB measuredreturn loss across the band, with no evidence of trapped-mode res-onances. The OMT is designed with an emphasis on performance,ease of tuning and manufacturability since a large number ofunits are needed for the array application. Extensive parametricanalyses were carried out, and nominal dimensions have beenset to ensure the devices exceed RF specifications provided theparts are machined to within specified tolerances. With excellentwideband performance and a simplified manufacturing process,the proposed OMT would be amenable to much larger futurearray projects.

Index Terms—Orthomode transducers (OMTs), radio as-tronomy, reflector antenna feeds, ridge waveguides.

I. INTRODUCTION

T HE very large array (VLA) of 27 radio telescopes, lo-cated on the Plains of San Agustin, New Mexico, USA,

was originally constructed over several years, and completed in1980 [1]. The expanded very large array (EVLA) project is asignificant upgrade to modernize the electronics of the existing27 radio telescopes to greatly improve sensitivity, spectral res-olution and frequency bandwidth [2]. Eight new cryogenicallycooled EVLA receivers provide continuous frequency coveragefrom 1 GHz to 50 GHz for each antenna. Three of the new re-ceivers cover the 1 GHz to 8 GHz frequency range, which issubdivided into L-band (1 GHz to 2 GHz), S-Band (2 GHz to 4GHz), and C-band (4 GHz to 8 GHz).

The three aforementioned low frequency receivers require po-larizers that operate over a full octave bandwidth to detect cir-cularly polarized signals. The octave bandwidth circular polar-izers consist of a quadruple-ridge orthomode transducer (OMT)followed by a quadrature hybrid [2], [3]. The OMT first sepa-rates the orthogonal linearly polarized signal components. Thequadrature hybrid then adds the orthogonal linearly polarized

Manuscript received November 25, 2009; revised March 30, 2010; acceptedApril 28, 2010. Date of publication March 03, 2011; date of current version June02, 2011. The National Radio Astronomy Observatory is a facility of the Na-tional Science Foundation operated under cooperative agreement by AssociatedUniversities, Inc.

The author is with the National Radio Astronomy Observatory, Socorro, NM87801 USA (e-mail: [email protected]).

Digital Object Identifier 10.1109/TAP.2011.2122220

components together with the appropriate phase shift to detectcircularly polarized astronomical signals [2], [3].

Numerous wideband OMT designs have been previously re-ported, which focus on improving manufacturability and re-ducing size. Coupled waveguide based designs operate over acomplete waveguide band, and perform well in terms of inser-tion loss and cross-polarized isolation [4]–[6]. The Bøifot OMT[6] is designed for ease of machining by using a split-block de-sign that can be bolted together. In [7], the Bøifot OMT has beenmodified by replacing the septum and inductive posts of the de-sign in [6] with a double-ridged waveguide junction to simplifymanufacturing of a physically small device that operates in the2 mm band. A double-ridge OMT with orthogonal double-ridgesections separated along the propagation direction has been re-ported in [8], which performs well from 16 GHz to 26 GHz.

Quadruple-ridge OMT designs have been reported whichhave the advantage of wider single-mode bandwidth com-pared to OMT designs based on square and circular waveguides[9]–[12]. The wideband quadruple-ridge OMT of [9] has a widerbandwidth than the OMTs of [4]–[8], however the cross-po-larized isolation performance is limited. The quadruple-ridgeOMT of [10] does not meet the EVLA 15 dB return loss speci-fication over an octave bandwidth, and has evidence of in-bandtrapped-mode resonances. A quadruple-ridged horn has beenreported in [11] which has broadband performance, but wouldnot meet the EVLA return loss specification. In [12], trappedmode resonances are mitigated by offsetting the ridge tapersalong the propagation direction, and device performs well from2.4 GHz to 4 GHz.

The EVLA L-Band OMT offers full octave bandwidth perfor-mance with 18.8 dB or better return loss and 35.7 dB or bettercross-polarized isolation across the 1 GHz–2 GHz bandwidth,while suppressing potentially occurring in-band trapped-moderesonances. The insertion loss and return loss are well matchedbetween the two orthogonal polarizations. Consequently, theEVLA L-Band OMT has a close amplitude match that mini-mizes errors when adding the phase-shifted orthogonal signalstogether for circular polarization detection.

The EVLA S-Band OMT was reported in [3], which was de-signed with an emphasis on manufacturability since the projectrequires 30 units (one for each of the 27 antennas and 3 spares).Ease of manufacture, however, is limited due to a number ofsmall components that must be assembled in a complex process.The new L-Band OMT has been designed to greatly simplifythe assembly procedure over previous designs. Although it isimportant to focus on manufacturability when designing com-ponents for the 27-element EVLA, the new L-band OMT wouldbe amenable to much larger future array projects.

U.S. Government work not protected by U.S. copyright.

COUTTS: OCTAVE BANDWIDTH ORTHOMODE TRANSDUCERS FOR THE EXPANDED VERY LARGE ARRAY 1911

Fig. 1. HFSS [13] model of the L-Band quadruple-ridge OMT. (a) L-BandOMT model (shaded regions are vacuum), (b) L-Band OMT side view (shadedregions are vacuum), (c) Coax to quadruple-ridge waveguide model with trans-parent ridge to illustrate shorting block structure and coaxial feeds. The centerconductor and dielectric of each coaxial feed extends across the ridge gap andinto a hole in the opposing ridge. (shaded regions are solid).

The L-Band OMT electromagnetic design, with its simplifiedmanufacturing process and octave bandwidth operation whilesuppressing trapped-mode resonances, is outlined in Section II.Section III discusses the manufacturability of the new OMT de-sign, including parametric analyses of critical parameters. Ex-perimental L-Band OMT results are presented in Section IValongside simulated results.

II. OMT DESIGN

This section provides a detailed description of the L-bandOMT design, which is based on the concept of the quadruple-ridge S-Band OMT of [3]. The new OMT is designed to operatefrom 1 GHz to 2 GHz, with a novel transition from quadruple-ridge waveguide to coaxial transmission lines that has improvedmanufacturability compared to previous designs.

The octave-bandwidth EVLA OMT designs are subject tostringent specifications. The devices must match a circularwaveguide input, which connects to the antenna feed, to twoorthogonal 50 ohm coaxial outputs with a return loss betterthan 15 dB. To prevent a loss of receiver sensitivity at some

frequencies, higher-order modes and trapped-mode resonancesare suppressed over the entire octave bandwidth.

The OMT structure may be subdivided into three sections asshown in Fig. 1. The first OMT section is the circular-to-squaretransition that connects to the circular waveguide at the receiverinput. Next is the square to quadruple-ridge tapered transition,followed by the coaxial to quadruple-ridge waveguide transi-tion. The total OMT length is 26.4 inches.

A square OMT cross-section was chosen since the squarestructure is easier to fabricate. Four outer shell sections and fourridges may be fabricated separately and bolted together. Thesquare waveguide cross-sectional dimension adjacent to the cir-cular-to-square transition is 6.5 on each side. The /TEmode cutoff frequency is therefore 908 MHz, whereas the 7.5diameter circular waveguide that connects to the antenna feedhorn has a 923 MHz cutoff frequency. The circular to squaretransition is a 2.4 long section of rounded square waveguide.

A. Octave-Band Impedance Match

The wide OMT bandwidth presents an interesting challenge:to design a device that is well matched in terms of impedance,while at the same time suppresses higher order modes. The 908MHz square waveguide cutoff frequency is close to the 1 GHzlower band edge to minimize the number of in-band higher ordermodes that may propagate. As a result of operating close tocutoff, the square waveguide impedance varies between 1113ohms at 1 GHz and 524 ohms at 2 GHz, and must simultane-ously be matched to the 50 ohm coaxial outputs.

For impedance matching purposes, the square waveguidenear the OMT input transitions to a chamfered quadruple-ridgewaveguide along a smooth taper as shown in Fig. 1(a),(b). Thequadruple-ridge waveguide dimensions at the coaxial transition,shown in Fig. 2, consist of a 45 degree ridge chamfer, having awidth varying between 500 mil and 156 mil. The gap betweenopposing ridges is 240 mil. The chamfered profile enables thegaps between the four ridges to be sufficiently small to havewaveguide impedance values close to 50 ohms for the dominant

and modes. The overall 500 mil ridge width waschosen to provide a sufficient overlapping chamfer area toobtain a quadruple-ridge waveguide impedance close to 50ohms, while maintaining a relatively large ridge gap comparedto the manufacturing tolerances. This enables fabrication of anOMT that can be bolted together without the need to adjust theridge gap during the tuning procedure.

At 1 GHz and 2 GHz, the quadruple-ridge waveguideimpedance is 59.6 ohms and 51.4 ohms, respectively, for thedominant modes. The percentage impedance variation acrossthe band is significantly reduced in comparison to the squarewaveguide since the quadruple-ridge waveguide operates far-ther from cutoff.

The coaxial to quadruple-ridge waveguide transition is shownin Fig. 1(c), where orthogonally positioned coaxial probes ori-ented along the y and x axes excite the and modesrespectively. The mode has x-oriented electric fields asshown in Fig. 2(a), whereas the mode has y-oriented elec-tric fields as shown in Fig. 2(b). Each pair of opposing ridges isconnected together with a series of 4 shorting blocks, as shownin Fig. 1(c). The blocks are 156 mil wide and 160 mil long


Fig. 2. Quadruple-ridge waveguide cross-sectional view, in the OMT coaxialtransition region, showing the electric fields for (a) the �� mode, (b) the�� mode, and (c) the �� mode. Opposing ridges are connected togetherby shorting blocks, whereas the light and dark ridge pairs are RF isolated.

Fig. 3. HFSS simulated quadruple-ridge�� mode cross-polarized transmis-sion past a single shorting block.

(in the direction of propagation). The orthogonal blocks arespaced 250 mil apart. The x-oriented blocks short-circuit thex-oriented mode electric field lines as can be seen inFig. 1(c). Similarly, the y-oriented blocks reflect the y-oriented

mode wave. The offset positions of the x and y orientedshorting blocks provide impedance matching to the offset or-thogonal coaxial feeds. There is a 1.859 distance between eachcoaxial feed and closest co-polarized shorting block.

For the mode, the signal must first propagate past oneorthogonal x-oriented block before being reflected by the co-po-larized y-oriented block. Ansoft high-frequency structure sim-ulator (HFSS) [13] simulations were first carried out to ensurethat the mode signal was not significantly attenuated. Ascan be seen in Fig. 3, the attenuation of a mode signalpast an orthogonally oriented shorting block is 0.0052 dB orbetter across the band, or better than 0.0104 dB for a reflectedsignal that is attenuated twice by the block. This small attenua-tion value will not significantly affect the OMT performance.

The center conductor and dielectric of each coaxial feedextends across the gap and into a hole in the opposing ridge,acting as an open-circuited transmission line matching stub.The coaxial to quadruple-ridge waveguide transition, shownin Fig. 1(c), has a simulated response with two resonancesas shown in Fig. 4. At the higher frequency resonance in theresponse of Fig. 4, the 0.9 open-circuited stub is close to onequarter wavelength, and the feed couples magnetically to the

Fig. 4. HFSS simulated reflection response of the coax to quadruple-ridgewaveguide transition, square to quadruple-ridge waveguide taper, and the entireOMT response.

TABLE ISQUARE TO QUADRUPLE-RIDGE TAPER

waveguide. The lower frequency resonance is due primarily toelectric field coupling from the coax feed to the waveguide.

The square waveguide to quadruple-ridge taper profileis designed to provide a smooth impedance transition fromthe square waveguide near the OMT input to the chamferedquadruple-ridge waveguide at the coaxial transition. Fig. 4shows the simulated taper performance, which has aresponse. The position of the first null of the taper response(which occurs at 983 MHz as shown in Fig. 4) is determined bythe overall taper length, and is critical to optimizing the low-fre-quency OMT performance. Table I lists the square waveguideto quadruple-ridge waveguide taper profile dimensions, withposition zero at the beginning of the OMT square waveguidesection.

In addition to adjusting the overall taper length, the low fre-quency OMT performance is further enhanced by setting the rel-ative position of the abrupt circular to square transition (shownin Fig. 1(a),(b)). The distance between the circular to squaretransition and the beginning of the quadruple-ridge waveguidetaper was found to be 1.18 through optimization using HFSS.Fig. 5 shows the improvement in OMT performance at the lowerband edge when adding the abrupt circular to square transitionat the input.

Combining the three aforementioned OMT components intoa single HFSS model yields the overall simulated performance,from the circular waveguide input to the coaxial output, shown


Fig. 5. Simulated OMT coaxial probe reflection response with the OMT ter-minated in a square waveguide, and the OMT simulated response after addingthe abrupt circular-to-square transition, with the OMT terminated in a circularwaveguide. The ‘front probe’ and ‘back probe’ refer, respectively, to the x-ori-ented and y-oriented coaxial feeds as shown in Fig. 1(c).

in Fig. 4. The simulated response meets the return loss specifi-cation by a wide margin.

B. Suppression of Trapped-Mode Resonances

The possibility of higher order modes propagating at somepoint along the OMT is unavoidable due to the wide bandwidth.As previously discussed, the fundamental orthogonal modes arethe and modes, with electric field patterns shown,respectively, in Figs. 2(b) and (a). The next two higher ordermodes in quadruple-ridge waveguide are the andmodes [14]–[16].

A scheme for mitigating issues related to mode exci-tation is implemented in the coax to waveguide transition. Asshown in Figs. 1(c) and 2, offset shorting blocks connect op-posing ridges together as previously discussed. The function ofthese blocks is twofold. In addition to providing an offset shortto the orthogonal coaxial feeds, the overlapping geometry of theblocks maintains RF isolation between the x-oriented and y-ori-ented ridge pairs. As can be seen in Fig. 2(c), the modemay propagate past the shorting blocks since the two ridge pairsare at different potentials even though each pair of opposingridges is shorted together. As long as the ridge pairs are iso-lated, the mode will propagate past the blocks due to theelectric field symmetry shown in Fig. 2(c).

The interleaved arrangement of small blocks, as shown inFig. 1(c), was chosen to minimize the difference in electricallength between the two orthogonal signals. The phase differencebetween orthogonal signals is therefore small (of the order of10 degrees at mid-band). To detect circularly polarized signals,coaxial cables that connect the OMT to an external quadraturehybrid must be manually trimmed to compensate for this phasedifference. A small phase difference minimizes the amplitudemismatch since the coaxial cable length difference for each or-thogonal signal is small. The phase-matched coaxial cables typ-ically have a length difference of less than 4 mm.

As shown in Fig. 1(c), absorbing material is placed next tothe shorting blocks, on the opposite side from the coaxial feeds.The absorbing material is isolated from the desired fundamentalmodes since the shorting blocks reflect the andwaves. The higher order mode, however, will propagate

Fig. 6. HFSS Simulated quadruple-ridge�� mode transmission past the se-ries of 8 offset shorting blocks.

Fig. 7. HFSS simulated quadruple-ridge �� /TE mode isolation throughthe shorting blocks, for a single block and for a series of four blocks, aligned inthe same direction as the fundamental mode electric field lines.

past the blocks and through to the absorber as shown in Fig. 1(c).This is used to suppress trapped-mode resonances that wouldotherwise occur due to in-band propagation. Fig. 6 showsthe HFSS simulated transmission for a mode signal prop-agating past the series of 8 blocks as shown in Fig. 1(c). Theinsertion loss is better than 0.029 dB, confirming that themode signal will propagate through to the absorber.

Fig. 7 shows the HFSS simulated isolation through a set ofshorting blocks that are oriented along the same direction as theelectric field lines. For a single block, the fundamental modeisolation varies between 17.7 dB at 1 GHz and 10.5 dB at 2 GHz,averaging 13.5 dB across the band. A single block would thus beinsufficient to adequately isolate the absorber from the desiredfundamental mode signal. Consequently, a series of 4 blocks isused for each polarization. As shown in Fig. 7, the isolation fora series of 4 blocks varies between 51.6 dB at 1 GHz and 42.3dB at 2 GHz, averaging 46.5 dB across the band.

The simulated plots of Fig. 8 illustrate the effectiveness ofthe absorbing material in suppressing trapped-mode reso-nances. When the OMT is simulated with no absorbing materialbehind the offset shorting blocks, two sharp resonances appearat 1.12 GHz and 1.27 GHz in the isolation response betweenthe two coaxial ports, as shown in Fig. 8. When the same OMTstructure is simulated with the absorber in place, Fig. 8 shows


Fig. 8. HFSS simulated isolation between OMT coaxial ports with and withoutthe absorbing material.

Fig. 9. Simulated modal analysis along the L-Band OMT transition showingattenuation constant (nepers per meter) vs. frequency. Each curve represents aquadruple ridge waveguide of constant cross-section corresponding to varyingdimensions along the OMT transition. The solid black line and light dashed linesare �� mode attenuation constants along the OMT transition. The dashedblack line and dotted light lines represent the�� mode attenuation constantsalong the OMT transition.

that the trapped-mode resonances disappear, providing furtherconfirmation of TE mode suppression.

Since no closed-form solution exists for analyzing thequadruple-ridge waveguide modes, a modal analysis along theL-band OMT tapered transition was carried out using AnsoftHFSS. The plots of Fig. 9 show the attenuation constant vs. fre-quency for the higher order and modes along thetapered transition. The frequency at which the attenuation con-stant drops to zero corresponds to the modal cutoff frequency.The lowest mode cutoff frequency occurs at 572 MHz, inthe coaxial to quadruple-ridge transition region, with a 240 milridge gap and 3 outer waveguide dimension. As the ridge gapincreases, the mode cutoff frequency tends to increaseas shown in Fig. 9. It is important to note that the highest

mode cutoff frequency along the transition occurs at thesquare waveguide end, having a 1.27 GHz value. At frequenciesbetween 1 GHz and 1.27 GHz, trapped-mode resonancesare possible since this mode is at or below cutoff somewherealong the transition, and will be reflected back towards theshorting blocks. The aforementioned OMT resonances, shownin Fig. 8, fall within the range of possible trapped-mode res-onance frequencies since the mode is below cutoff atsome point along the OMT. As previously described, the offsetshorting blocks and absorbing material effectively eliminatethe trapped-mode resonances.

The mode does not have a field symmetry that maybe used for trapped-mode resonance suppression [14]–[16].Consequently, it is not only necessary to design the taperedsquare waveguide to quadruple-ridge waveguide transition forimpedance matching, but also to ensure that the modedoes not propagate anywhere along the OMT at any in-bandfrequency.

The mode cutoff frequency, as shown in Fig. 9, ishighest at the coaxial to quadruple-ridge transition region. Thishighest cutoff frequency is 4.73 GHz, which is significantlyhigher than the upper band edge. The mode cutoff fre-quency, however, tends to decrease as both the ridge gap and theouter waveguide dimension increase. The lowest modecutoff frequency of 2.03 GHz occurs along the tapered transi-tion where the ridge gap is 3.45 , with a 6.43 outer waveguidedimension. For the frequency range between 1.27 GHz and 2.03GHz, the OMT does not have any possible trapped-mode reso-nances. As can be seen in Fig. 14, the first trapped-moderesonance occurs at 2.094 GHz according to HFSS simulationsand 2.093 GHz according to measured results.

III. DESIGN FOR MANUFACTURABILITY

The EVLA L-Band OMT design builds on the concepts ofthe S-Band OMT reported in [3], which was designed with anemphasis on manufacturability. The design of [3], however, hasa more complex offset shorting structure compared to the offsetblocks of the new L-Band design. The OMT of [3] requires theinstallation of fourteen beryllium-copper offset shorting pins,held in place by fifty-six set screws. There is one set screw oneither side of each pin, as well as an additional pair of set screwsfor each pin to lock the first pair in place. The assembly, po-sitioning of the beryllium-copper shorting pins and set-screwtightening adds significant time to the assembly process. Fur-thermore, there is a risk of misaligning the pins as well as insuffi-ciently tightening the set screws. These assembly errors may notbe detectable in room-temperature RF measurements. When thedissimilar metals are cryogenically cooled, however, the afore-mentioned assembly errors appear as degradation in receivernoise temperature due to some of the desired modesignals propagating through to the absorber as a result of poorcontact between the shorting pins and ridges. It would requireseveral days to warm the receiver, remove the OMT for rework,re-assemble and re-cool the receiver.

The offset shorting blocks of the L-band OMT, shown inFig. 10 avoid the assembly errors associated with using shortingpins. The four ridges are machined separately. Four shortingblocks are integrated into two of the ridges, and the structuresimply bolts together, held in place by a precision locator blockas shown in Fig. 10. The locator block sets the spacing and main-tains symmetry, which is essential for high isolation betweenorthogonal modes. The outer walls of the waveguide consist offour outer shell sections, that are machined in two pieces each,which bolt together. Once the four-ridge assembly has beenbolted together as shown in Fig. 11(a), the outer shell sectionsare added to complete the structure as shown in Fig. 11(b),(c).

Since the OMT bolts together as a fixed structure, the onlydegree of freedom for tuning is to set the length of the open-cir-cuited coaxial feed. Parametric analyses were carried out to de-


Fig. 10. Photograph of the partially assembled L-Band OMT, with one ridgeremoved, to show the orthogonal coaxial feed holes (left) offset orthogonalshorting blocks (center to right) and precision locator block and absorbing ma-terial (right).

Fig. 11. Fabricated L-Band OMT assembly. (a) Four ridges bolted to the pre-cision locator block. (b) Partly assembled OMT. (c) Fully assembled OMT.

termine the critical dimensions and set nominal values accord-ingly, to ensure that the fabricated devices meet RF performancespecifications provided parts are manufactured to within speci-fied tolerances.

The most critical dimensions of the OMT are in the coax toquadruple-ridge waveguide transition, which has the smallestdimensions of the structure. The simulated plots of Fig. 12 showthe effect of changing the relative distance from the shortingblocks to the coaxial feed by mil from the nominal value.The traces of Fig. 12 show that the most significant changesoccur at the upper and lower band edges, but the reflection at thecoaxial feed is still better than dB at 1 GHz and 2 GHz. Theworst-case reflection value degrades by 0.3 dB at the mid-band,to dB with a coax feed position of mil from thenominal value. Since the manufacturing tolerance is onlymil for the relative feed position, the effect of this variation onOMT performance is minimal.

The simulated tuned OMT reflection plots of Fig. 13 show theeffect of varying the ridge gap between 224 mil and 256 mil, in4 mil increments. The Fig. 13 plots show the maximum rangeover which the L-Band OMT can be tuned. The worst-case re-flection is dB at 1.53 GHz for a 224 mil ridge gap, and

dB at 2.0 GHz for a 256 mil ridge gap. The mid-range240 mil gap is chosen to be the nominal value. The expected gapvariation, based on manufacturing tolerances, is only mil.The OMT meets the impedance match specification by a wide

Fig. 12. HFSS simulated tuned OMT response with varying position of thecoaxial feed relative to the shorting blocks. The nominal position is 1.859 be-tween the coaxial feed and shorting block.

Fig. 13. HFSS simulated tuned OMT front probe response with ridge gapsvarying between 224 mil and 256 mil. These plots show the maximum rangeof gap values over which the L-band OMT can be tuned. The nominal 240 milvalue is chosen at the center of this range.

margin with the nominal dimensions. Fabricated devices thushave a high probability of meeting specifications provided partsare machined to within specified tolerances. Furthermore, thisOMT design has a sufficient RF performance margin to improvethe possibly of meeting specifications in the event of minor man-ufacturing errors.

IV. EXPERIMENTAL RESULTS

The L-Band OMT was measured using an Agilent E8364BPNA Series network analyzer. The waveguide OMT port wasterminated in a sliding load. The measurements shown inFigs. 14–16 show the performance of a fabricated OMT basedon initial assembly. The only required tuning was to set thelength of the open-circuited coaxial feeds. The L-Band OMTcan be tuned in less than one hour for both polarizations.

The plot of Fig. 14 shows the HFSS simulated and measuredL-Band OMT reflection at the two orthogonal coaxial ports. Themeasured OMT reflection is dB (at 2 GHz) or betteracross the entire octave bandwidth, whereas the simulated re-sponse is dB (at 1.2 GHz) or better across the band. Themeasured L-Band OMT response exceeds our dB reflec-tion specification by a wide margin. The plots of Fig. 14 alsoshow the OMT performance beyond the band edges. As previ-ously discussed, low-frequency performance is limited by wave-guide cutoff, and high-frequency performance is limited by the

trapped-mode resonances. The simulated OMT perfor-mance meets the dB reflection specification from 962.5


Fig. 14. HFSS Simulated and Measured L-Band OMT reflection.

Fig. 15. HFSS Simulated and Measured L-Band OMT transmission.

Fig. 16. HFSS Simulated and Measured L-Band OMT isolation.

MHz to 2.075 GHz. The measured OMT performance meetsspecifications from 982.5 MHz to 2.048 GHz.

The plot of Fig. 15 shows the HFSS simulated and measuredOMT insertion loss at room temperature. The measured inser-tion loss results are averaged from two OMT units connectedback-to back. As expected, the measured insertion loss is higherthan the ideal simulated values. The maximum simulated in-sertion loss of 0.093 dB occurs at 1 GHz, and the measuredvalue is 0.20 dB (at 1.47 GHz) or better across the entire octaveband. The average measured room-temperature insertion loss,as shown in Fig. 15, is 0.12 dB for both polarizations, across the1 GHz–2 GHz band.

Fig. 16 shows the HFSS simulated and measured OMTisolation between the orthogonal coaxial ports. The simulatedisolation is 48.8 dB at 2 GHz or better, whereas the measuredisolation is 35.7 dB at 1 GHz, or better across the entire

band, averaging 42.5 dB from 1 GHz–2 GHz. Moreover, thereis no evidence of trapped-mode resonances in the isolationplot of Fig. 16, or the reflection and transmission plots ofFigs. 14 and 15, respectively. As predicted from simulateddata, the fabricated L-Band OMT exhibits good performancein terms of bandwidth, return loss, transmission, isolation andtrapped-mode resonance suppression, exceeding specificationsby a wide margin.

V. CONCLUSION

A new octave-bandwidth OMT has been designed for theEVLA 1 GHz–2 GHz L-Band receiver. Hardware has been fab-ricated and tested, and exceeds specifications by a wide marginin terms of return loss performance and trapped mode reso-nance suppression. The new OMT design focuses on manu-facturability, and is significantly easier to assemble than pre-vious designs. The individually machined parts simply bolt to-gether, and the only required tuning is to adjust the open-cir-cuited coaxial feed length for each polarization. Tuning takesless than one hour for both polarizations. Extensive parametricanalyses were carried out, and nominal dimensions have beenset to ensure the devices exceed the RF specifications providedthe parts are machined to within specified tolerances. With itsexcellent wideband performance and simplified manufacturingprocess, the proposed OMT would be amenable to much largerfuture array projects.

ACKNOWLEDGMENT

The author would like to thank H. Dinwiddie for his work onthe mechanical implementation of the design and P. Madiganfor machining the first prototypes.

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Gordon M. Coutts (S’93–M’99) was born inOttawa, ON, Canada. He received the B.A.Sc. andM.A.Sc. degrees in electrical engineering and thePh.D. degree in electrical and computer engineering(with a specialization in frequency-steerable antennaarrays and RF MEMS on flexible substrates) fromthe University of Waterloo, Waterloo, ON, Canada,in 1998, 1999, and 2007, respectively. His mastersresearch concentrated on nonlinear propagationcharacteristics of high-temperature superconductingplanar transmission lines and applications.

For four years he was with Agilent Technologies, Rohnert Park, CA, andPenang, Malaysia. In 2003, he joined the Centre for Integrated RF Engi-neering, University of Waterloo. In August 2007, he joined the National RadioAstronomy Observatory, Socorro, NM, where he is currently an AssociateScientist/Research Engineer involved with the upgrade of cryogenic receiverfront-ends for the next-generation expanded very large array (EVLA).

Documents

Octave Bandwidth Orthomode Transducers for the EVLA