Novel PWM line conditioner with fast output voltage control

B.-D. Min B.-H. Kwon

Indexing terms: Line conditioner, Pube-width modulution, AC choppers, Commutation

Abstract: A novel pulse-width modulated (PWM) line conditioner with fast output voltage control is proposed. The line conditioner is made up of a PWM AC chopper and a transformer for series voltage compensation. In the PWM AC chopper, a proper switching operation for solving the commutation problem is achieved. Power semiconductor modules which are commercially available can be used in this circuit and regenerative DC snubbers are attached directly to power semiconductor modules to absorb the energy stored in line stray inductance. These regenerative DC snubbers have a very simple structure consisting of a capacitor only with no discharging resistors or complicated regenerative circuit for snubber energy. Therefore the proposed AC chopper gives high efficiency and high reliability. The output voltage of the line conditioner is controlled using rapid sensing of the output voltage. The optimal gains of the controller giving an output with small overshoot and adequate damping are designed using the time-weighted quadratic performance index. It is also shown using simulation and experimental results that the proposed line conditioner gives good dynamic and steady state performances with high quality output voltage.

drop of more than 15% cannot be tolerated for more than 30 cycles (or 500ms), as shown in Table 1. Simi- larly, a 35% voltage drop (can be tolerated for only one cycle (or 16.7ms). Currently, most AC line conditioners still rely on thyristor technology (phase-controlled AC controllers). This is due to the reliability and large power handling capability of thyristor switches. Such conditioners, however, have slow response and need large input-output filters to reduce low-order harmon- ics. The standard AC chopper requires bidirectional switches with their inherent commutation problems. The commutation of switches is critical and an alter- nate current path has to be provided when current paths are changed. This alternate current path is imple- mented using additional bidirectional switches or snub- bers [4-81. Such topologies are difficult and expensive to realise and the voltage: stress of the switch is also high, resulting in reduced reliability.

Table 1: Typical range of input power quality and load parameters of major computer manufacturers

Parameters Range or maximum

Voltage regulation, steady state

Voltage disturbances

Momentary u ndervoltag e

Transient overvoltage

1 Introduction

Solid-state line conditioners have been used in industry to provide regulation and to protect sensitive loads. Power line disturbances on sensitive equipment such as computers, communication equipment, and process control systems can often lead to valuable data loss, interruption to communication services and lengthy shutdowns of production [l]. The recent increase in the use of nonlinear loads has caused serious concern for power quality and consequently on the disturbances that can be tolerated by sensitive electronic loads. The IEEE standard 446-1987 [2, 31 describes the voltage tol- erance limits for sensitive loads, such as computer power supplies. In IEEE standard 446-1987 a voltage

0 IEE, 1998 ZEE Proceed&p online no. 19981 532

Paper first received 11 th November 1996 The authors are with the Department of Electronic and Electrical Engi- neering, Pohang University of Science and Technology, San 31 Hyoja- dong, Pohang 790-784, Kyungbuk, South Korea

IEE ProcElectr. Power Appl., Vol. 145. No. 2, March 1998

Voltage harmonic distortion

Noise

Frequency variation

Frequency rate of change

34, phase voltage unbalance

34, load unbalance

Power factor Load demand

+5, -10 to +IO%,-15% (ANSI C84.1-1970 is +6, -13%)

-25 to -30% for less than 0.5s, with -100% acceptable for 4 2 0 m s

+I50 to 200% for less than 0.2ms 3-5% (with linear load)

No standard

60Hz k 0.5Hz to k 1 Hz

1 Hz/s (slew rate)

2.55%

5-20% maximum for any one phase

0.8-0.9

0.75-0.85 (of connected load)

In this paper, a novel PWM line conditioner with fast output voltage control is proposed. The line condi- tioner is made up of a PVVM AC chopper and a trans- former for series voltage compensation. Since the power circuit of the AC chopper uses only two stand- ard switch modules and DC regenerative snubbers without discharging resistors, and a proper switching operation for solving the commutation problem is achieved, the line conditioners are reliable and applica-

8 5

line stray inductance r - h

L

t

I I I b

Fig. 1 Power circuit ofthe singleplzasse buck AC clzoppers a Pnoposed AC chopper using power semiconductor modules and regenerative DC snubbers h Conventional AC chopper using bidirectional switches and AC snubbers

2 Description and analysis of the AC chopper

The power circuit of the PWM AC chopper using insu- lated gate bipolar transistors (IGBTs) is shown in Fig. la. Power semiconductor modules which are com- mercially available can be used in this circuit and regenerative DC snubbers are attached directly to power semiconductor modules to absorb the energy stored in line stray inductance. These DC snubbers fea- ture a very simple structure consisting of a capacitor only with no need for discharging resistors or for a complicated regenerative circuit for snubber energy. This is because the snubber energy is regenerated dur- ing the active mode. The conventional AC chopper in Fig. 16 uses bidirectional switches capable of bidirec- tional current control. Therefore, the snubber circuit of the bidirectional switch must be an AC snubber circuit that can absorb bidirectional turn off spike energy due to line stray inductance. Further, since each bidirec- tional switch is turned on or off at high frequency, a circuit for regenerating the snubber energy needs to be provided to enhance the conversion efficiency of the equipment. However, the proposed AC chopper regen- erates snubber energy without using any additional regenerative circuit. In the AC chopper, the inductor current iL conducts through the input and output side, providing energy to the output during the active mode, and freewheels through the output side during the free- wheeling mode. However, the inductor current is bypassed through the input side or output side depend-

86

ing on its direction during the bypass (dead-time) mode. Fig. 2 shows the three possible modes during one switching cycle. The bold lines denote possible cur- rent passes

CiJ - - L i - -1 53 i T T I 1 I I

U

b

RL

RL

C i

C

Fig.2 a Active mode b Freewheeling mode L Bypass mode for v, > 0

Operation modes of the single-phase buck AC chopper

Active mode During this mode, the switches SI and S2 are turned on and the inductor current i, conducts through SI and the diode across S2 for iL > 0 or Sz and the diode across SI for iL < 0, as shown in Fig. 2a. As a result, the input voltage is connected to the output, providing energy. Freewheeling mode This mode is complementary to the active mode. Dur- ing this mode, the switches S3 and S4 are turned on so that the inductor current freewheels through the output side. The current passes of this mode are shown in Fig. 26. Bypass mode To avoid current spikes arising from use of nonideal switches, the dead time t d is required and at the same time a current path through the inductive load to avoid voltage spikes must be established. Thus the bypass mode is necessary. Two additional switches are turned on for safe commutation and they are used in this mode. When the voltage v , ~ is positive, the switches S2 and S4 are turned on for safe commutation. If the

IEE ProcElectr. Power Appl., Vol. 145, No 2, March 1998

inductor current iL is positive, the inductor current is bypassed through the output side using S4 and the diode across S3. If the inductor current iL is negative, the inductor current is bypassed through the input side using S2 and the diode across S1. When the voltage v, is negative, the switches S1 and S3 are turned on for safe commutation. Thus, a current path for the inductor current always exists in every current direction during the bypass mode. The current passes in this mode for v, > 0 are shown in Fig. 2c.

It is assumed that the following conditions are satis- fied:

1 W L << RL << -

w c

where w = 2nfand w, = 2zf, are the angular frequency of the source voltage and angular switching frequency, respectively. These inequalities are satisfied in the prac- tical filter design. These conditions ensure that the fun- damental component is transferred to the output load RL while harmonic components are not. The input voltage is defined as follows:

U&) = JZV, cos(wt) ( 3 ) where V, is the RMS voltage of the source voltage. The switching function is given by

00 2 sin(kD7r) cos(kw,t) (4)

k=l

where D is the duty ratio. Using eqns. 3 and 4, the PWM voltage v,(t) is given by

U T @ ) = S(LJs+Js(t) 00

= &DVs C O S ( W ~ ) + AV,, COS[ (~W, f w ) t ] k=l

( 5 ) where Vrk is the RMS harmonic component of the PWM voltage as follows:

V, sin(kD7r) k7r V T k =

The output filter reduces the harmonics of the inductor current and capacitor output voltage. Their total har- monic distortion factors (THDs) are defined as follows:

where V,,cf and VrCk are the magnitude of the funda- mental and harmonic components of the output volt- age v,, of the AC chopper, and ILf and ILk are the magnitude of the fundamental and harmonic compo- nents of the inductor current iL, respectively. From the equivalent circuit of the AC chopper shown in Fig. 3, the transfer function of the output voltage V,,(s) with respect to the PWM voltage V,(s) is given by

I,

-LA Fig.3 Equivalent circuit ofthe buck AC chopper

The fundamental component Vr,. of the output voltage is approximated using the assumption (eqn. 1) as fol- lows:

RLDVS mt2 + (1 - w2LCi2R; V c f = -

M DV, (9) Using the assumption (eqn. 2) and kw, f 0 = kws, the harmonic component VrCk of the output voltage iS approximated as

From eqns. 9 and 10, THD, is given by

where

Therefore the total harmonic distortion factor of the output voltage is independent of the load resistor RL. However, the inductor and capacitor have a great effect on THD, The transfer function of the inductor current IL(s) with respect to the PWM voltage V,(s) is given by

(13) I L ( S ) - 1 + SCRL

V,(S) si5 + (1 + S2LC)RL Similarly, the fundamental component ILf and har- monic components ILk of the inductor current are approximated as follows:

I L ~ M - kw, L

From eqns. 14 and 15, THDI is given by

where

(17) O0 sin2(kD.ir) TH2=Adz k4

Thus the total harmonic distortion factor of the induc- tor current is not dependent on the output capacitor C. The filter parameters can be designed within THD val-

87 IEE Proc.-Electr. Power Appl., Vol. 145, No. 2, March 1998

ues required in the system. From eqns. 12 and 17, the inductor and capacitor are designed as follows:

(18)

(19)

1002/ZRr, TH2 L = TW* T H D I

10Oa TH1 c=-- TWZL T H D v

Numerical calculations of TH, and TH, with respect to the duty ratio D to design the inductor and capacitor are shown in Fig. 4.

d u t y ra t io D Fig.4 TH, and TH, with respect to the duty ratio D

1 I

a

1 - I b

Fig.5 tioner U Power circuit h Equivalent circuit

Power circuit and equivalent circuit of the proposed line condi-

3

The proposed line conditioner using an AC chopper is shown in Fig. 5a. The output voltage is controlled through the series connected transformer to the line source. From the equivalent circuit of the proposed line conditioner as shown in Fig. 5b, the differential equa- tions of this system are obtained as follows:

Analysis of the line conditioner

88

where vo(t) is the output voltage, vC(t) is the series com- pensation voltage, u(t) is the control input voltage of the AC chopper, and n (= &/NI) is the turns ratio of the series transformer. The voltage and current under steady state are denoted by subscript s and disturbance value by subscript d as follows:

i L ( t ) = ILS + i L d ( t )

us@) = v s s + u s d ( t )

210 ( t ) = vo, + Vod ( t ) vc( t ) = Ks + & d ( t )

u ( t ) = U, + U d ( t ) (22) From eqns. 20-22 the disturbance model of the pro- posed line conditioner is obtained as follows:

When a step disturbance input v,d is applied to the sys- tem, the output voltage vod needs to be regulated to zero with no steady state error and small overshoot under the limited input. As a controller that satisfies such performance requirements and is easily imple- mented, the following proportional-integral (PI) con- troller is used:

U d ( t ) = kP2/,d(t) + ki J' % d ( t ) d t (25 )

where kp and ki are the constant gains to be deter- mined. This controller provides no steady state error for the step response because of the integral action of the output error. To determine the optimal PI gains, the following time-weighted performance index is adopted:

N

J = 2 1- (t) v , " ~ + yu2dt (26)

where tr is used to give relatively larger weighting on a sustained output error after a given time t,, and yis the input weighting factor. The time-weighted quadratic perfomance index provides an increasingly heavy pen- alty for sustained error and this gives better transient performance characteristics in the time domain com- pared to the conventional quadratic performance index (N = 0) [9-111. The time-weighted quadratic perform- ance index can be expected to give a response having small overshoot and adequate damping. The control problem is to determine the controller (eqn. 25), which minimises the time-weighted quadratic performance index (eqn. 26). In a practical determination of the PI gains, the time-weighting factor N is adjusted to give

IEE Proc.-Electr. Power Appl., Vol. 145, No 2, Murch 1998

Table 2: Optimal PI gains, settling time, maximum overshoot, and maximum undershoot

~

Time-weighting Optimal Gains 5% settling Maximum Maximum factor N kP k, time (ms) overshoot (%) undershoot (%)

0

4

9

0.795 0.16 1.42 11 15

0.450 0.14 1.26 10 7

0.281 0.12 0.85 6 3

an adequately damped response and y selected not to exceed the maximum input voltage. Based on this prin- ciple, a better output response of the system can be obtained using the following procedure: (i) The input weighting factor y is selected to be as small as possible, so as not to exceed the maximum input voltage with N = 0. This is conventional optimal control which gives a response with a smaller rise time while the input does not exceed its maximum limit. (ii) Select the appropriate time-weighting scale t,. As a rule of thumb, t, is chosen to be between the rise time and the time which gives the maximum overshoot. (iii) Obtain an adequately damped response increasing the time-weighting factor N. A general guideline is that an increase of the time-weighting factor N gives a more damped response.

The optimal PI gains derived from a computer simu- lation with t , = 0.3ms and y = 1 are shown in Table 2. The simulated step responses of the compensation volt- age v, for the weighting factor N are shown in Fig. 6. The output response for N = 9 is superior to the others for the same settling time. The following parameters are used in the simulation: source disturbance voltage v,d = -2OV, inductance L = 1.5mH, capacitance C = lOpF, load resistance R, = 10Q.

=-- U " 5

25r

10

0.0005 0.001 0 0.0015 0.0020 time,s

Fig.6 weighting factor N

Step responses of the Compensation voltage according to the time-

reduced, the more the discharging speed is increased, but the waveform of the sensed signal has greater rip- ple component. A fast peak voltage detector is pro- posed to operate on a sinusoidal input signal. This detector is made up of a phase shifter, two multipliers, and an adder as shown in Fig. 76. The proposed detec- tor uses the following simple trigonometric principle:

(27) sin 2 ut -I- cos2 w t = 1

The phase shifter delays its input waveform by 90°, i.e. a cosine input waveform is converted to a sine wave- form, and each waveform is multiplied and added.

Cl

analogue multiplier

phase shifter

multiplier

L-r!FJ b

Fig. 7 Peak voltage detectors a Conventional peak voltage detector h Proposed peak voltage detector

O u t P u t r L

4 Experimental results

For fast output voltage control of the line conditioner, a fast output voltage sensing technique is required. In general, the conventional peak voltage detector with diodes, capacitor, and resistor is used as a voltage sens- ing circuit as shown in Fig. 7a. When the input signal is decreased, the capacitor is discharged through the resis- tor, and when it is increased, the capacitor is charged directly. Therefore the charging speed is faster than the discharging speed. The more the resistor value is

IEE Proc.-Electr. Power Appl., Vol. 145. No. 2, March I998

time Fi .8 J a g e detector Upper trace: output response (ZV/div) Lower trace: input waveform (100V/div)

Experimental input waveform and output response of the peak

89

Then the sensed output v,, of the proposed detector is the magnitude of the output voltage as follows:

& = c;~(v: sin2 ut + V: cos2 ut)

where k, and V, are the sensing gain of the output volt- age and the peak value of the output voltage, respec- tively. The experimental waveforms of input and output signals of the output voltage detector for 20% line voltage disturbance are shown in Fig. 8. This peak voltage detector gives a rapid response to the voltage sag due to line voltage disturbance, To show the valid- ity of the proposed line conditioner, the PWM AC chopper shown in Fig. 1 was implemented with the fol- lowing parameters:

= k,2V2 (28)

v, = 110v C, = 10pF L = 1.5mH C = 10pF f = 60Hz RL = 5 0 R fs 15.6 kHz t d = 2 ps n = 0.3

The operation of the proposed AC chopper is shown in Figs. 9 and 10. Figs. 9u-d show the waveforms of the input source voltage, drain-source voltage, intermedi- ate voltage, and inductor current of the AC chopper.

1 I I a

vs 1 2 k b

Vi-

C

time d

Fig. 9 a v8, IOOV/div b Drain-source voltage of S,, 100V/div c v,, IOOVidiv d i ~ , 2Aidiv

PVuv~$onrrs of the input and output vmiubles

The gate signals of the switches are shown in Figs. 1 0u-d. The fast output voltage correction characteristic of the proposed line conditioner is shown in Figs. 11 and 12. The step responses of the compensation voltage v, are shown in Fig. 1 I at the optimal gains obtained in Section 3. The overshoot and settling time of the com- pensation voltage are shortest for the optimal gain N = 9. Although the 20% sudden voltage drop in input source voltage occurs for 3.5 periods of the source volt- age, the output voltage is very rapidly corrected in the proposed line conditioner as shown in Fig. 12.

90

I c 1

's5 1

t

c a

I + I

b

"ss3 , I J

time d

Fig. 10 Scale SVidiv

Gate-source voltage of ihe switches

a

t b

tlme C

Fig. 11 Ex erimentul step responses of the compensation voItage according to t le optimal gains Scale lOV/div a N = O b N = 4 c N = 9

5 Concluding remarks

In this paper, a novel PWM line conditioner with fast output voltage control is proposed. The line condi- tioner is made up of a PWM AC chopper and a trans- former for series voltage compensation. Since the proposed AC chopper uses only two standard switch- ing modules, a proper switching technique for solving the commutation problem, and has regenerative DC snubbers without discharging resistors or complicated regenerative circuit for snubber energy, the line condi- tioners are reliable and applicable to high power. The

IEE Proc.-Electr. Power Appl., Vol. 145, No. 2, March 1998

I also shown via simulation and exDerimenta1 results that

“S

0

“0

0

time Fig. 12 posed line conditioner Upper trace: input voltage, 100V/div Lower trace: output voltage, lOOV/div

Experimentul input and output voltage wuvejorms oj the pro-

the line conditioner gives goodA dynamic and steady state performances.

6 References

1 BHAVARAJU, V.B., and ENJETI, P.: ‘A fast active power filter to correct line voltage sags’, iEEE Trans. Ind. Electron., 1994, 41,

CLARK, J.K.: ‘AC power conditioner: design and applications’ (Academic, New York, 1990) The IEEE Orange Book, IEEE Standard 446-1987: ‘IEEE recom- mended practice for emergency and standby power systems for industrial and commercial applications’. IEEE publication, 1987

4 VINCENTI, D., JIN, H., and ZIOGAS, P.D.: ‘Design and implementation of a 25-kVA three-phase PWM AC line condi- tioner’, IEEE Trans. Power Electron., 1994, pp. 384-389

5 SALAZAR, L., VASQUEZ, C., and WIECHMANN, E.: ‘On the characteristics of a PWM AC controller using four switches’. IEEE PESC Record, PESC’93, 1993, pp. 307-313

6 JOOS, G., and ZIOGAS, P.D.: ‘A PWM AC controller-based high current power supply’. Proceedings of IECON’91, IEEE, 1991, pp. 554-559

7 HAMED, S.A.: ‘Steady-state modeling, analysis, and perform- ance of transistor-controlled ac power conditioning systems’, iEEE Truns. Power Electron., 1990, pp. 305-313 ADDOWEESH. K.E.. and MOHAMADEIN. A.L.: ‘Microuroc-

pp. 333-338 2

3

8 essor based harmonic ’elimination in chopper type ac voltage^ reg- ulators’, IEEE Trans. Power Electron., 1990, pp. 191-200

9 KWON, B.H., YOUN, MJ . , and BIEN, Z.: ‘Optimal constant feedback with time-multiplied performance index for discrete-time linear system’, IEEE Truns., 1985, AC-30, pp. 497499

10 KWON, B.H., and YOUN, M.J.: ‘Optimal observers using time- weighted performance index with prespecified eigenvalues’, Truns. ASME, J. Dyn. Syst. Meas. Control, 1986, 108, pp. 366-368

11 KWON, B.H.: ‘Design of a highly stable electromagnet power supply’, IEEE Trans. Ind. Electron., 1992, 39, pp. 149-158

line conditioner is controlled in the equal PWM pattern which is efficient and simple to implement. A fast out- put voltage detector is also proposed to ensure that the line conditioner gives very fast Output voltage Correc- tion. The optimal gains that give a response with small overshoot and adequate damping are obtained using the time-weighted quadratic performance index. It is

IEE Proc.-Elecfr. Power Appl.. Vol. 145, No. 2, March 1998 91