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I AD-A255 438 CII-
Millimeter-Wave Quasi-Optical
and Spatial Power Combining Techniques
Final Report
* by
Kai Chang DTICAugust 10, 1992 o 4 1992
U. S. Army Research Office S DContract, No.: DAALO3-89-K-0085
IDepartment of Electrical Engineering
ITexas A & M University
College Station, Texas 77843-3128
III
APPROVED FOR PUBLIC RELEASE;
* DISTRIBUTION UNLIMITED
II 92-24392
I
Millimeter-Wave Quasi-Optical
and Spatial Power Combining Techniques
Final Report
by
Kai Chang
August 10, 1992
U. S. Army Research Office
Contract No.: DAAL03-89-K-0085
Department of Electrical Engineering
Texas A & M University
College Station, Texas 77843-3128IAccesion For
NTIS CRA&ID1IC TABU ';ou ,'-,M'dJ.5i'!hCdtl(:, .. .. .. . .
B y . ... . . . .
Di..: ib . ,
APPROVED FOR PUBLIC RELEASE; z
I DISTRIBUTION UNLIMITED
REPOT DCUM NTATON AGEForm Approv.ed
REPORT~~~ DOU ETTINPGOMB N~o 0704-0188"'.. 'eaor-ng ousoen ' !sCO tO nf ormton 'Ssi.'ated TO i.erage' 10W 1~ "1 '~ lesTTls PC tt* 'or Tev.e-"ng *Tstr,ons. sear.Tnq e.%ng cata o'-,.the,-Mq AMC -mrnannq he C4At needeC, And C~Oe1IMq AMd r"ew-9.n !the OIeCTiOfl Of -foIO n .Cfld cOrn l f e,'9~~gt'1b~dnetraIo n t~raetO *
cOIfttn f no~nafOn. nC1~.nqi y.uqeltionj 'or FeduCinq thq OoreM to VVatt,,nqrOM H#4Caaa rte', se-~ces, cirectorate '''nfolhat~oh Ooe,fit.Ons and Reoorti. 11t5 ~lert'on,ae I~f.y ~. O ,AIMq'0n, 'A 22 2_. 40 an "o teO'ffC 4,0qn~n and ., "MCt P O eC Ad On 'ieet (07G4.O 198), Washnqto. OC 20S0331. AGENCY USE ONLY (Leave blank) 2. REPORT DATE 3.REPORT TYPE AND DATES COVERED
IAugust_10,_1992 ?_9_-_30_e________I.-
4. TITLE AND SUBTITLE 5. FUNDING NU"ERSMillimeter-Wave Quasi-Optical and Spatial Power Combining
Techn iques
6. AUTHOR(S) ~-9A -OS
Kai Chang
7. PERFORMING ORGANIZATION NAME(S) AND ADDRESS(ES) 8. PERFORMING ORGANIZATIONUDepartment of Electrical EngineeringREOTNMRTexas A&M University3 College Station, Texas 77843-3128
9. SPONSORING/ MONITORING AGENCY NAME(S) AND ADDRESS(ES) 10. SPONSORING/ MONITORING
IU. S. Army Research Office AEC EOTNMEP. 0. Box 12211 tResearch Triangle Park, NC 27709-2211
11. SUPPLEMENTARY NOTESThe view, opinions and/or findings contained in this report are those of theauthor(s) and should not be construed as an official Department of the Armyposition, policy, or decision, unless so designated by other documentation.
12&. DISTRIBUTION / AVAILABILITY STATEMENT 12b. DISTRIBUTION CODE
Approved for public release; distribution unlimited.
13. A TRACT (Ma.,mumn 200 words)
\.,/ This report summarizes the research activities carried out in Electromagnetics andU Microwave Laboratory, Department of Electrical Engineering, Texas A&M University.
The project was sponsored by the U.S. Army Research Office under contract No. DAALO3-
89-K-0085. The topics of investigation included active antenna elements, spatialpower combining and injection-locking, quasi-optical components, novel planar slot-line and coplanar waveguide circuits, and other circuit developments and analyses.
A list of publications and a report of invention are included.
14. SUBJECT TERMS 15. NUMBER OF PAGES
Quasi-Optical Techniques, Microwaves, Millimeter-Waves, Power 65
Combining, Active Antennas 1.PIECD
17. SECURITY CLASSIFICATION 18. SECURITY CLASSIFICATION 19. SECURITY CLASSIFICATION 20. LIMITATION OF ABSTRACTOF REPORT j OF ABSTRACT
UNCLASSIFIED UNCLASSIFIED I UNCLASSIFIED UL
NSN 7540-01-280-5500 Stantdard Form 298 (Rev 2-39)
8e 'Ci Deld ton ANSI Std 139-18
Millimeter-Wave Quasi-Optical
and Spatial Power Combining Techniques
Final Report
by
Kai Chang
August 10, 1992
U. S. Army Research Office
Contract No.: DAAL03-89-K-0085
Department of Electrical Engineering
Texas A & M University
College Station, Texas 77843-3128
APPROVED FOR PUBLIC RELEASE;
DISTRIBUTION UNLIMITED
I
THIIWOIINADO IDNSCNANDI HSRPRARIHS FTEATO()AD HUDNTB OSRE SAOFIILDPRMNIFTE RYPSTOPLCODEIININESS EINTE YOHRDCMNAIN
I
FOREWORD
This report summarizes the research activities carried out in the Electromagnetics and
Microwave Laboratory, Department of Electrical Engineering, Texas A & M University.
The project was sponsored by the U.S. Army Research Office under contract No. DAAL03-
89-K-0085. The period of performance was from April 1, 1989 through June 30, 1992. The
topics of investigation included active antenna elements, spatial power combining and
injection-locking, quasi-optical components, novel planar slotline and coplanar waveguide
circuits, and other circuit developments and analyses.
Table of Contents
Page
1. Introduction and Problem s Studied ............................................... 1
2. Active A ntenna Elem ents ......................................................... 4
2.1 Active Patch Antenna Elements ............................................. 4
2.2 Active Notch Antennas ..................................................... 4
2.3 Active Annular Ring Microstrip Antennas ................................... 7
2.4 Active Image - Line Antennas ............................................... 7
3. Spatial Power Combining and Injection-Locking ................................... 7
3.1 Combiners Using Active Patch Antenna Elements ............................ 7
3.2 Combiners Using Active Notch Antenna Elements ............................ 9
4. Quasi - Optical Com ponents ...................................................... 9
4.1 35 GHz Rectifying Antennas ................................................ 9
4.2 Frequency Selective Surface Filters ......................................... 11
4.3 Low - Loss Open Resonator Coupling and Filters .......................... 13
5. Novel Planar Slotline and Coplanar Waveguide Circuits .......................... 13
5.1 Coplanar Waveguide Tunable and Switchablie Filters ....................... 15
5.2 Coplanar W aveguide Gunn VCO ........................................... 15
5.3 Coplanar Waveguide Fed Slotline Ring Resonators ......................... 17
5.4 Conductor - Backed Coplanar Waveguide .................................. 17III
6. O ther Circuits and Analyses ..................................................... 19
6.1 Wide Inductive and Capacitive Strip Discontinuities ........................ 19
6.2 Superconductive Strip Discontinuities ....................................... 19
6.3 Microwave/Millimeter-Wave Impedance Measuring Scheme Using a Three-Probe M icrostrip Circuit .................................................. 21
R eferences .......................................................................... 23
List of P ersonnel ................................................................... 25
R eport of Invention ................................................................ 26
List of Publications Under ARO Support ........................................... 27
A ppendices ........................................................................ 31
Appendix 1
K. Chang, K.A. Hummer and J.L. Klein, "Experiments on Injection-Locking ofActive Antenna Elements for Active Phased Arrays and Spatial Power Combin-
ers," IEEE Trans. on Microwave Theory and Techniques, Vol. 37, July 1989, pp.1078-1084.
Appendix 2
J.A. Navarro, Y.H. Shu and K. Chang, "Broadband Electronically Tunable PlanarActive Radiating Elements and Spatial Power Combiners Using Notch Antennas,"IEEE Trans. on Microwave Theory and Techniques, Vol. 40, Feb. 1992, pp. 323-328.
Appendix 3
T. Yoo and K. Chang, "Theoretical and Experimental Development of 10 and 35
GHz Rectennas," IEEE Trans. on Microwave Theory and Techniques, Vol. 40,June 1992, pp. 1259-1266.
Appendex 4
Y.H. Shu, J.A. Navarro and K. Chang, "Electronically Switchable and TunableCoplanar Waveguide-Slotline Bandpass Filters," IEEE Trans. on Microwave The-ory and Techniques, Vol. 39, Mar. 1991, pp. 548-554.
ii
1. Introduction and Problems Studied
For the past decade, microwave/millimeter-wave activities have been accelerating for
both civil and military applications including telecommunications, radar, sensors, seekers,
navigation, remote sensing, electronic warfare and space technology. These applications
have created the urgent need for low cost, high performance solid-state integrated systems
using planar integrated circuit technology. These integrated systems normally consist of
high power transmitters, low noise receivers, antennas, and control devices. To lower the
losses and cost, quasi-optical technique is attractive for millimeter-wave frequencies. In
many applications, frequency agility is required, and electronic tunability of the system is
desirable. The output power from a single active solid-state device is limited by funda-
mental thermal and impedance problems. To meet the high power requirements for many
applications, it is necessary to combine many devices to achieve high power output.
Combining power from many active devices is a difficult task at millimeter-wave fre-
quencies due to the moding and size problems [1]. The maximum number of devices that
can be combined in a conventional resonant or hybrid-coupled power combiner decreases
3 as the operating frequency is increased. To overcome these problems, spatial and quasi-
optical power combining techniques have been proposed to combine power in free space at
high frequencies [2]. In these combiners, the active device is integrated with a radiating
element to form an active element. Many of these active elements are then combined in
3_! free space or in an open resonator. For efficient power combining, all source elements must
be coherent in phase and frequency. Injection-locking through open resonator, space or
-- mutual coupling can be used to achieve the coherency.
In addition to application in active antenna elements and power combiners, quasi-
optical techniques can be used for filters, receivers, detector or imaging arrays, power
conversitjn arrays, etc. The techniques are especially attractive for millimeter-wave fre-
quencies due to the advantages of low cost, large size and good performance. Several
3 quasi-optical components were developed with excellence performance under this project.
Along with the research of the quasi-optical components and spatial power combining
3techniques, work was also carried out on the novel planar slotline and coplanar waveguide
U!I
circuit developments. The slotline and coplanar waveguide have the center conductor and
ground planes on the same side of the substrate. They are truly uniplanar microwave
integrated circuits that have advantage of easy implementation of solid-state devices.
Although most of our experimental work was concentrated at microwave frequencies,
the circuits and analyses developed can be applied to millimeter-waves. The problems
studied and major accomplishments made are summarized below:
1. A single active radiating element with a Gunn diode mounted directly on the mi-5crostrip path antenna has been developed with over 9 percent electronic tuning range.
2. A spatial power combiner has been demonstrated in free space with over 90 percent5 combining efficiency using two directly integrated Gunn-patch antenna elements.
3. A spatial power combiner has been demonstrated using two aperture coupled mi-
3 crostrip antenna elements.
4. Injection-locking has been achieved for spatial power combiners through space and
mutual coupling.
5. An active radiating element using an FET source integrated with a microstrip patch
antenna has been developed.
6. Active planar notch antenna elements were invented with a wide electronic tuning
bandwidth. A power output of 14.5 ± 0.8 dBm over a tuning range from 8.9 to 10.2
GHz was achieved. This is equivalent to 14 percent of electronic tuning bandwidth.
The circuit is small and can be fabricated at low cost.
7. Power combining of two varactor-tuned active notch antenna elements has achieved
over 70% combining efficiency through the 14% tuning range. This is the first varactor
tunable power combiner ever reported.
8. An integrated active antenna using annular ring microstrip antenna and Gunn diode
was demonstrated with over 70 mW output power at 7 GHz.
9. An integrated image-line steerable active antenna was demonstrated with a 2 percent
electronic tuning range corresponding to a frequency steerable range of 4 degrees.
10. A quasi-optical rectifying antenna was developed at 35 GHz with over 39 percent RF
to DC conversion efficiency. The device was designed as a large-signal detector.
3 2
11. A quasi-optical filter was used to reduce the second harmonic radiation of the rectifying
antenna.
12. A low loss quasi-optical open resonator filter was devised using a Fabry-Perot resonator
and a narrow slot opening in the coupling waveguide. The filter has an insertion
loss of less than 1 dB and should have applications in many millimeter-wave and
submillimeter-wave systems.
13. Electronically switchable and tunable coplanar waveguide-slotline bandpass filters
were invented with over 20% electronic tuning range. These active filters should
have many applications in radar, communications and EW systems.
14. A novel varactor tunable coplanar waveguide Gunn VCO was invented with 300 MHz
tuning range at 10 GHz. The circuit is small, lightweight and offers low cost, good
reproducibility and excellent performance.
15. A coplanar waveguide fed slotline ring resonator was invented using varactor diodes
to create an electronically tunable planar resonator and filter. The resonator has over
23% tuning range and 4.5 + 1.5 dB of insertion loss.
16. A multiple dielectric structure was studied for conductor-backed coplanar waveguide
to suppress the leakage effects.
17. A simple microstrip to dielectric waveguide transistion has been developed with an
insertion loss of less than 1 dB.
18. An optimum dielectric overlay thickness was found for achieving the equal even- and
odd- mode phase velocities in coupled microstrip lines. A theoretical analysis based
on variational method was developed.
19. An analysis was developed to calculate the susceptance due to a wide inductive strip
and a wide resonant strip. The theory was based on a variational method and moment
method.
20. A novel resistance measurement technique was devised to characterize high-temperature
superconducting materials at microwave frequencies. The method uses a floating res-
onant strip and the measurement agree very well with the theoretical prediction. The
technique can be used for both bulk and thin-film materials.
21. A simple low cost microwave/millimeter-wave impedance measuring technique using
3
pL
a three-probe microstrip circuit was developed. The method provides good accuracy and
ease of calibration.
12. Active Antenna Elements
5 In an active antenna, an active solid-state device is directly integrated with a planar
antenna. The circuit can be used as a low cost, small transmitter. Many of these elements
3 can be aligned to form a spatial or quasi-optical power combiner. Under this project, we
have dcveloped active antenna elements using path, notch, annular ring, and image-line
antennas. The solid-state devices used were Gunn devices and FETs.
2.1 Active Patch Antenna Elements
A Gunn diode was integrated with a patch antenna as shown in Figure 1. The place-
5mnent D of the active device was chosen as such that the device impedance was matched
to the input impedance of the patch. The output power was calculated using Friis trans-
I mission equation.
At X-band, a 3 dB tuning range of 839 MHz was achieved from 9.278 to 10.117 GHz
3 using bias tuning. This tuning range is equivalent to a 9 percent bandwidth, which is much
wider than that of a single passive patch. The results are summarized in the literature [3]
( Sce Appendix 1).
Integration of an FET device with a patch a ntenna was also demonstrated earlier
[4]. The circuit is shown in Figure 2. The circuit consists of a patch antenna, aml FET,
and matching and bias circuits. The oscillator is formed by placing a frequency selective
network, such as a resonator which is a )atch antenna, in the feedback loop of an amplifier.
If the feedback is positive at the operating frequency, oscillation occurs. The patch acts as
both a resonator and radiator.
2.2 Active Notch Antennas
The use of a notch antenna has many advantages of )road bandwidth, beamwidth
design flexibility, relative high gain, and easy active device integration. Figure 3 shows
the varactor-tunable active notch antenna integrated with a Gunn diode [5). The circuit
exhibits a tuning bandwidth from 8.9 to 10.2 GHz with an output power of 14.5 + 0.8
4
!L
I
!W
IGunn Diode
Il DC Bias Line
Figure 1. An active patch antenna integrated with a Gunn device
1 5
PATCHANTENNA
IT
3--f \E
500l
I_ TUNING
STUBS
14 DC BIAS
Figure 2. An active patch antenra integrated with an FET deviceiVARACTOR
I I
CPW RESONATOR- -- GUNN
Figure 3. An active notch antenna integrated with a Gunn VCO
6
dBm. This is equivalent to ovor 14% tuning bandwidth. The results are given in the
literature [51 (See Appendix 2).
2.3 Active Annular Ring Microstrip Antennas
The use of an annula" ring antenna integrated with a Gunn device was investigated
[6]. The circuit is shown in Figure 4. A computer program based on theoretical analysis
was used to determined the placement location of the Gunn diode for optimum impedance
matching. A passive ring was first built and tested to verify the design procedure. Injection-
locking the active antenna was achieved using an external source.
2.4 Active Image-Line Antennas
At high frequencies (above 100 GHz), conductor losses become notable and microstrip
circuits become so small that using microstrip is difficult. To overcome the radiative and
conductor losses, an active image-line grating antenna was proposed. A Gunn diode was
placed in an image-line grating antenna with a resonant cap as shown in Figure 5 [7]. A
varactor was placed between the Gunn and the shorted end of the oscillator, as close as
possible to the Gunn. The varactor gave 1% tuning bandwidth which corresponds to a 4
degrees beam scanning.
3. Spatial Power Combining and Injection-Locking
The spatial power combiner utilizes the correct phase relationship of many radiating
elements to combine power in free space. For efficient combining, all the source elements
nmust be coherent in phase and frequency. Injection-locking technique can be used for
frequency alignment. Feasibility of injection-locking and power combining of using a two-
I element array has been established.
I 3.1 Combiners Using Active Patch Antenna Elements
Figure 6 shows a two-element power combiner. The two active devices in the two-
element array injection-locked each other through mutual coupling. Injection-locking could
also be achieved using an external source. A combining efficiency of over 90 percent has
beer successfully demonstrated [1] (See Appendix 1).
7
Bias line
Figure 4. An active annular ring antenna integrated with a Gunn device
I
Top plae
4,,,-, 25 r
abort +Biu lead r .-- n ' .OlMM Aborber
/ Ground plans
I Figure 5. An active image-line grating antenna integrated with a Gunn VCO
I
II
I
I3.2 Combiners Using Active Notch Antenna Elements
To demonstrate the feasibility of broadband, tunable power combiners, two active
notch antennas were set up in a broadside array at A/4 separation. The active notch
antenna elements were injection-locked to each other through nmtual coupling. Power
combining experiments of two injection-locked, varactor-tuned active notch antennas were
conducted throughout the electronic tuning range at 100 MHz increment over a 1.3 GHz
bandwidth at X-band. All power calculations were based on the Friis Transmission Equa-
tion. The increase in, gain of the notch and the array beam sharpening has been included
in the calculation. Over 70% combining efficiency was observed throughout the varactor
I tuning range. These results represent the first varactor-tunable power combiner over a
wide tuning range reported in literature [5] (See Appendix 2).
A four-element and a sixteen-element array using FETs are currently under develop-
ment. Preliminary results showed good combining efficiency.
4. Quasi-Optical Components
Many components can be built using quasi-optical techniques. At millimeter-wave
frequencies (above 100 GHz), the quasi-optical components have many advantages of low
cost, low losses, high performance, and ease of fabrication.
Under this project, several useful quasi-optical components have been developed.
These include a 35 GHz rectifying antenna, a frequency selective surface filter, and a
low loss Fabry-Perot resonator.
4.1 35 GHz Rectifying Antenna
A quasi-optical rectifying antenna (rectenna) converts the RF power into DC power.
It could be used in microwave power transmission or in microwave imaging array.
The basic structure of a rectenna is shown in Fig 7. The low pass filter inserted
between the antenna and rectifying circuit is designed so that the fundamental frequency
can be passed and a significant portion of the higher order harmonics generated from the
nonlinear rectifying circuit be rejected back to the rectifying circuit. The rectifying circuit
I consists of an single diode shunt-connected across the transmission lines.
II
Ii
Gunn Diodes
4II
DC Bias Lines
i Figure 6. Two-element active antenna array
I
I
I
Figure 7. Block diagram of rectenna circuit
10
i
Ifile blasic principle of the Iicrowave power conversion by this rectifying circuit is
aiialogous to a diode clamping circuit or a large signal mixer at microwave frequencies.
The power conversion efficiency is inaxiinized by substantially confining all the higher order
liarnonics between the low pass filter and the DC pass filter, using an efficient diode and
imatching the diode's input impedance to antenna's input impedance.
A GaAs Schottky diode (Model DMIK6606, Alpha Industries) used in Ka-band mixers
%was selected for the 35 GHz rectenna. A 35 GHz rectenna was designed using a microstrip
dipole as the antenna element. The input impedance of the dipole as an element in an
I infinite array was calculated using Method of Moment. The impedances calculated froni
the fundaunental frequency to the seventh order harmonic were used as an input data file
to LIBRA. Tile length and the width of a resonant dipole at the fundamental frequency
have been determined as 0.46 A0 and 0.02 A0 respectively. The effective impedance of the
diode was assumed to be 50 Q considering that the output DC voltage should be less than
4.5 volts (half of Vhr) with an operating power level of 100 InW.
Figure 8 shows the circuit layout for the 35 GHz rectenna. Over 39% conversion
efficiency was achieved. The results agreed fairly well with the theoretical prediction using
a nonlinear circuit analysis [8] (See Appendix 3).
4.2 Frequency Selective Surface Filter
Frequency selective surfaces (FSS) have been used for a number of applications in-
cluding reflector antenna systeins and dielectric radome designs. A new application of an
FSS is to reject the harmonics produced by a rectifying antenna (rectenna). A rectenna is
a receiving antenna that converts a microwave bean into useful DC power. Harmonics of
significant power levels are prodiced by a diode that converts the RF to DC power. This
harmonic power is then radiated into free space which may cause potential problems. If
the rectenna were placed abroad a satellite, the harmonic radiation could interfere with
coIIIriiunication signals. It is desirable to suppress these harmonics by use of an FSS.
j A gridded square FSS has been developed to act as a band stop filter. The filter was
placed in front of a rectifying antenna array as shown in Figure 9. The filter was designed
to plss the fundamental frequcncy and reject the harmonics.
11I
SI+I
IFigure 8. 35 GHz rectenna circuit layout
IFSSRectenna Plane
I ) Reflecting Plane
I
Figure 9. A rectenna array integrated with a frequency selective surface
II
I4.3 Low-Loss Open Resonator Coupling and Filters
Fabry-Perot or open resonators have been used at microwave/millinmeter-wave fre-
quencies for diplexers, dielectric characterization, antennas, and power combining. Few of
these applications have addressed the problem of efficiently coupling power into and out
of an open resonator.
For a Fabry-Perot resonator to be made into a useful device, a method of coupling
power into and out of the resonator must be devised. Previous methods employed include
meshes, small apertures, waveguide apertures, dielectric launchers, and microstrip patch
antennas. Most of these methods introduce high losses due to poor coupling efficiencies.
To overcome this problem, a novel, simple, and low-loss coupling method using narrow
slots was developed. The feed for the antenna is a waveguide opening covered by a thin
half-wavelength resonant slot. The results show that the slot is a very efficient method for
coupling power into and out of an open resonator and that this coupling method can be
I used to build a low-loss filter.
Figure 10 shows an open resonator bandpass filter built using this coupling mechanism.
An insertion loss of less than 1 dB was achieved for the passband. The filters should have
applications in many millimeter-wave and submillimeter-wave systems [9].
15. Novel Planar Slotline and Coplanar Waveguide Circuits
In recent years slotline and coplanar waveguide (CPW) have emerged as an alternative
to microstrip in microwave and millimeter-wave integrated circuits (MIC). The fact that
th,- "onter cond, lctor and ground planes are on the same side of the substrate allows series
and shunt connections of passive and active solid-state devices. Use of CPW also avoids
I the need for via holes to connect the center conductor to ground which should help to
reduce processing complexity and increase yield in monolithic implementations.
Several novel slotline and CPW components have been developed under this project.
These include a tunable filter, a switchable filter, a Gunn VCO, and a tunable ring res-
onator. The details are given in the following.
I13
I7
I
IIiue1 Slote oT oa o PENINGe
OPNNIIII
II1
II
5.1 Coplanar Waveguide Tunable and Switchable Filters
A novel coplanar waveguide (CPW)-slotline banipass filter and a microstrip-to-slotline
transition has been developed. The circuit is fully planar and allows easy integration of
active and passive semiconductor devices in series and in shunt. To test the filter, a new
mnicrostrip to slotline transition was designed and two of these transitions exhibited an
insertion loss of less than 1.0 dB over the 2.0 to 4.0 GHz range.
Figure 11 shows the schematic diagram of the switchable filter with three sections and
three PIN diodes. Each section consists of a CPW resonator. For the tunable filter, the
PIN diodes were replaced by varactor diodes.
The three-section CPW-slotline bandpass filter demnonstrated an insertion loss of less
than 0.2 dB over a 300 MHz bandwidth centered at 2.9 GHz. The three-section CPW-
slotline switchable bandpass filter integrated with three PIN diodes was developed with 25.0
dB isolation and 0.7 dB insertion loss. The three-section CPW-slotline varactor-tunable
filter integrated with three varactor diodes was demonstrated with a 2.0 dB insertion loss
and over 20% electronic tuning range [10] (See Appendix 4).
5.2 Coplanar Waveguide Gunn VCO
Considerable effort has been directed toward development of microwave and millimeter-
wave receivers in hybrid and monolithic integrated circuit forms. To fully realize the ben-
efits of integration, the local oscillator should be included. This requires a local oscillator
in planar form. In addition, for many applications, long-term frequency stability and elec-
tronical frequency tunability are also desirable. To achieve these requirements, a high
Q resonator, a varactor and a Gunn device are incorporated into a truly planar circuit
c(nlfigurat ion.
The configuration exhibits a tuning bandwidth from 10.2 to 10.55 GHz with an output
power of 16.3 ± 0.45 dBm. The spectral purity and signal stability are comparable to
waveguide and dielectric resonator stabilized inicrostrip oscillator [11].
Figure 12 shows the novel varactor tunable CPW-slotline Gunn oscillator picture.
The circuit consists of a CPW-slotline resonator, a microstrip to slotline transition and a
slotline lowpass filter. A Gunn diode is placed on the box at one of the open ends of
I15I
I S s s sLI. c
UIO L2L~
PIN Diodes I
I Figure 11. Configuration of the PIN diode switchable CPW-slotline bandpass filter
Mcrostrlp toI Slotitne Tr'nsiton
I CPW-Resonator L2, LmL4 5
Gun Lowpass Filter
Micrst
Fiur 1. hevaacortuedCP-sotin GnnVC
I1
the resonator and a varactor at the other end. The circuit offers many advantages of low
cost, simplicity, small size, excellent performance, good reproducibility and fully planar
configuration. The circuit is amenable to monolithic iml)lementation.
5.3 Coplanar Waveguide Fed Slotline Ring Resonators
The search for new uniplanar MIC resonators is of great interest. Microstrip ring
resonators can be used for filters and allow easy series insertion of solid-state devices for
tuning, switching, and RF power generation. These rings have not been realized in truly
planar transmission lines such as slotline or CPW to take advantage of easy shunt device
hybrid integration. These truly planar designs should also reduce processing complexity
in monolithic applications. Under this project, a CPW fed slotline ring resonator was
developed with over 23% varactor tuning range and 4.5 ± 1.5 dB of insertion loss. A
wider tuning range can be obtained for a greater insertion loss variation. The tuning
range is wider and the circuit has greater flexibility over the microstrip ring. For instance,
the microstrip requires high impedance lines to apply DC biasing and the position of the
va.ractor is fixed at a series gap. Shunt varactor placement on a microstrip ring requires
drilling through the substrate. The slotline ring circuit, on the other hand, has inherent
DC biasing pads and shunt placement of the varactor can be optimized on the ring. The
configuration also lends itself to other device integration for filtering, switching, tuning and
RF power generation. Figure 13 shows the circuit configuration of this novel resonator [12].
5.4 Conductor-Backed Coplana" Waveguide
I CPWs with conductor backing to support the structure (Figure 14) offer an attractive
alternative to microstrip as a transmission media for microwave integrated circuit. The
usefulness of the conductor backing CPW was questioned due to the mode coupling and
I energy leakage.
Conventional conductor-backed (grounded) coplanar waveguides (CBCPW) lose en-
ergy from its dominant mode in the form of a. parallel plate mode (PPM) (leaky wave)
propagating in the substrate at all frequencies which can generate undesirable coupling
effects (moding problems) and produce an ineffective circuit. A multiple dielectric struc-
17
I ,-0.48 mm Slotline
I / / / / / / pling Gap(g)
IFigure 13. Tunable CPW-fed slotline resonator
I z
Is w Wg
Th2 (E r2)
38..1mm
/ conductor
Figure 14. Conductor-backed coplanar waveguide
18
ture with appropriate dimensions was designed to suppress the leaky wave. The spectral
domain method was used to predict the cutoff frequency for the waveguide. Experimental
data verifies this procedure and demonstrates the significance of the moding problems.
The results should be useful for CBCPW design [13].
6. Other Circuits and Analyses
In addition to the research in active antenna elements, spatial power combiners, quasi-
optical components, and coplanar waveguide circuits, work was carried out in the analyses
and developments of other microwave circuits and components. This section gives a brief
summary of these circuits.
6.1 Wide Inductive and Capacitive Strip Discontinuities
An analysis based on the variational method and moment method has been developed
to calculate the discontinuity susceptance due to a single or multiple inductive strips in
a rectangular waveguide as shown in Figure 15(a) and (b). The strips could have wide
widths located unsymmetrically on the transverse plane of the waveguide. The current
distribution on the strips has been determined by solving - of linear equations. The
theoretical results agree closely with experiments r1 1
The analysis was extended to calculate the susceptance of a wide resonant strip on
the transverse plane of a rectangular waveguide as s!,o:,in Figure 15(c) [15]. Almost
any value of susceptance can be obtained at a specified operating frequency by properly
choosing the strip depth and strip width.
6.2 Superconductive Strip Discontinuities
A two-gap electrically floating resonant strip was used for surface resistance measure-
ments of the high-temperature superconductor, YBa 2 Cu307- 6 as shown in Figure 16. The
method used has the advantages of simplicity, no electrical contact, operation at various
resonant frequencies, and of requiring only a small sample. An analysis was used that
allows for the accurate design of the strip dimensions to produce a desired resonant fre-
quency. Experimental measurements on resonant frequencies in X and Ku-band (8-18 GHz)
agree well with the calculations. The method allows one to extract the normalized surface
19
Iyb --
w_ .- --
(a)
I c ____
0
Z --I 2 X , C-M -.M
i b ....
i(b) . .
0 aN
XC I XC2 a cM (7
Figure 15. Strip discontinuities in waveguid-: (a) a wide inductive strip, (b) multiple
inductive strips, (c) a wide resonant strip
1 20
resistance of the sample from transmission coefficient measurements at the resonant fre-
quency. These normalized values were found to compare favorably to the Mattis-Bardeen
theory taken in the local limit. The resonant strip in waveguide should have applications
in high-temperatu:e superconductive material characterization and in the development of
waveguide superconductive filters [16].
6.3 Microwave/Millimeter-Wave Impedance Measuring Scheme Using a Three-Probe
Microstrip Circuit
A simple, compact and low cost three-probe microstrip impedance measuring scheme
has been developed. The impedance of an unknown load can be easily determined by
measuring the coupling power levels at the three-probes as shown in Figure 17. The
coupling coefficients for the three-probes can he unequal and arbitrary, and only one power
meter is required by using a single-pole three-throw switch. A prototype device has been
built at X-band. The measured results agree very well with those obtained from an HP
8510 automatic network analyzer [17]. The technique has been recently extended to the
S-parameter measurements [18].
I!I
II
I
Wavegulde Adhesive
sample
0 0
I Figure 16. A superconductive strip inside waveguide
P2
1 2
jSource Z-Z2
3
P3 Ip 1
I Figure 17. Microstrip three-probe measurement system
1 22
I
References
1. K. Chang and C. SUi, "Millimeter-Wave Power Combining T'echniques," IEEE Trans.
on Microwave Theory and Techniques, Vol. MTT-31, Feb. 1983, pp. 91-107.
2. J.W. Mink, "Quasi-Optical Power Combining of Solid-State Millineter-Wave Sources,"IEEE Trans. on Microwave Theory and Techniques, Vol. MTT-34, Feb. 1986, pp.273-279.
3 K. Chang, K.A. Hummer and J.L. Klein, "Experiments on Injection-Locking of Ac-tive Antenna Elements for Active Phased Arrays and Spatial Power Combiners,"IEEE Trans. on Microwave Theory and Techniques, Vol. MTT-37, pp. 1078-1084,July 1989.
4. K. Chang, K.A. Hummer, G.K. Gopalakrishnan, "Active Radiating Element UsingFET Source Integrated with Microstrip Patch Antenna," Electronics Letters, Vol. 24,No. 21, pp. 1347-1348, Oct. 1988.
5. J.A. Navarro, Y.H. Shu and K. Chang, "Broadband Electronically Tunable Planar
Active Radiating Elements and Spatial Power Combiners Using Notch Antennas,"IEEE Trans. on Microwave Theory and Techniques, Vol. 40, Feb. 1992, pp. 323-328.
6. R.E. Miller and K. Chang, "Integrated Active Antenna Using Annular Ring MicrostripAntenna and Gunn Diode," Microwave and Optical Technology Letters, Vol. 4, No.1, Jan. 1991, pp. 72-75.
7. A.M. Kirk and K. Chang, "Integrated Image-Line Steerable Active Antennas," Inter-national Journal of Infrared and Millimeter-Waves, Vol. 13, ,June 1992, Ptp- 841-850.
S. T. Yoo and K. Chang, "Theoretical and Experimental Development of 10 and 35 GHzRectennas," IEEE Trans. oni Microwave Theory and Techniques, Vol. 40, June 1992,pp. 1259-1266.
9. J.C. McCleary and K. Chang, "Low-Loss Quasi-Optical Open Resonator Filters,"1991 IEEE-MTT Microwave Symposium Digest Technical Papers, Boston, MA., June
1991, pp. 313-316.
10. Y.H. Shu, J.A. Navarro and K. Chang, "Electronically Switchable and Tunable Copla-nar Waveguide-Slotline Bandpass Filters," IEEE Trans. on Microwave Theory and Tech-niques, Vol. 39, Mar. 1991, pp. 548-554.
11. J.A. Navarro, Y.H. Shu and K. Chang, "A Novel Varactor Tunable Coplanar Waveguide-Slotline Gunn VCO," 1991 IEEE-MTT Microwave Symposium Digest Technical Pa-
ers, Boston, MA., June 1991, pp. 1187-1190.
I23
12. J.A. Navarro, L. Fan, and K. Chang, "The Coplanar Waveguide-Fed Electronically
Tunable Slotline Ring Resonator," 1992 IEEE-MTT Microwave Symposium DigestTechnical Papers, Albuquerque, New Mexico, June 1992, pp. 951-954.
13. M.A. Magerko, L. Fan and K. Chang, "Multiple Dielectric Structures to EliminateModing Problems in Conductor-Backed Coplanar Waveguide MIC's," IEEE Microwaveand Guided Waves Letters, Vol. 2, June 1992, pp. 257-259.
14. B.H. Chu and K. Chang, "Analysis of Wide Transverse Inductive Metal Strips ina Rectangular Waveguide," IEEE Trans. on Microwave Theory and Techniques, Vol.37, July 1989, pp. 1138-1141.
15. B.H. Chu and K. Chang, "Analysis of a Wide Resonant Strip in Waveguide," IEEETrans. on Microwave Theory and Techniques, Vol. 40, Mar. 1992, pp. 495-498.
1 16. M.K. Skrehot and K. Chang, "New Resistance Measurement Technique Applicable toHigh-Temperature Superconducting Material at Microwave Frequencies," IEEE Trans.on Microwave Theory and Techniques, Vol. 38, Apr. 1990, pp. 434-437.
17. K. Chang, M.Y. Li and T.A. Sauter, "Low Cost Microwave/Millimeter-Wave ImpedancejMeasuring Scheme Using a Three-Probe Microstrip Circuit," IEEE Trans. on Micro-wave Theory and Techniques, Vol. 38, Oct. 1990, pp. 1455-1460.
18. M.Y. Li and K. Chang, "A Three-Probe Microstrip Measuring System for S-ParameterMeasurements," Electronics Letters, Vol. 27, No. 10, May 9, 1991, pp. 836-837.
24
List of Personnel
1. Principal Investigator: Kai Chang
2. Research Associates: M. LiY. Shu
3. Research Assistalits: B. Chu J.C. McCleary/Students G. Gopalakrishnan J.O. McSpadden
M. Ho J.A. MillerK.A. Hummer R.E. MillerA.M. Kirk J.A. NavarroW.K. Leverich M.K. SkrehotG. Luong T. YooM.A. Magerko
4. Degree Awarded
Master of Science: J.A. Miller August 1989R.E. Miller August 1989M.K. Skrehot August 1989J.A. Navarro December 1990A.M. Kirk May 1991J.C. McCleary December 1991
Ph.D: G. Gopalakristnan December 1990
B. Chu August 1991
25
I
Report of Invention
IA patent application entitled "Planar Active Endfire Radiating Elements and Copla-
nar Waveguide Filters with Wide Electronic Tuning Bandwidth," has been filed. This
patent is now in the final stage of approval by the U.S. Patent Office. K. Chang, J.A.
Navarro and Y.H. Shu are co-inventors of this patent.
I
26
List of Publications Under ARO Support
A. Journal Publications
1. B.H. Chu and K. Chang, "Analysis of Wide Transverse In(uctive Metal Strips ina Rectangular Waveguide," IEEE Trans. on Microwave Theory and Techniques, Vol.37, July 1989. pp. 1138-1141.
2 K. Chang, K.A. Hummer and J.L. Klein, "Experiments on Injection-Locking of Ac-tive Antenna Elements for Active Phased Arrays and Spatial Power Combiners,"IEEE Trans. on Microwave Theory and Techniques, Vol. 37, July 1989, pp. 1078-1084.
3. J.A. Miller, M.Y. Li and K. Chang. "A Simple Microstrip to Dielectric WaveguideTransition," Microwave and Optical Technology Letters, Vol. 2, Nov. 1989, pp. 393-398.
4. J.L. Klein and K. Chang, --Optimun Dielectric Overlay Thickness for Equal Even-and Odd- Mode Phase Velocities in Coupled Microstrip Circuits," Electronics Letters,Vol. 26, No. 5, Mar. 1990, pp. 274-276.
5. M.K. Skrehot and K. Chang. "New Resistance Measurement Technique Applicable toHigh-Temperature Superconducting Material at Microwave Frequencies," IEEE Trans.on Microwave Theory and Techniques, Vol. 38, Apr. 1990, pp. 434-437.
6. K. Chang, M.Y. Li and T.A. Sauter, "'Low Cost Microwave/Millimeter-Wave ImpedanceMeasuring Scheme Using a Three-Probe Microstrip Circuit," IEEE Trans. on Micro-wave Theory and Techniques, Vol. 38, Oct. 1990, pp. 1455-1460.
7. M.IN. Skrehiot and K. Chang, "Analysis and Modeling of a High-Temperature Super-conducting Floating Resonant Strip in Waveguide," International .Journal of Infraredand Millimeter-Waves, Vol. 11, Dec. 1990, pp. 1355-1376.
S. J.A. Navarro, K.A. Humner and 1K. Chang, "Active Integrated Antenna Elements,"(Invited Paper) Microwave Journal. Vol. 34..Jan. 1991. pp. 115-126.
9. R.E. Miller and K. Chang, "Integrated Active Antenna Using Annular Ring MicrostripAntenna and Gunn Diode," Microwave and Optical Technology Letters, Vol. 4, No.1. Jan. 1991, pp. 72-75.
10. Y.H. Shu, J.A. Navarro and K. Chang, " Electronically Switchable and Tunable Copla-nar Waveguide-Slotline Bandpass Filters," IEEE Trans. on Microwave Theory and Te-chniques, Vol. 39, Mar. 1991. pp. 548 554.
27
11. J.A. Navarro, Y.H. Shu and K. Chang, "Broadband Electronically Tunable PlanarActive Radiating Elements and Spatial Power Combiners Using Notch Antennas,"IEEE Trans. on Microwave Theory and Techniques, Vol. 40, Feb. 1992, pp. 323-328.
12. H.B. Chu and K. Chang, "Analysis of a Wide Resonant Strip in Waveguide," IEEETrans. on Microwave Theory and Techniques, Vol. 40, Mar. 1992, pp. 495-498.
13. T. Yoo and K. Chang, "Theoretical and Experimental Development of 10 and 35 GHzRectennas," IEEE Trans. on Microwave Theory and Techniques, Vol. 40, June 1992,pp. 1259-1266.
14. M.A. Magerko, L. Fan and K. Chang, "Multiple Dielectric Structures to EliminateModing Problems in Conductor-Backed Coplanar Waveguide MIC's," IEEE Microwaveand Guided Waves Letters, Vol. 2, June 1992, pp. 257-259.
15. A.M. Kirk and K. Chang, "Integrated Image-Line Steerable Active Antennas," International Journal of Infrared and Millimeter-Waves, Vol. 13, June 1992, pp. 841-850.
16. G. Luong and K. Chang, "Interconnect of Microstrip Lines Through a Narrow Rectan-
gular Slot in the Common Ground Plane," Microwave and Optical Technology Letters,Vol. 5, July 1992, pp. 388-393.
17. J.O. McSpadden, T. Yoo, and K. Chang, "Theoretical and Experimental Investigation
of a Rectenna Element for Microwave Power Transmission," Accepted by IEEE Trans.on Microwave Theory and Techniques, Vol. 40, To appear in Dec. 1992.
IS. J.A. Navarro and K. Chang, "Broadband Electronically Tunable Integrated CircuitActive Radiating Elements and Power Combiners," (Invited Paper) Microwave Jour-nal. Vol. 35. To appear in Oct. 1992.
B. Conference Papers
1. J.L. Klein and K. Chang, "Theoretical and Experimental Studies of Dielectric OverlayMicrostrip Circuits for Directional Couplers," 14th International Conference on In-frared and Millimeter-Waves, Wurzburg, Germany, Oct. 1989. pp. 145-146.
2. NI.K. Skrehot and K. Chang, "'Surface Resistance Measurements of High-Tc Super-conductors in X and Ku-Bands," 14th International Conference on Infrared and Mill-imeter-Waves, Wurzburg, Germany, Oct. 1989, pp. 521-522.
3. J.A. Navarro, Y.H. Shu and K. Chang, "Active Endfire Antenna Elements and Power
Combiners Using Notch Antennas," 1990 IEEE-MTT Microwave Symposium DigestTechnical Papers, May 1990, pp. 793-796.
28
I4. K. Chang, "Rece nt Developments of Microwave/Millimeter- Wave Integrated Active
Antenna Elements," 15th International Conference on Infrared and Millimeter-Waves,Orlando, Florida, Dec. 1990, pp. 520-522.
5. M.Y. Li and K. Chang, "Scattering Parameter Measurements Using a Three-Probe Mi-crostrip Circuit," 15th International Conference on Infrared and Millimeter-Waves, Or-lando, Florida, Dec. 1990, pp. 708-710.
6. K. Chang, "Integrated Circuit Active Antenna Elements for Monolithic Implementa-tion," (Invited Paper) SPIE Conference on Monolithic Microwave Integrated Circuits,paper 1475-19, Apr. 1991.
7. J.C. McCleary and K. Chang, "Low-Loss Quasi-Optical Open Resonator Filters,"1991 IEEE-MTT Microwave Symposium Digest Technical Papers, Boston, MA., June1991.
S. J.A. Navarro, Y.H. Shu and K. Chang, "Wideband Integrated Varactor-Tunable Ac-tive Notch Antennas and Power Combiners," 1991 IEEE-MTT Microwave SymposiumDigest Technical Papers, Boston, MA., June 1991.
9. J.A. Navarro, Y.H. Shu and K. Chang, "A Novel Varactor Tunable Coplanar Waveguide-Slotline Gunn VCO," 1991 IEEF-MTT Microwave Symposium Digest Technical Pa-pers, Boston, MA., June 1' .
10. J.C. McCleary and K. Chang, "Characterization of Slot-Fed Quasi-Optical Resonators,"Progress in Electrrinagnetics Research Synposium, Boston, MA., July 1991, p. 410.
11. J.A. Navarre. Y.H. Shu and K. Chang, "A Ka-Band Integrated Active Notch An-tenna," 16th International Conference on Infrared and Milliietcr-Nkaves, Lausanne,Switzerland, Aug. 1991, pp. 290-291.
12. A.M. Kirk and K. Chang, "inage-Line Voltage Controllable Active Antennas," 16thInternational Conference on Infrared and Millimeter-Waves, Lausanne, Switzerland,Aug. 1991, pp. 288-289.
13. M.Y. Li and K. Chang, "Techniques for S-Parameter Measurements Using a Three-Probe Circuit," 16th International Conference on Infrared and Millimeter-Waves, Lau-sanne, Switzerland, Aug. 1991, pp. 362-363.
14. T. Yoo, J.O. McSpadden, and K. Chang, "35 GHz Rectennas Implemented with aPatch and a Microstrip Dipole Antenna," 1992 IEEE-MTT Microwave Symposium Di-gest Technical Papers, Albuquerque, New Mexico, June 1992, pp. 345-348.
29
15. J.O. McSpadden, T. Yoo and K. Chang, "Diode Characterization in a Microstrip Mea-surement System for High Power Microwave Power Transmission," 1992 IEEE-MTTMicrowave Symposium Digest Technical Papers, Albuquerque, New Mexico, June 1992,pp. 1015-1018.
16. J.A. Navarro, L. Fan and K. Chang, "The Coplanar Waveguide-Fed ElectronicallyTunable Slotline Ring Resonator," 1992 IEEE-MTT Microwave Symposium DigestTechnical Papers, Albuquerque, New Mexico, June 1992, pp. 951-954.
30
Appendices
Appendix 1
K. Chang, K.A. Hummner and J.L. Klein, "Experiments on Injection-Locking ofActive Antenna Elements for Active Phased Arrays and Spatial Power Combin-ers," IEEE Trans. on Microwave Theory and Techniques, Vol. 37, July 1989, pp.1078-1084.
Appen(dix 2
J.A. Navarro, Y.H. Shu and K. Chang, "Broadband Electronically Tunable PlanarActive Radiating Elements aid Spatial Power Combiners Using Notch Antenas,"IEEE Trans. omi Microwave Theory and Techniques, Vol. 40, Feb. 1992, pp. 323-328.
Appendix 3
T. Yoo and K. Chang, "Theoretical and Experimental Development of 10 and 35GHz Rectennas," IEEE Trans. on Microwave Theory and Techniques, Vol. 40.June 1992, pp. 1259-1266.
Appendex 4
Y.H. Shu, J.A. Navarro and K. Chang, "Electronically Switchable and TunableCoplanar Waveguide-Slotline Bandpass Filters," IEEE Trans. on Microwave The-ory and Techniques, Vol. 39, Mar. 1991, pp. 548-554.
31
Appendix 1lWN III I RAN A( I IONs ON 1,1( Ri)OVA\I HIOR, ANt) I (HNIQULI, VOl 37. NO 7, Ili[ y 99
Experiments on Injection Locking of ActiveAntenna Elements for Active PhasedArrays and Spatial Power CombinersKAI CHANG. SENIOR MEMBER, Irtr-, KENNETH A. HUMMER, SrUD)-NI MIMBI:R. IEEL,
AND JAMES L. KLEIN
.4hAbtracti -1o pes of active antenna elements have been studied L
experimentall*. One ",pe uses a microstip antenna with an active devicemounted directl) on the antenna- The other uses ar, active device coupledto a microstrip patch antenna through an aperture. Microstrip active Dantenna elements and two-element arrays have been demonstrated for both Wrpe,, of circuit,. Injection locking of the antenna elements has been
achieved through space and mutual coupling. The circuit Q factor wascalculated based on the locking gain and the locking bandwidth. The poweroutput from tsso elements has been successfulli combined in free space"ith a combining efflcenc% of over 90 percent. For a single active antenna Guni Diodewith a Gunn diode mounted directl% on the patch, an electronic tuning C Bias Linerange exceeding 9 percent has been achieved bv var.ing the dc bias. There-,,lt should have man% applications in low-cost active arrays. active (a)iranmitier, and spatial power combiners.
I Gunn Diodes
I. INTRODUCTION AND BACKGROUND
R ECENT DEVELOPMENTS in solid-state devices andmicrossave, millimeter-wave integrated circuits have
made it possible to combine the active devices with planarantennas to form active arrays. Manv elements can becombined to build an active phased array or a spatialpo,,er combiner. These techniques allowh monolithic imple- -mentation b. fabricating both active devices and antennason I simle semiconductor substrate The circuits can bemade at lo.\ cost and should have many applications in D iLradar, communication, and EW systems.
(b)Txso types of active array configurations have been Fiinestigated. The first type uses active devices mounted Jig 1 ia Single- and dh) two-element acnve antenna etments withdes, ices mounted directl\ on antennasdirectl\ on planar antennas (shown in Fig. I). The secondJ'pe uses a transmit- receive (T/R) module mounted di-rectl\ behind an antenna element (shown in Fig. 9). In tions due to the moding and size problems. Consequentl,.these circuits, each element acts as a low-cost transmitter. the maximum number of devices that can be combined is\tan\ of these elements can be combined to form a spatial limited. To overcome these problems, spatial or quasi-opti-posser combiner or a quasi-optical power combiner. cal combiners have been proposed to combine the powser
\ian\ power-combining approaches have been demon- in free space or in a Fabry'-Perot resonator 131. An activestrated in the microwave and millime:er-wave frequency device mounted directly on a planar antenna forms arange 111, [21. Most of these techniques have serious limita- module for spatial or quasi-optical combiners. In order to
achieve coherenc and effective combining in free space.Manuscript recei\ed March 25. I9X8, revised Januarv 23. 1981) This the modules will be injection locked to each other throughsvork a, upported in part h% the ArmN Research Offict and b TR\\ mutual coupling or through an external master source.K Chang and K A Hummer are with the Department of Electrical u
t neinerin. Texas A&M Universit,. College Station. TX 77S43-3129 This paper reports the design and measurements of aI t Klein \ka, vith the Department of Electrical Engineering. Texas single active patch antenna with a Gunn diode oscillator-\&N1 I nt~er~ii\. College Station. TV fi[e is no\4 wi,,th Texas Instrument,,In, Dallas, T\ 2 Stao T integrated directly on the patch. The circuit forms an
IFIt Lop Number 8929198 element for spatial or quasi-optical combiners. The output
0018-)480) 8910700-1078S01.0( , 1989 IEEE32
, II\ t ,/Ni i I XII H 1%1F NI. I Nil III. II) KIN'. i,,
p,,c found c' Xaloparablle to ;I w;'tCI?,RtIl t-iritit usi, I CIrC
t e N' lm e 411111 diod es I he , i\e ,Iiken l Can l be iiade I l l 1 t i .i C d U J I A rieI .
tunable over a 1) percent andwidith 1,, \,ir\ing l tihtijil coid tictice of tie two edges of file ml-Ihis ,Idl luiing ralnge colmpared to a pjdssIvt patch,antcnni st a IoVt.er loaded Q factor lhe (listince front ei a feed
i)g dlkhdllqu" inpu t impedance l e l llell a t the feedIV\, o of these active antentlas were successfullv con,- location,
hi ed to form in arrav. Received output power W-a.s ap- X guided %ka' elecnth.tOs, ma.te[, doubled, indicating a combining efficiencv of
over 1h) percent. i-xperinental results also showed that the For a rectangular patch wit ll idth It' ). 3A,, and lengthalra antelind pattern broke froim a single beam into t(Vt1 /. - 2 2A/. (1, and (G_ are given b, James ct alseparate beanis as the dc bias voltage of one of the Illjsantnnas ,xas varied. This resulted in a tuning range ofabout 1 percent, which is lower than that of a single active ,I
patch alltenlla.I he second type of circuit configuration (shown in Fig.
9t uses a 1,, R module mounted directly behind an antenna 0.32. (3eleiment. Iwo commonly used feeding arrangements arethe space-feed and corporate-feed techniques. Many T/R Here ,., is the equivalent length to account for open-endmodules using FEr's have been developed for these appli- fringing capacitance. The input impedance /, w.as setcations 141 161. An aperture-coupled microstrip to patch equal to the magnitude of the active device resistance.antenna circuit is suitable to connect the T/R module to assumed to be 8 Q2 1121.the antenna element. The circuit can also be used in spatial The two-element active array is also show n in Fig. 1power combiners if the T/R mod, ule Is replaced by ainap ower combinesigtn T is replupon a ced bnalysis asing Each element was designed to have the same dimensions +t,oscillator. The design is based upon an analysis using the single antenna. Two antenna elements were separatedatperture coupling theory and an S-parameter matrix [7]. A tl igeatna w nen lmnswr eaaeatuecouling heor y anricd and teter atr t 11.4 A by one quarter of the guided wavelength. The Gunn diodetwo-element array was fabricated and tested at aboult 2.4 i~apcae iltp id rmMACM tpo
Gliz Inecton lckig troug muualcoupingwasis :t packaged pill-type diode from M/A COM. It pro-Gill,,. Injection locking through mutual coupling was duced 10-25 mW of output power at 10 GHz in an
demonstrated and good power-combining efficiency was o ted wav0guide cirut.achived.optimized wavegude circuit.
;ii.hi eyed.
II. PA( it AN IFNNA WI in A(cIviv Drvict 11.. R SUIP lS FOR A S N( .F Acinvi:
MOtN tEt) DIRI rl Y ON AN IINNA PAilil A NENNAl o measure the power output from the active antenna, aiw-cost transmitters and power combiners can be niule standard horn antenna was used as a receiver. The acti, e
h, mount tin the active devices directly on antenna ele-
n ents. Single-element microstrip active patch antennas patch antenna has an output power P and a gain of G \ha~ ben rpored y Toma era!.181andPerins1~ standard horn antenna with a gain of G, was placed .1ha\cedtPdistance R away from the active antenna. The received
using Gunn and IMPAUT diodes. Power combining using d
an open resonator has also been reported 1101. However. power P, can be measured using a power meter. The
no attempt was made to injection lock two antenna ek'- output power P can be calculated using the followingments through mutual or spatial coupling, to measure tile equation:
factor. or to electronically tune the active antenna 47R Ielement. Furthermore, no attempt was made to combine P,,= P - - (4)the output power from two active antennas in free space. ", i2
Both the single patch antenna and the two-element array Here it is assumed that the two antennas are well alignedwere constructed on Duroid 5870 substrate with a thick- and matched, and the polarization loss is neglected.ness of 1.524 mm. The circuits were designed at X-band For a single active patch antenna, the maximum outputusing Gunn diodes and patch antennas. The circuit config- power is about 15 mW at 10.1 0Hz. This power is calcu-urations are shown in Fig. 1. lated using (4) based on a 15.4 pW received power. The
The antenna dimensions were determined by equations output power and frequency as a function of dc bias aregiven bv James et al. [11]. The placement of the active shown in Fig. 2. A 3 dB tuning range of 839 MHz ,wasdevice was chosen such that the device impedance was achieved from 9.278 to 10.117 GHz. This tuning range ismatched to the input impeda-ce of the patch. The diode equivalent to a 9 percent bandwidth, which is much widerplacement location D is given by than that of a single passive patch.
AR _GThe E-plane antenna patterns for several bias voltages1) 2_ cos 2Z,,,G, 1 ) (1) are shown in Fig. 3. This shows a beam width of 90'. The
large peak present at - 60' is most likely due to radiation
I I I I I I I I3
1080 IEEL TPANSACrIONS ON MICROWAVE THEORY AND ILCHNIOULS. voi- 37. NO. 7, JULY 1989
10 2. :6
o 98. *0
U 96 azW
0 9. 4o
00 FQEQ 2.
D. C. BIAS VOLTAGE (volts)
Fig 2 Output power as a function of frequency and bias voltage for asingle active antenna.
(a)
BIAS VOLTAGE
V-8.99 VOLTS
.... V=10.21 VOLTS
SV=12.00 VOLTS0.
-5-
-10.z
-15
-80 -40 0 40 80
ANGLE (degrees)
Fig 3 E-plane antenna patterns for several different bias voltage levelsfor a single active antenna. (b)
Fig. 4. Signal spectra (a) before and (b) after the injection lockingVertical: 10 dB/di. Horizontal: 500 kHz/Div Center frequentc:
from the dc bias line that is required to bias the Gunn 9.967 GHzdiode. The plot also shows that the pattern changes verylittle as the bias voltage (and thus the frequency and 70
power) 1s varied.The single patch antenna can be injection locked by an 60-
external signal incident on the patch. Fig. 4 shows the50
frequency spectrum before and after the injection locking.The frequency stabilization and the noise suppression pro. 40
vided by injection locking are evident. Fig. 5 shows the Ri 30
injection-locking bandwidth versus locking gain (which is
defined as P0 /IP,). The external Q can be found from 20Adler [131:
*02f° p 22 ?4 26 28 30 32 34 36
Q, = (5) GAIN (dB)Fig. 5. Injection-locking bandwidth as a function of locking gain
where
Q, =external Q factor, Q factor measured here compares favorably with thosef, = operating frequency, previously reported for microstrip oscillators [14], [151.If = injection-lock bandwidth,P, = injection-lock signal power, IV. RESULTS FOR TwO-ELEMENT ACTIVE ARRAY
P0 = free-rutining oscillator power. The two active devices in the two-element array will
For the results shown in Fig. 5, the external Q value is injection lock each other through mutual coupling. Injec-about 20. This low Q value explains why a wider tuning tion locking may also be obtained by the use of an externalrange was achieved for a single active patch antenna. The source.
34
(HANG e al. I XPIRI4I NIS ON INJI.CIION LOCKING 108i
.... A ON D C BIAS VOLTAGE
--- 3 ON- 1OTH ON - V 1451
---. V = 1587v
0 0
3-'5, .. * . .
:4 40) 80
-80 -40 0 40 8o 4oA'lGLE (degrees)
Fig. 6 Measured E-plane power pattern for a two-element array. Fig. 8. Pattern broken up above 15.45 V.
o
!0 5- -35 31.
3 .... -3 To Active Element
10.4- ................
-25 .0-23
20 0 Microstrip Line
u 0.2 (a~-'5.0.1 Patch Antenna -Q 3O.
. (a)
FREQUENCY Z .........
14.0 1 5. O .5 16.0 16.5
0. C. BIAS VOLTAGE (volts) m Patch Antenna
Fig. 7 Frequency and received power as a function of bias voltage ofone diode.
The measured E-plane pattern for the array is shown inFig. 6 Also shown are the patterns with either antenna"off." It can be seen that when both antennas are "on," PlT/a Ground
the beam width is narrower and the gain is thus higher. P-Gnd
The patterns were normalized to the peak radiation power.T he bias to either diode was optimized individually to v .....achieve the maximum output power. The received bore-sight power and frequency as a function of dc bias on one ZLdiode are given in Fig. 7. An output power level of 30 mW Microstrip Linewas achieved at 10.42 GHz. The power was calculated by (b
using (4) with a two-element array antenna gain. This Fig. 9. Aperture-coupled microstrp to patch antenna circuit for activeoutput power level is about twice that from a single patch array (a) Top view. (b) Cross-sectional view.
active antenna. This demonstrates good combining effi-ciency. It can be seen that the boresight power and the phenomenon has recently been reported by Stephan andoperation frequency of the array experience a severe drop Young [161 for a different type of circuit.at a bias voltage of 15.45 V. To investigate this phe- V APERTURE-COUPLED PATCH ANTENNA CIRCUITSnomenon, antenna patterns were made for bias voltages Vabove and below 15.45 V. These results are shown in Fig. One practical difficulty with active arrays using T/R8. It was found that the radiation pattern broke from a modules is in isolating the input and output signals andsingle beam into two separate beams above this bias volt- maintaining the stability of the array. Another problem isage. The useful electronic tuning range is about 1 percent maintaining unidirective radiation and avoiding spuriousdue to the breakup phenomenon. It is believed that this radiation from feed lines.breakup is caused by the loss of phase lock. This phe- To overcome these problems, a two-sided substrate cir-nomenon should not pose any serious problem in practical cuit has been proposed (171. As shown in Fig. 9, the activeapplications for narrow-band systems. A similar breakup circuits, which include the oscillators, amplifiers, and phase
35
1082 It 1. ItKANSACTIONS ON MICROWAVE THEORY AND T-.ECHNIQUtS. VOL 37. NO 7. mit y I 989
TABLE IDIMENSIONS i-OR Two APERTUiRE-Cotiri.Li) AN If NNAS Points Frequency
antennat widtq29en29yparameter (mm) Patch A Patch B 1 2.2000 GHz
antenna width 29.5 29 5 2 2.3365 GHz
antenna length 39.5 39.0aperture size 10.25 10 0 Antenna Aline length 500 50.0 1 Antenna B
stub length 19.0 19.0
Edge separation between elements is 19.5 mm.
shifters. are located on the bottom side of the groundplane. The antenna elements are located on the top side of
the ground plane. The ground plane provides a good heat
sink for the active devices. The coupling between the two Fig. 10 Input impedance measurements on the vo antenna-s using an
layers is accomplished by circular apertures in the ground HP 8510 network anal\-zcrplane. The ground plane separates the radiating aperture
from the feed network, eliminating the possibility of spuri-ous signal radiation from the source. Because of the goodisolation between the radiating antenna and active device, 40the antenna and active circuits can be optimized sepa-30rately. Furthermore, since two substrates are used, one can t, 20use a low-dielectric-constant substrate for the antennas to . 10increase the efficiency. and a high-dielectric-constant sub- 0 Frequency (GHz)strate (such as GaAs) for the active circuitry. These fea- o 2.20 2.35 2.50Z -10tures have made aperture-coupled patch antenna circuits a0very attractive structure for active array applications. I
The design of the aperture-coupled patch antenna was W -30
lased on the analysis reported by Gao and Chang [7]. A -40six-port network was used to model the coupling circuitbased on the aperture-coupling theory and an S-parameter Fig. 11. Mutual coupling between two aperture-coupled antennasmatrix. The input impedance as a function of frequencycan be calculated using this analysis .. A ON
The spacing between the elements is a prime considera- - OTH ON
tion in the design of the array and is chosen based upon .... B ON
the coupling requirement for injection locking. For the o -structure considered here, one antenna is to be connected **-" ,to a sweeper. and the other to a free-running oscillator. -5-
The free-running oscillator will be injection locked to thesweeper signal through mutual coupling. The couplingbetween antennas was designed at 20 dB (thus providing Z -
20 dB of injection-locking gain). Using the data fromJedlicka. Poe, and Carver 1181, and assuming E-planeantenna coupling, it can be seen that 20 dB couplingcorresponds to a one-quarter-wavelength edge separation. -so -40 o 4'C' s0
ANGLE (degrees)
VI. TwO-ELEMENT ACTIVE ARRlAtY USING Fig. 12 E-plane power pattern of a two-element aperture-coupled
APERTURE-COUPLED PATCH ANTENNAS active arras
To demonstrate the concept, a two-element active arraywas designed and built on Duroid 5870 substrate operating 19.4 dB, very close to the desired value of 20 dB. Theat around 2.36 GHz. The dimensions for the two antenna bandwidth was also measured. The bandwidth foraperture-coupled antennas are given in Table I. an input VSWR of less than 2.0 is 28.5 MHz for one
Smith charts showing the input impedance for the two antenna and 31.5 MHz for the other.antennas are given in Fig. 10. It can be seen that the two The antenna pattern and injection-locking bandwidthantennas have similar impedance characteristics and res- were measured in an anechoic chamber. One antenna wasonate at almost the same frequency. The coupling between connected to a sweeper and the other to a free-runningantenna elements was also measured and is given in oscillator. The oscillator was a transistor oscillator manu-Fig. 11. It can be seen that maximum coupling is about factured by EMF Systems Inc. The sweeper was used to
36
iA , ; I \I'I kI %II% 1' 1)% IN]I( IION I (X KIN(, O
;N %~as measured. This beam wkidth was reduced to 65' when... ~.'*Nhoth sources %kere "on." Furthermore. the power w~ith both
- i J, I N.source" operating was about 2 dlB higher than the powerwith anv single source operating. This shows that the arrayis exhihiting good power-combhining properties. With bothsources on, the main lobe is centered at roughlN 25'. Thisoff-center condition is due ito the different lengths of thetransmission I nes used to connect antennas and sources. A
4 . phase difference thus exists between the t'wo sources. Thisdifference can be adjusted and overcome b,, the use of atransmission line section or phase shifter.
Fig. 13 shows the if-plane pattern of the arraN. It can be4 C) seen that no matter which sources are operating. the an-
ANG' I (degrees) tenna pattern remains relatively unchanged. This Is, to beFig 1 /-;)line pattern At a two-ctemeni aperture- coupled active expected since the antennas are arranged for F-plane
array, Coupling. Very little li-plane coupling can be expected.The antenna H-plane beam width was abc-'at 1 20'.
The injection locking through mutual coupling was alsodemonstrated. Fig. 14 shows the oscillator spectra beforeand after the injection locking. It can be seen that injectionlocking has a dramatic effect in reducing oscillator noise.The locking bandwidth was measured to be 2.15 MHz.Assuming an injection-locking gain of 2-0 dB. the lockingbandwidth corresponds to an external Q factor of 217. Thenarrow locking bandwidth and high Q factor are believedto be due to the high-Q transistor oscillator.
VII. CONCLUSONS
Two types of active antenna elements have been investi-gated. The first type uses active devices directly mountedon the antennas. The second type uses an apertu re-cou pled
(a) microstrip to patch antenna circuit which can be used toaccommodate a transmit -receive module. A t%%o-elementarray was built and demonstrated in both cases. Injectionlocking was achieved by using either mutual coupling or anexternal master source. Good power-combining efficiencywas achieved for both circuits. An electronictnn'ag
of over 9 percent was achieved for the single active an-tenna element and of about I percent for a two-elementarray.
ACKNOWLEDGMENT
The authors would like to thank X. Gao and Dr. R. D.Nevels for many helpful discussions.
(11 K. J Russell. "Microwave powAer combining technique,.- IEETrans Virwv Theory Tech., sot MT-T-2'. pp 4-1 41s. MaN
(hi 1979)-1 .pctr htfat (I fie lckig 121 K Chang and C. Sun. -Mi ltimeter- wave power combining, tech.
f:~~ 1 (~Iatr ii ad iflietlonniques.- IEEE Tranm Aficroware rter Tech . ..,IT-1.pcriic:al 1 iii t% ; Horizontal IN~) kiz/div (-enter frequency- NITT - 7pebI9p
2~~3 J.4~d; ~ A'W Mink. "Quasi-optical powei combining of Notid-Ntaic rnittimc-ter-\4a'. sources."JEEE- Trans Microwave Thcri Te~ i . %ol
injetio lok te fee-unnng ocilato bymutal ou- MT. 34. pp. 273-279, Feb. 19R6injctin lck he reeruningoscllaor y mtua co- 1 It C Johnson, R. 1F Marx. A. SanchC7. and F Mskictsn. "A
pling. circutart'. potarized actise antenna arl"as using miniature ,~Fig. 12 shox the I~paepattern of the active array [FT amplifiers." in 1984 IEH&.41T7-S bit kfuti. ii p
%xhe on,, he vecer ,, on."whe onv te ocilat(' - DIV . pp 2601 262.whe on\ te s~erie ;- "o." he ony te ocilatr is 1 1 C R Green vi al., "A 2 watt 6aAs TX/RX mt-dulc ssiih incepraij
"on," and v hen bo 11 sourco. are "on.'' When one of ithe canirl circuitrN for S-hand phased arias radars.- in IvN 11 1-souress %& a'f dlmnage , ;i I 1P beam 'Aidtt, of about 90' fi I .% /int kfir,wtve Svimi, Dig. pp 933-9t
37
l~~l~~4 tilt I KANSA. MI tN(IN SiRR OWAVI IltOh',ANt)II( lINIOIIS.1Vol 17. No 7. ji) lyxI
161 1 Piciits aind R. (louse. "An ultratnsAiureS W 1(fill. 2?V 'A' iiglics Aiici.ift ( ioitpafls. Torrance, (A. where he %a, insssls~d tir the,.115511 ms Iodule n for ii .ic apertlure appi i~aio55 il I VS Il1 1i rcxca.r~h andi dcvcelipinent of millsimeter w ,.is des iccs andi circuits 11 hi
it I I N\ lilt tit, ,, ae n il? Xsp ,)5 pil 5441 944 i i resultedi in state-of- the. art IMP1A-TI oscillator anrd power corn('7 \ (,a,, acid K Chang, -Nctwisrk miodeling si all ape ine Ckonpling biner performance at 4., 140. and 217 Gill Oi, tr actis sties inclubded
h0"tC1 een sIrOstIFp line. and pitch iflteflta filr a~tiie arS ia applica. si icon .mnd gallium arsenide I MPA 1-1 diode design and computer sinstls-/Ins I I1 lrsii Abi5 rstav Jisesrs lt ,s sol 10. p p ) 5 51 ;. ri ( Linln owcl lo~ r developnmentI. and ronopulse comnparator a nd phi se
Mar 19)88 shifter developmntn From 1981 to) I 5)5 he Aorked for TRW Flectron,.,X1It I I hoinas. 1) [ Fstidge. andi 0 Morris. -(,"nil 550T nc t and I )c.fense. Redondo Recach. (CA. a,, a section head in the Millimeter
g r a td w lilt a in11 erostrip patch,.4tift ril~ai & Rfp X 7 Kc. Tell~ W .iic Iechnislogiv I )s.partnrent. developing state of the ir I mill meterI.s e), % in tegi ai tcircuits anti subssstitts i nclutding tii ixt V(O % . Irian,
9 1 1) I 'rk ins. "Acts e iscrost np ci rculm pJLth .inicrinl a 'it 551ittcs'rs and nitpi i iers. modiula isrs. upi-cs nserter . muiishs.n ntipher,,,,,,J pp 110I 117. Mar 19s87 rcs.cii ers. ant transceilers lie joined the Fli-ctrical Fngifleeting Depart
11i ~S Ni sitng andl K 1) Stephan. "Stabilizatsionr ind poiwer comnbinriing men! sif I cxai A&M tLnsvcrsitv in August I 98 is an Asswiatc Prssfessoofp na rt srw is e ihdlatorS with an irpen rtnaii.in /9H antsis &a ps~roote t.d to Professor in 1 988 Ill, current interests it C in
/I I I / fil t 'i~ In, s tfnotta Sinip D~ig. pp 185 I 81 nm.rssw ac and millimeter-wave devices ind citmliit. rnscrimW ii c rpir. .s
1111J Rlancs.P SHal, ad 1 \X~xI.kltrmtp 4ieta Io~n inte ictsions. andi radar siostemsI I'l OcugIinos)I S liiia. an K) Pcrcgi n rsssi. c
4iiit D/is- r ('lang serves as the editor of lte Haiiko,iI A 's/ 'ii ,r,Ut ;,?,/
'121 k (Chiang ri a1/ . "I -hand iow noise integricd rcciser, //,/~ I )Isis l ( oniposisii i. tor be published bN Wi!-s & Suons IlIc is the eshitor of1,,vi,~ ~ ~ ~ ~ ~ ~ ~ ~ ~ ~~~~st %It Sit-i c hoiIe o TT 1 p1t '4 c hO Uu ad O pii u Tesiintdlog Letters titc has published iirc tisin
I )is sIt rtIM) iesr e o MT 1 p1(s-5.Vb~ technical papecrs in the are-as of rnicrsswa-e and nmirctcr-w,;\c
dc\ ices and circuitsji 11 R Adler. A studs sif Iscking phenomenain sosscill 1r. P'rot Il-
%As 4. pp 31 I 5 7. June 1946i 4l P )~ en et al . '.Mlillimeter- w se I MPA1'I oicrostrip oscillator,- in
/V' o [I.f 'sf77 S t it rimsie Nvrnp !lsg . pp 1394- 14115J K ( hang is a/ $ "i-band (75- 1144 Gill) mierostip euumponenits."?
11I1) frsus 's455rtsur'e Theisri Te's . vol MT-T-33. pp 1375 1382.
116tJ K 1) Stephanand S Y'oung. Mode stahbilitv of radiation-couplediliesn loc t ked toscillators for integrated phased arras,- 11.1i 11,ass'st %ftiH5,' Thcti let/i . vol 36. pp 921-924. Ma" 1988 Kenneth A. Ilummner jS'96) was born in Rento.
1171 1) Psolat and 1) H Schauhert. -Comparison of architectures for NV, on October 24. 1964 lie rcessed the B Smnhtsbc phased arras antennas.Ali MrowuicJ ,pp Y3- VA,. M ar and M S degrees in electrical engineering fromnsI t)%q Texas A&M Uniwirsits. College Station. TX lieRj 1) Jedlicka. MI T Poe and K P Carver. "Measured mutual is current1y pursuing the Phl 1) degree at Texsctpling between mierostrip antennas. IEF Tranis 4niestos A&M
Ir %s o sI AP-294. pp 147- 149. Jan 1991
Kai Chang lS'7S-M'7s SM'85) receised theB S E E degree from Natiosnal Taiwan Uniser-sit,,. Taipei. Taiw an. the MI S degree from the
State Unisersitv of New York at Slsrn Brook.
and the Ph D degree from the University, of Janw% L Klein received the Il S and MI S dc-Michigan. Ann Arbor. in 1970, 1972. and 1976, grees in electincal engsncenng from Texas A&NIrespee tivels University. College Station. TX. in 1986, ind
From 1972 to 197t6 he worked for the Mi- -1988. respeetives.crowave Solid-State Circuits (Group in the In June 1988. he joined Texas Instrumenr,Cooley Electronics tLaborators of the Uni-ersits -inc . Dallas. as a Miemo\%ase Design Engineerof Michigan as a research assistant From 1976 lie is currently responsible for the decsign and
I Q-S h \ was employed bs Shared Applications. Ann Arbor. where he development of monolithic microwase integrasted~sitrkCLd in etirrputer simulation of microwave circuits and mic:rowoave circuits for use in satellite and airborne phased.
ubes frsm 1978 to 1981. he was with the Electron Dynamics Division. arrav systems
Broadband Electronically Tunable Planar ActiveRadiating Elements and Spatial Power
Combiners Using Notch Antennas*Julio A. Navarro. Yone-H-ut ShU. Mid ii 01,1a1c. t-r11,. 11
Abstrac-A G.unn device has been integrated with two tspes plane tor efficient heat sinking. and an inexpensive methodof active planar notch antennas. The first It*ype uses a coplanar to create a icrowxave source. Howkever, the activc patchwa-,eguide UPCM'p resonator and a stepped-notch antenna " ith anen w xiie r ar% unrn~ s %ith w
bias tuning to achieve a bandwidth of 275 NIH7 centered at anen9.3e3iie e nro uig ags he(;Ii Aith a power output of 14.2 + 1.5 dBm. The second t~p cross-polarization lev els, and %%ide po'%ker outpu! de% ja-uses a (P%% resonator with a %aractor for frequencN tuning to tions. Futhermore. the structure has inheremtdtceceachieve a bandwidth of over 1.3 (;Hz centered at 9.6 Guli with for active millimeter-%% ave operation due to its small patcha power output of 14.5 + 0.8 dBm. Thii is equivalent to over dimensions, and it does not allo.% eas integration of other14%~ electronic tuning bandwidth. Both configurations exhibit soi-tedvcs.uhasheartr orwebna verN clean and stable output signal. A theoretical circuit model soi-tedvcsuhasheaatr.or ieanw4as developed to facilitate the design. The model agrees well electronic frequencN tuning. Previously. varactor tuningwAith experimental results. Injection-locking experiments on the has been accomplished using %Aaveguide and microstnpsecond configuration show a locking gain of 30 dB with a lock- Gunn oscillators I I i . 112]. but no attempt has been madeing bandwidth of 30 M1Hz at 10.2 GHz. Power combining ex- to inteerao. the varactor-tuned Gunn oscillator directltperiments of two varactor-tuned CPW active notch antenna it h nen.A lentv prahfrat~ celements in a broadside configuration have achieved well over beoteanen.A accmplied broc usin athe notch
70,combining efficienci throughout the wide tuning range. ment integration canbeacmlsd ui!thnohThe circuits hase advantages of small size, low cost, and eicel- antenna.lent performance. The notch antenna has man\ desirable characteristic.,
wvhich include broad impedance matchine bandwAidth andplanar A-ature. as wvell as good reproducibtlit\ . and ease
1. I NI H(DL'CtION of integration to passive and active devices. F'urthermore.
I C ONSIDERABLE etffort has been directed toward the the length of the notch can be increased to create a tra -
development of microwave and millimeter-wvave hv- elling,-svave antenna like the linear-tapered slot antennabrid and monolithic integrated circuits. Recent deeo- (LTSA) i i ', . fhe main design paranmeter of the notch an-
:-ctha\ . madc it possible to combhine active de,-ices tenna is the flare of the slot as gPisen b\ the exponent iall\ -
\thplanair antennas to create active radiating elements tapered Vivaldi antenna 1141. Other important work hasor ac'e quaisi-optical transmlitters. Due to the poAer lim- )ecn retnorted in the characteristics of the antenna imiped-tatlons kit aci e solid-state radiating element. quasi- ance an d radiation 1151j-1171. The flexibility of this tt pe
*~ptal nd patal pwercomtnig tchniueshav ben of antenna makes it ideal tot rtsn diverse apphlitons.Npkle: an16. The eneal posercobi m in techniquesbn An FET has been integrated \xi th a notch antenna 1181
ha\ e been re\ ie's ed in 161. The spatial power combining usng slotline ai~d CPW to create a 20 GHz receiver front-techniques create a single, coherent and high er-power en.Notratmphsben adtouehedv-\IL1al from man\ lowA-po\'ser radiating sources. Further- tages of the notch for an active quasi-optical transmitter.
more Taualor uas-opica powr cmhiingis ot im- This paper presents two novel conhgurations for the in-modin prolemsand llo's th com i egatton of a Gunn oscillator to a notch antenna:ted h\ ,ze or moigpolm n lo~ h ob- I I )Biased-tuned Gunn oscillator in the CPA' resonator
naton t'dgreternumber of active radiatine elements. uigasepdnthatna h oc sculdtThe mnicrostrip patch antenna has been used for an ac- usi cng tepped-notch aena.o Thech is coukplertic.planar. integrated. low-cost radiating element Il1, thendcnter o ah CPW renatrhc sphsc.prrI 7-I 10 1. T h e 'm ic ro stn p p a tc h a n te n n a p ro v id es a p 2n ic u a r a cto t e Gtn. s i l t r i a C N e o a oresonant structure for the Gunn diode to oscillate, a ground 2 aatrtndGn siltri P eoao
using a smooth-tapered notch antenna. The notch is cou-nu~q'i~ee~eJNo~mbe 2~ t~t. e~e Sercmcrpled to one end of' a CPWK resonator w hich is ph\ sicali\
Th rI' Ljppored in part hthe tU S Ar. Rcecarci (ML in colinear to the slotlineN Shu ~iih pvlon t.arnhda FVlecitonic Corporaton. (',rnc~a, IL 'The first configuration exhibits a clean, stable bias-
N'.r,' nl K Chang are "ith the tDepartment of Elericil FLngi tuned1 signal from 9.2 to 9 47 GHi wAith a poWeT oultput
_in Tej N&N Vn~ri\ oflg tt \-8332 14.2 + 1.5 dBm. How e% er. bias tuning creates a %A ide'Patet'l rvndinle djeviation in po\,er output and the resonator orientation
MXISi 94Si 92S01 WX 1Q')2 tt:FF
39
.7'- IFE-- TRANSACTIONS ON MI(CROWAVI'F THFOR'N AND I t-HNIQUhS. VOL 40. NO 2. FEBRUARY 1992
ma% introduce a strong cross-polarization component. For _p___
Resonatormost actiie antenna applications, a low-cost method o Ipro% iding a clean, stable signal with moderate and con- I - Stepped
C 'N' 0 Ntchsant power output over a wider electronic tuning band- i.Diode.% idth is needed. This can be accomplished by introducinga karactor in the oscillating circuit. The second configu-railont. "ith varactor tuning. exhibits a tuning bandwidth [ig I c'PW Stepped notch antenna clr ui conthguralion
from 8 9) and 10.2 GHz with an output power of 14.5 +0 S dBin. The spectral puritN and tuning range are corn- Frequency Powerparable to state-of-the-an results achieved in waveguide 9.5 ° -16
and microstrip oscillators. The theoretical tuning curve oobtained from a circuit model agrees fairly well with ex- 9.45, 14
perinental results. .Z 9.4. ',.XZ " 12Injection-locking experiments were also conducted a . * E
showk ine a locking gain of (30 dB with a locking bandwidth .10 .9.3.of 30 %IH, at 10.2 GHz. Power combining experiments 9." % .8
of t o varactor-tuned active notch antenna elements, in a 9 9.25- 0broadside configuration at a distance of 8 mm( X/4 at " 9.• 6 ,
"C 9.206 6 GHz) apart, have achieved well over 70% combining .4 0
etficiencv throughout the tuning range with a maximum F 9.15.-
oi 129.2 '/ at 97 GHz. To the best of the authors' knowl- 9.1-edge. this is the first varactor tunable power combiner ever 9.051 5 0reported. 10 11 12 13 14 15 16 17 18
The circuits offer man, advantages of low cost. sim- Bias Voltage (Vofts)
plicit. . small size. wide electronic tuning range, good sta- Fig 2 Frequency and po%,,er output %,ersu, bias soltage o the active CPW'bilt . and excellent performance. Although Gunn and stepped-noich antenna
\aracto. diodes were used here. other active devices canalso he used. Due to their planar nature. these circuits are mized using a transmission line equivalent circuit model.amenable to monolithic implementation and offer many The lengths of the transformer sections were optimizedapplications in radar. communications, and electronic for minimum return-loss throughout X-band. The circuitx\artare s,stems. was fabricated on a 60 mil thick, RT-Duroid 5870 sub-
strate. To test the passive circuit, an SMA connector was
11 Tiu A(-riF CPW STILPPED-NOTCH A.;TENNA soldered on to the CPW resonator and the measurementswere performed on an HP-8510 Network Analyzer. The
SI shows the active CPW stepped-notch antenna stepped-notch antenna gain was measured at 9.3 and 9.6configurition. The circuit consists of a stepped-notch an- GHz and found to be 7. I and 7.7 dBi. respectively. In-tenna coupled to a CPW resonator via slotline. A Gunn tegration of an active device may change the antenna ef-diode is placed in a heat-sink at the open terminals of the ficiencv and introduce degenerate modes at the oscillatingresonator. The notch is formed by many step transformers frequency. However. one can use this method to approx-k hich match the slotline impedance to free space. imate the oscillator power output. The gain measurements
The resonator is the essential design element for im- were used to calculate the oscillator power output with thepro'ed oscillation and stability. Considering the planar Friis transmission equation:nature. Q-factor. and ease of integration with active de-,ices. CPW was chosen for the resonator. The CPW slots p, p.(47R - (l(Iwere designed to be 0.3 mm with a 3.5 mm separation. X G, G_This arrangement provides a 50 UI characteristic imped- whereance and mates well with the 3.5 mm cap of the Gunn P, = Power received.diode The length of the resonator is about 0.5X at 10GHz A DC block was incorporated at the shorted end for P, = Power transmitted from the active notch an-
biasing purposes. The circuit is mounted on an L-shaped tenna.
metal block which serves as a heat-sink and holder. R = Wavelength of operation.The notch antenna design was accomplished by using G, = Gain of the transmit antenna.
cascading slotlines which act as impedance transformers = Gain of the ransit antenna.from the coupling point to free space. The low dielectricconstant of 2.3 allows efficient antenna radiation. The in- A Gunn device from M/A COM was integrated withput impedance of the notch was matched to the resonator the stepped-notch antenna. This Gunn diode producesat the coupling point. 72 mW in an optimized waveguide circuit. The bias volt-
.Matching from free space to the resonator was opti- age vs. frequency and power output is shown in Fig. 2.
40
s XK R\, "li "W H\\I l Wi l (N \, \if I 5 4 1 r%\1\, FlI %\I i\1,
Ihe 3 dB bias-tuning bandwidth was 275 MHz centered F -VARACTORat 9.33 (Hz with a maximum power outpUt of 37.5 mWat 9 328 (;Hi.
I1 lot 1 V\Rt( 1)k-11 N (PNV ,-\( I\i NoiII
t-L. 3tal sho\s the varactor-tunable CPW active notch CPW RESONATOR
antenna configuration. The circuit consists of a notch an-iennla integrated with a %aractor-tuned CPW resonator. .11
[hc notch antenna couple., to the resonator via slotline. Z4 sections of.\ Gunn and a varactor diode are placed at either end of K t- LA -1 slotline
the CPW resonator. This arrangement provides strong 0:1Z2 Siotlnecoupling between the Gunn diode and the taractor diode Vc 25 28 to Z29 RI SPACt
for increased tuning bandwidth. Gunn diode 377 OHMS
The CPW resonator allows multiple dc-biased devices 21
to be integrated due to its inherent dc blocks used for sep- -arate biasing. The input impedance of the antenna wasmatched at the coupling point with the resonator and the Z= i60 Ohm.. L = 0.845 mm
12 180 Ohms, 2 3.529 mmflare of the notch antenna was modified from the stepped Z ai, ohms. tA 3.700 mmZ.4 110ri Oh
, ,, - LA 7.200 m~m
to a smooth taper for improved bandwidth performance. 5 = 122 Ohm. L5 =2540 mm
A theoretical model was developed to facilitate the de- ,h)sign. The equivalent circuit can be represented as shown Fig 3 The aractor-tunable CPW nolt.h antenna iii (ir.uit onriour,'
in Fig. 3(b). The modeling of the CPW junction is based tion h) Equialent circuit
on reference 119). Neglecting the junction discontinuityeffects, the transformer ratio n is set equal to I. The flare R.- -8.00 Ohmstif the notch antenna was approximated bs 24 sections of C. - 1.05 pFLslotlines of different widths and impedances. The equiv- C, - 0.25 pF Cpalent circuits of the Gunn and varactor diodes used for L., 0.30 nHcalculations are shown in Fig. 4. The diode parasitics for Rg Cthe Gunn and varactor are given by the vendor. The fo--lowing conditions were used to determine the oscillating titfrequencies at various varactor bias levels:
!Re (Z,,,,)J - ]Re (Z.,,,,,)1 (2)
Im tZi,,4 Im (Z,...) = 0 (3) C.-,, 0.85 to 1.60 pF Ls
%khere Re (Z,,,,) is assumed to be -8 1 1201 and the os- R, -0.05 p Cpcillations occur when condition I is satisfied. The oscil- R,., 2.81 Ohmslating frequencies are found from condition 2. L,- 0.30 nH
The overall dimensions of the circuit board are I x 2inches. The circuit was etched on a 60 mil thick RT-Duroid 5870 substrate. To test the passive circuit, a coax- Fig 4 Deice Equi alent C.rc'um, j) Gunn diodc i" % a.r Jdc
ial connection was soldered on to the notch and measuredon an HP-8510 Network Analyzer. The measured SWRskas less than 2: 1 throughout the X-band range (8-12.4 tuning curve obtained for varying varactor bias levels fromGHz). The passive notch was then tested in an anechoic 0 to 30 V. Experimental results are shown in the samechamber for the field patterns and relative gain. The rel- figure for comparison. Considering the large deviation otalive gain (G,,) of the passive notch antenna measured at active device modelling parameters. the theoretical model9.0. 9.6. and 10.2 GHz were 8.0. 8.2. and 9.0 dBi. re- agrees fairly well with the experimental results. A fre-spectively. The gain was later used in ( 1) to determine the quency tuning range of 8.9 to 10.2 GHz was achieved foractise notch output power. varactor voltages of 0 to 30 V. This is equivalent to over
A Gunn and varactor diode from M/A COM were in- 14% electronic tuning bandwidth. There are no modetegrated into the resonator and coupled to the notch an- jumps and the signal spectrum remains clean and ve- sta-tenna via slotline. This Gunn diode (MA49106) produces ble. with an output power variation of +0.8 dBm80 mW in an optimized waveguide circuit while the var- throughout the frequency tuning range. The spectrum ofactor (MA46602F) is rated for 1.6 pF at 0 V. Theoretical the received signal from the varactor-tunable notch an-results were calculated using the circuit model shown in tenna is comparable to the spectrum of the active stepped-Fig. 3(b) and (2) and (3). Fig. 5 shows the theoretical notch. The E and H-field patterns as well as the cros,,-
41
IEEE rRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES. VOL 40. NO 2 FEBRUARYt 199,
10.4 olanzation patterns are shown in Fig, 6 for the varactor-110-2 Theory tunable active notch antenna at 10.2 GHz. The back ra-
~ - -- - - -diation remains belo%% - 10) dB throughout the tuning
t2 9.6'
8.8In ection-locking experiments with an externial HP-8 6 8690B Sweep Oscillator source were pert-Ormed to deter-6%.4; ~ o mine the locking-gain and lock ing -bandw idth of' the ac-
Vurector Voluve (Volts) tive notch antenna configuration. The test measurementset-up is, shown in Fig-. 7. Equation (I) %%as used to de-
10.4' --- -- '1-T8 termiine the P, from P. and P, -rorii P,. Injection-lockine,10.2- 17expenments were perforniled throughout the electronic
10 - :- 16 tuning range. TIhe locking-gain (P,, /P, )vs. locking, band-S9.8 a width (AF) results are shown in Fig. 8 and the are corni-
96 + '14 parable to previous patch antenna experiments Il11. A0 I locking gain of 30 dB with a locking bandwidth ot 3(0
94. - '3kZ MHz was obtained at 10.2 GHz. The Q-factor of the cir-
9-2 0 '2 cuit is found to be 2 1.5 and can be calculated with 12119o 1
0 5 10 15 20o 2_5 30 2F, P
(hi The locking gain in dB is defined aslg5 Aciic sajractor-Iunahke CPwA notch antenna results ia The
theoretical and experimental oaractor tuning rcquenc versus voltage hb) P"The experitnental poiher output of the active varactor-turiahle CPw noich G,= 10 log - (5)anicnn n,
where
wow -6oov Q, = External Q-factor.E F%"F, = Operating frequency.
.24' AF = lnJection-lockine bandwidth..P, =Injection-lock signal power.
-P,, Free-running oscillator rpower.
v 'Quasi-optical combiners using Fabrv-Perot resonator.,cx .32 E Pt- c-1and spatial power combiners have the potential ol- corni-34. bining many solid-state des ices at millimeter-w.ave fre-
38 01 0 ~ ~ 7 quencies. To demonstrate the feasibiliy of the spatial38go. ~ - power combiner. two notch antennas Were set up ina-90-0070 60 50 0 30-0 -0 010 0 0 4 5060 0 go9
"ie (09rnws( broadside array at 8 mm separation. To achieve etlicient(d) power-combining, the active notch antenna elements must
.10- ------ - - injection-lock to each other through mutual coupling.
m pu- Power combining experiments of two Inject ion -locked.-is- varactor-tuned active notch antennas were conducted
I throughout the electronic tuning range at 100 MHz incre-m-20 ments. The power combining efficiency is detined b\0-25
Efficienc\ IX'
-36. P.- c-where
P, Power of active notch #1 (Use G,,,= G_,_0W-5 0 3 20 -t0O 10 20 3040 50OSo 70 sogo G,, is the antenna gain of a single passive
Angle(Degries)notch antenna).
ic Vield patterns (it the aclixe saractor-tunable CPW notch antennaP.= owrfacienth#(UeG, GLat '10 2 6111 a) f, plane pattern and cross-polarization measurements (hi Pcombifler = Power of injection-locked. powver-coin-H- plane patterno and cross-polariiation measurements bined signal (Use G,,, = 2G,,).
42
NA% ARRO et al BROA()BAND ELECTRONICAI.IY TUNABLE RADIATING FIMENTS
I Spectrum AnalyzerI
I P ,
1Aci. Notch Coupler Power MeteriActive Notch G
SPower R lSupply I-] [ P -8650
Horn Variable SweepP,I Attenuator Oscillator
Fig. 7 The measuremcnt set-up for injectiton-locking expcrimcnts using an external source.
50 quencies can be attributed to improved impedance match-
5- ing in two mutually-coupled oscillators as compared to asingle oscillator which is not fully optimized. Similar re-
* suits have been reported by another power combiner [22].Z The H-plane field pattern and cross-polarization measure-6 3o! ments at 9.6 GHz are shown in Fig. 9 for the combiner.
S25, , The 3 dB beamwidth of the array was 53 degrees com-pared to 78 degrees for a single element. To the best of
S I our knowledge. these results represent the first widebandis, . varactor-tunable power combiner reported in literature.
-~10'
5.
3 4 50 6 ,0 V. CONCLUSION
25 30 34045055 60 6 07Locking Gain(dB) The Gunn device has been integrated with a stepped-
Fig. 8 The injection locking-gain versus locking-bandwidth at 10.2 GHz notch antenna using a CPW resonator. The configurationwas modified to incorporate a varactor diode and form avaractor-turned active antenna element. Over 14% elec-
" i.,. ,ronic tuning range was achieved with a fairly constantpower output. Two of these varactor-tuned active antenna
I elements were successfully injection-locked to each othero' 0i via mutual coupling and power combined throughout the
wide electronic tuning range with over 70% combining
.~('\efficiency.l H , -C-,o These circuits offer a small, simple, lightweight. low-
cost, reproducible, and truly planar active wideband tun-/ able source for many microwave applications. Using these
.2, ___ elements in planar arrays with injection-locking and power-9 0 8 0 7 0 6 0 -5 0 4 0 3 0 -2 0 -4 0 0 1 0 2 0 i 0 4 0 5 0 W 0 7 0 8 0 9 0
Anglo , d"10) combining techniques will enable higher power levels.The wide varactor tuning range should prove useful for
Fig. 9. The H-plane and cross-polarization meausurements of Iwo mu-
tually -coupled injection-locked, power combined active notch antennas at frequency modulated communication links, radar. and9 6 GHz. electronic warfare applications. The circuits are amenable
to monolithic circuit integration for mass production and
All power calculations are calculated from (1). For the should have many applications in frequency-agile trans-
P, and P, calculations, the gain of a single passive an- mitters at microwave and millimeter-wave bands.
tenna was used for the transmitting active antenna gainGo,. For the Pcombier calculations, the array gain was as-sumed to have twice the gain of a single passive antenna. ACKNOWLEDGMENT
The power combining efficiencies measured at 9.4. 9.7.and 10 GHz were 90.0. 129.2. and 75.0%. respectively. The authors would like to thank Mr. R. C. Waits forThe combining efficiency of over 100% at certain fre- his instruction and use of the fabrication facilities. The
43
A 1IF [-1 TRANSACTIONS ON NutCktWAvi.t THEOIhY AND TECHNIQIF~S. V'Of. 40. NO 2. FEBRUARY 1992
authors would also like to thank Mr. J. C. NcClear\ ajnd Julio A. Navarro %Aas horn in Cordoba. Argen-Mr .A u mrfor marNs aluable discussions and tina (in Norsember 13. 19t-4 He receised the
Mr. . A Hum erB S 1: F and Nil S F 17 front Texas A&M Unisu1l gestiofls. - scrsili in 198S and 0I90. respct:isel%
lie "a% a Co op Student "iii General D) narris, -ort VS tih f rot Mali 1985 toi Februars I 90
Rt~~~A Gene tI tM Crail 1) lnaics, he ixorked in As ionij S-,Ri I I Ri I s tins Iestimn Ads anced Technolog,, and S %terns
Engineering. Limtiters and Intelligence. AntennaI IiK (ang IsSi ummr, ad 11. lei. Ispeirtintsut tii'c ___Ss sietits anid Radit (ross Set ion Research He i,11 ('ang K HUnme, ad J1, lei. I\petinnt,',f tileAOLpresetitk a learching anid Research Assistant at
ionlokin i actr anena eetieiisfiraiisephsedatas'ai~l I exas .'S&N As a Research Assistant, he ha. deindaractor tunablesputa1
posser coitriiners.' Il-lb Jri,ii Aioi ,iir ?hios -h . etidtfire radiating clertients. sss litchahle and tunable untplanar tilter, and \ar\,,I ',. no _. pp IWiS 10S84. Juil\ 1080' actor tunable (Gunn V'CO, lie has also deseloped KIsa-and aperture-cou
21R Ai ikrk and R C (iirtiptiiii QOUasi optical ptisse cotihbi!t pled circular path antenna, lit a NASA ILessis lesas Instrumient, priricciusig iutui Iss 5 rrrii/d ts~ t ~iir arass. II I 1 an 'iH ~ IHis current research interests include high-Q1 uniplanar resitnatirs and quasi-
itliii 1hi,- ,i h/ "i Il it" ' p lp lif/Si Iii/ Junie 1001Q optical prisser cotmbitners Hec horlds an NSI- Ugne ingelloisship andr ,~/ Hf topos t- R %I Wklc 11 %1 Kim. K A Pottev Lit, 1 H)I curreni pursuing the Ph 1) deeree under the direstion it Dr Kai Chan-,Rutledge H*flt grid itscillaitirs. I'fff Iram Alt, riii lb- t
1,i , r b ll ',i noi c pp 225 2/i ; ti % lJT Iu)
1 I kk ',linik Quasi otiical pitssr ciririinitig it ohrld-talc milli longhui Shu "as hiirti itt Jiangsvu. China. o
mteter ssasc sourcs Ifff 7r~oii A/it riliti ItiiiFi 11 1it! . %klI April 24. 11.57 He rceisedl (tic Hf S and %11 SN111H 14. pp 2 1 2 '9 Febh 98it degrees in electrical engvineering troim the Nantin,
SI ) oung and Is 1) Stephan. ' Stabiliation and potter coiinine ot Institute oft rechnilg% - Nanuing. China in 1splanar tticriroasc o scil lators s ith an ripen resornattir i n lVS 7 If! I1 n 95 epcie\
Af/7 ht AtcratAtm Di: .pp 85-ISSFront Mlarch 1985 to Juk 1489. hie Aoirked tot!JiIrs C hang and C Sun. - -Millimetier \4ase prissr comitbining tech the Deparitment itt Radio Engineering it Nanine_
niq4ues. If.ff Iran, 'i/it Ciii i Acr let J . ol M1TT 31. nii Institute tit Techniilog icurrent name. Southeast2 P pp It- if e I'?, Untisersitsii as a Lecture' From. Siugu' IL1s i
SI homas, D) L F-udge. and G Morris. 'Gunn source integrated £ Jul\ 1990. he ss as iltith the Department oif Ele,ssit/i a inicrostrip patch.' Atiiriisave raid RI, sri1
24. pp 87 -8). J & tric al Engineering nI Texas A&Ml lUntsersits a, ~IIets I Qs S Research Associate In August 19911. he joined Epsilon Lambda Electron
IS1 1 0i Perkin,. -Actise microstrip circular patch antenna.' Altirii ics Corporation. Genesa. IL. as a NMicrrisase Design Engineer His current,iiii J . o iW3t. pip I I10- 1 17, Mar 11)87 interests are in tticrossasc and ntillirteter-\4as e des ices, circuits and sur
ilI I Hi rkeland and T Itoh. 'Planar FET oscillators using perio0dic [i i % steinsrilstrip patch antennas. l/fft/ Iraiiiritiiif im on tiirtirrii Ihc-un . ir Shu has published 210 technical papers fit tticrtrssase and mil micee
it/e, hniqtii. sol l1. nit 8. pp l2-12 ;6, 'Sue 1'4S) ssase areasIill K C'tinriev. K A Huttitter. and G~ Gipalak ris/inan Sitseradiating
cicie en usinti FE! source integ rated "sit/i itticrirsirip path antenna,..~
V-11,117 -- : . sitl 24. nit 21. pp 134-- 1 34S. Oct 13. l9S8 Kai (Chang 15*75-N176-SNM'85-F*9I i recei,;:,1 11 S Jiishi. WAide-hand saractitrtuned X-Hand Gunn Oscillators in the R.S E EV degree 11rom the Nalional Tamrssi
tull height saseud ast .Itff! /rri AM~ol~w ii iier Ii h /inisversit . Taipei. Taiman. the NI S. degree frommit MT1 21 , nii A. pp 13' 139, Mar 19" 1'f the State Linis-ersits Nof Ness York at Sions' Brootk.
12 1 1) Rubiti 'S aractisriuned millinteier-ssase %11C oiscillatoir.' Il-F! and the Ph D degree fromn the tiniserstt \ 'i/ 'it Ali, [(ir-Ir.n -(I/ . sol I %IT -"12.. no I I. pp, Stiti-hot. MnIichigan- Ann A'rboir. in IQ0. 110-:. and 10>_NH\ i ', respectis el
I jK s) 'tics es, I 1. Isorzcnitisski. Y S Kurt. L -. 1.killberg Fromt 1072 ito 19lth. he "sorked lor the Mi, ',.n.1-1 Jiihanson Thle taperedt slut antenna a ness integrate '-I d c itasie Solid- State Circuits Grioup. ('iiie~ Fie,
tiill itlttltimeter-ssase applications." 1IF!! T'rra lf rotittui tritnics Laboratorii of the Ltniserstl of Mfichig.r.1 t s, " ri 2 ppF,0 It41'I Febt 101 as a Rcsear..h A'Ssistatti I To utu t> to 10 -
-~ ~ ~~~h -'tI tisr %eSis aldi ASerial. - -it /nt,, vti 1 at,,p anur~o %h, Pt, %,as ettploi\ed h% Shared Applications. Inc .Anti Arbiir. %here he ss ur/.c,'it lo j'gr 1ttIK It)0") pp IMi I0/S in coittputer sittulatiiin ~o mticrissa% c circuits and tmicroi"sae tubes Fr.''
1< Jraa allis - L) 11 'ichaubecrt arnd I) MI Pitar, 'itrals sis (it the IQ7'M t 198 1. he %ssirked ftir the Electritt D% rtantcs Di s isin. Hughes ASi'I r~irtsserse Lie~troiittu Mode Linearts , a pered Slot Atienna. craft Coiipn\. Tirrance. CA , ssthere he Asas insolsed in the research an.'Ra':, tilts- oI 21,- nit 5. pp '1) h05/4. Sept 0,1l IIS8 deselopment (t irtlltmeter-ssas-e sotlid-state des ices and circuits. puss er
I Il I ] iv etnd F /hibit. 'Analstsi tit \ is aldi antennas,' itt Proit cittrbinerN. oiscillatoirs and transnrittmr Fromt 181I ti 1985. he ss rrked i-II! /iwi un If at t-ttuiii n p Fft Iit Pt I c i I .05 pp the FIA E lectritnics and Detetnse. Redondot Beach. CA . as a Sec:t,.-
Wtis~tt ardI)1 chuet aaceurt ed. des eliping state -if tt c O arl nil linieter sAas e integrated circuits anuK Jwj-mi in 1)H chjhcr 'Uhaacr tipedanse Oft . subs%. terns including mrixers. SC/I N. transnritters. a nipliuters. troiurdu,
it to ine- tn to,., perntiiis substratesc Iff.[ rruai .'itr,- fotrs uptnserters, sssitches. miultipliers. relceisers. and transceisers He-;i lht- tnt,, t it , sol MITT 14, pp W/51 Wif12. ASug 1050I tuined the Electrical Engineerintg [)epartmen t i Tesas A& I Urisersits irt kcult i/i Planar integrated 20 I (H. reicici e in slut line and coi August 1985 as an Associate Professor and "J' aspromoi'd 10 a Priofessor tr
plantar s as gu idc technique. -- i/i nrrirt, an rrtl pti ra/le ttiis 1088 In Januarn I lQ/I. he "sas appotinted 1- S~ stetrt Endrissed PriulessirI ,:;, ts i I,2 nit I I pip 4(04- 11/i Ntis 1980 (it Electrical Engineering His current interests are in micrirsase and mt.
I1I01 K C G;upta. R Garg., and I J Bahl. Muicrostrip Lines and Slotilnes, litreter-ssase devices and circuits. micrirssase integrated circuits. mien"tsstuttil %1 A -iech Horuse. 1l)"J. p 2041 ssase optical interacitions. and antennas
't11 Ks (hank - K Loiuie A5 J (iroit. R S Tahfini. N1I J Ninar. G Ii ir Chang sensed as the editor it the Itiurs irlutrt' Handbootk it N1Has ashibara. and (' Sin. V% band hiss niu intemraed circuit re crirssase and optical Coimponents - published h\~ John WA ies & Sons. In,,ci rt ift Iran l iuitt -iriotrt-i Thetint% it~ .t %o s r IT ;1. pp in I 950 and 199(0 He is the edittir irl the Nfitrrissas e and Optical Tecti1-4) 1-. Feb lo4st nilog\ Letters, andi the ilesc Booik Series in Micrimias and Optical Er-
I I1 R -Adler A - studs tit ticking phenoimetta in tiscillatrirs., in Prmt gineering He has published riser I 10 technical papers and sescral bur1,IRI \,I 14 pp t I 3 57. June 114t, chapters in the areas (it micrrisase and tnllimeter-ssase des ices and cit
- - \ \Ilitlit andi T huh. - A periodic planar Giunn driode prissr cuits Oft Chang receused the Special Achiceenent Aard Itrim TRW inItt/st net illattir - /fft , rn AitC rut r I/tiiri /t-t/ , sri 18, - 04. the Hallihutun Ptrofessorr *Sssard in 149 and the Distinguishedi
PP it SI(- Jan 1Q0/ Ieacltung 'Sisard in 1080 tromtu the I enas A&M Uitersit.\
44
Appendix 3[tl ~ ~ ~ ~ ~ ~ ~ ~ ~ ~ ~ ~ ~ ~ t , \R \.
I )xt" .1 ' \ ~ l)h \,' 1 ( .)[~ 1 , \(t il , i ,.i \j , I "j( \ '1 r),'.
Theoretical and Experimental Development of 10 and35 GHz Rectennas
Tae-Whan Yoo and Kai Chang, Fellow. IEEE
Abstract-A 35 GHz rectenna has been developed with 39% output ratio 171. The typical power conversion efficiencyconversion efficiency. The rectenna uses a microstrip dipole an- achieved is 85% at 2.45 GHz.tenna and a commercially available mixer diode. Over 60% The rectifying process is a nonlinear process. It is dif-conversion efficiency was demonstrated using this diode at 10 Thcult tfigrout ho t rcnnar rc It is izGHz. A theoretical analysis was derived to predict the perform- ficult to figure out how the rectenna circuit is optimizedance of the rectenna. The analysis is a useful tool for device for the maximum conversion efficiency. There were sev-and circuit design. The theoretical and experimental results eral theoretical analyses to solve this problem. Theseshould have many applications in microwave power transmis- analyses can be divided into two methods. One is to di-sion and detection. rectly simulate the rectenna circuit in time domain 181.
The other is to find a closed-form equation which can ex-plain the relationship between diode parameters and the
T HE AVAILABILITY of power for use in space is a conversion efficiency 191. [101. A computer simulation inkey requirement for future space activities. The cur- [8] sucessfully showed the conversion efficiency lager than
rent method of power generation with solar panels or bat- 80% at 2.5 GHz. A mathematical model in 19] was de-teries on board of each satellite has many difficulties in rived to relate the power conversion efficiency with thepackaging the power system. unfolding them in space. and conduction period of diode. A fixed relationship betweenachieving a high output power. Power becomes a limiting the dc output voltage and the RF input voltage was as-factor in the design of most systems. To overcome these sumed and the conduction period was unknown. A closed-problems, a potential method may be to deploy a Utility form for efficiency in 1101 was derived by expanding thePower Satellite (UPS) in space. The UPS will generate current-voltage (I-V) equation of diode with the diodepower using solar, nuclear or other techniques. The power voltage. But this is valid only for a high-efficiency rec-vould then be converted into microwave or laser beams, tenna since short turn-on period was assumed.transmitted through a free space and converted back into All these studies were concentrated on 2.45 GHz. Con-a useful form of energy for users. sidering that the frequencies below 3 GHz are not strongly
The laser beam has an advantage of the small beam di- attenuated by the atmosphere even under a severe weathererency. However, the efficiencies in generating the laser condition [11], 2.45 GHz is thought of as a proper fre-
beam and converting it back into electrical energy are low quency for the application of power transmission betweencompared with those of microwave. Therefore, the de- ground-to-ground. ground-to-space. and space-to-ground.
elopment of power transmission system by microwave However, for the space-to-space application the operatingheamsr is attracting new attention in space application, frequency can be increased to allow power transmission
The rectenna (rectifier + antenna) which receives and over much longer distances with the smaller antenna andconverts microwave power into dc power is a key element rectenna. Although the efficiencies of rectenna and gen-of this power transmission system. Since the rectenna was erator are low at 35 GHz, the advantages of size reductionin'ented [I I it has been improved through studies [21, [31 and longer transmission distance overcome the low deviceand used for various applications such as the microwave conversion efficiencies. The overall system efficiency ispovered helicopter [4], the receiving array for Solar thus higher at 35 GHz than 2.45 GHz for long distancePower Satellite [51. and the experiment on the microwave transmission using the same size of antenna and rectennapo,%ered aircraft which was recently conducted by Can- [121.ada under the project name of SHARP (Stationary High In this paper we report theoretical analysis and experi-Altitude Relay Platform) [61. As a result, the structure of mental results on 10 and 35 GHz rectenna. A closed-formrectenna has been evolved from a bulky bar-type to a equation for the conversion efficiency has been derived toplanar thin-film type greatly reducing the weight to power analyze the diode for the high frequency rectenna. To de-
Manurcnpt received May 22. 1991 revised October 24. 1991. This work rive the closed-form equation we assumed that the effectsa .upported in part by the NASA Center for Space Power and the Army of harmonics higher than or equal to the second order were
R.e,ejrch Office small and the forward voltage drop of the diode didn'tThe authors are with the Department of Electincal Engineenng. Texas
A&M University. College Station. TX 77843-3128. change during the turn-on period. The closed-form equa-IEEE Log Number 9107446. tion was validated by comparing it with a nonlinear circuit
0018-9480/92503 00 1992 IEEE
45
I , IF-. iRANSA('IIONS ()N MICRO"V,%I- 1H-()RY AN[) ri-HNIQUIES VOL .1() N) b. JlrNF I992
simulation computer program called LIBRA developed by RectfyinglEsof Inc. The maximum efficiency limited by the par- circitasi"ic series resistance and the junction capacitance of adiode has been calculated using the closed-form equation. Antenna Low PassIt ,,as possible to interpret the cutoff frequency of a diode ,as a parameter to limit the highest operating frequency ofthe diode in the power conversion circuit as a result of Fig 1 Block diagram ul a recrenna circuri.this calculation. This result of closed-form equation willbhe 10eful to select or design a diode for rectenna at any 1arbitrar- frequency. I Higher Order
A 35 GHz dipole rectenna has been designed with the Harmonics Effect
closed-fomi equation and LIBRA which uses a HarmonicBalance method to solve a nonlinear circuit. The overallconversion efficiency of rectenna was measured using theka\cuidce array simulator. The dc power was 39% of the Diode Maximuminput power of 51 mW and the reflected power was 12% Efficiency Curve 1
of the incident power. The incident power includes boththe input power to the rectenna diodes and the reflected _ _ _ _ _
powzer. The power conversion efficiency of the diode at vI1) GHz was measured by sending the power through a 4 RL Input Powercoaial line. The highest efficiency was measured as 60%, Fig. 2 General relationship between microwave to dc power conversion
which was consistent with the simulation result with efficiency and inpui power.
LIBRA. The devices used were commerically availablemixer diodes which are not optimized for this application.It is believed that higher efficiency could be achieved with The efficiency sharply decreases as the voltage swing at
optimited devices. the diode exceeds the breakdown voltage (Vb,) of thediode. The critical input power where the breakdowneffect becomes dominant is expressed as V2,/4RL in
II CLOiSED-FORM EQtATION FOR THE CONVERSION Fig. 2.EFFICIENCY The three parameters of diode such as Vh, zero-biased
junction capacitance (C,). and series resistance (Rs) de-.4. Basic Circuit Structure of Rectenna termine the power conversion efficiency of rectenna.
The basic structure of a rectenna is shown in Fig. 1. These parameters are related with each other due to theThe Iov, pass filter inserted between the antenna and rec- diode material properties 1131. The breakdown voltage oftif. ns circuit is designed so that the fundamental fre- a diode used for a 2.45 GHz rectenna is on the order ofquenc\ can be passed and a significant portion of the 60 V. The C, and Rs with this breakdown voltage arehi,01er order harmonics generated from the nonlinear rec- several pF and less than I Q2 respectively which are smalltift \im circuit be rejected back to the rectifying circuit. enough to operate at 2.45 GHz. The breakdown voltageThe rectifying circuit consists of a single diode shunt-con- is much larger than the junction voltage of the diode whichtheed across the transmission lines, is about 0.8 V. Therefore the efficiency of the efficiency
The basic principle of the microwave power conversion vs input power curve in Fig. 2 can increase nearly to 100%b\ th rectifying circuit is analogous to a diode clamping before reaching at the region where the breakdown effectcircuit or a large signal mixer at microwave frequencies. is dominant. For 35 GHz rectenna. C,, should be reducedThe power conversion efficiency is maximized by sub- to the order of 0.1 pF. But this reduction results in thestantially confining all the higher order harmonics be- increase of Rs and the decrease of Vbr. The typical valuetween the low pass filter and the dc pass filter, using an of Vs, of a Ka-band mixer diode is 10 V with Rs of 4-8 0efficient diode and matching the diode's input impedance and C,, of 0.1 pF. The efficiency would be low with thisto antenna's input impedance. small value of V, because the junction voltage effect is
The power conversion efficiency of a diode changes as still not negligible even at the maximum efficiency pointthe operating power level changes. The power conversion of the efficiency vs input power curve as illustrated in Fig.etficiencN of a rectenna generally changes with the input 2. Consequently, it is important to have a good trade-offpoer as shown in Fig. 2. V1 , Vt,. and R, are the forward between the diode parameters in designing a diode. Thevoltage drop (junction voltage), the breakdown voltage of closed-form equation derived in the next subsection willthe diode, and the dc load resistance of the rectenna, re- be useful to design a diode for a high frequency rectenna.spectiel,.x The efficiency is small in the low power re-gion because the voltage swing at the diode is below or B. Derivation of Closed-form Equation for Conversioncomparable with the forward voltage drop of the diode. Efficienc'The efficiency increases as the power increases and levels The equivalent circuit of the Schottky diode is shownoff i, ith the generation of strong higher order harmonics. in Fig. 3. The diode consists of a series resistor (Rs). a
46
YtO) AND (IAN(I D) VIOPMFNr OF II AN) S u RE'CTE.NNAS
A cording toR- V 4 +---
R, Vi (3)
B where r = R,/RL and R is the dc load resitance as shownV ,R L DC( LoDMV R13 D in Fig. 3. Vd.Dc introduced here is the average value of
R Vd CJ T the waveform V4 , and Vm in (2) is the dc component of
the waveform Vd in turn-off period. Therefore, V. DC isderived by taking an average of Vd over one full period as
- - ------------
C 0ffSDC= A) I + -- sin0f - V - 4)
Fig 3 Equivalent circuit if a Ka-band beamlead Schottky diode used for where 0of is the phase angle where the diode is turned offthe derivation of a closed-form equation. R, and C, model the intnnstc junc-tion for the diode. R5 is the parasitic series resistance of the diode and R, as shown in Fig. 4. The phase variable 0 is defined asis the dc load. 0 = wt - 0. Since the switching of the diode occurs when
V4 is equal to the forward voltage drop V of the diode,
nonlinear resistor (R,) described by the I-V relationship 8 is calculated by
of a diode at dc, and a nonlinear junction capacitance (C).C V1 + VO(The reactive parasitic elements of the physical diode are Vd5excluded from this model because they can be zuned outwithout affecting the rectifying efficiency. The closed- On the other hand, the equation for the current flowingform expressions for the rectifying efficiency and the ef- through Rs is written as follows when the diode is off:fective microwave impedance of the diode are derived d(C, Vd)
with the following assumptions: Rs d V - Vd . (6)
1) The voltage V applied between node A and C con-sists of the dc term and the fundamental frequency Since C is a monotonic increasing funtion of V4, C, canonly. be expanded as follows:
2) The current due to the junction capacitance is neg- C =
ligible when the diode is forward biased. Co + C cos (cat - 4)3) The forward voltage drop across the intrinsic diode + C2 cos (2wt - 20) + . (7)
junction is constant during the forward-bias period. Substituting C, into (6) with a Fourier series of (7) and
The first assumption is reasonable considering that the neglecting the terms higher than the second harmonic, theI magnitude of the higher order harmonics is usually small equation becomescompared to those of the dc and fundamental components.The second assumption is based on the fact that the change tRs(C1 Vo - Co1'41 ) sin (ot -4)of V, in Fig. 3 is small during the forward-bias period. Va - V0 + (V1 cos 4) - VdI) cos (WI -)
Under these assumpt~ons, the voltage waveform of Vand V, can be expressed as follows: - V, sin 4 sin (w t - d)). (8)
V V + V, Cos (Wt) Since the above equation should hold during the off periodof the diode, each term should separately zero:-, + Vdt cos (wot - 4)), if diode is off
V=V l/, (a)V f = Vr
if diode is on.
V(2 = V, cos 4 (9b)
The voltage Vo in (1) is the output dc voltage and the volt- V, sin 4 = wRs(CoVdi - C, Vo + C2V ). (9c)age Vt is the peak voltage of an incid- t microwave. Vo The phase delay is obtained from (9) as follows:and V,1 are the dc and the fundamental frequency com-
ponents of diode junction voltage Vd, respectively, when arcan ( Co cos + Cthe diode is off. V1 is the forward voltage drop of the diode Ls Co 1+ vfwhen the diode is on. Vo is known by specifying the out-put dc power. The efficiency and the diode effective = arctan (wRSCf) (10)
impedance are calculated by relating the other variables where vf = Vf/V 0 . From (3), (4), (5), and (9a). the re-such as V1, Vfo, Va1, and 4 to the known variable of V0. lationship between 0,f and r is derived as
By applying Kirchoff's voltage law along the dc-passloop shown as a dotted line in Fig. 3, the dc voltage ir tan Off - Ooff (11)(- V
4 , of Vj is related to the dc component of V ac- I + Vf
47!I
262 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES. VOL 40. NO 6. JUNE 1992
+ ( "- V1 dO
ISv -01
(- V,) cos (0 + o) dO7rR. H~
+ ' cos (0 + 0) dO
Fig 4 Simplified time-domain waveform of voltage V and Vd. ,'1 is the Ih -I' - I, ) sin (0 + o) dOvoltage between itrinsic diode. V is the voltage between A and C of r J -s.,Fig 3. Vanable Iis defined as 0 = - -, (V- )si(O O) OI . (14)
0 off is determined by the ratio of Rs to RL and the ratio of + . , V s (6 + 0) dO
the forward voltage drop Vf to the dc output voltage V0. The input admittance of the diode at the fundamental fre-Vf is solved with the I-V relation of the diode: quency is calculated as
VV1., exp \--V 1 R (12) Y = ,- A l (15)
V, in (12) is a function of 0,f and therefore Vf cannot be If the susceptance part resonates with the dc pass filter insolved without first knowing 0off. However, since the left Fig. 1, the input impedance to be matched to the antennaterm is of exponential form Vf does not vary significantly becomes the inverse of real part of Y.from the cut-in voltage of the diode which is about 0.6 to To validate the closed-form equation the conversion ef-0.8 V for a GaAs-Pt Schottky diode. Therefore. 0o, is ficiency calculated with the closed-form equation wassolved from (11) substituting Vf with a typical value of compared with the results simulated by a nonlinear circuitthe cut-in voltage of the diode. computer program (LIBRA). The efficiency of the diode
Once 0off is solved from a given V0 , Rs, and RL, the as defined by the ratio of output dc power to the net powerefficiency and the input impedance can be calculated from transmitted to the rectenna is compared in Fig. 5 for 10the time domain waveform V and Vd. These waveforms and 35 GHz. The diode parameters used for the calcula-are expressed as a function of 0of., diode parameters, and tion are summarized in Table I. Since the effect of Vb, wasV0 or V1. not included in the closed-form equation, the efficiency
V2 calculation using the closed-form equation terminatedPDC = R- when the negative peak of the diode junction voltage
reached Vr. At 10 GHz the efficiency calculated with the
P = LOSSC.R, + LOSSn.diode closed-form equation is close to the efficiency calculated
+LOSSff~dd with LIBRA. At 35 GHz there is relativly big differencein efficiency showing approximatly 10% of difference in
± (' .... (V - Vf)2- the region where the efficiency calculated with LIBRA isLOSS,.R = 2- ,s dO not affected by the finite Vb. Increase of the difference at
35 GHz is considered to be caused by the effect of non-
1 S'" (V - linear junction capacitance which becomes dominant atLOSS.d 2 7r -,, Rs the high frequency. The closed-form equation is not ac-
S 041 curate enough to be used for the design of the high fre-
LOSSfI R I ,, (V - Vd) 2 dO quency rectenna. However, it can be used to estimate ther --2 ,qf Rs preliminary efficiency of rectenna with any kind of diode
I C2? -0,, (V - Vd)Vd at my arbitrary frequency. This is important because itLOSSoffdoe = 2-R dO piedicts the efficiency level achievable through the non-
-64f Rlinear circuit optimization which usually takes a long time.PDC It also explains how the efficiency is decreased by break-
efficienCY - (13) ing down the power dissipation mechanism in the diode.P °" + P1 In the next subsection we calculated the power conversion
where the variable 0 is substituted for wt - 46. The current efficiency of an ideal diode using the closed-form equa-(I) flowing through Rs can be expressed as follows: tion.
I = + 11, cos (wt) + 11, sin (wt) C. Power Conversion Efficiency of Ideal Diode
Io = (V - Vf) dO The power conversion efficiency of an ideal diode with21rRs -0Eo a finite Rs and C, was calculated using the closed-form
48
C1 "I~ ,v~ i I \P1141111,\1s INJI( lION l(X KlN(i 1l)XI
.... , N %&as measured. 'his beam width was reduced to 65 ° wheni both ,ources were "on." Furthermore. the pot~er with bothI ilti ONsources operating was about 2 dB higher than the powe-
Aith any single source operating. This shows that the arrayis exhibiting good power-combining properties. With bothsources on, the main lobe is centered at roughly 25° . Thisoff-center condition is due to the different lengths of thetransmission lines used to connect antennas and sources. A
*phase difference thus exists between the two sources. Thisdifference can be adjusted and overcome by the use of atransmission line section or phase shifter.
Fig. 13 shows the H-plane pattern of the arra,. It can be-4, seen that no matter which sources are operating. the an-
,NG'-E (degrees) tenna pattern remains relatively unchanged. This is to bei I , /I -planc pattern of a to-element apertur. coupled active expected since the antennas are arranged for E-plane
arra,, coupling. Very little H-plane coupling can be expected.The antenna H-plane beam width was about 120'.
The injection locking through mutual coupling Aas alsodemonstrated. Fig. 14 shows the oscillator spectra beforeand after the injection locking. It can be seen that injectionlocking has a dramatic effect in reducing oscillator noise.The locking bandwidth was measured to be 2.15 MHz.Assuming an injection-locking gain of 20 dB, the lockingbandwidth corresponds to an external Q factor of 217. Thenarrow locking bandwidth and high Q factor are believedto be due to the high-Q transistor oscillator.
VII. CONCLUSIONS
Two types of active antenna elements have been investi-gated. The first type uses active devices directly mountedon the antennas. The second type uses an aperture-coupled
(a) microstrip to patch antenna circuit which can be used toaccommodate a transmit-receive module. A t%%o-elementarray was built and demonstrated in both cases. Injectionlocking was achieved by using either mutual coupling or anexternal master source. Good power-combining efficiencywas achieved for both circuits. An electronic tuning rangeof over 9 percent was achieved for the single active an-tenna element and of about 1 percent for a two-elementarray.
ACKNOWLEDGMENT
The authors would like to thank X. Gao and Dr. R. D.4 T Nevels for many helpful discussions.
REFERENCES
[11 K. J. RusscIl. "Microwave power combining technique,." IEEETrans. Wi'rowave Theor- Tech.. vol MTT-27. pp 4-2 -4"c Ma'.
(b) i 979
I iz 14 Oscillator spectra 1a) before and (b) after injection locking. n21 i. Chang and C. Sun. Milimeter-wave power combining tech-\ criical 10 dB, di, Horizontali 10 kHz/div Center frequency: niques." IEEE Trans. Murowave rheon' Tech. %ol MTT-.1, pp" x4t;Hz91-107, Feb. 1983.
354 oHz 131 J W Mink, "Quasi-optical power combining of solid-statc millime-
ter-wa.e sources." IEEE Trans 4 icrowae Theor Teth . %olinjection lock the free-running oscillator by mutual cou- 4lMTT-34. pp. 273-279. Feb 1986
1- If C Johnson, R. 1. Marx. A Sanchez. and E N.l'kietn. "Apling. circularly polarized actise antenna arra, using miniature GaAs
Fig. 12 sho%% s the L-rlane pattern of the active array FET amplifiers. in 1984 lEE- 41T7-S IpN u bhrn,Dtiq pp 260) 26' "
\hen only the sweeer , "on." when only the oscillator is Is! C R 6reen e al. "A 2 att GaAs TXR \ m dule 262th mnczral
-_In.' and \%hen b h si-urces are "on." When one of the conirol ircuitr' for S-band -.lased array radars. in , w- Ij 17,ources tkas disen,agel. a I JP beam widtf, of about 90' if l TS Int Mir,,wa!'e Sim,' Dig. pp 933 Q ,
49
1264 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES. VOL 40. NO 6. JUNE 1992
The low pass filter which consists of three transmission TABLE 1I
line sections was optimized using TOUCHSTONE. The OPTIMIZED LENGTHS OF COPLANAR STRIP LINES OF 35 GHz RECTENNA
cutoff frequency of the low pass filter was located be- LI L2 L3 L4tween the fundamental and the second order harmonic. (mm) (mm) (mm) (mm)Because the coplanar strip line was not available inTOUCHSTONE, an ideal transmission line was substi-tuted for the optimization. The ideal transmission line was Zh,h: 245 0 Width: 0.2 mm Gap: 0.4 mm
converted to the physical transmission line using the Zo: 130f? Width: 0.5mm Gap: 0.1 mm
closed-form equation in [151. The dimensions of thetransmission lines are shown in Table II. The Duroid sub- Ia 1 L
strate was 10 mils thick with a 2.2 dielectric constant. - MR
The efficiency and the reflected power have been cal- -- - Zt oculated through the nonlinear circuit simulation (LIBRA). bPThe circuit model for the simulation is shown in Fig. 7. lmpdu, c, Rd RL
The efficiency was maximized by adjusting L4 of the dcpass filter. The optimum length of L4 was 0.022 X.. Thecircuit configuration and the photograph of the actual cir- Fig. 7. Circuit model of 35 GHz dipole rectenna for the nonlinear circuit
cuit are shown in Fig. 8. The circuit was fabricated on a computer simulation (LIBRA). Z,( f): Antenna input impedance at the ithharmonic frequency. The real part of the antenna input impedance is usedsubstrate having a thin (10 usm) copper layer because the as a source resistance for the power source. The dual port power sampler
smallest gap of the circuit was 100 Am. This thickness is (DPWRSMP) is used to measure the input and the reflected power.
still 30 times larger than the substrate skin depth at 35GHz. The circuit was etched using the conventionalmethod and a beamlead diode was mounted on the circuit L4
with a silver epoxy. L I
C. 35 GHz Rectenna Measurements
The overall efficiency is defined as the ratio of the out-put dc power to the incident power. The incident power (a)includes both input power to the rectenna diode and re-flection power. A waveguide array simulation technique[161 was used to measure the overall efficiency. This tech-nique permits the generation of high power densities andthe accurate measurement of efficiency. The measurementsetup is shown in Fig. 9. A special waveguide expanderwas fabricated to appropriately simulate the rectenna ar-ray. The smaller side of a Ka-band waveguide (WR-28)was enlarged to make the cross section at the end of thewaveguide expander be square. The longer side was notexpanded in order for the mainlobe of the simulated arrayto be located in the same direction as that of two planewaves constructing the TE1 o mode. The area of the ex-panded cross section was 0.28 inch by 0.28 inch. Thisarea is 40% larger than the effective area of a dipole an- (b)
tenna located a quarter wavelength above a ground plane. Fig. 8. 35 GHz dipole retenna. (a) Circuit configuration. (b) Photograph
Therefore. the actual reflection of the rectenna might be of an actual circuit.
smaller than the value measured with this setup. A photoof waveguide expanded section, a quarter wavelengthspacer and a reflecting ground plane are shown in Fig. 10. r, ,,The quarter wavelength spacer was placed between therectenna and the reflecting plane. 0111111i
The measured data is plotted in Fig. 11. The circles 35 Oft
show the measured efficiencies which are defined as the -
ratio of the dc output power to the input power into the b AP '4 -diode. The simulated results have been plotted together W "--
for comparison. The theoretical efficiency with 100 Q1 dc Fig. 9. Experimental setup for the measurement of the overall efficiency
loads are better than those with 400 (1 loads. The highest and the reflected power of a 35 GHz rectenna.
50
YOO AND CHANG. DEVELOPMENT OF 10 AND 35 GHz RECTENNAS 1265
Pt 120 ,,imq,.
Fig. 12. Experimental setup for the measurement of the conversion effi-ciency of a Ka-band beam-lead diode at 10 GHz.
Diode: Model DOK66OG(Alpha). Rs=4.80, CjoO .13pF, Vbr=9V
--a- Nonlinea Sknuktion
Fig. 10. A waveguide expanded section assembly consisting of a wave-
guide expander, a quarter wavelength spacer, and a reflecting plane.
70. Diode: Model OMK6506(Alpho). R=4-an, CJo=0.13pF, Vbr=9V 40
RL = 400 Ohms C
so - 400(MEASURED) Z 0
400(LA SMUL) 20
40-
Z 0~S30- 0. 20 40 0 O 100 120 10
Input Power [mW]
20- "Fig. 13. 10 GHz to dc conversion efficiency of a Ka-band diode measured10 - with 120 Q dc load.
10-
0 400 0 load and 21% of the incident power at 100 0 load.0 20 40 6 so 100 120 140 'go 10 2wo The reflection can be reduced in the array environment.
INPUT POWER [mW]
(a) D. Measurement of Power Conversion Efficiency at-Diode: Model DMK6606(Alpho). Rs=4.8f1,Cjo--O.13pF, Vbr=9V 10 GHz
RL = 100 Ohms The power conversion efficiency of a Ka-band diodeSC -'--- .5R F2
(Model DMK6606, Alpha Industries) was measured at 1050 - ---- --_ --- GHz. A coaxial cable directional coupler and 50 0 coaxial0'. -- cables were used in the measurement as shown in Fig. 12.
--- The circuit consisted of a 50 0 microstrip transmission- line and a beam-lead diode. One end of the beam-lead
diode is connected to the end of microstrip line and the7. other end is connected to the ground plane through a hole.
The dc output power was extracted using a bias-T net-work. The efficiency was measured with and without theopen stub tuner illustrated in Fig. 12. The measured effi-
0 ciencies are plotted in Fig. 13. There was approximatly0 20 0 60 so ,o0 120 140 ,6 '8o 200 10% of improvement with the open stub tuner. The effi-
INPUT POWER [mWI ciency with the open stub tuner approaches 60% which is
Fig. I1. 35 GHz to dc power conversion efficiency. (a) With a 400 0 dc agreed with the results from nonlinear computer simula-load. (b) With a 100 Q1 dc load. The efficiencies calculated by LIBRA are tion results.plotted together to compare with the measured result. The diode parametersused for simulation are listed in Table I. IV. CONCLUSION
A 35 GHz rectenna has been developed with 39% ofefficiency measured was 39%. The measured maximum power conversion efficiency. The rectenna used a micro-efficiencies for 100 and 400 0 loads are about the same. strip dipole and a Ka-band mixer diode. A computer pro-But the reflection power is 12% of the incident power at gram LIBRA was used to optimize the rectenna for the
51
12I, Itt-}1 R..\'s,~\i IliN1 4F I R('0 s 0" ... '.F IHM)R4 AND t ('HNIQt'-S. VO(L 44. ) NO I J NE 14
high conversion efficiency. The net power conversion et- 1121 K, Chang. J. C. McCleary. and M. A. Pollock. "Feasibility study of35 GHz microwave power transmission in space," Space Power. volficiencv of the same diode was measured at 10 GHz using 8. no 3, pp 365-370. June 1989.
a simple rectifying circuit. The net conversion efficiency 1131 S. M. Sze. Phvsics of Semiconductor Devices. New York Wile%.
was 60%. If the impedance of a antenna is matched to a 1981. ch 2 and 5
rectifying circuit, 60% efficiency would be possible at 1141 S M. Wright. "Etticient analysis ofinfinitemicrostriparraysonelc.tricallv thick substrate.' Ph D dissertation. University of Illinois atGHz with this diode. A closed-form equation was derived Urbana-Champaign. 1984.
for the analysis of the high frequency rectenna by neglect- 1151 I J. Bahl. "'Transmission lines,'" in Handbook of Microwave andtor the higheris orde hai cs. f ency order to e tete Optical Components. K. Chang Ed.. vol. I. New York: Wiley. 989.ing the higher order harmonics. In order to estimate the p 35
maximum e-fficiency limited by a finite Rs and C,,, a net 1161 P w Hannan and M. A Ballour. "Simulation of a phase-array an-conersion etficiency of an ideal diode with the same Rs tenna in a waveguide." IEEE Trans. Antennas Propagat.. vol. AP-
and C, v'as calculated using the closed-form equation. 13. p 342-353. May 1965
With the Ka-band diode used for 35 GHz rectenna. themaximum efficiency was calculated to be approximatly601 at 35 GHz. The actual maximum efficiency is smallerthan this efficiency due to the effect of the finite forward Taewhan Yoo received the B.S. degree in nuclear
engineenng from Seoul National Universa' atvoltage drop and the breakdown voltage. Seoul. Korea in 1981. the M S. degree in phxsics
from Korea Advanced Institute of Science andTechnology at Seoul, Korea in 1983. He is cur-
ACKNOWLEDGMENT rently working on his doctoral dissertation at
The constant encouragement of Dr. A. D. Patton is ac- Texas A&M University. College Station. on mi-crowave power transmission.
knowledged. The authors would like to thank Mr. J. From 1983 to 1988 he was with Electronics andMcSpadden for many helpful suggestions and reviewing Telecommunication Research Institute at Taeton.
the manuscript. Mr. W. C. Brown of Raytheon Company Korea. His research area was the development ofo pto-electronic devices for fiber optic communication. His interests includemicrowave integrated circuit and its CAD design. wireless communication.
machining the hardware, and Dr. S. M. Wright for allow- and optoelectronic devices.
ing us to use his computer program for antenna inputimpedance. The authors would also like to thank review-ers for making many useful suggestions. Kai Chang (S'75-M'76-SM'85-F"1I) iceised
the B.S.E.E. degree from the National Taissan
REFERENCES University, Taipei. Taiwan. the M.S. degree tromthe State University of New York at Stony Brook.
(11 W C Brown et al. U.S. Patent 3 434 678. Mar. 25. 1969. and the Ph.D. degree from the University ot121 W C. Brown. "Expe ment in the transport of energy by microwave Michigan. Ann Arbor. in 1970, 1972. and 1976.
beam." in IEEE Int. Cony. Rec., vol. 12. pt. 2. Mar. 1964. pp. respectively.8-18 From 1972 to 1976. he worked for the Micro-
131 - . 'The history of power transmission by radio waves." IEEE wave Solid-State Circuits Group. Cooley Elec-Trans. Microwave Theors Tech.. vol. 32. no. 9. pp. 1230-1242. Sept. tronics Laboratory of the University ot Michigan1984 as a Research Assistant From I176 to I9-S, he
[41 -. *Expenments involving a microwave beam to power and po- was employed by Shared Applications. Inc.. Ann Arbor. where he sorked-,ition a helicopter." IEEE Trans Aerosp. Electron. Svst.. vol. AES- in computer simulation of microwave circuits and microwave tuhe, From5. no 5. pp. 692-702. Sept. 1969. 1978 to 1981. he worked for the Electron Dynamics Division. Hughes Ar
151 -. "'Solar Power Satellite Program Rev. DOE/NASA Satellite craft Company. Torrance. CA. where he was involved in the research andPower System Concept Develop. Evaluation Program.~ Final Proc. development of millimeter-wave solid-state devices and circuits. powerConf. 800491. July 1980 combiners, oscillators and transmitters. From 1981 to 1985. he worked lor
161 J Schlesak. A Alden and T Ohno, "A microwave powered high the TRW Electronics and Defense. Redondo Beach. CA, as a Section Head.altitude platform." in IEEE M7T-S Int. Microwave Symp. Dig.. 1988. developing state-of-the art millimeter-wave integrated circuits and subs s,-
pp 283-286 terns including mixers. VCO's. transmitters, amplifiers, modulators, upI- W C Brown. "Performance characteristics of the thin-film, etched- converters, switches, multipliers. receivers and transceivers He joined ihe
.ircuit rectenna." in IEEE MTT-S Int. Microwave Svmp. Dig_. 1984. Electrical Engineering Department ot Texas A&M Unisersit. in Auguipp 365-367 1985 as an Associate Professor and was promoted to Professor i In 8S0 In
181 1 J Nahas. "Modeling and computer simulation of a microwave-to- January 1990. he was appointed E-Systems Endowed Protessor ot Electri-dc energy conversion element," IEEE Trans. Microwave Theory cal Engineering. His current interests are in microwave and millimeter-Tech . vol. 23. no 12. pp. 1030-1035, Dec. 1975. wave devices and circuits. mtcrowase integrated circuits. micro.ae op
191 W C Brown and E. E. Eves. "A microwave beam power transfer tical interactions, and antennas.and guidance system for use in an orbital astronomy support facility." He served as the Editor of the four-volume Handbook olMicro'waiv andFinal Report (Phase 1I1 to NASA. NAS-8-25374. Sept. 1972. Optical Components. published by John Wiley & Sons. Inc in 148) and
1101 G P Boyakhchyan er al . "Analytical calculation of a high-effi- 1990. He ts the Editor of the Microwave and Optical Technoloes L,'rrericiencs microwave rectifier employing a Schottky-bamer diode." and the Wile: Book Series in Microwave and Optical Enigneerin He hasTelecom, Radio Engin ( English Translation of Elaktrosvyaz. and Ra- published over 150 technical papers and several book chapters in the area,diotekhnika) vol 37-38. no. 10. pp 64-66. Oct. 1983. of microwave and millimeter-wave devices and circuits. Dr Chane re-
11II W C Brown. ''All electronic propulsion key to future space ship ceised the Special Achievement Award from TRW in 1984. the Halliburtiondesign." presented at AIAA/ASMEISAE/ASEE 24th Joint Propulsion Professor Award in 1988 and the Distinguished Teaching Award in 1S410Conf. Boston. July 1988 from the Texas A&M University
52
AIM (Mn \\i M 41 1.l1 ,1I
Electronically Switchable and TunableCoplanar Waveguide-Slotline
Band-Pass Filters)I!in-l 11.1 u 111 11111 A. I iit \\ arro. and IKat C hang. I1wIL
fh~tract - X noeir coplanar waseizuidle If IIN I sluiline hand-pass filter -----
has been deselopcd. I he circuit allows planar integratioin of actise anidpassist- semiconductor drsioces both in series and in shunt. Yo test the %11 RoTI LoT1% 1)H %Ni % IM lfilter, a new microstrip to *.Iotline transition was desig~ned and two ofthese transition% eshihited in inserlion loss of less than L.0 dR o'er the Fig L. (jress-scctiiinal %c% ot microsirip line. N,inc irnd oiril,inar2M( to 4.0 Gilt range, % threv-section (P55 -slline hand-pass filter ssieguidcde-monstrattd an insertion loss of less than 0.2 dB 05cr tlit pasandcentered at 2.9 (Mi. A three-section CM~ -slotline switchahie hand-passfilter integrated with three p-i-n diodes vwas deseloped with a 01.7 dBlinsertion loss in the passband when the p-i-n diodes arc twt and 25.01 dB icristics make CPW important for mill imetcr-\ssas.c circuitsisolation across the entire hand when the pi-i-ri diodes are 4)N. A and have aroused considerable interest in rnicro),ka\c andthree-section UP55-slotline saractor-tunable filter integrated with three millimeter-wvave integrated circuit design [ 1J.Although'itractor diodes was demonstrated with a 2.0 dB insertion loss and oser sev~eral papers [4]-[tJ analyze the characteristics, and td% an-21V-1 electronic tuning range. Simple transmission line circuit models tages of CPWk. there are only a limited number of iP'\were used to optimize the design.
components available for circuit design. Thesec component,include directional couplers [6]-[7]. mixers 1[s\. [91. diplc\ers
I. IN I R(MiL C'I [(N 11) 111)l, and end-coupled resonant CPW filter,, [121. ThisHLRLis n icycsinedemnd o LNC mcroa~eand paper describes the integration of microstrip. slotline. and
T milRlis e-a,' incrine deand tonusetmic nerae d CPW to create a novel CPW slotline band-pass tilter ind a.1 mlliete-ssave xbrd ad mnolthicintgraed tr- new microstrip-to-slotline transitton.
t~l or mnan% system apiton.Because of their paaapplcatons plnar Fig. I shows a cross-sectional view of each of these trans -nature, these circuits offer cost, weight. reliabilitN, and re pro- mission lines. The filter uses CPW resonators interconnecteddUCIhilitx ads antagcs w~hen combined with photolithographic vavaslotline. The filter is planar and allows easy series andtec hniques. Microstrip has been the main transmission line shunt device mounting. Transmission line models were de-used in micro%%ave circuit dsg.partly because of the vastdesitin.%eloped to design the filters. To test the filters. a ncssaImount of design information waaiable. Although semicon - m icrost rip-to-slot line transition was developed. Thi- transi-Juclor de cscan bec readily integrated in series, shunt lion is unique and differs from previous transitions ',] due
,1~icsnistti mun~dh drilln hog h usrtt the advantage of providing dc blocks required frd~cEDrilio Ads cost and discontinuities which make the desien aigad o-asfles ordc h tasto osmorc difficult and deg'rade the Performance. especiall, bisnanto-as itr ordc tetasto os
A pair of micrustrip-to-slotline transitions was developedm~illrmeter-%%ase frequencies. Furthermore, the microstrip with a 1.0) dB loss throughout the 2.0 to 4.0) GHz band. A
impeanc ,id gide-waelegth harcteistcs re ery three-section CPW-slotline filter was developed with less'ens1~cit Ntsustrate thickness,, which further increases de- than (1.2 dB insertion loss in the passband and over 30.0) dBsi1gM probems. mi higher frequencies. isolation in the stopbands. Integrating three p-i-n diodes in
( oplanir vsaseguide (CMW) is an alternativ transmission the three-section filter created a switchable filter. When theinc that is trul\ planar and allow% easy series and shunt p I' dides were reversed-biased, the switchable filter exhib-
des c montig. (PW s ilt vey snsitve o sustrtited a 0.7 dB insertion loss in the passband and over 3(0.0 dBhickness and allows a wide range of impedance values on in the stop hands. When the p-i-n diodes were tom~ard-
rclativ-el\ thick substrates,. The radiatton loss in the CPW biased, the switchable filter exhibited over 25(0 dB insertionOdd mode is low for an open transmission line. These charac- loss. Integrating three varactor diodes in the three-sction
filter created a tunable band-pass filter. The %aractor-Ma~nuscript receised June 12. 11)(X): revised October 3. IWtl This tunable filter exhibited a 20%7- electronic tuning range with
wol js suppo~rted in part by the U.S. Armv Research Office arid the 1 )d neto osi hepsbn n vr2.)dresis Higher Education Coordinating Biard's Advanced Technology . Bisrinls nth asadadoe 00d
'uhu isolation in the stopbands.'i Ith \4,j, with the Department oif Electrical Engineering. Texas The circuit design was based on simple transmission line
\. 's1 Unisersits. College Station. TX fie is now with the Epsilon models. The insertion loss of the microstrip. slotline. andtjambdd Llectronics Corporation. Geneva. IL W1~t34. P wainld .Thdsctnuvefcsbtente
I.\ ssasirro aind K (Chang are with the D~epartmeni (if ElectricalCWwainld.Th dsctnuyefcsbtenteI ngincering. l-exas .A&M tUniversity. College Station. TX 77943-31t2K slotline and coplanar resonator were not included. The para-
Ift F tI.,& Numher OWN5 sitics of the devices were obtained from the vendor catalog.
(El.44O/~ itilOi~SX$4l.(1 1991 IEEE
53
Silt , Il ( OPLANAR wAVE GUI)I-SIOTLINE BAND-PASS FIITERS -c
(a) (b (c) (d)
i g 2 t)iftcrent ncronstrip-to-slot line transitions. (Solid line: rncrostrip line Dashed line slotlic I (al Solderedmiciosirip short and unitorm A /4 slotline. (b) Virtual short v ith unilomri A -4 open nlcroistrip and unilorm A '4 slolineIc) Soldered microstrip short and slotlh'e open circuit through nonunitorm circular slotlhnc. (dl Virtual shori 'lh
niiriunitorni circular microstrip and slotlIne open crc¢uit through noMunitm circular slot.lin
tion. C_, and C,,, represent the open circuit capacitance formicrostriy and slotline. respectively. The transmission linecharacteristic impedance ot microstrip and slotline are de-
L, t:0 ., L1 L i L -I -3, L L5 fined by Z,,,, and Z,, respectivel. Z,,,, is equal to 50 Q" - - while Z,, is equal to 60 !1. respectively. Z, and Z, are the
low and high impedance values used in the slotline low-passfilter design, respectively. The transformer ratio. N. is re-
7__ ( ported by 1141 to be
I (It (1)
where
(a) 1'(h)=- fh E, (h/t (2)
,, 2
C , C \. L.where h is the thickness of the substrate. b12 is the lengthof the microstrip feed to the slotline, [, is the voltage across, -+-,Z Lv ,, , . cthe slot, and E,(h) is the electric field of the slotline on the
. Lidielectric surface on the opposite side. From Cohn's analysis
1151.Z~. A, N Z.
Si -,L* - iE,(h) ----- (cos ( h -cot(q,,)sin --L, 1 )) (3)
f ik: [hc no.el microstrip-to-slot line transition: (a) configuration where(b) Cqui, alent circuit
,,hikch gtcs typical values only. Even with these uncertain- Al--l arctan I (4)tics. the analysis did predict the performance of the transi-tions and filters. Better agreement between theory and ex- A,pcrtment could be achieved with more accurate information U= - (5)and analsis.
Ihese new filters offer low passband insertion loss andhigh stophand isolation. The theoretical and experimental Aresults agree well. These filters are amenable to monolithic V=11 7-- - (6)mipiermentation. The circuit configurations should have manyapplications in microwave and millimeter-wave systems. where A,, is the guided wavelength of the slotline.
A computer program based on the equivalent circuit ofII. MI('ROSTRIP-T(-SLOTLINF TRANSITION Fig. 3(b) was developed and used to analyze and optimize
hLYA-luss,. wide-band microstrip-to-slotline transitions have the transition's performance. The lengths of the microstripbeen reported in the literature. Schuppert 1131 reviews the open stubs (L,). cach section of the slotline low-pass filterdifferent transitions shown in Fig. 2. These transitions offer (L1.L. L , L. L,) and interconnecting slotlines (L,,) weregoo-d performance and are widely used for circuit design. optimized at fixed characteristic impedance values for theOing to the requirements of device biasing, we developed a required reflection (S11) and insertion (S,) characteristics.ne'k microstrip-to-slotline transition which incorporates a dc The program was optimized for maxinium insertion loss (S,1 )block and a slotline low-pass filter for an RF choke, of -0.5 dB with a minimum return loss (S,,) of -20 dB
Fig. 3(a) shows the microstrip-to-slotline transition circuit. throughout the 2.0 to 4.0 GHz range. Closed-form equationsThe circuit consists of two microstrip-to-slotline transitions, for microstrip 1161, 1171 and slotline [181 transmission linest so slotline low-pass filters, and two dc blocks for device were used to analyze the circuit parameters. The transitionsbiasing. Fig. 3(b) shows the equivalent circuit for the transi- were fabricated on a 50-mil-thick RT-Duroid 6010.5 sub-
54
ii Liv IRANSA4. I IO)NS ()N .Il R o5 A' Ir I 11lORY1 ANDI1) ilHNIOL IS, Vol 19 W,) 1.\R tAI
IAiBLEt I
1k I 1(N S I 1( ' N I-Fu,. Ila)L.
S 24S 11.) lll QI.,, 5(4 0! , 10 Ill) 12
\ruluhle 1I- I., I., L~ I .L, . LL,,c
lluil 2o.ill 491 1.901 0-.7h 2.53 1.i. 9 701
~9MAC,dB/o~
- -- - - - ----- --- - - ---------
-~Z 7- Zeri e
.10 (b)
.0 . 3.0. . _______________ 4i.0 ~ .~ i7 I.
Fie. -1~~~~~~~~~~~~~~~~~~ Thoeia n xeietlisrinls ftetasto. Fg .TenvlCWsoln bad-a fitr a ofgrto
th-piie ici iesosadFg hw h ho io, . ± ilj- (IG
retial nd xpermenal nsetionlos ofthetransition. Fig 5. Th5 no1e CP-sotin h=dps 6iler (1) Z,,n ura 01 ,, 0f .ua=501Fepive.y Thecl experimental insertion loss of te (b)si eVuivalee cirLcuL,,it. L'1 "
ran The ls parfraeters osieresed usin goonad snc TALEt
thle tmze circi-t-ot d ime nsiionsFi. twsow mcth he-SowINFGS
lu)-slotline transitions, and a 26 mm length of slotline. Over- From the equivalent circuit, a computer model using c'--.all. thc theoretical and experimental results agree well. This cading transmission lines was developed. The model a,-circuit was used to test the CPW-slotline filters. counts for all open and short termination effects. The modc
optimized the CPW resonators and interconnecting slotlinz111.Tii-_ CW-S-ortINEBANDPAS FiLERSlengths to achieve at least 30.0 dB insertion loss in thzIll.T~l CPWS~rINE ANDPASSFILERSstopbands (2.0-2.5: 3.3-4.0 GHz) and less than t0.2 dlB in thz
Fjig. 1 shows at cross-sectional view of the CPW transmis- passband (2.75-3.05 GHz). The closed-form equations to-Nion line. CPW allows the easy integration of series and CPW in [181 were used to analyze the circuit parameter-shunt devices on a planar transmission line. Previous CPW The characteristic impedances of the CPW resonators ancband-pass filters [11 used end-coupled resonators. Although interconnecting slotlines are 50 fl and 60 fl. respectively. Tcthese circuits show good performance, device mounting and minimize optimization variables and form a symmetrical con-biasing are difficult to incorporate. This section describes the figuration, the lengths of the first and third resonators arZuse of CPW resonators coupled via slotlines to create a new equal to each other. The optimization variables L,, L,:CPW-slotline band-pass filter. L, 1, L, 2 , L,., and L,, are shown in Fig. 5 and the optimizec
The most basic design element of a band-pass filter is the dimensions are given in Table 11.resonator. Resonators can be designed with a combination of The filter and transitions were fabricated on a 50-mil-thickopen and short terminations. Because of the ease of device RT-Durolid 6010.5 substrate and the S parameters wereintegration, our filter uses CPW resonators with an open and measured using standard SMA connectors with an HP-85 11a shorted end interconnected via slotline. Fig. 5 shows the network analyzer. Fig. 6 shows the theoretical and experi-novel CPW-slotline band-pass filter configuration and its mental insertion loss of the filter with the transitions. respec-equivalent circuit. The microstrip-to-slotline transition is used tively. In the passband. the theoretical insertion loss (includ-to test the filter. The insertion loss through a pair of transi- ing transition loss) is around 0.7 dB at a center frequency oft6ons is about 1.0 dB across the 2.0 to 4.0 GI-z range, as 2.9 GI-z. The experimental results of the CPW-slotline fihleTdescribed in Section 11. with two microst rip-to-slot line transitions show a center fre-
55
Situ'e a[ atOPLANAR WAVtFGUI)[ -SLOTtY.INIr tANI)-ASS [I. 111IRS 551
S-2 1 /14 1.9 MAGE .0 dB
1 1:.!3 dB/V 1. 8i dB
MARPER 1 T_ he.." L'
IlI
.31).
203.0 i-req-GHz 4.01II~,!,-
I-i 11: Iheoretical aind experimental insertion los of the novel Z,* .1.
CPVh -slotline band-pass filter. .. ~ '/
quencs of 2.9 GHz with an insertion loss oif less than 1.2 d13. NIA
The actual loss due to filter excluding the transition loss is -, . .
onis (0.2 dB3. The stophand isolation is greater than 30(0 dB3cscept for a microstrip feedline stub resonance from the(htrani-sition at 2.1 GHz. The feedline resonance is due to the FirI. 7 The p-- diodc ,itchahlc CPW-slotline huinc-paN, filter.quarter-wavelength stub (L_, in Fig. 3(a)) resonating at 2.1 (a) eOnfigUration: (b) equivalent circuit.GlIz. The effect could be eliminated by using a 'A stub. Thestub resonance was verified using the HP-851t0 time-domain functions,. The filter was isolated to show that the R,pike at 2.1 (if z was not due to the filter. The theoretical Lmiodcl cenerallv predicts the experimental performance. The Rr Ci L
design can be easily modified for different frequencies andHpassaind charaeteristtcs.
CP
l.i'-i-\ ID~im S's Itt( iit i CPWk Si (tt ti. it Iig S the cqui'alent cireuli dI I p-i-n diodc
BANi-I'Ass FititRsS- itchablc band-pass filters have many applications in TABLE Ill
mnicro%% ave systems. Martin et al. 1191 integrated p-i-n diodes Orit tzi/i Li\, us )I ioi TiiRi i-Si ui lo-. Tiiktii--- Dioni
into a ring resonator to form a switchable ring resonator. SMt( iiABIJ CPWV-Si On si B-\\io-P-,ss Fuii tR
I his section describes the integraiion of p-i-n diodes into the Stiiks. i\ Fi.. 7 .7=51I
PlanarI C.PW -sloiline filter to create a switchable microwave ZH = 245; 1. Z' )( 11.4," = 511 1). Z,,. = O1.,4;)1
filtetr The p-i-n diodes dre mounted over the open end of Variable L, L, L, L.. 1_1 L,the (TPW resonators. Fig. 7 shows the circuit configuration Trm u- s~ 99 3.15 9423and equivalent circuit of the three-section. three-diodeS\S itchable filter. The p-i-n diode has the equivalent circuitshi)'~n in Fig. 8. R, is the forward-biased series resistance of interconnecting lines are optimized to provide low insertionthe diode while R, is the reversed-biased series resistance. loss in the passband and high rejection in the stophands. TheC represents the junction capacitance and LPand C,, repre- p-i-n diodes are then for'sard-biased (()N state) to determineNent the package parasitics. L, accounts for the inductance if the circuit meets the isolation specification across theot the bonding wire to the semiconductor material in the whole band. The circuit was optimized for minimum inser-diode The p-i-n dioides used are M/A COM 47047's with tion loss in the passband when the p-i-n diodes areI... 10 ni. - C, (.1 pF. L, = 0.2 nH-I and C',=0.3 pF at reversed-biased and maximum isolation when the p-i-n diodes- 511 V and R, =1.3 Q at (W mA. Although R, is not are forward-biased.quoted. similar reversed-biased devices show a resistance of Table IlI shows the circuit dimensions for the swvitchable2 11. These values were used in the computer model analysis filter. The switchabic filter was fabricated on a 50-mil-thickand optimization. RT-Duroid 60110.i substrate and the S parameters were
In the design procedure. while the p-i-n diodes are re- tested using standard SMA connectors with an HP-8510%ersed-biased ((OFu state). the lengths of the resonators and network analyzer. Fig. () shows the theoretical and experi-
56
If tI~ 2\5 Ii INSI'A I ](il 0 \\l ]III ('IN \\I) it ti~tuif I,,\()[ i, \~k
"2 1- MAI S 2 1/IQ log MAOREF0. 0d91 10.0 dB3/
tiA " F*' t .OH -30.026 dB "s~
MARKER 1 ..
2.97 GH4 - E~periment
- .- ---.-x .' -.-- ---
___________________Fig. 10 A* photograph of the %arzictor-tunahle CflPW - lot inc ixind-Pit-.20 3.0 Frqiiz 4.0 filter
f g I [he dieoreticol and experinmental reSUltS Of the p-i-n diode,\A itchahie harid-pass filter. R jM L
melresult ,,oth \wh bfilter. From the theoreticalR 5 C() Lrsl,.the center frequency is 3.0 GuIz with an insertion
loss (includine the transition loss) of [.0) dB when the p-i-n
d10iods arce nift-. When the p-i-n diodes are ON, the isolationis, it least 30.f) dI3 across the 2.0) to 4.1) GHz range. Thecxpertmental results indicated an insertion loss of 1.7 dB inthe passbhand and over 25-0 dB isolation except for a spike at21I (I I i. Ihe reasons for the spike are explained in Section C111. 1bis loss Includes the 1.0 dB loss caused by the twotralstttiotls: therefore. the actual filter loss is only 10.7 dB. Fig ftI The equiwalent circuit of a varactor diode.
[his, planar integrated circuit can be easily assembled at5cRx lov. cost. ['he pood switching performance has manyssstcni applications. Because of the planar nature, this cir- 0 to 31) V. The varactor diodes used arc from NI A\ (UNI
comtil 'i menable ito monolithic implementation. (model 4600) I) with L, =0.2 nil, C, =1.5p F. and R -
S 1). C, varies from 0(.5 to 2.5 pF. These approximate \alte'
V V 't Nwl CW-SIOf I[Niwere used to optimize the circuit parameters in the mode!l
\. \\R5(i -h N.h Ft CPWSi(ftt The junction capacitance used in the circuit optimization %NJ,B \N-l~S5 Ft T~RS1.10 pF. The center frequency for the design optimization %kix
Electronically tunable microwave filters have many appli- set at 3.1) G1-l with a + 200) MHz bandwidth. The circuitcations in a wide range of microwave systems which require model optimized the passband to less than 1.5 dB insertiol)broaid tuning ranges and fast tuning speed. Although YIG- loss with more than 30.1) dB isolation in the stophands.tuned filters offer vet-y wide tuning ranges, they are limited Table IV shows the optimized circuit dimensions for theIII tLuntng speed. Varactors can provide wide tuning ranges varactor-tunable filter. The varactor-tunable filter wAas fabri-ss ith high tuning speed for frequency agile systems. Hunter cated on a Sli-mil-thick RT-Duroid 6101.5 substrate. Part-.aind Rhodes [21)1 have developed varactor-tunable combline (a) and (b) of Fig. 12 show the theoretical and experimentalfilters, wkhile Makimoto and Sagawa [21] have developed results of the varactor-tunable filter. The frequency tuninvsaractor-t unable microstrip ring filters. This sectton de- range agrees well but there is a discrepancy in the band-,cribes the integration of three varactors with the CPW- width. The theoretical bandwidth increases at the lower endslotline filter to create a varactor-tunable band-pass filter, while the experiment shows an increase at the higher end.The '.aractors are mounted over the open end of the cpw [his discrepancy could be due to the simplified model usedresonator tn the same manner as the p-i-n diodes of the in the analysis. The model does not account for discontino-sositchable filter. Fig. 10 shows a pholtograph of the ities of the CPW to slotline junctions, the variation of the Q2%aractor-tunfable CPW-slotline filter. The circuit configura- factor of the varactor as a function of voltage. and thetion and equivalent circuit are the same as in Fig. 7. except variation of the device parasitics.that the p-i-n diodes are replaced by varactor diodes. The A tuning bandwidth of 6W0 MHz was achieved for varactorsaraCtor equivalent circuit used in the circuit model is shown bias voltages of 0 to 25 V. The maximum insertion loss ii?
in Fig. 11. R, is the series resistance of the diode w~hile C , 2.15 dB (excluding the 1.0 dB transition loss) occurs at therepresents the package capacitance. L, accounts for the low-end frequency of 2.7 6Hz and decreases to 0.7 dB at theinductance of the bonding wire to the package. The junction high-end frequency of 3.3 GFHz. The passband \aries fromc:apacitance. ( V). varies as a function of bias voltage from 2Sf) MHz in the low end to 450 MI-z in the high end
57
3 slit ci a? COPLANAR W~AVE( UIDE -SIOTINU BlAND)-PASS 1 1i1- it R
WO -%rrf i'o ith loiv insertion loss tin the passband and high isolation in25 00 - . li stophands.
-i ----- A simple CAD model and a computer program have been0 ~~c~r (5 ____ des eloped to optiiic the various parameters for the re-
- -- ar~Ctr '5 i quired filter design. T hese circuits ot'ler man\ advantages.S ~~,(r 255among them low cost. loss loss, high isolation, and ease of
F-- - --- series and shunt device inteieration. The filters have man\- applications in various microwxase systems- The circuits are
0 000 I ______amenable to monol!ithic circuit integration and can be scalede- to the titill imeter-wave frequency range.
A(-~t ,N(MI)([ Xi N~I
I'The authors would like to thank R. C'. Waits for his___0_---- instruction and use of the f'abrication facilities.
2 o0C 1.000' FREG-GiCZ 4 000
(a)
1 5..7 0i It] 4[1 M. Houdart. Coplanar lines: Application to broadband mi-
' M~kE iT 7 ~R. A. Pucel." Design considerations for monolithic microw.ase
(31 D. R. C'hen and D. R. Decker. "MMICs-The next generationof microwave components." MicroH'ate J.. pp. 67-68. May.
74-NI. 1975.T. Hatsuda. -Computation of coplanar-type stripline character-istics b% relaxation method and its application to microwavecircuits'." IEEE Trants. Microwai e T/ieorv Tech.. v-ol. MT-T-23.pp. 795-901. Oct,. 1N75.16] J. B. Knorr and B. Kuchler. "Analysis of coupled slots andcoplanar strips on dielectric substrates." IEEE Trants. Mi,-
45 cro~ai c T/won- Tec/h., vol. MTT-23. pp. 541 -548. July 1975.17] C. P. Wen. 'Coplanar-waveguide directional couplers.IkE
Trans. Vicroita, e T/itors Tch/_ vol. M'FT- t 8. pp. 31t8-331.1 i 2 - I tic \atactor -tunable CM sloiline band-pass tICtr. U) iheo- (81 D. F. Williams and S. E. Scbssarz. "A planar subbarmonicals-
rtL'ncal in'.riinn loss. (h) cspernmentail insertion lors pumped 71t GIltz receis er ss ith integral teed antenna." M~t. Ihi Iraresi and Millieter tia~,' i c it %o .I no. 2.pp. IN t13 .t986.
191 L. Yuan. J. Paul. and P. Ylen. -1-40 (jli quasi-optical planarTFABLE IV miser." in IEEE TiT-S Init. Micrisaics .Ssip.. 1982. pp.
) 51IILi silos isi 111i \MR-1-~i ult CPW-Sil-1 .7437
B\\O'PA55,F It-u iok N j, NT [It)qY C. Shih. L. Q. Bui. and T. ttoh. 'Milimeter-\save diplexersDniisi Ri i'i ~ii [in VRA(-ToR Diois w ith printed circuit elements.IEEE Trns. ,ificricate Theor\
Z 45 1. Z 0011.Z,, 5011.Z,,,= 6 12 Z, - 5 1Tec/h.. .ol. MTT-33. pip. 1465-1409. Dec. 1985.[il D24 F.. Wilim an) S. E.~ 6))z -Des5ign an and performance of
Thor Tch ol t1. p. S'56,Juts 1983.riniIm. )Q) S.r 1 01 1 2 K. uta R. (iarg. and 1. J. Bahl,. Mi cro.%trtp Li.nes and
s [)ht~s edham. MA: Ariech House. 1979. cb. h.1131 B cuer."Microstrip /slot line transitions-Modeling and
U~nequal passbands have been reported before for other experimental investigation." IEEE Tran, . Microtiaie T/meunt\pcs. of saractor-tuned filters [1.7ec/i., 3. pp. 1272'-1282. Aug. 198S.
1141 J. B Knorr, "Sloiline transitions."IEEE Trapo. Microw14aveTheory Tecih_ vol. \ITT-"2. pp. 548-5s4. 1974.
1151 S. 13 Cohn. 'Slottine field components." IEEE Train. Mi-V1. CNCLUSON'Scrowai e Theors Tee/i.. vol. MT F-20t. pp. 172-1743. 1472.INos\el C'PW-slodtinL band-pass filters and MiCro)strip-bo- 1161 E. Hamnmerstad and 0. Jensen. 'Accurate models for mi-
slotline transitions have been developed. Two transitions crostrip computer'aided design.- in IEEE: MTT-.S Inut. Mi-blocs ad thee-ectin sotlie lw-pas 171 roisaIe 5%mp, Dig.. 1980. pp. 407-40)9,ncorporating dc bokanthe-eto lti owps (11R. Garg and K. C. Gupta. "Expressions for %\aselength and
fiters were designed to achieve 1.0 dB insertion loss over the impedance oif stotline,' IEEE Trans. Microwai c T/u-or Tech..2.0 to 401 6Hz range. A novel band-pass filter was developed vol. MTr24. p. 532. 1976.ising three coplanar waveguide resonators cascaded by slot- [181 G. Ghinoe and C. Naldi. "Analytical formulas for coplanar
line. Te fiter wer mae sitchbleand unale b-inor- lines in hybrid and monolithic MIC's.'* Electron. Len.. vol. 201.po in the f i w r a d e sa c habdioe s an tun e bysnctor-. no. 4, pp. 179- 187. 16 Feb. 1984.
the -i- an vaactr dodesin he esoatos. (19] T. S. Martin. F. Wang. and K. Chang, "Theoretical and experi-respectively. Wide-band switching and tuning were achieved mental investigation of varacior'tuned switchable microstrip
* 58
I ,',, Islii , )1i k citcit'. It,. I,,oi III, 'ma , /he, -I Ii it III(i iaid pilecis 1 II\i)si l \I 11 .itl [ sC aract, 1r (In'thlec end-i Ire' ,I,. Jii p I I .141), D~c' I Q.'N. [i.LIllk i C loniIilii it. s5 i diil iild tIIlItAb ( P\\ filter,. ind saralc-
I(. I (iIeI(r tiid I I AOdc. I lCcuIi .ills I W11blo! 1111151 1s Ic Or-iiiiiCd (. '%% OSCIllatlOrs [IiCIIs )CIdIhnQ I. lie ha5 iAso develoipedl'iipistllers. IL 1 1 1ot 1,11t 1/i 1, s I/1"o In it sl. K- .Id Ki-biMnd ApItie-Csiiilcd P).iCh .iiiieria for a NASA.
I [- pp 1) 1 4 rsi Sep W . (,1 19s " I,,:%% I, - I I list frjiinit- priojcct, I Ic is cur rent k completing thcf,Il \1 \tiits1111 Ind \t 'Kasifar~rilled h~lll.1 Ill- ICquirCenICIti Iiir (he M S.1-1 degree and wijll PUrsueC 131h.D. tui.
tcrs ii'.ine miicrssrrip-hiin rc~siratsi'. n Ill I 1_- fill. Unider thie directioin Of Dr Kmi (iiie~Iisissu Us ii u, th . T. Ip 41 - - 14I
Yoing-Hui Shu Aas born in JinioChina. on -t) hn (7~f~\N5F4 eApnl24,195. H reeuse th 135. fldMS.ceised the BSEE deeree from the National
degree-, in electrical engineering fromt the Taiwa Lni\ersiiN. Talipei. Taiwoan. the MSNanjing Institute of Technology. Nanjing. degree from the State lUjniersits iii NessChina in 1982 Lind 1)85;. respeetisely. York atl Slow, Brook. and the PhD dcree
From March 1985 to Jul\ 1981), he wast, with fo h fMcia.AnAhrthe epatmen ofRadi) Eginerin ofin 197(). 1972, and 197h. respectisel>.
Naning Institute of Technology (current From 1972 to 1976. he worked tor the
naom Suthst n98r9tx aoJb \is wLeturr Microwave Solid-State Circuits Group. Coo-FrmAgsr98 oJll9(1&ewswt leo, Electronics Lahorators Of the Lnisersitsthe Department of Electrical Engineering of ot: Michivan a, a Research Assistant. From
ITexas A&M Umniersith as at Research Associate. In August 19904. he 1970 to 1978. he was \kitI.h Sha-red Applications. Inc.. Ann Arbor.Joined Epsilon Lambda Electronics Corporation. Cienesa. IL. d's at ,h ere h e worked on computer simulation of microwase circuits andNlicrowase Design Engineer. His current interests are in microwave microwave tubes. From 1978 to 19M1. he was with the Electronand millimeter-wase devices, circuits. and subssstems. Dynamics Disision. Hughes Aircraft (iompan\v. Torrance, CA. "here
Mr. Shu has published 201 technical papers in mucrowase and he was involved in the research and development oft millimeter-\Aasemillimeter-%wave areas. solid-state dev ices and circuits. power combiners,. oscillators. ind
transmitters. From 1981I to 19S5. he wais with TRW' Electronics. andDefense, Redondo Beach. CA. as, a Section Head deselopin staite-of-the-art millimeter-wave integrated circuits and subssstems includ-ing mixers. VCO's. transmitters. amplifiers, modulators, up-con-
+ serters. switches, multipliers. receivers. and transceisers. He joinedthe Electrical Engineering Department of Texas A&MI liniersits inA-ugust 1985 as an Associate Professor and ajs promoted to Profes-sor in 1988. In ianuar-, 19901. he \&as appointed E-Svstcnis EndowkedProfessor of Electrical Engineering. His current interests, arc irnmicrow as and millimneter-wave devices and circuits.miowa
Julio A-. Nakarro wkas born in oirdolua. Ar- integrated circuits, microwave optical Interaction'. .ind antennlas.izent ma, on Nosember 13. lI 44 Hie reccused Dr. (hani! served as the editor iot the four-volumne Hantdbookthe 1.S.E E. devree from Texas .- &M Umi- tr%%aiii andti Optical Components. published b% John Wiley &
ecrsit's fi MN 1988. He has been at coopera- Sons. Inc.. in 1989 and 199(0. He is the editor of Microiau Cand
tise education student with General Dvnam- Optical Technoho~-u- Letters and of the Wiley, Book Series on Mli-- cs-Fort Worth since Ma\ l1-485. erovave and Optical Engineering. He has published over 1001 tech-
lie has, worked in as ionic ,%stems design. nical papers and several book chapters in the areas of miicroasecadsanced technologs and s\stemN engineer- and millimeter-w-ave devices and circuits. Dr. Chang received theimg. field engineering, antenna s\stems. and Special Achievement Aw&ard fromt TRW in 1984. the Halliburton
4radar cross .section research. As a master's Professor Award in 1988. and the Distinguished Teaching Award incandid~itc at Texas .A&M Lnisersit. he has 198,Q from the Texas A&M lUni'sersits.
59
