Medium-Voltage Impedance Measurement Unit for Assessing

IEEE TRANSACTIONS ON ENERGY CONVERSION, VOL. 32, NO. 2, JUNE 2017 829

Medium-Voltage Impedance Measurement Unit forAssessing the System Stability of Electric Ships

Marko Jaksic, Member, IEEE, Zhiyu Shen, Member, IEEE, Igor Cvetkovic, Student Member, IEEE,Dushan Boroyevich, Fellow, IEEE, Rolando Burgos, Member, IEEE, Christina DiMarino, Student Member, IEEE,

and Fang Chen, Student Member, IEEE

Abstract—This paper describes the design and implementationof the first medium-voltage impedance measurement unit (IMU)capable of characterizing in situ source and load impedances ofdc and ac networks (4160 V ac, 6000 V dc, 300 A, 2.2 MVA) inthe frequency range of 0.1 Hz–1 kHz. The IMU comprises threepower electronics building blocks (PEBBs), each built using 10-kVSiC MOSFET H-bridges. The modularity of the PEBBs allows forboth series and shunt perturbation injection modes to be realized,as both injection modes are needed to accurately predict the sta-bility of the electrical system. The effectiveness of the proposedimpedance identification approach is experimentally verified onmedium voltage power grid.

Index Terms—AC–DC power converter, impedance mea-surements, interleaved converter, more electric ship, multi-level converter, silicon carbide (SiC), stability analysis, systemidentification.

I. INTRODUCTION

THE requirements for improved reliability and high surviv-ability of shipboard power systems have steered the devel-

opment of medium-voltage ac (MVAC), and medium-voltagedc (MVDC) systems as direct replacements of the conventionallow-voltage generation and distribution practice [1]. This is pre-dominantly the case with MVDC, as it is seen as a key enablingtechnology for all-electric ships [2].

However, the increased use of power electronics, althoughoffering necessary means for an advanced and flexible energyutilization, is fundamentally changing the nature of the ship-board power system sources and loads, inflicting low and highfrequency dynamic interactions that did not exist in the sys-tem before. Although power electronics converters make loadsmore robust to variations of voltage and frequency, they presenta negative incremental resistance behavior, which may initiatesundesirable low-frequency dynamic interactions [3]. In order tobetter understand design, and dynamically control future elec-tronic systems on all-electric ships, it is required to develop

Manuscript received June 16, 2016; revised December 9, 2016; acceptedFebruary 15, 2017. Date of publication April 7, 2017; date of current versionMay 18, 2017.

M. Jaksic is with the General Motor,, Detroit, , MI 48265 USA (e-mail:[email protected];).

Z. Shen, I. Cvetkovic, D. Boroyevich, R. Burgos, C. DiMarino, and F. Chenare with the Virginia Tech, Blacksburg, VA 24061 USA (e-mail: [email protected];[email protected]; [email protected]; [email protected]; [email protected]@vt.edu;[email protected] ).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org

Digital Object Identifier 10.1109/TEC.2017.2692275

innovative concepts that offer great insights into converter andsystem level behavior.

Recently a lot of research is being conducted in the area ofdesigning advanced control concepts to regulate micro-grids[4]–[6], which are penetrating classical power systems andchanging the nature of power grid. Due to the more complex in-teractions among newly installed smart converters, potential in-stability issues might arise. Therefore, impedance measurementconcepts for the online assessment of the micro-grid stabilityare equally applicable to the renewable micro-grids as well asto the next generation ship power systems.

One such concept is a small-signal stability assessment[7], [8]. Specifically, the system stability can be determinedat any ac interface by means of the Generalized Nyquist stabil-ity Criterion (GNC) [9], [10] based on the characteristic loci ofthe return ratio matrix L(s). For three-phase electrical systemsthis impedance is preferably expressed in the synchronous d-qframe. At dc interfaces, the equivalent, simpler formulation isgiven with a ratio of source and load impedances. In both in-stances, the concept requires measuring the source-output (ZS )and load-input (ZL = 1/YL ) impedances at the desired inter-face point. For a stable system, the respective loci of (1) and (2)must not encircle the critical point (–1, 0).[λ1(s)λ2(s)

]= eig (L(s)) = eig (ZS(s) · YL(s)) (1)

ZS (s) =

[Zdd (s) Zdq (s)Zqd (s) Zqq (s)

]; YL (s) =

[Ydd (s) Ydq (s)Yqd (s) Yqq (s)

]

(2)

In order to design a robust power system and to guaranteestable operation for a wide operating range, impedance inter-actions at various interfaces should be analyzed with an in-situimpedance measurement unit (IMU). Although, system inter-actions in ac power systems can be analyzed in several dif-ferent ways, in the essence all the methods rely on measuringsmall-signal source and load impedances [11]–[16]. In a dualmanner, it is also possible to examine the small-signal stabil-ity of autonomous hybrid power systems using a time-domainapproach [17].

Several automated IMU have already been presented in theliterature. It has been shown that voltage amplifiers can be usedto inject shunt current via transformers [18]. The identification

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830 IEEE TRANSACTIONS ON ENERGY CONVERSION, VOL. 32, NO. 2, JUNE 2017

TABLE IIMU SPECIFICATION

AC Impedance Measurements DC Impedance Measurements

System frequency: 60 and 400 Hz System frequency: DCMeasurement frequency range:0.1 Hz – 1 kHz

Measurement frequency range:0.1 Hz – 1 kHz

System voltage: 4160 V (line-to-line) System voltage: 6000 VSystem current: 300 A (rms) System current: 300 A

of the small-signal impedances in this case is performed throughthe commercially available network analyzer and automationof the process is achieved through PC computer and custominterface board.

An automated IMU capable of shunt current and series volt-age injection using a chirp signal was presented in [19], [20]. Thesmall-signal identification of the impedances was performedvia Welch’s cross-correlation technique, which is coded in theunit’s computer and automated through a custom measurementplatform. In this IMU, the wide-bandwidth signal is used tocharacterize a 480 V ac power system, showing the reductionin measurement time and extraction of impedances at the in-creased number of frequency points. However, the presentedsolution uses a three-phase two-level boost converter for thegeneration of injection currents and voltages, which is not anideal candidate for the use in medium voltage systems. In thepast several years, a lot of the research activities were focused onidentifying small signal impedances of power converter and/orsystems [21]–[26].

Additionally identification of fundamental line impedances inmicro-grid systems is being researched as well [27]. Similarly,it is possible to apply FFT algorithm to characterize commonmode impedance of inverter feed motor drives as described in[28]. Wide bandwidth signals are very popular solution for theimpedance identification as the measurement time is signifi-cantly reduced. Therefore, the wide bandwidth injection sig-nals are widely used for the characterization of small-signalimpedances at dc interfaces as well [29].

In order to characterize both source and load precisely, itis necessary to have the capability to inject shunt current aswell as voltage in series. If a big mismatch between source andload impedance magnitudes is present in a frequency range,then shunt current injection provides more precise characteri-zation of small impedances. On the other hand, series voltageinjection provides more precise characterization of the largeimpedances, as most of the injection signal would excite thatside of the system.

This paper presents the first hardware implementation ofmeasurement equipment capable of identifying small-signal dqimpedances of MVAC and MVDC systems using a single-phasewide-bandwidth injection algorithm.

II. IMU SYSTEM ARCHITECTURE

The IMU specifications are listed in the Table I . To extractimpedances from a system, the IMU has to be connected at thedesired interface point as shown in Fig. 1, where it would inject

Fig. 1. Illustration showing IMU insertion into the all-electric ship MVDCdistribution system (simplified).

Fig. 2. Simplified functional block diagram of the IMU.

small perturbations, measure voltage and current responses, andcalculate impedances of both source and load sides while thesystem is operating at a point of interest.

Fig. 2 shows the simplified functional block diagram of theIMU, where the four main groups are linked via Ethernet com-munication, and analog/digital signals. After inserting the IMUinto the system, the host computer (HC) generates the perturba-tion reference and loads it into the embedded PIU controller. ThePIU controller then directs all three power electronics buildingblocks (PEBBs) to inject perturbation into the system. The sys-tem response is captured by high-bandwidth voltage and currentsensors on the source and load side. The signal interface unit(SIU) receives these measurements and sends them to the PXIindustrial computer and HC for post-processing and impedanceidentification.

The core of the IMU is the perturbation injection unit (PIU),which comprises three PEBBs connected in series or parallel,depending on the desired mode of operation. In the shunt cur-rent injection mode, the three PEBBs are configured in seriesand operate as a single-phase cascaded multi-level converter asshown in Fig. 3. During this mode, the PIU injects about 5% ofthe system current between system phase A and phase B in acnetworks, or dc+ and dc- for dc networks. This configurationgives significantly better results when characterization of thesource output impedance is of interest, since the majority of theperturbation current flows into the source.

In the series voltage injection mode, three PEBBs are con-nected in parallel, forming a single-phase modular interleavedconverter as shown in Fig. 4. In this mode of operation, theIMU injects maximum 5% of the nominal system line-to-neutralvoltage. Contrary to the shunt current injection mode, the series

JAKSIC et al.: MEDIUM-VOLTAGE IMPEDANCE MEASUREMENT UNIT 831

Fig. 3. Shunt current configuration mode of the perturbation injection unit.

Fig. 4. Series voltage configuration mode of the perturbation injection unit.

voltage injection mode typically yields better results for the loadinput impedance.

The PEBBs constructed in this work, are each comprisingtwo 10 kV, 120 A SiC MOSFET half-bridge modules [30] inthe H-bridge configuration, gate drivers, decoupling and dc-linkcapacitors, arm inductors, and a 6.5 kV IGBT that serves as aprotection device. Fig. 5(a) shows the topology of the powerstage. Each PEBB also features a high-speed digital controller,an uninterruptable power supply, and liquid cooling. The as-sembled PEBB is shown in Fig. 5(b). The developed PEBBhas a volume of 0.69 m3 , and a power density of 1MW/m3 .This high power density is greatly due to the fast switching of

Fig. 5. PEBB (a) topology, and (b) hardware.

Fig. 6. Assembled IMU cabinet in the power lab environment.

the 10 kV SiC MOSFET module. The modularity of PEBBs,combined with the fast-switching and high-efficiency of 10 kVSiC MOSFETs,results in high-power-density, light-weight, low-maintenance, versatile units that are ideal for the power systemsof next-generation electric ships. In this work, three PEBBs,each consisting of a 10 kV, 120 A SiC MOSFET H-bridge, weredesigned, constructed and tested. The PEBBs can be operated inseries and parallel to scale the voltage and current, respectively.

Fig. 6 shows the final assembled IMU and in the left cabinetPIU is assembled. The top part is bus structure for reconfiguringbetween shunt and series injection mode.TableII The middlepart is PEBBs and the bottom part is energy storage capacitorsto support low frequency injection. The SIU and PXI computerare located in the IMU control cabinet. The host computer isoutside the area at operator’s room.


TABLE IIPEBB COMPONENT PARAMETERS

Output inductance 2.4 mH Switching frequency 10 kHz

DC capacitance 1.8 mF Output current 120 A rmsAverage DC voltage 4 kV Peak DC voltage 4.7 kV

III. CONTROLS OF MODULAR INJECTION CONVERTER

The controls strategies of the proposed injection convert-ers are presented and analyzed in depth in this section. Twooperational modes exist for the MV converter, shunt currentinjection and series voltage injection, both requiring regula-tion of injection signals and high frequency bandwidth to allowprecise generation of high frequency components. In addition,precise regulation of dc voltage of H-bridge modules is also re-quired. Although analysis for the both controllers is presented ins-domain, time delays of digital modulator and one switchingperiod delay due to microcontroller implementation are addedfor the completeness of the results. The equivalent z-domaincontrollers, which are coded into the converter’s controller, arecalculated using Tustin transformation. Finally, modulation sig-nals of PEBBs are phase interleaved to reduce switching ripplein the injection signals, yielding measurement results that areless corrupted with unwanted background noise.

A. Controller of Shunt Current Injection Converter

In order to extend operational limits of the converter, multi-level structure is used in shunt current injection mode. The mainbenefit is that a multi-level converter increases the equivalentvoltage applied to output inductors by a factor of N, where Nis the number of H-bridge modules. This is specifically impor-tant for the injection of high frequency perturbation current.During the generation of high frequency current perturbationsignal, the impedance of ac filter inductor is high, requiringhigh voltage output from H-bridges to compensate for the volt-age drop across filter inductors. On the other hand, the outputof H-bridge switches is directly proportional to the modulationsignal, which is limited in its nature due to the comparisonwith carrier signal. The purpose of shunt current injection con-verter is to generate excitation frequencies, with the minimumnumber of side harmonics. Therefore, the operation of the con-verter in the over-modulation region should be avoided, as itresults in the generation of a large number of sideband har-monics, which can create unwanted cross-talk, resulting in theerroneous impedance measurements.

The implemented control strategy requires regulation of injec-tion current, as well as frequency synchronization of injectioncurrent with the line frequency of the ship. The synchroniza-tion of the injection current and the voltage line frequency isachieved via phase locked-loop (PLL). Additionally, it is neces-sary to regulate dc voltages of the H-bridge PEBBs to guaranteestable operation during the injection. The complete block dia-gram of implemented control for shunt current injection modeis shown in Fig. 7.

Fig. 7. Shunt current injection control scheme with outer dc voltage loop,high frequency inner current loop, PLL synchronization, and feed forward part.

As it can be observed, the dc voltage regulator Cvdc(s) regu-lates average value of the PEBB’s dc voltages. The PLL blockprovides a sinusoidal signal that is multiplied with the output ofthe dc regulator, where the input of the PLL is a voltage withpoint of common coupling vpcc . This term will provide a linefrequency reference to a current controller, which is directlyresponsible for the regulation, charging and discharging of dccapacitors. In steady state operation, the line frequency term ofthe inductor current should be just large enough to cover thesemiconductor losses of the H-bridges as well as the losses oc-curring in the passive components. The plant transfer functionfor the dc voltage loop can be derived from a power balancemodeling. The power drawn at the point of common couplingwill be equal to the sum of the converter’s power losses and thepower dissipated on the dc side. Assuming low power lossesoccurring in the converter, the previous power balance principlecan be expressed as

NVdcIdc = VsrmsIlf rms (3)

Where N is the number of H-bridge modules placed in series,Vdc is the dc voltage of the capacitors, Idc is equivalent dc currentdrawn by each H-bridge module. The plant transfer function,which models dynamics from the inductor ilf magnitude to thesum of dc voltages is obtained by linearization of the powerbalance principle.

Gvdci (s) =vdc (s)ilf m (s)

=Vsrms√2NVdc

Zdc (s) (4)

Tvdci (s) = Gvdci (s)Cvdc (s) (5)

Where controller Cvdc(s) is typically designed to have anintegrator with gain of kivdc , together with a single zero ωzvdc

and single pole ωpvdc . The dc voltage loop is designed to havelow bandwidth, thus the dynamic of the inner current loop canbe approximated with an ideal transfer function that has unitygain and zero phase delay.

The generation of the perturbation current is controlledthrough the high frequency current loop. Signal per is usedto provide current reference to the controller, so that convertercan perturb the system. The bandwidth of the control loop isset at 3 kHz to ensure high loop gain up to 1 kHz, which is


the maximum injection frequency specified for this design. Thecurrent loop gain in the first approximation can be expressedanalytically as written below.

Gid(s) =NVdc

3sLf(6)

Tilf (s) = Gid(s)Cilf (s) =NVdc

3sLf

ki

s

(s + ωz1) (s + ωz2)(s + ωp1) (s + ωp2)

(7)

Where Lf is the inductance of the ac output inductors. Plantduty cycle to output transfer function Gid(s) is approximated byassuming static inductive ac filter load. The current compensatorCilf (s) is a typical controller, which consists of an integratorwith gain ki , two low frequency zeros ωz1 , ωz2 and two highfrequency poles ωp1 , ωp2 to set a desired phase margin. A moreaccurate current loop gain expression of the digital controller isobtained if the digital modulator delays are added.

Tilf del(s) = Gdel(s)Tilf (s) (8)

Gdel(s) = e−Td e l s ; Tdel = 1.5Tsw (9)

The second order Pade’s approximation is a sufficiently pre-cise simplification of the delay transfer function, as it providesan accurate estimation of the digital modulator delays in thefrequency range of interest.

Gdel(s) =1 − Td e l

2 s + T 2d e l

12 s2

1 + 1Td e l

2 s + T 2d e l

12 s2(10)

The specified injection signal, which can be either sinusoidal,chirp or multi-tone, is generated automatically on the HC andcommunicated via Ethernet protocol to the control card. The per-turbation reference for the current compensator block Cilf (s) isset by an internal signal ilf per . In this way, the current controlwill ensure generation of line frequency signal, as well as spec-ified sinusoidal or wide-bandwidth injection signals. In order toensure frequency decoupling of the dc voltage and current loops,and prevent frequency talk between the two loops, it is desiredto keep the bandwidth of the dc voltage loop significantly lowerthan the bandwidth of the current loop.

The balancing control part needs to compensate for themismatch between the H-bridge modules and the other non-idealities in the hardware that could cause different propagationtime delays, while providing the same dc voltage on each of theconverter’s capacitors. Straight forward implementation of thebalancing controller utilizes a classical PI controller.

Cbal(s) = kpbal +kibal

s(11)

Based on the sign of the injection current, the controller willdetermine the small delta values of the duty cycles to maintainbalancing between the dc voltages. It should be noted here thatonly two balancing controllers are needed as the third delta valueof the duty cycle is equal to the negative sum of the first twodelta duty cycle values.

Δd3 = − (Δd1 + Δd2) (12)

TABLE IIICONTROLLER PARAMETERS FOR THE SHUNT CURRENT INJECTION MODE

Balancingcontroller

Current controller DC voltage controller

kp b a l = 0.008; ki = 6036.4; kiv d c = 15.386;kib a l = 0.25; ω z 1 = 85.8 rad/s,

ω z 2 = 1741 rad/ sω z v d c = 4.319 rad/s;

ωp 1 = 7720 rad/s,ωp 2 = 2.425 105 rad/s;

ωpv d c = 186.2 rad/s;

Performance of the balancing controller is extensively testedin simulations using the equivalent switching model of the con-verter, while the final controller’s values for proportional andintegrator coefficients are fine-tuned experimentally. The con-troller parameters that are used to regulate injection converterin the shunt current mode are given in Table III.

B. Controller of Series Voltage Injection Converter

The objective of this section is to show control structure and todemonstrate effectiveness of the proposed decoupling control.The output of the converter is series inserted between a sourceand load via a capacitor Cf . Desired mode of the operation forthe converter is to inject a perturbation voltage to the system un-der test, while conducting system current. Additionally, controlneeds to regulate dc voltage of each PEBBs, providing a steadyoperating point for the converter. Ideally, the injection convertershould not modify system current and voltages, it should justtake a small amount of power to cover for its losses. The typi-cal control of transformer-less dynamic voltage restorers, whichconsists of single H-bridge modules, applies similar decouplingprinciple [31], [32].

In order to satisfy before mentioned requirements, a decou-pling controller is implemented as shown in Fig. 8. It relies onsensing the converter’s states, which are used as feedback sig-nals, to achieve cancellation of converter’s dynamic. However,due to the delays of digital control, converter’s parameters un-certainty, complete cancellation of converter’s dynamic is notpossible.

The control of average value dc voltages is achieved via thecontroller Cvdc(s). The output of the controller is multipliedwith phase information extracted from system current isys , pro-viding a voltage reference. This part of the control will createline frequency component in inserted voltage vcf , which willbe aligned with system current, making sure that active poweris drawn for maintaining the dc capacitor’s voltages.

The rest of the controller can be divided into two parts. Decou-pling control is implemented with controller Cvcf (s), Cilf (s)and by three dividers. The balancing part for dc voltages is im-plemented with two balancing controller Cbal(s), where outputsof the controllers will determine small delta values for internalcurrent references. The outer voltage loop of the seris insertedvoltage vcf is designed to have a bandwidth of 2 kHz, while in-ner current loops are designed to be around 1 kHz. The controlof dc voltages is frequency decoupled from the rest of control,by setting its bandwidth to be around 1 Hz.


Fig. 8. Decoupling control scheme for series voltage injection mode with outer dc voltage loop, high-frequency inner current loop, high frequency ac voltageloop, PLL synchronization, and vdc voltage divider.

A detailed design and analysis of the proposed decouplingcontroller for the series voltage injection is already presented onthe scaled-down hardware prototype [33]. Due to the modularityof converter, the designed controller can be easily scaled for themedium voltage grids.

It should be noted that power converter is designed to supportboth shunt current and series voltage injection into three-phaseac and dc networks, making it suitable for use in the future moreelectric ship power systems.

IV. IMPEDANCE IDENTIFICATION ALGORITHM

This paper extends the previous work done on single-phaseinjection [34], and proposes the injection of chirp and multi-tone currents and voltages via the modular injection converter.In addition, the cross correlation algorithm used with wide-bandwidth signals reduces the background noise and yields moreaccurate impedance results.

The main task of the identification algorithms, is to ex-tract useful information from the sensed current and voltageresponses and perform accurate identification of small-signalcharacteristics in the presence of background noise. Therefore,this section presents the implemented identification algorithms,which is based on cross-correlation identification using theWelch’s method [35].

The advantages of the cross-correlation algorithm are itsavailability in a number of computational software packages,and its relatively straight-forward implementation. In addition,Welch’s method is based on windowing and overlapping per-turbation responses and averaging calculated FFTs. In this way,background noise in the measurements can be effectively re-duced, yielding a precise identification of dq impedances withwide-bandwidth signals.

In order to identify small-signal dq impedances, two sets ofindependent perturbation signals need to be injected into thethree-phase ac interface.

Welch’s method is used to characterize the voltage and currentresponses with respect to either the injected current or voltageperturbation. The current perturbation is generated by the con-verter’s control using the clean signal reference. In this way, itis possible to identify the voltage and current responses with re-gard to the ideal current reference, requiring less measurementsand removing the uncorrelated noise. The signal references canbe an ideal sinusoidal, multi-tone or chirp signal generated inthe digital form inside a DSP card. The reference has a cleanspectrum which is not corrupted with noise and is thus suitablefor the response identification via cross-correlation methods.

In order to identify the small-signal impedances, two indepen-dent perturbations are generated, and the corresponding voltageand current responses are collected. The general relationshipbetween the steady-state responses and reference signals in thesynchronous reference frame is written below.

[vd1 (s)vq1 (s)

]=

[Gv11 (s) Gv12 (s)Gv21 (s) Gv22 (s)

] [rsin (s)rcos (s)

](13)

[vd2 (s)vq2 (s)

]=

[Gv11 (s) Gv12 (s)Gv21 (s) Gv22 (s)

] [rsin (s)−rcos (s)

](14)

Where rsin(s) and rcos(s) represent the Laplace transform ofthe used reference signal in both injections, which can be ei-ther sinusoidal, chirp of multi-tone signal as described above. Ifwide-bandwidth signals are used for the injection, then severalexcitation frequencies can be calculated simultaneously usingthe described algorithm. Due to the nature of the single-phaseinjection, both reference vectors consist of two orthogonal com-ponents, resulting in two independent perturbation vectors. Inorder to identify four small-signal transfer functions Gv11(s),Gv12(s), Gv21(s) and Gv22(s), the following transfer functions


are identified first using Welch’s method.

Gvd1sm (s) =vd1 (s)rsin(s)

= Gv11 (s) + Gv12 (s)rcos (s)rsin (s)

Gvd1cm (s) =vd1 (s)rcos(s)

= Gv11 (s)rsin (s)rcos (s)

+ Gv12 (s) (15)

Gvq1sm (s) =vq1 (s)rsin(s)

= Gv21 (s) + Gv22 (s)rcos (s)rsin (s)

Gvq1cm (s) =vq1 (s)rcos(s)


+ Gv22 (s) (16)

Similarly, one more set of transfer functions is identified atthe injection frequency points capturing voltage responses ofthe second perturbation and using the corresponding orthogonalset of references rsin(s) and rcos(s).

Gvd2sm (s) =vd2 (s)rsin(s)

= Gv11 (s) − Gv12 (s)rcos (s)rsin (s)

Gvd2cm (s) =vd2 (s)rcos(s)


− Gv12 (s) (17)

Gvq2sm (s) =vq2 (s)rsin(s)

= Gv21 (s) − Gv22 (s)rcos (s)rsin (s)

Gvq2cm (s) =vq2 (s)rcos(s)


− Gv22 (s) (18)

In this way, the digital reference signal is used as the input,and the uncorrelated noise present in the responses is attenuated.Finally, the previous set of equations are used to calculate eightsmall-signal transfer functions. It should be noted that all eightsmall-signal transfer functions require capturing time domaindata of voltage or current responses in the synchronous referenceframe, and saving the time domain data of the used orthogonalset of reference signal. The identification of the desired fourtransfer functions is obtained by algebraic manipulations of thealready-identified transfer functions.

Gv11 (s) = Gvd1sm (s) + Gvd2sm (s)

Gv12 (s) = Gvd1cm (s) − Gvd2cm (s) (19)

Gv21 (s) = Gvq1sm (s) + Gvq2sm (s)

Gv22 (s) = Gvq1cm (s) − Gvq2cm (s) (20)

Additionally using the same procedure, the source and loadcurrent responses with respect to the reference are character-ized using the cross-correlation technique. Therefore, the small-signal dq impedances of source and load Zsdq (s) and Zldq (s)can be calculated in the following way.

Zsdq (s) =[Gv11 (s) Gv12 (s)Gv21 (s) Gv22 (s)

] [Gis11 (s) Gis12 (s)Gis21 (s) Gis22 (s)

]−1

(21)

Zldq (s) =[Gv11 (s) Gv12 (s)Gv21 (s) Gv22 (s)

] [Gil11 (s) Gil12 (s)Gil21 (s) Gil22 (s)

]−1

(22)

The presented identification algorithm is derived for the shuntcurrent injection case. However, from the identification algo-rithm point of view, the series injection mode is dual to theshunt current injection mode. Thus, in the series injection mode,the algorithm would derive the current matrix transfer functionGi(s) as a common part for the source and load impedances.In addition, after identifying two more voltage matrices Gvs(s)and Gvl(s), the source and load dq impedance matrices arecalculated in an analog way as below.

Zsdq (s) =[Gvs11 (s) Gvs12 (s)Gvs21 (s) Gvs22 (s)

] [Gi11 (s) Gi12 (s)Gi21 (s) Gi22 (s)

]−1

(23)

Zldq (s) = −[Gvl11 (s) Gvl12 (s)Gvl21 (s) Gvl22 (s)

] [Gi11 (s) Gi12 (s)Gi21 (s) Gi22 (s)

]−1

(24)

Both presented algorithms are coded into the impedance mea-surement unit and used for online impedance calculation. Theclassical cross-correlation algorithm is modified to allow extrac-tion of dq impedances using the single-phase injection concept.

V. MEDIUM VOLTAGE TEST RESULTS

The experimental verification of medium voltage IMU wascarried out at Center for advanced power systems (CAPS) usingthe test facility [36]. The CAPS testing facility can be usedfor the development and testing of state-of-the-art MVDC andMVAC electric ship’s power system components. The developedIMU is used to characterize a variable voltage source (VVS)feeding a resistive load at medium voltage ranges.

A. System Architecture

The acquisition of the currents and voltages for the sourceand load sides is done via six high precision current sensors IT600-S and six high precision voltage sensors DV 2800/SP4. Fur-thermore, the acquisition unit uses a PXI controller 8135 for theautomation of the measurement process, and a high-precisionPXI 6368 data acquisition card for the real-time monitoring ofthe voltage and current responses.

Three voltage and current responses on both the source andload sides are oversampled with 2 MHz clock inside the acqui-sition card and down-sampled to 20 kHz using a digital filter inthe monitoring code, reducing the noise in the sensed signals.The entire measurement process is further automated via Eth-ernet communication between the PXI controller and the DSPcontroller of the injection converter. As the master controller,the PXI sends the commands to the injection controller and con-trols the type, magnitude and frequency of the injection signals.In this way, the unit is suitable to monitor and perform the on-line estimation of the impedances in the contemporary mediumvoltage power systems.

B. Experimental IMU Waveforms

An example of series voltage injection waveform, which isgenerated by the IMU converter operating in the series voltage


Fig. 9. Series inserted voltage vp that is injected into the system: (a) timedomain and (b) frequency domain.

mode, is show in Fig. 9(a). In the presented test-case 6 frequencypoints, namely 362, 372, 382, 390, 400 and 410 Hz, are usedas the excitation signal. It is interesting to check the frequencyspectrum of the injection signal, which is plotted in Fig. 9(b).Beside the injection frequency components, there is a dominantsystem frequency component, 60 Hz frequency that is used tomaintain dc capacitor voltages for each H-bridge.

Due to the usage of the switching converter as the injectiondevice, sideband harmonics are generated as well, clearly no-ticeable in the shown picture. Although, the primary goal ofthe injection converter is to support the operation and systemcharacterization at the medium-voltage range, undesired side-band and switching harmonics are generated as a product of theconverter’s switching nature. Equivalent switching frequencyis increased six times to 60 kHz by applying the interleavedmodulation signals to three H-bridges. Additionally, due to thedead-time insertion for the phase-leg switching, low frequencysideband harmonics are inevitable. For that reason sweeping al-gorithm is used in the characterization process, injecting 5 to6 frequency points between two line harmonics and coveringcomplete measurement range.

In this way, cross talk between excitation frequencies andsideband harmonics is minimized to an acceptable level. Itshould be noted here that due to the hardware imperfections,generation of the unwanted sideband harmonics cannot beavoided. Therefore, the converter has been overdesigned withrespect to the typical power converter designs, making it more

Fig. 10. Load dq voltage response to the multitone excitation.

Fig. 11. Spectrum of load dq voltages: (top) d voltage, and (bottom) q voltage.

suitable for the injection of “cleaner” perturbation signals. Es-sentially, the precision of IMU is increased, which is a criticalunit property to perform accurate measurements in the mediumvoltage range.

Voltage response of the load side to the discussed excitation,which is obtained via IMU acquisition subsystem, is shown inFig. 10. IMU is designed to excite single-phase of the systemunder test, yielding the unbalanced system response in abc co-ordinates. The injection asymmetry generates more side-bandharmonics and shows up in both d and q axis simultaneously. Asa direct consequence, the system responses are more pollutedwith the additional frequency components. Spectrum compo-nent that are present in both load d and q axis voltages, areshown in Fig. 11.

Obviously, single-phase injection generates an increasednumber of frequency components due to the asymmetrical ex-citation. Nevertheless, the identification algorithm used to cal-culate small-signal dq impedances is capable of suppressing thebackground noise and providing relatively “clean” impedanceresults. Partially, due to the oversampling in the acquisition unitand also due to the usage of cross-spectrum correlation algo-rithm, as both are reducing the influence of background noise.Even though, the power converter is not the most ideal injectiondevice for the impedance measurements, due to the developmentof high voltage SiC switches it can definitely perturb mediumvoltage systems with relatively “clean” signals. Thus, SiC de-vices are potentially bringing new application to the switching


Fig. 12. Identified Zsdd (s), Zsdq (s), Zsqd (s), and Zsq q (s) source small signal dq impedances with shunt current injection.

Fig. 13. Identified Zldd (s), Zldq (s), Zlq d (s), and Zlq q (s) load small signal dq impedances with shunt current injection.


Fig. 14. Identified Zsdd (s), Zsdq (s), Zsqd (s), and Zsq q (s) source small signal dq impedances with series voltage injection.

Fig. 15. Identified Zldd (s), Zldq (s), Zlq d (s), and Zlq q (s) load small signal dq impedances with series voltage injection.


power converters, medium voltage injection amplifiers for theonline impedance measurements.

C. Experimental Small-Signal dq Impedances

The IMU was first connected in shunt current injection mode,while the VVS was supplying a resistive load with 2.5 kV,60 Hz three-phase line voltage. The injection converter wasused to excite the system with multi-tone currents in the desiredfrequency range. Basically, the frequency sweeping algorithmis used to perturb the source and load with 5 to 6 frequencypoints in each frequency range, yielding the impedance resultsat 105 points. The source and load impedances are extractedin the specified frequency range (5 Hz–1 kHz) as plotted inFigs. 12 and 13. Although, there is some noise present in themeasured impedances, it is obvious that the obtained resultsare very clean. Due to the fact that shunt current would mainlyexcite the source side, the cleaner source impedance waveformsare extracted.

It should be noted here that the total time for IMU to per-form the described signal injection, charging and dischargingthe converter, signal sweeping and post processing is around15 minutes. Due to the fact that multi-tone injection, which ex-cites five to six points simultaneously, is used significant timereduction is achieved. In comparison, sweeping just pure sinu-soidal signal would take 5 to 6 longer time to obtain completesmall-signal impedances.

In the second test case, the impedance measurements at2.8 kV, 60 Hz line voltage have been achieved using the IMU inthe series connection mode. Similarly, multi-tone voltage wasinserted between the source and load, providing a small-signalperturbation to the system. Extracted source and load impedanceresults of the system under test are shown below in Figs. 14and 15, respectively. As expected, the series voltage injectionyields cleaner results of the load dq impedances. As the sourceimpedance is relatively low in the low frequency range, there isa noticeable error present in the impedance results. Therefore,the series injection mode should be used only for the character-ization of the high impedance values, which is a load side in theused test case.

As stated before, simple resistive network with large inductorsare used as a load to medium-voltage VVS. Passive load thatconsists of the resistance R and inductance L can be describedanalytically as provided below.

ZR (s) =[R + sL −ωL

ωL R + sL

](25)

Where ω represents angular frequency of the power systemunder test. It should be noted that both resistance (633.14 Ω) andinductance (27.9 mH) values of the passive load are capturedwith the developed IMU. The extracted values for the passiveload are very close to the specified values, experimentally ver-ifying usability of the developed and built IMU. On the otherhand VVS source exhibits typical power system impedance,which would be mainly inductive in the nature.

ZS (s) =[Rs + sLs −ωLs

ωLs Rs + sLs

](26)

Where Rs and Ls represent parasitic resistance and induc-tance of the source. In a similar manner, extracted impedanceresults can be used to identify Rs and Ls values, using iden-tification toolbox in a number of the commercially availablesoftware packages. As the next step in a validation of the de-veloped IMU, more complex state of the art ship architecturescould be used.

To the best of the authors’ knowledges, this is the first report ofonline system impedance measurements using medium voltageIMU, which was achieved using 10 kV SiC MOSFETs designedas part of PEBBs.

VI. CONCLUSION

This paper proposes the injection of single-phase wide band-width signals for identification of small-signal dq impedancesin new generation electric ships. The identification algorithm isbased on cross correlation technique, providing effective wayto reduce the noise influence in comparison to classical FFTalgorithm. The developed impedance measurement unit, pro-vides a capability to perform in-situ identification of systemdq impedances at various three-phase ac and dc interfaces. Inaddition, the single-phase injection of perturbation current orvoltage is performed using a modular SiC converter, resultingin minimized hardware size, weight and complexity. Due to themodularity and scalability of the proposed injection convertersolution, the developed IMU can be used at several different lowand medium voltage levels.

Furthermore, the proposed solution is capable of injectingsinusoidal, chirp or multi-tone signals, providing flexibility interms of measurement speed, SNR and identification precision.The injection of wide-bandwidth signals reduces the measure-ment time significantly, while yielding accurate dq impedanceresults. Finally, IMU is experimentally validated on mediumvoltage VVS supplying a resistive load with the injection ofsingle-phase wide-bandwidth signals.

The medium voltage IMU developed in this work can un-doubtedly aid the design of advanced Navy shipboard platformswith contemporary all-electric architecture. The IMU is a nec-essary technology for the development of cutting-edge hybridMVAC and MVDC power systems for future electric ships thatfeature high reliability and survivability without compromisingstability of the system.

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Marko Jaksic (S’10–M’15) received the Dipl.Ing.and M.S. degrees from the University of Belgrade,Belgrade, Serbia, in 2007 and 2010, respectively, andthe Ph.D. degree from Virginia Polytechnic Instituteand State University (Virginia Tech), Blacksburg, VA,USA, in 2014.

From 2009 to 2014, he was a Research as-sistant in the Center for Power Electronics Sys-tems, Virginia Polytechnic Institute and State Uni-versity. Since 2014, he has been a Specification An-alyst/Power Electronics Engineer in General Motors

Global Propulsion Systems, Pontiac, MI, USA. His research interests includemodeling of dc/dq impedances of power converters, wide-bandwidth signalinjection, impedance identification, control and stability analysis of power sys-tems, and design and analysis of SiC traction power inverter modules for thenext generation electric vehicles.

Zhiyu Shen (S’11–M’13) received the B.S. and M.S.degrees in electrical engineering from Tsinghua Uni-versity, Beijing, China and the Ph.D. degree in elec-trical engineering from Virginia Tech, Blacksburg,VA, USA, in 2004, 2007, and 2013, respectively.

He is currently a Research Scientist in the Cen-ter for Power Electronics System, Virginia Tech.His research interests include three-phase ac sys-tem impedance measurement, three-phase ac systemsmall signal stability, high-frequency high-densityconverter design, and control system architecture in

high-power converters.

Igor Cvetkovic (S’10) received the Dipl.Ing. degreein power systems from the University of Belgrade,Belgrade, Serbia, in 2004, and the M.S. degree fromthe Center for Power Electronics Systems, VirginiaPolytechnic Institute and State University (VirginiaTech), Blacksburg, VA, USA, in 2010, where he iscurrently working toward the Ph.D. degree.

Before joining Virginia Tech for higher education,he was in the Electric Power Industry of Serbia. Hisresearch interests include dc-electronic power distri-bution systems stability and design, as well as power

electronics systems modeling and control.


Dushan Boroyevich (S’81–M’86–SM’03–F’06) re-ceived the Dipl.Ing. degree from the University ofBelgrade, Belgrade, Serbia, in 1976, the M.S. degreefrom the University of Novi Sad, Novi Sad, Serbia,in 1982, and the Ph.D. degree, in 1986, from VirginiaPolytechnic Institute and State University (VirginiaTech), Blacksburg, VA, USA.

From 1986 to 1990, he was an Assistant Professorand the Director of the Power and Industrial Elec-tronics Research Program, Institute for Power andElectronic Engineering, University of Novi Sad, and

later, became the Head of the Institute. He then joined the Bradley Department ofElectrical and Computer Engineering, Virginia Tech as an Associate Professor,where he is currently the American Electric Power Professor at the Departmentand a Co-Director of the Center for Power Electronics Systems, Virginia Tech.His research interests include multiphase power conversion, electronic powerdistribution systems, power electronics systems modeling and control, and in-tegrated design of power converters.

Dr. Boroyevich was the President of the IEEE Power Electronics Societyfor 2011–2012. He received the IEEE William E. Newell Power ElectronicsTechnical Field Award, and is a Member of the US National Academy of Engi-neering.

Rolando Burgos (S’96–M’03) received the B.S. de-gree in electronics engineering, the Electronics Engi-neering Professional degree, and the M.S. and Ph.D.degrees in electrical engineering from the Universityof Concepcion, Concepcion, Chile, in 1995, 1997,1999, and 2002, respectively.

In 2002, he joined as a Postdoctoral Fellow, theCenter for Power Electronics Systems (CPES), Vir-ginia Tech, Blacksburg, VA, USA, and become a Re-search Scientist in 2003 and a Research AssistantProfessor in 2005. In 2009, he joined ABB Corpo-

rate Research, Raleigh, NC, USA, as a Scientist, become a Principal Scientist in2010. In 2010, he joined, as an Adjunct Associate Professor in the Department ofElectrical and Computer Engineering, North Carolina State University, Raleigh,working at the Future Renewable Electric Energy Delivery and ManagementSystems Center. In 2012, he returned to Virginia Tech, where he is currentlyan Associate Professor in the Bradley Department of Electrical and ComputerEngineering and CPES faculty. His research interests include multiphase mul-tilevel power conversion, grid power electronics systems, stability of ac and dcpower systems, high-power density power electronics, modeling, and controltheory and applications.

Dr. Burgos is a Member of the IEEE Power Electronics Society, where hecurrently serves as an Associate Editor of the IEEE Transactions on Power Elec-tronics, the IEEE Power Electronics Letters, and the IEEE Journal of Emergingand Selected Topics in Power Electronics. He is also the Vice-Chair of the Powerand Control Core Technologies Committee of the Power Electronics Society.He is also a member of the IEEE Industry Applications Society and of the IEEEIndustrial Electronics Society.

Christina DiMarino (S’13) received the Bachelor’sdegree in engineering from James Madison Univer-sity, Harrisonburg, VA, USA, in 2012, the Master’sdegree from Center for Power Electronics Systems(CPES) at Virginia Tech, Blacksburg, VA, USA, forher work on the high-temperature characterization ofsilicon carbide transistors. She joined CPES in thefall of 2012, where she is currently working towardthe Ph.D. degree in the area of high-power siliconcarbide modules.

Fang Chen (S’13) received the Bachelor’s and Mas-ter’s degrees in electrical engineering from ZhejiangUniversity, Hangzhou, China, in 2009 and 2012, re-spectively. He is currently working toward the Ph.D.degree in the Center for Power Electronics Systemsat Virginia Polytechnic Institute and State University(Virginia Tech), Blacksburg, VA, USA. From 2015to 2016, he was a Guest Student in the Departmentof Energy Technology, Aalborg University, Aalborg,Denmark.

His research interests include operation and con-trol of dc power distribution systems, net-zero-energy buildings, and design ofac–dc converters.

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Medium-Voltage Impedance Measurement Unit for Assessing