LINEAR TECHNOLOGY LINEAR TECHNOLOGY · 2018. 3. 20. · output-buffer amplifier, a 2.048V voltage reference, and an easy-to-use, cascadable three-wire serial inter-face. The single

OCTOBER 1993 VOLUME III NUMBER 3

IntroductionThe new LT1300 and LT1301

micropower DC-to-DC convertersprovide improvements in both elec-trical and physical efficiency, two keyareas of battery-based power-supplydesign. Housed in 8-lead DIP or SOICpackages, the devices feature a 1Aon-chip switch with a VCESAT of just170mV. The internal oscillator fre-quency is set at 155kHz, allowing theuse of tiny, 5mm diameter surfacemount inductors along with standardD-case size tantalum capacitors. Acomplete 2-cell to 12V, 5V, or 3.3Vconverter can fit in less than 0.4square inches of PC board area.

The devices use Burst ModeTM op-eration to maintain high efficiency atlight load. The quiescent current isonly 120µA. It can be further reducedto 10µA by taking the SHUTDOWNpin high, which also disables the

New LT1300 and LT1301Micropower DC-to-DCConverters

device. The output voltage of theLT1300 can be set at either 5V or 3.3Vvia the logic controlled SELECT pin,and the LT1301 output can be set ateither 5V or 12V using the same pin.The ILIM pin allows the reduction ofpeak switch current, normally 1A, toapproximately 400mA, increasing ef-ficiency and allowing the use of evensmaller components in lighter loadapplications.

OperationFigure 1 is a block diagram of the

LT1300/1301. Refer also to Figure 2for associated component hook-up.When A1’s negative input, relatedto the SENSE pin voltage by the appro-priate resistor -divider ratio,is higher than the 1.25V referencevoltage, A1’s output is low. A2, A3, and

continued on page 12

IN THIS ISSUE . . .

COVER ARTICLENew LT1300 and LT1301Micropower DC-to-DCConverters ....................... 1Steve Pietkiewicz

Editor's Page ................... 2Richard Markell

DESIGN FEATURESThe LTC1257 Provides aComplete, Single-Supply,12-Bit D/A in anSO-8 Package .................. 3Robert Reay

LT1204: High-Speed VideoMultiplexer with CableDriver has −90dBCrosstalk......................... 5John Wright

The World’s Lowest-NoiseDual-JFET Op Amp, theLT1113, Debuts ............... 8Alexander Strong

A New Family of High-Speed, Low-PowerOperational Amplifiers .. 10George Feliz

DESIGN IDEAS .......... 17–28(complete list on page 17)

DESIGN INFORMATIONBook Review:Power Electronics—Circuits, Devices andApplications .................. 29

New Device Cameos ....... 30

LTC in the News ............ 31

Design Tools .................. 32

Sales Offices ................. 32

by Steve Pietkiewicz

S1300_1.eps

–

+

18mV

VIN

+–

R2 700Ω

SW

Q2 3×

PGND

R1 3Ω

Q1 500×

ILIM

8500Ω

Q3

DRIVER

BIAS

OSCILLATOR 5.3µs ON 1.2µs OFF

–

+

A1 SLOW

COMPARATOR

ENABLE

500k

SENSESHUTDOWN

144k

161k

GND SELECT

1.25V REFERENCE

A2 CURRENT

COMPARATOR

A3

Figure 1. LT1300/LT1301 block diagram

Burst ModeTM is a trademark of Linear Technology Corporation.

LINEAR TECHNOLOGY LINEAR TECHNOLOGY LINEAR TECHNOLOGY

2 Linear Technology Magazine • October 1993

find the source of too much outputripple? How about trying to find outwhy the last two bits on an A/D con-verter system toggle? The creativeaspect of the thinking process is thesame in each endeavor.

This issue is packed with articleson new products from LTC. StevePietkiewicz and Dale Eagar providecircuits and architectural insights intothe new LT1300 and LT1301 mi-cropower DC-to-DC converters. Steveprovides the designer’s perspective andDale provides some novel circuits forthe LT1300 series.

George Feliz describes his newseries of high-speed operationalamplifiers, which combine the highslew rates of current-feedback ampli-fiers with the benefits of traditionaloperational amplifiers to create theLT1354 through LT1365 series of am-plifiers. In the related area of videoproducts, John Wright presents his70MHz, four-input video multiplexer,the LT1204. This product provides anincredible 90dB of isolation at 10MHz.

Earwigs in the Drip Watering System?Parsnips for Dinner!

Bob Reay offers a thorough intro-duction to LTC’s first digital-to-analogconverter, the LTC1257. This 12-bitserial D/A converter typically achieves1/4LSB DNL and 2LSBs INL errorwithout trimming. The part comple-ments our line of serial A/Ds quitenicely.

Alexander Strong introduces theJFET-input LT1113 operationalamplifier. The LT1113 is the world’slowest-noise dual-JFET op amp; thepart was designed to amplify high-impedance capacitive transducers. TimSkovmand discusses 2-cell power man-agement techniques and provides acomplete 2-cell to 3.3V, 5V and 12Vpower-management system.

Also, there are many Design Ideas:an LCD bias generator, a linear-phasebandpass filter, a high-efficiency bat-tery charger, a +5V to +3.3V converterfor desktop computers, and a LT1087GTL terminator.

1 See the bottom of page 3 for the answer, “How theygot into the pipes.”

by Richard Markell

I’ve been doing a lot of thinkingabout earwigs lately. Earwigs are thosebugs that look like little lobsters, withthe little pincers on their tails (orderDermaptera). They completely blockeda filter in my drip watering system sothat I had to clean it out every week.Not good. I spent many a sleeplessnight trying to reason out how they gotinto the pipes to be swept down to thepoint where they were trapped by thefilter. Was there a hole underground?Were earwigs spontaneously gener-ated inside the pipes? Were theydelivered as protein-enriched waterfrom my small water company? A sortof “big bug” theory of evolution?1

We all solve engineering problems,both at work and at home. Jim Wil-liams fixes Tektronix 547s; I engineersystems in the garden that water myparsnips. Is it the same thing? I got tothinking about the creative process.I’ve been tracing through the drip sys-tem in my mind, trying to figure outhow the earwigs got into a closedsystem. Is this different from tracingthrough a buck regulator circuit to

FAE Cameo: David DinsmoreLTC now has twenty Field Applica-

tion Engineers worldwide to assist ourcustomers in the design and selectionof circuits available from LTC. All ofour FAEs are available by phone and,in certain situations, in person to helpyou design your circuitry. This spacewill profile one FAE per issue.

David Dinsmore works out of LTC’sSoutheast Sales office. He covers thestates of Texas, Oklahoma, Alabama,Louisiana, Arkansas, Mississippi andthe western half of Tennessee. David’sexpertise is in the areas of data-acqui-sition systems and servo-motor control.

Dave relates, “One of my most inter-esting projects was a 3/4HPmotor-control system for a blood-ana-lyzer centrifuge. It had to be precisely

positioned in either direction for load-ing and unloading the blood samples.The 20-inch-diameter rotor then hadto be accelerated to 5000 RPM at adefined rate in 5.0 seconds. In addi-tion, the rotational speed had to beheld to 720 RPM ±0.1 RPM during thedata-acquisition period to define thesample rate in an A/D conversion sys-tem. This Analog-to-digital controlsystem comprised F/V and D/A con-verters, precision ampli-fiers and filters,high-voltage H-bridge drivers, and amicrocontroller with digital filtering.”

Dave recently talked to an engi-neer who had a very interestingapplication. He was building ahydrophone amplifier to track themovements of schools of fish. He had

gone so far as to have a taxidermistmake a plaster mold of a rainbowtrout. He and his co-workers spenthours painting the prototype troutto look very realistic. (This may betaking precision to the extreme.)He mounted the electronics (which,of course, use LTC op amps for signalconditioning) inside the plaster fish.

David and his wife, Donna, havebeen married for seven years. Theyhave a 16-month-old baby girl, Dana,who, like her father, is already takingthings apart to see how they work.

He enjoys golf, yard work (really?),and playing with his daughter. Youcan reach David through the LTCSoutheast Sales office listed on theback of this magazine.

EDITOR'S PAGE

Linear Technology Magazine • October 1993 3

The LTC1257 Provides a Complete, Single-Supply, 12-Bit D/Ain an SO-8 Package by Robert Reay

IntroductionThe new 8-pin SO LTC1257 re-

quires no external components andprovides one of the smallest, easiest-to-use 12-bit D/A (DAC) systemsavailable. The part includes anoutput-buffer amplifier, a 2.048Vvoltage reference, and an easy-to-use,cascadable three-wire serial inter-face. The single supply voltage canrange from 4.75V to 15.75V and anexternal reference can be used to over-ride the internal reference to ex-tend the output voltage range to0V–12V. The power supply current is alow 350µA max when operating froma 5V supply, and the differential non-linearity (DNL) is less than 1/2LSB(see Figure 1), with no missing codes.

Circuit Topology

Digital SectionThe block diagram and pin dia-

gram of the LTC1257 are shown inFigure 2. The digital section consistsof a 12-bit shift register, control logic,DAC register, and 5V logic regulator.Signal lines include inputs CLK, DIN,and LOAD, and output DOUT. Thedata on the DIN pin is clocked into theshift register and the last bit in theshift register is shifted to DOUT througha buffer on the rising edge of the clock.The MSB is loaded first and the LSBlast. The DAC register loads the datafrom the shift register when LOAD ispulled low, and remains transparentuntil LOAD is pulled high andthe data is latched. The logic regulatorlimits the internal logic’s voltageswing to 5V, reducing clock noise inthe analog section when VCC is at15V. However, the DOUT pin will swingfrom GND to VCC.

DESIGN FEATURES

1257_1.eps

CODE0

–0.5

DN

L ER

RO

R (L

SB's

)

0.0

0.5

512 4098358430722580204815361024

Figure 1. LTC1257 differential nonlinearity (DNL) plot

How they got into the pipesThe earwigs got in through the opening in theanti-siphon valve. I solved the problem by put-ting a fine mesh (my wife’s old stocking) over thevalve.

Figure 2a. LTC1257 block diagram

1257_2a.eps

DAC

–

+

DAC REGISTER

12-BIT SHIFT REGISTER

5V REGULATOR

2.048V REFERENCE

12

12

INTERNAL LOGIC SUPPLY

CLKDIN

GND

LOAD

VCC

DOUT

VOUT

REF

Figure 2b. LTC1257 pin diagram

8

7

6

54

3

2

1CLK

D

LOAD

DOUT GND

VREF

IN VOUT

VCC

1257_2b.eps

Analog SectionThe 12-bit DAC ladder, the 2.048V

reference, and an op amp connectedas a voltage follower make up theanalog section. The DAC has beenoptimized for excellent DNL perfor-mance in a small area. Figure 3 showsthe DAC ladder topology. A string ofequal-valued resistors is connectedbetween the reference pin and ground.

Multiple LTC1257s can be daisy-chained by connecting the DOUT pin ofone chip to the DIN pin of the next, withthe CLK and LOAD signals remainingcommon to all chips. Theserial data is clocked to all of theDACs and the LOAD signal is pulledlow to update all of them simulta-neously. The serial interface can beclocked at any speed up to 1.4MHz.


DESIGN FEATURES

VREF

–

+

1257_3.eps

Figure 4. Auto-ranging, 8-channel ADC with shutdownFigure 3. Equivalent DAC topology

and to the REF pin. The reference is abootstrapped bandgap circuit with atypical temperature coefficient of20ppm/°C. The voltage reference out-put is turned off when the pin is forcedabove 2.5V, allowing an external, high-precision reference to be connected tothe REF pin and DAC resistor ladder.By using the external reference, thefull scale voltage of the DAC can beextended up to 12V. The external ref-erence must be greater than 2.5V, lessthan VCC − 2.7V, and must be capableof driving the 10kΩ minimum DACresistor ladder.

The op amp buffer is connected as avoltage follower and has a common-mode range that extends from groundto within 2.7V of VCC while sourcing2mA. An internal NMOS transistorwith a 200Ω equivalent impedancepulls the output to ground. The out-put is protected against short circuitsand is able to drive a capacitive load ofup to 500pF without oscillation. Off-set is 4mV max over temperature andthe settling time is 6µs maximum.

A switch connects each resistor tap tothe input of the op amp. For any givencode, the corresponding switch isturned on, and the resistor ratio de-termines the fraction of the referencevoltage appearing at the output of thebuffer op amp. This topology guaran-tees no missing codes and the integralnonlinearity is determined by thematching of the resistors within thestring.

Because a 12-bit DAC would re-quire 4096 resistors and a large layout,we have developed a patented schemein which only the MSBs are deter-mined by the resistor string and theLSBs are decoded by a modified inputstage in the op amp. This design dra-matically reduces the number ofresistors and the size of the DAC andmaintains excellent DNL performance.The typical DNL error is1/4LSB while the typical INL error is2LSBs, without any costly trimming.

The internal voltage reference pro-vides a constant 2.048V, making 1LSBequal to 500µV. The reference isconnected to the DAC resistor ladder


ApplicationsThe LTC1257 is intended for appli-

cations where small size, low externalparts count, low supply current, single-supply operation, and excellent DNLperformance are needed. Applicationsinclude portable instrumentation, digi-tally controlled calibration, servocontrols, process-control equipment,and automatic test equipment.

Two LTC1257s and a LTC1296 A/Dcan be used to build an auto-ranging,8-channel ADC system with shut-down, as shown in Figure 4. TwoLTC1257s are cascaded together andprovide the zero and full-scale refer-ence voltages for the LTC1296 A/Dconverter. The DAC outputs are fil-tered to remove the small amount ofdigital noise from the outputs. VCC forthe DACs is supplied via a PNP tran-sistor. The microprocessor writes thetwo 12-bit full-scale and zero words tothe DACs via the DIN and CLK inputs,then pulls LOAD low to update theDAC outputs. With zero and full-scaleset, an A/D conversion can be made.

1257_4.eps

CLK

LOAD

DOUTGND

VREFVCC

VOUT

µP

LTC1257

CLK

LOAD

DOUTGND

VREFVCC

VOUT LTC1257

CS

DOUT

CLK

DIN

VCCCH0

CH7

COM

REF+ SSOREF–

5V22µF

8 ANALOG INPUT CHANNELS

74HC04

50k50k5V

0.1µF

100Ω

0.1µF

100Ω

0.1µF

LTC1296

DIN

DIN

2N3906


LT1204: High-Speed Video Multiplexerwith Cable Driver has −90dB Crosstalk

by John Wright

IntroductionThe LT1204 is the first in a series of

products developed by Linear Tech-nology to solve difficult video switchingand distribution problems in the newmultimedia products. This new 4-input multiplexer includes a 75MHzcurrent-feedback amplifier that willdirectly drive 75 ohm cables. We havedeveloped a novel circuit technique toexpand the number of multiplexers forlarge routing systems without degrad-ing the signal integrity. This techniquesimplifies cable terminations as thenumber of inputs increases. Videospecifications, such as differential gainand phase, gain flatness, and switch-ing transients are at professional videolevels. Unlike many multiplexers,which have 1V–2V input ranges, theLT1204 supports large input and out-put signal levels, which make it idealfor general-purpose analog signal se-

Figure 1. LT1204 block diagram

Table 1. LT1204 truth tableChannel

A1 A0 ENABLE SHUTDOWN Selected0 0 1 1 VIN0

0 1 1 1 VIN1

1 0 1 1 VIN2

1 1 1 1 VIN3

X X 0 1 High-Z OutputX X X 0 OFF

DESIGN FEATURES

1204_1a.eps

VIN 0

–

+

VIN 1

VIN 2

VIN 3

GND

REF

GND

1

2

3

4

5

6

7

8

75Ω

75Ω

75Ω

75Ω

6.8k 6.2k–15V

VIN 0

VIN 1

VIN 2

VIN 3 +1

+1

+1

+1

GND

V+ 16

15

14

13

12

11

10

9

V0

V–

FB

SHUTDOWN

ENABLE

A1

A0

LOGIC

CFA

RF 1k–15V

75Ω

RG 1k

VOUT

+15V

lection and multiplexing. The LT1204is available in 16-lead PDIP and 16-lead SOL packages.

Figure 1 shows a block diagram ofthe LT1204. Its truth table appears asTable 1. The input buffers are actuallylow-insertion-loss tee switches de-signed to give the excellent crosstalkperformance required for professionalvideo. The switches are internally con-nected to the noninverting input of theCFA to reduce capacitance and im-prove the AC characteristics of thesignal path; the inverting input is ex-ternal for easy gain adjustment. Thelogic interface decodes channel select,shutdown, and enable/disable, thelast of which puts the CFA into a truehigh output impedance state. A refer-ence (Pin 8) is available for optimizingthe internal logic circuitry for differentinput signal ranges.

The Challenge ofVideo Multiplexers

Before HDTV, CD Interactive, andthe proliferation of video products,source selection was made during theblanking period, and the effects ofswitching transients were not visible.As multimedia, new special effects,and picture processing become popu-lar, it has become necessary to switchvideo “in picture”; in such cases thenature of the switching transient iscritical. Switching techniques thatworked in the past now cause prob-lems. Older bipolar ICs that switchlateral PNP transistors in the signalpath take several microseconds tosettle, blurring the transition betweenpictures. CMOS multiplexers, whichare bidirectional, suffer from poor out-put-to-input isolation and causetransients to feed to the inputs. CMOSMUXs have been built with break-before-make switches to eliminate thetalking between channels, but thesesuffer from output glitches largeenough to interfere with the sync cir-cuitry. By contrast, the LT1204 isfabricated on LTC’s complementary-bipolar process to attain good switchingcharacteristics, buffering, crosstalk,and speed. Let’s look at these areasone-by-one to see how they stack up.


DESIGN FEATURES

Switching CharacteristicsSwitching between channels is a

make-before-break condition whereboth inputs are on momentarily. Theinput with the largest positive voltagedetermines the output level. If bothinputs are equal, there is only 40mVof error at the input of the CFA duringthe transition. The reference adjust(Pin 8) allows the user to trade positiveinput voltage range for switching time.For example, on ±15V supplies, set-ting the voltage on pin 8 to −6.8Vreduces the switching transient dura-tion to 50ns and the positive inputrange from +6V to +2.35V; the nega-tive input range is independent of thesetting on pin 8 and remains at −6V.When switching composite video “inpicture,” this short (50ns) transient isimperceptible, even on high qualitymonitors. The reference pin has noeffect when the LT1204 is operatingon ±5V, and should be grounded inthis situation. Figure 2 is a scopephotograph of the output switchingtransient with a 2MHz sine wave con-nected to VIN0 and VIN1.

Input BuffersThe buffers isolate the inputs when

the make-before-break switchingoccurs. The design of these input buff-ers included special attention to theirDC matching and dynamic character-istics. The DC input-offset matchbetween channels is more importantto the video engineer than is theactual value of the input offset. A DCmismatch as small as 3mV betweenchannels is just visible on a qualityvideo monitor. The typical VOS mis-match between channels on theLT1204 is about 300µV.

CrosstalkThe crosstalk (more accurately, all

hostile crosstalk) is measured by driv-ing signal into any three of the fourinputs and selecting the fourth inputwith logic control. This fourth input iseither shorted to ground or termi-nated in an impedance. All hostilecrosstalk is defined as the ratio in dBof the signal, at the output of the CFA,to the signal on the three driven in-

Figure 4. All hostile crosstalk of LT1204 in PDIP and SOIC

1204_4.epsFREQUENCY (MHz)

1–120

ALL

HO

STIL

E CR

OSS

TALK

(dB)

–100

–20

10

–40

–60

–80

100

PDIP

SOIC

VS = ±15V RL = 100Ω RS = 10Ω

Figure 3. Tee switch

1204_3.eps

VIN

VOUT

I1

I2

–

+

FB

V–

Q2

Q1

V+

Q3

RG

RF

CFATO LOGIC

1204_2.eps

VOUT PIN 15

LOGIC PIN 9

Figure 2. VIN0 and VIN1 connected to a 2MHzsinewave, pin 8 voltage = –6.8V

puts. Disable crosstalk is measuredwith all four inputs driven and thepart disabled. Crosstalk is critical inmany applications where video multi-plexers are used. In professional videosystems, a crosstalk figure of −72dB isa desirable specification.

The key to the outstandingcrosstalk performance of the LT1204is the tee switch shown in Figure 3.When the tee switch is on (Q2 off) Q1and Q3 are a pair of emitter-followerswith excellent AC response for drivingthe CFA. When the decoder turns offthe tee switch (Q2 on), the emitter-base junctions of Q1 and Q3 becomereverse biased while the Q2 emitterabsorbs current from I1. Not only dothe reverse-biased emitter -base

junctions provide good isolation, butany signal at VIN0 coupling to the Q1emitter is further attenuated by theshunt impedance of the Q2 emitter.Current source I2 routes current toany ON switch.

Crosstalk performance is stronglyaffected by the IC package and the PCboard layout, as well as by the circuitdesign. The die layout uses groundsbetween the inputs to isolate adjacentchannels, and the output and feed-back pins are on opposite sides of thedie from the inputs. Laying out a PCboard that provides 90dB of isolationfrom all crosstalk at 10MHz is not atrivial task. That crosstalk level corre-sponds to a 30µV output below a 1Vinput at 10MHz. We have fabricated a


DESIGN FEATURES

demonstration board to show the com-ponent and ground placementrequired to attain these crosstalk num-bers. Figure 4 is a graph of all hostilecrosstalk for both the PDIP and SOpackages. It has been found empiri-cally from these PC boards thatcapacitive coupling across the pack-age of greater than 3fF (yes, that is0.003 picofarads) will diminish therejection; we recommend that you usethis proven layout in your designs.

Multiplexer ExpansionLT1204s can be paralleled by short-

ing their outputs together to expandthe number of MUX inputs. This newmultiplexer uses a novel circuit (patentpending) to ensure that the unselectedamplifiers do not load or alter thecable termination and that there is noshoot-through current when the out-puts of two or more amplifiers areshorted together. (Shoot-through cur-rent is a spike of power-supply currentcaused by both amplifiers being on atthe same time.) When the LT1204 isdisabled (pin 11 low), the output stageis turned off and an active buffersenses the output voltage and drivesthe feedback pin of the CFA (Figure 5).This bootstraps the feedback resis-tors and raises the true outputimpedance of the circuit. For the con-dition where RF = RG = 1kΩ, the outputimpedance is typically raised to 25kΩduring disable. If the part is shutdown, however, the bootstrapping isinoperative, and the feedback resis-tors will load the output (ROUT = 2kΩ).If the CFA is operated at a gain of +1,the feedback resistor will not load theoutput even in shutdown mode, be-cause there is no resistive path toground; however, there will be a 6dBloss through the cable.

Figure 6 is a frequency-responseplot showing the effect of using thedisable feature versus using the shut-down. In this example, four LT1204swere connected together at their out-puts to form a 16-to-1 MUX. The plotshows the effect of the bootstrappingcircuit, which eliminates the impropercable termination due to the feedbackresistors loading the cable.

Figure 5. Active buffer drives FB node during disable

1204_5.eps

–

+

VIN3

VIN0

FB

CFA “OFF”

V+

V–

75Ω

VOUT

RF

RG

CABLE

75Ω

75ΩLT1204 “ON”

TEE SWITCH

TEE SWITCH

TEE SWITCH

TEE SWITCH

V–

V+

V–

Continued on page 16

Figure 6. 16-to-1 multiplexer response using disable feature versus shutdown feature

1204_6.epsFREQUENCY (MHz)

1

GAI

N (d

B)

–6

+2

10

0

–2

–4

100

DISABLE

SHUTDOWN

VS = ±15V RL = 150Ω


IntroductionThe LT1113 joins the LTC family of

low-noise op amps as the lowest-noisedual-JFET op amp available. This opamp combines the low current noise(less than 10fA/(Hz)1/2) of a FET opamp with a maximum voltage noise of6.0nV/(Hz)1/2. In addition, the LT1113is stable for a gain of +1 and willhandle 1000pF load capacitances. Allof this is available in an 8-pin small-outline surface mount package withthe standard pinout. Most op ampsrequire a relaxation of specificationsin the surface mount package due toassembly shifts. For the LT1113CS8,spec relaxation is not necessary. TheLT1113CS8 specs are identical to theLT1113CN8. Table 1 highlights someof the guaranteed specifications forthe low-cost grade.

The World’s Lowest-Noise Dual-JFET OpAmp, the LT1113, Debuts by Alexander Strong

DESIGN FEATURES

Voltage Noise @ 1kHz 6.0nV/√Hz MaxVOS 1.8mV MaxAVOL (RL = 10kΩ) 1000V/mV MinGBWP @ 100kHz 4.5MHz MinSlew Rate 2.5V/µS MinISUPPLY per amp 6.5mA Max

Table 1. LTC1113CS8 guaranteed specifica-tions: 100% tested, VS = ±15V, TA = 25°C

Hydrophones RequireHigh Input Impedance

The combination of low voltage noiseand low current noise makes theLT1113 suitable in applications wherelow level signals need to be amplifiedfrom high impedance capacitive trans-ducers. Photo diodes, hydrophones,and accelerometer transducers ex-hibit high impedances, which makethe op amp current noise dominatethe total output noise:

Vn =

AV √Vn2(OP AMP) + 4kT • 2RTRANS + 2q IB • 2(RTRANS)2

Current noise is derived from theDC value of the FET input bias current,or

In(fA/(Hz)1/2 ) = (2qIB)1/2.

where q = 1.6E–19For a 300pA IB, a 9.8fA/(Hz)1/2

current noise is multiplied by thetransducer impedance, which adds tothe total output voltage noise of theamplifier. The best low-noise bipolarop amps cannot match the per-formance of the LT1113 where high-impedance transducers are used.Figure 1 shows a comparison betweenthe LT1113 and the LT1124 bipolarop amp. The LT1124 has 40% lessvoltage noise at 1kHz than the LT1113and is a better choice for transducerimpedances less than 1kΩ, but fortransducer impedances over 100kΩ,the LT1113 is clearly the champ. Thedashed lines on Figure 1 show thetotal noise with a parallel 1000pFcapacitance (since most hydrophoneand accelerometer transducers arecapacitive by nature). The graph showsthat when the LT1113 is used, thetotal noise is dominated by the trans-ducer impedance and not the amplifier,as in the case of the LT1124.

Figure 2. Noninverting and inverting gain configurations

–

+OUTPUT

CF

CB RB

CB = CFCS RB = RFRS

RF

CS RS

TRANSDUCER

1113_2b.eps1113_2a.eps

–

+

R2

OUTPUT

RB

CB

R1

CS RS

CB ≅ CS RB = RS RS > R1 OR R2

TRANSDUCER

The two basic gain configurationsare shown in Figure 2. The noninvert-ing gain configuration is used forvoltage-mode transducers such as hy-drophones, whereas the inverting gainconfiguration is used for charge trans-ducers such as accelerometers. Ineach example, a source-balancing im-pedance is added to improve the overallperformance of the amplifier. RB isselected to cancel the voltage offsetdue to the input bias current flowing

Figure 1. Comparison of LT1113 and LT1124total output 1kHz voltage noise versus sourceresistance

SOURCE RESISTANCE (RS IN OHMS)1001

10

1k

1K 100M

1113_1.eps

100K

100

10M10K 1M

BOTH SOURCE RESISTORS ONLY

TOTA

L 1k

Hz

VOLT

AGE

NO

ISE

DEN

SITY

(nV/

√Hz)

Vn = √Vn2

(OP AMP) + 4kT • 2RSOURCE + 2q IB • 2(RSOURCE)2

+

–

CSRS

VO

CS

RS

LT1124*LT1113*

LT1124*

LT1113*

* PLUS BOTH SOURCE RESISTORS † SOURCE RESISTORS 1000pF SOURCE CAPACITANCE

LT1113†

LT1124†


DESIGN FEATURES

into the source impedance. This isespecially important as the operatingtemperature increases, since FETinput bias currents are usuallyuncancelled, and double for every 10degrees C. A parallel capacitor CBcancels the pole that is caused by theamplifier input capacitance and RB.

ApplicationsFigure 3 shows a low-noise hydro-

phone amplifier with a DC servo. Hereone half of the LT1113 is configured inthe noninverting mode to amplify avoltage signal from the hydrophone,and the other half of the LT1113 nullserrors due to voltage and current off-sets of the amplifier and to impedancemismatches. The value of C1 depends

on the capacitance of the hydrophone,which can range from 200pF to8000pF. The time constant of the servoshould be larger than the time con-stant of the hydrophone capacitanceand the 100MΩ source resistance.This will prevent the servo from can-celing the low-frequency signals fromthe hydrophone.

Another popular charge-outputtransducer is the accelerometer. Sinceprecision accelerometers are chargeoutput devices, the inverting mode isused to convert the transducer chargeto an output voltage. Figure 4 is anexample of an accelerometer with a DCservo. The charge from the transduceris converted to a voltage by C1, whichshould equal the transducer capaci-

tance plus the input capacitance of theop amp. The noise gain will be 1 + C1/CT. The low frequency bandwidth willdepend on the value of R1 (orR1 × (1+R2/R3) for a Tee network). Aswith the hydrophone example, the timeconstant of the servo should be largerthan the time constant of the amplifier.

ConclusionThe LT1113 is not just another op

amp. As the lowest-noise dual-JFETop amp in the market place, theLT1113 should be the first choicewhere high-impedance transducersare used. The 8-pin surface mountpackage will allow the LT1113 tosatisfy the most demanding boardspace requirements.

Figure 3. Low-noise hydrophone amplifier with DC servo

1113_3.eps

+

–100MΩ

C1

A 1/2 LT1113200Ω

3.9k

1MΩ

CT HYDROPHONE

100MΩ

1µF

OUTPUT

100k1MΩ

+

–B

1/2 LT11131MΩ

C1 = CT = 200pF TO 8000pF DC OUTPUT ≤ 4mV AT TA < 70°C OUTPUT VOLTAGE NOISE = 130nV/√Hz AT 1kHz POWER SUPPLY RANGE = ±5V TO ±15V

V+

V–

8

1

43

2

6

5

7

Figure 4. Accelerometer circuit with DC servo

1113_4.eps

ACCELEROMETER CT

R2 18k

C4 2µF

+

–

3

2

8

4

1

+

–1/2 LT1113

1/2 LT1113

C5 2µF

5

6

R5 20M

R4 20M

7

R3 2k

R1 100MΩ

C1 1250pF

OUTPUT = 0.8µV/pC*= 8.0mV/G**

DC OUTPUT ≤ 2.7mV OUTPUT NOISE = 6µV/√Hz AT 1kHz

*PICOCOULOMBS **EARTH’S GRAVITATIONAL CONSTANT


IntroductionA new family of high-speed opera-

tional amplifiers from LTC utilizes anovel circuit topology that blendsthe high slew rate of current-feed-back amplifiers with the benefits oftrue voltage-feedback amplifiers. Com-pared to other devices with similarbandwidths, these amplifiers offerlower supply current, higher slewrate, better DC specifications, fastersettling times, and lower input noise.The family is built on LTC’s comple-mentary bipolar process andencompasses bandwidths from 12MHzto 70MHz. The fastest parts are theLT1363, LT1364, and LT1365, whichhave 70MHz gain bandwidth and1000V/µs slew rate, and consumeonly 6mA of supply current per am-plifier. The lowest power devices arethe LT1354, LT1355, and LT1356,which draw only 1mA of supply cur-rent and provide 12MHz of bandwidthand 400V/µs slew rate. In betweenare the LT1357, LT1358, and LT1359,which have 25MHz gain bandwidthand 600V/µs slew rate and consume2mA, and the LT1360, LT1361, andLT1362, which have 50MHz band-width and 800V/µs slew rate, andconsume 4mA.

Table 1 summarizes the impor-tant specifications of each device. All

A New Family of High-Speed,Low-Power Operational Amplifiers

Figure 1. Traditional high-speed amplifier—simplified schematic

1363_1.eps

+IN–IN

V–

V+

OUTPUT

OUTPUT STAGE

CT

Q5

Q7Q6

BIASQ3Q2Q1

IO IO

IO

Q4R R

+1

devices operate over a wide supplyvoltage range of ±2.5V to ±18V andare available in dual, single, andquad versions. The single and dualversions are available in 8-pin DIPsand surface mount packages. Thequad versions are available in 14-pinDIPs and narrow, S16 surface mountpackages.

Parameter Conditions LT1354/5/6 LT1357/8/9 LT1360/1/2 LT1363/4/5 UnitsVOS (max) VS = ±5V, ±15V 1 0.6 1 1.5 mVIB (max) VS = ±5V, ±15V 0.3 0.5 1 2 µAIOS (max) VS = ±5V, ±15V 70 120 250 350 nAAVOL (min) VS = ±5V, 500Ω 10 10 3 3 V/mV

VS = ±15V, 1kΩ 20 20 4.5 4.5 V/mVGBW VS = ±15V 12 25 50 70 MHzSR VS = ±15V 400 600 800 1000 V/µstSETTLE 10V step, 0.1% 240 170 60 50 nsCMRR (min) VS = ±15V 86 86 86 86 dB

VS = ±5V 80 80 79 78 dBPSRR (min) VS = ±2.5V to ±15V 92 92 97 90 dBNoise voltage 10kHz 10 8 10 10 nV/√HzNoise current 10kHz 0.6 0.8 1 1.2 pA/√HzOutput swing VS = ±15V, 500Ω 12.5 12.5 13 13 V

VS = ±5V, 150Ω 3 3 3.2 3.4 VISUPPLY VS = ±15V 1 2 4 6 mA

Table 1. Important specifications; LT1354, LT1357, LT1360, and LT1363 op ampfamilies

BackgroundThe new topology changes the rela-

tionship between slew rate and supplycurrent found in traditional voltage-feedback amplifiers. In order tounderstand the circuit, let’s review atraditional high-speed, voltage-feed-back design, as shown in Figure 1.The input pair Q1 and Q2 is resis-tively degenerated and feeds PNPcascodes Q3 and Q4. Differential-to-single-ended conversion is performedby the mirror Q5-Q7. The collectors ofQ4 and Q5 form the high-impedancegain node with capacitance CT, estab-lishing the dominant pole. Aunity-gain buffer follows the gainnode. The slew rate is determined bythe current available to charge CT,which, in this case, is IO. Since theslew rate is IO/CT, it is directly propor-tional to supply current. In order toobtain high slew rates, the circuit’squiescent currents must be large. Forinstance, a 1mA current is required toobtain 250V/µs with 4pF of capaci-tance. The bandwidth of the amplifier

DESIGN FEATURES

by George Feliz


Figure 3. Large signal step response: LT1354,LT1357, LT1360, LT1363

DESIGN FEATURES

is determined by R and CT and is givenby 1/(2πRCT). The useable band-width is typically limited by otherpoles in the frequency response, so Ris selected to obtain adequate stabil-ity. For R = 1kΩ and CT = 4pF, thebandwidth is 40MHz. Once the slewrate and bandwidth are chosen, theDC performance is set by the degen-eration of the input pair. With IO =1mA and R = 1kΩ, a resistor mis-match of only 0.2% gives 1mV ofinput offset voltage. The open-loopgain is also reduced by the inputdegeneration, as it is inversely pro-portional to R. For high-slew-rate de-signs, the input bias current ishigh because it is also directly pro-portional to IO. In our example, if thetransistor beta is 100, the input biascurrent would be 5µA and the offsetcurrent would be about 500nA. Inputnoise is also degraded due to theresistor noise and reduced gain.

The New TopologyFigure 2 is a simplified schematic

of the new circuit. The circuit lookssimilar to a current-feedback ampli-fier, but both inputs are highimpedance as in a traditional volt-age-feedback amplifier. A comple-mentary cascade of emitter followersQ1–Q4 buffers the noninverting in-put and drives one side of a resistor.The other side of the resistor is driven

by Q5–Q8, which form a buffer forthe inverting input. The input voltageappears across the resistor, generat-ing a current that is mirrored byQ9–Q14 into the high impedancenode. Q15–Q18 form an output stage.In a current-feedback amplifier, therewould be an external resistor con-nected between the output and theemitters of Q3 and Q4, which wouldbe the inverting input.

The bandwidth, as before, is1/(2πRCT), but the current availableto slew CT is the differential inputvoltage divided by 2R, so the slew rateis independent of the operating cur-rents of the input stage (which canthen be reduced). For a 4V inputstep, a 2kΩ resistor, and a 4pF ca-pacitor, the slew rate is ideally 500V/µs. The high slew rate and balancedinput stage significantly reduce set-tling time. As with current-feedbackam-plifiers, the RC time constant setsboth the small and large signal re-sponses. Figure 3 shows the largesignal response of the family ofamplifiers.

The new topology provides signifi-cantly better DC performance. Theoffset does not rely on tight resistormatching as it does in the conven-tional circuit, but rather on transistormatching and errors in the currentmirrors. A mismatch of 4% betweenthe input devices gives 1mV of input

Figure 2. New high-speed amplifier—simplified schematic

1363_2.eps

–IN

V–

V+

CT

Q5Q6

Q8

Q7 Q3

2RQ2

Q1+IN

Q13 Q14

Q4

Q12

Q9

Q11Q10

Q15

Q16

Q17

Q18

OUTPUT

RCCC

offset voltage; this degree of match-ing is more readily achieved than the0.2% resistor matching required forthe conventional circuit. A 1µA mir-ror error will show up as 2mV of VOSfor R = 2kΩ. It is practical to reducethe value of R in this circuit, increas-ing the open-loop gain, decreasingoffset contributions from the currentmirrors, and reducing noise. Inputbias currents are reduced in two ways:the input devices can run at lowercurrents because the slew rate isindependent of their operating cur-rent, and the NPN and PNP basecurrents tend to cancel. For example,if the input devices run at 200µA, theinput bias current will be less than1µA, even if the betas of the NPN andPNP mismatch by a factor of two. Thebalanced inputs of the new topologyalso provide excellent rejection ofpower-supply and common-modevariations.

The circuit has one other notewor-thy feature—it is able to drivecapacitive loads and remain stable.The RC–CC network across the outputstage is bootstrapped when the am-plifier is driving a light or moderateload and has no effect under normaloperation. When driving a capacitiveload, the network is incompletelybootstrapped and adds to the com-pensation network. The addedcapacitance provided by CC slowsdown the amplifier and the zero cre-ated by RC adds phase margin,


LT1354

LT1357

LT1360

LT1363


the oscillator are turned off, drawingno current. Only the reference and A1consume current, typically 120µA.When the voltage at the SENSE pindecreases enough to overcome A1’s6mV hysteresis, A1’s output goeshigh, enabling the oscillator, A2, andA3. Quiescent current increases to2mA as the device prepares for high-current switching. Q1 then turns onin a controlled saturation for(nominally) 5.3µs or until current com-parator A2 trips, whichever comesfirst. After a fixed off-time of (nomi-nally) 1.2µs, Q1 turns on again. Q1’sswitching causes current to alternatelybuild up in L1 and dump into outputcapacitor C1 via D1, increasing theoutput voltage. When the output ishigh enough to cause A1’s output to golow, switching action ceases. C1 is leftto supply current to the load until VOUTdecreases enough to force A1’s output

Figure 3, trace B. The reduced peakswitch current reduces I2R losses inQ1, L1, C1, and D1. You can increaseefficiency by doing this, provided theoutput-current reduction is accept-able. Lower peak currents also extendalkaline battery life, due to the alka-line cells’ high internal impedance.

Five Volts from Two CellsFigure 2’s circuit provides 5V from

a 2-cell input. Shutdown is effectedby taking the SHUTDOWN pin high.VIN current drops to 10µA in thiscondition. This simple boost topologydoes not provide output isolation,and in shutdown the load is stillconnected to the battery via L1 andD1. Figure 4 shows the efficiency ofthe circuit with a range of inputvoltages, including a fresh battery(3V) and an “almost-dead” battery(2V). At load currents below a fewmilliamperes, the 120µA quiescent cur-

LT1300, continued from page 1 high, and the entire cycle repeats. Ifswitch current reaches 1A, causingA2 to trip, switch on-time is reducedand off-time increases slightly. Thisallows continuous mode operationduring bursts. A2 monitors thevoltage across 3Ω resistor R1. Q2’scollector current is set by the emitter-area ratio to 0.6% of Q1’s collectorcurrent. When R1’s voltage drop ex-ceeds 18mV, A2’s output goes high,truncating the on-time portion of theoscillator cycle and increasing off-timeto about 2µs, as shown in Figure 3,trace A. Eighteen millivolts across R1corresponds to a switch current of 1A.This peak current can be reduced bytying the ILIM pin to ground, causing15µA to flow through R2 into Q3’scollector. Q3’s current causes a10.4mV drop in R2, so that only7.6mV is required across R1 to turnoff the switch. This corresponds to a400mA switch current, as shown in

DESIGN FEATURES

S1300_2.eps

VINSW

LT1300

GND PGNDILIM SENSE

SHDN

100µF

2× AA CELL

SHUTDOWN

C1 100µF

L1* 10µH

D1 1N5817

+

+

SELECT

*SUMIDA CDS 4-100LC (708) 956-0666 COILCRAFT 3316-223 (800) 322-2645

5V OUTPUT 200mA

NC

5µs/DIVS1300_3.eps

TRACE B 1A/DIV

TRACE A 1A/DIV

Figure 3. Switch pin current with ILIM floating orgrounded

LOAD CURRENT (mA)1

74

EFFI

CIEN

CY (%

)

90

10 500

S1300_4.eps

100

88

86

84

82

80

78

76

VIN = 4.0V

VIN = 3.0V

VIN = 2.5V

VIN = 2.0V

20µs/DIVS1300_5.eps

C = 1A/DIV

A = 20mV/DIV AC COUPLED

B = 5V/DIV

Figure 2. 2-cell to 5 volt DC/DC converter delivers >200mA with a 2.0 volt input

Figure 4. Efficiency of Figure 2’s circuit

Figure 5. Burst ModeTM operation in action


BATTERY

TIME (HOURS)0

0

OU

TPU

T/BA

TTER

Y VO

LTAG

E (V

)

5.0

11

S1300_6.eps

9651

1.0

2.0

4.5

4.0

3.5

3.0

2.5

1.5

0.5

2 3 4 7 8 10

2× L91 LITHIUM

2× E91 ALKALINE

OUTPUT

DESIGN FEATURES

BATTERY

TIME (HOURS)0

0

OU

TPU

T/BA

TTER

Y VO

LTAG

E (V

)

5.0

5.5

S1300_7.eps

4.53.02.50.5

1.0

2.0

4.5

4.0

3.5

3.0

2.5

1.5

0.5

1.0 1.5 2.0 3.5 4.0 5.0

2× E91 ALKALINE

OUTPUT

2× L91 LITHIUM

VIN = 2.5V

VIN = 3V

VIN = 2V

LOAD CURRENT (mA)1

74

EFFI

CIEN

CY (%

)

82

90

10 100

S1300_9.eps

88

86

84

80

78

76

TIME (HOURS)0

0

OU

TPU

T/BA

TTER

Y VO

LTAG

E (V

)

5.0

22

S1300_10.eps

1812102

1.0

2.0

4.5

4.0

3.5

3.0

2.5

1.5

0.5

4 6 8 14 16 20

2× L91 LITHIUM

2× E91 ALKALINE

OUTPUT

BATTERY

24

rent of the device becomes significant,causing the fall-off in efficiency de-tailed in the figure. At load currents inthe 20mA to 200mA range, efficiencyflattens out in the 80% to 88% range,depending on the input. Figure 5 de-tails circuit operation. VOUT is shown intrace A. The burst-repetition pattern isclearly shown as VOUT decays, thensteps back up due to switching action.Trace B shows the voltage at the switchnode. The damped, high-frequencywaveform at the end of each burst isdue to the inductor “ringing off,” form-ing an LC tank with the switch anddiode capacitance. It is not harmfuland contains far less energy than thehigh-speed edge which occurs whenthe switch turns off. Switch current isshown in trace C. The current com-parator inside the LT1300 controlspeak switch current, turning off the

S1300_8.eps

VINSW

LT1300

GND PGNDILIM SENSE

SHDN

47µF

2× AA CELL

SHUTDOWN

33µF

L1* 22µH

D1 MBRS140T3

+

+

SELECT

*COILCRAFT 1608-223 (800) 322-2645

5V OUTPUT 50mA

Figure 8. Lower-power applications can use smaller components. L1 is tallestcomponent at 3.1mm

Figure 6. Two Eveready L91 lithium AA cellsprovide approximately twice the life of E91alkaline cells at a 100mA load current

Figure 7. Doubling load current to 200mAcauses E91 Alkaline battery life to drop by2/3; L91 lithium battery shows 2.5:1difference in operating life

Figure 9. Efficiency of Figure 8’s circuit Figure 10. 50mA load and reducedswitch current are kind to E91 AAalkaline battery; the advantages ofL91 lithium are not as evident

switch when the current reaches ap-proximately 1A.

Although efficiency curves presentuseful information, a more importantmeasure of battery-powered DC/DCconverter performance is operating life.Figures 6 and 7 detail battery life testswith Figure 2’s circuit at load currentsof 100mA and 200mA, respectively.Operating-life curves are shown usingboth Eveready E91 alkaline cells andnew L91 “Hi-Energy” lithium cells.These lithium cells, new to the market,are specifically designed for high-drainapplications. The performance advan-tage of lithium is about 2:1 at 100mAload current (Figure 6), increasing to2.5:1 at 200mA load (Figure 7). Alka-line cells perform poorly at high drainrates because their internal imped-ance ranges from 200mΩ to 500mΩ,causing a large voltage drop within the

cell. The alkaline cells feel quite warmat 200mA load current, the result ofI2R losses inside the cells.

The reduced-power circuit shownin Figure 8 can generate five volts atcurrents up to 50mA. Here the ILIM pinis grounded, reducing peak switchcurrent to 400mA. Lower profile com-ponents can be used in this circuit.The capacitors are C-case size solidtantalum and inductor L1 is the tallestcomponent at 3.2mm. The reducedpeak current also extends battery life,since the I2R loss due to internal bat-tery impedance is reduced. Figure 9details efficiency versus load currentfor several input voltages and Figure10 shows battery life at a 50mA load.Note that the L91 lithium battery lastsonly about 40% longer than the alka-line. The higher cost of the lithiumcells makes the alkaline cells more


DESIGN FEATURES

cost-effective in this application. A pairof Eveready AAA alkaline cells (typeE92) lasts 96.6 hours with 5mA load,very close to the rated capacity of thebattery.

A 4-Cell ApplicationA 4-cell pack is a convenient,

popular battery size. Alkaline cellsare sold in 4-packs at retail storesand four cells usually providesufficient energy to keep battery re-placement frequency reasonable.Generating 5V from four cells, how-ever, is a bit tricky. A fresh 4-cellpack has a terminal voltage of 6.4V,but at the end of its life, the pack’sterminal voltage is around 3.2V;hence, the DC/DC converter muststep the voltage either up or down,depending on the state of the batter-

ies. A flyback topology with a costly,custom-designed transformer couldbe employed, but Figure 11’s circuitgets around these problems by usinga flying-capacitor scheme along witha second inductor. The circuit alsoisolates the input from the output,allowing the output to go to zero voltsduring shutdown. The circuit can bedivided conceptually into boost andbuck sections. L1 and the LT1300switch comprise the boost or step-upsection, and L2, D1, and C3 comprisethe buck or step-down section. C2 ischarged to VIN and acts as a level shiftbetween the two sections. The switchnode toggles between ground and VIN+ VOUT, and the L2–C2–diode nodetoggles between −VIN and VOUT + VD.Figure 12 shows efficiency versusload current for the circuit. All four

energy-storage elements must handlepower, which accounts for the lowerefficiency of this circuit compared tothe simpler boost circuit in Figure 2.Efficiency is directly related to theESR and DCR of the capacitors andinductors used. Better capacitors costmore money. Better inductors do notnecessarily cost more, but they dotake up more space. Worst-case RMScurrent through C2 occurs at mini-mum input voltage and measures0.4A at full load with a 3V input. C2’sspecified maximum RMS currentmust be greater than this worst-casecurrent. The Sanyo capacitors notedspecify a maximum ESR of 45mΩwith a maximum ripple current rat-ing of 2.1A. The Gowanda inductorsspecify a maximum DCR of 58mΩ.

S1300_11.eps

VINSW

LT1300

GND PGND

ILIM

SENSESHDN

C1** 100µF

4× AA CELLS

SHUTDOWN

L2* 27µH

C3** 100µF

5V OR 3.3V 220mA

L1* 27µH

C2** 100µF

1N5817

5V/3.3V

*L1, L2 = **C1, C2, C3 =

+

+

+

NC

SELECT

GOWANDA GA20-272K (716) 532-2234 SANYO OS-CON 16SA100M (619) 661-6835

Figure 13. LT1301 Delivers 12V from 3.3V or 5V input

S1300_13.eps

VINSW

LT1301

GND PGNDILIM SENSE

SHDN

100µF

SHUTDOWN

47µF

L1* 22µH

1N5817

+

+

SELECT

SUMIDA CD75-220K (708) 956-0666

*L1 =

12V OUTPUT

3.3V OR 5V INPUT

LOAD CURRENT (mA)1

64

EFFI

CIEN

CY (%

)

84

10

S1300_12.eps

100

78

76

74

72

70

68

66

80

82

VIN = 3V

VIN = 4V

VIN = 5V

VIN = 6V

Figure 12. Efficiency of up-downconverter in Figure 11

Figure 11. 4-Cell to 3.3V or 5V converter output goes to zero when in shutdown.Inductors may have, but do not require coupling; a transformer or two separateunits can be used


DESIGN FEATURES

LT1301 OutputsFive or Twelve Volts

The LT1301 is identical to theLT1300 in every way except outputvoltage. The LT1301 can be set to a5V or 12V output via its SELECT pin.Figure 13 shows a simple 3.3V or 5Vto 12V step-up converter. It can gen-erate 120mA at 12V from either 3.3Vor 5V inputs, enabling the circuit toprovide VPP on a PCMCIA cardsocket. Figure 14 shows the circuit’sefficiency. Switch voltage drop is asmaller percentage of input voltageat 5V than at 3.3V, resulting in thehigher efficiency at 5V input.

ConclusionThe new LT1300 and LT1301 are

full-function, micropower step-upDC/DC converters optimized forbattery-powered operation. The con-verters have been optimized for acomplete surface mount solution withhigh efficiency, low quiescent cur-rent, and a low parts count.

LTC1257, continued from page 4

When the LTC1296 receives a shut-down command, the SSO pin of theADC goes high, the PNP turns off, andthe system shuts off.

Figure 5 shows how an LT1021-10can be used to override the LTC1257’sinternal reference. The supply voltagefor both the reference and the DAC isset to 15V, and the full-scale voltagefor the DAC becomes 10V.

The circuit in Figure 6 is a 12-bit,single 5V-supply temperature-controlsystem with shutdown. An externaltemperature is monitored by a J-type

thermocouple. The LT1025A providesthe cold junction compensation for thethermocouple and the LTC1050 chop-per op amp provides signal gain. The47kΩ, 1µF capacitor filters the chop-ping noise before the signal is sent tothe A/D converter. The LTC1297 A/Dconverter uses the reference of theLTC1257 after it has been filtered toset full scale. After the A/D measure-ment is taken, CS is pulled high andeverything except the LTC1257 is pow-ered down, reducing the system supplycurrent to about 350µA. A word can

then be written to the LTC1257 and itsoutput can be used to as a tempera-ture-control signal for the system beingmonitored.

ConclusionThe LTC1257 is one of the smallest,

easiest-to-use 12-bit DACs availabletoday. With its cascadable 3-wire serialinterface, built-in reference and voltagebuffer, single supply operation, low sup-ply current, and small package size, thechip is a natural choice to reduce sys-tem complexity and cost.

1257_6.eps

µP

CLK

LOAD

DOUTGND

VCCVREF

VOUT LTC1257 DAC

CS

DOUT

CLK

VCC–IN

+IN

GND

5V

LTC1297 ADC

10µF

VREF

1µF

10k100k

–

+100k

47k

0.1µFVIN

GND COMM

LT1025A

1µF

1k

1µF

–+

2N3906

LTC1050

74k

CONTROL OUTPUT

DAC LOAD

DATA

CLK

CS/ POWER DOWN

DIN

J

1257_5.eps

CLK

LOAD

DOUTGND

VREFVCC

VOUTCONTROL

OUTPUT

INLT1021-10

OUT

GND0.1µF

15V

µPLTC1257

DIN

Figure 6. 12-bit single 5V control system with shutdownFigure 5. DAC with external reference

VIN = 3.3V

VIN = 5V

LOAD CURRENT (mA)1

74

EFFI

CIEN

CY (%

)

82

90

10 100

S1300_14.eps

88

86

84

80

78

76

Figure 14. Efficiency of Figure 13’s circuit


LT1204, continued from page 7 Table 2. LT1204 Performance

Parameter Conditions Typical Value

Bandwidth AV = +2, RL = 150Ω 70MHz0.1dB gain flatness AV = +2, RL = 150Ω, no peaking >30MHzSlew rate AV = +10 1000V/µsDifferential gain AV = +2, RL = 150Ω, NTSC 0.04%Differential phase AV = +2, RL = 150Ω, NTSC 0.06°Channel select time AV = +10, VIN = 0.5V 120nsEnable time AV = +10, VIN = 0.5V 100nsDisable time AV = +10, VIN = 0.5V 50nsInput voltage range Pin 8 = 0V, VS ≥ ±10V ± 6VInput offset voltage 5mVOutput swing RL = 400Ω 13.5VSupply current 19mASupply current in shutdown Pin 12 = 0V 1.5mAOutput limit current 55mA

DESIGN FEATURES

Switchable Gain AmplifierAn example of the flexibility of the

LT1204 can be seen in a switchablegain amplifier (Figure 7), which canuse either the shutdown or disablefeature. This circuit maintains a rela-tively constant output voltage of 1VPP±330mV over a 128-to-1 change ininput level. When LT1204 #1 is se-lected, an input attenuator alters the

1204_7.eps

1

–

++

++

357

131.5k

100

124Ω

124Ω

249Ω

499Ω

VIN

LT1204 #1

1

–

++

++

357

13

LT1204 #2

1.5k

VOUT = 1VPP

Figure 7. Switchable Gain amplifierAV = 1, 2, 4, 8

LT1363, continued from page 11

1363_5.eps

LT1360

LT1357

LT1354

LT1363

1363_4.eps

LT1360

LT1357

LT1354

LT1363

input signal by 1, 0.5, 0.25, or 0.125to form an amplifier with a gain of 16,8, 4, or 2. LT1204 #2 is connected tothe same attenuator, and when it isenabled (LT1204 #1 disabled), it has again of +1 instead of +16. The secondLT1204 is used to extend the gainrange to 1, 0.5, 0.25, and 0.125.

PerformanceTable 2 summarizes the major per-

formance specifications of the LT1204.

ConclusionsThe LT1204 combines a

fast-switching multiplexer with a high-speed current-feedback amplifier forgain adjustment and cable driving.Switching transients have been greatlyreduced in amplitude and duration, allhostile and disable crosstalk have beenreduced below –90dB at 10MHz, andthe unique disable feature eases sys-tem expansion. The high performanceof this multiplexer makes it ideal forthe newest multimedia products.

Figure 5. Large signal step response, CLoad = 10,000pF:LT1354, LT1357, LT1360, LT1363

Figure 4. Small signal step response, CLoad = 200pF:LT1354, LT1357, LT1360, LT1363

ConclusionThe new topology offers significant

improvement in circuit performance.It achieves higher slew rates with lowersupply currents and improves DCspecifications and noise with higherinput-stage transconductance, lower

operating currents, and balanced in-put stages. The family of parts isalso stable with capacitive loads andcan be used in any voltage feedbackapplication.

ensuring stability. Figure 4 shows thefamily of amplifiers driving a 200pF loadand Figure 5 shows the large signal re-sponse with a 10,000pF load. Note thatthe slew rate when driving a large capaci-tive load is limited by the short-circuitcurrent limit of the amplifier.


The LT1300: Two-Cells-to-Real-World Interface by Dale Eagar

DESIGN IDEAS...

The LT1300: Two-Cells-to-Real-World Interface . 17Dale Eagar

Using a Fast AnalogMultiplexer to Switch VideoSignals for NTSC“Picture-in-Picture”Displays ........................ 21Frank Cox

High-Efficiency (>90%)NiCad Battery-ChargerCircuit Programmable for1.3A Fast Charge or 100mATrickle Charge .............. 23Brian Huffman

An LTC1087-Based1.2V GTL Terminator ..... 24Mitchell Lee

LTC1163: 2-CellPower Management ....... 25Tim Skovmand

A High-Efficiency, 5V to3.3V/5A Converter ......... 26Randy G. Flatness

A Dual-Output LCD-BiasVoltage Generator ......... 27Jon A. Dutra

A Linear-PhaseBandpass Filter for DigitalCommunications ........... 28Philip Karantzalis

IntroductionThe LT1300 micropower, high-speed

step-up DC/DC converter opensup many new applications to theuser, such as those requiring highefficiency in battery-operated equip-ment. The LT1300 can be used toproduce high voltages for many spe-

DESIGN IDEAS

Figure 1. Flame detector1300a_1.eps

VIN

SHUTDOWN

SW

ILIM SENSE

SEL GND PGND

U1 LT1300

2 1 8

5 4

7

NC

3

6

C2 0.01

R1 100k

PULSE

C1 47pF

D2 1N4148

3V

V1 R2868

3V+C3 100µF 6.3V

D1 MUR1100

D5 1N5718

D4 1N5718

C4 0.47µF

R2 1k

Q1 ZTX788

D3 1N758A

3V

V1 =

T1 =

Q1 =

C1 =

HAMAMATSU R2868 FLAME SENSOR HAMAMATSU (408) 261-2022 COILTRONICS CTX02-12186 COILTRONICS (407) 241-7876 ZETEX ZTX788 ZETEX (516) 543-7100 47pF > 500V

T1

cialized tasks with high efficiency. Hereare three such applications. In the firstapplication, the LT1300 is used toproduce 325VDC while drawing a mere200 microamps from two C-size cells.

Flame SensorAn interesting characteristic of flame

is that it emits short-wave-length ultraviolet light (<260nm). Thisshort-wavelength light falls into a win-dow of the light spectrum that isrelatively empty. Tungsten light, fluo-rescent light, and sunlight below theatmosphere are almost totally devoidof spectral energy in this window. Thecircuit shown in Figure 1 usesa photoelectric sensor with a suffi-ciently high cathode work function tomake it blind to anything with a wave-length longer than 260nm (such asnormal UV, visible light, or infrared.)Cathode work function is a measureof how hard it is to free an electronfrom an atom; when related to lightilluminating a cathode, it specifies theminimum energy of a photon that can

liberate an electron. UV photonshave higher energy than visible light.

Theory of OperationThe LT1300 and transformer T1

form a flyback converter to step up thevoltage from 3V to 325V. The second-ary winding of T1 connects throughD1 (an MUR1100) to C1, a holdingcapacitor for the 325VDC, which, inturn, is applied to the anode of thephotoelectric sensor tube V1. TheLT1300 SW pin senses the volt-age on C1, as scaled by the turnsratio, through T1. The voltage on theprimary winding is programmed to be10.6V, translating to 325V on C1.When C1 has charged to 325V, the

1300a_2.eps

+VLOGIC

INTR

VN2222PULSE

10k

Figure 2. FET inverter/microprocessorinterface


The tube V1 has 325V across its ter-minals to get sufficient energy intoa liberated photo electron to ionizethe gas that fills the tube. Once thegas in the tube ionizes, there are moreelectrons available; they cause a chainreaction in the tube that causes thetube to avalanche. When the tubeavalanches, most of the charge on C1is transferred to C2 and the voltageacross C1 drops to a fraction of itsoriginal 325V. When C2 has chargedto 3.6V, all the excess charge residingin C1 gets dumped through D2 intothe battery. The voltage across C2 isthe output signal called PULSE. PULSEasserts the shutdown pin of theLT1300, allowing the plasma in thephotoelectric tube to quench. Figure 2shows an interface circuit that en-ables the PULSE signal to interrupt amicroprocessor.

1300a_3.eps

–

+U1B

LT1179

+

–U1C

LT1179

–

+U1A

LT1179

–

+U1D

LT1179

U1

R21 5.1Ω

3V

R22 180Ω

C7 4.7µF

R8 100kPOT 1

SENSITIVITY ADJ 1M

3V

R7 3M

R9 30k

R10 1M

C6 1µF

FLAME

D1 1N4148

ALARM 3V 3V

LED1

3V ALARM DEVICE

D2 1N4148

Q1 2N2222

R20 10k

A1R13 560k

R12 390k

REPLACE

R11 560k

3VSMOOTHED

U2 LT1004-1.2

D3 1N4148R18

470kR19 100k

3V

R17 390k

R16 510k

R15 750k

R14 1.2M

3V

R6 1M

R5 10M

R4 10k

C3 0.15

R3 300k

C2 1µF

R2 300k

C1 0.68

R1 300k

PULSE

C4 0.1

U1 = A1 =

LT1179 ARCHER 273-065A

C8 0.1

C5 0.01

+

3V

4

11

For you analog purists, Figure 3shows a discriminator circuit withlow battery detect for a complete 3Vflame alarm. The discriminator isneeded because the photo detectoroccasionally detects a cosmic rayor some rare room-light photon.The discriminator consists of foursections.

Filter Section(U1a, R1–R6, C1–C4)

In this section, the PULSE signal isfiltered with a third-order Gaussianlowpass filter. Each PULSE is con-verted into a Gaussian-shaped pulseabout 300ms wide. These Gaussian-shaped pulses are superimposed andcan be seen on the SMOOTHED testpoint. The net effect of this filtering isto accumulate PULSEs that are closetogether (<300ms) into a single DC

feedback loop comprised of D3, R2,and Q1 kicks in and charges C4through D4. When the voltage at C4exceeds 3.3V, the LT1300 goes intoits wait mode. In wait mode theLT1300 consumes only 100µA of cur-rent. The LT1300 stays in wait modeuntil the voltage on C4 falls below3.3V, at which time the LT1300 turnson to burst recharge both C1 and C4.Burst ModeTM operation ensures 30Hzoscillation in this system. This rate isdetermined by the value of C4, theinternal sense resistance to groundin the LT1300 (approximately 1 MΩ),and the amount of overcharge C4gets when charging.) D5 is a Schottkycatch diode to keep reverse currentout of U1.

When illuminated with a photon ofsufficient energy, the photoelectrictube’s cathode liberates an electron.

Figure 3. Discriminator circuitry

DESIGN IDEAS


voltage. When the input PULSE hap-pens frequently, the SMOOTHED testpoint goes positive, indicating manyrecently detected photons.

Threshold-Setting Section(U1b, POT1, R1–R9, C5)

POT1 sets the threshold of thealarm; U1b compares the thresholdwith the SMOOTHED signal from theprevious section and pulls its outputlow when it detects more voltage onSMOOTHED than on the threshold.This output is the test point labelledFLAME.

Pulse-Stretcher Section:(D1, R10–R13, C6, U1c)

When FLAME goes low, C6 dis-charges through D1 and is slowlyrecharged through R10. U1c is acomparator with hysteresis that out-puts a high signal (test point ALARM)when FLAME goes low. Because of thetime constant of R10 and C6, ALARMstays high for about one second afterFLAME goes high. The ALARM outputalso drives the base of Q1 through D2to sound the sounding device A1.

Low-Battery-Detect Circuit(R14–R19, C7, D3, U1d)

This sub-circuit is a divider/com-parator/oscillator that is activatedwhen the battery voltage drops to 2V.The output is a positive-going 1/4-second signal called REPLACE.REPLACE initially occurs every threeseconds at a battery voltage of 2V, butthe frequency of repetition increasesas the battery voltage drops. At abattery voltage of 1.5V, the REPLACEsignal frequency is approximately 2Hz.REPLACE also sounds the soundingdevice A1 through R20 and Q1. R21and C8 form a trash com-pactor to decouple U1.

The complete circuit, including thedetector (Figure 1) and the discrimina-tor (Figure 3) consume a mere300µA of supply current from 3V. Thecircuit lends itself very well to bothbattery operation and two-wire re-mote operation. Battery life is morethan two years when powered by twoC-size alkaline cells. In full sun, the

DESIGN IDEAS

and the input offset voltage is lessthan 0.05 volts. The output voltagetracks the input voltage from 0V to520V. For safety (and to isolate theinput capacitance) a 100MΩ resistoris placed in series with the input,but with the ±570pA of input biascurrent (over temperature) for theLT1097, this translates into only±57mV of additional offset. The inputimpedance of this buffer measuresfour trillion ohms when measuredwith a 100-to-400 volt input. Thedetailed circuit is shown in Figure 5.

Theory of OperationU1 monitors the voltage difference

between the circuit’s noninvert-ing input and output and attemptsto make it zero. If the voltage on thenoninverting input is less than thevoltage on the noninverting output,U1’s output goes positive, turning Q1on slightly. Q1 acts as a currentsource, discharging C3. When the

detector can easily “see” a cigarettelighter 30 feet away and still discrimi-nate it from the sun.

Infinite-Input-ImpedanceVoltage Buffer

In the flame detector circuit (Fig-ure 1), it is difficult to measure thevoltage across C1 because almost anyload invalidates the meter reading.This next application for the LT1300is a voltage buffer that overcomesthis measurement problem. This is afour-terminal, unity-gain buffer, asshown functionally in Figure 4. Theinput impedance is essentially infi-nite, the input bias current is negligible,

Figure 5. Voltage buffer schematic

Figure 4. Voltage buffer block diagram

1300a_4.eps

+

+

–E

ISOLATION

+IN

–IN

+OUT

–OUT

1300a_5.eps

VIN

SENSE

SWILIM

SHUT- DOWN

U2 LT1300

T1

2 1 8

4

5 7

6

3

C6 100µF 6.3V

+

NC

R9 5.1Ω

3V

Z1 15V

D1 1N4148

C5 100µF

6.3V

+

C4 220pF

D2 1N4148

3V

R8 1M

C3 0.01

R7 1k

Q1 2N3904

R6 10k

R5 10k

+

–

3

2

6

U1 LT1097

R4 100k

C2 0.01

R2 1M

R3 1M

C1 1000pF

R1 100M

+ IN

+ OUT

C8 0.1

R11 20Ω

3V

7

4

–OUT

–IN

R10 100M

C7 1000pF D3

MUR1110

1.5V

1.5V

SW1

3V

T1 =

R1, R10 =

COILTRONICS CTX02-12179 COILTRONICS (407) 241-7876 VICTOREEN SLIM-MOX100 VICTOREEN (216) 248-9300

U1 LT1097


Z1 dissipates the energy stored inT1’s inductance. During the flybacktime, C4 charges C3 through D1.The voltage across C3 exceeds0.6V, shutting down U2. U2 staysshut down until Q1 discharges C3 torestart the sequence.

When the +output voltage is morepositive than the +input voltage, theoutput of U1 goes low, Q1 stays off, R8keeps C3 charged to more than 0.6V,and U2 stays shut down. The parallelcombination of R10 and the load re-sistance (e.g., 10MΩ in a handheldvoltmeter) discharges C7 and the+output and the +input voltages areagain equal. The current output ofthis circuit is limited to a safe value(1mA at 50V, 0.1mA at 500V), evenwhen the +input is attached to +500V.We do not recommend increasingthe value of C7, because at highervoltages it may become a shockhazard. Battery life 40 hours for apair of AA alkaline batteries driving10MΩ at 500V.

Cold-CathodeFlorescent Lamp Driver

CCFLs seem to be the latest craze;they offer high brightness, long life,small size, and produce white light.Figure 6 shows a CCFL driver circuit.

Theory of OperationThis is a forward/flyback inverter

optimized for minimum parts count.When enabled, U1 charges the pri-mary winding of T1 to 1 amp, and letsgo. T1 then flies back, exciting manyhundreds of volts across its second-ary winding, which, in turn, ionizesthe CCFL. Because the initial currentthrough the CCFL is only in one direc-tion, C2 takes on a DC potential. Asthe circuit runs, the voltage across C2stabilizes at about 100VDC.Additionally, C2 removes the DC com-ponent from the tube current,extending tube life. The nonlinear V/Icharacteristic of the CCFL, in con-junction with C2, forces the converterto run in both forward and flybackmodes simultaneously. The light in-tensity can be pulse-width modulated


DESIGN IDEAS

Figure 7. Electronic light stick controllerFigure 6. CCFL driver

1300a_7.eps

+

–LT1178

C4 0.1R7

10Ω

3V

8

4

U1A LT11783

2

R4 200k

R2 270k

R3 270k

R1 300k

1

3V

+

–U1B

LT11785

6

C2 470µF

R5 1k R6

1M7

3V

SAWTOOTH

C3 0.01

PULSE WIDTH OUT

D1 2N3904

LIGHT LEVEL PROGRAM

D1 1N4148

ONSUNSET

OFF +

R7 100M

C1 0.022

1300a_6.eps

VIN

SHUTDOWN

SW

ILIM SENSE

SEL GND PGND

U1 LT1300

2 1 8

5 4

7

NC

3

6

PWM IN

C1 100µF

6.3V

3V

CCFL LAMP =

T1 =

C2 =

JKL BF650-20B JKL (800) 897-3056 COILTRONICS CTX02-12189 COILTRONICS (407) 241-7876 15pF, 500V

T1

+

D1 1N5718

C2 15pF

CCFL

voltage on C3 falls below approxi-mately 0.6V, U2 is enabled. When it isenabled, U2 turns its switch on (U2pin 7 pulls low, to near 0V). Thiscauses approximately 3V to be im-posed across the primary winding ofT1. The magnetizing inductance ofthe primary winding of T1, acrosswhich a voltage is applied, requires asteadily increasing current. At thesame time, C4 is charging throughD2. When the current flowing throughthe switch of the LT1300 reaches 1amp, the LT1300 switches off. Themagnetizing inductance of the pri-mary winding of T1, seeing that theLT1300 is attempting to discontinuecurrent flow, takes over by swingingpositive in voltage until it finds some-thing that will take the 1 amp ofmagnetizing current. While the pri-mary winding is finding somewhereto put the magnetizing current, thesecondary winding takes it upon it-self to do the same, but due to itsturns ratio with the primary winding,it moves 100 times faster and 100times as far as the primary winding.T1’s secondary dumps a significantportion of the magnetizing energyinto C7 via D3, thus forming a flybackinverter.

by modulating the shutdown pin.When the shutdown pin is pulledhigh, the LT1300 goes into its shut-down mode, where it draws only 10µAof input current.

Electronic Light StickCamping in November with my

kids has its own unique problems,even if we aren’t camping in six feet ofsnow. Although we had the usuallight sources, something was miss-ing, namely a light that simulates thenatural sunset at bedtime to wind thekids down for the night. The circuit inFigures 6 and 7 (see explanationbelow) details a high efficiency fluo-rescent lantern with a built in sunsetfeature.

The function of the circuit is asfollows: To turn on: switch SW1 into the

ON position. To turn off fast: switch SW1 into

the OFF position. To simulate sunset: 1. turn light

ON. 2. switch SW1 into theSUNSET position.This application uses the circuitry

of both Figure 6 and Figure 7. Thepulse-width output of Figure 7 drivesthe pulse-width input of Figure 6.


Using a Fast Analog Multiplexerto Switch Video Signals for NTSC“Picture-in-Picture” Displays

DESIGN IDEAS

IntroductionThe majority of production1 video

switching consists of selecting onevideo source out of many for signalrouting or scene editing. For thesepurposes, the video signal is switchedduring the vertical interval in order toreduce visual switching transients.The image is blanked during this time,so if the horizontal and vertical syn-chronization and subcarrier lock aremaintained, there will be no visibleartifacts. Although vertical-intervalswitching is adequate for most rout-ing functions, there are times when itis desirable to switch two synchro-nous video signals during the active(visible) portion of the line to obtainpicture-in-picture, key, or overlay ef-fects. Picture-in-picture or active videoswitching requires signal-to-signaltransitions that are both clean andfast. A clean transition should have aminimum of pre-shoot, over-shoot,ringing, or other aberrations com-monly lumped under the term“glitching.”

Using the LT1204A quality, high-speed multiplexer

amplifier can be used with good

results for active video switching. Theimportant specifications for this ap-plication are small, controlledswitching glitch, good switching speed,low distortion, good dynamic range,wide bandwidth, low path loss, lowchannel-to-channel crosstalk, andgood channel-to-channel offset match-ing. The LT1204 specifications matchthese requirements quite well, espe-cially in the areas of bandwidth,distortion, and channel-to-channelcrosstalk (which is an outstanding90dB at 10MHz). The LT1204 wasevaluated for use in active videoswitching with the test setup shownin Figure 1. Figure 2 shows the videowaveform of a switch between a 50%white level and a 0% white level about30% into the active interval and backagain at about 60% of the activeinterval. The switch artifact is briefand well controlled. Figure 3 is anexpanded view of the same waveform.When viewed on a monitor, the switchartifact is just visible as a very fineline. The lower trace is a switch be-tween two black level (0V) video signalsshowing a very slight channel-to-channel offset, which is not visible on

the monitor. Switching between twoDC levels is a worst-case test, asalmost any active video will haveenough variation to totally obscurethis small switch artifact.

Figure 1. “Picture-in-picture” test setup

COX_1.eps

MONITOR

OFF AIR VIDEO SOURCE OR VIDEO

PATTERN GENERATOR

SYNC STRIPPER, SAMPLE PULSE

GENERATOR

OUTPUT

75

75

75

75

LT1204

75Ω

SCOPE

SAMPLE PULSE

50%

INPUTS

LOOP THROUGH

Figure 2. Video waveform switched from 50%white level to 0% white level and back

Figure 3. Expaneded view of rising edge ofLT1204 switching from 0% to 50% (50nshorizontal division)

by Frank Cox


50ns/DIVCOX_p3.eps

COX_5.eps

RISE TIME ≈ 1 2fC

DELAY ≈ N 2fC

(WHERE N IS ORDER OF FILTER)

1 fC

Figure 5. Pulse response of an ideal sharp-cutoff filter at frequency fC

Figure 4. Expanded view of “brand-x” switch0%–50% transition

Some Definitions—“Picture in picture” refers to theproduction effect in which one videoimage is inserted within the bound-aries of another. The process maybe as simple as splitting the screendown the middle or it may involveswitching the two images along acomplicated geometric boundary.In order to make the compositepicture stable and viewable, both videosignals must be in horizontal andvertical sync. For compositecolor signals, the signals must also bein subcarrier lock.

“Keying” is the process of switchingamong of two or more video signals,triggering on some characteristic ofone of the signals. For instance, achroma keyer will switch on thepresence of a particular color.Chroma keyers are used to insert aportion of one scene into another.In a commonly used effect, the TVweather person (the “talent”) ap-pears to be standing in front of acomputer generated weather map.Actually, the talent is standing infront of a specially colored back-ground; the weather map is aseparate video signal, which hasbeen carefully prepared to containnone of that particular color. Whenthe chroma keyer senses the keyingcolor, it switches to the weathermap background. Where there isno keying color, the keyer switchesto the talent’s image.

DESIGN IDEAS

troublesome (that is, more visible)than one that has more amplitudebut decays quickly. The LT1204 hasa switching glitch that is not only lowin amplitude but well controlled andquickly damped. Refer to Figure 4,which shows a video multiplexer thathas a long, slow-settling tail. This sortof distortion is highly visible on avideo monitor.

Composite video systems, such asNTSC, are inherently band-limitedand thus edge-rate limited. In asharply band-limited system, the in-troduction of signals that containsignificant energy higher in frequencythan the filter cutoff will cause distor-tion of transient waveforms (see Figure5). Filters used to control the band-width of these video systems shouldbe group-delay equalized to minimizethis pulse distortion. Additionally, ina band-limited system, the edge ratesof switching glitches or level-to-leveltransitions should be controlled toprevent ringing and other pulse aber-rations that could be visible. Inpractice, this is usually accomplishedwith pulse-shaping networks (Besselfilters are one example). Pulse-shap-ing networks and delay-equalizedfilters add cost and complexity tovideo systems and are usually found

Video-Switching CaveatsIn a video processing system that

has a large bandwidth compared tothe bandwidth of the video signal, afast transition from one video level toanother (with a low-amplitude glitch)will cause minimal visual disturbance.This situation is analogous to theproper use of an analog oscilloscope.In order to make accurate measure-ment of pulse waveforms, theinstrument must have much morebandwidth than the signal in ques-tion (usually five times the highestfrequency). Not only should the glitchbe small, it should be otherwise wellcontrolled. A switching glitch thathas a long settling “tail” can be more



High-Efficiency (>90%) NiCad Battery-Charger Circuit Programmable for 1.3AFast Charge or 100mA Trickle Charge

by Brian Huffman

Battery-charger circuits are ofuniversal interest to laptop,notebook, and palmtop computermanufacturers. High efficiency is de-sirable in these applications tominimize the power dissipated in thesurface mount components. The cir-cuit shown in Figure 1 is designed tocharge four NiCad cells at a 1.3A fastcharge or a 100mA trickle charge,with efficiency exceeding 90%. Thiscircuit can be modified easily to handleup to eight NiCad cells.

The circuit uses an LTC1148 in astep-down configuration to controlthe charge rate. The LTC1148 is asynchronous switching-regulatorcontroller that drives external,complementary power MOSFETs us-ing a constant off-time current modearchitecture. When the LTC1148’sP-drive output pulls the gate of Q1

low, the P-channel MOSFET turns onand connects one side of the inductorto the input voltage. During this pe-riod, current flows from the inputthrough Q1, through the inductor,and into the battery. When theLTC1148 P-drive pin goes high, Q1 isturned off, and the voltage on thedrain of Q1 drops until the clampdiode is for-ward biased. The diodeconducts for a very short period oftime, until the LTC1148 internal cir-cuitry senses that the P-channel isfully off, pre-venting the simultaneousconduction of Q1 and Q2. Then the N-drive out-put goes high, turning onQ2, which shorts out D1. Now theinductor current flows through the N-channel MOSFET instead of throughthe diode, increasing efficiency. Thistype of switching architecture is knownas synchronous rectification.

During the fast-charge interval,the resistor divider network (R4 andR5) forces the LTC1148’s feedbackpin (VFB) below 1.25V, causing theLTC1148 to operate at the maximumoutput current. R3, a 100mΩ resis-tor, senses the current and sets it atapproximately 1.3A, according toequation 1. When the batteries aredisconnected, the error amplifierforces the feedback pin to 1.25V,limiting the output voltage to 8.1V.Diode D2 prevents the batteries fromdischarging through the divider net-work when the charger is shut down.In shutdown mode, the circuit drawsless than 20µA from the input supply.

The dual-rate charging is controlledby Q3, which can be toggled betweenfast charge and trickle charge. Thetrickle-charge rate is set by resistorR1. Figure 2 is a graph showing the

DESIGN IDEAS


Brian_1.eps

SHUTDOWN

ITH

CT

PDRIVE

SENSE+

SENSE–

VFB

NDRIVE

C5 200pF NPO

C4 3300pF X7R

R2 1k

R1 51

Q3 VN2222LL

C7 100pF

Q2 Si9410DY

C6 0.01µF

Q1 Si9430DY

C2 0.1µF

C1 1µF

VIN 8V - 15V

0V = >1.5V =

NORMAL SHUTDOWN

VIN

3

1

8

7

9

14

1211

10

6

4

SGND PGND

LTC1148

C3 22µF x2 25V

+

VBATT 4 CELLS

D1 MBRS140T3

L1 50µH R3

100mΩ

R4 274k 1%

R5 49.9k 1%

C8 220µF 10V

D2 MBRS340T3

“1” TRICKLE CHARGE

(TA) AVX (TA) TPSD226K025R0200 ESR = 0.200 IRMS = 0.775A AVX (TA) TPSE227M010R0100 ESR = 0.100 IRMS = 1.149A SILICONIX PMOS BVDSS = 20V RDSON = 0.125 CRSS = 400pF QG = 25nC θJA = 50°C/W SILICONIX NMOS BVDSS = 30V RDSON = 0.050 CRSS = 160pF QG = 50nC θJA = 50°C/W MOTOROLA SCHOTTKY VBR =40V KRL SP-1/2-A1-0R100J Pd = 0.75V COILTRONICS CTX50-4 DCR = 0.175 IDC = 1.350A KOOL Mµ CORE ALL OTHER CAPACITORS ARE CERAMIC COILTRONICS (407) 241-7876 KRL (809) 668-3210

C1 = C3 = C8 = Q1 = Q2 =

D1, D2 = R3 = L1 =

VOUT = 1.25V • (1 + R4/R5) = 8.1V FAST CHARGE = 130mV/R3 = 1.3A (EQ. 1) TRICKLE CHARGE = 100mA (SEE FIGURE 2)

+

1

4

2

3 VOUT

Figure 1. Schematic diagram: 4 cell, 1.3 amp battery charger implemented in surface mount technology


Theory of OperationU1a, R1–R4, and C1 form a

sawtooth oscillator for pulse-widthmodulating the light (implementinglight levels less than 100%). U1b actsas a comparator, comparing thesawtooth output of the oscillator withthe programmed light level (as seenon the + terminal of C2). C2 is theholding cap that programs the lightlevel; when it is charged to 2.5V, thelight is on 100% of the time. Asthe voltage on C2 drops below 2.5V,the overall light level decreases,because the light is being pulse-widthmodulated. When the voltage on C2

LT1300, continued from page 20

which form an exponential RC timeconstant. The second exponential as-pect of the SUNSET decay isimplemented because R6 is driven byU1b pin 7, whose duty factor ischanging,causing the off time to de-crease exponentially as the light levelfades. The output of U1b is a pulse-width-modulated level gating the lightdriver on and off. The lamp is illumi-nated when U1b’s output is low. C3 isa trash compactor and R7 and C4 forma trash compactor to decouple U1from the high-frequency ripple gener-ated by the switcher.

is at or below 1 volt, the light is off.D1 and R5 charge and hold C2 whenSW1 is in the ON position. R5 andSW1 discharge and hold C2 whenSW1 is in the OFF position. The combi-nation of D2, R6, and U1b dischargeC2 when SW1 is in the SUNSET posi-tion. The discharging of C2 when inthe SUNSET mode is doubly exponen-tial, causing the tail end of thesimulated sunset to go very slowly (agood idea, because kids have a loga-rithmic response to light). The firstexponential aspect of the SUNSET de-cay is implemented by R6 and C2,

DESIGN IDEAS

by Mitchell Lee

A recent development in high-speeddigital design has resulted in a newfamily of logic chips called GunningTransition Logic (GTL). Because of thespeeds involved, careful attention mustbe paid to the transmission-linecharacteristics of the interconnec-tions between these chips; activetermination is required.

The termination voltage is 1.20V,and currents of several amperes arecommon in a complete system. Onemethod of generating 1.2V is to use alinear regulator operating from 3.3Vor 5V. Unfortunately, this methodsuffers from two major drawbacks.First, the minimum adjust voltage,

without the aid of a negative supply,is 1.25V for most adjustable linearregulators. Second, most low-voltagelinear regulators do not feature lowdropout characteristics, renderingthem unusable on a 3.3V input. TheLT1087 solves both of these problemswith an output that can be adjustedto less than the reference voltage anda low-dropout architecture.

Figure 1 shows the complete cir-cuit. The LT1087 features feedbacksense, which, in its original applica-tion, was used for remote Kelvinsensing. In the GTL terminator circuit,the sense pins are used to adjust theinternal 1.25V reference downward.

value of R1 for a given trickle-chargeoutput current. The trickle-charge cur-rent can also be varied by using an opamp to force the threshold pin voltagewithin its 0V–2V range. Figure 3 showsthe output current as a function ofthreshold pin voltage.

NiCad, continued from page 23

R1 (kΩ)0

0

I OU

T (m

A)

200

1400

2 4

Brian_2.eps

400

600

800

1000

1200

1 3THRESHOLD PIN VOLTAGE (V)

00

I OU

T (m

A)

200

1400

1.0 2.0

Brian_3.eps

400

600

800

1000

1200

0.5 1.5

Figure 3. LTC1148 output current versusforced threshold pin voltage

Figure 2. LTC1148 output current voicetrickle charge set resistance (R1)

An LT1087-Based 1.2V GTL TerminatorThe result is a 1.20V, 5A regulator,with 2% output tolerance over all con-ditions of line, load, and temperature.To minimize power dissipation, a 3.3Vinput source is recommended.

Figure 1. Schematic diagram: 1.2 volt GTLtermination voltage generator

GTL_1.eps

LT1087CT

22µF Ta

10µF

VIN = 3.3V

+

1.21kΩ 1%

4.42Ω 1%

VOUT

VIN

+SENSE

–SENSE

GND

1.20V/5A 2%

+


LTC1163: 2-CellPower Management

Figure 2 Complete 2-cell to 3.3V, 5V, and 12V power management system

1163_2.eps

+

4 5

8

3

2147

100µH

LT1173CS8 39k

24k 220µF 6.3V

3.3V

MBRS120T3

+

4

10µF 20V

12V

MBRS120T3

22µH

31

7 LT1109CS8-12

LTC1163CONTROL LOGIC OR µP

2-CELL BATTERY PACK

8

+

4

22µF 16V

5V

MBRS120T3

22µH

31

7 LT1109CS8-5 8

IN1

IN2

IN3

OUT1

OUT2

OUT3

VS

GND

+100µF 6.3V

RFD14N05LSM

RFD14N05LSM

RFD14N05LSM

+

The LTC1163 1.8V to 6V high-sideMOSFET driver allows inexpensiveN-channel switches to be used toefficiently manage power in 2-cell sys-tems such as palmtop computers,portable medical equipment, cellulartelephones and personal organizers.

Any supply voltage above 3V, suchas: 3.3V, 5V or 12V, can be generatedby step-up converters powered from a2-cell supply. Step-up regulators aretypically configured as shown inFigure 1. An inductor is connected

directly to the 2-cell battery packand switched by a large (1A) switch.The inductor current is then passedthrough a low-drop Schottky rectifierto charge the output capacitor to avoltage higher than the input voltage.Unfortunately, when the regulator isshut down, the inductor and dioderemain connected and the load mayleak significant current in standby.

One possible solution to this prob-lem is to add a low RDS(ON) MOSFETswitch between the battery pack and

the input of the regulator to com-pletely disconnect it and the loadfrom the battery pack. MOSFETswitches, however, cannot operatedirectly from 2-cell battery sup-plies because the gate voltage islimited to 3V with fresh cells and 1.8Vwhen the cells are fully discharged.

The LTC1163 solves this problemby generating gate drive voltageswhich fully enhance high-sideN-channel switches when poweredfrom a 2-cell battery pack as shownin Figure 2. The standby currentwith all three drivers switched offis typically 0.01µA. The quiescentcurrent rises to 85µA per channelwith the input turned on and thecharge pump producing 10V (aboveground) from a 3V supply. Thesurface mount MOSFET switchesshown are guaranteed to be lessthan 0.1Ω with VGS = 5V and less than0.12Ω with VGS = 4V and thereforehave extremely low voltage drops.

1163_1.eps

+

L1

STEP-UP SWITCHING REGULATOR

COUT

D1

2-CELL BATTERY PACK

+ +CIN

LOAD

SHUTDOWN

Figure 1. Typical step-up converter topology

by Tim Skovmand

DESIGN IDEAS


The next generation of notebookand desktop computers is incorporat-ing more 3.3V ICs alongside 5Vdevices. As the number of devicesincreases, the current requirementsalso increase. Typically, a high-current 5V supply is already available.Thus, the problem is reduced to de-riving 3.3V from 5V efficiently in asmall amount of board space.

High efficiency is mandatory inthese applications, since converting5V to 3.3V at 5A using a linear regula-tor would require dissipatingover 8W. This wastes power andboard space for heat sinking.

The LTC1148 synchronous switch-ing-regulator controller accomplishesthe 5V to 3.3V conversion with highefficiencies over a wide load-currentrange. The circuit shown in Figure 1provides 3.3V at efficiencies greaterthan 90% from 5mA to 5A (overthree decades of load current). The

A High-Efficiency,5V to 3.3V/5A Converter

efficiency of the circuit in Figure 1 isplotted in Figure 2.

At an output current of 5A theefficiency is 90%; this means only1.8W are lost. This lost power isdistributed among RSENSE, L1, andthe power MOSFETs; thus heat sink-ing is not required.

The LTC1148 series of controllersuse constant off-time current-modearchitecture to provide clean start-up, accurate current limit, andexcellent line and load regulation. Tomaximize the operating efficiency atlow output currents, Burst ModeTM

operation is used to reduce switchinglosses. Synchronous switching, com-bined with Burst ModeTM operation,yields very efficient energy conver-sion over a wide range of load currents.

The top P-channel MOSFETs inFigure 1 will be on 2/3 of the timewith an input of 5V. Hence, thesedevices should be carefully examined

to obtain the best performance. TwoMOSFETs are needed to handle thepeak currents safely and enhancehigh-current efficiency. The LTC1148can drive both MOSFETs adequatelywithout a problem. A single N-channelMOSFET is used as the bottomsynchronous switch, which shuntsthe Schottky diode. Finally, adap-tive anti-shoot-though circuitryautomatically prevents cross conduc-tion between the complementaryMOSFETs, which can kill efficiency.

The circuit in Figure 1 has a no-load current of only 160µA. Inshutdown mode, with pin 10 heldhigh (above 2V), the quiescent cur-rent reduces to less than 20µA withall MOSFETs held off DC. Althoughthe circuit in Figure 1 is specifiedat a +5V input voltage, the circuitwill function from 4V to 15Vwithout requiring any componentsubstitutions.

by Randy G. Flatness

Figure 2. Efficiency for 5V to 3.3V synchronousswitcher

DESIGN IDEAS

1148_2.eps

IOUT (mA)1

70

EFFI

CIEN

CY (%

)

100

10 10000100

90

1000

80

Figure 1. LTC1148-3.3 high-efficiency 5V to 3.3V/5A step-down converter

1148_1.eps

TANTALUM SANYO (OS-CON) 20SA100M ESR = 0.037Ω IRMS = 2.25A AVX (TA) TPSE227K01R0080 ESR = 0.080Ω IRMS = 1.285A SILICONIX PMOS BVDSS = 20V DCRON = 0.100Ω Qg = 50nC SILICONIX NMOS BVDSS = 30V DCRON = 0.050Ω Qg = 30nC MOTOROLA SCHOTTKY VBR = 30V KRL NP-2A-C1-0R020J Pd = 3W KOOL Mµ CORE, 16 GAUGE COILTRONICS (408)241-7876 KRL BANTRY (603) 668-3210 SILICONIX (800) 554-5565

C1 = C3 = C6 =

Q1, Q2 = Q3 = D1 = R2 = L1 =

SHUTDOWN

ITH

CT

LTC1148-3.3

VIN

SENSE +

SENSE –

+

C7 0.01µF

+

C2 0.1µF

R1 470Ω

C4 3300pF

C5 680pF NPO

10

6

4

1

8

7

Q2 Si9430DY

Q3 Si9410 D1

MBRS140T3

C3 100µF 20V X2

L1 27µH

R2 20mΩ

VOUT 3.3V/5A

C6 220µF

10V X2

3

11

12

PDRIVE

NDRIVESGND PGND

+

Q1 Si9430DY

C1 1µF

VIN 5V

0V = >2V =

NORMAL SHUTDOWN


With the many different kinds ofLCD displays available, systemsmanufacturers often want theoption of deciding the polarity of theLCD-bias voltage at the time ofmanufacturing.

The circuit in Figure 1 uses theLT1107 micropower DC-to-DC con-verter with a single inductor. TheLT1107 features an ILIM pin that en-ables direct control of maximuminductor current. This allows the useof a smaller inductor without the riskof saturation. The LT1111 could alsobe used, with a resulting reduction inoutput power.

Circuit OperationThe circuit is basically an AC-

coupled boost topology. The feedbacksignal is derived separately from theoutputs, so loading of the outputsdoes not affect loop compensation.Since there is no direct feedback fromthe outputs, load regulation perfor-mance is reduced. With 28 volts out,from 10% to 100% load (4mA to40mA), the output voltage sags by

A Dual-Output LCD-BiasVoltage Generator by Jon A. Dutra

about 0.65 volts. From 1mA to 40mAload, the output voltage sags by about1.4 volts. This is acceptable for mostdisplays.

Output noise is reduced by usingthe auxiliary gain block (AGB) in thefeedback path. This added gain effec-tively reduces the hysteresis of thecomparator and tends to randomizeoutput noise. With low-ESR caps forC2 and C4, output noise is below30mV over the output load range.Output power increases with VBATTERY,from about 1.4 watts out with 5 voltsin to about 2 watts out with 8 volts ormore. Efficiency is 80% over a broadoutput-power range.

If only a positive or negative outputvoltage is required, the two diodesand two capacitors associated withthe unused output can be eliminated.The 100kΩ load is required on eachoutput to load a parasitic voltagedoubler created by the capacitance ofdiodes D2 and D4. Without this mini-mum load, the output voltage can goup to almost 50% above the nominalvalue.

LCD_1.eps

++

+ +

++

VIN 3V to 12V

10µF 16V

100k 30Ω 10µF 16V

VBATTERY 4V to 16V

L1

L1 =

C1, C2, C3 and C4 =

=

COILTRONICS CTX 50-4

22µF, 35V LOW ESR

1N5819 or MBR140

(OPTION SEE TEXT)

C1

C2 100k

+VO 24V TO 32V (0mA TO 40mA)

SHUTDOWN IN “1” = OFF

C3

C4 100k

–VO –24V TO –32V (0mA TO 40mA)

1N4148 10k

0.01

1N4148

2.32M

100k1.43M

AO

FB

SW1

SET

VIN ILIM

SW2 GND

5V CONTRAST ADJ. 1M POT.

LT1107CS8

1.25V

D3

D4

D1

D2

Component SelectionThe voltage at the switch pin SW1

swings from zero volts to VOUT plustwo diode drops. This voltage is ACcoupled to the positive outputthrough C1 and D1 and to the nega-tive output through C3 and D3. Thefull output current flows through C1and C3. Most tantalum capacitors arenot rated for current flow, and theiruse can result in field failures. Use arated tantalum or a rated electrolyticfor longer system life. At lower outputcurrents or higher frequencies, usingmonolithic ceramics is also feasible.

One could replace the 1N5819Schottky diodes with 1N4148 typesfor lower cost, with a reductionin efficiency and load-regulationcharacteristics.

ShutdownThe circuit can be shut down in

several ways. The easiest is to pull theset pin above 1.25 volts; however, thisconsumes 300µA in shutdown condi-tions. A lower power method is toturn off VIN to the LT1107 by meansof a high-side switch or by simplydisabling a logic supply. This dropsquiescent current from the VBATTERYinput below 10µA. In both cases VOUTdrops to zero volts. In the that event+VOUT does not need to drop to zero,C1 and D1 can be eliminated.

Output Voltage AdjustmentThe output voltage can be adjusted

from any voltage above VBATTERY up to46 volts with proper passive compo-nents. Output voltage can becontrolled by the user with DAC,PWM, or potentiometer control. Bysumming currents into the feedbacknode, the output voltage can be ad-justed downward.

This design idea originally appeared in EDNmagazine.

DESIGN IDEAS

Figure 1. Schematic diagram LT1107 dual-output LCD-bias generator


Bandpass filters with linear pass-band phase are useful for a variety ofdata communications tasks, the mostnoteworthy of which may be inmodulation-demodulation (modem)circuitry. Modems generate signalsthat must be processed without phasedistortion to allow error-free trans-mission and reception of information(or the closest approach to that idealwe can achieve).

Figure 1 shows a linear-phasebandpass filter using the LTC1264high-frequency, universal switched-

capacitor filter building block. Thisfilter is an eighth-order narrowbandpass filter, centered at 50kHz fora 1MHz clock input, with flat groupdelay in its passband. The fCLK-to-fCENTER frequency ratio is 20:1. Figure2 shows the filter’s narrow-band gainresponse and Figure 3 shows thepassband group delay.

An interesting feature of linear-phase bandpass filters is that theirresponse to a step input produces ashort transient sine wave burst witha symmetrical envelope. Figure 4

A Linear-Phase Bandpass Filter forDigital Communications by Philip Karantzalis

shows a comparison of the transientresponses to a step input for thelinear phase bandpass filter of Figure1 and a bandpass filter with a similarpassband and nonlinear phase re-sponse. The response of a bandpassfilter to a step input is a simple qualita-tive test for determining the linearity ofits phase response, although indata transmission systems the mea-surements are usually made witheye-diagrams and constellationdisplays.

The maximum clock frequency forthe filter is 2MHz with ±7.5 volt sup-plies. This allows bandpass filterswith center frequencies up to 100kHzto be realized without significantphase distortion in the passband.

Capacitor C, across R4 in sectionsC and D, minimizes gain and phasevariations when the filter is used withclock frequencies greater than1.4MHz. For ±5 volt supplies, themaximum clock frequency is 1.6MHz.Use the Table 1 as a guide for theselection of capacitor C.

1264_1a.eps

R2A, 10.7k

R3A, 12.4k

R4A, 11.5k

0.1µF

R4B, 13.3k

R3B, 29.4k

R2B, 10.7k

R2D, 10k

R3D, 29.4k

R4D, 10.7k

R4C, 10.7k

R3C, 37.4k

R2C, 12.4k

RlB, 30.1k

RhB, 107k

CC

0.1µF–7.5V

1MHz

CD

RhD, 27.4k

VOUT

7.5V

R1B, 60.4k VIN

1INV B

2HP B

3BP B

4LP B

5SB

6GND

7V+

8SA

9LP A

10BP A

11HP A

12INV A

24INV C

23HP C

22BP C

21LP C

20SC

19V–

18FCLK

17SD

16LP D

15BP D

14HP A

13INV D

LTC1264

RlD, 100k R1A, 16.2k

Figure 1. LTC1264 linear-phase, 8th order bandpass filter

VS fCLK CC = CD

±7.5V 1.8MHz 3pF±7.5V 2.0MHz 5pF±5V 1.6MHz 5pF±5V 1.4MHz 3pF

Table 1. Capacitor selection guide

DESIGN IDEAS

1264_2.eps

FREQUENCY (kHz)10

–90

GAI

N (d

B)

10

100 200

–80

–70

–60

–50

–40

–30

–20

–10

0

Figure 4. Step response—Top: nonlinear-phase filterBottom: linear-phase filterFigure 2. Filter gain versus frequency Figure 3. Filter group delay versus frequency

1264_3.eps

FREQUENCY (kHz)46

50

DEL

AY (µ

s)

100

47 54

55

60

65

70

75

80

85

90

95

48 49 50 51 52 53


Book Review:Power Electronics—Circuits, Devices and Applicationsby Muhammad H. Rashid, Englewood Cliffs, New Jersey,Prentice Hall Publishing. Second Edition, 1993.

DESIGN INFORMATION

This is the first book review toappear in the pages of Linear Technol-ogy. As is our custom, we shall presentthe reader with tools that help him todo his job. We hope that by reviewingand recommending a few really use-ful books each year, we will help you,our customers, become even betterdesign engineers.

While such subjects as power andDC/DC converters are certainly wor-thy of life-long study, one has to startsomewhere. I am a novice in this area,but I chose to review this book inorder to bring an unbiased viewpointto the subject.

The book is quite a good funda-mental text on power electronics.Power electronics is the study andapplication of solid-state electronicsfor the control and conversion of elec-tric power. Rashid walks the readerthrough many of the basic to not-so-basic concepts inherent in the designof all types of power circuits, fromlarge “electric company” type circuitryto DC/DC converters. For example,Rashid carefully covers the impor-tant parameters of power transistors

in his chapter on these devices.I noted, in particular, his discussionson the storage-time constant and thesaturating charge in bipolar transis-tors. His treatment of the gate-driverequirements for MOSFETs will helpthe non-expert designer gain the ex-pertise required for complex switcherdesigns.

The text also includes chapters onDC/DC converters, Advanced Modu-lation Techniques, and ResonantConverters. There is a chapter onProtection of Devices and Circuitsthat should be required reading for alldesigners, whether of power suppliesor A-to-D converters.

Simulation of circuits, a topic oftengiven the “black magic” treatmentuntil graduate school or one’s firstjob, receives some ink in Rashid’stext. Rashid examines device modelsin PSPICE for diodes, thyristors,power transistors, and other activedevices as well as for circuitsranging from buck regulatorsto pulse-width-modulated inverters.These simulations are included toverify design examples and to

introduce the reader to SPICE.This is an excellent way to introducenot only the concept of modeling butthe idea that each SPICE model rep-resents a device or circuit composedof devices in the real world. Designersmust build circuits from these real-world devices, so the model mustmatch the device; though such mod-els are simplifications and may omitsome real-world complexities.

I would advise the reader to bread-board any circuits gleaned from thistext. Although it is quite a good intro-duction to power circuitry, no textcan bring home the problems en-countered when real circuits areswitching amperes in the vicinity ofnodes that must sense millivolts.

Rashid’s book is a good, concisetext that includes all the righttopics at a technical level that willbe useful for many power-supplydesigners. The book also is a goodintroduction to power electronicsfor would-be power supply experts.It deserves a look.

——RM

important. When the LT1204 is usedfor active video switching betweentwo flat-field video signals (a verycritical test) the switching artifactsare nearly invisible. When the LT1204is used to switch between two livevideo signals, the switching artifactsare invisible.

1 Video production, in the most general sense,means any purposeful manipulation of the videosignal, whether in a television studio or on adesktop PC.

only on expensive equipment. Wherecost is a determining factor in systemdesign, the exceptionally low ampli-tude and brief duration of theLT1204’s switching artifact make itan excellent choice for active videoswitching.

ConclusionActive video switching can be ac-

complished inexpensively and withexcellent results when care is takenwith both the selection and applica-tion of the high-speed multiplexer.Both fast switching and small, well-controlled switching glitches are

“Picture-in-Picture” continued from page 22


LT1161: Quad, Protected,High-Side MOSFET Driver

High-side switching in hostile en-vironments, such as industrialcontrol, avionics, and automotiveapplications, requires a ruggedizedN-channel MOSFET driver. TheLT1161 provides a full 100% opera-ting voltage margin in 24V systemsand can survive supply voltages from−15V to +60V (+75V on the gate)without damage.

Each of the four LT1161 switchchannels has a completely self-contained charge pump to fullyenhance an N-channel MOSFETswitch with no external components.Also included in each switch channelis a drain-sense comparator which isused to sense switch current.

The LT1161 independently pro-tects and restarts each MOSFET.Latch-off current limit can also beimplemented with the LT1161.

The LT1161 carries an industrialtemperature grade and is available ineither 20-lead plastic DIP or 20-leadSO surface mount packages.

The LTC1165 Triple, Inverting1.8V to 6V MOSFET Driver

The LTC1165 triple, inverting 1.8Vto 6V gate driver makes it possibleto switch either supply- or ground-referenced loads through low RDS(ON),N-channel switches from as little as1.8V (two discharged cells). The in-verting inputs make it possible todirectly replace P-channel MOSFETswitches while maintaining system-drive polarity. The LTC1165 containsthree on-chip charge pumps, sothat less expensive, lower RDS(ON)N-channel MOSFETs can be used toreplace high-side P-channel switches.

The three charge pumps have beendesigned to be very efficient and re-quire no external components. Thestandby current with the three in-puts switched off is typically 0.01µA.The quiescent current rises to 95µAper channel with the input turned onand the charge pump producing 11V

New Device Cameosfrom a 3.3V supply. The LTC1165 isavailable in both 8-pin DIP and 8-pinSOIC packaging.

LTC1143: Dual, 3.3V/5V High-Efficiency Step-Down Regulator

The LTC1143 is a dual-outputswitching-regulator controller in anarrow, 16-lead SOIC package thatminimizes board area. The LTC1143utilizes two independent current-mode regulator blocks to providesimultaneous 3.3V and 5V outputswith individual shutdown controls.

The LTC1143 extends battery lifeby providing high efficiencies at loadcurrents ranging from milliamps toamps. Both LTC1147-based regulatorblocks use current-mode architecturewith Burst ModeTM operation.

The LTC1143 is ideal for low-costapplications requiring 3.3V and 5Vsimultaneously, with high conver-sion efficiencies and the smallestcomponent count and board space.

LTC1142: Dual, 3.3V/5V Ultra-High-Efficiency SynchronousStep-Down Regulator

The LTC1142 is a dual-output,synchronous switching-regulatorcontroller in a compact, 28-lead SSOP.Two independent regulator blockssimultaneously provide 3.3V and 5Voutputs with individual shutdowncontrols.

Both LTC1148-based regulatorblocks use current-mode architecturewith Burst ModeTM operation toprovide extremely high operating effi-ciency, typically greater than 90%,over the entire load range. TheLTC1142 extends battery life by pro-viding high efficiencies at load currentsranging from milliamps to amps.

LT1203/LT1205: 170MHz Dualand Quad Video Multiplexers

The LT1203 is a wideband, two-input video multiplexer designed forsub-pixel switching and broadcast-quality routing. The LT1205 is afour-input, two-output multiplexer

NEW DEVICE CAMEOS

designed for multi-input expansion.These multiplexers act as SPDT videoswitches with 10ns transition times attoggle rates up to 30MHz. The −3dBbandwidth is 170MHz; gain flatnessis −0.1dB to greater than 60MHz.Many parts can be tied together attheir outputs by using a shutdownfeature that reduces the power dissi-pation and raises the outputimpedance to 5MΩ. Output capaci-tance in shutdown mode is only 1pFand the LT1203 peaks less than 2dBwith a 50pF load. Channel crosstalkand shutdown isolation are greaterthan 80dB at 10MHz. An on-chiplogic buffer interfaces to fast TTL orCMOS edges to reduce switching tran-sients to 50mV with a 20ns duration.The LT1203 and LT1205 outputs areprotected against shorts to ground.

The LT1203 is available in 8-leadPDIP and SO packages and the LT1205is available in the 16-lead SO package.

LTC1327, LTC1337, LTC1349,and LTC1350 Ultra-Low-PowerRS232 Transceivers

Four new RS562/RS232 trans-ceivers for notebook and palmtopcomputers feature the industry’s low-est power consumption. Eachtransceiver is a three-driver/five-receiver device that offers a completePC serial port solution for 3V or 5Vsystems and draws only 300µA qui-escent supply current.

On-chip charge pumps allow op-eration from standard logic supplies.The charge pumps require only fourspace-saving 0.1µF capacitors andcan supply up to 12mA of extra cur-rent for external circuitry. Thetransceivers operate up to 120k baudwith a 1000pF/3kΩ loads. The partsare pin compatible with the LT1137.

All transceivers feature LTC’s pro-prietary output structure, whichwithstands repeated ESD strikes of upto ±10kV using the human body model.

The LTC1327 is an RS562 trans-ceiver designed for 3V systems. In theSHUTDOWN mode, the supply cur-


rent drops to less than 1µA. TheLTC1337 is the RS232 version of theLTC1327—it operates from a 5Vsupply.

The LTC1349 is a 5V-supply RS232transceiver that features two low-power receivers that remain activewhile in the SHUTDOWN mode. Thetwo receivers together draw only 30µAof supply current. The LTC1350 isthe RS562, 3V-supply version of theLTC1349.

All four circuits are available in 28-pin DIP, SSOP, and SOIC packages.

Four New Products CombineLow-Power RS562 and RS485Transceivers On One Chip

Four new RS562/RS485 trans-ceivers each contain two interfaceports that can operate in either RS485mode, RS562 mode, or in a combina-tion of the two. A loopback modeperforms a diagnostic self test; ashutdown mode reduces the supplycurrent to 15µA. The typical unloadedsupply current is only 600µA innormal operation.

The RS562 transceivers operate to120k baud and are fully compliantwith RS562 overvoltage specifications.The RS485 transceivers operate to10M baud and are fully compliantwith all RS485 specifications. All driveroutputs feature short-circuit protec-tion and thermal shutdown. An enablepin allows the RS485 driver outputsto be forced into three-state operation,which is maintained when the out-puts are forced beyond the supplyrails or the power is off. Both receiverinputs and driver outputs feature±10kV ESD protection.

A +5V supply for VDD and VCC anda −5V supply for VEE are required forRS562 operation. However, if RS232voltage levels are required, VDD can beraised to +(6.5 to 10V) and VEE low-ered to −(6.5 to 10V).

The LTC1321 can be configured astwo RS562 transceivers, as one RS562transceiver and one RS485 trans-ceiver, or as two RS485 transceivers.The LTC1322 can configured as fourRS562 transceivers, as two RS562transceivers and one RS485 trans-

ceiver, or as two RS485 tranceivers.The LTC1334 and LTC1335 are pincompatible with the LTC1321 andLTC1322 respectively, except that VCCis internally connected to VDD and theunused pin becomes a receiver-output enable.

All four circuits are available in24-pin DIP and SOIC packages.

LT1413 Dual, Single-SupplyPrecision Op Amp

The LT1413 is an improved, low-cost version of Linear Technology’sindustry-standard LT1013 dual,single-supply op amp. The LT1413 isoptimized for single 5V applications:the input goes below ground and theoutput swings to ground while sink-ing current (no output pull-downresistors are needed). Phase-reversalprotection circuitry keeps the properphase at the output, even when theinput is significantly below ground.

The specifications achieved at sup-ply voltages of V+ = +5V, V− = 0V withthe low-cost plastic DIP and SO-8grades are 280µV maximum offsetvoltage (380µV maximum in the SO-8), 0.8nA maximum offset current,1.4 million voltage gain, 0.5µV/°Cdrift, 950kHz gain-bandwidth prod-uct, 0.55µV peak-to-peak noise from0.1Hz to 10Hz, and 140dB channelseparation. The output delivers inexcess of 10mA load current (sourc-ing or sinking), even though the sup-ply current per amplifier is only 330µVquiescent. A full set of specifications isalso provided on the LT1413 at±15V supplies.

LTC in the News . . .

Fiscal 1993 Again a RecordOn July 20, 1993, Linear Tech-

nology Corporation announcedthat net sales for its fiscal yearended June 27, 1993 were a record$150,867,000, an increase of 26%over fiscal 1992. The company alsoreported record net income for theyear of $36,435,000 or $0.99 pershare.

Net sales for the fourth quarterof 1993 were a record $42,813,000,a 31% increase over the fourthquarter of the previous year. Netincome for the quarter was$10,838,000 or $0.29 per share,an increase of 49% over the fourthquarter of the previous year. Acash dividend of $0.05 per sharewill be paid on August 18, 1993 toLTC shareholders of record onAugust 2, 1993.

According to Robert H.Swanson, President and CEO,“1993 was another excellent yearfor us. Bookings, revenues, profits,and cash generated from the busi-ness all grew handsomely. Weintroduced a record number ofnew products, brought up asecond fabrication line, and com-menced construction on our firstoffshore plant in Singapore. Thisbalance of financial performance,product support and introduction,and sales penetration continues tobe our major strategic focus.”

More Rankings and RatingsRecognition of Linear Tech-

nology's performance continues.California Business ranked LTC as43rd in its Top 500 companies inCalifornia, and also ranked LTCamong its Top 50 in Technology.

The Los Angeles Times includedLinear Technology in its Top 100Companies in California. Locally, LTCwas included in the San FranciscoChronicle’s and the San FranciscoExaminer’s Top 100 Bay Area Com-panies, and in the San Jose MercuryNews Silicon Valley Top 150.

NEW DEVICE CAMEOS

For further information on theabove or any other devicesmentioned in this issue of LinearTechnology, use the reader servicecard or call the LTC literature-service number: 1-800-4-LINEAR.Ask for the pertinent data sheetsand application notes.

Information furnished by Linear Technology Cor-poration is believed to be accurate and reliable.However, no responsibility is assumed for its use.Linear Technology makes no representation thatthe circuits described herein will not infringe onexisting patent rights.

Burst ModeTM is a trademark of Linear TechnologyCorporation.


DESIGN TOOLSApplications on DiskNOISE DISKThis IBM-PC (or compatible) progam allows the user tocalculate circuit noise using LTC op amps, determine thebest LTC op amp for a low noise application, display thenoise data for LTC op amps, calculate resistor noise, andcalculate noise using specs for any op amp.Available at no charge.

SPICE MACROMODEL DISKThis IBM-PC (or compatible) high density diskette containsthe library of LTC op amp SPICE macromodels. Themodels can be used with any version of SPICE for generalanalog circuit simulations. The diskette also contains work-ing circuit examples using the models, and a demonstrationcopy of PSPICETM by MicroSim.Available at no charge.

Technical Books1990 Linear Databook — This 1440 page collectionof data sheets covers op amps, voltage regulators,references, comparators, filters, PWMs, data conversionand interface products (bipolar and CMOS), in both com-mercial and military grades. The catalog features well over300 devices. $10.00

1992 Linear Databook Supplement — This 1248 pagesupplement to the 1990 Linear Databook is a collection ofall products introduced since then. The catalog contains fulldata sheets for over 140 devices. The 1992 Linear DatabookSupplement is a companion to the 1990 Linear Databook,which should not be discarded. $10.00

Linear Applications Handbook — 928 pages full ofapplication ideas covered in depth by 40 Application Notesand 33 Design Notes. This catalog covers a broad range of“real world” linear circuitry. In addition to detailed, systems-oriented circuits, this handbook contains broad tutorialcontent together with liberal use of schematics and scopephotography. A special feature in this edition includes a 22page section on SPICE macromodels. $20.00

1993 Linear Applications Handbook Volume II —Continues the stream of “real world” linear circuitry initiatedby the 1990 Handbook. Similar in scope to the 1990 edition,the new book covers Application Notes 41 through 54 andDesign Notes 33 through 69. Additionally, references andarticles from non-LTC publications that we have founduseful are also included. $20.00

Interface Product Handbook — This 200 page handbookfeatures LTC’s complete line of line driver and receiverproducts for RS232, RS485, RS423, RS422 and AppleTalk

applications. Linear’s particular expertise in this area in-volves low power consumption, high numbers of driversand receivers in one package, 10kV ESD protection ofRS232 devices and surface mount packages.Available at no charge.

Monolithic Filter Handbook — This 232 page book comeswith a disk which runs on PCs. Together, the book and diskassist in the selection, design and implementation of theright switched capacitor filter circuit. The disk containsstandard filter responses as well as a custom mode. Thehandbook contains over 20 data sheets, Design Notes andApplication Notes. $40.00

SwitcherCAD Handbook — This 144 page manual, in-cluding disk, guides the user through SwitcherCAD—apowerful PC software tool which aids in the design andoptimization of switching regulators. The program can cutdays off the design cycle by selecting topologies, calculat-ing operating points and specifying component values andmanufacturer's part numbers. $20.00

Linear Technology Corporation1630 McCarthy BoulevardMilpitas, CA 95035-7487Phone: (408) 432-1900FAX: (408) 434-0507

AppleTalk is a registered trademark of Apple Computer, Inc.

LINEAR TECHNOLOGY CORPORATION1630 McCarthy BoulevardMilpitas, CA 95035-7487

(408) 432-1900Literature Department 1-800-4-LINEAR

FRANCELinear Technology S.A.R.L.Immeuble “Le Quartz”58 Chemin de la Justice92290 Chatenay MallabryFrancePhone: 33-1-46316161FAX: 33-1-46314613

GERMANYLinear Technology GMBHUntere Hauptstr. 9D-8057 EchingGermanyPhone: 49-89-3194710FAX: 49-89-3194821

JAPANLinear Technology KK5F YZ Building4-4-12 Iidabashi Chiyoda-KuTokyo, 102 JapanPhone: 81-3-3237-7891FAX: 81-3-3237-8010

KOREALinear Technology Korea BranchNamsong Building, #505Itaewon-Dong 260-199Yongsan-Ku, SeoulKoreaPhone: 82-2-792-1617FAX: 82-2-792-1619

SINGAPORELinear Technology Pte. Ltd.101 Boon Keng Road#02-15 Kallang Ind. EstatesSingapore 1233Phone: 65-293-5322FAX: 65-292-0398

TAIWANLinear Technology CorporationRm. 801, No. 46, Sec. 2Chung Shan N. Rd.Taipei, Taiwan, R.O.C.Phone: 886-2-521-7575FAX: 886-2-562-2285

UNITED KINGDOMLinear Technology (UK) Ltd.The Coliseum, Riverside WayCamberley, Surrey GU15 3YLUnited KingdomPhone: 011-44-276-677676FAX: 011-44-276-64851

InternationalSales Offices

© 1993 Linear Technology Corporation/ Printed in U.S.A./20K

World Headquarters

U.S. AreaSales OfficesCENTRAL REGIONLinear Technology CorporationChesapeake Square229 Mitchell Court, Suite A-25Addison, IL 60101Phone: (708) 620-6910FAX: (708) 620-6977

NORTHEAST REGIONLinear Technology CorporationOne Oxford Valley2300 E. Lincoln Hwy., Suite 306Langhorne, PA 19047Phone: (215) 757-8578FAX: (215) 757-5631

NORTHWEST REGIONLinear Technology Corporation782 Sycamore Dr.Milpitas, CA 95035Phone: (408) 428-2050FAX: (408) 432-6331

SOUTHEAST REGIONLinear Technology Corporation17060 Dallas ParkwaySuite 208Dallas, TX 75248Phone: (214) 733-3071FAX: (214) 380-5138

SOUTHWEST REGIONLinear Technology Corporation22141 Ventura BoulevardSuite 206Woodland Hills, CA 91364Phone: (818) 703-0835FAX: (818) 703-0517

Documents

LINEAR TECHNOLOGY LINEAR TECHNOLOGY · 2018. 3. 20. · output-buffer amplifier, a 2.048V voltage reference, and an easy-to-use, cascadable three-wire serial inter-face. The single