LETTER A compact multi-mode multi- matching networks for

A compact multi-mode multi-band power amplifier withharmonic-suppressionmatching networks for GSM/TD-SCDMA/LTE terminals

Jun-ming Lin1, Gary Zhang1a), Zhi-hao Zhang1, Yao-hua Zheng1,Jia-jin Li1, Li-xiang Ou1, and Zu-hua Liu1,21 School of Information Engineering, Guangdong University of Technology,

No. 100 Waihuan XiRd, Guangzhou 510006, People’s Republic of China2 Foshan Innovative Micro-IC Technology Inc.,

The HTZK Center of Science & Technology, 528000, People’s Republic of China

a) [email protected]

Abstract: This paper presents a fully-integrated and compact multi-band

and multi-mode power amplifier for commercial quad-band GSM, dual-band

TD-SCDMA and mono-band LTE cellular terminals. To reach a compromise

among the performance in terms of efficiency, harmonic suppression and

miniaturization, a Class-E PA and a linear PA cores are employed with

harmonic-trap immersed output matching networks for low and high band,

respectively. Experimental results suggest that the PA generated an output

power higher than 33 and 30.5 dBm for low and high band, respectively,

with considerable PAE across the operating bands when applied CW signals,

as well as favorable ACPR performance for GMSK/EDGE, QPSK TD-

SCDMA and 10MHz QPSK TD-LTE signals.

Keywords: power amplifier, multi-mode, multi-band, quad-band GSM,

dual-band TD-SCDMA, LTE

Classification: Integrated circuits

References

[1] T. Furuta, et al.: “Compact 1.5GHz to 2.5GHz multi-band multi-mode poweramplifier,” IEICE Electron. Express 8 (2011) 854 (DOI: 10.1587/elex.8.854).

[2] J. C. Clifton, et al.: “Wideband high efficiency multi-band, multi-mode (LTE/WCDMA/GSM) power amplifier for mobile terminals,” 2013 EuropeanMicrowave Conference (2013) 1495.

[3] S. Kang, et al.: “A multi-mode multi-band reconfigurable power amplifier forlow band GSM/UMTS handset applications,” 2013 Power Amplifiers forWireless and Radio Applications (2013) (DOI: 10.1109/PAWR.2013.6490174).

[4] C. Zhang and A. E. Fathy: “A novel reconfigurable power amplifier structurefor multi-band and multi-mode portable wireless applications using areconfigurable die and a switchable output matching network,” 2009 IEEEMTT-S International Microwave Symposium Digest (2009) (DOI: 10.1109/MWSYM.2009.5165846).

© IEICE 2016DOI: 10.1587/elex.13.20161033Received October 21, 2016Accepted October 31, 2016Publicized November 28, 2016Copyedited December 25, 2016

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LETTER IEICE Electronics Express, Vol.13, No.24, 1–12

http://dx.doi.org/10.1587/elex.8.854




http://dx.doi.org/10.1109/PAWR.2013.6490174




http://dx.doi.org/10.1109/MWSYM.2009.5165846





[5] J. Lin, et al.: “A multi-mode multi-band power amplifier for qua-band GSM,dual-band TD-SCDMA, and TDD LTE band 39 cellular applications,” 2015Asia-Pacific Microwave Conference (2015) (DOI: 10.1109/APMC.2015.7411817).

[6] H. Motoyama, et al.: “Stacked FET structure for multi-band mobile terminalpower amplifier module,” 2013 IEEE Microwave Symposium Digest (IMS)(2013) (DOI: 10.1109/MWSYM.2013.6697530).

[7] S. C. Cripps: RF Power Amplifier for Wireless Communications (Artech HousePublishers, 2006).

[8] K. Narendra and T. YewKok: “A design technique to improve harmonicsuppression in high efficiency wideband Class E RF power amplifier,” IEICEElectron. Express 11 (2014) 20130824 (DOI: 10.1587/elex.11.20130824).

[9] N. O. Sokal and A. D. Sokal: “Class E—a new class of high efficiency tunedsingle-ended switching power amplifiers,” IEEE J. Solid-State Circuits 10(1975) 168 (DOI: 10.1109/JSSC.1975.1050582).

[10] Z. Zhang, et al.: “A novel body self-biased technique for enhanced RFperformance of a SP8T antenna switch in partially depleted CMOS-SOItechnology,” 2014 IEEE Solid-State and Integrated Circuit Technology(ICSICT) (2014) (DOI: 10.1109/ICSICT.2014.7021253).

1 Introduction

Explosive-growth demand for information, inevitably motivate modern wireless

communication systems to upgrade the existing obsolete devices by employing

more complicated technologies (3G/3.9G/4G) with spectral efficient modulation

schemes to meet the strict requirements of large data service, but also using many

of given limited spectrum fragmentations for multi-use applications, which de-

mands stringent abilities of multi-band and multi-mode (MMMB) [1, 2, 3].

Although there is no need to design a ubiquitous-coverage phone to cover excess

frequency bands due to the minimum worldwide multi-mode requirements (2/2.5/

3/4G) and region frequency multiplexing techniques, this increased complexity

becomes a critical issue, especially for integrated RF front ends [4]. Moreover, PA

design meets an inaccessible compromise because of the complicated combination

of envelope-varying modulation schemes and multiple access techniques [5]. Over

the years, many researchers have tried fulfilling strict MMMB specifications with

single core by means of switchable or broadband matching schemes to provide

matching optimum for specified bands [4, 6]. Nonetheless, data is still scare rear

and further studies are still essential to those mixed-mode applications, thus that

separating PAs respectively dedicated to low and high band, is a distinct possibility

to commercial applications among the reported works.

In this work, a miniaturized MMMB PA is elaborated for customized terminals

with two separated PA cores to cover the frequency band 824 to 915MHz (LB) and

1710 to 2025MHz (HB). The proposed two cores are fabricated together by an

InGap/GaAs HBT technology with off-chip harmonic-suppression matching net-

works. Finally, the MMMB PA is assembled with a CMOS controller and a SP8T

radio frequency (RF) switch on a microwave laminate for experimental evaluation.© IEICE 2016DOI: 10.1587/elex.13.20161033Received October 21, 2016Accepted October 31, 2016Publicized November 28, 2016Copyedited December 25, 2016

2

IEICE Electronics Express, Vol.13, No.24, 1–12

http://dx.doi.org/10.1109/APMC.2015.7411817












http://dx.doi.org/10.1109/JSSC.1975.1050582




http://dx.doi.org/10.1109/ICSICT.2014.7021253




2 A compact structure of RF front ends

Advanced multifunctional cellular phones should support at least four operation

modes and seven frequency bands. In China, China Mobile Communication

Corporation (CMCC) employs Time Division Duplexing (TDD) multiple access

techniques for either 3G (TD-SCDMA) or 4G (TD-LTE) systems. Directed at the

requirements of CMCC, a novel custom-built structure of RF front ends is

presented and ameliorated in accordance with proposal improvements that du-

plexers can be replaced by radio frequency switches basing on the recognized

standard of TDD techniques that either transmitting or receiving operation is not

simultaneous. The compact scenario is shown in Fig. 1 and contains a MMMB

PA to cover all TX paths with two separated cores, two low pass filter to filter out

the out-band leakage, eight band pass filters (SAW), and a SP8T switch. Specially,

the GSM 1800 and TD-SCDMA Band 34 are integrated together to share a

common receiver path. Both factors have made it possible to help with the

complexity and cost reduction.

3 Circuit design of proposed MMMB PA

As depicted in Fig. 2, the graphical MMMB PA module is made up of two PA

cores, a commercial CMOS controller to control the bias condition (off/on), and a

RF switch to choose the internal flow orientation of signals. By means of switch-

able and broadband techniques, the PA is designed to cover seven frequency bands

which are grouped into two different classifications, respectively named LB

(DCS1800/PCS1900) and HB (DCS1800/PCS1900, TD-SCDMA bands 34/39,

and TDD LTE band 39).

To enhance linearity performance, a load-line broadband technique [7] was

directly improved by replacing the mentioned user-switchable or electricity-tunable

capacitors with LC tanks to suppress higher harmonics. Considering that only two

low GSM bands are allocated to LB PA, high efficiency scheme, class-E PA, is

applied to take full advantage of the large parasitic capacitance at the output of

LB PA core. To enhance the drain efficiency, adding capacitor is shunted out of chip

to tune the optimum for Class-E operation to short higher harmonics effectively for

Fig. 1. The schematic block diagram of the front end for a quad-bandGSM, dual-band TD-SCDMA and mono-band LTE cellularphone.


3


LB. The characteristics of a Class E PA can be determined by finding its steady-

state drain voltage and current waveforms with the help of simulation platforms

[7, 8]. To keep the peak voltage below a certain level referred to the DC supply, and

considering the trade-off between PUF and efficiency, a mediate Class-AB bias

condition is set by choosing the conduction angle of the LB PA to be around 1:35�.

For HB operation, since LTE and TD-SCDMA lead to a variable envelope,

necessitating a linear PA, the HB PA cores is design with linear PA architecture

with several efficiency enhanced techniques. The HB PA is biased at a deep

Class AB mode that even lower than that of the LB PA. The gain stage is biased

to a quiescent point beyond the light Class AB condition for high linearity, and

respectively Class AB for the drive stage with a half-wave rectified output voltage,

so that high gain and high efficiency performance can be obtained from the power

stage [7]. To achieve favorable harmonic suppression performance, a second

harmonic trap is shunted at the output of the HB PA core.

4 Design considerations of class-E operation

To the best of our knowledge, although class-E PA is a highly nonlinear device,

it has the highest efficiency among class A, AB, B and F, essentially the same

saturated output power. In essence, the presence of significant drain capacitance of

high power device makes class E useful in low-frequency applications.

In this paper, we present a creative matching network to suppress the out-band

harmonic products without excessively deteriorate the performance of Class E

which is approximated by providing the drain with the fundamental frequency

impedance and preferably one or more of the harmonic impedances.

The graphical class-E implementation of LB PA is straightforward illustrated

in Fig. 3. It employs a cumbersome transistor operated as a switch, resulting in a

shaped drain-current waveform which is the linear superposition of the DC and RF

currents charging the drain-shunt capacitance. Basically, for a given bandwidth,

an optimum class E delivers a full output power of Pout to an optimized load Ropt in

the presence of a drain shunt susceptance Y and a drain series reactance B. The

approximate solutions of Y and B can be quoted from [9] as:

Fig. 2. Schematics block diagram of the proposed MMMB PA module.


4


Y ¼ Qnet � 0:66

QnetðQnet � 2:08ÞRopt; B ¼ ðQnet � 5:447ÞRopt; and Pout ¼ 0:577

Vdd2

Roptð1Þ

Where Qnet is the quality factor of the output network and must be smaller than that

typically required by the carrier bandwidth, and Vdd is the drain bias voltage.

5 Designs of output matching networks

For broadband application, each output path of PAs is linked with several broad-

band matching sections (low-pass filters) which are immersed with harmonic traps

or blocks to prevent the higher splatter swarming into adjacent channels. The HB

PA to meet the requirements of high PAPR signals (LTE/TD-SCDMA) is linear

without qualification that the load should be optimized and converted directly from

50Ohm, by considering the circuit shown in Fig. 4(a).

The output matching network (OMN) of HB PA consists of four second

harmonic traps, a third parallel-connected harmonic trap intending to suppress

second and third harmonic due to their significant impact to linearity and efficiency

of PAs. The equivalent circuit of Fig. 4(a) at fundamental frequency is shown in

Fig. 4(b). The series-connection traps in the shunt are equivalent to capacitance at

fundamental frequency, whereas the parallel-connection trap in the series is treated

as inductance, and the inductance L11 can be divided into two series-connection

inductance L1 and L2.

Fig. 3. Basic model of class E stage combing a drain shunt susceptanceY and a drain series reactance B.

Fig. 4. (a) The OMN with 2nd short trap for HB operation, (b) theequivalent circuit of (a) at fundamental frequency, (c) the OMNwith capacitance-tuned trap for LB operation, (d) the equivalentcircuit of (c) at fundamental frequency.


5


The matching strategy corresponding to such complex matching network is

ideally and proportionally depicted in Fig. 5(a) for intuitionistic analysis. The

output matching network contains three L-C sections, indicating that the matching

network is highly broadband. However, the first section (C1-L1) is opposite to the

others, thus that the overall bandwidth is deteriorated and become slightly smaller,

but good harmonic suppression performance can be obtained by means of shunt

capacitor, C1.

The point B plotted in Fig. 5(a) refers to the optimized impedance (Ropt) of the

output matching network. The point A at the left of point B is a random point and

has a big impact on the overall bandwidth of output matching network. Skipping

the tedious derivation of algebra [7], the minimal quality factor of Fig. 4(a), and

the resistance in the equivalent circuit shown in Fig. 4(b) can be calculated by the

followed expressions:

Qmin ¼ffiffiffiffiffiffiffiffiffiffiffik � 1

p; k ¼

ffiffiffiffiffiffiRL

R1

3

r; R3 ¼ RL

k; R2 ¼ RL

k2and Qp ¼

ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiRopt

R1

� 1

rð2Þ

Where RL and R1 are the resistance at point B and point A, respectively, k is

transformation ratio, Qp is the minimum quality factor of C1-L1 section. After a

series of algebra, all components can be derived as:

CF1 ¼ 8

9� Qmin

!RL; LF1 ¼ 1

9!2CF1; LD1 ¼ 3

4� QminRL

!ð1 þ Qmin2Þ ; CD1 ¼ 1

4!2LD1ð3Þ

CB1 ¼ 3

4� Qmink

2

!RL; LB1 ¼ 1

4!2CB1; CA1 ¼ 3

4� Qp

!ð1 þ Qp2ÞR1

; LA1 ¼ 1

4!2CA1ð4Þ

L11 ¼ QpR1

!þ QminRL

!k2ð1 þ Qmin2Þ ; L3 ¼ QminRL

!kð1 þ Qmin2Þ ð5Þ

For LB operation, the output matching network is rather multifunctional because of

the expectation of a shunt susceptance Yand a drain series reactance B for optimum

class-E PA. The OMN of LB is similar with HB, but the drain-shunt network (LA1

(a) (b)

Fig. 5. (a) Matching topology to give improved performance betweenbandwidth and harmonic suppression for HB applications;(b) Matching topology to give improved performance betweenbandwidth and harmonic suppression for LB applications.


6


and LC1) is used for capacitance tuning and the transferred point A is coincidence

with point B, as shown in Fig. 4(c) and Fig. 5(b). The equivalent circuit of

Fig. 4(c) at fundamental frequency is shown in Fig. 4(d). With the addition margin

for insertion loss of matching, the LB PA is designed with the target output power

of 35 dBm resulting in an optimized Ropt ¼ 2:24Ω.

Since the quality factor of the three LC matching sections is definitely small

(about 1.35), the bandwidth margin is still plentiful and the overall quality factor

Qnet ¼ 6:5 is selected and used for estimating the drain impedance of LB PA,

resulting in a bandwidth 134MHz, a drain impedance of (3:41 þ j2:12) Ohm at the

fundamental frequency, �j11:29 at the second harmonic, and proportionately

smaller capacitive reactance at higher harmonics. Furthermore, due to the large

peripheral of power stage of LB PA, the drain capacitance is too significant to be

neglect, thus that the CAD simulation platform ADS is used to characterize and a

capacitance of about 3.2 pF is obtained with base shorted.

The simulated input reflection coefficient of the output matching network of

HB and LB with the load RL ¼ 50Ω is plotted in Fig. 6(a) and (b), respectively.

The optimized values are illustrated in Table I. The temporary resistance R1 for HB

(2Ω) and LB (2.24Ω), is selected respectively. To HB, although the overall

bandwidth is decreased approximately by 11%, the impedance maintains constant

around 3Ω across 1.88 to 1.92GHz, and near zero across 3.76 to 3.84GHz.

The comparisons of forward transmission coefficients among the kinds of

matching topotype, including a simple and multiple lower-order networks, and

our works, are depicted in Fig. 7. In this case, the quality factor for one (low-order),

two (low-order) and multiple sections (our works) with load 50Ω are 3.96, 1.76 and

1.35, respectively, which indicates an improvement of about 193% and 30.4%

compared to multiple-section OMN. The fairly and conclusive summary among

above matching topotypes is attractive for scientific and engineering applications

that the unwished second and third harmonics are blocked by the proposed multiple

matching networks without overly degrade the bandwidth characteristics, resulting

in two minimum points, and less power is dissipated at higher harmonic products.

The second harmonic is obviously suppressed into a deep level compared against

the third harmonic. As a preliminary consideration, it is not available to design firth

or higher harmonic traps, because the desired inductance is dramatically small

which makes the design more challenge. The simulated results of charging current

for transistor and shunt capacitor C1 are illustrated in Fig. 7(b), respectively. The

Table I. Optimized values of output matching network of HB/LB PA

ModeFre. LA1 LB1 LC1 LD1 LF1 L11

(MHz) (nH) (nH) (nH) (nH) (nH) (nH)

0.118/ 0.118/ 0.344/ 1.490/ 0.377/ 0.351/0.5 0.29 0.8 3.29 0.85 1.05

HB/LB 1890/870CA1 CB1 CC1 CD1 CF1 L3

(pF) (pF) (pF) (pF) (pF) (nH)

14.81/ 14.90/ 5.096/ 1.177/ 2.066/ 0.679/8.0 29.31 10.41 2.55 4.38 1.55


7


transient current would charge into shunt capacitor C1 (as well as the parasitic

capacitance of transistor) with transistor off. On the contrary, the overall current

would flow into the transistor with the statue on. The above action leads to an

orthogonal overlap between transient current and voltage, thus that less power is

dissipated by the transistor and high efficiency is obtained.

6 Fabrication and measurement results

The front end is designed to meet the needs of both LB and HB with a target output

power of 33 and 31 dBm, respectively. Fig. 8(a) is a die photograph of the MMMB

PAwhich is fabricated by an InGap/GaAs HBT technology. The size of the chip is

(a) (b)

Fig. 6. (a) The simulated results of input reflection coefficient of theoutput matching network for HB operation with optimizedRopt ¼ 3Ω and temporary impedance R1 ¼ 2Ω across 1 to10GHz; (b) The simulated results of input reflection coefficientof the output matching network for LB operation withoptimized Ropt ¼ 2:24Ω and network quality factor Qnet ¼ 6:5.

(a) (b)

Fig. 7. (a) The simulated results of forward transmission of the outputmatching network for LB and HB operations with optimizedRopt ¼ 3Ω and temporary impedance R1 ¼ 2Ω across 10MHzto 10GHz. (b) Voltage and current waveforms for simulatedcircuit shown in Fig. 2 with the optimum LB output matchingnetwork shown in Fig. 4(c) with a target output power of35.1 dBm.


8


2:1 � 1:1 � 0:82mm3 including all bonding pads. The area of LB PA core is

approximately twice as big as HB.

All elements of the MMMB module, such as PA die, commercial CMOS

controller and RF SP8T antenna switch [10], are assembled in a four-layer laminate

(5mm � 5mm) on which the output matching networks is fabricated with trans-

mission line for tunability, as shown in Fig. 8(b). Most of the inductors are realized

by either bonding wires or transmission lines, since the performance of RF PAs is

primarily determined by the quality of the passive used and high quality inductor

are certainly difficult to realize in chip.

The module is tested under a number of prospective frequencies (824, 880, 915

and 1710, 1810, 1910MHz). More detailly, the MMMB PA is biased with a drain

voltage of 3.5V and a bias reference voltage of 2.9V, under which the quiescent

drain current is approximately 630mA (LB) and 510mA (HB). The measured

reflection coefficient (S11) and forward transmission parameters (S21) are illustrated

in Fig. 9. Each S21 curves has a sunken point at around second frequency, and

higher broadband performance is obtained at HB at the expense of forward signal

gain.

(a) (b)

Fig. 8. Physical configurations for test, (a) Die photograph of theMMMB PA; (b) Photograph of the laminate for proposedMMMB PA including a commercial CMOS controller and aSP8T switch.

(a) (b)

Fig. 9. Measured Scatter parameters of MMMB module: (a) reflectioncoefficient (S11); (b) Forward transmission parameters (S21).


9


The measured results of max power gain and PAE are shown in Fig. 10, the

proposed PA module generated a PAE higher than 30% for both bands meeting

max output power of higher than 33 and 30.5 dBm for LB and HB, respectively.

Due to the lower frequency-parasitic component, better PAE performance is

obtained at LB compared to HB.

Fig. 11 shows how power gain, output power and PAE (Power Added Effi-

ciency) vary with CW input signals. When LB operations are active, the HB PA

core will be shut down by the CMOS controller, and vice versa. The noticeable

trend of power gain is similar with each other with maximum output nearly

33.5 dBm. In contrast to LB, the power gain of HB at low input power region is

slightly bumpy, which indicates that the HB PA is not broadband enough to cover

all the frequencies due to the reluctant matching topologies with high-frequency

parasitic components, and further improvements are needed. Based on Fig. 11(b),

the specification of PAE among modes can be evaluated. Between 824 and

2010MHz, a reduction of 19% with maximum output power and 30% in 6 dB

back-off region will be achieved. This means that the cellular phone will meet

a maximum increment of power consumption of 49% while transfer the mode to

4G from 2G. To indicate additionally, some auxiliary techniques can be used to

ameliorate the efficiency performance, such as envelope-tracking techniques.

Fig. 10. Measured result of Pmax and PAE at Pmax for LB and HB.

(a) (b)

Fig. 11. Measured result of proposed MMMB PA for LB and HB:(a) Power gain and output power versus Pin; (b) PAE versusPin.


10


The key test results of harmonic and linearity are summarized in Table II.

Benefited from the second- and third-harmonic traps, the PA suppresses the second

harmonic at the maximum degree and shows harmonic suppression of better than

64 dBc across all operation bands.

The last set of data summarized in Table II also shows that the linearity of

EDGE (25% duty cycle) is significantly considerable in accordance with relative

frequency offsets. When measured at �200, �400 and �600KHz offsets, the PA

generates an ACPR of better than −34.5 and −35.0 dBc, −57.4 and −61.4 dBc and−71.2 and −74.9 dBc with a maximum linear output power 27.2 and 26 dBm for

LB and HB, respectively. When tested with TD-SCDMA signals at 1.6 and

3.2MHz offsets, the PA module shows an ACPR of −50 and −61 with an EVM

of 4% and a linear output power of 26.5 dBm. Finally, the PA module is imbued

with TD-LTE source (10MHz 50 RB QPSK signals), and generates an

E_UTRA_ACPR of −40 dBc and UTRA_ACPR of −43 dBc compared to commer-

cial specifications.

For multi-mode and multi-band applications, many works had been reported, to

demonstrate the feasibility of our work, the comparisons with previous work are

summarized in Table III. This paper focus only on the technologies to miniaturize

a MMMB PA module. Some limitations of this study are obviously emerged into

the PAE of 3G/4G when compared to switchable designs. Notwithstanding its

limitation, this work does realize a MMMB PA module with considerable size.

Table II. Measured result of harmonic and ACPR for LB and HB

Fre. Pout Har. (dBc)ACPR1 (dBc) & Fre. Offset

EVMMode

(MHz) (dBm)(KHz)

2nd 3rd 200 400 600

824 33.5 −69.1 −70.9 −35.3 −61.1 −71.7

LB 880 33.3 −66.8 −70.5 −34.7 −58.7 −71.8915 33.1 −65.3 −71.2 −34.6 −57.5 −71.2 5%

1710 31.5 −64 −70.6 −35.7 −61.5 −78.5

HB 1810 31.3 −63.5 −67.3 −35.2 −61.7 −76.21910 31.0 −67.2 −62.2 −35 −61.7 −74.9

TD-SCDMA21910 26.5 - - −50 −61 - 4%

2010 24.5 - - −44 −62 - 4%

TD-LTE3 1910 25.0 - - −40 −43 - 5%1ACPR measured with 1.6 and 3.2MHz offset at 26.5 and 24.5 dBm for TD-SCMDA;2Measured using ETSI TS 125.102 UL reference measurement channel (12.2 kbps), 16%duty cycle;3Measured using ETSI TS 136.101 UL reference measurement channel, 10MHz, QPSK,50RB per channel.


11


7 Conclusion

In summary, this paper concerns the developments of a single-chip MMMB module

and investigates the harmonic-suppressed broadband matching networks to design

a compact MMMB PA for quad-band GSM, dual-band TD-SCDMA, and mono-

band TD-LTE commercial applications using two resizable PA cores. The PA

was fabricated by an InGap/GaAs HBT technology with the aim of scaling the

characteristics between LB and HB. The experimental investigations reveal that the

propose PA module generates sufficient output power with considerable compro-

mise between PAE and ACPR.

Acknowledgments

This work was supported by National Natural Science Foundation of China (Grant

61574049 and 61404032), Leading-Talent Specific Foundation of Guangdong

Province, China (Grant 400130002). Furthermore, the authors wish to acknowledge

the assistance and support of all the other staff of Guangzhou Junheng Micro-

electronics Technology Co. Ltd. and Foshan Zhenzhi microchip technology Co.

Ltd., China.

Table III. Comparisons with previous works

Ref. [2] [3] [4] SKY77910 This work

Mode 2/3/4 2/2.5/3/4 2/3 2/2.5/3/4 2/2.5/3/4

Band HB/LB HB/LB HB/LB HB/LB HB/LB

Frequency 1710–2050/ 1710–1980/1800/800

1710–2025/ 1710–2025/(MHz) 698–915 824–915 824–915 824–915

PAE2G 39.5–42.8 50.2–53.4 48 35–40 33–35

(%)3G 44 40.1–41.1 - 15 17

4G 32 34.3–35.4 - 13 15

No. of Core 2 2 3 2 2

No. of Path 2 5 2 2 2

Size (mm2) 5 � 7 5 � 7 - 5:5 � 5:3 5:0 � 5:0

NotePHEMT,Switchablematching.

SwitchablePA

ReconfigurableDie,

Switchablematching.

HBT,Broadbandmatching.

HBT,Broadbandsuppression.


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Documents

LETTER A compact multi-mode multi- matching networks for