Jan-Geralt Bij de Vaate ASTRON - Onderzoekpeletier/RFsystemen2.pdf2 Basic Detection Techniques – “RF systems” JGBdV 2006 Content: • Introduction RF design • Basic principles

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Basic Detection Techniques – “RF systems”

JGBdV 2006

RF SYSTEMS

Jan-Geralt Bij de Vaate

ASTRON

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JGBdV 2006

Content:

• Introduction RF design• Basic principles

– Non linearity– Noise– Sensitivity– Dynamic range

• RF – building blocks– PLL– Oscillators– Mixers– Amplifiers– De-modulators– Filters

• Examples– Receiver architectures– Transceiver– WSRT– LOFAR

• Practicum– Use of spectrum analyzer

• Gain• IP2/IP3• Noise

– Use of noise meter

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Introduction RF designRF design is still a design bottleneck:• Multiple-disciplines

– Communication theory– Signal processing– IC technology– Etc.

• RF design hexagon

Trade off :

NoiseDC Voltage

Frequency Amplification

Power supplyLinearity

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• Design tools

– Reasonable circuit simulators • Accuracy depends on model quality

– Good Electro Magnetic simulators • 2.5 D of 3D• (computer) Time consuming

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Element Duplexer LNA BPF Mixer AMP #1 BPF AMP #2 AMP #3 Mixer BPF AMPGain (dB) -3 20 -2.7 -7 20 -17.4 30 30 -6 -6 45Noise Figure (dB) 2.7 1.5 2.7 7.5 4 17.4 4 6 6 6 5NFO (dB) 2.7 4.2 4.3 4.7 5.1 5.3 5.6 5.6 5.6 5.6 5.6Power (dBm) -125.3 -103.9 -107 -114.4 -94.1 -111.5 -81.5 -53.2 -58.9 -65 -19.9SNR (dB) 6.5 4.9 4.9 4.5 4 3.9 3.6 3.6 3.6 3.6 3.6TOI @ Output (dBm) 3.2 0.5 -7.5 8 -9.4 9.1 24.2 15.8 9.8 20

EXAMPLE RF DESIGN

Budget analyses

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Basic principles RF design

• Non linearity– Harmonic distortion– Gain compression– Blocking– Cross modulation– Inter-modulation (IP2/IP3)

• Random processes noise– Noise figure– Noise figure in cascaded circuits

• Sensitivity• Dynamic range

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Non linearity

...)()()()( 33

221 +++≈ txtxtxty ααα

• Linearity in analogue (RF) systems does not exists• Amplifier y=A tanh(Bx)• Or in a Tailor series:

• Effects of non-linearity• Generation of harmonics• Compression• Blocking• Cross modulation• Inter modulation

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Harmonics• Suppose input signal:

• Follows:

DC-term, fundamental and harmonicsDifferential amplifiers cancel even harmonicsDe nth harmonics grows with An

( )tAtx ωcos)( =

( )

)3cos(4

)2cos(2

)cos(4

32

)(cos)(coscos)(3

32

23

31

22

333

2221

tAtAtAAA

tAtAtAty

ωαωαωααα

ωαωαωα

++⎟⎟⎠

⎞⎜⎜⎝

⎛++=

++=

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Gain compression• For small input signals the amplifiers has a

more or less linear behavior• For large input signals

– 1dB compression point:

20logAout

A1dB 20log Ain0

1 dB

3

11 145.0

αα

=dBA

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Blocking• Weak signals can be blocked by a strong

interferer

• If A1 << A2:

)cos()cos()( 2211 tAtAtx ωω +=

...)cos(23)( 11

2231 +⎟

⎠⎞

⎜⎝⎛ += tAAty ωαα

The gain of the wanted signal depends on α3

α3 < 0 reduces the gain for increasing A2

blocking

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Cross modulation• Weak signals can modulated by a strong

interferer

• If A1 << A2:

• For a modulation of the strong interferer:


...)cos(23)( 11

2231 +⎟

⎠⎞

⎜⎝⎛ += tAAty ωαα

( ) )cos()cos(1 22 ttmA m ωω+

)cos()cos(2)2cos(22

123)(

22221311 ttmtmmAAAty mm ωωωαα ⎥

⎦

⎤⎢⎣

⎡

⎭⎬⎫

⎩⎨⎧

++++=

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Inter-modulation• The output of an amplifier:

• Inter-modulation is generated when two signals ‘mix’

...)()()()( 33

221 +++≈ txtxtxty ααα


[ ][ ]

[ ] ..)cos()cos(

)cos()cos(

)cos()cos()(

322113

222112

22111

++

++

++=

tAtA

tAtA

tAtAty

ωωα

ωωα

ωωα

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This results in second and third inter-modulation products

ω1 ω2 ω ω1 ω2 ω2ω1− ω2 2ω2− ω1

ω2− ω1 ω1+ ω2

tAAtAA

tAAtAA

tAAtAA

)2cos(4

3)2cos(4

3

)2cos(4

3)2cos(4

3

)cos()cos(

121

223

121

223

212

213

212

213

2121221212

ωωαωωα

ωωαωωα

ωωαωωα

−++

+−++

+−++

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The distortion products change different with inputs levels

f f2f - f1 2 1 2

Powerin dB

33

2 12f - f

2 2

f2-f1 f1+f2

On a logarithmic: ∆2 becomes 2∆

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IP2 en IP3• Example IP3

Out

put p

ower

, dB

m

0

-20

-40

-60

-80

-100

-60 -30 0 +30IIP3

First order

Thirdorder

Input power [dBm]

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IP 2 and IP 3

• Following the math:

3

∝

∝

αα

αα

1

2

1

343

2

IP

IP

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Noise

• External noise– Man-made noise: generated by equipment– Atmospheric noise: e.g caused by lightning– Space noise: e.g. the sun

• Internal noise

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Internal noise

• Thermal noise produced by random movement of electrons

Noise power: PN = kTBT = absolute temperature in Kk = Boltzmann’s: 1.38x10-23 J/KB = bandwidth in Hz

• Shot Noise – random variations in currents in active elements.

• Partition Noise - caused by multi patch effects• Excess Noise (1/f noise) – caused by density

variations in components.• Transit-Time Noise – high frequency noise

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Noise Spectrum of electronic components

DeviceNoise

Shot and Thermal Noises

Excess orFlicker Noise

Transit-Time orHigh-FrequencyEffect Noise

1 kHz fhcf

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Signaal to Noise ratio

• A very important parameter in de communication theory is the signal to noise ratio (SNR of S/N).

• Expressed in dB’s:

N

S

N

S

VVlog20

PP log 10 dB)(

NS

==

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Noise factor & Noise figure

Noise factor Fn = SNRin/SNRout

Noise figure NF (dB) = 10 log Fn= SNRin (dB) - SNRout (dB)

Equivalent noise temperature, Te = (Fn -1) Towith To = 290 K

0 1 2 3 4

NF in dB

0 75 170 290 438

Effective noise temperature in K

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Noise factor in cascaded stages

• For amplifiers in cascade, the noise measurebecomes:

12121

3

1

21 ...

1...11

−

−++

−+

−+=

n

nM GGG

FGG

FG

FFF

G2F2

G3F3

G1F1

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Example noise calculations

F1=6dB=4 F2=3dB=2 F3=16dB=40 G1=20dB=100 G2=-3dB=0.5 G3=60dB=1000000

With pre-amp:

Without pre-amp:

Verlies =3 dB

G3= 60dBF3=16dB

G1 = 20 dBF1 = 6dB

Pre-amp Cable Receiver

dBF 8.679.45.0*100

140100

1240 ==−

+−

+=

dBF 19805.0

14020 ==−

+=

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Example gain and noise calculation

Noise temperature of the amplifier: T3=(F3-1)T0=2610KReceiver noise temperature: T=T2+ T3/G2=125+2610/100 = 151KTotal noise power = G2 G3 k(Tg+T)B = 1010k(60+151)106=-45.3 dBW

We can calculate the required transmit power given a SNR of 20dB:

Pz(dBW)+6-190+40+20+80=-45.3+20(SNR) Pz = 18.6 dBW =73 Watt

G2 = 20 dBT2 = 125K

G3 = 80 dBF3 = 10dB

Antenna

Tg=60K, G=40dB

Pre-amplifier amplifier

Satellite, G=6dB

Loss = 190dB

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Sensitivity• Defined as the minimal signal that the system

can detect with an acceptable SNR

• Noise floor:

minmin, log10/174 SNRBNFHzdBmPin +++−=

BNFHzdBmPruisvloer log10/174 ++−=

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Dynamic range• Defined as the ratio between the maximum

signal that can be handled by the circuit and the minimum input signal.

could be defined as:DR= Dynamic rangen = orderIPin = input inter modulation Intercept pointMDS = minimum detectable signal

• Also SFDR = spurious free Dynamic Range

nMDSIPn

DR innn

))(1( , −−=

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RF building blocks

• Amplifiers• Oscillators• PLL’s• Filters• Mixers• Modulation / demodulation

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Amplifiers

• Example of design parameters:– Noise figure NF (e.g 2 dB)– Input IP3 (e.g. –10 dBm)– Amplification (e.g. 20 dB)– Input impedance (50 Ohm)– Output impedance (50 Ohm)– Input return loss (-15 dB)– Output return loss (-15 dB)– Reverse isolation (25 dB)

• Practicum: Determine amplification, noise figure and IP2 / IP3

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Oscillators - principle

• Conditions for oscillation– The phase shift in the

complete feedback loop should be 0o of 360o

– The loop gain |BAv| = 1, With B = attenuation of feedback circuit, and Av = amplifier gain

B

AvVout

Basic elements of an oscillator

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Basic Wien-Bridge Oscillator

_

+

C2

C1 R4

R3

R1

R2

Voltage divider

Lead-lagcircuit

Vout+

_

R1

R4

R2

R3 C2

C1

Vout

Two versions of identical principle

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Wien-Bridge Oscillator

• At the resonance frequency generates the lead-lag circuit a positive feedback with a attenuation of 1/3 if R3=R4=XC1=XC2.

• For oscillation, requires a non-inverting amplifier with a gain of 3x. Possible whenR1 = 2R2

• If R3 = R4 = R, en C1= C2 = C, then the resonance frequency equals:

RCfr π2

1=

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Phase shift Oscillator

+

_

Rf

C1 C2 C3

R1 R2 R3

Vout

Each RC section causes 60 degrees phase shift. The total attenuation in thethree-section RF feedback circuit is B = 1/29.

If: R1 = R2 = R3 = R,C1 = C2 = C3 = C,Then resonancefrequency equals:

RCfr 62

1π

=

293

==RR

A fcl

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Colpitts Oscillator

+VDD

R2 C5

C3

R1 R3

L

C1 C2

C4

VoutVAC

CB 1

1

2 ==

Tr LC

fπ2

1=

met21

21

CCCCCT +

=

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Crystal Oscillator• For a stable and accurate oscillator

piezoelectric crystal (e.g. quartz) can be used in the feedback loop.

• Piezoelectric effect: For a changing mechanical stress at the crystal, a voltage will be generated. Or reverse, when an AC signal is applied, the crystal will resonate (vibrate) at the frequency of that signal. The largest resonance will occur at the resonance frequency.

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Symbol and electric equivalent circuit

XTAL

Cp Cs

Ls

Rs

Symbol Electricalequivalent

• A crystal can have a series and a parallel resonance.

• Crystals have a very high Q

• Resonance frequencies depend on dimensions, types, temperature etc.

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Basic Crystal Oscillator

C1R2 R4

+VCC

R1 R3

C2

CCXtalC1

R1 R2

R3

C2

C3Vo

C4

Vo

+VCC

C5

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Voltage-Controlled Oscillator (VCO)• VCOs are applied in many systems e.g. AFC,

PLL, frequency tuning, etc.• The principle is based on changing a capacitor

of a varactor diode in a resonance circuit. • The approximate total capacity of the diode

depends on the bias spanning:

b

oV V

CC21+

=

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Phase noise

ωωc

Ideal oscillator

ωωc

∆ωPractical oscillator

))(cos()( ttAtx nc φω +=

Example: •Carrier power = -2dBm•Noise power in a 1kHz bandwidth with 1MHz offset is –70dBm•Phase noise = -70 dBm + 2 dBm (carrier) – 30 dB (bandwidth)

= -98 dBc/Hz

φn(t)= phase noise

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Phase-Locked Loop• The PLL is the building block for modern

synthesizers. • The block diagram for a simpel PLL:

PhaseDetector LPF Loop

Amplifier VCOfr fo

Vp

Divider :n

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The PLLInitial a PLL is not locked; the VCO is on the free-running frequency, fo.If fo is not equal to frequency fr , a Vp will be generated by the phase detector.This voltage Vp will go through the filter before amplification and subsequently applied to the VCO. A stable system will be: fo = fr. The PLL is in phase lock.

( if n=1 )

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PLL Frequency Specifications

Free-RunningFrequency

Catch range

Lock range

fofLCfLL fHC fHLf

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Potential problems with PLL’s

• Phase noise• Spurious • Lock issues

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Filter design

• Filters are used for many reasons:– Suppression of harmonics– Suppression of image frequencies– Selectivity– Demodulation– Etc…

• Parameters for the design of filters :– Attenuation– Bandwidth– Center frequency– Cutoff frequency– Delay (differential and groups-)

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Filter design

– Insertion lossFor wanted and un-wanted signals.

– Pass band– Pass band ripple– Phase behavior– Poles– Q-factor– Return loss– Shape factor

The shape factor is defined as de ratio between bandwidth at -60 dB and the bandwidth at -3 dB

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Low Pass Filter Response

Vo

fcf0

10.707

BW

Gain (dB)

0

-20

-40

-60

fc 10fc 100fc 1000fc

-20 dB/dec

-40 dB/dec

-60 dB/dec

LPF with different roll-off ratesBasic LPF response

f

Ideal

Pass band

BW = fc

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High Pass Filter Response

Vo

fc f0

1

0.707

Gain (dB)

0

-20

-40

-60

0.01fc 0.1fc fc

-20 dB/dec

-40 d

B/de

c-6

0 dB

/dec

Pass band

Basic HPF response HPF with different roll-off rates

f

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Band-Pass Filter Response

Vout

1

0.707

ffofc1 fc2

BW

BW = fc2 - fc1

21 cco fff =Center frequency:

Quality factor:BWfQ o=

Q is an indication for the Selectivity of a BPF.Small BPF: Q > 10.Wide-band BPF: Q < 10.

Attenuation Factor: QDF 1=

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Gain (dB)

0-3

ffofc1 fc2

BW

Pass band

Band-Stop Filter Response

• Also known as band-reject, of notch filter.

• Used for suppression of RFI

example

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Application for notch filters

Nu:2*TV-Drenthe - NL3 = 347 MHz2*TV-Drenthe - NL2 = 323 MHzTV-Drenthe – NL1 = 320 MHz

f181 MHZ 501 MHZ 655 MHZ 679 MHZ

NL1 TV-D NL3 NL2

92 cm-band310-390 MHz

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Filter Response characteristics

Av

f

Chebyshev

Butterworth

Bessel

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Filter characteristics• Butterworth: flat amplitude response in the

pass band. Slope -20 dB/dec/pole; Phase response non-linear.

• Chebyshev: Slope > -20 dB/dec/pole; ripples in the pass band; Phase response very non-linear.

• Bessel: linear phase response, Slope < -20 dB/dec/pole.

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Mixers

• A mixer is a non linear circuit combining two signals, generating a plus and a minus signal

Vrf

VLO

VIF

ttV

ttVtVtV

RFLORFLOIF

RFLOIF

RFRF

LOLO

)cos(21)cos(

21

)sin().sin()sin()sin(

ωωωω

ωωωω

+−−=

===

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Balanced Mixers• A balanced mixer suppresses the input signals.

Circuit symbol:

f1

f2

f1+ f2

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Single-balanced Mixer

LO

IF

D1

D2

RF

...)cos(1)cos(1)sin(21

+−++−= tttV RFLORFLORFIF ωωπ

ωωπ

ω

-The RF signal is still present in the IF

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Double balanced Diode ring mixer

LO

IF

D1

D2

RF

D4

D3

ttV RFLORFLOIF )cos(2)cos(2 ωωπ

ωωπ

−++−=

-None of the input signals is present in the IF

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Dual-Gate (active) MOSFET Mixer

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Image reject mixer• Also used in the Westerbork array

90o

90ofilter

filter

Input

LO

Output

)cos()cos()( lluu tttx φωφω +++=))cos(()(),)cos(()(

lLOlLO

LOuLOu

tAtyoftAty

φφωωφφωω

−+−=−+−=

low highIF

(SSB)LO

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Modulation en demodulation

• Analogue modulation– Amplitude modulation– Single Side Band modulation– Phase en frequency modulation

• Digital modulation– Binary modulation (e.g. FSK)– Quad modulation (e.g. QPSK)– …

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AM wave form

ec = Ec sin ωctem = Em sin ωmt

AM signaal:es = (Ec + em) sin ωct

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Modulation Index• The level of amplitude modulation: modulation

index:

minmax

minmax

EEEEof

EEm

c

m

+−

=

If Em = Ec , m =1 of 100% modulation.

Over-modulation, if Em>Ec : distortion

With, Emax = Ec + Em; Emin = Ec - Em

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Effects of the Modulation Index

m = 1 m > 1

For more then one frequency component:

222

21 ... nT mmmm +++=

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AM in the frequency domain

• The formulation for a AM signal:es = (Ec + em) sin ωct

is equal to:

es = Ec sin ωct + ½ mEc[cos (ωc-ωm)t-cos (ωc+ωm)t]

• The AM signal consists of original carrier, an lower side frequency flsf = fc - fm, and an upper side frequencyfusf = fc + fm.

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AM Spectrum

ffc

Ec

fusf

mEc/2mEc/2

flsf

fmfm

fusf = fc + fm ; flsf = fc - fm ; Esf = mEc/2Band width, B = 2fm

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AM power• Total average (rms) power of an AM signal is:

PT = Pc + 2Psf , withPc = carrier power; and Psf = side band power

• For a load R, gives this: Pc = Ec2/(2R); and

Psf = m2Pc/4. So,

)2

1(2mPP cT +=

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Complex AM wave form• For complex AM signals with more frequency

components, the modulation index m will be substituted by a new index mT.

)2

1(2

TCT

mPP +=

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AM modulation

LO

IF RF

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AM demodulation

cos(ωct)

xAM(t)LPF

Draw backs of AM:

• Band width is 2x more then required• Power in de carrier is lost energy

• Single side band modulation (SSB): USB / LSB

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Phase modulation

[ ])(cos)( tmxtAtx BBccPM += ωm = phase modulation index

tmAtxttx

ccPM

BB

)cos()()(

αωα

+=⇒=

A slope generates a frequency shift

(Linear)

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Frequency modulation

⎥⎦

⎤⎢⎣

⎡+= ∫

∞−

t

BBCCFM dttxmtAtx )(cos)( ω

m = frequency modulation index

tmAAtxAtx

ccFM

BB

)cos()()(

+=⇒=

ω

A DC input generates a frequency shift

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Phase en frequency wave form

t

xBB(t)

t

xPM(t)

t

xFM(t)

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Frequention modulation/demodulation

xBB(t)xFM(t)

VCO

C

R

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Examples• Receiver architectures

• 50 MHz transceiver

• Westerbork array

• LOFAR, low frequency array

FSKFSK

CWCWPLL SynthesizerPLL Synthesizer

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Receiver architectures

• Direct conversion receiver (also known as homodyne receiver)

• Super heterodyne receiver• Image –reject receiver• Direct sampling receiver

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Homodyne receiverSimple type: direct to Baseband mixing (direct conversion)

A0cosωot

ωω0

LDF

ω0

LNA

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(Super)heterodyne ontvanger

A0cosωot

ωω1

BPF

ωω2

12 ωωω −= LO

Band selection filter is easy (low Q)

ω1

Problem:

ωLO ωimageωIF ωIF

image

• Solution: filter before the mixerImage reject filter

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Example: the single super

Modefilter

LFHF

filterDet.

LO

MFHF

Draw backs:

•For a MF 455 MHz, the image frequency will be 900 MHz•Could still be in the HF band•Sharp filters required

The double super

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Example: the double super

• High first MF• Second LO fixed

MFfilter

LF

HFfilter Det.

LO 1

2e MFHF

Modefilter

LO 2

1e MF

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Image reject receivers

• Avoid the requirement for image reject filters:

– Hartley architecture– Weaver architecture

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Hartley architecture

90o

LPF

LPF

IF outputRF input sinωlot

cosωlot

A

C

B

ω-ωLO ωLO0

image

Wanted signal

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0

ω

0

+j/2 +j/2

-j/2-j/2

A

0

0

ω

ω

ωB

0

ω0

C

ω

900

0 ω

IF output

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Direct sampling ReceiverOver-sampling Receiver

HF

Band selection with digital filter and mixer

Requires high data rate A to Digital Converter

LPF LPF

LO

A

D

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Direct sampling ReceiverSub-sampling Receiver

HF

Under sampling: lower sample rate ADCs required

Band selection with analogue filter

BDFA

D

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50 MHz Transceiver

• Specifications– Modes : CW en FSK– Image suppression 60 dB– Sensitivity receiver: 0.15 µV with 10 dB SNR

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Possible design

FSKFSK

CWCW

PLL SynthesizerPLL Synthesizer

50-51 MHz

Image reject filter

Receiver

Transmitter

LO

IFMixer

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Some calculations

• Sensitivity of 0.15 µV and SNR of 10 dB– For SSB : BW = 2.5 kHz– Noise floor P=4kTB = -134 dBm– 10 dB SNR minimum signal = -124 dBm– 0.15 µV @ 50 Ω -123.46 dBm– With other words: the noise figure of the receiver

should better then 0.6 dB

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The Westerbork Array• WSRT - Westerbork Synthesis Radio Telescope

–Equivalent 93 m telescope for area–Equivalent 2700 m telescope for resolution–14 telescopes of 25 m diameter with 2700 m baseline

• Synthesis–Phase coherent combinations–Correlation of all signals–Earth rotation, 12 uur–4 telescopes moveable

• Coherent addition–Fanbeam: 0,5 x 0,004 degrees(1420MHz)

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The signal path, block diagram

Recv.Recv.

Equal.Equal.

IVCIVC

ADCADC

Recv.Recv.

Equal.Equal.

IVCIVC

ADCADC

Receivers in 14 telescopeswest east

Coaxial cables

Central building

TADU

ΣTADU

Σ

LOLO LOLOVLBIVLBI PuMaPuMa

CorrelatorCorrelator

Ref. Ref.

∆φ ∆φ∆τ∆τ

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The receiverMFFE:Multi Frequency Front End

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Frequency bands (1)Wave length Frequency-

band92 cm 310-390 MHz

49 cm 560-620 MHz

21 cm 1200-1450 MHz

18 cm 1590-1750 MHz

13 cm 2215-2375 MHz

6 cm 4770-5020 MHz

3,6 cm 8150-8650 MHz

UHF low (120-65 cm)

250-460 MHz

UHF high (43-25 cm)

700-1200 MHz

Two polarizations for all frequencies

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Frequency bands (2)3.6 cm6 cm13 cm18 cm21 cm49 cm92 cmUHF highUHF low

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Astronomical demands

• Sensitivity

• Clean bands (no RFI)

• Stable system

τBTTT recsky+

≈∆

Requires low noise receivers…

…long integration time…

…and large bandwidth.

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MFFE Block diagram (1)UHFlow UHFhigh 92 49 13 18/21 6 3.6

swsw

LO1 low Synth.

LO1 low Synth.

swsw

1200-2200 MHz

1 GHzIF1 IF1

LO1 high Synth.

LO1 high Synth.

swsw

2200-9600 MHz

1 GHz

1 GHz

LO2 Synth.

LO2 Synth.

900 MHz

2x IF out100 ± 80 MHz

Cryogenic

94


JGBdV 2006

MFFE Cryostat

95


JGBdV 2006

MFFE Block diagram (2)UHFlow UHFhigh 92 49 13 18/21 6 3.6

swsw

LO1 low Synth.

LO1 low Synth.

swsw

1200-2200 MHz

1 GHzIF1 IF1

LO1 high Synth.

LO1 high Synth.

swsw

2200-9600 MHz

1 GHz

1 GHz

LO2 Synth.

LO2 Synth.

900 MHz

2x IF out100 ± 80 MHz

96


JGBdV 2006


Recv.Recv.

Equal.Equal.

IVCIVC

ADCADC

Recv.Recv.

Equal.Equal.

IVCIVC

ADCADC


coaxial cables

Central building

TADU

ΣTADU

Σ

LOLO LOLO


Ref. Ref.


97


JGBdV 2006

The Equalizer

Equalizer

-50

-40

-30

-20

-10

0

10

20

30

10 30 50 70 90 110 130 150 170 190

Freq (MHz)

Gain (dB)

98


JGBdV 2006


Recv.Recv.

Equal.Equal.

IVCIVC

ADCADC

Recv.Recv.

Equal.Equal.

IVCIVC

ADCADC


coaxial cables

Central building

TADU

ΣTADU

Σ

LOLO LOLO


Ref. Ref.


99


JGBdV 2006

IF to Video Converter (IVC)• Converts (IF) to

baseband (Video)• Splits 160 MHz band in

8x20 MHz bands

We have 14+2 IF channels, with each 2 polarizations divided in 8 bands: totals 16x2x8=256 modules!

100


JGBdV 2006

Block diagram IVC

div

div

8filters

BW=20 MHz

BW=156 kHz

20-180 MHzinput

LO1240-400 MHz

LO2200 en 220 MHz

200-220 MHzIRM video

out

LO1LO1 LO2LO2LO2

LO1

0 20 180 220 400200 240

80 100 30010MHzref.

10MHzref.

101


JGBdV 2006

IVC Modules

Converter module

Filter module

LO module

102


JGBdV 2006


Recv.Recv.

Equal.Equal.

IVCIVC

ADCADC

Recv.Recv.

Equal.Equal.

IVCIVC

ADCADC


coaxiale cables

Central building

TADU

ΣTADU

Σ

LOLO LOLO


Ref. Ref.


103


JGBdV 2006

The A/D Converter system

• Conversion from baseband to digital

• Bandwidth is 20 MHz, so for Nyquist sampling 40 MHz is required

• The signal is noise: only 1 bit coding required.

• 2 bits sampling used

104


JGBdV 2006

Block diagram ADC

samplersampler delaydelay adderadder

formatterformatter

timingtiming

256 analoginputs

Referenceclock

Total power information

32 digital outputs to correlator

3 bits S,M,V

Shifted samplingclock

Sampling clock32 summedoutputs to tied array applications

25 nsec

128 steps of 0.2 nsec

Max. delay 1.6 msecin steps of 0.2 nsec

105


JGBdV 2006

Sampler Module

Analogue part Digital part

106


JGBdV 2006


Recv.Recv.

Equal.Equal.

IVCIVC

ADCADC

Recv.Recv.

Equal.Equal.

IVCIVC

ADCADC


coaxial cables

Central building

TADU

ΣTADU

Σ

LOLO LOLO


Ref. Ref.


107


JGBdV 2006

The Correlator

• Combinations of all telescopes are made• Also ‘combinations’ in the time domain: spectral

information

• Pulsar gating

108


JGBdV 2006

Correlator Board

32 custom correlatorchips

10 custom crossbar chips

2 DSP chips

13 Mb srammemory

109


JGBdV 2006

LOFAR Radio Ontvanger Systeem

AnalogueProcessing

AnalogueTo

Digital

DigitalProcessing

PostProcessing

Visualization

Storage

Antennas

110


JGBdV 2006

LOFAR Radio receiver system

Documents

Jan-Geralt Bij de Vaate ASTRON - Onderzoekpeletier/RFsystemen2.pdf2 Basic Detection Techniques – “RF systems” JGBdV 2006 Content: • Introduction RF design • Basic principles