A Widely Tunable Active Duplexing Transceiver withSame-Channel Concurrent RX/TX and 30dB RX/TX Isolation

Dong Yang and Alyosha Molnar

Cornell University, Ithaca, New York, 14853, USA

Abstract — We present a widely tunable passive mixer-first duplexing transceiver which employs baseband noise-canceling, duplexing LNAs. The LNAs buffer transmittedsignals to the mixer while canceling those signals in receivepath. The transmitted signals are up-converted by the samemixer used for receiver down-conversion. The transceiveroperates over a frequency range of 0.1-1.5GHz with -18dBmtransmitted power. A 33dB linear isolation between receiveand transmit (separated by 135kHz) as well as suppressionof transmit-induced noise and nonlinearity are achieved.

Index Terms — Duplexer, wideband transceiver, passivemixer, software-defined radio (SDR), low noise amplifier.

I. INTRODUCTION

Recent developments in communication theory indicate

that system performance can be significantly enhanced by

using transceivers that support same-channel full-duplex

links among local nodes [1]. Particularly, a low noise long-

range down-link combined with short-range full-duplex

relaying has applications in interference alignment [2],

distributed MIMO [3], and other cooperative schemes. For

short-range network, even relatively low power transmitted

signals can provide reasonable link margin.

In this paper, we present an 8-phase passive mixer-first

full-duplex RF front-end with baseband noise-canceling

LNAs as duplexers. With the transparency property of

passive mixers [4], active duplexer circuitry can be moved

to baseband to provide easier tuning of circuit parameters.

The single mixer performs both up-conversion and down-

conversion at the same time, using a single LO chain.

A differential load sharing technique enables cancella-

tion of the transmitted signals within baseband duplexing

LNAs. The complex impedance on the RF port is balanced

by a tunable complex feed-forward network maintaining

high receive/transmit isolation across frequency.

II. TRANSCEIVER ARCHITECTURE

Fig. 1 shows the system architecture of the transceiver.

A wideband frequency divider generates 8-phase nonover-

lapping pulses, which drive passive mixer switches [4].

The baseband ports of the mixer are fed to four differential

baseband duplexing LNAs which provide isolation from

the transmitted signals and tunable impedance matching.

The outputs of duplexing LNAs are further amplified

by differential amplifiers providing buffering and gain

Wide Range Frequency Divider

0o 180o

90o 270o

45o 225o

135o 315o

÷4 and

phase split

+ LO -

Baseband Amplifiers

Passive Mixers

RF Antenna

0o

45o

90o

135o

180o

225o

270o

315o

Duplexing LNA

0o

180o

TX Input

45o

225o

90o

270o

135o

315o

0o

180o

45o

225o

90o

270o

135o

315o

BB Output

Fig. 1: System architecture of transceiver with LO generation,passive mixer, baseband duplexing LNAs and amplifiers.

control. The 8-phase baseband transmitter input signals are

buffered by the LNAs and then up-converted by the same

passive mixer, saving LO power and complexity.

III. NOISE-CANCELING BASEBAND DUPLEXING LNA

A conventional RF noise-canceling LNA consists of a

common-gate and a common-source amplifier with their

inputs tied together [5]. The same mechanism for noise-

canceling can also cancel a signal injected to the gate of

common-gate element [6]. In our design, as in Fig. 2(a),

the gate of M1 is used to inject the baseband transmitted

signals instead of being held at a fixed DC bias.

A. Impedance MatchingThe input of each duplexing LNA is connected to one of

the 8 passive mixer switches. Looking into the RF port, the

in-band impedance will be Rin ≈ Rsw +8/gm1 while the

out-of-band impedance will be Rin ≈ Rsw. Conversely,

the LNA sees an effective source impedance of:

Rs ≈ 8(Ra +Rsw) (1)

In our design, Rsw ≈ 10Ω, which translates to Rs ≈480Ω for a standard RF source impedance of Ra = 50Ω.

B. Rejection of Transmitted SignalsFor the received signal, the single-ended input generates

differential outputs given by:

V +o,RX = gm1R1 ·RXin, V

−o,RX = −gm2R2 ·RXin (2)

978-1-4799-3864-3/14/$31.00 © 2014 IEEE 2014 IEEE Radio Frequency Integrated Circuits Symposium

RTU2A-3

321

���� M1

M2

���

R2R1

����

����

Rs

(a)

Rs

����

M1

M2

R2

M2

R1-R2

R2

R1-R2

M1

�����

�����

��� ���

�

����

Rs

(b)

Fig. 2: (a) Single-ended noise-canceling duplexing LNA. (b)Fully differential duplexing LNA with load sharing.

To generate fully differential signals, the condition is:

gm1R1 = gm2R2 (3)

For the transmitted signal, the baseband input to the gate

of M1 will generate in-phase signals at the outputs:

V +o,TX = − gm1R1

1 + gm1Rs· TXin (4)

V −o,TX = −gm1gm2R2Rs

1 + gm1Rs· TXin (5)

Equating (4) and (5) gives the condition for cancellation

of transmitted signals:

R1 = gm2R2RS (6)

With input impedance matched, RS = 1/gm1, this con-

dition is the same as in (3). With R1 = 2.20kΩ and R2

= 630Ω, the gain of duplexing LNA is 13.2dB.

C. Noise Analysis

When the condition for transmit cancellation is met, the

noise contribution of M1 is also automatically canceled

[5]. For differential outputs, ignoring noise from R1 and

R2, the noise factor is:

F = 1 +v∗2n,M2

v∗2n,RS

= 1 +RS

gm2(7)

To minimize NF, it requires gm2 < RS . When the input

impedance is matched, this condition becomes gm1 < gm2

and therefore, R1 > R2.

D. Differential Duplexing LNA with Load Sharing andComplex Feed-Forward

One of the drawbacks of this simple noise-canceling

LNA architecture for in-band duplex is that the transmitted

signals are still present at the outputs as common mode,

and thus may saturate the outputs. However, by merging

R2 into R1 and employing a fully differential structure,

as shown in Fig. 2(b), the common mode signals are also

��

R2

R1-R2

R2

R1-R2

���

�� �

��

���

R1-R2 R1-R2

����

����

���

Rs RsRs

��� ���

� �� ��

�

���

���

���

���Positive Complex

Feed-ForwardNegative Complex

Feed-Forward

R2 R2

Rs

Fig. 3: Positive and negative complex feed-forward transconduc-tance applied to the baseband duplexing LNAs.

suppressed. The voltage drop across R3 (R3 = R1 −R2)

and R2 due to the transmitted signals will be:

V ±R3= −gm1 · TX±

in

1 + gm1RS(R1 −R2) (8)

V ±R2=

(−gm1 · TX±

in

1 + gm1RS− gm1gm2RS · TX∓

in

1 + gm1RS

)R2 (9)

such that the LNA outputs are:

V ±out = −gm1R1

1 + gm1RS·TX±

in−gm1gm2R2RS

1 + gm1RS·TX∓

in (10)

Therefore, with the same condition in (3), the transmitted

signals are canceled at the outputs. The difference in the

bias currents of M1 and M2 is compensated by a PMOS

bias current. The M2 transistors of differential LNAs are

configured as a differential pair for ease of biasing.

In this architecture, especially at high frequencies, the

antenna impedance can have reactive components, such

that RS becomes complex. To still satisfy the condition for

cancellation of transmitted signals and noise for complex

values in ZS , we added transconductance paths between I

and Q as in Fig. 3, effectively replacing gm2 with a com-

plex transadmittance ym2. In this case, (6) changes to:

R1 = ym2R2ZS = (gm2 + jbm2)R2(RS + jXS) (11)

The complex feed-forward paths are capable of providing

both positive and negative bm2 with different polarity. Thus

with properly selected bm2, the canceling condition can

still be satisfied.

IV. MEASUREMENT RESULTS

This transceiver was fabricated in a 65nm CMOS pro-

cess with an area of 1.5mm2, as seen in Fig. 4. It functions

across frequency of 0.1-1.5GHz with a power consumption

of 43mW to 56mW (7-20mW for LO generation with a

1.2V supply and 36mW for baseband with a 2.5V supply).

For receiver performance, Fig. 5 shows the NF and

receive gain of the transceiver across its entire range of

322

Fig. 4: Microphotograph of implemented transceiver.

0 500 1000 1500

5

6

7

8

9

Frequency, MHz

DS

B N

oise

Fig

ure,

dB

0 250 500 750 1000 1250 1500

40

44

48

52

56

Rec

eive

Gai

n, d

B

Fig. 5: Measured NF and receive gain across RF frequency.

operating frequencies. It achieves 5-8dB NF and maintains

a gain of 53±2dB with maximum gain setting.

Fig. 6 shows that the frequency of optimum S11 tracks

LO frequency and is maintained across a wide frequency

range. The input impedance can be tuned by varying the

bias current of M1 as shown in Fig. 7.

The out-of-band and in-band linearity of the transceiver

were also characterized. We measured the out-of-band IIP3

to be +22.5dBm (dominated by the transistors in the mixer

[4]) and the in-band IIP3 to be -32.7dBm.

We also characterized the transceiver in full duplex:

transmitting and receiving concurrently, separated by

135kHz. Fig. 8 shows the effect of the transmitted power

on the isolation between transmit and receive. To quantify

this isolation, we calculated the expected receiver output if

the transmitted signals were directly injected into the RF

port, and defined the isolation as the difference in ampli-

tude between this and the actual receiver outputs due to the

transmitted signals. Averaged across all four differential

output channels, the isolation remained >33.5dB as the

transmitted power reached -17.3dBm, and for the channel

with worst transmit leakage, it remained >28.3dBm.

However, this linear isolation is not necessarily the

most critical measure of duplexing performance, since

such leakage can easily be canceled in subsequent DSP,

by more than 50dB [1]. More critical are mechanisms

by which the transmitter degrades the received signals

unpredictably. Fig. 9 shows the effect of transmitted power

on the NF and receive gain. The receive gain is compressed

by 1.3dB at maximum transmitted power of -17.3dBm,

while the NF gradually increases with the transmitted

power before a rapid degradation starting at ∼-25dBm. For

comparison, the same specifications (compression and NF

0 200 400 600 800 1000 1200 1400 1600

−50

−40

−30

−20

−10

0

Frequency, MHz

S11

, dB

Fig. 6: Measured S11 for varying LO frequencies.

680 690 700 710 720

−50

−40

−30

−20

−10

0

Frequency, MHz

S11

, dB

340μA320μA300μA280μA260μA240μA

Fig. 7: Measured S11 for varying bias current of M1 at 700MHz.

−60 −55 −50 −45 −40 −35 −30 −25 −20 −15

25

30

35

40

Transmitted Power, dBm

TX

/RX

Isol

atio

n, d

B

Average across OutputsOutput with Maximum TX Leakage

Fig. 8: Measured TX/RX isolation for varying TX power at700MHz.

degradation) are measured using a second RF input at the

same frequency and power level as the transmitted signals.

For the same NF degradation, the transmitted power can

be ∼28dB larger than the equivalent received power, while

for similar compression, this difference is ∼35dB.

We also measured the TX/RX isolation across RF

frequency and characterized the effect of complex feed-

forward transconductance bm2 in Fig. 10. A proper choice

of bm2 improves the isolation by up to 14.3dB. To further

confirm the utility of complex feed-forward, the NF and

receive gain were measured while varying bm2 in the

presence of a -22dBm transmitted signal, as in Fig. 11. The

optimum NF, receive gain and isolation are all achieved

with similar settings of bm2. Note that while bm2 (and

potentially gm2) must be adapted as frequency or antenna

impedance vary, this setting can easily be adjusted on the

fly by monitoring the TX/RX leakage and adapting bm2 to

minimize it. Furthermore, since leakage can be monitored,

other slow-changing parameters, such as process and tem-

perature variation, can similarly be canceled.

323

−60 −55 −50 −45 −40 −35 −30 −25 −20 −15

5

7

9

11

13

Transmitted Power, dBm

DS

B N

oise

Fig

ure,

dB

−60 −55 −50 −45 −40 −35 −30 −25 −20 −15

25

30

35

40

45

Rec

eivi

ng G

ain,

dB

Fig. 9: Measured NF and receive gain for varying TX power at700MHz (dashed: two-tone injection of RX, solid: duplexing).

0 200 400 600 800 1000

5

15

25

35

45

Frequency, MHz

TX

/RX

Isol

atio

n, d

B

With Complex Feed−ForwardWithout Complex Feed−Forward

Fig. 10: Measured TX/RX isolation across RF frequency withand without complex feed-forward engaged.

0.5 0.55 0.6 0.65 0.7 0.75 0.8 0.85 0.9 0.95 1

15

25

35

45

Bias Control Voltage of Feed−Forward Amplifiers, V

TX

/RX

Isol

atio

n, d

B

Average across OutputsOutput with Maximum TX Leakage

Fig. 11: Measured TX/RX isolation for varying bm2 at 800MHz.

To characterize the nonlinear interactions between re-

ceived and transmitted signals, we performed a series of

in-band 2-tone tests with different combinations of receive

and transmit tones. For input tones at frequencies f1and f2, we measured the IM3 tone at 2f1 − f2, where

the signal power of input-referred IM3 is expected to

be proportional to P 21P2. To characterize the effect of

transmitter suppression on TX-RX intermodulation, we

compute the ratio P 21P2/PIM3 for different combinations

of in-band transmit and receive tones: without transmit

isolation, this ratio would be the same in all cases. As

seen in Table 1, the ∼33dB linear rejection of transmitted

signals results in a ∼30dB reduction in the transmitter

contribution to TX-RX intermodulation.

V. CONCLUSION

In this work, we presented a full-duplex transceiver

capable of wideband operation and high isolation between

0.5 0.55 0.6 0.65 0.7 0.75 0.8 0.85 0.9 0.95 1

8

10

12

14

16

Bias Control Voltage of Feed−Forward Amplifiers, V

DS

B N

oise

Fig

ure,

dB

0.5 0.55 0.6 0.65 0.7 0.75 0.8 0.85 0.9 0.95 1

25

30

35

40

45

Rec

eive

Gai

n, d

B

Fig. 12: Measured NF and receive gain for varying bm2 at800MHz.

TABLE I:3RD ORDER NONLINEARITY WITH TWO-TONE INJECTION

Tone 1 Tone 2 Nonlinear PerformanceRX/TX P1(dBm) RX/TX P2(dBm) PIM3(dBm) P2

1P2/PIM3(dB)RX -59.8 RX -59.8 -114 -65.4RX -52.8 TX -25.9 -97.9 -33.6TX -25.9 RX -52.8 -102 -2.60TX -33.9 TX -33.9 -98.0 -3.70

same-channel transmit and receive. We proposed a base-

band duplexing LNA scheme which generates differential

receiver outputs and in-phase transmitter outputs. We have

also developed differential load sharing techniques to can-

cel the transmitted signals in a single stage. Finally, we

proposed a complex feed-forward network to counteract

the effect of complex RF port impedance.

ACKNOWLEDGMENTS

This material is based upon work supported by the NSF

under Grant No. 1247915; fabrication was provided by the

TSMC University Shuttle Program.

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