Forward-Flyback Mixed ZVS DC-DC Converter

With Non-Dissipative LC Snubber Circuit Frank Chen Emil Auadisian John Shen Issa Batarseh

School of Electrical Engineering and Computer Science University of Central Florida

Orlando, FL 32816, USA Frankchen@eecs.ucf.edu

Abstract— This paper presents a zero-voltage switching (ZVS) forward-flyback DC-DC converter, which is able to process and deliver power efficiently over very wide input voltage variation. The proposed ZVS forward flyback DC/DC converter is part of a Micro-inverter to perform input voltage regulation to achieving maximum power point tracking for Photo-voltaic panel. The converter operates at boundary between current continuous and discontinuous mode to achieve ZVS. Variable frequency with fixed off time is used for reducing core losses of the transformer, achieving high efficiency. In addition, non-dissipative LC snubber circuit is used to get both benefits: 1) the voltage spike is restrained effectively while the switch is turned off with high current in primary side; 2) the main switch still keeps ZVS feature. Finally, experiment results provided from a 200W prototype (30Vdc-50Vdc input, 230Vdc output) validate the feasibility and superior performance of the proposed converter.

I. INTRODUCTION

Due to the environmental concerns about the global warming, the fossil fuel exhaustion and the carbon dioxide reduction requirements, a solar micro-inverter called module integrated converter (MIC), one of renewable energy resources, has been catching more and more attention from government, academia and industry in recent years [1-3]. Commonly, the output voltage of a single PV array employing in the renewable energy is comparatively low and varies in a wide range, which needs high power density and high step up dc-dc converters to boost the low voltage to a high one for the grid-connected power applications. Generally speaking, the isolated step up dc-dc converter for MIC application should own these features of high efficiency, process capability of high input current, high voltage conversion ratio, the power rating typically from 200Watts to 300Watts and last but not least low cost. The topologies with galvanic isolation suitable for this application can be categorized into two groups: single switch topology and multi-switches topologies. Single switch topology mainly includes fly-back, forward. Multiple-

switches topology includes half bridge & full bridge. Recently, LLC resonant topology has been attracted by the academy and industry due to unique performance of high efficiency and natural ZVS/ZCS commutation features, which is widespread utilized for step down DC/DC converter such as application of telecom, adapter, etc [4-6]. Unfortunately, conventional LLC resonant topology as the first stage of step-up DC-DC converter is hardly implemented for MIC application due to difficultly maintain high efficiency over a wide input range with different load conditions.

According to the power rating of the MIC application for a single PV panel, compared to the multiple switches topology, single switch flyback and forward topologies are good candidates for step up DC-DC converter application due to their simplicity and low cost, while achieving high efficiency and the wide voltage operation range[7-10]. However, firstly the size of the transformer for the flyback converter is a major concern, especially while the power rating exceeds 100 Watts; Secondly, it is difficult to handle the current stress in the primary side of the transformer while the output voltage of single PV panel is less than 30 volts. On the other hand, the forward converter needs additional circuits or an auxiliary winding to reset the magnetizing current of the transformer.

Many researchers have been done for forward-flyback converter as shown in Figure.1, over the past several decades [11-13]. This topology, usually running at continuous current mode, is widely used for high input voltage and low output voltage application. In order to achieve zero voltage switching (ZVS) of primary switch S1, an auxiliary switch is necessary such as active clamp circuit, but it already said in the aforementioned problems [13] [14].

In this paper, a 200 watts ZVS forward-flyback step-up DC-DC converter with wide input voltage, from 30Vdc to 50Vdc, is proposed while the forward-flyback converter is designed at boundary current mode (BCM) of the output inductor L1 and discontinuous current mode (DCM) of the magnetizing inductor of the transformer that is shown in figure 1 [15]. This simple ZVS control scheme is proposed by

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employing the BCM&DCM without any extra hardware cost. The switching loss and reverse recovery current of the output rectifier diodes are greatly reduced. In addition, the on resistance of the low voltage Mosfet, with the rapid development of the semiconductor industry, has a very good performance so far; for example, where the on resistance of the 150 volts Mosfet is already less than 10 milliohms in D2Pak package. It is obvious that the conduction loss does not increase much under this control scheme of DCM&BCM compared to conventional continuous current mode (CCM). Unfortunately, the voltage spike across the Mosfet is very high at turn off time causing by the leakage inductance of the transformer [15]. In order to suppress voltage stress of the Mosfet, a non-dissipative LC snubber circuit is proposed in this paper. Section II analyzes the operation principle of the proposed ZVS Forward-flyback converter. Section III discusses the features of the converter steady state operation. Design and consideration of the ZVS Forward-flyback converter is presented in Section IV. Section V shows detailed experimental results to verify the operation principle of the proposed ZVS Forward-flyback converter.

Figure 1. Forward-Flyback Converter with LC Circuit

II. OPERATION MODE

The following assumptions are made: all passive components are considered to be ideal, which implies the ESRs of inductors and capacitors are neglected. S1, D1, D2 and D3 are all ideal semiconductor that are linearly switch on or off. Figure 2 shows operation mode of the ZVS forward-flyback converter with non-dissipative LC snubber circuit.

Figure 2. Operation Mode of ZVS Forward Flyback Converter

Lm

C1

S1

Tr D1

D2

C2

D3

RL

VinL1

N1 N2

N3

Vout

CossCs

Ls1

Ds1

Llk

Ds2

Lm

C1

S1

Tr D1

D2

C2

D3

RL

VinL1

N1 N2

N3

Vout

CossCs

Ls1

Ds1

t0 -t1

Im

Llk

IS1

iLs

Ds2

iL1

Lm

C1

S1

Tr D1

D2

C2

D3

RL

VinL1

N1 N2

N3

Vout

CossCs

Ls1

Ds1

t1 -t2

Im

Llk

IS1

iLs

Ds2

iL1

Lm

C1

S1

Tr D1

D2

C2

D3

RL

VinL1

N1 N2

N3

Vout

CossCs

Ls1

Ds1

t2 -t3

Im

Llk

ICoss

iLs

Ds2

iL1

Lm

C1

S1

Tr D1

D2

C2

D3

RL

VinL1

N1 N2

N3

Vout

CossCs

Ds1

t3 -t4

Im

Llk

ICoss

iLs

Ds2

iL1

Lm

C1

S1

Tr D1

D2

C2

D3

RL

VinL1

N1 N2

N3

Vout

CossCs

Ds1

t4 –t6

Im

Llk

ICoss

iLs

Ds2

iD2

Lm

C1

S1

Tr D1

D2

C2

D3

RL

VinL1

N1 N2

N3

Vout

CossCs

Ds1

t6 –t7

Im

Llk

ICoss

iLs

Ds2

iD2

2133

[t0, t1] - The main switch S1 is turned on at t0 under ZVS condition, and the forward-flyback converter runs at forward mode. The main power is delivered to load through the transformer and the output inductor L1. Meanwhile, the current of the inductor of the snubber circuit (Ls) is discharging at this period.

· · · (1)

Where

· · · · (2)

· · · · · (3)

[t1, t2] - The main switch S1 is still turned on. While the current via Ls reaches to zero, the diode DS2 will be naturally blocked. The operation mode is the same as the previous interval for the main power delivering. [t2, t3] -The main switch S1 is turned off at t2. The output capacitor of the MOSFET and Cs are charged by the current of the primary side. This interval is end at t3 while the voltage across the MOSFET is equal to input voltage Vin plus the output voltage over the turns ratio N3/N1. [t3, t4] - The main switch S1 is still off. The current through D3 delivers energy stored in the transformer to load as a conventional flyback converter. During this interval, the voltage across Ls is same as the primary winding of the transformer, so the snubber inductor is charging until the current through D3 is gradually decreased to zero. [t4, t5] - Natural resonance occurs between the output capacitance of Mosfet S1 and the magnetizing inductance Lm of the transformer. The voltage across Mosfet S1 drops to Vin. [t5, t6] – The secondary windings of N2 transformer is shorted because the D2 is still conducted. The voltage across Mosfet S1 is clamped to input voltage during this period, keeping the transformer’s primary voltage at zero. The mode will enter into another state until the current through L1 is decreased to zero at t6. [t6, t7] – D2 is naturally blocked. The winding N2 of the secondary transformer is not shorted by D1 and D2. So, another resonant mode begins again between the output capacitance of Mosfet and the magnetizing inductance of transformer. Based on this analysis of operation mode, the voltage spike can be largely reduced while the value of the snubber capacitor Cs is selected several Nano-farads. Meanwhile, the output ripple voltage can be greatly reduced compared to flyback design because of the output inductor. Meanwhile, the winding of N3 is used to demagnetize the transformer and recycle the energy stored in the core to load while the switch S1 is turned off.

III. FEATURES OF ZVS FORWARD-FLYBACK CONVERTER

In this section, the steady state operation of the converter is analysed. The following assumptions are considered for the derivations:

1) The converter losses are neglected, which implies that the input power of the converter equals to the output power.

2) All passive components are considered to be ideal, which implies the ESRs of inductors and capacitors are neglected. And all switches (Mosfet &Diode) are also considered to be ideal.

3) The voltage ripple across the output filter capacitance is very small.

A. Transfer Function Expression

The voltage gain of the proposed ZVS forward-flyback converter is derived by · · · · · ·· (4)

From the above equation, the leakage inductance of the transformer will cause the decrease of the voltage gain because of duty cycle loss at the resonant period. While the load resistance is approaching to infinite, the voltage gain is simplified without consideration of the resonant mode.

· · · · (5)

B. Voltage Stress Calculation

Ignoring the voltage ripple on the input capacitor and the output capacitor, from the steady-state operational analysis, the voltage stress of the primary power MOSFET and the secondary output diodes can be derived; and equations are shown in the table 1. From the table 1, it can be found that the voltage stress of the output diode is determined by input &output voltage and the turns ratio of the transformer. The output diode voltage stress decreases with the turns ratio reducing. But the voltage stress of the primary switch S1 is increase while decreasing the turns ratio. It needs balance to select components between primary mosfet and output diode considering the voltage stress.

Table 1. Voltage Stress of the Output Diode and Switch

Device Voltage Stress

S1(switch) Vin + Vout•N1/N3

D1 Vout•N2/N3

D2 Vin•N2/N1

D3 Vin•N3/N1+Vout

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C. Current Stress Calculation

Table2 shows the Root Mean Square (RMS) current stress’s equation of the output diode and primary switch. From the table 2, it can be seen that the current stress of the primary power MOSFETs and the output diodes has a relationship with the turns ratio, duty cycle, switching frequency, the magnetizing inductance of the transformer and the value of the output inductor.

Table 2. Voltage Stress of the Output Diode and Switch

Device Current Stress

S1 (Switch)

√√3 • 1

D1 √3 • √ 1

D2 √3 • 1

D3 1 13 • •

Figure 3. diode current stresses for different input

voltages and load condition

Figure 3 shows the current stress of three diodes for different input voltages at output voltage Vout=230V and different load. From the figure seen, the RMS current of D2 is gradually increased with input voltage increase; but the RMS current of D1 and D3 are both decreased. In addition, according to the power deliver of two parts between forward and flyback, the power from flyback part is delivered at low input voltage that is more than at high input voltage according to the figure of the D3 RMS current.

IV. DESIGN AND CONSIDERATION

In order to deeply understand the forward-flyback converter and achieve high efficiency, several considerations, such as the voltage ripple calculation of the output capacitor, variable frequency control and control scheme, considerations of the freewheel mode during off time and power distribution of forward & flyback mode are discussed below.

A. Soft Switching Condition

From the steady-state operation of the proposed converter, ZVS soft switching performance is achieved for the primary power MOSFET, which reduces the switching losses greatly. Refer to the two modes [t1-t2] and [t3-t4], the condition to achieve ZVS of the primary MOSFET S1 is: firstly the flyback runs at discontinuous mode; secondarily the current of the output inductor L1 of the forward part also runs at boundary mode between discontinuous and continuous that makes the voltage across S1 decreases; third, the energy stored in the leakage inductor of the transformer should be greater than the energy

of the parasitic capacitance of the MOSFET S1 is shown in this equation.

(6)

Where Llk is the leakage inductance of the transformer, Coss is the output capacitance of the primary power MOSFET.

B. Voltage Ripple of the Output Capacitor

Figure 4 illustrates the voltage & current ripple of the output capacitor C2. From the figure 4 seen, during the period t1-t3, the output capacitor C2 is charged by the total current of the output inductor and ID3. While the load current (Iout) is greater than the current sum of the IL1 and ID3. So the output capacitor voltage is discharged after t3. The voltage discharging of the output capacitor will be end until next cycle while the load current is equal to the total current of IL1 and ID3. In addition, the meaning of electrical current can be broken down into change in charge per change in time, i.e.

(7)

By integrating this, we obtain

2135

(8)

From which we see that charge equals the integral of current over time. So, the output capacitor starts to charge while the output current of the forward is greater than the load current. The total charge of Q(t)2 for the output capacitor is expressed before the sum of the output inductor (iL1) and the D3 current (iD3) of the flyback during the freewheel mode is equal to the load current (Iout). The current ripple can be expressed as the followed equation at charging & discharging period:

(9)

Where,

ic is the current through the output capacitor;

iL1 is the current of the output inductor L1;

iD3 is the current of the flyback at freewheel mode;

iout is the load current;

Figure 4. Voltage & Current ripple of the output capacitor

The Voltage ripple of the output capacitor can be expressed as the followed equation:

· _ · _ _ · _ _ (10)

C. Adaptive Variable Frequency Control

The presupposition to achieve ZVS in Mosfet S1 is the output inductor L1 operating the boundary conduction mode and discontinuous current mode of the magnetizing inductance of transformer. Two restraint conditions are assumed: one is ·· , and the other is _ _ which means the discharging time of flyback mode is less than the forward mode to make the circuit work well.

The expression of the off time can be derived according to the equation.

_ 2 (11)

Through the deduction in previous, the equation (11) discloses the off time is mainly determined by the load while L1, Lm and turns ratio are fixed. Therefore, the variable frequency control is proposed to adaptive change with the variation of output power and input voltage. Based on the previous deduction of the fixed off time, figure 5 shows a block diagram of the variable control scheme for this proposed ZVS forward-flyback DC/DC converter. High frequency current transformer is series with the output inductor to detecting the output current of the forward part. The triangular signal of the current transformer, as a saw waveform, compares to the output error of the voltage loop, which determines the turning on time. While the triangular signal is equal to the error of the output voltage loop, the primary Mosfet is turned off. While the triangular signal is equal to zero, the Mosfet will turn on again to start a next cycle. The digital control scheme is verified by a 16 bits Digital signal processor (DSPIC33FJ16GS504) which has internal comparators.

Figure 5. Control diagram for ZVS forward-flyback converter

D. Power Distribution between Forward and Flyback

Followed the section C, the proportion of delivering energy in fly-back mode is caused by different values of the magnetizing inductance of the transformer. The smaller magnetizing inductance of the transformer, the more energy is delivered from fly-back part. But, the smaller magnetizing inductance, the core loss is higher. Therefore, how to select the optimize point, trade-off between the value of Lm and core loss, which is also determined by the power distribution between flyback mode and forward mode. Followed the previous deduction, the ratio δ is average current of output inductor over the current through the third winding N3, the equation is listed below.

Vin

S1

D3

N3

N2

D1

D2

L1

2136

Figure 6. The ratio of power distribution between forward and flyback

is vary with different load and input voltage

(12)

· · 1 (13)

Forward part: ; Flyback part: ; . The bigger δ, the more delivering power is through forward part. Based on the above equation power distribution between forward and flyback, under different input voltage and load condition, are describled in the figure 6.

V. EXPERIMENTAL RESULTS

A 200W prototype is built to verify the performance of the proposed ZVS forward-flyback converter with non-dissipative LC snubber circuit. The specifications of the prototype are listed in table 3.

Table 3. The Detail Parameters of the Prototype

Input voltage 30V-50V Output power 200W

Output voltage 230V Switching frequency 60-220KHz

Turns number N1:N2:N3 5:60:24

Magnetizing inductance 35uH

Leakage inductance 290nH Output inductor L1 810uH

Primary Mosfet S1

IRFS4115PBF(2PCS)

Output diode

D1, D2 (forward) STTH312S

Output capacitor C2 10uF

Output diode

D3 (flyback) STTH310S

Trans Core Size RM14 Snubber Parameters Ls1=128uH

Cs=4.7nF

Switching waveforms under different input voltage and power rating are shown from figure 7 to figure 9 (CH1: voltage stress across Mosfet, CH2: output voltage 230V, CH3: output current, CH4: gate signal of Mosfet). From the figure7 seen, under this condition of 200Watts output at 30V input, ZVS is achieved before the gate signal turns on. Switching frequency is running at 62KHz. The voltage spike across the Mosfet is limited less than 100Volts thanks to the LC snubber circuit. Even in the maximum input 50volts, voltage across the Mosfet is decreased down to 20 Volts at 200 watts output before the switch is turning on; which effectively reduces switching loss.

Figure 7. waveforms in full power (200W) at 30V and 50V input

Figure 8 shows switching waveforms at half load condition of 100 watts output, which the range of the switching frequency is from 75 KHz to 145 KHz.

Figure 8. waveforms in half power (100W) at 30V and 50V input

Figure 9. waveforms in 10% power (20W) at 30V and 50V input

Figure 9 shows switching waveforms at 10% load of 20 watts output, which switching frequency is variable from 157 KHz to 220 KHz. From the figure 9 shown, even at very light loads, only 10% power output, ZVS is still achieved at 30 volts input. Figure 10 (CH1: voltage stress across Mosfet, CH2: voltage stress across D1, CH3 output current of the forward inductor L1, CH4: gate signal), shows that the voltage spike across D1 is less than 700V, and the voltage spike across Mosfet is 134V. Meanwhile, the ZVS performance is achieved at 43 volts input at full power condition.

CH1:Vds (20V/div)CH2:VOut (50V/div)CH3:IOut (0.5A/div)CH4:Vgs (10V/div)

CH1:Vds (20V/div)CH2:VOut (50V/div)CH3:IOut (0.5A/div)CH4:Vgs (5V/div)

CH1:Vds (20V/div)CH2:VOut (50V/div)CH3:IOut (0.5A/div)CH4:Vgs (5V/div)

CH1:Vds (20V/div)CH2:VOut (50V/div)CH3:IOut (0.5A/div)CH4:Vgs (5V/div)

CH1:Vds (20V/div)CH2:VOut (50V/div)CH3:IOut (0.1A/div)CH4:Vgs (5V/div)

CH1:Vds (20V/div)CH2:VOut (50V/div)CH3:IOut (0.1A/div)CH4:Vgs (5V/div)

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Figure 10. ZVS waveform in 200W load at 43V input

Figure 11. Waveforms of Non-disspative snubber capacitor

Figure 11 (CH1: voltage stress across snubber capacitor Cs, CH2: output voltage 230V, CH3 output current of the forward inductor L1, CH4: voltage stress across Mosfet), shows voltage stress between Cs and Mosfet under high voltage input. The voltage spike is reduced to less than 140 volts, which low voltage Mosfet can be used.

CONCLUSIONS

A ZVS forward–flyback converter has been proposed in this paper. Compared with the continuous current mode of the output inductor of forward topology, there is no recovery current in rectifier diodes, significantly reducing the turn-on losses of switch S1. Compared with flyback-type, the proposed converter has smaller output voltage ripples, significantly reducing the value of output capacitor. Non-dissipative LC snubber circuit is used to reduce the voltage spike across the primary switch at turn off time, significantly improve the reliability, the conversion efficiency and power density.

ACKNOWLEDGMENT The author would like to thank Mr. Charlie Jourdan for his help. This work was supported by the United States Department of Energy under Award DE-EE0003176.

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CH1:Vds(20V/div)CH2:VD1(100V/div)CH3:IL1(0.5A/div)CH4: Vgs (10V/div)

CH1:Vds(50V/div) CH2:Vout (50V/div)CH3:IL1(0.5A/div) CH4: VCs (50V/div)

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